MIC3230YML-TR [MICROCHIP]
SWITCHING CONTROLLER, 1000kHz SWITCHING FREQ-MAX, PDSO12;
| 型号： | MIC3230YML-TR |
| 厂家： | 
MICROCHIP |
| 描述： | SWITCHING CONTROLLER, 1000kHz SWITCHING FREQ-MAX, PDSO12 开关 光电二极管 |
| 文件： | 总19页 (文件大小：719K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
MIC3230/1/2
Constant Current Boost Controller for
Driving High Power LEDs
General Description
The MIC3230/1/2 are constant current boost switching
controllers specifically designed to power one or more
strings of high power LEDs. The MIC3230/1/2 have an
input voltage range from 6V to 45V and are ideal for a
variety of solid state lighting applications.
Bringing the Power to Light™
Features
• 6V to 45V input supply range
The MIC3230/1/2 utilizes an external power device which
offers a cost conscious solution for high power LED
applications. The powerful drive circuitry can deliver up to
70W to the LED system. Power consumption has been
minimized through the implementation of a 250mV
feedback voltage reference providing an accuracy of ±3%.
The MIC323x family is dimmable via a pulse width
modulated (PWM) input signal and also features an enable
pin for low power shutdown.
• Capable of driving up to 70W
• Ultra low EMI via dithering on the MIC3231
• Programmable LED drive current
• Feedback voltage = 250mV ±3%
• Programmable switching frequency (MIC3230/1) or
400kHz fixed frequency operation (MIC3232)
• PWM Dimming and separate enable shutdown
• Frequency synchronization with other MIC3230s
• Protection features:
Multiple MIC3230 ICs can be synchronized to a common
operating frequency. The clocks of these synchronized
devices can be used together in order to help reduce noise
and errors in a system.
Over Voltage Protection (OVP)
Over temperature protection
Under-voltage Lock-out (UVLO)
• Packages:
An external resistor sets the adjustable switching
frequency of the MIC3230/1. The switching frequency can
be between 100kHz and1MHz. Setting the switching
frequency provides the mechanism by which a design can
be optimized for efficiency (performance) and size of the
external components (cost). The MIC323x family of LED
drivers also offer the following protection features: Over
voltage protection (OVP), thermal shutdown and under-
voltage lock-out (UVLO).
N/C
VIN
1
2
3
4
5
6
7
8
16 N/C
15 VDD
14 DRV
13 PGND
12 OVP
11 IS
VIN
EN
1
2
3
4
5
6
12 VDD
11 DRV
10 PGND
VIN
EN
1
2
3
4
5
10 VDD
EN
9
8
7
6
DRV
PGND
OVP
IS
PWMD
COMP
IADJ
FS
PWMD
COMP
IADJ
FS
PWMD
COMP
IADJ
9
8
7
OVP
IS
10 SYNC/NC
EPAD
SYNC/NC
EPAD
AGND
9 N/C
MIC3232
10-pin MSOP
MIC3230/1
12-pin MLF®
MIC3230/1
16-pin TSSOP
The MIC3231 offers a dither feature to assist in the
reduction of EMI. This is particularly useful in sensitive EMI
applications, and provides for a reduction or emissions by
approximately 10dB.
• –40°C to +125°C junction temperature range
The MIC3232 is a 400kHz fixed frequency device offered
in a small 10-pin MSOP package. The MIC3230/1 are
offered in both the EPAD 16-pin TSSOP package and the
12-pin 3mm × 3mm MLF® package.
Applications
• Street lighting
• Solid state lighting
• General illumination
• Architectural lighting
• Constant current power supplies
Datasheets and support documentation can be found on
Micrel’s web site at: www.micrel.com.
Bringing the Power to Light is a trademark of Micrel, Inc.
MicroLeadFrame and MLF are registered trademarks of Amkor Technology.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
M9999-030311-D
March 2011
Micrel, Inc.
MIC3230/1/2
Typical Application
L
D1
47µH
VIN
VOUT
CIN
4.7µF/50v
R8
100k
VIN
R2
100k
COUT
4.7µF
100V
LED 1
ENABLE
PWMD
EN
OVP
DRV
PWMD
SYNC
FS
Synch to other MIC3230
Q1
LED X
MIC3230/31
R9
4.33k
ILED Return
RSLC
51
COMP
VDD
AGND
IS
IADJ
RCS
VFB = 0.25V
RFS
16.5k CCOMP
10nF
1/2W
C3
10µF
10V
RADJ
PGND
EPAD
1/4W
Analog ground
Power ground
Figure 1. Typical Application of the MIC3230 LED Driver
Product Option Matrix
MIC3230
MIC3231
MIC3232
6V to 45V
No
Input Voltage
Synchronization
Dither
6V to 45V
6V to 45V
Yes
No
No
Yes
No
Frequency Range
Adj from 100kHz to 1MHz
Adj from 100kHz to 1MHz
Fixed Freq. = 400kHz
16-pin EPAD TSSOP
16-pin EPAD TSSOP
Package
10-pin MSOP
12-pin 3mm × 3mm MLF®
12-pin 3mm × 3mm MLF®
Ordering Information
Part Number
Temperature Range
Package
Lead Finish
MIC3230YTSE
MIC3230YML
MIC3231YTSE
MIC3231YML
MIC3232YMM
–40° to +125°C
–40° to +125°C
–40° to +125°C
–40° to +125°C
–40° to +125°C
16-pin EPAD TSSOP
Pb-Free
12-pin 3mm x 3mm MLF®
16-pin EPAD TSSOP
12-pin 3mm x 3mm MLF®
10-pin MSOP
Pb-Free
Pb-Free
Pb-Free
Pb-Free
M9999-030311-D
March 2011
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Micrel, Inc.
MIC3230/1/2
Pin Configuration
N/C
VIN
1
2
3
4
5
6
7
8
16 N/C
15 VDD
14 DRV
13 PGND
12 OVP
11 IS
VIN
EN
1
2
3
4
5
6
12 VDD
11 DRV
10 PGND
VIN
EN
1
2
10 VDD
EN
9
8
7
6
DRV
PGND
OVP
IS
PWMD
COMP
IADJ
FS
PWMD
COMP
IADJ
FS
PWMD 3
9
8
7
OVP
COMP
IADJ
4
5
IS
EPAD
SYNC/NC
10 SYNC/NC
EPAD
AGND
9 N/C
10-Pin MSOP (MM)
MIC3232
12-Pin 3mmx3mmMLF® (ML)
MIC3230, MIC3231
16-Pin TSSOP (TSE)
MIC3230, MIC3231
See Product Option Matrix for selection
See Product Option Matrix for selection
Pin Description
Pin Number Pin Number Pin Number
Pin Name
Pin Function
MLF®
TSSOP
MSOP
--
1
2
1
2
3
--
1
2
NC
VIN
EN
No Connect.
Input Voltage (power) 6V to 45V.
Enable Control (Input). Logic High (≥1.5V) enables the
regulator. Logic Low (≤0.4V) shuts down the regulator.
Connect a 100kΩ resistor from EN to VIN.
3
4
3
PWMD
PWM Dimming Input. Logic Low will disable the brightness
control of the LED drivers.
4
5
6
5
6
7
4
5
--
COMP
IADJ
FS
Compensation (output): for external compensation.
Feedback (input).
Frequency Select (input). Connected to a Resistor to
determine the operating frequency.
--
--
7
8
9
--
--
--
AGND
NC
Analog Ground.
No Connect.
10
SYNC
Sync (output). Connect to another MIC3230 to synchronize
multiple converters.
8
9
11
12
6
7
IS
Current Sense (input). Connected to external current sense
resistor which in turn is connected to the source of the external
FET as well as an external slope compensation resistor.
OVP
OVP divider connection (output). Connect the top of the
divider string to the output. If the load is disconnected, the
output voltage will rise until OVP reaches 1.25V and then will
regulate around this point.
10
11
12
13
14
15
8
9
PGND
DRV
Power Ground.
Drive Output: connect to the gate of external FET (output).
10
VDD
VDD Filter for internal power rail. Do not connect an external
load to this pin. Connect 10µF to GND.
--
--
16
--
--
--
NC
No Connect.
EPAD
Connect to AGND.
M9999-030311-D
March 2011
3
Micrel, Inc.
MIC3230/1/2
Absolute Maximum Ratings(1)
Operating Ratings(2)
Supply Voltage (VIN).....................................................+48V
Enable Pin Voltage...........................................-0.3V to +6V
IADJ Voltage ..................................................................+6V
Lead Temperature (soldering, sec.)........................... 260°C
Storage Temperature (Ts)..........................-65°C to +150°C
ESD Rating(3)
Supply Voltage (VIN)......................................... +6V to +45V
Junction Temperature (TJ)........................–40°C to +125°C
Junction Thermal Resistance
MSOP (θJA) ...................................................130.5°C/W
EPAD TSSOP (θJA).........................................36.5°C/W
MLF® (θJA).......................................................60.7°C/W
MIC3230 ....................................... 1500V HB, 100VMM
MIC3232 ........................................... 2kV HB, 100VMM
MIC3231 ....................................... 1500V HB, 150VMM
Electrical Characteristics(4)
VIN = 12V; VEN = 3.6V; L = 47µH; C = 4.7µF; TJ = 25°C, Bold values indicate –40°C≤ TJ ≤ +125°C, unless noted.
Symbol Parameter
Condition
Min
6
Typ
Max
45
Units
V
VIN
Supply Voltage Range
UVLO
IVIN
Under Voltage Lockout
Quiescent Current
3.5
4.9
3.2
5.5
10
V
VFB > 275mV (to ensure device is not
switching)
mA
ISD
Shutdown Current
VEN = 0V
30
250
250
1.2
2
µA
mV
mV
µA
%
VIADJ
Feedback Voltage (at IADJ)
Room temperature (3%)
–40°C≤ TJ ≤ +125°C (5%)
VFB = 250mV
242.5
257.5
262.5
3
237.5
IADJ
Feedback Input Current
Line Regulation
VIN = 12V to 24V
VOUT to 2 × VOUT
Load Regulation
2
%
DMAX
VEN
Maximum Duty Cycle
MIC3230 & MIC3232
MIC3231
90
88
%
%
Enable Threshold
Turn ON
Turn OFF
1.5
1.15
1.1
V
V
0.4
IEN
Enable Pin Current
PWMD Threshold
VEN = 3.3V
REN = 100kΩ
17
30
µA
VPWM
Turn ON
Turn OFF
1.5
0
0.75
0.7
V
V
0.4
fPWMD
fSW
PWMD Frequency Range
Note 5 (L = 47µH; C = 4.7µF)
500
Hz
Programmable Oscillator
Frequency
RFREQ = 82.5kΩ
109
400
950
kHz
kHz
kHz
RFREQ = 21kΩ
360
360
440
440
RFREQ = 8.25kΩ
fSW
Fixed Frequency Option
Low EMI (MIC3231)
(MIC3232YMM)
400
±12
0.45
250
kHz
%
FDITHER
VSENS
ISENSE
Notes:
Frequency dither shift from nominal
RSENSE = 390Ω
Current Limit Threshold Voltage
ISENSE Peak Current Out
0.315
0.585
V
RSENSE = 390Ω
µA
1. Exceeding the absolute maximum rating may damage the device.
2. The device is not guaranteed to function outside its operating rating.
3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF.
4. Specification for packaged product only.
5. Guaranteed by design
M9999-030311-D
March 2011
4
Micrel, Inc.
MIC3230/1/2
Electrical Characteristics (Continued)
Symbol Parameter
Condition
Min
Typ
Max
Units
VOVP
Over Voltage Protection
1.203
1.24
1.277
3.5
V
Driver Impedance
Sink
Source
2.4
2
Ω
Ω
VDRH
TJ
Driver Voltage High
VIN = 12V
7
9
11
V
Over-Temperature Threshold
Shutdown
150
°C
Thermal Shutdown
Hysteresis
5
°C
M9999-030311-D
March 2011
5
Micrel, Inc.
MIC3230/1/2
Typical Characteristics
M9999-030311-D
March 2011
6
Micrel, Inc.
MIC3230/1/2
Load Regulation
12.2
12.15
12.1
12.05
12
11.95
11.9
VIN = 3.6V
11.85
11.8
0
25 50 75 100 125 150
LOAD (mA)
M9999-030311-D
March 2011
7
Micrel, Inc.
MIC3230/1/2
The MIC3230/1/2 features a low impedance gate driver
capable of switching large MOSFETs. This low impedance
helps provide higher operating efficiency.
Functional Description
A constant output current converter is the preferred
method for driving LEDs. Small variations in current have a
minimal effect on the light output, whereas small variations
in voltage have a significant impact on light output. The
MIC323x family of LED drivers are specifically designed to
operate as constant current LED Drivers and the typical
application schematic is shown in Figure 1.
The MIC323x family can control the brightness of the
LEDs via its PWM dimming capability. Applying a PWM
signal (up to 500Hz) to the PWMD pin allows for control of
the brightness of the LED.
Each member of the MIC323x family employs peak current
mode control. Peak current mode control offers
advantages over voltage mode control in the following
manner. Current mode control can achieve a superior line
transient performance compared to voltage mode control
and through small signal analysis (not shown here),
current mode control is easier to compensate than voltage
mode control, thus allowing for a less complex control loop
stability design. Figure 2 shows the functional block
diagram.
The MIC323x family is designed to operate as a boost
controller, where the output voltage is greater than the
input voltage. This configuration allows for the design of
multiple LEDs in series to help maintain color and
brightness. The MIC323x family can also be configured as
a SEPIC controller, where the output voltage can be either
above or below the input voltage.
The MIC3230/1/2 have a very wide input voltage range,
between 6V and 45V, to help accommodate for a diverse
range of input voltage applications. In addition, the LED
current can be programmed to a wide range of values
through the use of an external resistor. This provides
design flexibility in adjusting the current for a particular
application need.
Figure 2. MIC3230 Functional Block Diagram
M9999-030311-D
March 2011
8
Micrel, Inc.
MIC3230/1/2
Output Over Voltage Protection (OVP)
Power Topology
The MIC323x provides an OVP circuitry in order to help
protect the system from an overvoltage fault condition.
This OVP point can be programmed through the use of
external resistors (R8 and R9 in Figure 1). A reference
value of 1.245V is used for the OVP. Equation 3 can be
used to calculate the resistor value for R9 to set the OVP
point.
Constant Output Current Controller
The MIC323x family is peak current mode boost
controllers designed to drive high power LEDs. Unlike a
standard constant output voltage controller, the MIC323x
family has been designed to provide a constant output
current. The MIC323x family is designed for a wide input
voltage range, from 6V to 45V. In the boost configuration,
the output can be set from VIN up to 100V.
R8
Eq. (3)
R9 =
(V
/1.245) −1
OVP
As a peak current mode controller, the MIC323x family
provides the benefits of superior line transient response as
well as an easier to design compensation.
LED Dimming
The MIC323x family of LED drivers can control the
brightness of the LED string via the use of pulse width
modulated (PWM) dimming. A PWM input signal of up to
500Hz can be applied to the PWM DIM pin (see Figure 1)
to pulse the LED string ON and OFF. It is recommended to
use PWM dimming signals above 120Hz to avoid any
recognizable flicker by the human eye. PWM dimming is
the preferred way to dim a LED in order to prevent
color/wavelength shifting, as occurs with analog dimming.
The output current level remains constant during each
PWMD pulse.
This family of LED drivers features a built-in soft-start
circuitry in order to prevent start-up surges. Other
protection features include:
•
Current Limit (ILIMIT) - Current sensing for over current
and overload protection
•
Over Voltage Protection (OVP) - Output over voltage
protection to prevent operation above a safe upper
limit
•
Under Voltage Lockout (UVLO) – UVLO designed to
prevent operation at very low input voltages
Oscillator and Switching Frequency Selection
Setting the LED Current
The MIC323x family features an internal oscillator that
synchronizes all of the switching circuits internal to the IC.
This frequency is adjustable on the MIC3230 and MIC3231
and fixed at 400kHz in the MIC3232.
The current through the LED string is set via the value
chosen for the current sense resistor, RADJ. This value can
be calculated using Equation 1:
0.25V
In the MIC3230/1, the switching frequency can be set by
choosing the appropriate value for the resistor, R1
according to Equation 4:
I
=
Eq. (1)
LED
R
ADJ
Another important parameter to be aware of in the boost
controller design is the ripple current. The amount of ripple
current through the LED string is equal to the output ripple
voltage divided by the LED AC resistance (RLED – provided
by the LED manufacturer) plus the current sense resistor
(RADJ). The amount of allowable ripple through the LED
string is dependent upon the application and is left to the
designer’s discretion. This equation is shown in Equation
2:
1.035
⎛
⎞
7526
⎜
⎜
⎟
⎟
Eq. (4)
RFS (kΩ) =
FSW (kHz)
⎝
⎠
SYNC (MIC3230 Only)
Multiple MIC3230 ICs can be synchronized by connecting
their SYNC pins together. When synchronized, the
MIC3230 with the highest frequency (master) will override
the other MIC3230s (slaves). The internal oscillator of the
master IC will override the oscillator of the slave part(s)
and all MIC3230 will be synchronized to the same master
switching frequency.
V
OUTRIPPLE
Eq. (2)
ΔILED ≈
(RLED + RADJ
)
ILED × D ×T
Where
VOUT
=
The SYNC pin is designed to be used only by other
MIC3230s and is available on the MIC3230 only. If the
SYNC pin is being unused, it is to be left floating (open). In
the MIC3231, the SYNC pin is to be left floating (open).
RIPPLE
COUT
Reference Voltage
The voltage feedback loop of the MIC323x uses an
internal reference voltage of 0.25V with an accuracy of
±3%. The feedback voltage is the voltage drop across the
current setting resistor (RADJ) as shown in Figure 1. When
in regulation the voltage at IADJ will equal 0.25V.
M9999-030311-D
March 2011
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Micrel, Inc.
MIC3230/1/2
Dithering (MIC3231 Only)
Current Sense IS
The MIC3231 has a feature which dithers the switching
frequency by ±12%. The purpose of this dithering is to help
achieve a spread spectrum of the conducted EMI noise.
This can allow for an overall reduction in noise emission by
approximately 10dB.
The IS pin monitors the rising slope of the inductor current
(m1 in Figure 5) and also sources a ramp current
(250µA/T) that flows through RSLC that is used for slope
compensation. This ramp of 250µA per period, T,
generates a ramped voltage across RSLC and is labeled VA
in Figure 3. The signal at the IS pin is the sum of VCS + VA
(as shown in Figure 3). The current sense circuitry and
block diagram is displayed in Figure 4. The IS pin is also
used as the current limit (see the previous section on
Current Limit).
Internal Gate Driver
External FETs are driven by the MIC323x’s internal low
impedance gate drivers. These drivers are biased from the
VDD and have a source resistance of 2ꢀ and a sink
resistance of 3.5ꢀ.
VDD
VDD is an internal linear regulator powered by VIN and VDD
is the bias supply for the internal circuitry of the MIC323x.
A 10µF ceramic bypass capacitor is required at the VDD pin
for proper operation. This pin is for filtering only and should
not be utilized for operation.
Current Limit
The MIC323x family features a current limit protection
feature to prevent any current runaway conditions. The
current limit circuitry monitors current on a pulse by pulse
basis. It limits the current through the inductor by sensing
the voltage across RCS. When 0.45V is present at the IS
pin, the pulse is truncated. The next pulse continues as
normally until the IS pin reaches 0.45V and it is truncated
once again. This will continue until the output load is
decreased.
Figure 3. Slope Compensation Waveforms
Soft Start
The boost switching convertor features a soft start in order
to power up in a controlled manner, thereby limiting the
inrush current from the line supply. Without this soft start,
the inrush current could be too high for the supply. To
prevent this, a soft start delay can be set using the
compensation capacitor (CCOMP in Figure 1). For switching
to begin, the voltage on the compensation cap must reach
about 0.7V. Switching starts with the minimum duty cycle
and increases to the final duty cycle. As the duty cycle
increases, VOUT will increase from VIN to its final value. A
6µA current source charges the compensation capacitor
and the soft start time can be calculated in Equation 7:
Select RCS using Equation 5:
0.45
Eq. (5)
RCS =
(
VOUT −VIN
)
× D
MAX
MIN
+ IL
PK _ LIMIT
L× FSW
Slope Compensation
CCOMP ×VCOMP_STEADY_STATE
Eq. (7) TSOFTSTART
≈
The MIC323x is a peak current mode controller and
requires slope compensation. Slope compensation is
required to maintain internal stability across all duty cycles
and prevent any unstable oscillations. The MIC323x uses
slope compensation that is set by an external resistor,
RSLC. The ability to set the proper slope compensation
through the use of a single external component results in
6μA
VCOMP_STEADY_STATE is usually between 0.7V to 3V, but can
be as high as 5V.
Eq. (8) VCOMP _ STEADY _ STATE = Ai×
VA +VcsPK
PK
IRAMP
Where: VA
=
× RSLC × D×T and
design flexibility. This slope compensation resistor, RSLC
can be calculated using Equation 6:
,
PK
T
V
= IL
× RCS
CSPK
_ PK
V
−V
×R
CS
OUTMAX
INMIN
Eq. (6) R
=
SLC
Ai = 1.4 V/V
L×250μA×F
SW
D = Duty cycle (0 to1)
T = period
where VIN_MAX and VOUT_MAX can be selected to system
specifications.
A 10nF ceramic capacitor will make this system stable at
all operating conditions.
M9999-030311-D
March 2011
10
Micrel, Inc.
MIC3230/1/2
Leading Edge Blanking
2
(
IIN _ PP
)
2
Large transient spikes due to the reverse recovery of the
diode may be present at the leading edge of the current
sense signal. (Note: drive current can also cause such
spikes) For this reason a switch is employed to blank the
first 100ns of the current sense signal. See Figure 6.
Eq. (10)
Eq. (11)
IIN _ AVE
=
(
IIN _RMS
)
−
12
IIN _ PP
IIN _ PEAK = IIN _ AVE
+
2
Note: If IIN_PP is small then IIN_AVE nearly equals IIN_RMS
VOUT × IOUT
Eq. (9)
IIN _RMS =
eff ×VIN
VIN
L1
D1
S
R
Clock
DRV
Q
IL
250µa/T
VA
+RSLC–
PWM Comparator
IS
VCS
Ai
VA = IRAMP × RSLP
+
RCS
–
VCS = IL × RCS
Current Limit
0.45V
VC
0.45V
IADJ
RCOMP = 10k
COMP
CCOMP
Figure 4. Current Sense Circuit (An explanation of the IS pin)
T
Clock
(1-D)T
DT
PWM
IL_PK = IL_AVE + 1/2 IL_PP
VC
m2
IL_AVE = IIN_AVE
IL_PP
m1
IL
0
VC
IL_AVE = IIN_AVE
IFET_RMS
IFET
0
VC
IDIODE
IOUT
0
Figure 5. Current Waveforms
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March 2011
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Micrel, Inc.
MIC3230/1/2
Figure 6. IS Pin and VRCS (Ch1 = Switch Node, Ch2 = IS Pin, Ref1 = VCS
)
Design Procedure for a LED Driver
Symbol Parameter
Input
Min
Nom
Max
Units
VIN
IIN
Input Voltage
Input current
8
12
14
2
V
A
Output
LEDs
VF
Number of LEDs
Forward voltage of LED
Output voltage
5
6
3.5
21
7
3.2
16
4.0
28
V
V
VOUT
ILED
LED current
0.33
0.35
40
0.37
A
IPP
Required I Ripple
PWM Dimming
mA
%
V
PWMD
OVP
0
100
Output over voltage protection
30
System
FSW
Switching frequency
Efficiency
500kHz
80
eff
%
V
VDIODE
Forward drop of schottky diode
0.6
Table 2. Design Example Parameters
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March 2011
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Micrel, Inc.
MIC3230/1/2
L
D1
47µH
VIN
VOUT
CIN
4.7µF/50v
R8
100k
VIN
R2
100k
COUT
4.7µF
100V
LED 1
ENABLE
EN
OVP
DRV
PWMD
PWMD
SYNC
FS
Synch to other MIC3230
Q1
LED X
MIC3230/31
R9
4.33k
ILED Return
RSLC
51
COMP
VDD
AGND
IS
IADJ
RCS
VFB = 0.25V
RFS
16.5k CCOMP
1/2W
10nF
C3
10µF
10V
RADJ
PGND
EPAD
1/4W
Analog ground
Power ground
Figure 7. Design Example Schematic
These can be calculated for the nominal (typical) operating
conditions, but should also be understood for the minimum
and maximum system conditions as listed below.
Design Example
In this example, we will be designing a boost LED driver
operating off a 12V input. This design has been created
to drive six LEDs at 350mA with a ripple of about 12%.
We are designing for 80% efficiency at a switching
frequency of 500kHz.
(Voutnom − eff ×Vinnom + Vschottky
Voutnom + Vschottky
)
Dnom
Dmax
Dmin
=
(Voutmax − eff ×Vinmin + Vschottky
Voutmax + Vschottky
)
=
Select RFS
To operate at a switching frequency of 500kHz, the RFS
resistor must be chosen using Equation 3.
(Voutmin − eff ×Vinmax +Vschottky
Voutmin +Vschottky
)
=
(
7526 1.035
)
Therefore DNOM =56% DMAX = 78% and DMIN = 33%
RFS
(
kΩ
)
=
= 16.6kΩ
500
Inductor Selection
Use the closest standard value resistor of 16.5kꢀ.
First, it is necessary to calculate the RMS input current
(nominal, min and max) for the system given the operating
conditions listed in the design example table. This minimum
value of the RMS input current is necessary to ensure proper
operation. Using Equation 9, the following values have been
calculated:
Select RADJ
Having chosen the LED drive current to be 350mA in this
example, the current can be set by choosing the RADJ
resistor from Equation 1:
0.25V
RADJ
=
= 0.71Ω
V
×I
OUT _max OUT _max
0.35A
The power dissipation in this resistor is:
RADJ
= I2 * RADJ = 87mW
I
=
=
=
= 1.64A _ rms
= 0.78A _ rms
= 0.48A _ rms
IN _ RMS _max
eff ×V
IN _min
V
×I
OUT _ nom OUT _ nom
P
(
)
I
IN _ RMS _ nom
eff ×V
IN _ nom
Use a resistor rated at ¼ watt or higher. Choose the
closest value from a resistor manufacture.
V
×I
OUT _min OUT _min
I
IN _ RMS _min
eff ×V
IN _max
Operating Duty Cycle
Iout is the same as ILED
The operating duty cycle can be calculated using
Equation 12 provided below:
Selecting the inductor current (peak-to-peak), IL_PP, to be
between 20% to 50% of IIN_RMS_nom, in this case 40%, we
obtain:
(Vout − eff ×Vin +Vdiode
Vout +Vdiode
)
Eq. (12)
D =
I
= 0.4I
= 0.4 * 0.78 = 0.31A
in _rms _nom P−P
in _PP _nom
M9999-030311-D
March 2011
13
Micrel, Inc.
MIC3230/1/2
0.45 = IRAMP × RSLC × D + IL _ pk × RCS
(see the current waveforms in Figure 5).
Eq. (14a)
Limit
It can be difficult to find large inductor values with high
saturation currents in a surface mount package. Due to
this, the percentage of the ripple current may be limited
by the available inductor. It is recommended to operate
in the continuous conduction mode. The selection of L
described here is for continuous conduction mode.
To calculate the value of the slope compensation resistance,
RSLC, we can use Equation 5:
V
−V
× RCS
OUTMAX
INMIN
RSLC
=
L × 250μA ×FSW
First we must calculate RCS, which is given below in
Equation 15:
V
× D ×T
IN
Eq. (13)
L =
I
0.45
VOUTMAX −VINMIN ×Dmax
L×FSW
Eq. (15)
in _ PP
RCS
=
(
)
+ I
Using the nominal values, we get:
L _ pkLimit
12V × 0.56 × 2μs
L =
= 43μH
Therefore;
0.31A
0.45
0.50
Select the next higher standard inductor value of 47µH.
RCS
=
= 179mΩ
(
28v − 8v
)
×
(
)
+ 1.9A
Going back and calculating the actual ripple current
gives:
47μH × 500kHz
Using a standard value 150mꢀ resistor for RCS, we obtain
the following for RSLC
V
×D
×T
12v ×0.56×2us
47uh
IN _ nom
nom
Eq. (13a)
:
I
=
=
= 0.29A
PP
in _PP
L
28 − 8
47μH × 250μA × 500kHz
)
×150mΩ
RSLC
=
= 511Ω
The average input current is different than the RMS input
current because of the ripple current. If the ripple current
is low, then the average input current nearly equals the
RMS input current. In the case where the average input
current is different than the RMS, Equation 10 shows the
following:
Use the next higher standard value if this not a standard
value. In this example 511ꢀ is a standard value.
Check: Because we must use a standard value for Rcs and
R
SLC; I
may be set at a different level (if the calculated
L_pkLimit
2
IIN _ PP
value isn’t a standard value) and we must calculate the
2
Eq. (13b)
IIN _ AVE _ max
=
(IIN _ RMS _ max
)
−
12
actual IL_pk
value (remember IL_pk is the same as
Limit
Limit
2
0.29
2 /12 ≈ 1.64A
)
Iin_pk ).
IIN _ AVE _max
=
(
1.64
)
−
(
Limit
The Maximum Peak input current IL_PK can found using
equation 11:
Rearranging Equation 14a to solve for
:
L _ pkLimit
I
(0.45 − I
× R
× D)
SLC
RAMP
I
=
IL _ PK _ max = IIN _ AVE _ max + 0.5× IL _ PP _ max =1.78A
in _ pkLimit
R
CS
The saturation current (ISAT) at the highest operating
temperature of the inductor must be rated higher than
this.
(0.45 − 250ua × 511×0.75)
I
=
= 2.34A
in _ actualLimit
.150
This is higher than the initial 1.2×IL _PK _max = 1.9A limit
The power dissipated in the inductor is:
= Iin _RMS _max2 × DCR
because we have to use standard values for RCS and for
Eq. (13c)
P
INDUCTOR
RSLC. If I
is too high than use a higher value for
in_actualLimit
RCS. The calculated value of RCS for a 1.9A current limit was
179mꢀ. In this example, we have chosen a lower value
which results in a higher current limit. If we use a higher
standard value the current limit will have a lower value. The
designer does not have the same choices for small valued
resistors as with larger valued resistors. The choices differ
from resistor manufacturers. If too large a current sense
resistor is selected, the maximum output power may not be
able to be achieved at low input line voltage levels. Make
sure the inductor will not saturate at the actual current limit
Current Limit and Slope Compensation
Having calculated the IL_pk above, We can set the current
limit 20% above this maximum value:
I
= 1.2 ×1.6A = 1.9A
L _ pkLimit
The internal current limit comparator reference is set at
0.45V, therefore when VIS _ PIN = 0.45, the IC enters
current limit.
Eq. (14)
0.45 = VA +VcsPK
PK
Iin_actual
.
Limit
Where VA is the peak of the VA waveform and
PK
Perform a check at IIN=2.34Apk.
VcsPK is the peak of the Vcs waveform
V
= 250μA×
0.78 × 511Ω + 2.34A×150mΩ = 0.45V
)
IS _PIN
M9999-030311-D
March 2011
14
Micrel, Inc.
MIC3230/1/2
Maximum Power dissipated in RCS is;
Input Capacitor
2
The input current is shown in Figure 5. For superior
performance, ceramic capacitors should be used because of
their low equivalent series resistance (ESR). The input ripple
current is equal to the ripple in the inductor plus the ripple
voltage across the input capacitor, which is the ESR of CIN
times the inductor ripple. The input capacitor will also
bypass the EMI generated by the converter as well as any
voltage spikes generated by the inductance of the input line.
Eq. (17)
Eq. (18)
P
= I
× RCS
RCS
RCS _ RMS
2
⎛
⎜
⎞
⎟
IL _ PP
2
IR
= IFET _ RMS _ max
=
D IIN _ AVE _ max
+
CS _RMS _max
⎜
⎝
⎟
⎠
12
2
⎛
⎞
⎟
0.26
12
2
⎜
I
=
0.78 1.64
+
= 1.44A _ rms
RCS _ RMS
⎜
⎝
⎟
⎠
For a required VIN_RIPPLE
:
PR = 1.252 ×.15 = 0.31watt
CS
Eq. (21)
Use a 1/2 Watt resistor for RCS.
I
(
0.28A
)
IN _ PP
C
=
=
= 1.4μF
IN
8×V
× F
8× 50mV × 500kHz
Output Capacitor
IN _ RIPPLE
SW
In this LED driver application, the ILED ripple current is a
more important factor compared to that of the output
ripple voltage (although the two are directly related). To
find the COUT for a required ILED ripple use the following
calculation:
This is the minimum value that should be used. The input
capacitor should also be rated for the maximum RMS input
current. To protect the IC from inductive spikes or any
overshoot, a larger value of input capacitance may be
required and it is recommended that ceramic capacitors be
used. In this design example a value of 4.7µF ceramic
capacitor was selected.
For an output ripple ILED
= 20% of ILED
nom
ripple
ILED
= 0.2× 0.35 = 70mA
ripple
MOSFET Selection
ILED
* D
*T
nom
nom
In this design example, the FET has to hold off an output
voltage maximum of 30V. It is recommended to use an 80%
de-rating value on switching FETs, so a minimum of a 38V
FET should be selected. In this design example, a 75V FET
has been selected.
Eq. (19)
C
=
out
ILED
* (R
+ R
)
ripple
adj
LED _total
Find the equivalent ac resistance RLED _ ac from the
datasheet of the LED. This is the inverse slope of the
ILED vs. VF curve i.e.:
The switching FET power losses are the sum of the
conduction loss and the switching loss:
ΔVF
Eq. (20)
RLED _ ac =
Eq. (22)
P
= P
+ P
FET
FET _COND FET _SWITCH
ΔILED
The conduction loss of the FET is when the FET is turned
on. The conduction power loss of the FET is found by the
following equation:
In this example, use RLED_ac = 0.1ꢀ for each LED.
If the LEDs are connected in series, multiply
RLED_ac = 0.1ꢀ by the total number of LEDs. In this
2
Eq. (23)
P
= I
× R
, where
FET _COND
FET _ RMS
DSON
example of 6 LEDs, we obtain the following:
R
= 6× 0.1Ω = 0.6Ω
2
LED _ total
⎛
⎞
⎟
IL _ PP
2
⎜
IFET _ RMS = D IIN _ AVE
+
ILED
* D
*T
nom
nom
⎜
⎟
12
C
=
= 4.1uF
out
⎝
⎠
ILED
* (R
+ R
)
LED _ total
ripple
adj
The switching losses occur during the switching transitions
of the FET. The transition times, ttransition, are the times when
the FET is turning off and on. There are two transition times
per period, T. It is important not to confuse T (the period)
Use the next highest standard value, which is 4.7μF.
There is a trade off between the output ripple and the
rising edge of the PWMD pulse. This is because
between PWM dimming pulses, the converter stops
pulsing and COUT will start to discharge. The amount that
COUT will discharge depends on the time between PWM
Dimming pluses. At the next PWMD pulse COUT has to
be charged up to the full output voltage VOUT before the
desired LED current flows.
with the transition time, ttransition
.
1
Eq. (24)
Eq. (25)
T =
Fsw
PFET _ SWITCH _max = IFET _ AVE _ max ×VOUT _ max × ttransition _max × FSW
To find t
:
transition _max
M9999-030311-D
March 2011
15
Micrel, Inc.
MIC3230/1/2
voltage stress on the diode is the max VOUT and therefore a
diode with a higher rating than max VOUT should be used. An
80% de-rating is recommended here as well.
Qg
Eq. (26)
ttransition _max
≈
Igatedrv
where Qg is the total gate charge of the external
Eq (28)
Pdiode≈VSCHOTTKY×IOUT_ max
MOSFET provided by the MOSFET manufacturer and
the Qg should chosen at a VGS≈10V. This is not an
Pdiode ≈VSCHOTTKY × ILED _ max
Pdiode ≈ 0.25W
exact value, but is more of an estimate of ttransition_max
.
Eq. (29)
The FET manufacturers’ provide a gate charge at a
specified VGS voltage:
MIC3230 Power Losses
Q
G
C
=
In _FET
The power losses in the MIC3230are:
@V
GS
Eq.(30)
P
= Q
×V
× F + I ×Vin
gate Q
This is the FET’s input capacitance. Select a FET with
RDS(on) and QG such that the external power is below
about 0.7W for a SO-8 or about 1W for a PowerPak
(FET package). The Vishay Siliconix Si7148DP in a
PowerPak SO-8 package is one good choice. The
internal gate driver in the MIC3230/1/2 is 2A. From the
Si7148DP data sheet:
MIC3230
gate
where Qgate is the total gate charge of the external
MOSFET. Vgate is the gate drive voltage of the MIC3230.
F is the switching frequency. IQ is the quiescent current of
the MIC3230 found in the electrical characterization table.
IQ = 3.2mA . VIN is the voltage at the VIN pin of the MIC3230.
R
DS(on)_25°C=0.0145ꢀ
From Eq.(30)
Total gate Charge=68nC (typical)
P
= 68nF ×12× 500kHz + 3.2mA×14 = 0.45W
The RDS(on)(temp) is a function of temperature. As the
MIC3230
OVP-Over voltage protection
temperature in the FET increases so does the RDS(on)
.
To find RDS(on)(temp) use Equation 27, or simply double
the RDS(on)(25o C) for RDS(on)(125o C) .
Set OVP higher than the maximum output voltage by at least
one volt. To find the resistor divider values for OVP use
Equation 3 and set the OVP=30V and R8=100kꢀ:
o
(Temp−25o )
Eq. (27)
R
(temp) = R
(25 C)× (1.007
)
DS(on)
DS(on)
100kΩ×1.245
30 −1.245
R9 =
= 4.33kΩ
The RDS(on)(temp) at 125°C is:
RDSon (125oC) = 0.0145 × (1.007(125 −25 ) ) ≈ 30mΩ
From Equation 23:
PCB Layout
∗
o
1. All typologies of DC-to-DC converters have a reverse
recovery current (RRC) of the flyback or (freewheeling)
diode. Even a Schottky diode, which is advertised as having
zero RRC, it really is not zero. The RRC of the freewheeling
diode in a boost converter is even greater than in the Buck
converter. This is because the output voltage is higher than
the input voltage and the diode has to charge up to –VOUT
during each on-time pulse and then discharge to VF during
the off-time.
PFET _COND = 1.642 × 30mΩ = 62mW
Qg
68nC
2A
From Equation 26:
t
≈
=
= 34ns
transition
Igatedrv
I
= 1.64A
FET _ AVE _max
V
= 28V
OUT _max
From Equation 25:
2. Even though the RRC is very short (tens of nanoseconds)
the peak currents are high (multiple amperes). The high
RRC causes a voltage drop on the ground trace of the PCB
and if the converter control IC is referenced to this voltage
drop, the output regulation will suffer.
P
= 1.64A × 28V × 34ns × 500kHz = 0.78Watts
FET _SWITCH _max
From Equation 22
P
= 62mW + 0.78W = 0.84W
FET
This is about the limit for a part on a circuit board without
having to use any additional heat sinks.
3. It is important to connect the IC’s reference to the same
point as the output capacitors to avoid the voltage drop
caused by RRC. This is also called a star connection or
single point grounding.
Rectifier Diode
A Schottky Diode is best used here because of the lower
forward voltage and the low reverse recovery time. The
4. Feedback trace: The high impedance traces of the FB
should be short.
M9999-030311-D
March 2011
16
Micrel, Inc.
MIC3230/1/2
Package Information
10-Pin MSOP (MM)
M9999-030311-D
March 2011
17
Micrel, Inc.
MIC3230/1/2
12-Pin 3mm × 3mm MLF® (ML)
M9999-030311-D
March 2011
18
Micrel, Inc.
MIC3230/1/2
16-Pin Exposed Pad TSSOP (TSE)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
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Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
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© 2009 Micrel, Incorporated.
M9999-030311-D
March 2011
19
相关型号：
MIC3231YMLTR
SWITCHING CONTROLLER, 1000kHz SWITCHING FREQ-MAX, PDSO12, 3 X 3 MM, LEAD FREE, MLF-12
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