MC33215 [MOTOROLA]
Telephone Line Interface and Speakerphone Circuit; 电话线接口和扬声器电路型号: | MC33215 |
厂家: | MOTOROLA |
描述: | Telephone Line Interface and Speakerphone Circuit |
文件: | 总21页 (文件大小:294K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
Order this document by MC33215/D
52
1
FB SUFFIX
PLASTIC PACKAGE
CASE 848B
The MC33215 is developed for use in fully electronic telephone sets with
speakerphone functions. The circuit performs the ac and dc line termination,
2–4 wire conversion, line length AGC and DTMF transmission. The
speakerphone part includes a half duplex controller with signal and noise
monitoring, base microphone and loudspeaker amplifiers and an efficient
supply. The circuit is designed to operate at low line currents down to 4.0 mA
enabling parallel operation with a classical telephone set.
(TQFP–52)
• Highly Integrated Cost Effective Solution
• Straightforward AC and DC Parameter Adjustments
• Efficient Supply for Loudspeaker Amplifier and Peripherals
• Stabilized Supply Point for Handset Microphone
• Stabilized Supply Point for Base Microphone
• Loudspeaker Amplifier can be Powered and Used Separately
• Smooth Switch–Over from Handset to Speakerphone Operation
• Adjustable Switching Depth for Handsfree Operation
42
1
B SUFFIX
PLASTIC PACKAGE
CASE 858
(SDIP–42)
ORDERING INFORMATION
Operating
Temperature Range
Device
Package
MC33215FB
MC33215B
TQFP–52
SDIP–42
T = –20° to +70°C
A
Simplified Application
AC
Impedance
DC Offset
Line Current
Telephone
Line
Current
Splitter
1:10
DTMF
MF
HM
BM
V
Supply
CC
Handset
Microphone
Line
Driver
DC Slope
Base
Microphone
Attenuator
Duplex
Controller
Receive Signal
V
or
CC
External Supply
R
x
Handset Earpiece
Attenuator
LS
Base Loudspeaker
Auxiliary Input
This device contains 2782 active transistors.
This document contains information on a new product. Specifications and information herein
Motorola, Inc. 1997
Rev 0
are subject to change without notice.
MC33215
FEATURES
Line Driver and Supply
• Separate Input for DTMF and Auxiliary Signals
• Parallel Operation Down to 4.0 mA of Line Current
• AC and DC Termination of Telephone Line
• Adjustable Set Impedance for Real and Complex
Termination
• Efficient Supply Point for Loudspeaker Amplifier and
Peripherals
• Two Stabilized Supply Points for Handset and Base
Microphones
• Separate Supply Arrangement for Handset and
Speakerphone Operation
Speakerphone Operation
• Handsfree Operation via Loudspeaker and Base
Microphone
• Integrated Microphone and Loudspeaker Amplifiers
• Differential Microphone Inputs
• Loudspeaker Amplifier can be Powered and Used
Separately from the Rest of the Circuit
• Integrated Switches for Smooth Switch–Over from
Handset to Speakerphone Operation
• Signal and Background Noise Monitoring in Both
Channels
Handset Operation
• Transmit and Receive Amplifiers
• Differential Microphone Inputs
• Sidetone Cancellation Network
• Line Length AGC
• Adjustable Switching Depth for Handsfree Operation
•
Switch–Over
• Dial Tone Detector in the Receive Channel
• Microphone and Earpiece Mute
Figure 1. Pin Connections
1
2
V
42
41
40
39
38
37
36
PGD
LSO
VLS
LSB
CC
VLN
3
VHF
VMC
SLB
52
51
50
49
48
47
46
45
44
43
42
41
40
4
5
1
2
3
4
5
6
7
8
9
SLB
REG
SLP
MFI
LSF
LSF
BVO
PPL
39
38
REG
6
BVO
7
SLP
MFI
PPL 37
LSI 36
8
LSI 35
9
HM1
HM2
34
33
32
31
30
29
28
27
26
25
24
23
22
VOL
HM1
HM2
VOL 35
SWD 34
10
11
12
13
14
15
16
17
18
19
20
21
SWD
REF
AGC
Gnd
RLS
RSA
BM2
BM1
REF
AGC
33
32
BM2
BM1
SDIP–42
TQFP–52
V
DD
V
Gnd 31
RLS 30
RSA 29
RSE 28
DD
TSA
TSE
TBN
10 TSA
11 TSE
12 TBN
13 MUT
RSE
RBN
RXI
MUT
SPS
PRS
RBN
27
GRX
14
15
16
17
18
19
20
21
22
23
24
25
26
SWT
LSM
RXO
RXS
(Top View)
(Top View)
2
MOTOROLA ANALOG IC DEVICE DATA
MC33215
MAXIMUM RATINGS
Rating
Min
–0.5
–
Max
Unit
V
Peak Voltage at VLN
12
160
12
Maximum Loop Current
mA
V
Voltage at VLS (if Powered Separately)
Voltage at VHF (if Externally Applied)
Voltage at SPS, MUT, PRS, LSM
Maximum Junction Temperature
Storage Temperature Range
–0.5
–0.5
–0.5
–
5.5
7.5
150
150
V
V
°C
°C
–65
NOTE: ESD data available upon request.
RECOMMENDED OPERATING CONDITIONS
Characteristic
Min
2.4
4.0
2.4
2.4
0
Max
Unit
V
Biasing Voltage at VLN
10
130
8.0
5.0
5.0
70
Loop Current
mA
V
Voltage at VLS
Voltage at VHF (if Externally Applied)
Voltage at SPS, MUT, PRS, LSM
Operating Ambient Temperature Range
V
V
–20
°C
ELECTRICAL CHARACTERISTICS (All parameters are specified at T = 25°C, I
= 18 mA, VLS = 2.9 V, f = 1000 Hz,
line
PRS = high, MUT = high, SPS = low, LSM = high, test figure in Figure 17 with S1 in position 1, unless otherwise stated.)
Characteristic
Min
Typ
Max
Unit
DC LINE VOLTAGE
Line Voltage V
V
line
Parallel Operation, I = 4.0 mA
–
3.9
4.8
2.4
4.2
5.2
–
4.5
5.6
line
I
I
= 20 mA
= 70 mA
line
line
SUPPLY POINT V
DD
Internal Current Consumption from V
–
1.2
1.5
mA
DD
V
DD
= 2.5 V
SUPPLY POINT VMC
DC Voltage at VMC (= VMC0)
1.6
1.0
1.75
–
1.9
–
V
Current Available from VMC
VMC = VMC0 – 200 mV
mA
SUPPLY POINT VHF
DC Voltage at VHF (= VHF0)
2.6
–
2.8
1.4
3.0
2.0
V
Internal Current Consumption from VHF
VHF = VHF0 + 100 mV
mA
Current Available from VHF
VHF = VHF0 – 300 mV
2.0
–
–
mA
SUPPLY POINT V
CC
Current Available from V
13
–
15
–
mA
V
CC
V
CC
= 2.4 V, I
= 20 mA
line
DC Voltage Drop Between VLN and V
1.0
1.5
CC
I
= 20 mA
line
SUPPLY INPUT VLS
Internal Current Consumption from VLS
–
1.0
1.5
mA
3
MOTOROLA ANALOG IC DEVICE DATA
MC33215
ELECTRICAL CHARACTERISTICS (continued) (All parameters are specified at T = 25°C, I
= 18 mA, VLS = 2.9 V, f = 1000 Hz,
line
PRS = high, MUT = high, SPS = low, LSM = high, test figure in Figure 17 with S1 in position 1, unless otherwise stated.)
Characteristic
Min
Typ
Max
Unit
LOGIC INPUTS
Logic Low Level Pins PRS, MUT, SPS, LSM
Logic High Level Pins PRS, MUT, SPS, LSM
Internal Pull Up Pins PRS, MUT, LSM
Internal Pull Down Pin SPS
–
2.0
–
–
0.4
5.0
–
V
V
–
100
100
kΩ
kΩ
–
–
T CHANNEL, HANDSET MICROPHONE AMPLIFIER
x
Voltage Gain from V
to V
46
60
47
–
48
–
dB
dB
HM
line
V
HM
= 1.5 mVrms
Gain Reduction in Mute Condition
MUT = Low or PRS = Low or SPS = High
Input Impedance at HM1 and HM2
Common Mode Rejection Ratio
Total Harmonic Distortion at VLN
14
–
18
50
–
22
–
kΩ
dB
%
–
2.0
V
HM
= 4.5 mVrms
Psophometrically Weighted Noise Level at V
HM1 and HM2 Shorted with 200 Ω
–
–72
–
dBmp
dB
line
T CHANNEL, BASE MICROPHONE AMPLIFIER (SPS = HIGH, T MODE FORCED)
x
x
Voltage Gain from V
to V
53
55.5
58
BM
line
V
BM
= 0.5 mVrms
Input Impedance at BM1 and BM2
Common Mode Rejection Ratio
Total Harmonic Distortion at VLN
14
–
18
50
–
22
–
kΩ
dB
%
–
2.0
V
BM
= 1.5 mV
Psophometrically Weighted Noise Level at V
BM1 and BM2 Shorted with 200 Ω
–
–62
–
–
–
dBmp
dB
line
Gain Reduction in Mute Condition
60
MUT = Low or PRS = Low or SPS = Low
T CHANNEL, DTMF AMPLIFIER (MUT = LOW OR PRS = LOW)
x
Voltage Gain from V to V
34
35
36
dB
MF
line
V
MF
= 7.5 mVrms
Input Impedance at MFI
14
60
18
–
22
–
kΩ
Gain Reduction in Mute Condition
MUT = High or PRS = Low
dB
R CHANNEL, EARPIECE AMPLIFIER
x
Voltage Gain from V
to V
(Note 1)
EAR
23
60
24
–
25
–
dB
dB
RXI
V
line
= 20 mVrms
Gain Reduction in Mute Condition
MUT = Low or SPS = Low
Input Impedance at RXI
24
–
30
36
–
kΩ
Psophometrically Weighted Noise Level at V
130
µVrms
EAR
RXI Shorted to Gnd via 10 µF
Confidence Level During DTMF Dialing
10
15
–
20
–
mVrms
mVpp
mVpp
V
MF
= 7.5 mVrms, MUT = Low
Output Swing Capability into 150 Ω
THD ≤2%
680
Output Swing Capability into 450 Ω
1800
–
–
THD ≤2%, R
= 360 kΩ
RXO
NOTE: 1. Corresponding to –0.6 dB gain from the line to output RXO in the typical application.
4
MOTOROLA ANALOG IC DEVICE DATA
MC33215
ELECTRICAL CHARACTERISTICS (continued) (All parameters are specified at T = 25°C, I
= 18 mA, VLS = 2.9 V, f = 1000 Hz,
line
PRS = high, MUT = high, SPS = low, LSM = high, test figure in Figure 17 with S1 in position 1, unless otherwise stated.)
Characteristic
Min
Typ
Max
Unit
R CHANNEL, LOUDSPEAKER PRE–AMPLIFIER (SPS = HIGH, R MODE FORCED)
x
x
Voltage Gain from V
to V
(Note 2)
RLS
21
60
24
–
27
–
dB
dB
RXI
V
line
= 20 mVrms
Gain Reduction in Mute Condition
SPS = Low or MUT = Low
R CHANNEL, LOUDSPEAKER AMPLIFIER
x
Voltage Gain from V to V
25
26
27
dB
LSI
LSP
V
LSI
= 10 mVrms
Attenuation at Delta R
= 47 kΩ
–
–
32
–
–
dB
VOL
Psophometrically Weighted Noise Level at V
1.2
mVrms
LSP
RXI Shorted to Gnd via 10 µF
Confidence Level During DTMF Dialing
150
200
250
mVrms
V
MF
= 7.5 mVrms MUT = Low
Available Peak Current from LSO
110
1.8
–
–
–
–
mApeak
Vpp
Output Capability into 25 Ω
THD ≤2%, V = 55 mVrms
LSI
Output Capability into 25 Ω
2.7
60
–
–
–
–
Vpp
dB
THD ≤2%, V = 5.0 V, V = 90 mVrms
LS
LSI
Gain Reduction in Mute Condition
LSM = Low
R CHANNEL PEAK–TO–PEAK LIMITER
x
Peak–to–Peak Limiter Attack Time
–
–
–
–
–
300
–
5.0
–
ms
ms
%
V
LSI
Jumps from 40 mVrms to 120 mVrms
Peak–to–Peak Limiter Release Time
Jumps from 120 mVrms to 40 mVrms
V
LSI
THD at 10 dB Overdrive
= 120 mVrms
7.0
0.1
V
LSI
Peak–to–Peak Limiter Disable Threshold at PPL
–
V
AUTOMATIC GAIN CONTROL
Gain Reduction in Transmit and Receive Channel with Respect to I
= 18 mA
4.5
–
6.0
–
7.5
1.5
dB
dB
line
I
= 70 mA
line
Gain Variation in Transmit and Receive Channel with Respect to I
=18 mA with
line
AGC Disabled (AGC to V
)
DD
Highest Line Current for Maximum Gain
Lowest Line Current for Minimum Gain
BALANCE RETURN LOSS
–
–
20
50
–
–
mA
mA
Balance Return Loss with Respect to 600 Ω
SIDETONE
20
–
–
–
–
dB
dB
Voltage Gain from V
S1 in Position 2
to V
28
HM
EAR
LOGARITHMIC AMPLIFIERS AND ENVELOPE DETECTORS
Voltage Gain from RXI to RSA
18
17.5
40
20
18.5
–
22
19.5
–
dB
dB
dB
V
RXI
= 15 mVrms
Voltage Gain from BMI to TSA
= 0.5 mVrms
V
BM
Dynamic Range of Logarithmic Compression from TSA to TSE and RSA to RSE
and I from 2.5 µA to 250 µA
I
TSA
RSA
Envelope Tracking Between TSE and RSE and Between TBN and RBN
Maximum Source Current from TSE or RSE
–
±3.0
–
dB
0.3
0.4
0.5
µA
NOTE: 2. Corresponding to –0.6 dB gain from the line to output RLS in the typical application.
5
MOTOROLA ANALOG IC DEVICE DATA
MC33215
ELECTRICAL CHARACTERISTICS (continued) (All parameters are specified at T = 25°C, I
= 18 mA, VLS = 2.9 V, f = 1000 Hz,
line
PRS = high, MUT = high, SPS = low, LSM = high, test figure in Figure 17 with S1 in position 1, unless otherwise stated.)
Characteristic
Min
Typ
Max
Unit
LOGARITHMIC AMPLIFIERS AND ENVELOPE DETECTORS
Maximum Sink Current into TSE or RSE
100
0.7
100
–
–
–
1.3
–
µA
µA
Maximum Sink Current into TBN and RBN
1.0
–
Maximum Source Current from TBN or RBN
µA
Dial Tone Detector Threshold at V
20
4.5
–
mVrms
dB
line
Speech Noise Threshold Both Channels
ATTENUATOR CONTROL
–
–
Switching Depth
46
24
–
50
–
54
60
–
dB
dB
dB
µA
Adjustable Range for Switching Depth
Gain Variation in Idle Mode for Both Channels
Current Sourced from SWT
25
10
7.0
13
T Mode
x
Current Sunk into SWT
7.0
10
13
µA
R Mode
x
PIN FUNCTION DESCRIPTION
Pin
SDIP–42 TQFP–52
Name
Description
Supply Output for Loudspeaker Amplifier and Peripherals
Line Connection Input
1
2
47
48
49
50
51
52
1
V
CC
VLN
VHF
VMC
N/C
3
Supply Output for Speakerphone Section and Base Microphone
Supply Output for Handset Microphone
Not Connected
4
–
–
N/C
Not Connected
5
SLB
REG
SLP
MFI
SLP Buffered Output
6
2
Regulation of Line Voltage Adjustment
DC Slope Adjustment
7
3
8
4
DTMF Input
9
5
HM1
HM2
BM2
BM1
Handset Microphone Input 1
Handset Microphone Input 2
Base Microphone Input 2
10
11
12
13
14
15
16
17
–
6
7
8
Base Microphone Input 1
9
V
DD
Supply Input for Speech Part
Transmit Sensitivity Adjustment
Transmit Signal Envelope Timing Adjustment
Transmit Background Noise Envelope Timing Adjustment
Transmit and Receive Mute Input
Not Connected
10
11
12
13
14
15
16
17
18
19
TSA
TSE
TBN
MUT
N/C
–
N/C
Not Connected
18
19
20
21
SPS
PRS
SWT
LSM
Speakerphone Select Input
Privacy Switch Input
Switch–Over Timing Adjustment
Loudspeaker Mute Input
6
MOTOROLA ANALOG IC DEVICE DATA
MC33215
PIN FUNCTION DESCRIPTION (continued)
Pin
SDIP–42 TQFP–52
Name
N/C
Description
–
20
21
22
23
24
25
26
27
28
29
30
31
32
33
34
35
36
37
38
39
40
41
42
43
44
45
46
Not Connected
22
23
24
25
–
RXS
RXO
GRX
RXI
Receive Amplifier Stability
Receive Amplifier Output
Earpiece Amplifier Feedback Input
Receive Amplifier Input
N/C
Not Connected
–
N/C
Not Connected
26
27
28
29
30
31
32
33
34
35
36
37
38
–
RBN
RSE
RSA
RLS
Gnd
AGC
REF
SWD
VOL
LSI
Receive Background Noise Envelope Timing Adjustment
Receive Signal Envelope Timing Adjustment
Receive Sensitivity Adjustment
Receive Output for Loudspeaker Amplifier
Small Signal Ground
Line Length AGC Adjustment
Reference Current Set
Switching Depth Adjustment for Handsfree
Volume Control Adjustment
Loudspeaker Amplifier Input
Peak–to–Peak Limiter Timing Adjustment
Bias Voltage for Loudspeaker Amplifier Output
Loudspeaker Amplifier Feedback Input
Not Connected
PPL
BVO
LSF
N/C
–
N/C
Not Connected
39
40
41
42
–
LSB
VLS
LSO
PGD
N/C
Loudspeaker Amplifier Bootstrap Output
Supply Input for Loudspeaker Amplifier
Loudspeaker Amplifier Output
Power Ground
Not Connected
7
MOTOROLA ANALOG IC DEVICE DATA
MC33215
DESCRIPTION OF THE CIRCUIT
Based on the typical application circuit as given in
into a small part for biasing the internal line drive transistor
and into a large part for supplying the speakerphone. The
ratio between these two currents is fixed to 1:10. The dc set
impedance or dc setting of the telephone as created by the
line driver and its external components can be approximated
with the equivalent of a zener voltage plus a series resistor
according to:
Figure 18, the MC33215 will be described in three parts: line
driver and supplies, handset operation, and handsfree
operation. The data used refer to typical data of the
characteristics.
LINE DRIVER AND SUPPLIES
The line driver and supply part performs the ac and dc
telephone line termination and provides the necessary
supply points.
VLN
V
ILN x R
zener
slope
AC Set Impedance
With:
V
The ac set impedance of the telephone as created by the
line driver and its external components can be approximated
with the equivalent circuit shown in Figure 2.
R
REG1
0.2 x
1
10 µA x R
zener
REG1
R
REG2
ILN
R
I
– I
line
VDD
Figure 2. Equivalent of the AC impedance
R
R
SLP
11
REG1
x
1
Inductor
slope
R
REG2
Z
620
R
REG1
360 k
VDD
Z
bal
If RREG2 is not mounted, the term between the brackets
becomes equal to 1.
With the values shown in the typical application and under
the assumption that IVDD = 1.0 mA, the above formulas can
be simplified to:
C
10 n
C
100
R
2.2 k
C
220 n
R
REG
∞
VLN
VDD
µ
SLB
REG
Slope
R
SLP
11
VLN
3.8 V
3.8 V
I
– 1.0 mA x 20
Inductor
Slope
R
x C
x
line
REG1
REG
R
I
x 20
line
R
SLP
11
REG1
REG2
x
1
In the typical application this leads to a line voltage of 4.2 V
at 20 mA of line current with a slope of 20 Ω. Adding a 1.5 V
voltage drop for the diode bridge and the interruptor, the dc
voltage at tip–ring will equal 5.7 V.
If the dc mask is to be adapted to a country specific
requirement, this can be done by adjusting the resistors
RREG1 and RREG2, as a result, the zener voltage and the
slope are varied. It is not advised to change the resistor RSLP
since this changes many parameters. The influence of RREG1
and RREG2 is shown in Figure 4.
R
With the component values of the typical application, the
inductor calculates as 1.6 H. Therefore, in the audio range of
300 Hz to 3400 Hz, the set impedance is mainly determined
by ZVDD. As a demonstration, the impedance matching or
Balance Return Loss BRL is shown in Figure 3.
Figure 3. Balance Return Loss
40
35
30
25
20
15
10
5.0
0
Figure 4. Influence of RREG1 and RREG2
on the DC Mask
.
12
R
R
= 470 k
= 220 k
REG1
REG2
10
8.0
6.0
4.0
2.0
0
R
R
= 365 k
= 220 k
REG1
REG2
R
R
= 365 k
= Infinite
REG1
REG2
100
1000
10000
R
R
= 470 k
= Infinite
REG1
REG2
f, FREQUENCY (Hz)
The influence of the frequency dependent parasitic
components is seen for the lower frequencies (Inductor) and
the higher frequencies (CVLN) by a decreasing BRL value.
0
20
40
60
80
100
I
(mA)
line
DC Set Impedance
As can be seen in Figure 4, for low line currents below
10 mA, the given dc mask relations are no longer valid. This
is the result of an automatic decrease of the current drawn
The line current flowing towards the MC33215 application
is partly consumed by the circuitry connected to VDD while
the rest flows into Pin VLN. At Pin VLN, the current is split up
8
MOTOROLA ANALOG IC DEVICE DATA
MC33215
from Pin REG by the internal circuit (the 10 µA term in the
formulas). This built–in feature drops the line voltage and
therefore enables parallel operation.
The voltage over the line driver has to be limited to 12 V to
protect the device. A zener of 11 V at VLN is therefore the
maximum advised.
If, during parallel operation, a high current is required from
VMC, a 2.7 k resistor between VMC and VHF can be applied.
In Figure 5, the VMC voltage under different microphone
currents, is shown.
VHF Supply
VHF is a stabilized supply which powers the internal
duplex controller part of the MC33215, and which is also
meant to power the base microphone or other peripherals.
The base microphone however, can also be connected to
VMC, which is preferred in case of microphones with a poor
supply rejection. Another possibility is to create an additional
filter at VHF, like is shown in the typical application. The
supply capability of VHF is guaranteed as 2.0 mA for line
currents of 20 mA and greater.
Since in parallel operation not enough line current is
available to power a loudspeaker and thus having a
speakerphone working, the current internally supplied to VHF
is cut around 10 mA of line current to save current for the
handset operated part. A small hysteresis is built in to avoid
system oscillations.
When the current to VHF is cut, the voltage at VHF will
drop rapidly due to the internal consumption of 1.4 mA and
the consumption of the peripherals. When VHF drops below
2.0 V, the device internally switches to the handset mode,
neglecting the state of the speakerphone select Pin SPS.
In case an application contains a battery pack or if it is
mains supplied, speakerphone operation becomes possible
under all line current conditions. In order to avoid switch–over
to handset operation below the 10 mA, VHF has to be
supplied by this additional power source and preferably kept
above 2.4 V.
V
DD Supply
The internal circuitry for the line driver and handset
interface is powered via VDD. This pin may also be used to
power peripherals like a dialer or microcontroller. The voltage
at VDD is not internally regulated and is a direct result of the
line voltage setting and the current consumption at VDD
internally (IVDD) and externally (IPER). It follows that:
V
VLN – I
I
x R
set
DD
VDD
PER
For correct operation, it must be ensured that VDD is
biased at 1.8 V higher than SLP. This translates to a
maximum allowable voltage drop across ZVDD of
Vzener – 1.8 V. In the typical application, this results in a
maximum allowable current consumption by the peripherals
of 2.0 mA.
VMC Supply
At VMC, a stabilized voltage of 1.75 V is available for
powering the handset microphone. Due to this stabilized
supply, microphones with a low supply rejection can be used
which reduces system costs. In order to support the parallel
operation of the telephone set, the voltage at VMC will be
maintained even at very low line currents down to 4.0 mA.
Under normal supply conditions of line currents of 20 mA
and above, the supply VMC is able to deliver a guaranteed
minimum of 1.0 mA. However, for lower line currents, the
supply capability of VMC will decrease.
V
CC Supply
At VCC the major part of the line current is available for
Figure 5. VMC Under Different Microphone Loads
powering the loudspeaker amplifier and peripheral circuitry.
This supply pin should be looked at as a current source since
the voltage on VCC is not stabilized and depends on the
instantaneous line voltage and the current to and consumed
1.8
1.7
I
= 20 mA
line
from VCC
.
I
= 4.0 mA
line
1.6
1.5
1.4
1.3
1.2
1.1
1.0
2.7 k VMC–VHF
The maximum portion of the line current which is available
at VCC is given by the following relation:
10
11
I
x
I
– I
– I
– I
I
= 4.0 mA
VCC
line
VDD
VMC
VHF
line
This formula is valid when the voltage drop from VLN to
VCC is sufficient for the current splitter to conduct all this
current to VCC. When the drop is not sufficient, the current
source saturates and the surplus of current is conducted to
the power ground PGD to avoid distortion in the line driver. In
fact, when no current is drawn from VCC, the voltage at VCC
will increase until the current splitter is in balance. In Figure 6
this behavior is depicted.
0
0.2
0.4
0.6
0.8
(mA)
1.0
1.2
1.4
1.6
I
VMC
9
MOTOROLA ANALOG IC DEVICE DATA
MC33215
Figure 6. Available Current at VCC
100
90
80
70
60
50
40
30
20
10
0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
I
I
at 98% of
VCC
VCC(max)
I
I
at 50% of
VCC
VCC(max)
I
/l
(%)
VCC line
V
to VLS
Open
CC
V
CC
I
VCC(max) (mA)
0
20
40
60
80
100
0
20
40
60
80
100
I
(mA)
I
(mA)
line
line
A. Maximum Available Current at V
B. Voltage Drop to V
CC
CC
For instance, at a line current of 20 mA a maximum of
15 mA of current is available at VCC. If all this current is
taken, VCC will be 1.7 V below VLN. When not all this current
is drawn from VCC, but for instance only 1.0 mA for biasing of
the loudspeaker amplifier, the voltage at VCC will be 1.2 V
below VLN. Although the measurements for Figure 6 are
done with RREG1 = 365 k, the results are also globally valid
for other dc settings.
by adjusting the sensitivity of the handset microphone by
adjusting the resistors RHM1 and RHM2. It is not advised to
adjust the gain by including series resistors towards the Pins
HM1 and HM2.
A high pass filter is introduced by the coupling capacitors
C
HM1 and CHM2 in combination with the input impedance. A
low pass filter can be created by putting capacitors in parallel
with the resistors RHM1 and RHM2
.
As can be seen from Figure 6, the voltage at VCC is limited
by the voltage at VLN minus 1.0 V. This means that the
voltage at VCC is limited by the external zener at VLN. If it is
necessary to limit the voltage at VCC in order to protect
peripheral circuits, a zener from VCC to Gnd can be added. If
The transmit noise is measured as –72 dBmp with the
handset microphone inputs loaded with a capacitively
coupled 200 Ω. In a real life application, the inputs will be
loaded with a microphone powered by VMC. Although VMC
is a stablized supply voltage, it will contain some noise which
can be coupled to the handset microphone inputs, especially
when a microphone with a poor supply rejection is used. An
additional RC filter on VMC can improve the noise figure, see
also the base microphone section.
the supply of the loudspeaker VLS is also connected to VCC
it is advisable that VCC does not exceed 8.0 V.
,
The high efficiency of the VCC power supply contributes
to a high loudspeaker output power at moderate line
currents. More details on this can be found in the handsfree
operation paragraph.
Handset Earpiece Amplifier
The handset earpiece is to be capacitively connected to
the RXO output. Here, the receive signal is available which is
amplified from the line via the sidetone network and the Rx
and EAR amplifiers. The sidetone network attenuates the
receive signal from the line via the resistor divider composed
of RSLB and Zbal, see also the sidetone section. The
attenuation in the typical application by this network equals
24.6 dB. Then the signal from the sidetone network is
pre–amplified by the amplifier Rx with a typical gain of 6.0 dB.
This amplifier also performs the AGC and MUTE functions,
see the related paragraphs. Finally, the signal is amplified by
the noninverting voltage amplifier EAR. The overall receive
gain ARX from the line to the earpiece output then follows as:
HANDSET OPERATION
During handset operation, the MC33215 performs the
basic telephone functions for the handset microphone and
earpiece. It also enables DTMF transmission.
Handset Microphone Amplifier
The handset microphone is to be capacitively connected
to the circuit via the differential input HM1 and HM2. The
microphone signal is amplified by the HMIC amplifier and
modulates the line current by the injection of the signal into
the line driver. This transfer from the microphone inputs to the
line current is given as 15/(RSLP/11), which makes a total
transmit voltage gain AHM from the handset microphone
inputs to the line of:
V
R
R
RXO
RXO
GRX
A
A
x A
x
1
RX
ST
RXI
V
V
Z
x Z
set
line
line
HM
15
line
A
x
HM
V
Z
Z
R
11
set
line
With: AST = Attenuation of the Sidetone Network
RXI = Gain of the Pre–Amplifier Rx
SLP
With the typical application and Zline = 600 Ω the transmit
gain calculates as 47 dB.
A
For the typical application an overall gain from the line to
the earpiece is close to 0 dB.
In case an electret microphone is used, it can be supplied
from the stabilized microphone supply point VMC of 1.75 V
properly biased with resistors RHM1 and RHM2. This allows
the setmaker to use an electret microphone with poor supply
rejection to reduce total system costs. Since the transmit gain
The receive gain can be adjusted by adjusting the resistor
ratio RRXO over RGRX. However, RRXO also sets the
confidence tone level during dialing which leaves RGRX to be
chosen freely. A high pass filter is introduced by the coupling
capacitor CRXI together with the input impedance of the input
A
HM is fixed by the advised RSLP = 220 Ω and the constraints
of set impedance and line impedance, the transmit gain is set
10
MOTOROLA ANALOG IC DEVICE DATA
MC33215
RXI. A second high pass filtering is introduced by the
Automatic Gain Control
combination of CGRX and RGRX. A low pass filter is created
by CRXO and RRXO. The coupling capacitor at the output
RXO is not used for setting a high pass filter but merely for dc
decoupling.
In combination with dynamic ear capsules, the EAR
amplifier can become unstable due to the highly inductive
characteristic of some of the capsules. To regain stability, a
100 nF capacitor can be connected from RXS to Gnd in
those cases. An additional 10 nF at the RXI input, as shown
in the typical application, improves the noise figure of the
receiver stage.
To obtain more or less constant signal levels for transmit
and receive regardless of the telephone line length, both the
transmit and receive gain can be varied as a function of line
current when the AGC feature is used. The gain reduction as
a function of line current, and thus line length, is depicted in
Figure 8.
Figure 8. Automatic Gain Control
0
–1.0
R
= 20 k
AGC
–2.0
Sidetone Cancellation
R
= 30 k
AGC
The line driver and the receiver amplifier of the MC33215
are tied up in a bridge configuration as depicted in Figure 7.
This bridge configuration performs the so–called hybrid
function which, in the ideal case, prevents transmitted signals
from entering the receive channel.
–3.0
–4.0
–5.0
–6.0
Figure 7. Sidetone Bridge
0
10
20
30
40
(mA)
50
60
70
VLN
I
line
For small line currents, and thus long lines, no gain
reduction is applied and thus the transmit and receive gains
are at their maximum. For line currents higher than Istart, the
gain is gradually reduced until a line current Istop is reached.
This should be the equivalent of a very short line, and the
gain reduction equals 6.0 dB. For higher line currents the
gain is not reduced further. For the start and stop currents the
following relations are valid:
Z
//Z
Z
bal
line set
V
x 15
11
HM
RXI
Receive
Transmit
R
Gnd
SLP
Gnd
R
/11
R
SLB
SLP
SLP
1
I
stop
As can be seen from Figure 7 by inspection, the receiver
will not pick up any transmit signal when the bridge is in
balance, that is to say when:
R
11
SLP
20 µ x R
AGC
11
1
Z
Z
Z
I
–
set
11
start
bal
line
R
11
R
SLP
SLP
R
R
SLB
SLP
For the typical application, where RAGC = 30 kΩ, the gain
will start to be reduced at Istart = 20 mA while reaching 6.0 dB
of gain reduction at Istop = 50 mA. When AGC is connected to
The sidetone suppression is normally measured in an
acoustic way. The signal at the earpiece when applying a
signal on the microphone is compared with the signal at the
earpiece when applying a signal on the line. The suppression
takes into account the transmit and receive gains set. In fact
the sidetone suppression can be given as a purely electrical
parameter given by the properties of the sidetone bridge
itself. For the MC33215, this so–called electrical sidetone
suppression ASTE can be given as:
V
DD, the AGC function is disabled leading to no gain
reduction for any line current. This is also sometimes called
PABX mode.
The automatic gain control takes effect in the HMIC and Rx
amplifiers as well as in the BMIC amplifier. In this way the
AGC is also active in speakerphone mode, see the handsfree
operation paragraph.
Z
R
11
bal
SLP
Privacy and DTMF Mode
During handset operation a privacy and a DTMF mode can
be entered according the logic Table 1.
A
1 –
x
STE
R
Z
Z
set
SLB
line
Values of –12 dB or better, thus ASTE < 0.25, can easily be
reached in this way.
Table 1. Logic Table for Handset Mode
Logic Inputs
Amplifiers
DTMF
SPS
MUT
PRS
HMIC
On
BMIC
Off
R
RX
EAR
On
Mode
x
att
0
0
0
1
1
0
1
0
Handset Normal
Handset Privacy
Handset DTMF
Off
On
On
On
On
Off
Off
Off
Off
Off
Off
On
X
Off
Off
On
11
MOTOROLA ANALOG IC DEVICE DATA
MC33215
Table 2. Logic Table for Handsfree Mode
Logic Inputs
Amplifiers
DTMF
SPS
MUT
PRS
HMIC
Off
BMIC
On
R
RX
EAR
Off
Mode
x
att
1
1
1
1
1
0
1
0
Handsfree Normal
Handsfree Privacy
Handsfree DTMF
Off
On
On
On
On
Off
On
Off
Off
On
On
Off
X
Off
Off
Off
By applying a logic 0 to Pin MUT, the DTMF mode is
entered where the DTMF amplifier is enabled and where the
Rx amplifier is muted. A DTMF signal can be sent to the line
via the MFI input for which the gain ADTMF is given as:
With the typical application and Zline = 600 Ω the transmit
gain calculates as 55 dB.
The electret base microphone can be supplied directly
from VHF but it is advised to use an additional RC filter to
obtain a stable supply point, as shown in the typical
application. The microphone can also be supplied by VMC.
The transmit gain is set by adjusting the sensitivity of the
V
Z
x Z
set
line
MFI
3.75
SLP
line
A
x
DTMF
V
Z
Z
R
11
set
line
In the typical application, the gain equals 35 dB. The
DTMF gain can be controlled by a resistor divider at the input
MFI as shown in the typical application. The signal has to be
capacitively coupled to the input via CMFI which creates a
high pass filter with the input impedance. The line length
AGC has no effect on the DTMF gains.
base microphone by adjusting the resistors RBM1 and RBM2
It is not advised to adjust the gain by including series
resistors towards the Pins BM1 and BM2.
.
A high pass filter is introduced by the coupling capacitors
C
BM1 and CBM2 in combination with the input impedance. A
low pass filter can be created by putting capacitors in parallel
The signal applied to the MFI input is made audible at the
earpiece output for confidence tone. The signal is internally
applied to the GRX pin where it is amplified via the EAR
amplifier which is used as a current to voltage amplifier. The
with the resistors RBM1 and RBM2
.
Loudspeaker Amplifier
The loudspeaker amplifier of the MC33215 has three major
benefits over most of the existing speakerphone loudspeaker
amplifiers: it can be supplied and used in a telephone line
powered application but also stand alone, it has an all NPN
bootstrap output stage which provides maximum output
swing under any supply condition, and it includes a
peak–to–peak limiter to limit the distortion at the output.
The loudspeaker amplifier is powered at Pin VLS. In
telephone line powered applications, this pin should be
connected to VCC where most of the line current is available,
see the VCC supply paragraph. In an application where an
external power supply is used, VLS and thus the loudspeaker
amplifier can be powered separately from the rest of the
circuit. The amplifier is grounded to PGD, which is the circuits
power ground shared by both the loudspeaker amplifier and
the current splitter of the VCC supply. Half the supply voltage
of VLS is at BVO, filtered with a capacitor to VLS. This
voltage is used as the reference for the output amplifier.
The receive signal present at RLS can be capacitively
coupled to LSI via the resistor RLSI. The overall gain from
RLS to LSO follows as:
gain is therefore proportional to the feedback resistor RRXO
.
For RRXO = 180 kΩ the gain equals 6.0 dB. The confidence
tone is also audible at the loudspeaker output when the
loudspeaker amplifier is activated, see speakerphone
operation.
By applying a logic 0 to Pin PRS, the MC33215 enters
privacy mode. In this mode, both handset and handsfree
microphone amplifiers are muted while the DTMF amplifier is
enabled. Through the MFI input, a signal, for example music
on hold, can be sent to the line. In the same way, the MFI
input can also be used to couple in signals from, for instance,
an answering machine.
HANDSFREE OPERATION
Handsfree operation, including DTMF and Privacy modes,
can be performed by making Pin SPS high according Table 2.
The handset amplifiers will be switched off while the base
amplifiers will be activated. The MC33215 performs all the
necessary functions, such as signal monitoring and
switch–over, under supervision of the duplex controller.
With the MC33215 also a group listening–in application
can be built. For more information on this subject please refer
to application note AN1574.
V
R
LSO
RLS
LSF
A
–
x 4.0
LS
V
R
LSI
In the typical application this leads to a loudspeaker gain
LS of 26 dB. The above formula follows from the fact that the
overall amplifier architecture from RLS to LSO can be looked
at as an inverting voltage amplifier with an internal current
gain from LSI to LSF of 4. The input LSI is a signal current
summing node which allows other signals to be applied here.
Base Microphone Amplifier
A
The base microphone can be capacitively connected to
the circuit via the differential input BM1 and BM2. The setup
is identical to the one for the handset microphone amplifier.
The total transmit voltage gain ABM from the base
microphone inputs to the line is:
V
Z
x Z
set
line
BM
37.5
SLP
line
A
x
BM
V
Z
Z
R
11
set
line
12
MOTOROLA ANALOG IC DEVICE DATA
MC33215
Figure 9. Loudspeaker Output Stage
0.5 VLS
0
1.5 VLS
VLS
VLS
–0.5 VLS
0.5 VLS
Loudspeaker
LSB
VLS
C
LSO
T2
LSO
PGD
T1
VLS
0.5 VLS
0
Figure 10. Loudspeaker Amplifier Output Power with External Supply
140
120
100
80
300
R
R
= 25
= 50
Ω
Ω
LSP
250
R
= 25 Ω
LSP
200
150
100
60
40
20
0
LSP
R
= 50 Ω
LSP
50
0
2.0
4.0
6.0
2.0
3.0
5.0
7.0
3.0
5.0
7.0
8.0
4.0
6.0
8.0
VLS (V)
VLS (V)
A. Peak–to–Peak Limiter Active
B. Peak–to–Peak Limiter Disabled
The total gain from the telephone line to the loudspeaker
output includes, besides the loudspeaker amplifier gain, also
the attenuation of the sidetone network and the internal gain
from RXI to RLS. When in receive mode, see under duplex
controller, the gain from RXI to RLS is maximum and equals
24 dB at maximum volume setting. The attenuation of the
sidetone network in the typical application equals 24.6 dB
which makes an overall gain from line to loudspeaker of
25.4 dB. Due to the influence of the line length AGC on the Rx
amplifier, the gain will be reduced for higher line currents.
The output stage of the MC33215 is a modified all NPN
bootstrap stage which ensures maximum output swing under
all supply conditions. The major advantage of this type of
output stage over a standard rail–to–rail output is the higher
stability. The principle of the bootstrap output stage is
explained with the aid of Figure 9.
T2 to be supplied for output signals with positive excursions
up to VLS without distorting the output signal. The resulting
ac signal over the loudspeaker will equal the signal at LSO.
As an indication of the high performance of this type of
amplifier, in Figure 10, the output power of the loudspeaker
amplifier as a function of supply voltage is depicted for 25 Ω
and 50 Ω loads with both the peak–to–peak limiter active and
disabled. As can be seen, in case the peak–to–peak limiter is
disabled, the output power is roughly increased with 6.0 dB,
this at the cost of increased distortion levels up to 30%.
In a telephone line powered application, the loudspeaker
amplifier output power is limited not only by the supply
voltage but also by the telephone line current. This means
that in telephones the use of 25 Ω or 50 Ω speakers is
preferred over the use of the cheaper 8.0 Ω types. Figure 11
gives the output power into the loudspeaker for a line
powered application and two different dc settings with the
peak–to–peak limiter active. In case the peak–to–peak limiter
is disabled the output power will be increased for the higher
line currents up to 6.0 dB.
The output LSO is biased at half the supply VLS while the
filtering of the loudspeaker with the big capacitor CLSO
requires that LSB is biased at VLS. In fact, because of the
filtering, LSB is kept at VLS/2 above the LSO output even if
LSO contains an ac signal. This allows the output transistor
13
MOTOROLA ANALOG IC DEVICE DATA
MC33215
Figure 11. Loudspeaker Amplifier Output
Power when Line Powered
loudspeaker amplifier is muted which is needed for correct
handset operation.
100
90
The volume of the loudspeaker signal can be varied via a
potentiometer at VOL. A fixed current of 10 µA is running
through the potentiometer and the resulting voltage at VOL
is a measure for the gain reduction. The relation between
the voltage at VOL and the obtained gain reduction is given
in Figure 13.
R
R
R
= 365 k
= 220 k
REG1
REG2
LSP
= 25
Ω
80
R
R
R
= 365 k
= 220 k
REG1
REG2
LSP
70
R
R
R
= 365 k
= Infinite
= 50
Ω
REG1
REG2
LSP
60
50
40
30
20
10
0
= 25
Ω
R
R
R
= 365 k
= Infinite
REG1
REG2
LSP
Figure 13. Volume Reduction
0
= 50
Ω
–5.0
–10
0
20
40
60
80
100
–15
–20
I
(mA)
line
The quality of the audio output of the loudspeaker amplifier
is mainly determined by the distortion level. To keep high
quality under difficult supply conditions, the MC33215
incorporates a peak–to–peak limiter. The peak–to–peak
limiter will detect when the output stage gets close to its
maximum output swing and will then reduce the gain from LSI
to LSF. The attack and release of the limiter is regulated by
the CPPL capacitor. Figure 12 depicts the limiter’s attack
behavior with CPPL = 100 nF. The release time is given as
3 x CPPL x RPPL. In the typical application this leads to a
release time of 300 ms.
–25
–30
–35
–40
0
100
200
(mV), dA
300
(dB)
400
500
V
VOL
LSP
It can be seen from Figure 13 that a linear variation of
RVOL will give a logarithmic gain reduction which adapts
better to the human ear than a linear gain reduction.
During DTMF dialing, see Table 2, a confidence tone is
audible at the loudspeaker of which the level is proportional
to the feedback resistor RLSF only. At RLSF = 180 kΩ the gain
from MFI to LSO equals 28.5 dB.
Figure 12. Peak–to–Peak Limiter Dynamic Behavior
Half Duplex Controller
V
LSO
To avoid howling during speakerphone operation, a half
duplex controller is incorporated. By monitoring the signals in
both the transmit and receive channel the duplex controller
will reduce the gain in the channel containing the smallest
signal. A typical gain reduction will be between 40 dB and
52 dB depending on the setting, see below. In case of equal
signal levels or by detection of noise only, the circuit goes into
idle mode. In this mode the gain reduction in both channels is
halfway, leading to 20 dB to 26 dB of reduction.
V
PPL
V
in
0
2.0
5.0
In a speakerphone built around the MC33215, following
the signal path from base microphone to the line and via
sidetone, loudspeaker and acoustic coupling back to the
microphone, the loop gain can be expressed as a sum of the
gains of the different stages. However, since the transmit and
receive gains are dependent on AGC and the sidetone
suppression is dependent on matching with the different lines
we are mostly interested by the maximum possible loop gain
1.0
3.0
4.0
6.0
t, TIME (ms)
Figure 12 clearly shows that due to the action of the
peak–to–peak limiter, the output swing and thus the output
power is reduced with respect to the maximum possible as
already indicated in Figure 10. The peak–to–peak limiter can
be disabled by connecting the PPL pin to ground.
On top of the peak–to–peak limiter, the MC33215
incorporates a supply limiter, which reduces the gain rapidly
when the supply voltage VLS drops too much. This will
avoid malfunctioning of the amplifier and unwanted
oscillations. The voltage drop is detected via the BVO input
and for that reason the CBVO has to be connected to VLS
and not to Gnd.
The amplifier can be activated by making Pin LSM high. In
the typical application this pin is connected to SPS, which
activates the loudspeaker amplifier automatically when the
speakerphone mode is entered. When LSM is made low, the
A
LOOP(max). It follows:
A
LOOP(max) = ABMRX(max) + ARXBM(max) – ASWD (dB)
With: ABMRX(max) = Maximum gain from BM1 and BM2 to
RXI as a function of line length AGC and line
impedance matching
ARXBM(max) = Maximum gain from RXI to BM1 and
BM2 as a function of line length AGC and acoustic
coupling
14
MOTOROLA ANALOG IC DEVICE DATA
MC33215
ASWD = Switching depth as performed in the
attenuators
mode, due to the coupling of the high loudspeaker signal, is
automatically taken into account.
In the table, two particulars can be found. At first, the set
will go to idle mode if the signals are not at least 4.5 dB
greater then the noise floor, which calculates as a 13 mV
voltage difference in envelopes. This avoids continuous
switching over between the modes under slight variations of
the background noise due to, for instance, typing on a
keyboard. Second, a dial tone detector threshold is
implemented to avoid that the set goes to idle mode in
presence of a continuous strong receive signal like a dial
tone. The dial tone detector threshold is proportional to the
RRSA resistor. In the typical application with RRSA = 3.3 kΩ,
the threshold is at 1.26 mVrms at the input RXI or 20 mVrms
at the line. Line length AGC is of influence on the dial tone
detector threshold, increasing the level depending on the line
current with a maximum of 6.0 dB.
To avoid howling, the maximum possible loop gain should
be below 0 dB and preferably below –10 dB for comfort. In a
practical telephone design, both the ABMRX(max) and the
A
RXBM(max) will be less than 20 dB thus a switching depth of
50 dB will give a loop gain of less than –10 dB. An optimized
sidetone network is of high importance for handsfree
operation. The better the network matches with the
telephone line the less local feedback and the smaller the
switching range can be.
The amount of gain reduction ASWD obtained by the
duplex controller is set via resistor RSWD according:
2
3.6 x R
SWD
A
20 log
(dB)
SWD
R
REF
In order to perform a correct comparison between the
signal strengths, the sensitivity of the envelope detectors can
be adjusted via the resistors connected to TSA and RSA.
Based on the above, and on the fact that there is an effective
gain of 20 dB in the transmit monitor, it can be derived that for
stable operation the following two relations are valid:
In the typical application the gain reduction will be 50 dB.
To compare the transmit and receive signals with each
other, they have to be monitored. This is done by making a
signal envelope and a background noise envelope via the
TSE, CTBN capacitors for the transmit channel and via the
RSE, CRBN capacitors for the receive channel. In Figure 14,
C
C
20 log R
20 log R
– A
20 (dB)
a schematic behavior of the envelopes is depicted which is
equal for both transmit and receive.
TSA
20 log R
RSA
20 log R
BMRX(max)
– A
The voltage signal at the input is first transferred to a
current via the sensitivity adjust network. Then this current is
led through a diode which gives a logarithmic compression in
voltage. It is this voltage from which the signal envelope is
created by means of asymmetric charge and discharge of the
signal envelope capacitor. The noise envelope voltage then
follows in a similar way. Based on the envelope levels, the
MC33215 will switch to transmit, receive or idle mode
following Table 3. The fact that in receive mode the signal on
the base microphone is greater than it is in case of transmit
TSA
RSA
RXBM(max)
– A
20 (dB)
SW
By measuring the gains and choosing the RRSA, the limits
for RTSA follow. The choice for the sensitivity resistors is not
completely free. The logarithmic compressors and the
amplifier stages have a certain range of operation and, on the
receive side, the choice for RRSA is given by the desired dial
tone detector threshold. Figure 15 indicates the available
dynamic range for the selected value of the sensitivity
resistors.
Figure 14. Signal and Noise Envelopes
1.8 V
Internal
VHF
VHF
C
TSE
C
TSE
TBN
TBN
Microphone
Input Signal
TSA
R
TSA
C
TSA
15
MOTOROLA ANALOG IC DEVICE DATA
MC33215
Table 3. Logic Table for Switch–Over
TSE > RSE
TSE > TBN + 13 mV
RSE > V
RSE > RBN + 13 mV
Mode
DDT
1
1
0
0
0
1
0
X
X
1
0
0
X
X
X
1
Transmit
Idle
X
X
X
Receive
Receive
Idle
0
The resistors for the sensitivity setting have to be coupled
capacitively to the pins for dc decoupling, and also to create
a high pass filter to suppress low frequent background noises
like footsteps and 50 Hz.
The switch–over timing is performed by charging and
discharging the CSWT capacitor. The switch–over from
transmit to receive or vice versa is fast, on the order of
milliseconds, and is proportional to the value of CSWT. The
switch–over to idle mode is slow, in the order of a few
seconds, and is proportional to the product of the values of
RSWT and CSWT. Figure 16 depicts a typical switch–over
behavior when applying transmit and receive stimuli.
The electrical characteristics and the behavior of the
MC33215 are not the only factor in designing a handsfree
speakerphone. During the design the acoustics have to be
taken into account from the beginning. The choice of the
transducers and the design of the cabinet are of great
influence on the speakerphone performance. Also, to
achieve a proper handsfree operation, the fine tuning of the
components around the duplex controller have to be done
with the final choice of the cabinet and the transducers.
Figure 15. Compression Range of the Signal Monitors
100.0E–3
10.0E–3
1.0E–3
100.0E–3
Upper Limit of
Compression
Upper Limit of
Compression
Dial Tone
Threshold
10.0E–3
Lower Limit of
Compression
Lower Limit of
Compression
1.0E–3
100.0E–6
10.0E–6
100.0E–6
1000
10000
100000
1000
10000
100000
100
100
R
(Ω)
R
(Ω)
RSA
TSA
A. Receive Monitor
B. Transmit Monitor
Figure 16. Switch–Over Behavior
Receive
Transmit
VMC + 0.5
VMC – 0.5
SWT
16
MOTOROLA ANALOG IC DEVICE DATA
MC33215
Figure 17. Test Circuit
Z
VDD
620
Z
33 k
R
360 k
bal
REG
600
Vac
C
100
C
REG
220 n
VDD
µ
V
V
V
line
DD
I
line
R
2.2 k
SLB
Gnd
V
VLN
REG
DD
VMC
VHF
C
C
C
VMC
10
µ
Supply
1:10
Supply
MC33215
VHF
V
47
µ
CC
C
HM1
MHM
VCC
µ
33 n
HM1
HM2
470
Driver
SLB
SLP
1x
HMIC
BMIC
V
HM
R
SLP
0.2 V
MDF
220
C
HM2
AGC
MBM
33 n
C
BM1
33 n
BM1
BM2
C
MF1
47 n
MFI
C
BM2
V
T Attenuator
BM
x
DTMF
33 n
V
MF
SPS
MUT
PRS
LSM
AGC
C
470 n
R
TSA
2.2 k
TSA
V
SPS
TSA
TSE
MHM
MBM
MDF
MRX
MRA
T
Log–Amp
and
Envelope
Detectors
x
V
MUT
Logic
Control
Block
C
TSE
VHF
330 n
V
PRS
TBN
C
TBN
4.7
AGC
30 k
µ
V
M
LSM
EAR
Attenuator
Control
R
SWT
RBN
AGC
REF
C
4.7
RBN
µ
R
V
REF
SWT
20 k
AGC
R
Log–Amp
x
C
R
RSE
VHF
SWD
100 k
Analog
Control
Block
and Envelope
Detectors
330 n
SWD
VOL
RSE
RSA
C
RSA
470 n
R
3.3 k
RSA
R
47 k
MRX
VOL
RLS
2
RXI
R Attenuator
x
R
x
S1
C
47 n
1
RXI
V
V
RLS
BVO
VLS
V
RXI
AGC
C
BVO
MRA
220 n
M
EAR
VLS
V
LSP
25
C
EAR
V
LSB
LSO
10
µ
RXO
GRX
C
47
EAR
LSO
µ
LSP
Peak
Limiter
R
V
RXO
180 k
V
EAR
PGD
R
LSO
180 k
R
24 k
GRX
LSF
PPL
LSI
RXS
C
R
R
LSI
C
100 n
C
47 n
PPL
1.0 M
PPL
RXS
GRX
36 k
100 n
C
LSI
47 n
V
LSI
17
MOTOROLA ANALOG IC DEVICE DATA
MC33215
Figure 18. Typical Application
T1
R
REG2
Z
R
365 k
VDD
620
REG1
Z1
10 V
Z
bal
33 k
0.01
Hook
Switch
C
VDD
100
C
V
REG
220 n
DD
µ
R
2.2 k
SLB
Gnd
V
REG
VLN
DD
VMC
VHF
T2
C
10
VMC
µ
Supply
1:10
Supply
V
DD
MC33215
C
VHF
VMC
47
µ
V
CC
C
R
HM1
33 n
C
470
HM1
1.0 k
MHM
VCC
µ
HM1
HM2
Driver
SLB
SLP
1x
C
HM2
HMIC
33 n
R
220
SLP
0.2 V
R
HM2
1.0 k
VHF
AGC
MBM
Tip
C
BM1
33 n
R
1.0 k
BM1
1.0 k
MDF
BM1
BM2
C
MF1
47 n
MFI
C
BM2
33 n
T
Attenuator
10 µF
BMIC
x
DTMF
Ring
Dialer or
Microcontroller
R
1
4
7
*
2
5
8
0
3
6
9
#
BM2
1.0 k
SPS
MUT
AGC
R
TSA
470
TSA
TSE
MHM
MBM
C
TSA
1.0
Privacy
Button
Speakerphone
Button
T
Log–Amp
µ
F
x
and Envelope
Detectors
MDF
MRX
MRA
PRS
LSM
Logic Control
Block
C
TSE
VHF
330 n
TBN
C
TBN
4.7
C
VMC
SWT
M
µ
EAR
100 n
SWT
Attenuator
Control
R
AGC
30 k
AGC
REF
R
R
C
4.7
SWT
REF
20 k
RBN
µ
2.2 M
RBN
AGC
C
330 n
R
100 k
RSE
SWD
Analog
Control
Block
R
Log–Amp
x
VHF
SWD
VOL
RSE
RSA
and Envelope
Detectors
C
RSA
470 n
R
50 k
R
3.3 k
VOL
RSA
MRX
RLS
RXI
R
Attenuator
MRA
R
x
x
C
RXI
33 n
10 n
BVO
VLS
AGC
C
BVO
220 n
V
CC
M
EAR
C
EAR
10
LSB
LSO
µ
RXO
GRX
25
Ω
C
R
LSO
LSP
EAR
47
µ
R
Peak Limiter
RXO
180 k
150 Ω
PGD
LSF
LSO
180 k
R
24 k
GRX
PPL
LSI
RXS
C
C
47 n
R
C
100 n
RXS
GRX
PPL
1.0 M
PPL
100 n
C
R
RLS
33 n
LSI
36 k
18
MOTOROLA ANALOG IC DEVICE DATA
MC33215
OUTLINE DIMENSIONS
FB SUFFIX
PLASTIC PACKAGE
CASE 848B–04
(TQFP–52)
ISSUE C
B
B
L
39
27
26
40
–A–, –B–, –D–
DETAIL A
DETAIL A
–B–
–A–
L
F
J
N
14
13
52
1
BASE METAL
D
–D–
M
S
S
B
0.02 (0.008)
C
A–B
D
M
S
S
0.20 (0.008)
H
A–B
D
D
SECTION B–B
0.05 (0.002) A–B
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
V
M
S
S
0.20 (0.008)
C
A–B
2. CONTROLLING DIMENSION: MILLIMETER.
3. DATUM PLANE –H– IS LOCATED AT BOTTOM OF
LEAD AND IS COINCIDENT WITH THE LEAD WHERE
THE LEAD EXITS THE PLASTIC BODY AT THE
BOTTOM OF THE PARTING LINE.
4. DATUMS –A–, –B– AND –D– TO BE DETERMINED AT
DATUM PLANE –H–.
5. DIMENSIONS S AND V TO BE DETERMINED AT
SEATING PLANE –C–.
DETAIL C
M
C
E
6. DIMENSIONS A AND B DO NOT INCLUDE MOLD
PROTRUSION. ALLOWABLE PROTRUSION IS 0.25
(0.010) PER SIDE. DIMENSIONS A AND B DO
INCLUDE MOLD MISMATCH AND ARE DETERMINED
AT DATUM PLANE –H–.
7. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR PROTRUSION
SHALL BE 0.08 (0.003) TOTAL IN EXCESS OF THE D
DIMENSION AT MAXIMUM MATERIAL CONDITION.
DAMBAR CANNOT BE LOCATED ON THE LOWER
RADIUS OR THE FOOT.
DATUM
–H–
PLANE
0.10 (0.004)
H
SEATING
PLANE
–C–
M
G
MILLIMETERS
INCHES
DIM
A
B
C
D
E
MIN
9.90
9.90
2.10
0.22
2.00
0.22
MAX
10.10
10.10
2.45
0.38
2.10
MIN
MAX
0.398
0.398
0.096
0.015
0.083
0.013
0.390
0.390
0.083
0.009
0.079
0.009
U
F
0.33
G
H
J
K
L
0.65 BSC
0.026 BSC
–––
0.13
0.65
0.25
0.23
0.95
–––
0.005
0.026
0.010
0.009
0.037
R
Q
7.80 REF
0.307 REF
M
N
Q
R
S
T
U
V
5
0.13
0
0.13
12.95
0.13
0
12.95
0.35
1.6 REF
10
0.17
7
0.30
13.45
–––
–––
13.45
0.45
5
0.005
0
0.005
0.510
0.005
0
0.510
0.014
0.063 REF
10
0.007
7
0.012
0.530
–––
–––
0.530
0.018
K
T
W
X
DETAIL C
W
X
19
MOTOROLA ANALOG IC DEVICE DATA
MC33215
OUTLINE DIMENSIONS
B SUFFIX
PLASTIC PACKAGE
CASE 858–01
(SDIP–42)
ISSUE O
NOTES:
–A–
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
3. DIMENSION L TO CENTER OF LEAD WHEN
FORMED PARALLEL.
4. DIMENSIONS A AND B DO NOT INCLUDE MOLD
FLASH. MAXIMUM MOLD FLASH 0.25 (0.010).
42
22
21
–B–
INCHES
MILLIMETERS
1
DIM
A
B
C
D
F
MIN
MAX
1.465
0.560
0.200
0.022
0.046
MIN
36.45
13.72
3.94
0.36
0.81
MAX
37.21
14.22
5.08
0.56
1.17
L
1.435
0.540
0.155
0.014
0.032
H
C
G
H
J
K
L
0.070 BSC
0.300 BSC
1.778 BSC
7.62 BSC
0.008
0.115
0.015
0.135
0.20
2.92
0.38
3.43
–T–
SEATING
PLANE
0.600 BSC
15.24 BSC
N
G
M
N
0
15
0
0.51
15
1.02
M
F
K
0.020
0.040
J 42 PL
0.25 (0.010)
D 42 PL
M
S
T
B
M
S
0.25 (0.010)
T A
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the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and
specificallydisclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters which may be provided in Motorola
datasheetsand/orspecificationscananddovaryindifferentapplicationsandactualperformancemayvaryovertime. Alloperatingparameters,including“Typicals”
must be validated for each customer application by customer’s technical experts. Motorola does not convey any license under its patent rights nor the rights of
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applicationsintended to support or sustain life, or for any other application in which the failure of the Motorola product could create a situation where personal injury
ordeathmayoccur. ShouldBuyerpurchaseoruseMotorolaproductsforanysuchunintendedorunauthorizedapplication,BuyershallindemnifyandholdMotorola
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arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that
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