CRCW12061212F [NSC]
N-Channel FET Synchronous Buck Regulator Controller for Low Output Voltages; N沟道FET同步降压稳压器控制器的低输出电压
| 型号： | CRCW12061212F |
| 厂家： | 
National Semiconductor |
| 描述： | N-Channel FET Synchronous Buck Regulator Controller for Low Output Voltages |
| 文件： | 总22页 (文件大小：597K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
June 2003
LM2727/LM2737
N-Channel FET Synchronous Buck Regulator Controller
for Low Output Voltages
General Description
Features
n Input power from 2.2V to 16V
The LM2727 and LM2737 are high-speed, synchronous,
switching regulator controllers. They are intended to control
currents of 0.7A to 20A with up to 95% conversion efficien-
cies. The LM2727 employs output over-voltage and under-
voltage latch-off. For applications where latch-off is not de-
sired, the LM2737 can be used. Power up and down
sequencing is achieved with the power-good flag, adjustable
soft-start and output enable features. The LM2737 and
LM2737 operate from a low-current 5V bias and can convert
from a 2.2V to 16V power rail. Both parts utilize a fixed-
frequency, voltage-mode, PWM control architecture and the
switching frequency is adjustable from 50kHz to 2MHz by
adjusting the value of an external resistor. Current limit is
achieved by monitoring the voltage drop across the on-
resistance of the low-side MOSFET, which enhances low
duty-cycle operation. The wide range of operating frequen-
cies gives the power supply designer the flexibility to fine-
tune component size, cost, noise and efficiency. The adap-
tive, non-overlapping MOSFET gate-drivers and high-side
bootstrap structure helps to further maximize efficiency. The
high-side power FET drain voltage can be from 2.2V to 16V
and the output voltage is adjustable down to 0.6V.
n Output voltage adjustable down to 0.6V
n Power Good flag, adjustable soft-start and output enable
for easy power sequencing
n Output over-voltage and under-voltage latch-off
(LM2727)
n Output over-voltage and under-voltage flag (LM2737)
n Reference Accuracy: 1.5% (0˚C - 125˚C)
n Current limit without sense resistor
n Soft start
n Switching frequency from 50 kHz to 2 MHz
n TSSOP-14 package
Applications
n Cable Modems
n Set-Top Boxes/ Home Gateways
n DDR Core Power
n High-Efficiency Distributed Power
n Local Regulation of Core Power
Typical Application
20049410
© 2003 National Semiconductor Corporation
DS200494
www.national.com
Connection Diagram
20049411
14-Lead Plastic TSSOP
θJA = 155˚C/W
NS Package Number MTC14
EAO (Pin 8) - Output of the error amplifier. The voltage level
Pin Description
on this pin is compared with an internally generated ramp
signal to determine the duty cycle. This pin is necessary for
compensating the control loop.
BOOT (Pin 1) - Supply rail for the N-channel MOSFET gate
drive. The voltage should be at least one gate threshold
above the regulator input voltage to properly turn on the
high-side N-FET.
SS (Pin 9) - Soft start pin. A capacitor connected between
this pin and ground sets the speed at which the output
voltage ramps up. Larger capacitor value results in slower
output voltage ramp but also lower inrush current.
LG (Pin 2) - Gate drive for the low-side N-channel MOSFET.
This signal is interlocked with HG to avoid shoot-through
problems.
FB (Pin 10) - This is the inverting input of the error amplifier,
which is used for sensing the output voltage and compen-
sating the control loop.
PGND (Pins 3, 13) - Ground for FET drive circuitry. It should
be connected to system ground.
SGND (Pin 4) - Ground for signal level circuitry. It should be
connected to system ground.
FREQ (Pin 11) - The switching frequency is set by connect-
ing a resistor between this pin and ground.
VCC (Pin 5) - Supply rail for the controller.
SD (Pin 12) - IC Logic Shutdown. When this pin is pulled low
the chip turns off the high side switch and turns on the low
side switch. While this pin is low, the IC will not start up. An
PWGD (Pin 6) - Power Good. This is an open drain output.
The pin is pulled low when the chip is in UVP, OVP, or UVLO
mode. During normal operation, this pin is connected to VCC
or other voltage source through a pull-up resistor.
internal 20µA pull-up connects this pin to VCC
.
HG (Pin 14) - Gate drive for the high-side N-channel MOS-
FET. This signal is interlocked with LG to avoid shoot-
through problems.
ISEN (Pin 7) - Current limit threshold setting. This sources a
fixed 50µA current. A resistor of appropriate value should be
connected between this pin and the drain of the low-side
FET.
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2
Absolute Maximum Ratings (Note 1)
Infrared or Convection (20sec)
ESD Rating
235˚C
2 kV
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Operating Ratings
VCC
7V
21V
Supply Voltage (VCC
Junction Temperature Range
Thermal Resistance (θJA
)
4.5V to 5.5V
−40˚C to +125˚C
155˚C/W
BOOTV
Junction Temperature
Storage Temperature
Soldering Information
Lead Temperature
(soldering, 10sec)
150˚C
)
−65˚C to 150˚C
260˚C
Electrical Characteristics
VCC = 5V unless otherwise indicated. Typicals and limits appearing in plain type apply for TA=TJ=+25˚C. Limits appearing in
boldface type apply over full Operating Temperature Range. Datasheet min/max specification limits are guaranteed by design,
test, or statistical analysis.
Symbol
VFB_ADJ
VON
Parameter
Conditions
VCC = 4.5V, 0˚C to +125˚C
VCC = 5V, 0˚C to +125˚C
VCC = 5.5V, 0˚C to +125˚C
VCC = 4.5V, −40˚C to +125˚C
VCC = 5V, −40˚C to +125˚C
VCC = 5.5V, −40˚C to +125˚C
Rising
Min
Typ
0.6
0.6
0.6
0.6
0.6
0.6
4.2
3.6
Max
Units
0.591
0.591
0.591
0.589
0.589
0.589
0.609
0.609
0.609
0.609
0.609
0.609
FB Pin Voltage
V
UVLO Thresholds
V
Falling
SD = 5V, FB = 0.55V
Fsw = 600kHz
1
1.5
1.7
2
Operating VCC Current
mA
IQ-V5
SD = 5V, FB = 0.65V
Fsw = 600kHz
0.8
2.2
0.7
Shutdown VCC Current
SD = 0V
0.15
0.4
6
mA
µs
tPWGD1
tPWGD2
ISD
PWGD Pin Response Time
PWGD Pin Response Time
SD Pin Internal Pull-up Current
SS Pin Source Current
FB Voltage Going Up
FB Voltage Going Down
6
µs
20
µA
ISS-ON
SS Voltage = 2.5V
0˚C to +125˚C
8
5
11
11
15
15
µA
-40˚C to +125˚C
ISS-OC
SS Pin Sink Current During Over SS Voltage = 2.5V
Current
95
µA
µA
ISEN Pin Source Current Trip
Point
0˚C to +125˚C
35
28
50
50
65
65
ISEN-TH
-40˚C to +125˚C
ERROR AMPLIFIER
GBW
Error Amplifier Unity Gain
5
MHz
Bandwidth
G
Error Amplifier DC Gain
Error Amplifier Slew Rate
FB Pin Bias Current
60
6
dB
SR
IFB
V/µA
FB = 0.55V
0
0
15
30
2.8
0.8
1.2
3.2
100
155
nA
mA
V
FB = 0.65V
IEAO
VEA
EAO Pin Current Sourcing and
Sinking
VEAO = 2.5, FB = 0.55V
VEAO = 2.5, FB = 0.65V
Minimum
Error Amplifier Maximum Swing
Maximum
3
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Electrical Characteristics (Continued)
VCC = 5V unless otherwise indicated. Typicals and limits appearing in plain type apply for TA=TJ=+25˚C. Limits appearing in
boldface type apply over full Operating Temperature Range. Datasheet min/max specification limits are guaranteed by design,
test, or statistical analysis.
Symbol
GATE DRIVE
IQ-BOOT
Parameter
Conditions
Min
Typ
Max
Units
BOOT Pin Quiescent Current
BOOTV = 12V, EN = 0
0˚C to +125˚C
95
95
160
215
µA
-40˚C to +125˚C
RDS1
RDS2
Top FET Driver Pull-Up ON
resistance
@
BOOT-SW = 5V 350mA
3
2
3
2
Ω
Ω
Ω
Ω
Top FET Driver Pull-Down ON
resistance
@
BOOT-SW = 5V 350mA
RDS3
Bottom FET Driver Pull-Up ON
resistance
@
BOOT-SW = 5V 350mA
RDS4
Bottom FET Driver Pull-Down
ON resistance
@
BOOT-SW = 5V 350mA
OSCILLATOR
RFADJ = 590kΩ
50
300
600
600
1400
2000
90
RFADJ = 88.7kΩ
RFADJ = 42.2kΩ, 0˚C to +125˚C
RFADJ = 42.2kΩ, -40˚C to +125˚C
RFADJ = 17.4kΩ
500
490
700
700
fOSC
PWM Frequency
Max Duty Cycle
kHz
%
RFADJ = 11.3kΩ
D
fPWM = 300kHz
fPWM = 600kHz
88
LOGIC INPUTS AND OUTPUTS
VSD-IH
VSD-IL
SD Pin Logic High Trip Point
SD Pin Logic Low Trip Point
2.6
1.6
1.6
3.5
V
V
0˚C to +125˚C
1.3
1.25
-40˚C to +125˚C
FB Voltage Going Down
0˚C to +125˚C
VPWGD-TH-LO PWGD Pin Trip Points
VPWGD-TH-HI PWGD Pin Trip Points
0.413
0.410
0.430
0.430
0.446
0.446
V
-40˚C to +125˚C
FB Voltage Going Up
0˚C to +125˚C
0.691
0.688
0.710
0.710
35
0.734
0.734
V
-40˚C to +125˚C
VPWGD-HYS PWGD Hysteresis (LM2737 only) FB Voltage Going Down FB Voltage
Going Up
mV
110
Note 1: Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating ratings indicate conditions for which the device
operates correctly. Opearting Ratings do not imply guaranteed performance limits.
Note 2: The human body model is a 100pF capacitor discharged through a 1.5k resistor into each pin.
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Typical Performance Characteristics
Efficiency (VO = 1.5V)
Efficiency (VO = 3.3V)
FSW = 300kHz, TA = 25˚C
FSW = 300kHz, TA = 25˚C
20049412
20049413
VCC Operating Current vs Temperature
FSW = 600kHz, No-Load
Bootpin Current vs Temperature for BOOTV = 12V
FSW = 600kHz, Si4826DY FET, No-Load
20049415
20049414
Bootpin Current vs Temperature with 5V Bootstrap
FSW = 600kHz, Si4826DY FET, No-Load
PWM Frequency vs Temperature
for RFADJ = 43.2kΩ
20049416
20049417
5
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Typical Performance Characteristics (Continued)
RFADJ vs PWM Frequency
RFADJ vs PWM Frequency
(in 100 to 800kHz range), TA = 25˚C
(in 900 to 2000kHz range), TA = 25˚C
20049418
20049419
Switch Waveforms (HG Falling)
VIN = 5V, VO = 1.8V
IO = 3A, CSS = 10nF
FSW = 600kHz
VCC Operating Current Plus Boot Current vs
PWM Frequency (Si4826DY FET, TA = 25˚C)
20049423
20049420
Start-Up (No-Load)
VIN = 10V, VO = 1.2V
CSS = 10nF, FSW = 300kHz
Switch Waveforms (HG Rising)
VIN = 5V, VO = 1.8V
IO = 3A, FSW = 600kHz
20049424
20049421
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Typical Performance Characteristics (Continued)
Start-Up (Full-Load)
VIN = 10V, VO = 1.2V
IO = 10A, CSS = 10nF
FSW = 300kHz
Start Up (No-Load, 10x CSS
VIN = 10V, VO = 1.2V
CSS = 100nF, FSW = 300kHz
)
20049422
20049426
Start Up (Full Load, 10x CSS
VIN = 10V, VO = 1.2V
IO = 10A, CSS = 100nF
FSW = 300kHz
)
Shutdown
VIN = 10V, VO = 1.2V
IO = 10A, CSS = 10nF
FSW = 300kHz
20049425
20049427
Start Up (Full Load, 10x CSS
VIN = 10V, VO = 1.2V
IO = 10A, CSS = 100nF
FSW = 300kHz
)
Load Transient Response (IO = 0 to 4A)
VIN = 12V, VO = 1.2V
FSW = 300kHz
20049433
20049428
7
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Typical Performance Characteristics (Continued)
Load Transient Response (IO = 4 to 0A)
VIN = 12V, VO = 1.2V
Line Transient Response (VIN =5V to 12V)
VO = 1.2V, IO = 5A
FSW = 300kHz
FSW = 300kHz
20049429
20049430
Line Transient Response (VIN =12V to 5V)
VO = 1.2V, IO = 5A
Line Transient Response
VO = 1.2V, IO = 5A
FSW = 300kHz
FSW = 300kHz
20049431
20049432
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8
Block Diagram
20049401
Application Information
THEORY OF OPERATION
The LM2727 is a voltage-mode, high-speed synchronous
buck regulator with a PWM control scheme. It is designed for
use in set-top boxes, thin clients, DSL/Cable modems, and
other applications that require high efficiency buck convert-
ers. It has power good (PWRGD), output shutdown (SD),
over voltage protection (OVP) and under voltage protection
(UVP). The over-voltage and under-voltage signals are OR
gated to drive the Power Good signal and a shutdown latch,
which turns off the high side gate and turns on the low side
gate if pulled low. Current limit is achieved by sensing the
voltage VDS across the low side FET. During current limit the
high side gate is turned off and the low side gate turned on.
The soft start capacitor is discharged by a 95µA source
(reducing the maximum duty cycle) until the current is under
control. The LM2737 does not latch off during UVP or OVP,
and uses the HIGH and LOW comparators for the power-
good function only.
An application for a microprocessor might need a delay of
3ms, in which case CSS would be 12nF. For a different
device, a 100ms delay might be more appropriate, in which
case CSS would be 400nF. (390 10%) During soft start the
PWRGD flag is forced low and is released when the voltage
reaches a set value. At this point this chip enters normal
operation mode, the Power Good flag is released, and the
OVP and UVP functions begin to monitor Vo.
NORMAL OPERATION
While in normal operation mode, the LM2727/37 regulates
the output voltage by controlling the duty cycle of the high
side and low side FETs. The equation governing output
voltage is:
START UP
When VCC exceeds 4.2V and the enable pin EN sees a logic
high the soft start capacitor begins charging through an
internal fixed 10µA source. During this time the output of the
error amplifier is allowed to rise with the voltage of the soft
start capacitor. This capacitor, Css, determines soft start
time, and can be determined approximately by:
The PWM frequency is adjustable between 50kHz and
2MHz and is set by an external resistor, RFADJ, between the
FREQ pin and ground. The resistance needed for a desired
frequency is approximately:
9
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until VCC rises above 4.2V. As with shutdown, the soft start
capacitor is discharged through a FET, ensuring that the next
start-up will be smooth.
Application Information (Continued)
CURRENT LIMIT
Current limit is realized by sensing the voltage across the
low side FET while it is on. The RDSON of the FET is a known
value, hence the current through the FET can be determined
as:
MOSFET GATE DRIVERS
The LM2727/37 has two gate drivers designed for driving
N-channel MOSFETs in a synchronous mode. Power for the
drivers is supplied through the BOOTV pin. For the high side
gate (HG) to fully turn on the top FET, the BOOTV voltage
must be at least one VGS(th) greater than Vin. (BOOTV ≥
2*Vin) This voltage can be supplied by a separate, higher
voltage source, or supplied from a local charge pump struc-
ture. In a system such as a desktop computer, both 5V and
12V are usually available. Hence if Vin was 5V, the 12V
supply could be used for BOOTV. 12V is more than 2*Vin, so
the HG would operate correctly. For a BOOTV of 12V, the
initial gate charging current is 2A, and the initial gate dis-
charging current is typically 6A.
VDS = I * RDSON
The current limit is determined by an external resistor, RCS
,
connected between the switch node and the ISEN pin. A
constant current of 50µA is forced through Rcs, causing a
fixed voltage drop. This fixed voltage is compared against
VDS and if the latter is higher, the current limit of the chip has
been reached. RCS can be found by using the following:
RCS = RDSON(LOW) * ILIM/50µA
For example, a conservative 15A current limit in a 10A
design with a minimum RDSON of 10mΩ would require a
3.3kΩ resistor. Because current sensing is done across the
low side FET, no minimum high side on-time is necessary. In
the current limit mode the LM2727/37 will turn the high side
off and the keep low side on for as long as necessary. The
chip also discharges the soft start capacitor through a fixed
95µA source. In this way, smooth ramping up of the output
voltage as with a normal soft start is ensured. The output of
the LM2727/37 internal error amplifier is limited by the volt-
age on the soft start capacitor. Hence, discharging the soft
start capacitor reduces the maximum duty cycle D of the
controller. During severe current limit, this reduction in duty
cycle will reduce the output voltage, if the current limit con-
ditions lasts for an extended time.
20049402
During the first few nanoseconds after the low side gate
turns on, the low side FET body diode conducts. This causes
an additional 0.7V drop in VDS. The range of VDS is normally
much lower. For example, if RDSON were 10mΩ and the
current through the FET was 10A, VDS would be 0.1V. The
current limit would see 0.7V as a 70A current and enter
current limit immediately. Hence current limit is masked dur-
ing the time it takes for the high side switch to turn off and the
low side switch to turn on.
FIGURE 1. BOOTV Supplied by Charge Pump
In a system without a separate, higher voltage, a charge
pump (bootstrap) can be built using a diode and small ca-
pacitor, Figure 1. The capacitor serves to maintain enough
voltage between the top FET gate and source to control the
device even when the top FET is on and its source has risen
up to the input voltage level.
UVP/OVP
The LM2727/37 gate drives use a BiCMOS design. Unlike
some other bipolar control ICs, the gate drivers have rail-to-
rail swing, ensuring no spurious turn-on due to capacitive
coupling.
The output undervoltage protection and overvoltage protec-
tion mechanisms engage at 70% and 118% of the target
output voltage, respectively. In either case, the LM2727 will
turn off the high side switch and turn on the low side switch,
and discharge the soft start capacitor through a MOSFET
switch. The chip remains in this state until the shutdown pin
has been pulled to a logic low and then released. The UVP
function is masked only during the first charging of the soft
start capacitor, when voltage is first applied to the VCC pin. In
contrast, the LM2737 is designed to continue operating dur-
ing UVP or OVP conditions, and to resume normal operation
once the fault condition is cleared. As with the LM2727, the
powergood flag goes low during this time, giving a logic-level
warning signal.
POWER GOOD SIGNAL
The power good signal is the or-gated flag representing
over-voltage and under-voltage protection. If the output volt-
age is 18% over it’s nominal value, VFB = 0.7V, or falls 30%
below that value, VFB = 0.41V, the power good flag goes low.
The converter then turns off the high side gate, and turns on
the low side gate. Unlike the output (LM2727 only) the power
good flag is not latched off. It will return to a logic high
whenever the feedback pin voltage is between 70% and
118% of 0.6V.
SHUT DOWN
UVLO
If the shutdown pin SD is pulled low, the LM2727/37 dis-
charges the soft start capacitor through a MOSFET switch.
The high side switch is turned off and the low side switch is
turned on. The LM2727/37 remains in this state until SD is
released.
The 4.2V turn-on threshold on VCC has a built in hysteresis
of 0.6V. Therefore, if VCC drops below 3.6V, the chip enters
UVLO mode. UVLO consists of turning off the top FET,
turning on the bottom FET, and remaining in that condition
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10
In the case of a desktop computer system, the input current
slew rate is the system power supply or "silver box" output
current slew rate, which is typically about 0.1A/µs. Total input
capacitor ESR is 9mΩ, hence ∆V is 10*0.009 = 90 mV, and
the minimum inductance required is 0.9µH. The input induc-
tor should be rated to handle the DC input current, which is
approximated by:
Application Information (Continued)
DESIGN CONSIDERATIONS
The following is a design procedure for all the components
needed to create the circuit shown in Figure 3 in the Ex-
ample Circuits section, a 5V in to 1.2V out converter, capable
of delivering 10A with an efficiency of 85%. The switching
frequency is 300kHz. The same procedures can be followed
to create the circuit shown in Figure 3, Figure 4, and to
create many other designs with varying input voltages, out-
put voltages, and output currents.
In this case IIN-DC is about 2.8A. One possible choice is the
TDK SLF12575T-1R2N8R2, a 1.2µH device that can handle
8.2Arms, and has a DCR of 7mΩ.
INPUT CAPACITOR
The input capacitors in a Buck switching converter are sub-
jected to high stress due to the input current waveform,
which is a square wave. Hence input caps are selected for
their ripple current capability and their ability to withstand the
heat generated as that ripple current runs through their ESR.
Input rms ripple current is approximately:
OUTPUT INDUCTOR
The output inductor forms the first half of the power stage in
a Buck converter. It is responsible for smoothing the square
wave created by the switching action and for controlling the
output current ripple. (∆Io) The inductance is chosen by
selecting between tradeoffs in efficiency and response time.
The smaller the output inductor, the more quickly the con-
verter can respond to transients in the load current. As
shown in the efficiency calculations, however, a smaller in-
ductor requires a higher switching frequency to maintain the
same level of output current ripple. An increase in frequency
can mean increasing loss in the FETs due to the charging
and discharging of the gates. Generally the switching fre-
quency is chosen so that conduction loss outweighs switch-
ing loss. The equation for output inductor selection is:
The power dissipated by each input capacitor is:
Here, n is the number of capacitors, and indicates that power
loss in each cap decreases rapidly as the number of input
caps increase. The worst-case ripple for a Buck converter
occurs during full load, when the duty cycle D = 50%.
In the 5V to 1.2V case, D = 1.2/5 = 0.24. With a 10A
maximum load the ripple current is 4.3A. The Sanyo
10MV5600AX aluminum electrolytic capacitor has a ripple
current rating of 2.35A, up to 105˚C. Two such capacitors
make a conservative design that allows for unequal current
sharing between individual caps. Each capacitor has a maxi-
mum ESR of 18mΩ at 100 kHz. Power loss in each device is
then 0.05W, and total loss is 0.1W. Other possibilities for
input and output capacitors include MLCC, tantalum,
OSCON, SP, and POSCAPS.
Plugging in the values for output current ripple, input voltage,
output voltage, switching frequency, and assuming a 40%
peak-to-peak output current ripple yields an inductance of
1.5µH. The output inductor must be rated to handle the peak
current (also equal to the peak switch current), which is (Io +
0.5*∆Io). This is 12A for a 10A design. The Coilcraft D05022-
152HC is 1.5µH, is rated to 15Arms, and has a DCR of 4mΩ.
OUTPUT CAPACITOR
INPUT INDUCTOR
The output capacitor forms the second half of the power
stage of a Buck switching converter. It is used to control the
output voltage ripple (∆Vo) and to supply load current during
fast load transients.
The input inductor serves two basic purposes. First, in high
power applications, the input inductor helps insulate the
input power supply from switching noise. This is especially
important if other switching converters draw current from the
same supply. Noise at high frequency, such as that devel-
oped by the LM2727 at 1MHz operation, could pass through
the input stage of a slower converter, contaminating and
possibly interfering with its operation.
In this example the output current is 10A and the expected
type of capacitor is an aluminum electrolytic, as with the
input capacitors. (Other possibilities include ceramic, tanta-
lum, and solid electrolyte capacitors, however the ceramic
type often do not have the large capacitance needed to
supply current for load transients, and tantalums tend to be
more expensive than aluminum electrolytic.) Aluminum ca-
pacitors tend to have very high capacitance and fairly low
ESR, meaning that the ESR zero, which affects system
stability, will be much lower than the switching frequency.
The large capacitance means that at switching frequency,
the ESR is dominant, hence the type and number of output
capacitors is selected on the basis of ESR. One simple
formula to find the maximum ESR based on the desired
output voltage ripple, ∆Vo and the designed output current
ripple, ∆Io, is:
An input inductor also helps shield the LM2727 from high
frequency noise generated by other switching converters.
The second purpose of the input inductor is to limit the input
current slew rate. During a change from no-load to full-load,
the input inductor sees the highest voltage change across it,
equal to the full load current times the input capacitor ESR.
This value divided by the maximum allowable input current
slew rate gives the minimum input inductance:
11
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Rbypass and Cbypass are standard filter components de-
signed to ensure smooth DC voltage for the chip supply and
for the bootstrap structure, if it is used. Use 10Ω for the
resistor and a 2.2µF ceramic for the cap. Cb is the bootstrap
capacitor, and should be 0.1µF. (In the case of a separate,
higher supply to the BOOTV pin, this 0.1µF cap can be used
to bypass the supply.) Using a Schottky device for the boot-
strap diode allows the minimum drop for both high and low
side drivers. The On Semiconductor BAT54 or MBR0520
work well.
Application Information (Continued)
In this example, in order to maintain a 2% peak-to-peak
output voltage ripple and a 40% peak-to-peak inductor cur-
rent ripple, the required maximum ESR is 6mΩ. Three Sanyo
10MV5600AX capacitors in parallel will give an equivalent
ESR of 6mΩ. The total bulk capacitance of 16.8mF is
enough to supply even severe load transients. Using the
same capacitors for both input and output also keeps the bill
of materials simple.
Rp is a standard pull-up resistor for the open-drain power
good signal, and should be 10kΩ. If this feature is not
necessary, it can be omitted.
RCS is the resistor used to set the current limit. Since the
design calls for a peak current magnitude (Io + 0.5 * ∆Io) of
12A, a safe setting would be 15A. (This is well below the
saturation current of the output inductor, which is 25A.)
Following the equation from the Current Limit section, use a
3.3kΩ resistor.
RFADJ is used to set the switching frequency of the chip.
Following the equation in the Theory of Operation section,
the closest 1% tolerance resistor to obtain fSW = 300kHz is
88.7kΩ.
MOSFETS
MOSFETS are a critical part of any switching controller and
have a direct impact on the system efficiency. In this case
the target efficiency is 85% and this is the variable that will
determine which devices are acceptable. Loss from the ca-
pacitors, inductors, and the LM2727 itself are detailed in the
Efficiency section, and come to about 0.54W. To meet the
target efficiency, this leaves 1.45W for the FET conduction
loss, gate charging loss, and switching loss. Switching loss
is particularly difficult to estimate because it depends on
many factors. When the load current is more than about 1 or
2 amps, conduction losses outweigh the switching and gate
charging losses. This allows FET selection based on the
RDSON of the FET. Adding the FET switching and gate-
charging losses to the equation leaves 1.2W for conduction
losses. The equation for conduction loss is:
CSS depends on the users requirements. Based on the
equation for CSS in the Theory of Operation section, for a
3ms delay, a 12nF capacitor will suffice.
EFFICIENCY CALCULATIONS
A reasonable estimation of the efficiency of a switching
controller can be obtained by adding together the loss is
each current carrying element and using the equation:
PCnd = D(I2 * RDSON *k) + (1-D)(I2 * RDSON *k)
o
o
The factor k is a constant which is added to account for the
increasing RDSON of a FET due to heating. Here, k = 1.3. The
Si4442DY has a typical RDSON of 4.1mΩ. When plugged into
the equation for PCND the result is a loss of 0.533W. If this
design were for a 5V to 2.5V circuit, an equal number of
FETs on the high and low sides would be the best solution.
With the duty cycle D = 0.24, it becomes apparent that the
low side FET carries the load current 76% of the time.
Adding a second FET in parallel to the bottom FET could
improve the efficiency by lowering the effective RDSON. The
lower the duty cycle, the more effective a second or even
third FET can be. For a minimal increase in gate charging
loss (0.054W) the decrease in conduction loss is 0.15W.
What was an 85% design improves to 86% for the added
cost of one SO-8 MOSFET.
The following shows an efficiency calculation to complement
the Circuit of Figure 3. Output power for this circuit is 1.2V x
10A = 12W.
Chip Operating Loss
PIQ = IQ-V *VCC
CC
2mA x 5V = 0.01W
FET Gate Charging Loss
PGC = n * VCC * QGS * fOSC
The value n is the total number of FETs used. The Si4442DY
has a typical total gate charge, QGS, of 36nC and an rds-on of
4.1mΩ. For
a
single FET on top and bottom:
CONTROL LOOP COMPONENTS
2*5*36E-9*300,000 = 0.108W
The circuit is this design example and the others shown in
the Example Circuits section have been compensated to
improve their DC gain and bandwidth. The result of this
compensation is better line and load transient responses.
For the LM2727, the top feedback divider resistor, Rfb2, is
also a part of the compensation. For the 10A, 5V to 1.2V
design, the values are:
FET Switching Loss
PSW = 0.5 * Vin * IO * (tr + tf)* fOSC
The Si4442DY has a typical rise time tr and fall time tf of 11
and 47ns, respectively. 0.5*5*10*58E-9*300,000 = 0.435W
Cc1 = 4.7pF 10%, Cc2 = 1nF 10%, Rc = 229kΩ 1%. These
values give a phase margin of 63˚ and a bandwidth of
29.3kHz.
SUPPORT CAPACITORS AND RESISTORS
The Cinx capacitors are high frequency bypass devices,
designed to filter harmonics of the switching frequency and
input noise. Two 1µF ceramic capacitors with a sufficient
voltage rating (10V for the Circuit of Figure 3) will work well
in almost any case.
www.national.com
12
Input Inductor Loss
PLin = I2 * DCRinput-L
Application Information (Continued)
FET Conduction Loss
PCn = 0.533W
in
Input Capacitor Loss
2.822*0.007 = 0.055W
Output Inductor Loss
PLout = I2 * DCRoutput-L
102*0.004 = 0.4W
o
System Efficiency
4.282*0.018/2 = 0.084W
Example Circuits
20049403
FIGURE 2. 5V-16V to 3.3V, 10A, 300kHz
This circuit and the one featured on the front page have been
designed to deliver high current and high efficiency in a small
package, both in area and in height The tallest component in
this circuit is the inductor L1, which is 6mm tall. The com-
pensation has been designed to tolerate input voltages from
5 to 16V.
13
www.national.com
Example Circuits (Continued)
20049404
FIGURE 3. 5V to 1.2V, 10A, 300kHz
This circuit design, detailed in the Design Considerations
section, uses inexpensive aluminum capacitors and off-the-
shelf inductors. It can deliver 10A at better than 85% effi-
ciency. Large bulk capacitance on input and output ensure
stable operation.
20049405
FIGURE 4. 5V to 1.8V, 3A, 600kHz
The example circuit of Figure 4 has been designed for
minimum component count and overall solution size. A
switching frequency of 600kHz allows the use of small input/
output capacitors and a small inductor. The availability of
separate 5V and 12V supplies (such as those available from
desk-top computer supplies) and the low current further
reduce component count. Using the 12V supply to power the
MOSFET drivers eliminates the bootstrap diode, D1. At low
currents, smaller FETs or dual FETs are often the most
efficient solutions. Here, the Si4826DY, an asymmetric dual
FET in an SO-8 package, yields 92% efficiency at a load of
2A.
www.national.com
14
Example Circuits (Continued)
20049406
FIGURE 5. 3.3V to 0.8V, 5A, 500kHz
The circuit of Figure 5 demonstrates the LM2727 delivering a
low output voltage at high efficiency (87%) A separate 5V
supply is required to run the chip, however the input voltage
can be as low as 2.2
15
www.national.com
Example Circuits (Continued)
20049407
FIGURE 6. 1.8V and 3.3V, 1A, 1.4MHz, Simultaneous
The circuits in Figure 6 are intended for ADSL applications,
where the high switching frequency keeps noise out of the
data transmission range. In this design, the 1.8 and 3.3V
outputs come up simultaneously by using the same softstart
capacitor. Because two current sources now charge the
same capacitor, the capacitance must be doubled to achieve
the same softstart time. (Here, 40nF is used to achieve a
5ms softstart time.) A common softstart capacitor means
that, should one circuit enter current limit, the other circuit
will also enter current limit. In addition, if both circuits are
built with the LM2727, a UVP or OVP fault on one circuit will
cause both circuits to latch off. The additional compensation
components Rc2 and Cc3 are needed for the low ESR, all
ceramic output capacitors, and the wide (3x) range of Vin.
www.national.com
16
Example Circuits (Continued)
20049408
FIGURE 7. 12V Unregulated to 3.3V, 3A, 750kHz
This circuit shows the LM27x7 paired with a cost effective
solution to provide the 5V chip power supply, using no extra
components other than the LM78L05 regulator itself. The
input voltage comes from a ’brick’ power supply which does
not regulate the 12V line tightly. Additional, inexpensive 10uF
ceramic capacitors (Cinx and Cox) help isolate devices with
sensitive databands, such as DSL and cable modems, from
switching noise and harmonics.
20049409
FIGURE 8. 12V to 5V, 1.8A, 100kHz
In situations where low cost is very important, the LM27x7
can also be used as an asynchronous controller, as shown in
the above circuit. Although a a schottky diode in place of the
bottom FET will not be as efficient, it will cost much less than
the FET. The 5V at low current needed to run the LM27x7
could come from a zener diode or inexpensive regulator,
such as the one shown in Figure 7. Because the LM27x7
senses current in the low side MOSFET, the current limit
feature will not function in an asynchronous design. The
ISEN pin should be left open in this case.
17
www.national.com
TABLE 1. Bill of Materials for Typical Application Circuit
ID
Part Number
Type
Synchronous
Controller
N-MOSFET
Inductor
Size
Parameters
Qty.
Vendor
U1
LM2727
TSSOP-14
TSSOP-14
1
NSC
Q1, Q2
L1
Si4884DY
SO-8
7.1x7.1x3.2mm
0805
30V, 4.1mΩ, 36nC
1.5µH, 6.1A 9.6mΩ
10µF 6.3V
1
1
2
1
2
1
1
1
1
1
1
1
1
1
1
1
Vishay
TDK
RLF7030T-1R5N6R1
C2012X5R1J106M
C3216X7R1E105K
6MV2200WG
Cin1, Cin2
Cinx
MLCC
TDK
Capacitor
AL-E
1206
1µF, 25V
TDK
Co1, Co2
Cboot
Cin
10mm D 20mm H
1206
2200µF 6.3V125mΩ
0.1µF, 25V
Sanyo
Vishay
TDK
VJ1206X104XXA
C3216X7R1E225K
VJ1206X123KXX
VJ1206A2R2KXX
VJ1206A181KXX
CRCW1206100J
CRCW12066342F
CRCW12063923F
CRCW12061002F
CRCW12061002F
CRCW1206222J
Capacitor
Capacitor
Capacitor
Capacitor
Capacitor
Resistor
1206
0.1µF, 25V
Css
1206
12nF, 25V
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Cc1
1206
2.2pF 10%
180pF 10%
10Ω 5%
Cc2
1206
Rin
1206
Rfadj
Rc1
Resistor
1206
63.4kΩ 1%
392kΩ 1%
Resistor
1206
Rfb1
Rfb2
Rcs
Resistor
1206
10kΩ 1%
Resistor
1206
10kΩ 1%
Resistor
1206
2.2kΩ 5%
TABLE 2. Bill of Materials for Circuit of Figure 2
(Identical to BOM for 1.5V except as noted below)
ID
L1
Part Number
Type
Size
Parameters
Qty.
Vendor
RLF12560T-2R7N110 Inductor
12.5x12.8x6mm
2.7µH, 14.4A 4.5mΩ
1
TDK
Co1, Co2,
Co3, Co4
Cc1
10TPB100M
POSCAP
7.3x4.3x2.8mm
100µF 10V 1.9Arms
4
Sanyo
VJ1206A6R8KXX
VJ1206A271KXX
VJ1206A471KXX
CRCW12068451F
CRCW12061102F
Capacitor
Capacitor
Capacitor
Resistor
1206
1206
1206
1206
1206
6.8pF 10%
270pF 10%
470pF 10%
8.45kΩ 1%
11kΩ 1%
1
1
1
1
1
Vishay
Vishay
Vishay
Vishay
Vishay
Cc2
Cc3
Rc2
Rfb1
Resistor
TABLE 3. Bill of Materials for Circuit of Figure 3
ID
U1
Q1
Part Number
LM2727
Type
Synchronous
Controller
Size
TSSOP-14
SO-8
Parameters
Qty.
1
Vendor
NSC
@
Si4442DY
N-MOSFET
30V, 4.1mΩ, 4.5V,
1
Vishay
36nC
@
Q2
Si4442DY
BAT-54
N-MOSFET
SO-8
30V, 4.1mΩ, 4.5V,
1
Vishay
36nC
D1
Lin
L1
Schottky Diode
SOT-23
30V
1
1
1
Vishay
Coilcraft
Coilcraft
SLF12575T-1R2N8R2 Inductor
12.5x12.5x7.5mm
22.35x16.26x8mm
12µH, 8.2A, 6.9mΩ
1.5µH, 15A,4mΩ
D05022-152HC
Inductor
Aluminum
Electrolytic
Capacitor
Aluminum
Electrolytic
Capacitor
Capacitor
Capacitor
Cin1, Cin2
10MV5600AX
16mm D 25mm H
1206
5600µF10V 2.35Arms
1µF, 25V
2
1
2
Sanyo
TDK
Cinx
Co1, Co2,
Co3
C3216X7R1E105K
10MV5600AX
16mm D 25mm H
5600µF10V 2.35Arms
Sanyo
Cboot
Cin
VJ1206X104XXA
C3216X7R1E225K
VJ1206X123KXX
1206
1206
1206
0.1µF, 25V
2.2µF, 25V
12nF, 25V
1
1
1
Vishay
TDK
Css
Vishay
www.national.com
18
TABLE 3. Bill of Materials for Circuit of Figure 3 (Continued)
ID
Part Number
Type
Capacitor
Size
1206
1206
1206
1206
1206
1206
1206
1206
Parameters
4.7pF 10%
1nF 10%
Qty.
1
Vendor
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Cc1
Cc2
Rin
VJ1206A4R7KXX
VJ1206A102KXX
CRCW1206100J
CRCW12068872F
CRCW12062293F
CRCW12064991F
CRCW12064991F
CRCW1206152J
Capacitor
Resistor
Resistor
Resistor
Resistor
Resistor
Resistor
1
10Ω 5%
1
Rfadj
Rc1
Rfb1
Rfb2
Rcs
88.7kΩ 1%
229kΩ 1%
4.99kΩ 1%
4.99kΩ 1%
1.5kΩ 5%
1
1
1
1
1
TABLE 4. Bill of Materials for Circuit of Figure 4
ID
Part Number
Type
Synchronous
Controller
Size
Parameters
Qty.
Vendor
U1
LM2727
TSSOP-14
1
NSC
Q1/Q2
L1
Si4826DY
Asymetric Dual
N-MOSFET
Inductor
SO-8
30V, 24mΩ/ 8nC
Top 16.5mΩ/ 15nC
2.2µH, 6.1A, 12mΩ
1
1
Vishay
DO3316P-222
12.95x9.4x
5.21mm
7.3x4.3x3.1mm
7.3x4.3x3.1mm
1206
Coilcraft
Cin1
Co1
Cc
10TPB100ML
4TPB220ML
POSCAP
POSCAP
Capacitor
Capacitor
Capacitor
Capacitor
Capacitor
Resistor
Resistor
Resistor
Resistor
Resistor
Resistor
100µF 10V 1.9Arms
220µF 4V 1.9Arms
1µF, 25V
1
1
1
1
1
1
1
1
1
1
1
1
1
Sanyo
Sanyo
TDK
C3216X7R1E105K
C3216X7R1E225K
VJ1206X123KXX
VJ1206A100KXX
VJ1206A561KXX
CRCW1206100J
CRCW12064222F
CRCW12065112F
CRCW12062491F
CRCW12064991F
CRCW1206272J
Cin
1206
2.2µF, 25V
12nF, 25V
TDK
Css
Cc1
Cc2
Rin
1206
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
1206
10pF 10%
1206
560pF 10%
10Ω 5%
1206
Rfadj
Rc1
Rfb1
Rfb2
Rcs
1206
42.2kΩ 1%
51.1kΩ 1%
2.49kΩ 1%
4.99kΩ 1%
2.7kΩ 5%
1206
1206
1206
1206
TABLE 5. Bill of Materials for Circuit of Figure 5
ID
Part Number
Type
Synchronous
Controller
Size
Parameters
Qty.
Vendor
U1
LM2727
TSSOP-14
1
NSC
@
Q1
Q2
Si4884DY
Si4884DY
N-MOSFET
SO-8
SO-8
30V, 13.5mΩ, 4.5V
1
1
Vishay
Vishay
15.3nC
@
N-MOSFET
30V, 13.5mΩ, 4.5V
15.3nC
D1
Lin
BAT-54
Schottky Diode
Inductor
SOT-23
30V
1
1
1
1
Vishay
Pulse
Pulse
Sanyo
P1166.102T
P1168.102T
10MV5600AX
7.29x7.29 3.51mm
12x12x4.5 mm
16mm D 25mm H
1µH, 11A 3.7mΩ
1µH, 11A, 3.7mΩ
5600µF 10V 2.35Arms
L1
Inductor
Cin1
Aluminum
Electrolytic
Capacitor
Aluminum
Electrolytic
Capacitor
Capacitor
Capacitor
Cinx
Co1, Co2,
Co3
C3216X7R1E105K
16MV4700WX
1206
1µF, 25V
1
2
TDK
12.5mm D 30mm
4700µF 16V 2.8Arms
Sanyo
H
Cboot
Cin
VJ1206X104XXA
C3216X7R1E225K
VJ1206X123KXX
1206
1206
1206
0.1µF, 25V
2.2µF, 25V
12nF, 25V
1
1
1
Vishay
TDK
Css
Vishay
19
www.national.com
TABLE 5. Bill of Materials for Circuit of Figure 5 (Continued)
ID
Part Number
Type
Capacitor
Size
1206
1206
1206
1206
1206
1206
1206
1206
Parameters
4.7pF 10%
680pF 10%
10Ω 5%
Qty.
1
Vendor
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Cc1
Cc2
Rin
VJ1206A4R7KXX
VJ1206A681KXX
CRCW1206100J
CRCW12064992F
CRCW12061473F
CRCW12061492F
CRCW12064991F
CRCW1206332J
Capacitor
Resistor
Resistor
Resistor
Resistor
Resistor
Resistor
1
1
Rfadj
Rc1
Rfb1
Rfb2
Rcs
49.9kΩ 1%
147kΩ 1%
14.9kΩ 1%
4.99kΩ 1%
3.3kΩ 5%
1
1
1
1
1
TABLE 6. Bill of Materials for Circuit of Figure 6
ID
Part Number
Type
Synchronous
Controller
Assymetric Dual
N-MOSFET
Schottky Diode
Inductor
Size
Parameters
Qty.
Vendor
U1
LM2727
TSSOP-14
1
NSC
Q1/Q2
Si4826DY
SO-8
30V, 24mΩ/ 8nC
Top 16.5mΩ/ 15nC
30V
1
Vishay
D1
Lin
BAT-54
SOT-23
6.8x7.1x3.2mm
6.8x7.1x3.2mm
1812
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
Vishay
TDK
RLF7030T-1R0N64
RLF7030T-3R3M4R1
C4532X5R1E156M
C4532X5R1E156M
VJ1206X104XXA
C3216X7R1E225K
VJ1206X393KXX
VJ1206A220KXX
VJ1206A681KXX
VJ1206A681KXX
CRCW1206100J
CRCW12061742F
CRCW12061072F
CRCW120666R5F
CRCW12064991F
CRCW12061002F
CRCW1206152J
1µH, 6.4A, 7.3mΩ
3.3µH, 4.1A, 17.4mΩ
15µF 25V 3.3Arms
15µF 25V 3.3Arms
0.1µF, 25V
L1
Inductor
TDK
Cin1
Co1
Cboot
Cin
MLCC
Sanyo
Sanyo
TDK
MLCC
1812
Capacitor
Capacitor
Capacitor
Capacitor
Capacitor
Capacitor
Resistor
1206
1206
2.2µF, 25V
TDK
Css
Cc1
Cc2
Cc3
Rin
1206
39nF, 25V
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
1206
22pF 10%
1206
680pF 10%
1206
680pF 10%
1206
10Ω 5%
Rfadj
Rc1
Rc2
Rfb1
Rfb2
Rcs
Resistor
1206
17.4kΩ 1%
Resistor
1206
10.7kΩ 1%
Resistor
1206
66.5Ω 1%
Resistor
1206
4.99kΩ 1%
Resistor
1206
10kΩ 1%
Resistor
1206
1.5kΩ 5%
TABLE 7. Bill of Materials for 3.3V Circuit of Figure 6
(Identical to BOM for 1.8V except as noted below)
ID
Part Number
RLF7030T-4R7M3R4
VJ1206A270KXX
VJ1206X102KXX
VJ1206A821KXX
CRCW12061212F
CRCW12054R9F
CRCW12062211F
CRCW12061002F
Type
Inductor
Size
6.8x7.1x 3.2mm
1206
Parameters
4.7µH, 3.4A, 26mΩ
27pF 10%
Qty.
1
Vendor
TDK
L1
Cc1
Cc2
Cc3
Rc1
Rc2
Rfb1
Rfb2
Capacitor
Capacitor
Capacitor
Resistor
Resistor
Resistor
Resistor
1
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
1206
1nF 10%
1
1206
820pF 10%
12.1kΩ 1%
54.9Ω 1%
1
1206
1
1206
1
1206
2.21kΩ 1%
10kΩ 1%
1
1206
1
TABLE 8. Bill of Materials for Circuit of Figure 7
ID
Part Number
Type
Synchronous
Controller
Size
Parameters
Qty.
Vendor
U1
LM2727
TSSOP-14
1
NSC
www.national.com
20
TABLE 8. Bill of Materials for Circuit of Figure 7 (Continued)
ID
Part Number
Type
Voltage
Size
Parameters
Qty.
Vendor
U2
LM78L05
SO-8
1
NSC
Regulator
Q1/Q2
Si4826DY
Assymetric Dual
N-MOSFET
Schottky Diode
Inductor
SO-8
30V, 24mΩ/ 8nC
Top 16.5mΩ/ 15nC
30V
1
Vishay
D1
Lin
BAT-54
SOT-23
6.8x7.1x3.2mm
12.5x12.5x6.5mm
D: 10mm L:
12.5mm
1210
1
1
1
1
Vishay
TDK
RLF7030T-1R0N64
1µH, 6.4A, 7.3mΩ
4.2µH, 5.5A, 15mΩ
680µF 16V 3.4Arms
L1
SLF12565T-4R2N5R5 Inductor
TDK
Cin1
16MV680WG
Al-E
Sanyo
Cinx
Co1 Co2
Cox
C3216X5R1C106M
16MV680WG
MLCC
10µF 16V 3.4Arms
15µF 25V 3.3Arms
10µF 6.3V 2.7A
0.1µF, 25V
2.2µF, 25V
12nF, 25V
1
1
TDK
MLCC
1812
Sanyo
TDK
C3216X5R10J06M
VJ1206X104XXA
C3216X7R1E225K
VJ1206X123KXX
VJ1206A8R2KXX
VJ1206X102KXX
VJ1206X472KXX
CRCW12063252F
CRCW12065232F
CRCW120662371F
CRCW12062211F
CRCW12061002F
CRCW1206202J
MLCC
1206
Cboot
Cin
Capacitor
Capacitor
Capacitor
Capacitor
Capacitor
Capacitor
Resistor
Resistor
Resistor
Resistor
Resistor
Resistor
1206
1
1
1
1
1
1
1
1
1
1
1
1
Vishay
TDK
1206
Css
1206
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Cc1
1206
8.2pF 10%
1nF 10%
Cc2
1206
Cc3
1206
4.7nF 10%
32.5kΩ 1%
52.3kΩ 1%
2.37Ω 1%
Rfadj
Rc1
1206
1206
Rc2
1206
Rfb1
Rfb2
Rcs
1206
2.21kΩ 1%
10kΩ 1%
1206
1206
2kΩ 5%
TABLE 9. Bill of Materials for Circuit of Figure 8
ID
Part Number
Type
Synchronous
Controller
Size
Parameters
Qty.
Vendor
U1
LM2727
TSSOP-14
1
NSC
Q1
D2
Si4894DY
N-MOSFET
Schottky Diode
SO-8
SO-8
30V, 15mΩ, 11.5nC
30V, 3A
1
1
1
1
1
1
2
Vishay
ON
MBRS330T3
L1
SLF12565T-470M2R4 Inductor
12.5x12.8x 4.7mm
1812
47µH, 2.7A 53mΩ
20V 0.5A
TDK
D1
MBR0520
16MV680WG
Schottky Diode
ON
Cin1
Cinx
Co1, Co2
Al-E
1206
680µF, 16V, 1.54Arms
10µF, 16V, 3.4Arms
680µF 16V 26mΩ
Sanyo
TDK
C3216X5R1C106M
16MV680WG
MLCC
Al-E
1206
D: 10mm L:
12.5mm
1206
Sanyo
Cox
Cboot
Cin
C3216X5R10J06M
VJ1206X104XXA
C3216X7R1E225K
VJ1206X123KXX
VJ1206A561KXX
VJ1206X392KXX
VJ1206X223KXX
CRCW12062673F
CRCW12066192F
CRCW12067503F
CRCW12061371F
CRCW12061002F
CRCW1206122F
MLCC
10µF, 6.3V 2.7A
0.1µF, 25V
2.2µF, 25V
12nF, 25V
56pF 10%
3.9nF 10%
22nF 10%
267kΩ 1%
61.9kΩ 1%
750kΩ 1%
1.37kΩ 1%
10kΩ 1%
1
1
1
1
1
1
1
1
1
1
1
1
1
TDK
Capacitor
Capacitor
Capacitor
Capacitor
Capacitor
Capacitor
Resistor
Resistor
Resistor
Resistor
Resistor
Resistor
1206
Vishay
TDK
1206
Css
1206
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Vishay
Cc1
Cc2
Cc3
Rfadj
Rc1
Rc2
Rfb1
Rfb2
Rcs
1206
1206
1206
1206
1206
1206
1206
1206
1206
1.2kΩ 5%
21
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Physical Dimensions inches (millimeters) unless otherwise noted
TSSOP-14 Pin Package
NS Package Number MTC14
LIFE SUPPORT POLICY
NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT
DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL
COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:
1. Life support devices or systems are devices or
systems which, (a) are intended for surgical implant
into the body, or (b) support or sustain life, and
whose failure to perform when properly used in
accordance with instructions for use provided in the
labeling, can be reasonably expected to result in a
significant injury to the user.
2. A critical component is any component of a life
support device or system whose failure to perform
can be reasonably expected to cause the failure of
the life support device or system, or to affect its
safety or effectiveness.
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Support Center
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Fax: +49 (0) 180-530 85 86
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National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.
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