LM25085 [NSC]
42V Constant On-Time PFET Buck Switching Controller; 42V恒定导通时间PFET降压开关控制器
| 型号： | LM25085 |
| 厂家： | 
National Semiconductor |
| 描述： | 42V Constant On-Time PFET Buck Switching Controller |
| 文件： | 总24页 (文件大小：482K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
December 17, 2009
LM25085
42V Constant On-Time PFET Buck Switching Controller
General Description
Features
The LM25085 is a high efficiency PFET switching regulator
controller that can be used to quickly and easily develop a
small, efficient buck regulator for a wide range of applications.
This high voltage controller contains a PFET gate driver and
a high voltage bias regulator which operates over a wide 4.5V
to 42V input range. The constant on-time regulation principle
requires no loop compensation, simplifies circuit implemen-
tation, and results in ultra-fast load transient response. The
operating frequency remains nearly constant with line and
load variations due to the inverse relationship between the
input voltage and the on-time. The PFET architecture allows
100% duty cycle operation for a low dropout voltage. Either
the RDS(ON) of the PFET or an external sense resistor can be
used to sense current for over-current detection.
LM25085Q is an Automotive Grade product that is AEC-
■
Q100 grade 1 qualified (-40°C to 125°C operating junction
temperature)
Wide 4.5V to 42V input voltage range
■
■
Adjustable current limit using RDS(ON) or a current sense
resistor
Programmable switching frequency to 1MHz
■
■
■
■
No loop compensation required
Ultra-Fast transient response
Nearly constant operating frequency with line and load
variations
Adjustable output voltage from 1.25V
■
■
■
■
■
■
Precision ±2% feedback reference
Capable of 100% duty cycle operation
Internal soft-start timer
Integrated high voltage bias regulator
Thermal shutdown
Package
MSOP-8EP
■
■
■
MSOP-8
LLP-8 (3 mm x 3 mm)
Typical Application, Basic Step Down Controller
30079301
© 2009 National Semiconductor Corporation
300793
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Connection Diagrams
30079302
Top View
8-Lead MSOP-EP
30079304
Top View
8-Lead LLP
30079303
Top View
8-Lead MSOP
Ordering Information
Package
Type
NSC Package
Junction
Temperature Range
Order Number
Supplied As
Feature
Drawing
MUY08A
MUY08A
MUY08A
MUY08A
MUY08A
MUY08A
MUA08A
MUA08A
MUA08A
SDA08A
SDA08A
SDA08A
LM25085QMYE MSOP-8EP
LM25085QMY MSOP-8EP
LM25085QMYX MSOP-8EP
250 units on tape and reel
1000 units on tape and reel
3500 units on tape and reel
250 units on tape and reel
1000 units on tape and reel
3500 units on tape and reel
250 units on tape and reel
1000 units on tape and reel
3500 units on tape and reel
250 units on tape and reel
1000 units on tape and reel
4500 units on tape and reel
AEC-Q100 Grade 1
qualified. Automotive
Grade Production Flow*
-40°C to +125°C
LM25085MYE
LM25085MY
LM25085MYX
LM25085MME
LM25085MM
LM25085MMX
LM25085SDE
LM25085SD
MSOP-8EP
MSOP-8EP
MSOP-8EP
MSOP-8
MSOP-8
MSOP-8
LLP-8
-40°C to +125°C
LLP-8
LM25085SDX
LLP-8
*Automotive Grade (Q) product incorporates enhanced manufacturing and support processes for the automotive market, including defect detection methodologies.
Reliability qualification is compliant with the requirements and temperature grades defined in the AEC-Q100 standard. Automotive grade products are identified
with the letter Q. For more information go to http://www.national.com/automotive.
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2
Pin Descriptions
Pin
Name Description
No.
Application Information
1
2
3
ADJ
Current Limit Adjust
The current limit threshold is set by an external resistor from VIN to ADJ in
conjunction with the external sense resistor or the PFET’s RDS(ON)
.
RT
On-time control and shutdown
An external resistor from VIN to RT sets the buck switch on-time and switching
frequency. Grounding this pin shuts down the controller.
FB
Voltage Feedback from the
regulated output
Input to the regulation and over-voltage comparators. The regulation level is
1.25V.
4
5
GND Circuit Ground
Ground reference for all internal circuitry
ISEN Current sense input for current
limit detection.
Connect to the PFET drain when using RDS(ON) current sense. Connect to the
PFET source and the sense resistor when using a current sense resistor.
Connect to the gate of the external PFET.
6
7
PGATE Gate Driver Output
VCC Output of the gate driver bias
regulator
Output of the negative voltage regulator (relative to VIN) that biases the PFET
gate driver. A low ESR capacitor is required from VIN to VCC, located as close
as possible to the pins.
8
VIN
EP
Input supply voltage
Exposed Pad
The operating input range is from 4.5V to 42V. A low ESR bypass capacitor must
be located as close as possible to the VIN and GND pins.
Exposed pad on the underside of the package (MSOP-8EP and LLP only). This
pad is to be soldered to the PC board ground plane to aid in heat dissipation.
3
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VIN to VCC, VIN to PGATE
ESD Rating (Note 2)
Human Body Model
-0.3V to 10V
Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
2kV
Storage Temperature Range
-65°C to +150°C
VIN to GND
-0.3V to 45V
Operating Ratings (Note 1)
ISEN to GND
ADJ to GND
RT, FB to GND
-0.3V to VIN + 0.3V
-0.3V to VIN + 0.3V
-0.3V to 7V
VIN Voltage
4.5V to 42V
−40°C to + 125°C
Junction Temperature
Electrical Characteristics Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the
junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are guaranteed through test, design, or statistical
correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only.
Unless otherwise stated the following conditions apply: VIN = 24V, RT = 100 kΩ.
Symbol
VIN Pin
Parameter
Conditions
Min
Typ
Max
Units
IIN
IQ
Operating current
Shutdown current
Non-switching, FB = 1.4V (Note 6)
RT = 0V (Note 6)
1.25
175
1.75
300
mA
µA
VCC Regulator (Note 3)
VCC(reg)
VIN - VCC
Vin = 9V, FB = 1.4V, ICC = 0 mA
Vin = 9V, FB = 1.4V, ICC = 20 mA
Vin = 42V, FB = 1.4V, ICC = 0 mA
VCC increasing
6.9
7.7
7.7
7.7
3.8
8.5
V
V
V
V
UVLOVcc
VCC under-voltage lock-out
threshold
UVLOVcc hysteresis
VCC Current Limit
VCC decreasing
FB = 1.4V
260
40
mV
mA
VCC(CL)
PGATE Pin
VPGATE(HI)
20
PGATE High voltage
PGATE Low voltage
PGATE Pin = Open
PGATE Pin = Open
VIN -0.1
VIN -0.1
VIN
VCC
VIN
V
V
V
V
A
A
Ω
Ω
VPGATE(LO)
VPGATE(HI)4.5
VPGATE(LO)4.5
IPGATE
VCC+0.1
VCC+0.1
PGATE High Voltage at Vin = 4.5V PGATE Pin = Open
PGATE Low Voltage at Vin = 4.5V PGATE Pin = Open
VCC
1.75
1.5
Driver Output Source Current
Driver Output Sink Current
Driver Output Resistance
VIN = 12V, PGATE = VIN - 3.5V
VIN = 12V, PGATE = VIN - 3.5V
Source current = 500 mA
Sink current = 500 mA
RPGATE
2.3
2.3
Current Limit Detection
IADJ
ADJUST pin current source
VADJ = 22.5V
32
-9
40
0
48
9
µA
VCL OFFSET
Current limit comparator offset
VADJ = 22.5V, VADJ - VISEN
mV
RT Pin
RTSD
Shutdown threshold
RT Pin voltage rising
0.73
50
V
RTHYS
Shutdown threshold hysteresis
mV
On-Time
tON – 1
tON – 2
tON - 3
tON - 4
On-time
3.5
560
329
55
5
7.15
870
500
235
µs
ns
ns
ns
VIN = 4.5V, RT = 100 kΩ
720
415
140
VIN = 24V, RT = 100 kΩ
VIN = 42V, RT = 100 kΩ
Minimum on-time in current limit
VIN = 24V, 25 mV overdrive at ISEN
(Note 7)
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Symbol
Off-Time
Parameter
Conditions
Min
Typ
Max
Units
tOFF(CL1)
tOFF(CL2)
tOFF(CL3)
tOFF(CL4)
Off-time (current limit) (Note 7)
VIN = 12V, VFB = 0V
VIN = 12V, VFB = 1V
VIN = 24V, VFB = 0V
VIN = 24V, VFB = 1V
5.35
1.42
8.9
7.9
1.9
13
10.84
3.03
17.7
4.68
µs
µs
µs
µs
2.22
3.2
Regulation and Over-Voltage Comparators (FB Pin)
VREF
FB regulation threshold
FB over-voltage threshold
FB bias current
1.225
1.4
1.25
350
10
1.275
V
VOV
Measured with respect to VREF
Junction temperature rising
mV
nA
IFB
Soft-Start Function
tSS
Soft-start time
2.5
4.3
ms
Thermal Shutdown
TSD
Junction shutdown temperature
Junction shutdown hysteresis
170
20
°C
°C
THYS
Thermal Resistance
Junction to ambient, 0 LFPM air
flow (Note 5)
MSOP-8 package
MSOP-8EP package
LLP-8 package
126
46
°C/W
°C/W
θJA
54
Junction to case, 0 LFPM air flow MSOP-8 package
29
θJC
(Note 5)
MSOP-8EP package
LLP-8 package
5.5
9.1
Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the
device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics.
Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.
Note 3: VCC provides self bias for the internal gate drive.
Note 4: For detailed information on soldering plastic MSOP and LLP packages refer to the Packaging Data Book available from National Semiconductor
Corporation.
Note 5: Tested on a 4 layer JEDEC board. Four vias provided under the exposed pad. See JEDEC standards JESD51-5 and JESD51-7.
Note 6: Operating current and shutdown current do not include the current in the RT resistor.
Note 7: The tolerance of the minimum on-time (tON-4) and the current limit off-times (tOFF(CL1) through (tOFF(CL4)) track each other over process and temperature
variations. A device which has an on-time at the high end of the range will have an off-time that is at the high end of its range.
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Typical Performance Characteristics
Unless otherwise specified the following conditions apply: TJ = 25°C, VIN = 24V.
Efficiency (Circuit of Figure 4)
Input Operating Current vs. VIN
30079305
30079306
30079308
30079311
Shutdown Current vs. VIN
VCC vs. VIN
30079307
VCC vs. ICC
On-Time vs. RT and VIN
30079309
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Off-Time vs. VIN and VFB
Voltage at the RT Pin
30079312
30079313
ADJ Pin Current vs. VIN
Input Operating Current vs. Temperature
30079318
30079317
Shutdown Current vs. Temperature
VCC vs. Temperature
30079353
30079352
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On-Time vs. Temperature
Minimum On-Time vs. Temperature
30079354
30079355
Off-Time vs. Temperature
Current Limit Comparator Offset vs. Temperature
30079357
30079356
ADJ Pin Current vs. Temperature
PGATE Driver Output Resistance vs. Temperature
30079314
30079310
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Feedback Reference Voltage vs. Temperature
Soft-Start Time vs. Temperature
30079315
30079316
RT Pin Shutdown Threshold vs. Temperature
30079358
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Block Diagram
30079319
Sense resistor method shown for current limit detection.
Minimum output ripple configuration shown.
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sistors. The output voltage ripple is a result of the inductor’s
ripple current passing through the output capacitor’s ESR, or
through a resistor in series with the output capacitor. Multiple
methods are available to ensure sufficient ripple is supplied
to the FB pin, and three different configurations are discussed
in the Applications Information section.
Functional Description
OVERVIEW
The LM25085 is a PFET buck (step-down) DC-DC controller
using the constant on-time (COT) control principle. The input
operating voltage range of the LM25085 is 4.5V to 42V. The
use of a PFET in a buck regulator greatly simplifies the gate
drive requirements and allows for 100% duty cycle operation
to extend the regulation range when operating at low input
voltage. However, PFET transistors typically have higher on-
resistance and gate charge when compared to similarly rated
NFET transistors. Consideration of available PFETs, input
voltage range, gate drive capability of the LM25085, and ther-
mal resistances indicate an upper limit of 10A for the load
current for LM25085 applications. Constant on-time control is
implemented using an on-time one-shot that is triggered by
the feedback signal. During the off-time, when the PFET (Q1)
is off, the load current is supplied by the inductor and the out-
put capacitor. As the output voltage falls, the voltage at the
feedback comparator input (FB) falls below the regulation
threshold. When this occurs Q1 is turned on for the one-shot
period which is determined by the input voltage (VIN) and the
RT resistor. During the on-time the increasing inductor current
increases the voltage at FB above the feedback comparator
threshold. For a buck regulator the basic relationship between
the on-time, off-time, input voltage and output voltage is:
When in regulation, the LM25085 operates in continuous con-
duction mode at medium to heavy load currents and discon-
tinuous conduction mode at light load currents. In continuous
conduction mode the inductor’s current is always greater than
zero, and the operating frequency remains relatively constant
with load and line variations. The minimum load current for
continuous conduction mode is one-half the inductor’s ripple
current amplitude. In discontinuous conduction mode, where
the inductor’s current reaches zero during the off-time, the
operating frequency is lower than in continuous conduction
mode and varies with load current. Conversion efficiency is
maintained at light loads since the switching losses are re-
duced with the reduction in load and frequency.
If the voltage at the FB pin exceeds 1.6V due to a transient
overshoot or excessive ripple at VOUT the internal over-volt-
age comparator immediately switches off Q1. The next on-
time period starts when the voltage at FB falls below the
feedback reference voltage.
ON-TIME TIMER
The on-time of the PFET gate drive output (PGATE pin) is
determined by the resistor (RT) and the input voltage (VIN),
and is calculated from:
(1)
where Fs is the switching frequency. Equation 1 is valid only
in continuous conduction mode (inductor current does not
reach zero). Since the LM25085 controls the on-time inverse-
ly proportional to VIN, the switching frequency remains rela-
tively constant as VIN is varied. If the input voltage falls to a
level that is equal to or less than the regulated output voltage
Q1 is held on continuously (100% duty cycle) and VOUT is ap-
proximately equal to VIN.
(3)
where RT is in kΩ.
The minimum on-time, which occurs at maximum VIN, should
not be set less than 150 ns (see Current Limiting section). The
buck regulator effective on-time, measured at the SW node
(junction of Q1, L1, and D1) is typically longer than that cal-
culated in Equation 3 due to the asymmetric delay of the
PFET. The on-time difference caused by the PFET switching
delay can be estimated as the difference of the turn-off and
turn-on delays listed in the PFET data sheet. Measuring the
difference between the on-time at the PGATE pin versus the
SW node in the actual application circuit is also recommend-
ed.
The COT control scheme, with the feedback signal applied to
a comparator rather than an error amplifier, requires no loop
compensation, resulting in very fast load transient response.
The LM25085 is available in both an 8 pin MSOP package
and an 8 pin LLP package with an exposed pad to aid in heat
dissipation. An 8 pin MSOP package without an exposed pad
is also available.
In continuous conduction mode, the inverse relationship of
tON with VIN results in a nearly constant switching frequency
as VIN is varied. The operating frequency can be calculated
from:
REGULATION CONTROL CIRCUIT
The LM25085 buck DC-DC controller employs a control
scheme based on a comparator and a one-shot on-timer, with
the output voltage feedback compared to an internal refer-
ence voltage (1.25V). When the FB pin voltage falls below the
feedback reference, Q1 is switched on for a time period de-
termined by the input voltage and a programming resistor
(RT). Following the on-time Q1 remains off until the FB voltage
falls below the reference. Q1 is then switched on for another
on-time period. The output voltage is set by the feedback re-
sistors (RFB1, RFB2 in the Block Diagram). The regulated
output voltage is calculated as follows:
(4)
where RT is in kΩ, and tD is equal to 50 ns plus the PFET’s
delay difference. To set a specific continuous conduction
mode switching frequency (FS), the RT resistor is determined
from the following:
VOUT = 1.25V x (RFB2+ RFB1)/ RFB1
(2)
(5)
The feedback voltage supplied to the FB pin is applied to a
comparator rather than a linear amplifier. For proper opera-
tion sufficient ripple amplitude is necessary at the FB pin to
switch the comparator at regular intervals with minimum delay
and noise susceptibility. This ripple is normally obtained from
the output voltage ripple attenuated through the feedback re-
where RT is in kΩ. A simplified version of Equation 5 at VIN
12V, and tD = 100 ns, is:
=
11
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For VIN = 42V and tD = 100 ns, the simplified equation is:
When using equation 6 or 7, the tolerances for the ADJ pin
current sink and the offset of the current limit comparator
should be included to ensure the resulting minimum current
limit is not less than the required maximum switch current.
Simultaneously increasing the values of RADJ and RSEN de-
creases the effects of the current limit comparator offset, but
at the expense of higher power dissipation. When using a
sense resistor, the RSEN resistor value should be chosen with-
in the practical limitations of power dissipation and physical
SHUTDOWN
The LM25085 can be shutdown by grounding the RT pin (see
Figure 1). In this mode the PFET is held off, and the VCC
regulator is disabled. The internal operating current is re-
duced to the value shown in the graph “Shutdown current vs.
VIN”. The shutdown threshold at the RT pin is ≊0.73V, with
≊50 mV of hysteresis. Releasing the pin enables normal op-
eration. The RT pin must not be forced high during normal
operation.
size. For example, for a 10A current limit, setting RSEN
=
0.005Ω results in a power dissipation as high as 0.5W. Cur-
rent sense connections to the RSEN resistor, or to Q1, must
be Kelvin connections to ensure accuracy.
The CADJ capacitor filters noise from the ADJ pin, and helps
prevent unintended switching of the current limit comparator
due to input voltage transients. The recommended value for
CADJ is 1000 pF.
CURRENT LIMIT OFF-TIME
When the current through Q1 exceeds the current limit thresh-
old, the LM25085 forces an off-time longer than the normal
off-time defined by Equation 1. See the graph “Off-Time vs.
VIN and VFB”, or calculate the current limit off-time from the
following equation:
30079324
(8)
where VIN is the input voltage, and VFB is the voltage at the
FB pin at the time current limit was detected. This feature is
necessary to allow the inductor current to decrease sufficient-
ly to offset the current increase which occurred during the on-
time. During the on-time, the inductor current increases an
amount equal to:
FIGURE 1. Shutdown Implementation
CURRENT LIMITING
The LM25085 current limiting operates by sensing the voltage
across either the RDS(ON) of Q1, or a sense resistor, during the
on-time and comparing it to the voltage across the resistor
RADJ (see Figure 2). The current limit function is much more
accurate and stable over temperature when a sense resistor
is used. The RDS(ON) of a MOSFET has a wide process vari-
ation and a large temperature coefficient.
(9)
During the off-time the inductor current decreases due to the
reverse voltage applied across the inductor by the output volt-
age, the freewheeling diode’s forward voltage (VFD), and the
voltage drop due to the inductor’s series resistance (VESR).
The current decrease is equal to:
If the voltage across RDS(ON) of Q1, or the sense resistor, is
greater than the voltage across RADJ, the current limit com-
parator switches to turn off Q1. Current sensing is disabled
for a blanking time of ≊100 ns at the beginning of the on-time
to prevent false triggering of the current limit comparator due
to leading edge current spikes. Because of the blanking time
and the turn-on and turn-off delays created by the PFET, the
on-time at the PGATE pin should not be set less than 150 ns.
An on-time shorter than that may prevent the current limit de-
tection circuit from properly detecting an over-current condi-
tion. The duration of the subsequent forced off-time is a
function of the input voltage and the voltage at the FB pin, as
shown in the graph “Off-time vs. VIN and VFB”. The longer-
than-normal forced off-time allows the inductor current to
decrease to a low level before the next on-time. This cycle-
by-cycle monitoring, followed by a forced off-time, provides
effective protection from output load faults over a wide range
of operating conditions.
(10)
The on-time in Equation 9 is shorter than the normal on-time
since the PFET is shut off when the current limit threshold is
crossed. If the off-time is not long enough, such that the cur-
rent decrease (Equation 10) is less than the current increase
(Equation 9), the current levels are higher at the start of the
next on-time. This results in a further decrease in on-time,
since the current limit threshold is crossed sooner. A balance
is reached when the current changes in Equations 9 and 10
are equal. The worst case situation is that of a direct short
circuit at the output terminals, where VOUT = 0 volts, as that
results in the largest current increase during the on-time, and
the smallest decrease during the off-time. The sum of the
diode’s forward voltage and the inductor’s ESR voltage must
be sufficient to ensure current runaway does not occur. Using
Equations 9 and 10, this requirement can be stated as:
The voltage across the RADJ resistor is set by an internal 40
µA current sink at the ADJ pin. When using Q1’s RDS(ON) for
sensing, the current at which the current limit comparator
switches is calculated from:
ICL = 40 µA x RADJ/RDS(ON)
(6)
When using a sense resistor (RSEN) the thrshold of the current
limit comparator is calculated from:
(11)
ICL = 40 µA x RADJ/RSEN
(7)
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For tON in Equation 11 use the minimum on-time at the SW
node. To determine this time period add the “Minimum on-
time in current limit” specified in the Electrical Characteristics
(tON-4) to the difference of the turn-off and turn-on delays of
the PFET. For tOFF use the value in the graph “Off-Time vs.
VIN and VFB”, or use Equation 8, where VFB is equal to zero
volts. When using the minimum or maximum limits of those
specifications to determine worst case situations, the toler-
ance of the minimum on-time (tON-4) and the current limit off-
times (tOFF(CL1) through tOFF(CL4)) track each other over the
process and temperature variations. A device which has an
on-time at the high end of the range will have an off-time that
is at the high end of its range.
30079325
FIGURE 2. Current Limit Sensing
VCC REGULATOR
designing a PFET buck regulator the drain terminal is con-
nected to the switching node. This situation requires a trade-
off between thermal and EMI performance since increasing
the PC board area of the switching node to aid the PFET
power dissipation also increases radiated noise, possibly dis-
rupting the circuit operation. Typically the switching node area
is kept to a reasonable minimum and the PFET peak current
is derated to stay within the recommended temperature rating
of the PFET. The RDS(ON) of the PFET determines a portion
of the power dissipation in the PFET. However, PFETs with
very low RDS(ON) usually have large values of gate charge. A
PFET with a higher gate charge has a corresponding slower
switching speed, leading to higher switching losses and af-
fecting the PFET power dissipation.
The VCC regulator provides a regulated voltage between the
VIN and the VCC pins to provide the bias and gate current for
the PFET gate driver. The 0.47 µF capacitor at the VCC pin
must be a low ESR capacitor, preferably ceramic as it pro-
vides the high surge current for the PFET’s gate at each turn-
on. The capacitor must be located as close as possible to the
VIN and VCC pins to minimize inductance in the PC board
traces.
Referring to the graph “VCC vs. VIN”, the voltage across the
VCC regulator (VIN – VCC) is equal to VIN until VIN reaches
approximately 8.5V. At higher values of VIN, the voltage at
the VCC pin is regulated at approximately 7.7V below VIN.
The VCC regulator has a maximum current capability of at
least 20 mA. The regulator is disabled when the LM25085 is
shutdown using the RT pin, or when the thermal shutdown is
activated.
If the PFET RDS(ON) is used for current limit detection, note
that it typically has a positive temperature coefficient. At 100°
C the RDS(ON) may be as much as 50% higher than the value
at 25°C which could result in incorrect current limiting if not
accounted for when determining the value of the RADJ resistor.
The PFET Total Gate Charge determines most of the power
dissipation in the LM25085 due to the repetitive charge and
discharge of the PFET’s gate capacitance by the gate driver
(powered from the VCC regulator). The LM25085’s internal
power dissipation can be calculated from the following:
PGATE DRIVER OUTPUT
The PGATE pin output swings between VIN (Q1 off) and the
VCC pin voltage (Q1 on). The rise and fall times depend on
the PFET gate capacitance and the source and sink currents
provided by the internal gate driver. See the Electrical Chara-
teristics for the current capability of the driver.
PDISS = VIN x ((QG x FS) + IIN)
(12)
P-CHANNEL MOSFET SELECTION
where QG is the PFET's Total Gate Charge obtained from its
datasheet, FS is the switching frequency, and IIN is the
LM25085's operating current obtained from the graph "Input
Operating Current vs. VIN". Using the Thermal Resistance
specifications in the Electrical Characteristics table, the ap-
proximate junction temperature can be determined. If the
calculated junction temperature is near the maximum oper-
ating temperature of 125°C, either the switching frequency
must be reduced, or a PFET with a smaller Total Gate Charge
must be used.
The PFET must be rated for the maximum input voltage, with
some margin above that to allow for transients and ringing
which can occur on the supply line and the switching node.
The gate-to-source voltage (VGS) normally provided to the
PFET is 7.7 volts for VIN greater than 8.5V. However, if the
circuit is to be operated at lower values of VIN, the selected
PFET must be able to fully turn-on with a VGS voltage equal
to VIN. The minimum input operating voltage for the LM25085
is 4.5V.
Similar to NFETs, the case or exposed thermal pad for a
PFET is electrically connected to the drain terminal. When
13
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SOFT-START
The internal soft-start feature of the LM25085 allows the reg-
ulator to gradually reach a steady state operating point at
power up, thereby reducing startup stresses and current
surges. Upon turn-on, when Vcc reaches its under-voltage
lockout threshold, the internal soft-start circuit ramps the feed-
back reference voltage from 0V to 1.25V, causing VOUT to
ramp up in a proportional manner. The soft-start ramp time is
typically 2.5 ms.
A standard value 90.9 kΩ resistor is selected. Using equation
3 the minimum on-time at the PGATE pin, which occurs at
maximum input voltage (42V), is calculated to be 381 ns. This
minimum one-shot period is sufficiently longer than the mini-
mum recommended value of 150 ns. The minimum on-time
at the SW node is longer due to the delay added by the PFET
(57 ns). Therefore the minimum SW node on-time is 438 ns
at 42V. At the SW node the maximum on-time is calculated
to be 2.55 µs at 7V.
In addition to controlling the initial power up cycle, the soft-
start circuit also activates when the LM25085 is enabled by
releasing the RT pin, and when the circuit is shutdown and
restarted by the internal Thermal Shutdown circuit.
L1: The main parameter controlled by the inductor value is
the current ripple amplitude (IOR). See Figure 3. The minimum
load current for continuous conduction mode is used to de-
termine the maximum allowable ripple such that the inductor
current’s lower peak does not fall below 0 mA. Continuous
conduction mode operation at minimum load current is not a
requirement of the LM25085, but serves as a guideline for
selecting L1. For this example, the maximum ripple current is:
If the voltage at FB is below the regulation threshold value
due to an over-current condition or a short circuit at Vout, the
internal reference voltage provided by the soft-start circuit to
the regulation comparator is reduced along with FB. When the
over-current or short circuit condition is removed, VOUT re-
turns to the regulated value at a rate determined by the soft-
start ramp. This feature helps prevent the output voltage from
over-shooting following an overload event.
IOR(max) = 2 x IOUT(min) = 1.2 Amp
(13)
THERMAL SHUTDOWN
If an application’s minimum load current is zero, a good initial
estimate for the maximum ripple current (IOR(max)) is 20% of
the maximum load current. The ripple calculated in equation
13 is then used in the following equation to calculate L1:
The LM25085 should be operated such that the junction tem-
perature does not exceed 125°C. If the junction temperature
increases above that, an internal Thermal Shutdown circuit
activates at 170°C (typical) to disable the VCC regulator and
the gate driver, and discharge the soft-start capacitor. This
feature helps prevent catastrophic failures from accidental
device overheating. When the junction temperature falls be-
low 150°C (typical hysteresis = 20°C), the gate driver is
enabled, the soft-start circuit is released, and normal opera-
tion resumes.
(14)
A standard value 15 µH inductor is selected. Using this in-
ductance value, the maximum ripple current amplitude, which
occurs at maximum input voltage, calculates to 1.08 Ap-p.
The peak current (IPK) at maximum load current is 5.54A.
However, the current rating of the selected inductor must be
based on the maximum current limit value calculated below.
Applications Information
EXTERNAL COMPONENTS
The procedure for calculating the external components is il-
lustrated with the following design example. Referring to the
Block Diagram, the circuit is to be configured for the following
specifications:
VOUT = 5V
VIN = 7V to 42V, 12V Nominal
Maximum load current (IOUT(max)) = 5A
Minimum load current (IOUT(min)) = 600 mA (for continuous
conduction mode)
Switching Frequency (FSW) = 300 kHz
Maximum allowable output ripple (VOS) = 5 mVp-p
Selected PFET: Vishay Si7465
30079332
FIGURE 3. Inductor Current Waveform
RFB1 and RFB2: These resistors set the output voltage. The
ratio of these resistors is calculated from:
RSEN, RADJ: To achieve good current limit accuracy and avoid
over designing the power stage components, the sense re-
sistor method is used for current limiting in this example. A
standard value 10 mΩ resistor is selected for RSEN, resulting
in a 50 mV drop at maximum load current, and a maximum
0.25W power dissipation in the resistor. Since the LM25085
uses peak current detection, the minimum value for the cur-
rent limit threshold must be equal to the maximum load cur-
rent (5A) plus half the maximum ripple amplitude calculated
above:
RFB2/RFB1 = (VOUT/1.25V) - 1
For this example, RFB2 / RFB1 = 3. Typically, RFB1 and RFB2
should be chosen from standard value resistors in the range
of 1 kΩ to 20 kΩ which satisfy the above ratio. For this ex-
ample, RFB2 = 10 kΩ, and RFB1 = 3.4 kΩ.
RT, PFET: Before selecting the RT resistor, the PFET must be
selected as its turn-on and turn-off delays affect the calculated
value of RT. For the Vishay Si7465 PFET, the difference of its
typical turn-off and turn-on delays is 57 ns. Using equation 5
at nominal input voltage, RT calculates to be:
ICL(min) = 5A + 1.08A/2 = 5.54A
At this current level the voltage across RSEN is 55.4 mV.
Adding the current limit comparator offset of 9 mV (max) in-
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14
creases the required current limit threshold to 6.44A. Using
equation 7 with the minimum value for the ADJ pin current (32
µA), the required RADJ resistor calculates to:
where VSW is the absolute value of the voltage at the SW node
during the off-time, typically 0.5V to 1V depending on the
diode D1. Using a typical value of 0.65V, VA calculates to
4.81V. VA is the nominal DC voltage at the R3/C1 junction,
and is used in the next equation:
A standard value 2.1 kΩ resistor is selected. The nominal
current limit threshold calculates to:
where tON is the maximum on-time (at minimum input volt-
age), and ΔV is the desired ripple amplitude at the R3/C1
junction, typically 25 mVp-p. For this example
Using the tolerances for the ADJ pin current and the current
limit comparator offset, the maximum current limit threshold
calculates to:
R3 and C1 are then selected from standard value compo-
nents to produce the product calculated above. Typical values
for C1 are 3000 pF to 10,000 pF, and R3 is typically from 10
kΩ to 300 kΩ. C2 is then chosen large compared to C1, typ-
ically 0.1 µF. For this example, 3300 pF is chosen for C1,
requiring R3 to be 67.7 kΩ. A standard value 66.5 kΩ resistor
is selected.
The minimum current limit thresholds calculate to:
CIN, CBYP: These capacitors limit the voltage ripple at VIN by
supplying most of the switch current during the on-time. At
maximum load current, when Q1 is switched on, the current
through Q1 suddenly increases to the lower peak of the
inductor’s ripple current, then ramps up to the upper peak,
and then drops to zero at turn-off. The average current during
the on-time is the load current. For a worst case calculation,
these capacitors must supply this average load current during
the maximum on-time, while limiting the voltage drop at VIN.
For this example, 0.5V is selected as the maximum allowable
droop at VIN. Their minimum value is calculated from:
The load current in each case is equal to the current limit
threshold minus half the current ripple amplitude. The recom-
mended value of 1000 pF for CADJ is used in this example.
COUT: Since the maximum allowed output ripple voltage is
very low in this example (5 mVp-p), the minimum ripple con-
figuration (R3, C1, and C2 in the Block Diagram) must be
used. The resulting ripple at VOUT is then due to the inductor’s
ripple current passing through COUT. This capacitor’s value
can be selected based on the maximum allowable ripple volt-
age at VOUT, or based on transient response requirements.
The following calculation, based on ripple voltage, provides a
first order result for the value of COUT
:
A 33 µF electrolytic capacitor is selected for CIN, and a 1 µF
ceramic capacitor is selected for CBYP. Due to the ESR of
CIN, the ripple at VIN will likely be higher than the calculation
indicates, and therefore it may be desirable to increase CIN to
47 µF or 68 µF. CBYP must be located as close as possible to
the VIN and GND pins of the LM25085. The voltage rating for
both capacitors must be at least 42V. The RMS ripple current
rating for the input capacitors must also be considered. A
good approximation for the required ripple current rating is
IRMS > IOUT/2.
where IOR(max) is the maximum ripple current calculated
above, and VRIPPLE is the allowable ripple at VOUT
.
D1: A Schottky diode is recommended. Ultra-fast recovery
diodes are not recommended as the high speed transitions at
the SW pin may affect the regulator’s operation due to the
diode’s reverse recovery transients. The diode must be rated
for the maximum input voltage, and the worst case current
limit level. The average power dissipation in the diode is cal-
culated from:
A 100 µF capacitor is selected. Typically the ripple amplitude
will be higher than the calculations indicate due to the
capacitor’s ESR.
R3, C1, C2: The minimum ripple configuration uses these
three components to generate the ripple voltage required at
the FB pin since there is insufficient ripple at VOUT. A minimum
of 25 mVp-p must be applied to the FB pin to obtain stable
constant frequency operation. R3 and C1 are selected to
generate a sawtooth waveform at their junction, and that
waveform is AC coupled to the FB pin via C2. The values of
the three components are determined using the following pro-
cedure:
PD1 = VF x IOUT x (1-D)
where VF is the diode’s forward voltage drop, and D is the on-
time duty cycle. Using Equation 1, the minimum duty cycle
occurs at maximum input voltage, and is calculated to be
≊11.9% in this example. The diode power dissipation calcu-
lates to be:
PD1 = 0.65V x 5A x (1- 0.119) = 2.86W
Calculate VA = VOUT - (VSW x (1 – (VOUT/VIN(min))))
15
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CVCC: The capacitor at the VCC pin (from VIN to VCC) pro-
vides not only noise filtering and stability for the VCC regula-
tor, but also provides the surge current for the PFET gate
drive. The typical recommended value for CVCC is 0.47 µF. A
good quality, low ESR, ceramic capacitor is recommended.
CVCC must be located as close as possible to the VIN and
VCC pins. If the selected PFET has a Total Gate Charge
specification of 100 nC or larger, or if the circuit is required to
operate at input voltages below 7 volts, a larger capacitor may
be required. The maximum recommended value for CVCC is
1 µF.
maximum input voltage. The Total Gate Charge for the
Si7465 PFET is specified to be 40 nC (max) in its data sheet.
Therefore the total power dissipation within the LM25085 is
calculated to be:
PDISS = 42V x ((40 nC x 300 kHz) + 1.3 mA) = 559 mW
Using an MSOP-8EP package with a θJA of 46°C/W produces
a temperature rise of 26°C from junction to ambient.
Final Design Example Circuit
The final circuit is shown in Figure 4, and its performance is
presented in Figure 5 through Figure 8.
IC Power Dissipation: The maximum power dissipated in the
LM25085 package is calculated using Equation 12 at the
30079342
FIGURE 4. Example Circuit
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30079305
FIGURE 5. Efficiency vs. Load Current and VIN (Circuit of Figure 4)
30079344
FIGURE 6. Frequency vs. VIN (Circuit of Figure 4)
17
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30079345
FIGURE 7. Current Limit vs. VIN (Circuit of Figure 4)
30079346
FIGURE 8. LM25085 Power Dissipation (Circuit of Figure 4)
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18
voltage is passed to the FB pin via Cff. The ripple at VOUT can
be set as low as 25 mVp-p since it is not attenuated by RFB2
and RFB1. The minimum value for R4 is calculated from:
Alternate Output Ripple
Configurations
The minimum ripple configuration, using C1, C2 and R3, used
in the example circuit figure 4, results in a low ripple amplitude
at VOUT determined mainly by the characteristics of the output
capacitor and the ripple current in L1. This configuration al-
lows multiple ceramic capacitors to be used for VOUT if the
output voltage is provided to several places on the PC board.
However, if a slightly higher level of ripple at VOUT is accept-
able in the application, and distributed capacitance is not
used, the ripple required for the FB comparator pin can be
generated with fewer external components using the circuits
shown below.
where IOR(min) is the minimum ripple current, which occurs at
minimum input voltage. The minimum value for Cff is deter-
mined from:
a) Reduced ripple configuration: In Figure 9, R3, C1 and
C2 are removed (compared to Figure 4). A low value resistor
(R4) is added in series with COUT, and a capacitor (Cff) is
added across RFB2. Ripple is generated at VOUT by the
inductor’s ripple current flowing through R4, and that ripple
where tON(max) is the maximum on-time, which occurs at min-
imum VIN. The next larger standard value capacitor should
be used for Cff.
30079349
FIGURE 9. Reduced Ripple Configuration
b) Lowest cost configuration: This configuration, shown in
Figure 10, is the same as Figure 9 except Cff is removed.
Since the ripple voltage at VOUT is attenuated by RFB2 and
RFB1, the minimum ripple required at VOUT is equal to:
The minimum value for R4 is calculated from:
VRIP(min) = 25 mV x (RFB2 + RFB1)/RFB1
where IOR(min) is the minimum ripple current, which occurs at
minimum input voltage.
19
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30079351
FIGURE 10. Lowest Cost Ripple Generating Configuration
loop is that formed by CIN, Q1, L1, COUT, and back to CIN. The
second loop is that formed by D1, L1, COUT, and back to D1.
The connection from the anode of D1 to the ground end of
CIN must be short and direct. CIN must be as close as possible
to the VIN and GND pins, and CVCC must be as close as pos-
sible to the VIN and VCC pins.
PC Board Layout
In most applications, the heat sink pad or tab of Q1 is con-
nected to the switch node, i.e. the junction of Q1, L1 and D1.
While it is common to extend the PC board pad from under
these devices to aid in heat dissipation, the pad size should
be limited to minimize EMI radiation from this switching node.
If the PC board layout allows, a similarly sized copper pad can
be placed on the underside of the PC board, and connected
with as many vias as possible to aid in heat dissipation.
If the anticipated internal power dissipation of the LM25085
will produce excessive junction temperatures during normal
operation, a package option with an exposed pad must be
used (MSOP-8EP or LLP-8). Effective use of the PC board
ground plane can help dissipate heat. Additionally, the use of
wide PC board traces, where possible, helps conduct heat
away from the IC. Judicious positioning of the PC board within
the end product, along with the use of any available air flow
(forced or natural convection) also helps reduce the junction
temperature.
The voltage regulation, over-voltage, and current limit com-
parators are very fast and can respond to short duration noise
pulses. Layout considerations are therefore critical for opti-
mum performance. The layout must be as neat and compact
as possible with all the components as close as possible to
their associated pins. Two major current loops conduct cur-
rents which switch very fast, requiring the loops to be as small
as possible to minimize conducted and radiated EMI. The first
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20
Physical Dimensions inches (millimeters) unless otherwise noted
8-Lead MSOP-EP Package
NS Package Number MUY08A
8-Lead MSOP Package
NS Package Number MUA08A
21
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8-Lead LLP Package
NS Package Number SDA08A
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22
Notes
23
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Notes
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