LM2733YMFX [NSC]
0.6/1.6 MHz Boost Converters With 40V Internal FET; 0.6 / 1.6 MHz的升压转换器， 40V内部FET
| 型号： | LM2733YMFX |
| 厂家： | 
National Semiconductor |
| 描述： | 0.6/1.6 MHz Boost Converters With 40V Internal FET |
| 文件： | 总13页 (文件大小：505K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
February 2003
LM2733
0.6/1.6 MHz Boost Converters With 40V Internal FET
Switch in SOT-23
General Description
Features
n 40V DMOS FET switch
The LM2733 switching regulators are current-mode boost
converters operating fixed frequency of 1.6 MHz (“X” option)
and 600 kHz (“Y” option).
n 1.6 MHz (“X”), 0.6 MHz (“Y”) switching frequency
n Low RDS(ON) DMOS FET
n Switch current up to 1A
n Wide input voltage range (2.7V–14V)
n Low shutdown current ( 1 µA)
The use of SOT-23 package, made possible by the minimal
power loss of the internal 1A switch, and use of small induc-
tors and capacitors result in the industry’s highest power
density. The 40V internal switch makes these solutions per-
fect for boosting to voltages of 16V or greater.
<
n 5-Lead SOT-23 package
n Uses tiny capacitors and inductors
n Cycle-by-cycle current limiting
n Internally compensated
These parts have a logic-level shutdown pin that can be
used to reduce quiescent current and extend battery life.
Protection is provided through cycle-by-cycle current limiting
and thermal shutdown. Internal compensation simplifies de-
sign and reduces component count.
Applications
n White LED Current Source
n PDA’s and Palm-Top Computers
n Digital Cameras
Switch Frequency
X
Y
n Portable Phones and Games
n Local Boost Regulator
1.6 MHz
0.6 MHz
Typical Application Circuit
20055424
20055457
20055401
20055458
© 2003 National Semiconductor Corporation
DS200554
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Typical Application Circuit (Continued)
20055440
20055459
Connection Diagram
Top View
20055402
5-Lead SOT-23 Package
See NS Package Number MF05A
Ordering Information
Order Number Package Type Package Drawing
Supplied As
Package ID
S52A
LM2733XMF
1K Tape and Reel
3K Tape and Reel
1K Tape and Reel
3K Tape and Reel
LM2733XMFX
S52A
SOT23-5
MF05A
LM2733YMF
S52B
LM2733YMFX
S52B
Pin Description
Pin
1
Name
SW
Function
Drain of the internal FET switch.
Analog and power ground.
2
GND
FB
3
Feedback point that connects to external resistive divider.
Shutdown control input. Connect to VIN if this feature is not used.
Analog and power input.
4
SHDN
VIN
5
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2
Block Diagram
20055403
the Gm amplifier is derived from the feedback (which
samples the voltage at the output), the action of the PWM
comparator constantly sets the correct peak current through
the FET to keep the output volatge in regulation.
Theory of Operation
The LM2733 is a switching converter IC that operates at a
fixed frequency (0.6 or 1.6 MHz) using current-mode control
for fast transient response over a wide input voltage range
and incorporate pulse-by-pulse current limiting protection.
Because this is current mode control, a 50 mΩ sense resis-
tor in series with the switch FET is used to provide a voltage
(which is proportional to the FET current) to both the input of
the pulse width modulation (PWM) comparator and the cur-
rent limit amplifier.
Q1 and Q2 along with R3 - R6 form a bandgap voltage
reference used by the IC to hold the output in regulation. The
currents flowing through Q1 and Q2 will be equal, and the
feedback loop will adjust the regulated output to maintain
this. Because of this, the regulated output is always main-
tained at a voltage level equal to the voltage at the FB node
"multiplied up" by the ratio of the output resistive divider.
At the beginning of each cycle, the S-R latch turns on the
FET. As the current through the FET increases, a voltage
(proportional to this current) is summed with the ramp com-
ing from the ramp generator and then fed into the input of the
PWM comparator. When this voltage exceeds the voltage on
the other input (coming from the Gm amplifier), the latch
resets and turns the FET off. Since the signal coming from
The current limit comparator feeds directly into the flip-flop,
that drives the switch FET. If the FET current reaches the
limit threshold, the FET is turned off and the cycle terminated
until the next clock pulse. The current limit input terminates
the pulse regardless of the status of the output of the PWM
comparator.
3
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Absolute Maximum Ratings (Note 1)
FB Pin Voltage
−0.4V to +6V
−0.4V to +40V
−0.4V to +14.5V
SW Pin Voltage
Input Supply Voltage
Shutdown Input Voltage
(Survival)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Storage Temperature Range
Operating Junction
−65˚C to +150˚C
−0.4V to +14.5V
265˚C/W
θJ-A (SOT23-5)
Temperature Range
−40˚C to +125˚C
300˚C
ESD Rating (Note 3)
Human Body Model
Machine Model
Lead Temp. (Soldering, 5 sec.)
Power Dissipation (Note 2)
2 kV
Internally Limited
200V
Electrical Characteristics
Limits in standard typeface are for TJ = 25˚C, and limits in boldface type apply over the full operating temperature range
(−40˚C ≤ TJ ≤ +125˚C). Unless otherwise specified: VIN = 5V, VSHDN = 5V, IL = 0A.
Min
(Note 4)
2.7
Typical
(Note 5)
Max
(Note 4)
14
Symbol
Parameter
Input Voltage
Conditions
Units
VIN
ISW
V
A
Switch Current Limit
Switch ON Resistance
Shutdown Threshold
(Note 6)
1.0
1.5
R
DS(ON)
ISW = 100 mA
Device ON
Device OFF
VSHDN = 0
VSHDN = 5V
VIN = 3V
500
650
mΩ
SHDNTH
1.5
V
0.50
ISHDN
VFB
Shutdown Pin Bias Current
0
0
µA
2
Feedback Pin Reference
Voltage
1.205
1.230
1.255
V
IFB
IQ
Feedback Pin Bias Current
Quiescent Current
VFB = 1.23V
60
2.1
nA
mA
VSHDN = 5V, Switching "X"
VSHDN = 5V, Switching "Y"
VSHDN = 5V, Not Switching
VSHDN = 0
3.0
2
1.1
400
0.024
500
1
µA
FB Voltage Line Regulation
2.7V ≤ VIN ≤ 14V
0.02
%/V
MHz
FSW
DMAX
IL
Switching Frequency
Maximum Duty Cycle
Switch Leakage
“X” Option
1.15
0.40
87
1.6
0.60
93
1.85
0.8
“Y” Option
“X” Option
%
“Y” Option
93
96
Not Switching VSW = 5V
1
µA
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating the
device outside of the limits set forth under the operating ratings which specify the intended range of operating conditions.
Note 2: The maximum power dissipation which can be safely dissipated for any application is a function of the maximum junction temperature, T (MAX) = 125˚C,
J
the junction-to-ambient thermal resistance for the SOT-23 package, θ
= 265˚C/W, and the ambient temperature, T . The maximum allowable power dissipation
J-A
A
at any ambient temperature for designs using this device can be calculated using the formula:
If power dissipation exceeds the maximum specified above, the internal thermal protection circuitry will protect the device by reducing the output voltage as required
to maintain a safe junction temperature.
Note 3: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. The machine model is a 200 pF capacitor discharged
directly into each pin.
Note 4: Limits are guaranteed by testing, statistical correlation, or design.
Note 5: Typical values are derived from the mean value of a large quantity of samples tested during characterization and represent the most likely expected value
of the parameter at room temperature.
Note 6: Switch current limit is dependent on duty cycle (see Typical Performance Characteristics). Limits shown are for duty cycles ≤ 50%.
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4
Typical Performance Characteristics Unless otherwise specified: VIN = 5V, SHDN pin is tied to VIN
.
Iq VIN (Active) vs Temperature - "X"
Iq VIN (Active) vs Temperature - "Y"
20055410
20055442
Oscillator Frequency vs Temperature - "X"
Oscillator Frequency vs Temperature - "Y"
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20055443
Max. Duty Cycle vs Temperature - "X"
Max. Duty Cycle vs Temperature - "Y"
20055456
20055455
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Typical Performance Characteristics Unless otherwise specified: VIN = 5V, SHDN pin is tied to
VIN. (Continued)
Feedback Voltage vs Temperature
RDS(ON) vs Temperature
20055406
20055407
Current Limit vs Temperature
RDS(ON) vs VIN
20055409
20055423
Efficiency vs Load Current (VOUT = 12V) - "X"
Efficiency vs Load Current (VOUT = 15V) - "X"
20055414
20055445
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Typical Performance Characteristics Unless otherwise specified: VIN = 5V, SHDN pin is tied to
VIN. (Continued)
Efficiency vs Load Current (VOUT = 20V) - "X"
Efficiency vs Load Current (VOUT = 25V) - "X"
20055447
20055446
Efficiency vs Load Current (VOUT = 30V) - "X"
Efficiency vs Load Current (VOUT = 35V) - "X"
20055448
20055449
Efficiency vs Load Current (VOUT = 40V) - "X"
Efficiency vs Load (VOUT = 15V) - "Y"
20055435
20055450
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Typical Performance Characteristics Unless otherwise specified: VIN = 5V, SHDN pin is tied to
VIN. (Continued)
Efficiency vs Load (VOUT = 20V) - "Y"
Efficiency vs Load (VOUT = 25V) - "Y"
20055427
20055428
Efficiency vs Load (VOUT = 30V) - "Y"
Efficiency vs Load (VOUT = 35V) - "Y"
20055429
20055430
Efficiency vs Load (VOUT = 40V) - "Y"
20055432
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8
LAYOUT HINTS
Application Hints
High frequency switching regulators require very careful lay-
out of components in order to get stable operation and low
noise. All components must be as close as possible to the
LM2733 device. It is recommended that a 4-layer PCB be
used so that internal ground planes are available.
SELECTING THE EXTERNAL CAPACITORS
The best capacitors for use with the LM2733 are multi-layer
ceramic capacitors. They have the lowest ESR (equivalent
series resistance) and highest resonance frequency which
makes them optimum for use with high frequency switching
converters.
As an example, a recommended layout of components is
shown:
When selecting a ceramic capacitor, only X5R and X7R
dielectric types should be used. Other types such as Z5U
and Y5F have such severe loss of capacitance due to effects
of temperature variation and applied voltage, they may pro-
vide as little as 20% of rated capacitance in many typical
applications. Always consult capacitor manufacturer’s data
curves before selecting a capacitor. High-quality ceramic
capacitors can be obtained from Taiyo-Yuden, AVX, and
Murata.
SELECTING THE OUTPUT CAPACITOR
A single ceramic capacitor of value 4.7 µF to 10 µF will
provide sufficient output capacitance for most applications.
For output voltages below 10V, a 10 µF capacitance is
required. If larger amounts of capacitance are desired for
improved line support and transient response, tantalum ca-
pacitors can be used in parallel with the ceramics. Aluminum
electrolytics with ultra low ESR such as Sanyo Oscon can be
used, but are usually prohibitively expensive. Typical AI elec-
trolytic capacitors are not suitable for switching frequencies
above 500 kHz due to significant ringing and temperature
rise due to self-heating from ripple current. An output capaci-
tor with excessive ESR can also reduce phase margin and
cause instability.
20055422
Recommended PCB Component Layout
Some additional guidelines to be observed:
1. Keep the path between L1, D1, and C2 extremely short.
Parasitic trace inductance in series with D1 and C2 will
increase noise and ringing.
2. The feedback components R1, R2 and CF must be kept
close to the FB pin of U1 to prevent noise injection on
the FB pin trace.
3. If internal ground planes are available (recommended)
use vias to connect directly to ground at pin 2 of U1, as
well as the negative sides of capacitors C1 and C2.
SELECTING THE INPUT CAPACITOR
An input capacitor is required to serve as an energy reservoir
for the current which must flow into the coil each time the
switch turns ON. This capacitor must have extremely low
ESR, so ceramic is the best choice. We recommend a
nominal value of 2.2 µF, but larger values can be used. Since
this capacitor reduces the amount of voltage ripple seen at
the input pin, it also reduces the amount of EMI passed back
along that line to other circuitry.
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the external resistors R1 and
R2 (see Basic Application Circuit). A value of approximately
13.3 kΩ is recommended for R2 to establish a divider current
of approximately 92 µA. R1 is calculated using the formula:
R1 = R2 X (VOUT/1.23 − 1)
FEED-FORWARD COMPENSATION
SWITCHING FREQUENCY
Although internally compensated, the feed-forward capacitor
Cf is required for stability (see Basic Application Circuit).
Adding this capacitor puts a zero in the loop response of the
converter. Without it, the regulator loop can oscillate. The
recommended frequency for the zero fz should be approxi-
mately 8 kHz. Cf can be calculated using the formula:
The LM2733 is provided with two switching frequencies: the
“X” version is typically 1.6 MHz, while the “Y” version is
typically 600 kHz. The best frequency for a specific applica-
tion must be determined based on the tradeoffs involved:
Higher switching frequency means the inductors and capaci-
tors can be made smaller and cheaper for a given output
voltage and current. The down side is that efficiency is
slightly lower because the fixed switching losses occur more
frequently and become a larger percentage of total power
loss. EMI is typically worse at higher switching frequencies
because more EMI energy will be seen in the higher fre-
quency spectrum where most circuits are more sensitive to
such interference.
Cf = 1 / (2 X π X R1 X fz)
SELECTING DIODES
The external diode used in the typical application should be
a Schottky diode. If the switch voltage is less than 15V, a
20V diode such as the MBR0520 is recommended. If the
switch voltage is between 15V and 25V, a 30V diode such as
the MBR0530 is recommended. If the switch voltage ex-
ceeds 25V, a 40V diode such as the MBR0540 should be
used.
The MBR05XX series of diodes are designed to handle a
maximum average current of 0.5A. For applications exceed-
ing 0.5A average but less than 1A, a Microsemi UPS5817
can be used.
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Application Hints (Continued)
20055405
Basic Application Circuit
DUTY CYCLE
the inductor current to drop to zero during the cycle. It should
be noted that all boost converters shift over to discontinuous
operation as the output load is reduced far enough, but a
larger inductor stays “continuous” over a wider load current
range.
The maximum duty cycle of the switching regulator deter-
mines the maximum boost ratio of output-to-input voltage
that the converter can attain in continuous mode of opera-
tion. The duty cycle for a given boost application is defined
as:
To better understand these tradeoffs, a typical application
circuit (5V to 12V boost with a 10 µH inductor) will be
analyzed. We will assume:
VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V
Since the frequency is 1.6 MHz (nominal), the period is
approximately 0.625 µs. The duty cycle will be 62.5%, which
means the ON time of the switch is 0.390 µs. It should be
noted that when the switch is ON, the voltage across the
inductor is approximately 4.5V.
This applies for continuous mode operation.
The equation shown for calculating duty cycle incorporates
terms for the FET switch voltage and diode forward voltage.
The actual duty cycle measured in operation will also be
affected slightly by other power losses in the circuit such as
wire losses in the inductor, switching losses, and capacitor
ripple current losses from self-heating. Therefore, the actual
(effective) duty cycle measured may be slightly higher than
calculated to compensate for these power losses. A good
approximation for effctive duty cycle is :
Using the equation:
V = L (di/dt)
We can then calculate the di/dt rate of the inductor which is
found to be 0.45 A/µs during the ON time. Using these facts,
we can then show what the inductor current will look like
during operation:
DC (eff) = (1 - Efficiency x (VIN/VOUT))
Where the efficiency can be approximated from the curves
provided.
INDUCTANCE VALUE
The first question we are usually asked is: “How small can I
make the inductor?” (because they are the largest sized
component and usually the most costly). The answer is not
simple and involves tradeoffs in performance. Larger induc-
tors mean less inductor ripple current, which typically means
less output voltage ripple (for a given size of output capaci-
tor). Larger inductors also mean more load power can be
delivered because the energy stored during each switching
cycle is:
20055412
10 µH Inductor Current,
5V–12V Boost (LM2733X)
During the 0.390 µs ON time, the inductor current ramps up
0.176A and ramps down an equal amount during the OFF
time. This is defined as the inductor “ripple current”. It can
also be seen that if the load current drops to about 33 mA,
the inductor current will begin touching the zero axis which
means it will be in discontinuous mode. A similar analysis
can be performed on any boost converter, to make sure the
ripple current is reasonable and continuous operation will be
maintained at the typical load current values.
E =L/2 X (lp)2
Where “lp” is the peak inductor current. An important point to
observe is that the LM2733 will limit its switch current based
on peak current. This means that since lp(max) is fixed,
increasing L will increase the maximum amount of power
available to the load. Conversely, using too little inductance
may limit the amount of load current which can be drawn
from the output.
Best performance is usually obtained when the converter is
operated in “continuous” mode at the load current range of
interest, typically giving better load regulation and less out-
put ripple. Continuous operation is defined as not allowing
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10
Application Hints (Continued)
MAXIMUM SWITCH CURRENT
The maximum FET swtch current available before the cur-
rent limiter cuts in is dependent on duty cycle of the appli-
cation. This is illustrated in the graphs below which show
both the typical and guaranteed values of switch current for
both the "X" and "Y" versions as a function of effective
(actual) duty cycle:
The equation shown to calculate maximum load current
takes into account the losses in the inductor or turn-OFF
switching losses of the FET and diode. For actual load
current in typical applications, we took bench data for vari-
ous input and output voltages for both the "X" and "Y"
versions of the LM2733 and displayed the maximum load
current available for a typical device in graph form:
20055425
Switch Current Limit vs Duty Cycle - "X"
20055434
Max. Load Current vs VIN - "X"
20055426
Switch Current Limit vs Duty Cycle - "Y"
20055433
Max. Load Current vs VIN - "Y"
CALCULATING LOAD CURRENT
As shown in the figure which depicts inductor current, the
load current is related to the average inductor current by the
relation:
DESIGN PARAMETERS VSW AND ISW
The value of the FET "ON" voltage (referred to as VSW in the
equations) is dependent on load current. A good approxima-
tion can be obtained by multiplying the "ON Resistance" of
the FET times the average inductor current.
ILOAD = IIND(AVG) x (1 - DC)
Where "DC" is the duty cycle of the application. The switch
current can be found by:
FET on resistance increases at VIN values below 5V, since
the internal N-FET has less gate voltage in this input voltage
range (see Typical performance Characteristics curves).
Above VIN = 5V, the FET gate voltage is internally clamped
to 5V.
1
ISW = IIND(AVG) + ⁄
2
(IRIPPLE
)
Inductor ripple current is dependent on inductance, duty
cycle, input voltage and frequency:
IRIPPLE = DC x (VIN-VSW) / (f x L)
combining all terms, we can develop an expression which
allows the maximum available load current to be calculated:
11
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Application Hints (Continued)
The maximum peak switch current the device can deliver is
dependent on duty cycle. The minimum value is guaranteed
>
to be 1A at duty cycle below 50%. For higher duty cycles,
see Typical performance Characteristics curves.
THERMAL CONSIDERATIONS
20055413
At higher duty cycles, the increased ON time of the FET
means the maximum output current will be determined by
power dissipation within the LM2733 FET switch. The switch
power dissipation from ON-state conduction is calculated by:
Discontinuous Design, 5V–12V Boost (LM2733X)
The voltage across the inductor during ON time is 4.8V.
Minimum inductance value is found by:
P(SW) = DC x IIND(AVE)2 x RDSON
V = L X dl/dt, L = V X (dt/dl) = 4.8 (0.524µ/1) = 2.5 µH
There will be some switching losses as well, so some derat-
ing needs to be applied when calculating IC power dissipa-
tion.
In this case, a 2.7 µH inductor could be used assuming it
provided at least that much inductance up to the 1A current
value. This same analysis can be used to find the minimum
inductance for any boost application. Using the slower
switching “Y” version requires a higher amount of minimum
inductance because of the longer switching period.
MINIMUM INDUCTANCE
In some applications where the maximum load current is
relatively small, it may be advantageous to use the smallest
possible inductance value for cost and size savings. The
converter will operate in discontinuous mode in such a case.
INDUCTOR SUPPLIERS
Some of the recommended suppliers of inductors for this
product include, but not limited to are Sumida, Coilcraft,
Panasonic, TDK and Murata. When selecting an inductor,
make certain that the continuous current rating is high
enough to avoid saturation at peak currents. A suitable core
type must be used to minimize core (switching) losses, and
wire power losses must be considered when selecting the
current rating.
The minimum inductance should be selected such that the
inductor (switch) current peak on each cycle does not reach
the 1A current limit maximum. To understand how to do this,
an example will be presented.
In the example, the LM2733X will be used (nominal switch-
ing frequency 1.6 MHz, minimum switching frequency
1.15 MHz). This means the maximum cycle period is the
reciprocal of the minimum frequency:
SHUTDOWN PIN OPERATION
TON(max) = 1/1.15M = 0.870 µs
The device is turned off by pulling the shutdown pin low. If
this function is not going to be used, the pin should be tied
directly to VIN. If the SHDN function will be needed, a pull-up
resistor must be used to VIN (approximately 50k-100kΩ rec-
ommended). The SHDN pin must not be left unterminated.
We will assume the input voltage is 5V, VOUT = 12V, VSW
0.2V, VDIODE = 0.3V. The duty cycle is:
=
Duty Cycle = 60.3%
Therefore, the maximum switch ON time is 0.524 µs. An
inductor should be selected with enough inductance to pre-
vent the switch current from reaching 1A in the 0.524 µs ON
time interval (see below):
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12
Physical Dimensions inches (millimeters) unless otherwise noted
5-Lead SOT-23 Package
Order Number LM2733XMF, LM2733XMFX, LM2733YMF or LM2733YMFX
NS Package Number MF05A
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