LM2735YSDX/NOPB [NSC]
IC 3 A SWITCHING REGULATOR, 680 kHz SWITCHING FREQ-MAX, DSO6, LLP-6, Switching Regulator or Controller;
| 型号： | LM2735YSDX/NOPB |
| 厂家： | 
National Semiconductor |
| 描述： | IC 3 A SWITCHING REGULATOR, 680 kHz SWITCHING FREQ-MAX, DSO6, LLP-6, Switching Regulator or Controller 开关 |
| 文件： | 总42页 (文件大小：565K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
February 3, 2009
LM2735/LM2735Q
520kHz/1.6MHz – Space-Efficient Boost and SEPIC DC-DC
Regulator
General Description
Features
The LM2735 is an easy-to-use, space-efficient 2.1A low-side
switch regulator ideal for Boost and SEPIC DC-DC regulation.
It provides all the active functions to provide local DC/DC
conversion with fast-transient response and accurate regula-
tion in the smallest PCB area. Switching frequency is inter-
nally set to either 520kHz or 1.6MHz, allowing the use of
extremely small surface mount inductor and chip capacitors
while providing efficiencies up to 90%. Current-mode control
and internal compensation provide ease-of-use, minimal
component count, and high-performance regulation over a
wide range of operating conditions. External shutdown fea-
tures an ultra-low standby current of 80 nA ideal for portable
applications. Tiny SOT23-5, LLP-6, and eMSOP-8 packages
provide space-savings. Additional features include internal
soft-start, circuitry to reduce inrush current, pulse-by-pulse
current limit, and thermal shutdown.
Input voltage range 2.7V to 5.5V
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Output voltage range 3V to 24V
2.1A switch current over full temperature range
Current-Mode control
Logic high enable pin
Ultra low standby current of 80 nA in shutdown
170 mΩ NMOS switch
±2% feedback voltage accuracy
Ease-of-use, small total solution size
Internal soft-start
Internal compensation
Two switching frequencies
520 kHz (LM2735-Y)
1.6 MHz (LM2735-X)
Uses small surface mount inductors and chip capacitors
Tiny SOT23-5, LLP-6, and eMSOP-8 packages
LM2735Q is AEC-Q100 Grade 1 qualified and is
manufactured on an Automotive Grade Flow
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Applications
LCD Display Backlighting For Portable Applications
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OLED Panel Power Supply
USB Powered Devices
Digital Still and Video Cameras
White LED Current Source
Automotive
Typical Boost Application Circuit
20215801
Efficiency vs Load Current VO = 12202V15815
© 2009 National Semiconductor Corporation
202158
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Connection Diagrams
Top View
Top View
Top View
20215803
5-Pin SOT23
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20215805
6-Pin LLP
8-Pin eMSOP
Ordering Information
Order Number
Description
Package Type
Package
Drawing
Supplied As
Features
LM2735YMF
LM2735YMFX
LM2735YQMF
LM2735YQMFX
1000 units tape & reel
3000 units tape & reel
1000 units tape & reel
3000 units tape & reel
SOT23-5
MF05A
AEC-Q100 Grade 1
qualified. Automotive
Grade Production Flow*
520kHz
LM2735YSD
LM2735YSDX
LM2735YMY
LM2735YMYX
LM2735XMF
1000 units tape & reel
4500 units tape & reel
1000 units tape & reel
3500 units tape & reel
1000 units tape & reel
3000 units tape & reel
1000 units tape & reel
3000 units tape & reel
LLP-6
SDE06A
MUY08A
eMSOP-8
LM2735XMFX
LM2735XQMF
LM2735XQMFX
SOT23-5
MF05A
AEC-Q100 Grade 1
qualified. Automotive
Grade Production Flow*
1.6MHz
LM2735XSD
LM2735XSDX
LM2735XMY
LM2735XMYX
1000 units tape & reel
4500 units tape & reel
1000 units tape & reel
3500 units tape & reel
LLP-6
SDE06A
MUY08A
eMSOP-8
*Automotive Grade (Q) product incorporates enhanced manufacturing and support processes for the automotive market, including defect detection methodologies.
Reliability qualification is compliant with the requirements and temperature grades defined in the AEC-Q100 standard. Automotive grade products are identified
with the letter Q. For more information go to http://www.national.com/automotive.
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2
Pin Descriptions - 5-Pin SOT23
Pin
Name Function
1
SW
GND
FB
Output switch. Connect to the inductor, output diode.
Signal and power ground pin. Place the bottom resistor of the feedback network as close as possible to this
pin.
2
3
4
5
Feedback pin. Connect FB to external resistor divider to set output voltage.
Shutdown control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN +
0.3V.
EN
VIN
Supply voltage for power stage, and input supply voltage.
Pin Descriptions - 6-Pin LLP
Pin
1
Name Function
PGND Power ground pin. Place PGND and output capacitor GND close together.
2
VIN
Supply voltage for power stage, and input supply voltage.
Shutdown control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN +
0.3V.
3
4
EN
FB
Feedback pin. Connect FB to external resistor divider to set output voltage.
Signal ground pin. Place the bottom resistor of the feedback network as close as possible to this pin & pin
4.
5
AGND
SW
6
Output switch. Connect to the inductor, output diode.
Signal & Power ground. Connect to pin 1 & pin 5 on top layer. Place 4-6 vias from DAP to bottom layer GND
plane.
DAP
GND
Pin Descriptions - 8-Pin eMSOP
Pin
1
Name Function
No Connect
2
PGND Power ground pin. Place PGND and output capacitor GND close together.
3
VIN
EN
FB
Supply voltage for power stage, and input supply voltage.
Shutdown control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN +
0.3V.
4
5
6
7
8
Feedback pin. Connect FB to external resistor divider to set output voltage.
AGND Signal ground pin. Place the bottom resistor of the feedback network as close as possible to this pin & pin 5
SW
Output switch. Connect to the inductor, output diode.
No Connect
Signal & Power ground. Connect to pin 2 & pin 6 on top layer. Place 4-6 vias from DAP to bottom layer GND
plane.
DAP
GND
3
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Soldering Information
Infrared/Convection Reflow (15sec)
Absolute Maximum Ratings (Note 1)
220°C
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Operating Ratings (Note 1)
VIN
2.7V to 5.5V
3V to 24V
VIN
-0.5V to 7V
-0.5V to 26.5V
-0.5V to 3.0V
-0.5V to 7.0V
2kV
VSW
SW Voltage
FB Voltage
EN Voltage
ESD Susceptibility (Note 4)
Junction Temperature (Note 2)
Storage Temp. Range
VEN (Note 5)
0V to VIN
Junction Temperature Range
Power Dissipation
(Internal) SOT23-5
−40°C to +125°C
400 mW
150°C
-65°C to 150°C
Electrical Characteristics Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the
junction temperature range of (TJ = -40°C to 125°C). Minimum and Maximum limits are guaranteed through test, design, or statistical
correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only.
VIN = 5V unless otherwise indicated under the Conditions column.
Symbol
Parameter
Conditions
Min
Typ
Max Units
1.230 1.255 1.280
1.236 1.255 1.274
1.225 1.255 1.285
1.229 1.255 1.281
1.220 1.255 1.290
1.230 1.255 1.280
0.06
−40°C ≤ to TJ ≤ +125°C (SOT23-5)
0°C ≤ to TJ ≤ +125°C (SOT23-5)
−40°C ≤ to TJ ≤ +125°C (LLP-6)
−0°C ≤ to TJ ≤ +125°C (LLP-6)
−40°C ≤ to TJ ≤ +125°C (eMSOP-8)
VFB
Feedback Voltage
V
0°C ≤ to TJ ≤ +125°C (eMSOP-8)
VIN = 2.7V to 5.5V
Feedback Voltage Line Regulation
Feedback Input Bias Current
%/V
µA
ΔVFB/VIN
IFB
0.1
1
LM2735-X
1200 1600 2000
kHz
FSW
DMAX
Switching Frequency
Maximum Duty Cycle
Minimum Duty Cycle
Switch On Resistance
LM2735-Y
360
88
520
96
99
5
680
LM2735-X
%
%
LM2735-Y
91
LM2735-X
DMIN
LM2735-Y
2
SOT23-5 and eMSOP-8
LLP-6
170
190
3
330
350
RDS(ON)
mΩ
ICL
Switch Current Limit
Soft Start
2.1
A
SS
4
ms
mA
LM2735-X
LM2735-Y
All Options VEN = 0V
VIN Rising
VIN Falling
(Note 5)
7.0
3.4
80
2.3
1.9
11
7
Quiescent Current (switching)
IQ
Quiescent Current (shutdown)
Undervoltage Lockout
nA
V
2.65
0.4
UVLO
1.7
1.8
Shutdown Threshold Voltage
Enable Threshold Voltage
Switch Leakage
VEN_TH
V
(Note 5)
I-SW
I-EN
VSW = 24V
Sink/Source
1.0
µA
nA
Enable Pin Current
100
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Symbol
Parameter
Junction to Ambient
0 LFPM Air Flow (Note 3)
Conditions
LLP-6 and eMSOP-8 Package
SOT23-5 Package
Min
Typ
80
Max Units
θJA
°C/W
118
18
LLP-6 and eMSOP-8 Package
SOT23-5 Package
θJC
Junction to Case (Note 3)
°C/W
°C
60
TSD
Thermal Shutdown Temperature (Note 2)
Thermal Shutdown Hysteresis
160
10
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see Electrical Characteristics.
Note 2: Thermal shutdown will occur if the junction temperature exceeds the maximum junction temperature of the device
Note 3: Applies for packages soldered directly onto a 3” x 3” PC board with 2oz. copper on 4 layers in still air.
Note 4: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.
Note 5: Do not allow this pin to float or be greater than VIN +0.3V.
5
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Typical Performance Characteristics
Current Limit vs Temperature
FB Pin Voltage vs Temperature
20215806
20215807
Oscillator Frequency vs Temperature - "X"
Oscillator Frequency vs Temperature - "Y"
20215808
20215809
Typical Maximum Output Current vs VIN
RDSON vs Temperature
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20215811
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LM2735X Efficiency vs Load Current, Vo = 20V
LM2735Y Efficiency vs Load Current, Vo = 20V
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20215813
LM2735X Efficiency vs Load Current, Vo = 12V
LM2735Y Efficiency vs Load Current, Vo = 12V
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20215815
Output Voltage Load Regulation
Output Voltage Line Regulation
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20215817
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Simplified Internal Block Diagram
20215818
FIGURE 1. Simplified Block Diagram
at the falling edge of the reset pulse generated by the internal
oscillator. When this pulse goes low, the output control logic
turns on the internal NMOS control switch. During this on-
time, the SW pin voltage (VSW) decreases to approximately
GND, and the inductor current (IL) increases with a linear
slope. IL is measured by the current sense amplifier, which
generates an output proportional to the switch current. The
sensed signal is summed with the regulator’s corrective ramp
and compared to the error amplifier’s output, which is propor-
tional to the difference between the feedback voltage and
VREF. When the PWM comparator output goes high, the out-
put switch turns off until the next switching cycle begins.
During the switch off-time, inductor current discharges
through diode D1, which forces the SW pin to swing to the
output voltage plus the forward voltage (VD) of the diode. The
regulator loop adjusts the duty cycle (D) to maintain a con-
stant output voltage .
Application Information
THEORY OF OPERATION
The LM2735 is a constant frequency PWM boost regulator IC
that delivers a minimum of 2.1A peak switch current. The reg-
ulator has a preset switching frequency of either 520 kHz or
1.60 MHz. This high frequency allows the LM2735 to operate
with small surface mount capacitors and inductors, resulting
in a DC/DC converter that requires a minimum amount of
board space. The LM2735 is internally compensated, so it is
simple to use, and requires few external components. The
LM2735 uses current-mode control to regulate the output
voltage. The following operating description of the LM2735
will refer to the Simplified Block Diagram (Figure 1) the sim-
plified schematic (Figure 2), and its associated waveforms
(Figure 3). The LM2735 supplies a regulated output voltage
by switching the internal NMOS control switch at constant
frequency and variable duty cycle. A switching cycle begins
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20215819
FIGURE 2. Simplified Schematic
20215820
FIGURE 3. Typical Waveforms
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CURRENT LIMIT
The LM2735 uses cycle-by-cycle current limiting to protect
the internal NMOS switch. It is important to note that this cur-
rent limit will not protect the output from excessive current
during an output short circuit. The input supply is connected
to the output by the series connection of an inductor and a
diode. If a short circuit is placed on the output, excessive cur-
rent can damage both the inductor and diode.
20215824
Design Guide
FIGURE 4. Inductor Current
ENABLE PIN / SHUTDOWN MODE
The LM2735 has a shutdown mode that is controlled by the
Enable pin (EN). When a logic low voltage is applied to EN,
the part is in shutdown mode and its quiescent current drops
to typically 80 nA. Switch leakage adds up to another 1 µA
from the input supply. The voltage at this pin should never
exceed VIN + 0.3V.
THERMAL SHUTDOWN
A good design practice is to design the inductor to produce
10% to 30% ripple of maximum load. From the previous equa-
tions, the inductor value is then obtained.
Thermal shutdown limits total power dissipation by turning off
the output switch when the IC junction temperature exceeds
160°C. After thermal shutdown occurs, the output switch
doesn’t turn on until the junction temperature drops to ap-
proximately 150°C.
SOFT-START
This function forces VOUT to increase at a controlled rate dur-
ing start up. During soft-start, the error amplifier’s reference
voltage ramps to its nominal value of 1.255V in approximately
4.0ms. This forces the regulator output to ramp up in a more
linear and controlled fashion, which helps reduce inrush cur-
rent.
Where: 1/TS = FSW = switching frequency
One must also ensure that the minimum current limit (2.1A)
is not exceeded, so the peak current in the inductor must be
calculated. The peak current (ILPK ) in the inductor is calcu-
lated by:
INDUCTOR SELECTION
ILpk = IIN + ΔIL
The Duty Cycle (D) can be approximated quickly using the
ratio of output voltage (VO) to input voltage (VIN):
or
ILpk = IOUT / D' + ΔIL
When selecting an inductor, make sure that it is capable of
supporting the peak input current without saturating. Inductor
saturation will result in a sudden reduction in inductance and
prevent the regulator from operating correctly. Because of the
speed of the internal current limit, the peak current of the in-
ductor need only be specified for the required maximum input
current. For example, if the designed maximum input current
is 1.5A and the peak current is 1.75A, then the inductor should
be specified with a saturation current limit of >1.75A. There is
no need to specify the saturation or peak current of the in-
ductor at the 3A typical switch current limit.
Therefore:
Power losses due to the diode (D1) forward voltage drop, the
voltage drop across the internal NMOS switch, the voltage
drop across the inductor resistance (RDCR) and switching
losses must be included to calculate a more accurate duty
cycle (See Calculating Efficiency and Junction Temperature
for a detailed explanation). A more accurate formula for cal-
culating the conversion ratio is:
Because of the operating frequency of the LM2735, ferrite
based inductors are preferred to minimize core losses. This
presents little restriction since the variety of ferrite-based in-
ductors is huge. Lastly, inductors with lower series resistance
(DCR) will provide better operating efficiency. For recom-
mended inductors see Example Circuits.
INPUT CAPACITOR
An input capacitor is necessary to ensure that VIN does not
drop excessively during switching transients. The primary
specifications of the input capacitor are capacitance, voltage,
RMS current rating, and ESL (Equivalent Series Inductance).
The recommended input capacitance is 10 µF to 44 µF de-
pending on the application. The capacitor manufacturer
specifically states the input voltage rating. Make sure to check
any recommended deratings and also verify if there is any
Where η equals the efficiency of the LM2735 application.
The inductor value determines the input ripple current. Lower
inductor values decrease the size of the inductor, but increase
the input ripple current. An increase in the inductor value will
decrease the input ripple current.
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significant change in capacitance at the operating input volt-
age and the operating temperature. The ESL of an input
capacitor is usually determined by the effective cross sec-
tional area of the current path. At the operating frequencies
of the LM2735, certain capacitors may have an ESL so large
that the resulting impedance (2πfL) will be higher than that
required to provide stable operation. As a result, surface
mount capacitors are strongly recommended. Multilayer ce-
ramic capacitors (MLCC) are good choices for both input and
output capacitors and have very low ESL. For MLCCs it is
recommended to use X7R or X5R dielectrics. Consult capac-
itor manufacturer datasheet to see how rated capacitance
varies over operating conditions.
OUTPUT CAPACITOR
20215829
The LM2735 operates at frequencies allowing the use of ce-
ramic output capacitors without compromising transient re-
sponse. Ceramic capacitors allow higher inductor ripple
without significantly increasing output ripple. The output ca-
pacitor is selected based upon the desired output ripple and
transient response. The initial current of a load transient is
provided mainly by the output capacitor. The output
impedance will therefore determine the maximum voltage
perturbation. The output ripple of the converter is a function
of the capacitor’s reactance and its equivalent series resis-
tance (ESR):
FIGURE 5. Setting Vout
A good value for R1 is 10kΩ.
COMPENSATION
The LM2735 uses constant frequency peak current mode
control. This mode of control allows for a simple external
compensation scheme that can be optimized for each appli-
cation. A complicated mathematical analysis can be complet-
ed to fully explain the LM2735’s internal & external compen-
sation, but for simplicity, a graphical approach with simple
equations will be used. Below is a Gain & Phase plot of a
LM2735 that produces a 12V output from a 5V input voltage.
The Bode plot shows the total loop Gain & Phase without ex-
ternal compensation.
When using MLCCs, the ESR is typically so low that the ca-
pacitive ripple may dominate. When this occurs, the output
ripple will be approximately sinusoidal and 90° phase shifted
from the switching action .
Given the availability and quality of MLCCs and the expected
output voltage of designs using the LM2735, there is really no
need to review any other capacitor technologies. Another
benefit of ceramic capacitors is their ability to bypass high
frequency noise. A certain amount of switching edge noise
will couple through parasitic capacitances in the inductor to
the output. A ceramic capacitor will bypass this noise while a
tantalum will not. Since the output capacitor is one of the two
external components that control the stability of the regulator
control loop, most applications will require a minimum at 4.7
µF of output capacitance. Like the input capacitor, recom-
mended multilayer ceramic capacitors are X7R or X5R.
Again, verify actual capacitance at the desired operating volt-
age and temperature.
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the following equation where
R1 is connected between the FB pin and GND, and R2 is
connected between VOUT and the FB pin.
20215831
FIGURE 6. LM2735 Without External Compensation
One can see that the Crossover frequency is fine, but the
phase margin at 0dB is very low (22°). A zero can be placed
just above the crossover frequency so that the phase margin
will be bumped up to a minimum of 45°. Below is the same
application with a zero added at 8 kHz.
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20215829
FIGURE 8. Setting External Pole-Zero
20215832
FIGURE 7. LM2735 With External Compensation
The simplest method to determine the compensation compo-
nent value is as follows.
There is an associated pole with the zero that was created in
the above equation.
Set the output voltage with the following equation.
It is always higher in frequency than the zero.
A right-half plane zero (RHPZ) is inherent to all boost con-
verters. One must remember that the gain associated with a
right-half plane zero increases at 20dB per decade, but the
phase decreases by 45° per decade. For most applications
there is little concern with the RHPZ due to the fact that the
frequency at which it shows up is well beyond crossover, and
has little to no effect on loop stability. One must be concerned
with this condition for large inductor values and high output
currents.
Where R1 is the bottom resistor and R2 is the resistor tied to
the output voltage. The next step is to calculate the value of
C3. The internal compensation has been designed so that
when a zero is added between 5 kHz & 10 kHz the converter
will have good transient response with plenty of phase margin
for all input & output voltage combinations.
Lower output voltages will have the zero set closer to 10 kHz,
and higher output voltages will usually have the zero set clos-
er to 5 kHz. It is always recommended to obtain a Gain/Phase
plot for your actual application. One could refer to the Typical
applications section to obtain examples of working applica-
tions and the associated component values.
There are miscellaneous poles and zeros associated with
parasitics internal to the LM2735, external components, and
the PCB. They are located well over the crossover frequency,
and for simplicity are not discussed.
Pole @ origin due to internal gm amplifier:
FP-ORIGIN
PCB Layout Considerations
Pole due to output load and capacitor:
When planning layout there are a few things to consider when
trying to achieve a clean, regulated output. The most impor-
tant consideration when completing a Boost Converter layout
is the close coupling of the GND connections of the COUT ca-
pacitor and the LM2735 PGND pin. The GND ends should be
close to one another and be connected to the GND plane with
at least two through-holes. There should be a continuous
ground plane on the bottom layer of a two-layer board except
under the switching node island. The FB pin is a high
impedance node and care should be taken to make the FB
trace short to avoid noise pickup and inaccurate regulation.
The feedback resistors should be placed as close as possible
to the IC, with the AGND of R1 placed as close as possible to
the GND (pin 5 for the LLP) of the IC. The VOUT trace to R2
should be routed away from the inductor and any other traces
that are switching. High AC currents flow through the VIN, SW
This equation only determines the frequency of the pole for
perfect current mode control (CMC). I.e, it doesn’t take into
account the additional internal artificial ramp that is added to
the current signal for stability reasons. By adding artificial
ramp, you begin to move away from CMC to voltage mode
control (VMC). The artifact is that the pole due to the output
load and output capacitor will actually be slightly higher in fre-
quency than calculated. In this example it is calculated at 650
Hz, but in reality it is around 1 kHz.
The zero created with capacitor C3 & resistor R2:
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and VOUT traces, so they should be as short and wide as pos-
sible. However, making the traces wide increases radiated
noise, so the designer must make this trade-off. Radiated
noise can be decreased by choosing a shielded inductor. The
remaining components should also be placed as close as
possible to the IC. Please see Application Note AN-1229 for
further considerations and the LM2735 demo board as an ex-
ample of a four-layer layout.
that are obtainable via the National Semiconductor website.
The demonstration board consists of a two layer PCB with a
common input and output voltage application. Most of the
routing is on the top layer, with the bottom layer consisting of
a large ground plane. The placement of the external compo-
nents satisfies the electrical considerations, and the thermal
performance has been improved by adding thermal vias and
a top layer “Dog-Bone”.
Below is an example of a good thermal & electrical PCB de-
sign. This is very similar to our LM2735 demonstration boards
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Example of Proper PCB Layout
guidelines). The electrical design considerations outweigh the
thermal considerations. Other factors that influence thermal
performance are thermal vias, copper weight, and number of
board layers.
Definitions
Heat energy is transferred from regions of high temperature
to regions of low temperature via three basic mechanisms:
radiation, conduction and convection.
Radiation: Electromagnetic transfer of heat between masses
at different temperatures.
Conduction: Transfer of heat through a solid medium.
Convection: Transfer of heat through the medium of a fluid;
typically air.
Conduction & Convection will be the dominant heat transfer
mechanism in most applications.
RθJA: Thermal impedance from silicon junction to ambient air
temperature.
RθJC: Thermal impedance from silicon junction to device case
temperature.
20215840
CθJC: Thermal Delay from silicon junction to device case tem-
perature.
FIGURE 9. Boost PCB Layout Guidelines
CθCA: Thermal Delay from device case to ambient air tem-
perature.
Thermal Design
When designing for thermal performance, one must consider
many variables:
RθJA
& RθJC: These two symbols represent thermal
impedances, and most data sheets contain associated values
for these two symbols. The units of measurement are °C/
Watt.
Ambient Temperature: The surrounding maximum air tem-
perature is fairly explanatory. As the temperature increases,
the junction temperature will increase. This may not be linear
though. As the surrounding air temperature increases, resis-
tances of semiconductors, wires and traces increase. This will
decrease the efficiency of the application, and more power
will be converted into heat, and will increase the silicon junc-
tion temperatures further.
RθJA is the sum of smaller thermal impedances (see simplified
thermal model below). The capacitors represent delays that
are present from the time that power and its associated heat
is increased or decreased from steady state in one medium
until the time that the heat increase or decrease reaches
steady state on the another medium.
Forced Airflow: Forced air can drastically reduce the device
junction temperature. Air flow reduces the hot spots within a
design. Warm airflow is often much better than a lower am-
bient temperature with no airflow.
External Components: Choose components that are effi-
cient, and you can reduce the mutual heating between de-
vices.
PCB design with thermal performance in mind:
The PCB design is a very important step in the thermal design
procedure. The LM2735 is available in three package options
(5 pin SOT23, 8 pin eMSOP & 6 pin LLP). The options are
electrically the same, but difference between the packages is
size and thermal performance. The LLP and eMSOP have
thermal Die Attach Pads (DAP) attached to the bottom of the
packages, and are therefore capable of dissipating more heat
than the SOT23 package. It is important that the customer
choose the correct package for the application. A detailed
thermal design procedure has been included in this data
sheet. This procedure will help determine which package is
correct, and common applications will be analyzed.
20215841
FIGURE 10. Simplified Thermal Impedance Model
There is one significant thermal PCB layout design consider-
ation that contradicts a proper electrical PCB layout design
consideration. This contradiction is the placement of external
components that dissipate heat. The greatest external heat
contributor is the external Schottky diode. It would be nice if
you were able to separate by distance the LM2735 from the
Schottky diode, and thereby reducing the mutual heating ef-
fect. This will however create electrical performance issues.
It is important to keep the LM2735, the output capacitor, and
Schottky diode physically close to each other (see PCB layout
www.national.com
14
The datasheet values for these symbols are given so that one
might compare the thermal performance of one package
against another. In order to achieve a comparison between
packages, all other variables must be held constant in the
comparison (PCB size, copper weight, thermal vias, power
dissipation, VIN, VOUT, Load Current etc). This does shed light
on the package performance, but it would be a mistake to use
these values to calculate the actual junction temperature in
your application.
RθJA [Variables]:
•
•
•
•
•
Input Voltage, Output Voltage, Output Current, RDSon.
Ambient temperature & air flow.
Internal & External components power dissipation.
Package thermal limitations.
PCB variables (copper weight, thermal via’s, layers
component placement).
It would be wrong to assume that the top case temperature is
the proper temperature when calculating value. The
value represents the thermal impedance of all six sides
of a package, not just the top side. This document will refer to
a thermal impedance called
.
represents a thermal
impedance associated with just the top case temperature.
This will allow one to calculate the junction temperature with
a thermal sensor connected to the top case.
We will talk more about calculating the variables of this equa-
tion later, and how to eventually calculate a proper junction
temperature with relative certainty. For now we need to define
the process of calculating the junction temperature and clarify
some common misconceptions.
15
www.national.com
and loads. All loss elements will mutually increase the heat
on the PCB, and therefore increase each other’s tempera-
tures.
LM2735 Thermal Models
Heat is dissipated from the LM2735 and other devices. The
external loss elements include the Schottky diode, inductor,
20215843
FIGURE 11. Thermal Schematic
20215844
FIGURE 12. Associated Thermal Model
www.national.com
16
Calculating Efficiency, and Junction
Temperature
The complete LM2735 DC/DC converter efficiency (η) can be
calculated in the following manner.
The diode, NMOS switch, and inductor DCR losses are in-
cluded in this calculation. Setting any loss element to zero will
simplify the equation.
VD is the forward voltage drop across the Schottky diode. It
can be obtained from the manufacturer’s Electrical Charac-
teristics section of the data sheet.
The conduction losses in the diode are calculated as follows:
PDIODE = VD x IO
Depending on the duty cycle, this can be the single most sig-
nificant power loss in the circuit. Care should be taken to
choose a diode that has a low forward voltage drop. Another
concern with diode selection is reverse leakage current. De-
pending on the ambient temperature and the reverse voltage
across the diode, the current being drawn from the output to
the NMOS switch during time D could be significant, this may
increase losses internal to the LM2735 and reduce the overall
efficiency of the application. Refer to Schottky diode
manufacturer’s data sheets for reverse leakage specifica-
tions, and typical applications within this data sheet for diode
selections.
Power loss (PLOSS) is the sum of two types of losses in the
converter, switching and conduction. Conduction losses usu-
ally dominate at higher output loads, where as switching
losses remain relatively fixed and dominate at lower output
loads.
Losses in the LM2735 Device: PLOSS = PCOND + PSW + PQ
Conversion ratio of the Boost Converter with conduction loss
elements inserted:
Another significant external power loss is the conduction loss
in the input inductor. The power loss within the inductor can
be simplified to:
2
PIND = IIN RDCR
One can see that if the loss elements are reduced to zero, the
conversion ratio simplifies to:
The LM2735 conduction loss is mainly associated with the
internal NFET:
And we know:
Therefore:
PCOND-NFET = I2SW-rms x RDSON x D
20215852
FIGURE 13. LM2735 Switch Current
Calculations for determining the most significant power loss-
es are discussed below. Other losses totaling less than 2%
are not discussed.
A simple efficiency calculation that takes into account the
conduction losses is shown below:
(small ripple approximation)
PCOND-NFET = IIN2 x RDSON x D
17
www.national.com
PCONDUCTION = IIN2 x D x RDSON x 305 mW
Diode Losses
VD = 0.45V
PDIODE = VD x IIN(1-D) = 236 mW
The value for should be equal to the resistance at the junction
temperature you wish to analyze. As an example, at 125°C
and VIN = 5V, RDSON = 250 mΩ (See typical graphs for value).
Inductor Power Losses
RDCR = 75 mΩ
Switching losses are also associated with the internal NMOS
switch. They occur during the switch on and off transition pe-
riods, where voltages and currents overlap resulting in power
loss.
PIND = IIN2 x RDCR = 145 mW
Total Power Losses are:
TABLE 2. Power Loss Tabulation
5V
The simplest means to determine this loss is to empirically
measuring the rise and fall times (10% to 90%) of the switch
at the switch node:
VIN
VOUT
IOUT
VD
12V
500mA
0.4V
POUT
6W
PSWR = 1/2(VOUT x IIN x FSW x TRISE
)
PDIODE
236mW
PSWF = 1/2(VOUT x IIN x FSW x TFALL
PSW = PSWR + PSWF
)
FSW
TRISE
TFALL
IQ
1.6MHz
6nS
PSWR
PSWF
PQ
80mW
70mW
20mW
305mW
145mW
Typical Switch-Node Rise and Fall Times
5nS
VIN
3V
5V
3V
5V
VOUT
5V
TRISE
6nS
6nS
7nS
7nS
TFALL
4nS
5nS
5nS
5nS
4mA
RDSon
RDCR
D
PCOND
PIND
250mΩ
75mΩ
0.623
86%
12V
12V
18V
PLOSS
856mW
η
Quiescent Power Losses
IQ is the quiescent operating current, and is typically around
4mA.
PINTERNAL = PCOND + PSW = 475 mW
PQ = IQ x VIN
Calculating
and
Example Efficiency Calculation:
TABLE 1. Operating Conditions
VIN
VOUT
IOUT
VD
5V
12V
500mA
0.4V
FSW
IQ
1.60MHz
4mA
TRISE
TFALL
RDSon
RDCR
D
6nS
We now know the internal power dissipation, and we are try-
ing to keep the junction temperature at or below 125°C. The
5nS
next step is to calculate the value for
and/or
. This is
250mΩ
50mΩ
0.64
actually very simple to accomplish, and necessary if you think
you may be marginal with regards to thermals or determining
what package option is correct.
IIN
1.4A
The LM2735 has a thermal shutdown comparator. When the
silicon reaches a temperature of 160°C, the device shuts
down until the temperature reduces to 150°C. Knowing this,
ΣPCOND + PSW + PDIODE + PIND + PQ = PLOSS
Quiescent Power Losses
PQ = IQ x VIN = 20 mW
Switching Power Losses
one can calculate the
or the
of a specific application.
Because the junction to top case thermal impedance is much
lower than the thermal impedance of junction to ambient air,
the error in calculating
you will need to attach a small thermocouple onto the top case
of the LM2735 to obtain the value.
is lower than for
. However,
PSWR = 1/2(VOUT x IIN x FSW x TRISE) ≊ 6 ns ≊ 80 mW
PSWF = 1/2(VOUT x IIN x FSW x TFALL) ≊ 5 ns ≊ 70 mW
PSW = PSWR + PSWF = 150 mW
Internal NFET Power Losses
Knowing the temperature of the silicon when the device shuts
down allows us to know three of the four variables. Once we
calculate the thermal impedance, we then can work back-
wards with the junction temperature set to 125°C to see what
RDSON = 250 mΩ
www.national.com
18
maximum ambient air temperature keeps the silicon below
the 125°C temperature.
Procedure:
Place your application into a thermal chamber. You will need
to dissipate enough power in the device so you can obtain a
good thermal impedance value.
Raise the ambient air temperature until the device goes into
thermal shutdown. Record the temperatures of the ambient
air and/or the top case temperature of the LM2735. Calculate
the thermal impedances.
Example from previous calculations:
Pdiss = 475 mW
Ta @ Shutdown = 139°C
Tc @ Shutdown = 155°C
20215856
FIGURE 14. RθJA vs Internal Dissipation for the LLP-6
and eMSOP-8 Package
LLP = 55°C/W
LLP = 21°C/W
SEPIC Converter
LLP & eMSOP typical applications will produce
in the range of 50°C/W to 65°C/W, and will vary between
numbers
The LM2735 can easily be converted into a SEPIC converter.
A SEPIC converter has the ability to regulate an output volt-
age that is either larger or smaller in magnitude than the input
voltage. Other converters have this ability as well (CUK and
Buck-Boost), but usually create an output voltage that is op-
posite in polarity to the input voltage. This topology is a perfect
fit for Lithium Ion battery applications where the input voltage
for a single cell Li-Ion battery will vary between 3V & 4.5V and
the output voltage is somewhere in between. Most of the
analysis of the LM2735 Boost Converter is applicable to the
LM2735 SEPIC Converter.
18°C/W and 28°C/W. These values are for PCB’s with two
and four layer boards with 0.5 oz copper, and four to six ther-
mal vias to bottom side ground plane under the DAP.
For 5-pin SOT23 package typical applications, RθJA numbers
will range from 80°C/W to 110°C/W, and
will vary between
50°C/W and 65°C/W. These values are for PCB’s with two &
four layer boards with 0.5 oz copper, with two to four thermal
vias from GND pin to bottom layer.
Here is a good rule of thumb for typical thermal impedances,
and an ambient temperature maximum of 75°C: If your design
requires that you dissipate more than 400mW internal to the
LM2735, or there is 750mW of total power loss in the appli-
cation, it is recommended that you use the 6 pin LLP or the 8
pin eMSOP package.
SEPIC Design Guide:
SEPIC Conversion ratio without loss elements:
Note: To use these procedures it is important to dissipate an
amount of power within the device that will indicate a true
thermal impedance value. If one uses a very small internal
dissipated value, one can see that the thermal impedance
calculated is abnormally high, and subject to error. The graph
below shows the nonlinear relationship of internal power dis-
Therefore:
sipation vs .
.
Small ripple approximation:
In a well-designed SEPIC converter, the output voltage, and
input voltage ripple, the inductor ripple and is small in com-
parison to the DC magnitude. Therefore it is a safe approxi-
mation to assume a DC value for these components. The
main objective of the Steady State Analysis is to determine
the steady state duty-cycle, voltage and current stresses on
all components, and proper values for all components.
In a steady-state converter, the net volt-seconds across an
inductor after one cycle will equal zero. Also, the charge into
a capacitor will equal the charge out of a capacitor in one cy-
cle.
Therefore:
19
www.national.com
Applying Charge balance on C1:
Since there are no DC voltages across either inductor, and
capacitor C6 is connected to Vin through L1 at one end, or to
ground through L2 on the other end, we can say that
Substituting IL1 into IL2
VC1 = VIN
Therefore:
The average inductor current of L2 is the average output load.
This verifies the original conversion ratio equation.
It is important to remember that the internal switch current is
equal to IL1 and IL2. During the D interval. Design the converter
so that the minimum guaranteed peak switch current limit
(2.1A) is not exceeded.
20215863
FIGURE 15. Inductor Volt-Sec Balance Waveform
20215880
FIGURE 16. SEPIC CONVERTER Schematic
www.national.com
20
Steady State Analysis with Loss
Elements
20215866
Using inductor volt-second balance & capacitor charge bal-
ance, the following equations are derived:
20215890
Efficiencies for Typical SEPIC Application
SEPIC Converter PCB Layout
The layout guidelines described for the LM2735 Boost-Con-
verter are applicable to the SEPIC Converter. Below is a
proper PCB layout for a SEPIC Converter.
Therefore:
One can see that all variables are known except for the duty
cycle (D). A quadratic equation is needed to solve for D. A
less accurate method of determining the duty cycle is to as-
sume efficiency, and calculate the duty cycle.
20215872
FIGURE 17. SEPIC PCB Layout
21
www.national.com
LLP Package
The LM2735 packaged in the 6–pin LLP:
20215873
FIGURE 18. Internal LLP Connection
For certain high power applications, the PCB land may be
modified to a "dog bone" shape (see Figure 19). Increasing
the size of ground plane, and adding thermal vias can reduce
the RθJA for the application.
20215874
FIGURE 19. PCB Dog Bone Layout
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22
LM2735X SOT23-5 Design Example 1
20215875
LM2735X (1.6MHz): Vin = 5V, Vout = 12V @ 350mA
Part ID
Part Value
2.1A Boost Regulator
22µF, 6.3V, X5R
10µF, 25V, X5R
330pF
Manufacturer
NSC
Part Number
LM2735XMF
U1
C1, Input Cap
C2 Output Cap
C3 Comp Cap
D1, Catch Diode
L1
TDK
C2012X5R0J226M
C3216X5R1E106M
C1608X5R1H331K
STPS120M
TDK
TDK
0.4Vf Schottky 1A, 20VR
15µH 1.5A
ST
Coilcraft
Vishay
Vishay
Vishay
MSS5131-153ML
CRCW06031022F
CRCW06038662F
CRCW06031003F
R1
10.2kΩ, 1%
R2
86.6kΩ, 1%
R3
100kΩ, 1%
23
www.national.com
LM2735Y SOT23-5 Design Example 2
20215875
LM2735Y (520kHz): Vin = 5V, Vout = 12V @ 350mA
Part ID
Part Value
2.1A Boost Regulator
22µF, 6.3V, X5R
10µF, 25V, X5R
330pF
Manufacturer
NSC
Part Number
LM2735YMF
U1
C1, Input Cap
C2 Output Cap
C3 Comp Cap
D1, Catch Diode
L1
TDK
C2012X5R0J226M
C3216X5R1E106M
C1608X5R1H331K
STPS120M
TDK
TDK
0.4Vf Schottky 1A, 20VR
33µH 1.5A
ST
Coilcraft
Vishay
Vishay
Vishay
DS3316P-333ML
CRCW06031022F
CRCW06038662F
CRCW06031003F
R1
10.2kΩ, 1%
R2
86.6kΩ, 1%
R3
100kΩ, 1%
www.national.com
24
LM2735X LLP-6 Design Example 3
20215876
LM2735X (1.6MHz): Vin = 3.3V, Vout = 12V @ 350mA
Part ID
U1
Part Value
2.1A Boost Regulator
22µF, 6.3V, X5R
No Load
Manufacturer
NSC
Part Number
LM2735XSD
C1 Input Cap
C2 Input Cap
C3 Output Cap
C4 Output Cap
C5 Comp Cap
D1, Catch Diode
L1
TDK
C2012X5R0J226M
10µF, 25V, X5R
No Load
TDK
C3216X5R1E106M
330pF
TDK
ST
C1608X5R1H331K
STPS120M
0.4Vf Schottky 1A, 20VR
6.8µH 2A
Coilcraft
Vishay
Vishay
Vishay
DO1813H-682ML
CRCW06031022F
CRCW06038662F
CRCW06031003F
R1
10.2kΩ, 1%
R2
86.6kΩ, 1%
R3
100kΩ, 1%
25
www.national.com
LM2735Y LLP-6 Design Example 4
20215876
LM2735Y (520kHz): Vin = 3.3V, Vout = 12V @ 350mA
Part ID
U1
Part Value
2.1A Boost Regulator
22µF, 6.3V, X5R
No Load
Manufacturer
NSC
Part Number
LM2735YSD
C1 Input Cap
C2 Input Cap
C3 Output Cap
C4 Output Cap
C5 Comp Cap
D1, Catch Diode
L1
TDK
C2012X5R0J226M
10µF, 25V, X5R
No Load
TDK
C3216X5R1E106M
330pF
TDK
ST
C1608X5R1H331K
STPS120M
0.4Vf Schottky 1A, 20VR
15µH 2A
Coilcraft
Vishay
Vishay
Vishay
MSS5131-153ML
CRCW06031022F
CRCW06038662F
CRCW06031003F
R1
10.2kΩ, 1%
R2
86.6kΩ, 1%
R3
100kΩ, 1%
www.national.com
26
LM2735Y eMSOP-8 Design Example 5
20215877
LM2735Y (520kHz): Vin = 3.3V, Vout = 12V @ 350mA
Part ID
U1
Part Value
2.1A Boost Regulator
22µF, 6.3V, X5R
No Load
Manufacturer
NSC
Part Number
LM2735YMY
C1 Input Cap
C2 Input Cap
C3 Output Cap
C4 Output Cap
C5 Comp Cap
D1, Catch Diode
L1
TDK
C2012X5R0J226M
10µF, 25V, X5R
No Load
TDK
C3216X5R1E106M
330pF
TDK
ST
C1608X5R1H331K
STPS120M
0.4Vf Schottky 1A, 20VR
15µH 1.5A
Coilcraft
Vishay
Vishay
Vishay
MSS5131-153ML
CRCW06031022F
CRCW06038662F
CRCW06031003F
R1
10.2kΩ, 1%
R2
86.6kΩ, 1%
R3
100kΩ, 1%
27
www.national.com
LM2735X SOT23-5 Design Example 6
20215878
LM2735X (1.6MHz): Vin = 3V, Vout = 5V @ 500mA
Part ID
Part Value
2.1A Boost Regulator
10µF, 6.3V, X5R
10µF, 6.3V, X5R
1000pF
Manufacturer
NSC
Part Number
LM2735XMF
U1
C1, Input Cap
C2, Output Cap
C3 Comp Cap
D1, Catch Diode
L1
TDK
C2012X5R0J106K
C2012X5R0J106K
C1608X5R1H102K
STPS120M
TDK
TDK
0.4Vf Schottky 1A, 20VR
10µH 1.2A
ST
Coilcraft
Vishay
Vishay
Vishay
DO1608C-103ML
CRCW08051002F
CRCW08053012F
CRCW06031003F
R1
10.0kΩ, 1%
R2
30.1kΩ, 1%
R3
100kΩ, 1%
www.national.com
28
LM2735Y SOT23-5 Design Example 7
20215878
LM2735Y (520kHz): Vin = 3V, Vout = 5V @ 750mA
Part ID
Part Value
2.1A Boost Regulator
22µF, 6.3V, X5R
22µF, 6.3V, X5R
1000pF
Manufacturer
NSC
Part Number
LM2735YMF
U1
C1 Input Cap
C2 Output Cap
C3 Comp Cap
D1, Catch Diode
L1
TDK
C2012X5R0J226M
C2012X5R0J226M
C1608X5R1H102K
STPS120M
TDK
TDK
0.4Vf Schottky 1A, 20VR
22µH 1.2A
ST
Coilcraft
Vishay
Vishay
Vishay
MSS5131-223ML
CRCW08051002F
CRCW08053012F
CRCW06031003F
R1
10.0kΩ, 1%
R2
30.1kΩ, 1%
R3
100kΩ, 1%
29
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LM2735X SOT23-5 Design Example 8
20215879
LM2735X (1.6MHz): Vin = 3.3V, Vout = 20V @ 100mA
Part ID
Part Value
2.1A Boost Regulator
22µF, 6.3V, X5R
4.7µF, 25V, X5R
470pF
Manufacturer
NSC
Part Number
LM2735XMF
U1
C1, Input Cap
C2, Output Cap
C3 Comp Cap
D1, Catch Diode
L1
TDK
C2012X5R0J226M
C3216X5R1E475K
C1608X5R1H471K
MBR0530
TDK
TDK
0.4Vf Schottky 500mA, 30VR
10µH 1.2A
Vishay
Coilcraft
Vishay
Vishay
Vishay
DO1608C-103ML
CRCW06031002F
CRCW06031503F
CRCW06031003F
R1
10.0kΩ, 1%
R2
150kΩ, 1%
R3
100kΩ, 1%
www.national.com
30
LM2735Y SOT23-5 Design Example 9
20215879
LM2735Y (520kHz): Vin = 3.3V, Vout = 20V @ 100mA
Part ID
Part Value
2.1A Boost Regulator
22µF, 6.3V, X5R
10µF, 25V, X5R
470pF
Manufacturer
NSC
Part Number
LM2735YMF
U1
C1 Input Cap
C2 Output Cap
C3 Comp Cap
D1, Catch Diode
L1
TDK
C2012X5R0J226M
C3216X5R1E106M
C1608X5R1H471K
MBR0530
TDK
TDK
0.4Vf Schottky 500mA, 30VR
33µH 1.5A
Vishay
Coilcraft
Vishay
Vishay
Vishay
DS3316P-333ML
CRCW06031002F
CRCW06031503F
CRCW06031003F
R1
10.0kΩ, 1%
R2
150.0kΩ, 1%
R3
100kΩ, 1%
31
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LM2735X LLP-6 Design Example 10
20215876
LM2735X (1.6MHz): Vin = 3.3V, Vout = 20V @ 150mA
Part ID
U1
Part Value
2.1A Boost Regulator
22µF, 6.3V, X5R
22µF, 6.3V, X5R
10µF, 25V, X5R
No Load
Manufacturer
NSC
Part Number
LM2735XSD
C1 Input Cap
C2 Input Cap
C3 Output Cap
C4 Output Cap
C5 Comp Cap
D1, Catch Diode
L1
TDK
C2012X5R0J226M
C2012X5R0J226M
C3216X5R1E106M
TDK
TDK
470pF
TDK
Vishay
Coilcraft
Vishay
Vishay
Vishay
C1608X5R1H471K
MBR0530
0.4Vf Schottky 500mA, 30VR
8.2µH 2A
DO1813H-822ML
CRCW06031002F
CRCW06031503F
CRCW06031003F
R1
10.0kΩ, 1%
R2
150kΩ, 1%
R3
100kΩ, 1%
www.national.com
32
LM2735Y LLP-6 Design Example 11
20215876
LM2735Y (520kHz): Vin = 3.3V, Vout = 20V @ 150mA
Part ID
U1
Part Value
2.1A Boost Regulator
10µF, 6.3V, X5R
10µF, 6.3V, X5R
10µF, 25V, X5R
No Load
Manufacturer
NSC
Part Number
LM2735YSD
C1 Input Cap
C2 Input Cap
C3 Output Cap
C4 Output Cap
C5 Comp Cap
D1, Catch Diode
L1
TDK
C2012X5R0J106K
C2012X5R0J106K
C3216X5R1E106M
TDK
TDK
470pF
TDK
Vishay
Coilcraft
Vishay
Vishay
Vishay
C1608X5R1H471K
MBR0530
0.4Vf Schottky 500mA, 30VR
22µH 1.5A
DS3316P-223ML
CRCW06031002F
CRCW06031503F
CRCW06031003F
R1
10.0kΩ, 1%
R2
150kΩ, 1%
R3
100kΩ, 1%
33
www.national.com
LM2735X LLP-6 SEPIC Design Example 12
20215880
LM2735X (1.6MHz): Vin = 2.7V - 5V, Vout = 3.3V @ 500mA
Part ID
Part Value
2.1A Boost Regulator
22µF, 6.3V, X5R
No Load
Manufacturer
NSC
Part Number
LM2735XSD
U1
C1 Input Cap
TDK
C2012X5R0J226M
C2 Input Cap
C3 Output Cap
10µF, 25V, X5R
No Load
TDK
C3216X5R1E106M
C4 Output Cap
C5 Comp Cap
2200pF
TDK
TDK
C1608X5R1H222K
C2012X5R1C225K
STPS120M
C6
2.2µF 16V
D1, Catch Diode
0.4Vf Schottky 1A, 20VR
6.8µH
ST
L1
L2
Coilcraft
Coilcraft
Vishay
Vishay
Vishay
DO1608C-682ML
DO1608C-682ML
CRCW06031002F
CRCW06031652F
CRCW06031003F
6.8µH
R1
R2
R3
10.2kΩ, 1%
16.5kΩ, 1%
100kΩ, 1%
www.national.com
34
LM2735Y eMSOP-8 SEPIC Design Example 13
20215881
LM2735Y (520kHz): Vin = 2.7V - 5V, Vout = 3.3V @ 500mA
Part ID
Part Value
2.1A Boost Regulator
22µF, 6.3V, X5R
No Load
Manufacturer
NSC
Part Number
U1
LM2735YMY
C1 Input Cap
TDK
C2012X5R0J226M
C2 Input Cap
C3 Output Cap
10µF, 25V, X5R
No Load
TDK
C3216X5R1E106M
C4 Output Cap
C5 Comp Cap
2200pF
TDK
TDK
C1608X5R1H222K
C2012X5R1C225K
STPS120M
C6
2.2µF 16V
D1, Catch Diode
0.4Vf Schottky 1A, 20VR
15µH 1.5A
ST
L1
L2
Coilcraft
Coilcraft
Vishay
Vishay
Vishay
MSS5131-153ML
MSS5131-153ML
CRCW06031002F
CRCW06031652F
CRCW06031003F
15µH 1.5A
R1
R2
R3
10.2kΩ, 1%
16.5kΩ, 1%
100kΩ, 1%
35
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LM2735X SOT23-5 LED Design Example 14
20215882
LM2735X (1.6MHz): Vin = 2.7V - 5V, Vout = 20V @ 50mA
Part ID
Part Value
2.1A Boost Regulator
22µF, 6.3V, X5R
4.7µF, 25V, X5R
0.4Vf Schottky 500mA, 30VR
15µH 1.5A
Manufacturer
NSC
Part Number
LM2735XMF
U1
C1 Input Cap
TDK
C2012X5R0J226M
C3216JB1E475K
MBR0530
C2 Output Cap
TDK
D1, Catch Diode
Vishay
Coilcraft
Vishay
Vishay
Vishay
L1
R1
R2
R3
MSS5131-153ML
CRCW080525R5F
CRCW08051000F
CRCW06031003F
25.5Ω, 1%
100Ω, 1%
100kΩ, 1%
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36
LM2735Y LLP-6 FlyBack Design Example 15
20215883
LM2735Y (520kHz): Vin = 5V, Vout = ±12V 150mA
Part ID
Part Value
2.1A Boost Regulator
22µF, 6.3V, X5R
10µF, 25V, X5R
Manufacturer
NSC
Part Number
U1
C1 Input Cap
C2 Output Cap
C3 Output Cap
Cf Comp Cap
D1, D2 Catch Diode
T1
LM2735YSD
C2012X5R0J226M
C3216X5R1E106M
C3216X5R1E106M
C1608X5R1H331K
MBR0530
TDK
TDK
10µF, 25V, X5R
TDK
330pF
TDK
0.4Vf Schottky 500mA, 30VR
Vishay
R1
Vishay
Vishay
Vishay
CRCW06031002F
CRCW06038662F
CRCW06031003F
10.0kΩ, 1%
86.6kΩ, 1%
100kΩ, 1%
R2
R3
37
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LM2735X SOT23-5 LED Design Example 16
VRAIL > 5.5V Application
202158a3
LM2735X (1.6MHz): VPWR = 9V, Vout = 12V @ 500mA
Part ID
Part Value
2.1A Boost Regulator
10µF, 6.3V, X5R
10µF, 25V, X5R
0.1µF, 6.3V, X5R
1000pF
Manufacturer
NSC
Part Number
LM2735XMF
U1
C1, Input Cap
TDK
C2012X5R0J106K
C3216X5R1E106M
C2012X5R0J104K
C1608X5R1H102K
STPS120M
C2, Output Cap
TDK
C3 VIN Cap
TDK
C4 Comp Cap
TDK
D1, Catch Diode
0.4Vf Schottky 1A, 20VR
3.3V Zener, SOT23
6.8µH 2A
ST
D2
L1
Diodes Inc
Coilcraft
Vishay
Vishay
Vishay
Vishay
BZX84C3V3
DO1813H-682ML
CRCW08051002F
CRCW08058662F
CRCW06031003F
CRCW06034991F
R1
R2
R3
R4
10.0kΩ, 1%
86.6kΩ, 1%
100kΩ, 1%
499Ω, 1%
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38
LM2735X SOT23-5 LED Design Example 17
Two Input Voltage Rail Application
202158a4
LM2735X (1.6MHz): VPWR = 9V in = 2.7V - 5.5V, Vout = 12V @ 500mA
Part ID
U1
Part Value
2.1A Boost Regulator
10µF, 6.3V, X5R
10µF, 25V, X5R
0.1µF, 6.3V, X5R
1000pF
Manufacturer
NSC
Part Number
LM2735XMF
C1, Input Cap
C2, Output Cap
C3 VIN Cap
C4 Comp Cap
D1, Catch Diode
L1
TDK
C2012X5R0J106K
C3216X5R1E106M
C2012X5R0J104K
C1608X5R1H102K
STPS120M
TDK
TDK
TDK
0.4Vf Schottky 1A, 20VR
6.8µH 2A
ST
Coilcraft
Vishay
Vishay
Vishay
DO1813H-682ML
CRCW08051002F
CRCW08058662F
CRCW06031003F
R1
10.0kΩ, 1%
R2
86.6kΩ, 1%
R3
100kΩ, 1%
39
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Physical Dimensions inches (millimeters) unless otherwise noted
6-Lead LLP Package
NS Package Number SDE06A
5-Lead SOT23-5 Package
NS Package Number MF05A
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40
8-Lead eMSOP Package
NS Package Number MUY08A
41
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