LM3488MMX_技术文档

May 2003

LM3488

High Efficiency Low-Side N-Channel Controller for

Switching Regulators

n

1.5% (over temperature) internal reference

General Description

n 5µA shutdown current (over temperature)

The LM3488 is a versatile Low-Side N-FET high perfor-

mance controller for switching regulators. It is suitable for

use in topologies requiring low side FET, such as boost,

flyback, SEPIC, etc. Moreover, the LM3488 can be operated

at extremely high switching frequency in order to reduce the

overall solution size. The switching frequency of LM3488 can

be adjusted to any value between 100kHz and 1MHz by

using a single external resistor or by synchronizing it to an

external clock. Current mode control provides superior band-

width and transient response, besides cycle-by-cycle current

limiting. Output current can be programmed with a single

external resistor.

Features

n 8-lead Mini-SO8 (MSOP-8) package

n Internal push-pull driver with 1A peak current capability

n Current limit and thermal shutdown

n Frequency compensation optimized with a capacitor and

a resistor

n Internal softstart

n Current Mode Operation

n Undervoltage Lockout with hysteresis

The LM3488 has built in features such as thermal shutdown,

short-circuit protection and over voltage protection. Power

saving shutdown mode reduces the total supply current to

5µA and allows power supply sequencing. Internal soft-start

limits the inrush current at start-up.

Applications

n Distributed Power Systems

n Notebook, PDA, Digital Camera, and other Portable

Applications

n Offline Power Supplies

n Set-Top Boxes

Key Specifications

n Wide supply voltage range of 2.97V to 40V

n 100kHz to 1MHz Adjustable and Synchronizable clock

frequency

Typical Application Circuit

10138844

Typical SEPIC Converter

DS101388

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Connection Diagram

10138802

8 Lead Mini SO8 Package (MSOP-8 Package)

Package Marking and Ordering Information

Order Number

Package Type

Package Marking

Supplied As:

LM3488MM

MSOP-8

S21B

1000 units on Tape and Reel

3500 units on Tape and Reel

LM3488MMX

MSOP-8

S21B

Pin Description

Pin Name

Pin Number

Description

I_SEN

1

Current sense input pin. Voltage generated across an external

sense resistor is fed into this pin.

COMP

FB

2

3

Compensation pin. A resistor, capacitor combination connected to

this pin provides compensation for the control loop.

Feedback pin. The output voltage should be adjusted using a

resistor divider to provide 1.26V at this pin.

Analog ground pin.

AGND

PGND

DR

4

5

6

Power ground pin.

Drive pin of the IC. The gate of the external MOSFET should be

connected to this pin.

FA/SYNC/SD

7

Frequency adjust, synchronization, and Shutdown pin. A resistor

connected to this pin sets the oscillator frequency. An external

clock signal at this pin will synchronize the controller to the

frequency of the clock. A high level on this pin for ≥ 30µs will turn

the device off. The device will then draw less than 10µA from the

supply.

V_IN

8

Power supply input pin.

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2

Absolute Maximum Ratings (Note 1)

Lead Temperature

MM Package

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Vapor Phase (60 sec.)

Infared (15 sec.)

DR Pin Voltage

I_LIMPin Voltage

215˚C

220˚C

−0.4V ≤ VDR ≤ 8V

600mV

Input Voltage

45V

< <

-0.4V V_FB7V

FB Pin Voltage

<

-0.4V

FA/SYNC/SD Pin Voltage

<

1.0A

V_FA/SYNC/SD7V

Operating Ratings (Note 1)

Supply Voltage

<

Peak Driver Output Current ( 10µs)

2.97V ≤ V_IN≤ 40V

Power Dissipation

Internally Limited

−65˚C to +150˚C

+150˚C

Junction

Storage Temperature Range

Junction Temperature

Temperature Range

Switching Frequency

−40˚C ≤ T_J≤ +125˚C

100kHz ≤ F_SW≤ 1MHz

ESD Susceptibilty

Human Body Model (Note 2)

2kV

Electrical Characteristics

Specifications in Standard type face are for T_J= 25˚C, and in bold type face apply over the full Operating Temperature

Range. Unless otherwise specified, V_IN= 12V, R_FA= 40kΩ

Symbol

V_FB

Parameter

Conditions

V_COMP= 1.4V,

Typical

Limit

Units

V

Feedback Voltage

1.26

2.97 ≤ V_IN≤ 40V

I_EAOSource/Sink

1.2507/1.24

1.2753/1.28

V(min)

V(max)

%/V

∆V_LINE

∆V_LOAD

V_UVLO

Feedback Voltage

Line Regulation

Output Voltage Load

Regulation

0.001

0.5

%/V (max)

Input Undervoltage

Lock-out

2.85

170

V

2.97

V(max)

mV

V_UV(HYS)

Input Undervoltage

Lock-out Hysteresis

130

210

mV (min)

mV (max)

kHz

F_nom

Nominal Switching

Frequency

R_FA= 40KΩ

400

370

420

kHz(min)

kHz(max)

Ω

R_{DS1 (ON)}

R_{DS2 (ON)}

V_{DR (max)}

D_max

Driver Switch On

Resistance (top)

I_DR= 0.2A, V_IN= 5V

I_DR= 0.2A

16

Driver Switch On

Resistance (bottom)

Maximum Drive

4.5

Ω

V

<

V_IN7.2V

V_IN

7.2

Voltage Swing(Note 6)

V_IN≥ 7.2V

Maximum Duty

Cycle(Note 7)

100

%

T_min(on)

Minimum On Time

325

nsec

nsec(min)

nsec(max)

mA

230

550

I_SUPPLY

I_Q

Supply Current

(switching)

(Note 9)

2.0

5

2.6

7

mA (max)

µA

Quiescent Current in

Shutdown Mode

Current Sense

Threshold Voltage

V_FA/SYNC/SD= 5V(Note

10), V_IN= 5V

µA (max)

mV

V_SENSE

V_IN= 5V

165

140/ 135

195/ 200

mV (min)

mV (max)

3

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Electrical Characteristics (Continued)

Specifications in Standard type face are for T_J= 25˚C, and in bold type face apply over the full Operating Temperature

Range. Unless otherwise specified, V_IN= 12V, R_FA= 40kΩ

Symbol

V_SC

Parameter

Conditions

V_IN= 5V

Typical

Limit

Units

mV

Short-Circuit Current

Limit Sense Voltage

325

235

395

mV (min)

mV (max)

mV

V_SL

Internal Compensation

Ramp Voltage

V_IN= 5V

92

50

52

132

mV(min)

mV(max)

V_OVP

Output Over-voltage

Protection (with

V_COMP= 1.4V

mV

mV(min)

mV(max)

mV

respect to feedback

voltage) (Note 8)

Output Over-Voltage

Protection

32/ 25

78/ 85

V_OVP(HYS)

V_COMP= 1.4V

60

800

38

20

110

mV(min)

mV(max)

µmho

Hysteresis(Note 8)

Error Ampifier

Gm

V_COMP= 1.4V

I_EAO= 100µA

(Source/Sink)

V_COMP= 1.4V

I_EAO= 100µA

(Source/Sink)

Source, V_COMP= 1.4V,

V_FB= 0V

Transconductance

600/ 365

1000/ 1265

µmho (min)

µmho (max)

V/V

A_VOL

Error Amplifier Voltage

Gain

26

44

V/V (min)

V/V (max)

µA

I_EAO

Error Amplifier Output

Current (Source/ Sink)

110

−140

2.2

80/ 50

140/ 180

µA (min)

µA (max)

µA

Sink, V_COMP= 1.4V, V_FB

= 1.4V

−100/ −85

−180/ −185

µA (min)

µA (max)

V

V_EAO

Error Amplifier Output

Voltage Swing

Upper Limit

V_FB= 0V

1.8

2.4

V(min)

V(max)

V

COMP Pin = Floating

Lower Limit

0.56

V_FB= 1.4V

0.2

1.0

V(min)

V(max)

msec

T_SS

T_r

Internal Soft-Start

Delay

V_FB= 1.2V, V_COMP

Floating

=

4

Drive Pin Rise Time

Cgs = 3000pf, V_DR= 0 to

25

ns

3V

T_f

Drive Pin Fall Time

Cgs = 3000pf, V_DR= 0 to

25

3V

VSD

Shutdown and

Output = High

1.27

0.65

V

V (max)

V

Synchronization signal

threshold (Note 5)

1.35

0.35

Output = Low

V (min)

µA

I_SD

Shutdown Pin Current

V_SD= 5V

V_SD= 0V

−1

+1

TSD

T_sh

Thermal Shutdown

Hysteresis

165

10

˚C

θ_JA

Thermal Resistance

MM Package

200

˚C/W

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4

Electrical Characteristics (Continued)

Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the device

is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics.

Note 2: The human body model is a 100 pF capacitor discharged through a 1.5kΩ resistor into each pin.

Note 3: All limits are guaranteed at room temperature (standard type face) and at temperature extremes (bold type face). All room temperature limits are 100%

tested. All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods. All limits are used to calculate

Average Outgoing Quality Level (AOQL).

Note 4: Typical numbers are at 25˚C and represent the most likely norm.

Note 5: The FA/SYNC/SD pin should be pulled to V through a resistor to turn the regulator off.

IN

Note 6: The voltage on the drive pin, V is equal to the input voltage when input voltage is less than 7.2V. V is equal to 7.2V when the input voltage is greater

DR

than or equal to 7.2V.

Note 7: The limits for the maximum duty cycle can not be specified since the part does not permit less than 100% maximum duty cycle operation.

Note 8: The over-voltage protection is specified with respect to the feedback voltage. This is because the over-voltage protection tracks the feedback voltage. The

over-voltage thresold can be calculated by adding the feedback voltage, V to the over-voltage protection specification.

FB

Note 9: For this test, the FA/SYNC/SD Pin is pulled to ground using a 40K resistor .

Note 10: For this test, the FA/SYNC/SD Pin is pulled to 5V using a 40K resistor.

5

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Typical Performance Characteristics Unless otherwise specified, V_IN= 12V, T_J= 25˚C.

I_Qvs Temperature & Input Voltage

I_Supplyvs Input Voltage (Non-Switching)

10138834

10138803

I_Supplyvs V_IN

Switching Frequency vs RFA

10138835

10138804

Frequency vs Temperature

Drive Voltage vs Input Voltage

10138854

10138805

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6

Typical Performance Characteristics Unless otherwise specified, V_IN= 12V, T_J= 25˚C. (Continued)

Current Sense Threshold vs Input Voltage

COMP Pin Voltage vs Load Current

10138862

10138845

Efficiency vs Load Current (3.3V In and 12V Out)

Efficiency vs Load Current (5V In and 12V Out)

10138859

10138858

Efficiency vs Load Current (9V In and 12V Out)

Efficiency vs Load Current (3.3V In and 5V Out)

10138860

10138853

7

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Typical Performance Characteristics Unless otherwise specified, V_IN= 12V, T_J= 25˚C. (Continued)

Error Amplifier Gain

Error Amplifier Phase

10138855

10138856

COMP Pin Source Current vs Temperature

Short Circuit Protection vs Input Voltage

10138857

10138836

Compensation Ramp vs Compensation Resistor

Shutdown Threshold Hysteresis vs Temperature

10138846

10138851

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Typical Performance Characteristics Unless otherwise specified, V_IN= 12V, T_J= 25˚C. (Continued)

Current Sense Voltage vs Duty Cycle

10138852

9

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Functional Block Diagram

10138806

The voltage sensed across the sense resistor generally

contains spurious noise spikes, as shown in Figure 1. These

spikes can force the PWM comparator to reset the RS latch

prematurely. To prevent these spikes from resetting the

latch, a blank-out circuit inside the IC prevents the PWM

comparator from resetting the latch for a short duration after

the latch is set. This duration is about 150ns and is called the

blank-out time.

Functional Description

The LM3488 uses a fixed frequency, Pulse Width Modulated

(PWM), current mode control architecture. In a typical appli-

cation circuit, the peak current through the external MOS-

FET is sensed through an external sense resistor. The volt-

age across this resistor is fed into the I_SENpin. This voltage

is then level shifted and fed into the positive input of the

PWM comparator. The output voltage is also sensed through

an external feedback resistor divider network and fed into

the error amplifier negative input (feedback pin, FB). The

output of the error amplifier (COMP pin) is added to the slope

compensation ramp and fed into the negative input of the

PWM comparator.

Under extremely light load or no-load conditions, the energy

delivered to the output capacitor when the external MOSFET

is on during the blank-out time is more than what is delivered

to the load. An over-voltage comparator inside the LM3488

prevents the output voltage from rising under these condi-

tions. The over-voltage comparator senses the feedback (FB

pin) voltage and resets the RS latch under these conditions.

The latch remains in reset state till the output decays to the

nominal value.

At the start of any switching cycle, the oscillator sets the RS

latch using the SET/Blank-out and switch logic blocks. This

forces a high signal on the DR pin (gate of the external

MOSFET) and the external MOSFET turns on. When the

voltage on the positive input of the PWM comparator ex-

ceeds the negative input, the RS latch is reset and the

external MOSFET turns off.

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Functional Description (Continued)

10138807

FIGURE 1. Basic Operation of the PWM comparator

>

SLOPE COMPENSATION RAMP

From the above equation, when D 0.5, ∆I₁will be greater

than ∆I_O. In other words, the disturbance is divergent. So a

very small perturbation in the load will cause the disturbance

to increase.

The LM3488 uses a current mode control scheme. The main

advantages of current mode control are inherent cycle-by-

cycle current limit for the switch, and simpler control loop

characteristics. It is also easy to parallel power stages using

current mode control since as current sharing is automatic.

To prevent the sub-harmonic oscillations, a compensation

ramp is added to the control signal, as shown in Figure 3.

Current mode control has an inherent instability for duty

cycles greater than 50%, as shown in Figure 2. In Figure 2,

a small increase in the load current causes the switch cur-

rent to increase by ∆I_O. The effect of this load change, ∆I₁, is

:

With the compensation ramp,

10138809

>

FIGURE 2. Sub-Harmonic Oscillation for D 0.5

11

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Functional Description (Continued)

10138811

FIGURE 3. Compensation Ramp Avoids Sub-Harmonic Oscillation

The compensation ramp has been added internally in

LM3488. The slope of this compensation ramp has been

selected to satisfy most of the applications. The slope of the

internal compensation ramp depends on the frequency. This

slope can be calculated using the formula:

In this equation, ∆V_SLis equal to 40.10^-6R_SL. Hence,

M_C= V_SL.F_SVolts/second

In the above equation, V_SLis the amplitude of the internal

compensation ramp. Limits for V_SLhave been specified in

the electrical characteristics.

In order to provide the user additional flexibility, a patented

scheme has been implemented inside the IC to increase the

slope of the compensation ramp externally, if the need

arises. Adding a single external resistor, R_SL(as shown in

Figure 4) increases the slope of the compensation ramp, M_C

by :

∆V_SLversus R_SLhas been plotted in Figure 5 for different

frequencies.

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Functional Description (Continued)

10138813

FIGURE 4. Increasing the Slope of the Compensation Ramp

10138851

FIGURE 5. ∆V_SLvs R_SL

FREQUENCY

this resistor is dependent on the off time of the synchroniza-

ADJUST/SYNCHRONIZATION/SHUTDOWN

tion pulse, T_OFF(SYNC). Table 1 shows the range of resistors

to be used for a given T_OFF(SYNC)

.

The switching frequency of LM3488 can be adjusted be-

tween 100kHz and 1MHz using a single external resistor.

This resistor must be connected between FA/SYNC/SD pin

and ground, as shown in Figure 6. Please refer to the typical

performance characteristics to determine the value of the

resistor required for a desired switching frequency.

TABLE 1.

T_OFF(SYNC)(µsec)

R_SYNCrange (kΩ)

5 to 13

1

2

3

The LM3488 can be synchronized to an external clock. The

external clock must be connected to the FA/SYNC/SD pin

through a resistor, R_SYNCas shown in Figure 7. The value of

20 to 40

40 to 65

13

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The FA/SYNC/SD pin also functions as a shutdown pin. If a

high signal (refer to the electrical characteristics for definition

of high signal) appears on the FA/SYNC/SD pin, the LM3488

stops switching and goes into a low current mode. The total

supply current of the IC reduces to less than 10µA under

these conditions.

Functional Description (Continued)

TABLE 1. (Continued)

T_OFF(SYNC)(µsec)

R_SYNCrange (kΩ)

55 to 90

4

5

70 to 110

Figure 8 and Figure 9 show implementation of shutdown

function when operating in Frequency adjust mode and syn-

chronization mode respectively. In frequency adjust mode,

connecting the FA/SYNC/SD pin to ground forces the clock

to run at a certain frequency. Pulling this pin high shuts down

the IC. In frequency adjust or synchronization mode, a high

signal for more than 30ms shuts down the IC.

6

85 to 140

7

100 to 160

120 to 190

135 to 215

150 to 240

8

9

10

It is also necessary to have the width of the synchronization

pulse narrower than the duty cycle of the converter. It is also

necessary to have the synchronization pulse width

300nsecs.

≥

10138816

FIGURE 6. Frequency Adjust

10138815

FIGURE 7. Frequency Synchronization

10138816

FIGURE 8. Shutdown Operation in Frequency Adjust Mode

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Functional Description (Continued)

10138817

FIGURE 9. Shutdown Operation in Synchronization Mode

SHORT-CIRCUIT PROTECTION

When the voltage across the sense resistor (measured on

I_SENPin) exceeds 350mV, short-circuit current limit gets

activated. A comparator inside LM3488 reduces the switch-

ing frequency by a factor of 5 and maintains this condition till

the short is removed.

Typical Applications

The LM3488 may be operated in either continuous or dis-

continuous conduction mode. The following applications are

designed for continuous conduction operation. This mode of

operation has higher efficiency and lower EMI characteristics

than the discontinuous mode.

ferred to the load and output capacitor. The ratio of these two

cycles determines the output voltage. The output voltage is

defined as:

BOOST CONVERTER

The most common topology for LM3488 is the boost or

step-up topology. The boost converter converts a low input

voltage into a higher output voltage. The basic configuration

for a boost regulator is shown in Figure 10. In continuous

conduction mode (when the inductor current never reaches

zero at steady state), the boost regulator operates in two

cycles. In the first cycle of operation, MOSFET Q is turned

on and energy is stored in the inductor. During this cycle,

diode D is reverse biased and load current is supplied by the

(ignoring the drop across the MOSFET and the diode), or

where D is the duty cycle of the switch, V_Dis the forward

voltage drop of the diode, and V_Qis the drop across the

MOSFET when it is on. The following sections describe

selection of components for a boost converter.

output capacitor, C_OUT

.

In the second cycle, MOSFET Q is off and the diode is

forward biased. The energy stored in the inductor is trans-

15

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Typical Applications (Continued)

10138822

FIGURE 10. Simplified Boost Converter Diagram (a) First cycle of operation. (b) Second cycle of operation

POWER INDUCTOR SELECTION

The inductor is one of the two energy storage elements in a

boost converter. Figure 11 shows how the inductor current

varies during a switching cycle. The current through an

inductor is quantified as:

10138824

FIGURE 11. A. Inductor current B. Diode current

If V_L(t) is constant, di_L(t)/dt must be constant. Hence, for a

given input voltage and output voltage, the current in the

inductor changes at a constant rate.

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16

PROGRAMMING THE OUTPUT VOLTAGE AND OUTPUT

CURRENT

Typical Applications (Continued)

The important quantities in determining a proper inductance

value are I_L(the average inductor current) and ∆i_L(the

inductor current ripple). If ∆i_Lis larger than I_L, the inductor

current will drop to zero for a portion of the cycle and the

converter will operate in discontinuous conduction mode. If

∆i_Lis smaller than I_L, the inductor current will stay above

zero and the converter will operate in continuous conduction

mode. All the analysis in this datasheet assumes operation

in continuous conduction mode. To operate in continuous

conduction mode, the following conditions must be met:

The output voltage can be programmed using a resistor

divider between the output and the feedback pins, as shown

in Figure 12. The resistors are selected such that the voltage

at the feedback pin is 1.26V. R_F1and R_F2can be selected

using the equation,

A 100pF capacitor may be connected between the feedback

and ground pins to reduce noise.

>

I_L∆i_L

The maximum amount of current that can be delivered at the

output can be controlled by the sense resistor, R_SEN. Current

limit occurs when the voltage that is generated across the

sense resistor equals the current sense threshold voltage,

V_SENSE. Limits for V_SENSEhave been specified in the elec-

trical characteristics. This can be expressed as:

*

I_sw(peak)R_SEN= V_SENSE

V_SENSErepresents the maximum value of the control signal

as shown in Figure 2. This control signal, however, is not a

constant value and changes over the course of a period as a

result of the internal compensation ramp (see Figure 3).

Therefore the current limit will also change as a result of the

internal compensation ramp. The actual command signal,

V_CS, can be better expressed as a function of the sense

voltage and the internal compensation ramp:

Choose the minimum I_OUTto determine the minimum L. A

common choice is to set ∆i_Lto 30% of I_L. Choosing an

appropriate core size for the inductor involves calculating the

average and peak currents expected through the inductor. In

a boost converter,

*

V_CS= V_SENSE− (D V_SL

)

V_SLis defined as the internal compensation ramp voltage,

limits are specified in the electrical characteristics.

The peak current through the switch is equal to the peak

inductor current.

and I_{L_peak}= I_L(max) + ∆i_L(max),

where

I_sw(peak)= I_L+ ∆i_L

Therefore for a boost converter

A core size with ratings higher than these values should be

chosen. If the core is not properly rated, saturation will

dramatically reduce overall efficiency.

Combining the three equation yields an expression for R_SEN

The LM3488 can be set to switch at very high frequencies.

When the switching frequency is high, the converter can be

operated with very small inductor values. With a small induc-

tor value, the peak inductor current can be extremely higher

than the output currents, especially under light load condi-

tions.

The LM3488 senses the peak current through the switch.

The peak current through the switch is the same as the peak

current calculated above.

17

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Typical Applications (Continued)

10138820

FIGURE 12. Adjusting the Output Voltage

CURRENT LIMIT WITH ADDITIONAL SLOPE

COMPENSATION

In the above equation, I_OUTis the output current and ∆I_Lhas

been defined in Figure 11

If an external slope compensation resistor is used (see

Figure 4) the internal control signal will be modified and this

will have an effect on the current limit. The control signal is

given by:

The peak reverse voltage for boost converter is equal to the

regulator output voltage. The diode must be capable of

handling this voltage. To improve efficiency, a low forward

drop schottky diode is recommended.

*

V_CS= V_SENSE− (D V_SL

)

POWER MOSFET SELECTION

Where V_SENSEand V_SLare defined parameters in the elec-

trical characteristics section. If R_SLis used, then this will add

to the existing slope compensation. The command voltage

will then be given by:

The drive pin of LM3488 must be connected to the gate of an

external MOSFET. In a boost topology, the drain of the

external N-Channel MOSFET is connected to the inductor

and the source is connected to the ground. The drive pin

(DR) voltage depends on the input voltage (see typical per-

formance characteristics). In most applications, a logic level

MOSFET can be used. For very low input voltages, a sub-

logic level MOSFET should be used.

*

V_CS= V_SENSE− (D ( V_SL+ ∆V_SL) )

Where ∆V_SLis the additional slope compensation generated

and can be calculated by use of Figure 5 or is equal to 40 x

10⁻⁶* R_SL. This changes the equation for R_SENto:

The selected MOSFET directly controls the efficiency. The

critical parameters for selection of a MOSFET are:

1. Minimum threshold voltage, V_TH(MIN)

2. On-resistance, R_DS(ON)

3. Total gate charge, Q_g

4. Reverse transfer capacitance, C_RSS

5. Maximum drain to source voltage, V_DS(MAX)

Therefore R_SLcan be used to provide an additional method

for setting the current limit.

The off-state voltage of the MOSFET is approximately equal

to the output voltage. V_DS(MAX)of the MOSFET must be

greater than the output voltage. The power losses in the

MOSFET can be categorized into conduction losses and ac

switching or transition losses. R_DS(ON)is needed to estimate

the conduction losses. The conduction loss, P_COND, is the

I²R loss across the MOSFET. The maximum conduction loss

is given by:

POWER DIODE SELECTION

Observation of the boost converter circuit shows that the

average current through the diode is the average load cur-

rent, and the peak current through the diode is the peak

current through the inductor. The diode should be rated to

handle more than its peak current. The peak diode current

can be calculated using the formula:

I_D(Peak)= I_OUT/ (1−D) + ∆I_L

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Typical Applications (Continued)

OUTPUT CAPACITOR SELECTION

The output capacitor in a boost converter provides all the

output current when the inductor is charging. As a result it

sees very large ripple currents. The output capacitor should

be capable of handling the maximum rms current. The rms

current in the output capacitor is:

where D_MAXis the maximum duty cycle.

The turn-on and turn-off transitions of a MOSFET require

times of tens of nano-seconds. C_RSSand Q_gare needed to

estimate the large instantaneous power loss that occurs

during these transitions.

Where

The amount of gate current required to turn the MOSFET on

can be calculated using the formula:

and D, the duty cycle is equal to (V_OUT− V_IN)/V_OUT

.

I_G= Q_g.F_S

The ESR and ESL of the output capacitor directly control the

output ripple. Use capacitors with low ESR and ESL at the

output for high efficiency and low ripple voltage. Surface

Mount tantalums, surface mount polymer electrolytic and

polymer tantalum, Sanyo- OSCON, or multi-layer ceramic

capacitors are recommended at the output.

The required gate drive power to turn the MOSFET on is

equal to the switching frequency times the energy required

to deliver the charge to bring the gate charge voltage to V_DR

(see electrical characteristics and typical performance char-

acteristics for the drive voltage specification).

P_Drive= F_S.Q_g.V_DR

Designing SEPIC Using LM3488

INPUT CAPACITOR SELECTION

Since the LM3488 controls a low-side N-Channel MOSFET,

it can also be used in SEPIC (Single Ended Primary Induc-

tance Converter) applications. An example of SEPIC using

LM3488 is shown in Figure 14. As shown in Figure 14, the

output voltage can be higher or lower than the input voltage.

The SEPIC uses two inductors to step-up or step-down the

input voltage. The inductors L1 and L2 can be two discrete

inductors or two windings of a coupled transformer since

equal voltages are applied across the inductor throughout

the switching cycle. Using two discrete inductors allows use

of catalog magnetics, as opposed to a custom transformer.

The input ripple can be reduced along with size by using the

coupled windings of transformer for L1 and L2.

Due to the presence of an inductor at the input of a boost

converter, the input current waveform is continuous and

triangular, as shown in Figure 11. The inductor ensures that

the input capacitor sees fairly low ripple currents. However,

as the input capacitor gets smaller, the input ripple goes up.

The rms current in the input capacitor is given by:

The input capacitor should be capable of handling the rms

current. Although the input capacitor is not as critical in a

boost application, low values can cause impedance interac-

tions. Therefore a good quality capacitor should be chosen

in the range of 100µF to 200µF. If a value lower than 100µF

is used, then problems with impedance interactions or

switching noise can affect the LM3478. To improve perfor-

mance, especially with V_INbelow 8 volts, it is recommended

to use a 20Ω resistor at the input to provide a RC filter. The

resistor is placed in series with the V_INpin with only a bypass

capacitor attached to the V_INpin directly (see Figure 13). A

0.1µF or 1µF ceramic capacitor is necessary in this configu-

ration. The bulk input capacitor and inductor will connect on

the other side of the resistor with the input power supply.

Due to the presence of the inductor L1 at the input, the

SEPIC inherits all the benefits of a boost converter. One

main advantage of SEPIC over boost converter is the inher-

ent input to output isolation. The capacitor CS isolates the

input from the output and provides protection against

shorted or malfunctioning load. Hence, the A SEPIC is useful

for replacing boost circuits when true shutdown is required.

This means that the output voltage falls to 0V when the

switch is turned off. In a boost converter, the output can only

fall to the input voltage minus a diode drop.

10138893

FIGURE 13. Reducing IC Input Noise

19

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Designing SEPIC Using LM3488

(Continued)

The duty cycle of a SEPIC is given by:

In the above equation, V_Qis the on-state voltage of the

MOSFET, Q, and V_DIODEis the forward voltage drop of the

diode.

10138844

FIGURE 14. Typical SEPIC Converter

SELECTION OF INDUCTORS L1 AND L2

POWER MOSFET SELECTION

As in boost converter, the parameters governing the selec-

tion of the MOSFET are the minimum threshold voltage,

V_TH(MIN), the on-resistance, R_DS(ON), the total gate charge,

Q_g, the reverse transfer capacitance, C_RSS, and the maxi-

mum drain to source voltage, V_DS(MAX). The peak switch

voltage in a SEPIC is given by:

Proper selection of the inductors L1 and L2 to maintain

constant current mode requires calculations of the following

parameters.

Average current in the inductors:

V_SW(PEAK)= V_IN+ V_OUT+ V_DIODE

The selected MOSFET should satisfy the condition:

>

V_DS(MAX)V_SW(PEAK)

I_L2AVE= I_OUT

The peak switch current is given by:

Peak to peak ripple current, to calculate core loss if neces-

sary:

The rms current through the switch is given by:

>

maintains the condition I_L

mode.

∆i_Lto ensure constant current

POWER DIODE SELECTION

The Power diode must be selected to handle the peak

current and the peak reverse voltage. In a SEPIC, the diode

peak current is the same as the switch peak current. The

off-state voltage or peak reverse voltage of the diode is V_IN

+ V_OUT. Similar to the boost converter, the average diode

current is equal to the output current. Schottky diodes are

recommended.

www.national.com

20

having high rms current ratings relative to size. Ceramic

capacitors could be used, but the low C values will tend to

cause larger changes in voltage across the capacitor due to

the large currents. High C value ceramics are expensive.

Electrolytics work well for through hole applications where

the size required to meet the rms current rating can be

accommodated. There is an energy balance between CS

and L1, which can be used to determine the value of the

capacitor. The basic energy balance equation is:

Designing SEPIC Using LM3488

(Continued)

Peak current in the inductor, to ensure the inductor does not

saturate:

Where

I_L1PKmust be lower than the maximum current rating set by

the current sense resistor.

is the ripple voltage across the SEPIC capacitor, and

The value of L1 can be increased above the minimum rec-

ommended to reduce input ripple and output ripple. How-

ever, once D_IL1is less than 20% of I_L1AVE, the benefit to

output ripple is minimal.

By increasing the value of L2 above the minimum recom-

mended, ∆_IL2can be reduced, which in turn will reduce the

output ripple voltage:

is the ripple current through the inductor L1. The energy

balance equation can be solved to provide a minimum value

for C_S:

where ESR is the effective series resistance of the output

capacitor.

Input Capacitor Selection

If L1 and L2 are wound on the same core, then L1 = L2 = L.

All the equations above will hold true if the inductance is

replaced by 2L. A good choice for transformer with equal

turns is Coiltronics CTX series Octopack.

Similar to a boost converter, the SEPIC has an inductor at

the input. Hence, the input current waveform is continuous

and triangular. The inductor ensures that the input capacitor

sees fairly low ripple currents. However, as the input capaci-

tor gets smaller, the input ripple goes up. The rms current in

the input capacitor is given by:

SENSE RESISTOR SELECTION

The peak current through the switch, I_SW(PEAK)can be ad-

justed using the current sense resistor, R_SEN, to provide a

certain output current. Resistor R_SENcan be selected using

the formula:

The input capacitor should be capable of handling the rms

current. Although the input capacitor is not as critical in a

boost application, low values can cause impedance interac-

tions. Therefore a good quality capacitor should be chosen

in the range of 100µF to 200µF. If a value lower than 100µF

is used, then problems with impedance interactions or

switching noise can affect the LM3478. To improve perfor-

mance, especially with V_INbelow 8 volts, it is recommended

to use a 20Ω resistor at the input to provide a RC filter. The

resistor is placed in series with the V_INpin with only a bypass

capacitor attached to the V_INpin directly (see Figure 13). A

0.1µF or 1µF ceramic capacitor is necessary in this configu-

ration. The bulk input capacitor and inductor will connect on

the other side of the resistor with the input power supply.

Sepic Capacitor Selection

The selection of SEPIC capacitor, CS, depends on the rms

current. The rms current of the SEPIC capacitor is given by:

The SEPIC capacitor must be rated for a large ACrms cur-

rent relative to the output power. This property makes the

SEPIC much better suited to lower power applications where

the rms current through the capacitor is relatively small

(relative to capacitor technology). The voltage rating of the

SEPIC capacitor must be greater than the maximum input

voltage. Tantalum capacitors are the best choice for SMT,

Output Capacitor Selection

The ESR and ESL of the output capacitor directly control the

output ripple. Use low capacitors with low ESR and ESL at

21

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mount tantalums, surface mount polymer electrolytic and

polymer tantalum, Sanyo- OSCON, or multi-layer ceramic

capacitors are recommended at the output for low ripple.

Output Capacitor Selection

(Continued)

the output for high efficiency and low ripple voltage. Surface

mount tantalums, surface mount polymer electrolytic and

polymer tantalum, Sanyo- OSCON, or multi-layer ceramic

capacitors are recommended at the output.

The output capacitor of the SEPIC sees very large ripple

currents (similar to the output capacitor of a boost converter.

The rms current through the output capacitor is given by:

The ESR and ESL of the output capacitor directly control the

output ripple. Use low capacitors with low ESR and ESL at

the output for high efficiency and low ripple voltage. Surface

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22

Other Application Circuits

10138843

FIGURE 15. Typical High Efficiency Step-Up (Boost) Converter

23

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Physical Dimensions inches (millimeters)

unless otherwise noted

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DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL

COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:

1. Life support devices or systems are devices or

systems which, (a) are intended for surgical implant

into the body, or (b) support or sustain life, and

whose failure to perform when properly used in

accordance with instructions for use provided in the

labeling, can be reasonably expected to result in a

significant injury to the user.

2. A critical component is any component of a life

support device or system whose failure to perform

can be reasonably expected to cause the failure of

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型号：	LM3488MMX
厂家：	National Semiconductor
描述：	High Efficiency Low-Side N-Channel Controller for Switching Regulators 高效低侧N通道控制器开关稳压器稳压器开关控制器
文件：	总24页 (文件大小：620K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

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