LM4960SQX [NSC]
IC 1 CHANNEL, AUDIO AMPLIFIER, PQCC28, PLASTIC, LLP-28, Audio/Video Amplifier;
| 型号： | LM4960SQX |
| 厂家： | 
National Semiconductor |
| 描述： | IC 1 CHANNEL, AUDIO AMPLIFIER, PQCC28, PLASTIC, LLP-28, Audio/Video Amplifier 驱动器 |
| 文件： | 总12页 (文件大小：346K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
October 2004
LM4960
Piezoelectric Speaker Driver
General Description
The LM4960 utilizes a switching regulator to drive a dual
audio power amplifier. It delivers 24VP-P mono-BTL to a
ceramic speaker with less than 1.0% THD+N while operating
on a 3.0V power supply.
Key Specifications
@
n VOUT VDD = 3.0 THD+N ≤ 1%
24VP-P (typ)
3.0 to 7V
1.6MHz (typ)
n Power supply range
n Switching Frequency
The LM4960’s switching regulator is a current-mode boost
converter operating at a fixed frequency of 1.6MHz.
Features
n Stereo BTL amplifier
Boomer audio power amplifiers were designed specifically to
provide high quality output power with a minimal amount of
external components. The LM4960 does not require output
coupling capacitors or bootstrap capacitors, and therefore is
ideally suited for mobile phone and other low voltage appli-
cations where minimal power consumption is a primary re-
quirement.
n Low current shutdown mode
n "Click and pop" suppression circuitry
n Low Quiescent current
n Unity-gain stable audio amplifiers
n External gain configuration capability
n Thermal shutdown protection circuitry
n Wide input voltage range (3.0V - 7V)
n 1.6MHz switching frequency
The LM4960 features a low-power consumption externally
controlled micropower shutdown mode. Additionally, the
LM4960 features and internal thermal shutdown protection
mechanism along with a short circuit protection.
Applications
n Mobile phone
n PDA’s
The LM4960 is unity-gain stable and can be configured by
external gain-setting resistors.
Connection Diagram
LM4960SQ
20076582
Top View
Order Number LM4960SQ
See NS Package Number
Boomer® is a registered trademark of National Semiconductor Corporation.
© 2004 National Semiconductor Corporation
DS200765
www.national.com
Typical Application
20076581
FIGURE 1. Typical Audio Amplifier Application Circuit
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2
Absolute Maximum Ratings (Notes 1, 2)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Junction Temperature
Thermal Resistance
θJA (LLP)
150˚C
˚C/W
See AN-1187 ’Leadless Leadframe Packaging (LLP).’
Supply Voltage (VDD
)
8.5V
Supply Voltage (V1)
Operating Ratings
Temperature Range
(Pin 27 referred to GND)
Storage Temperature
18V
−65˚C to +150˚C
−0.3V to VDD + 0.3V
Internally limited
2000V
TMIN ≤ TA ≤ TMAX
Supply Voltage (VDD
Supply Voltage (V1)
−40˚C ≤ TA ≤ +85˚C
3.0V ≤ VDD ≤ 7V
9.6V ≤ V1 ≤ 16V
Input Voltage
)
Power Dissipation (Note 3)
ESD Susceptibility (Note 4)
ESD Susceptibility (Note 5)
200V
Electrical Characteristics VDD = 3.0V (Notes 1, 2)
The following specifications apply for VDD = 3V, AV = 10, RL = 800nF+20Ω, V1 = 12V unless otherwise specified. Limits apply
for TA = 25˚C.
Symbol
Parameter
Conditions
LM4960
Typical Limit
(Note 6) (Notes 7, 8)
Units
(Limits)
IDD
ISD
Quiescent Power Supply Current
Shutdown Current
VIN = GND, No Load
85
30
5
150
100
40
mA (max)
µA (max)
mV (max)
V (max)
V (min)
ms
VSHUTDOWN = GND (Note 9)
VOS
Output Offset Voltage
VSDIH
VSDIL
TWU
Shutdown Voltage Input High
Shutdown Voltage Input Low
Wake-up Time
2
0.4
CB = 0.22µF
50
150
190
20
˚C (min)
˚C (max)
VP-P (min)
%
TSD
Thermal Shutdown Temperature
170
VO
Output Voltage
THD = 1% (max); f = 1kHz
24
0.04
90
THD+N
eOS
Total Harmomic Distortion + Noise VO = 3Wrms; f = 1kHz
Output Noise
A-Weighted Filter, VIN = 0V
VRIPPLE = 200mVp-p, f = 1kHz
µV
PSRR
VFB
Power Supply Rejection Ratio
Feedback Pin Reference Voltage
55
50
dB (min)
V (max)
1.23
Electrical Characteristics VDD = 5.0V (Notes 1, 2)
The following specifications apply for VDD = 5V, AV = 10, RL = 800nF+20Ω unless otherwise specified. Limits apply for TA
=
25˚C.
Symbol
Parameter
Conditions
LM4960
Typical Limit
Units
(Limits)
(Note 6) (Notes 7, 8)
45
IDD
ISD
Quiescent Power Supply Current
Shutdown Current
VIN = GND, No Load
mA (max)
µA (max)
V (max)
V (min)
s
VSHUTDOWN = GND (Note 9)
55
100
2
VSDIH
VSDIL
TWU
Shutdown Voltage Input High
Shutdown Voltage Input Low
Wake-up Time
0.4
CB = 0.22µF
50
150
190
˚C (min)
˚C (max)
TSD
VO
Thermal Shutdown Temperature
Output Voltage
170
THD = 1% (max); f = 1kHz
RL = Ceramic Speaker
24
20
VP-P (min)
THD+N
eOS
Total Harmomic Distortion + Noise VO = 3Wrms; f = 1kHz
0.04
90
%
Output Noise
A-Weighted Filter, VIN = 0V
VRIPPLE = 200mVp-p, f = 1kHz
µV
PSRR
VFB
Power Supply Rejection Ratio
Feedback Pin Reference Voltage
60
dB (min)
V (max)
1.23
Note 1: All voltages are measured with respect to the GND pin, unless otherwise specified.
3
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Electrical Characteristics VDD = 5.0V (Notes 1, 2) (Continued)
Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions which
guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit
is given, however, the typical value is a good indication of device performance.
Note 3: The maximum power dissipation must be derated at elevated temperatures and is dictated by T
, θ , and the ambient temperature, T . The maximum
A
JMAX JA
allowable power dissipation is P
= (T
− T ) / θ or the given in Absolute Maximum Ratings, whichever is lower. For the LM4960 typical application (shown
DMAX
JMAX A JA
in Figure 1) with V
= 12V, R = 4Ω stereo operation the total power dissipation is 3.65W. θ = 35˚C/W.
DD
L
J
A
Note 4: Human body model, 100pF discharged through a 1.5kΩ resistor.
Note 5: Machine Model, 220pF–240pF discharged through all pins.
Note 6: Typicals are measured at 25˚C and represent the parametric norm.
Note 7: Limits are guaranteed to National’s AOQL (Average Outgoing Quality Level).
Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis.
Note 9: Shutdown current is measured in a normal room environment. The Shutdown pin should be driven as close as possible to V
for minimum shutdown
DD
current.
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Typical Performance Characteristics
THD+N vs Frequency
THD+N vs Frequency
VDD = 3V, V1 = 9.6V, V0 = 3Vrms
VDD = 3V, V1 = 12V, V0 = 3Vrms
20076514
20076515
THD+N vs Frequency
THD+N vs Frequency
VDD = 3V, V1 = 15V, V0 = 3Vrms
VDD = 5V, V1 = 9.6V, V0 = 3Vrms
20076516
20076517
THD+N vs Frequency
THD+N vs Frequency
VDD = 5V, V1 =12V, V0 = 3Vrms
VDD = 5V, V1 =15V, V0 = 3Vrms
20076518
20076519
5
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Typical Performance Characteristics (Continued)
THD+N vs Output Power
VDD = 3V, V1 = 9.6V,
THD+N vs Output Power
VDD = 3V, V1 = 12V,
f = 100Hz, 1kHz, 10kHz
f = 100Hz, 1kHz, 10kHz
20076520
20076521
20076523
20076525
THD+N vs Output Power
VDD = 3V, V1 = 15V,
f = 100Hz, 1kHz, 10kHz
THD+N vs Output Power
VDD = 5V, V1 = 9.6V,
f = 100Hz, 1kHz, 10kHz
20076522
THD+N vs Output Power
VDD = 5V, V1 = 12V,
f = 100Hz, 1kHz, 10kHz
THD+N vs Output Power
VDD = 5V, V1 = 15V,
f = 100Hz, 1kHz, 10kHz
20076524
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Typical Performance Characteristics (Continued)
Power Dissipation vs Output Voltage
VDD = 3V, from top to bottom:
V1 = 15V, V1 = 12V, V1 = 9.6V
Power Dissipation vs Output Voltage
VDD = 5V, from top to bottom:
V1 = 15V, V1 = 12V, V1 = 9.6V
20076509
20076510
Supply Current vs Supply Voltage
from top to bottom:
VDD = 15V, VDD = 12V, VDD = 9.6V
Power Supply Rejection Ratio
VDD = 3V
20076513
20076511
Power Supply Rejection Ratio
VDD = 5V
20076512
7
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BOOST CONVERTER POWER DISSIPATION
Application Information
At higher duty cycles, the increased ON-time of the switch
FET means the maximum output current will be determined
by power dissipation within the LM2731 FET switch. The
switch power dissipation from ON-time conduction is calcu-
lated by Equation 2.
BRIDGE CONFIGURATION EXPLANATION
The Audio Amplifier portion of the LM4960 has two internal
amplifiers allowing different amplifier configurations. The first
amplifier’s gain is externally configurable, whereas the sec-
ond amplifier is internally fixed in a unity-gain, inverting
configuration. The closed-loop gain of the first amplifier is set
by selecting the ratio of Rf to Ri while the second amplifier’s
gain is fixed by the two internal 20kΩ resistors. Figure 1
shows that the output of amplifier one serves as the input to
amplifier two. This results in both amplifiers producing sig-
nals identical in magnitude, but out of phase by 180˚. Con-
sequently, the differential gain for the Audio Amplifier is
PDMAX(SWITCH) = DC x IIND(AVE)2 x RDS(ON)
(2)
where DC is the duty cycle.
There will be some switching losses as well, so some derat-
ing needs to be applied when calculating IC power dissipa-
tion.
TOTAL POWER DISSIPATION
AVD = 2 *(Rf/Ri)
The total power dissipation for the LM4960 can be calculated
by adding Equation 1 and Equation 2 together to establish
Equation 3:
By driving the load differentially through outputs Vo1 and
Vo2, an amplifier configuration commonly referred to as
“bridged mode” is established. Bridged mode operation is
different from the classic single-ended amplifier configura-
tion where one side of the load is connected to ground.
PDMAX(TOTAL) = [4*(VDD)2/2π2ZL] + [DC x IIND(AVE)2 xRD
(ON)] (3)
-
S
The result from Equation 3 must not be greater than the
power dissipation that results from Equation 4:
A bridge amplifier design has a few distinct advantages over
the single-ended configuration. It provides differential drive
to the load, thus doubling the output swing for a specified
supply voltage. Four times the output power is possible as
compared to a single-ended amplifier under the same con-
ditions. This increase in attainable output power assumes
that the amplifier is not current limited or clipped. In order to
choose an amplifier’s closed-loop gain without causing ex-
cessive clipping, please refer to the Audio Power Amplifier
Design section.
PDMAX = (TJMAX - TA) / θJA
(4)
For the LQA28A, θJA = 59˚C/W. TJMAX = 125˚C for the
LM4960. Depending on the ambient temperature, TA, of the
system surroundings, Equation 4 can be used to find the
maximum internal power dissipation supported by the IC
packaging. If the result of Equation 3 is greater than that of
Equation 4, then either the supply voltage must be in-
creased, the load impedance increased or TA reduced. For
the typical application of a 3V power supply, with V1 set to
12V and a 800nF + 20Ω load, the maximum ambient tem-
perature possible without violating the maximum junction
temperature is approximately 118˚C provided that device
operation is around the maximum power dissipation point.
Thus, for typical applications, power dissipation is not an
issue. Power dissipation is a function of output power and
thus, if typical operation is not around the maximum power
dissipation point, the ambient temperature may be increased
accordingly. Refer to the Typical Performance Characteris-
tics curves for power dissipation information for lower output
levels.
The bridge configuration also creates a second advantage
over single-ended amplifiers. Since the differential outputs,
Vo1 and Vo2, are biased at half-supply, no net DC voltage
exists across the load. This eliminates the need for an output
coupling capacitor which is required in a single supply,
single-ended amplifier configuration. Without an output cou-
pling capacitor, the half-supply bias across the load would
result in both increased internal IC power dissipation and
also possible loudspeaker damage.
AMPLIFIER POWER DISSIPATION
Power dissipation is a major concern when designing a
successful amplifier, whether the amplifier is bridged or
single-ended. A direct consequence of the increased power
delivered to the load by a bridge amplifier is an increase in
internal power dissipation. Since the amplifier portion of the
LM4960 has two operational amplifiers, the maximum inter-
nal power dissipation is 4 times that of a single-ended am-
plifier. The maximum power dissipation for a given BTL
application can be derived from Equation 1.
EXPOSED-DAP PACKAGE PCB MOUNTING
CONSIDERATIONS
The LM4960’s exposed-DAP (die attach paddle) package
(LD) provides a low thermal resistance between the die and
the PCB to which the part is mounted and soldered. The low
thermal resistance allows rapid heat transfer from the die to
the surrounding PCB copper traces, ground plane, and sur-
rounding air. The LD package should have its DAP soldered
to a copper pad on the PCB. The DAP’s PCB copper pad
may be connected to a large plane of continuous unbroken
copper. This plane forms a thermal mass, heat sink, and
radiation area. Further detailed and specific information con-
cerning PCB layout, fabrication, and mounting an LD (LLP)
package is found in National Semiconductor’s Package En-
gineering Group under application note AN1187.
2
PDMAX(AMP) = 4(VDD
)
/ (2π2ZL)
(1)
where
ZL = Ro1 + Ro2 +1/2πfc
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and is apt to create pops upon device enable. Thus, by
minimizing the capacitor value based on desired low fre-
quency response, turn-on pops can be minimized.
Application Information (Continued)
SHUTDOWN FUNCTION
In many applications, a microcontroller or microprocessor
output is used to control the shutdown circuitry to provide a
quick, smooth transition into shutdown. Another solution is to
use a single-pole, single-throw switch, and a pull-up resistor.
One terminal of the switch is connected to GND. The other
side is connected to the two shutdown pins and the terminal
of the pull-up resistor. The remaining resistance terminal is
connected to VDD. If the switch is open, then the external
pull-up resistor connected to VDD will enable the LM4960.
This scheme guarantees that the shutdown pins will not float
thus preventing unwanted state changes.
SELECTING BYPASS CAPACITOR FOR AUDIO
AMPLIFIER
Besides minimizing the input capacitor value, careful consid-
eration should be paid to the bypass capacitor value. Bypass
capacitor, CB, is the most critical component to minimize
turn-on pops since it determines how fast the amplifer turns
on. The slower the amplifier’s outputs ramp to their quies-
cent DC voltage (nominally 1/2 VDD), the smaller the turn-on
pop. Choosing CB equal to 1.0µF along with a small value of
Ci (in the range of 0.039µF to 0.39µF), should produce a
virtually clickless and popless shutdown function. Although
the device will function properly, (no oscillations or motor-
boating), with CB equal to 0.1µF, the device will be much
more susceptible to turn-on clicks and pops. Thus, a value of
CB equal to 1.0µF is recommended in all but the most cost
sensitive designs.
PROPER SELECTION OF EXTERNAL COMPONENTS
Proper selection of external components in applications us-
ing integrated power amplifiers, and switching DC-DC con-
verters, is critical for optimizing device and system perfor-
mance. Consideration to component values must be used to
maximize overall system quality.
SELECTING FEEDBACK CAPACITOR FOR AUDIO
AMPLIFIER
The best capacitors for use with the switching converter
portion of the LM4960 are multi-layer ceramic capacitors.
They have the lowest ESR (equivalent series resistance)
and highest resonance frequency, which makes them opti-
mum for high frequency switching converters.
The LM4960 is unity-gain stable which gives the designer
maximum system flexability. However, to drive ceramic
speakers, a typical application requires a closed-loop differ-
ential gain of 10. In this case a feedback capacitor (Cf2) will
be needed as shown in Figure 2 to bandwidth limit the
amplifier.
When selecting a ceramic capacitor, only X5R and X7R
dielectric types should be used. Other types such as Z5U
and Y5F have such severe loss of capacitance due to effects
of temperature variation and applied voltage, they may pro-
vide as little as 20% of rated capacitance in many typical
applications. Always consult capacitor manufacturer’s data
curves before selecting a capacitor. High-quality ceramic
capacitors can be obtained from Taiyo-Yuden, AVX, and
Murata.
This feedback capacitor creates a low pass filter that elimi-
nates possible high frequency oscillations. Care should be
taken when calculating the -3dB frequency because an in-
correct combination of Rf and Cf2 will cause rolloff before the
desired frequency
SELECTING OUTPUT CAPACITOR (CO) FOR BOOST
CONVERTER
POWER SUPPLY BYPASSING
A single 4.7µF to 10µF ceramic capacitor will provide suffi-
cient output capacitance for most applications. If larger
amounts of capacitance are desired for improved line sup-
port and transient response, tantalum capacitors can be
used. Aluminum electrolytics with ultra low ESR such as
Sanyo Oscon can be used, but are usually prohibitively
expensive. Typical AI electrolytic capacitors are not suitable
for switching frequencies above 500 kHz because of signifi-
cant ringing and temperature rise due to self-heating from
ripple current. An output capacitor with excessive ESR can
also reduce phase margin and cause instability.
As with any amplifier, proper supply bypassing is critical for
low noise performance and high power supply rejection. The
capacitor location on both V1 and VDD pins should be as
close to the device as possible.
SELECTING INPUT CAPACITOR FOR AUDIO
AMPLIFIER
One of the major considerations is the closedloop bandwidth
of the amplifier. To a large extent, the bandwidth is dictated
by the choice of external components shown in Figure 1. The
input coupling capacitor, Ci, forms a first order high pass filter
which limits low frequency response. This value should be
chosen based on needed frequency response for a few
distinct reasons.
In general, if electrolytics are used, we recommended that
they be paralleled with ceramic capacitors to reduce ringing,
switching losses, and output voltage ripple.
High value input capacitors are both expensive and space
hungry in portable designs. Clearly, a certain value capacitor
is needed to couple in low frequencies without severe at-
tenuation. But ceramic speakers used in portable systems,
whether internal or external, have little ability to reproduce
signals below 100Hz to 150Hz. Thus, using a high value
input capacitor may not increase actual system perfor-
mance.
SELECTING INPUT CAPACITOR (Cs1) FOR BOOST
CONVERTER
An input capacitor is required to serve as an energy reservoir
for the current which must flow into the coil each time the
switch turns ON. This capacitor must have extremely low
ESR, so ceramic is the best choice. We recommend a
nominal value of 4.7µF, but larger values can be used. Since
this capacitor reduces the amount of voltage ripple seen at
the input pin, it also reduces the amount of EMI passed back
along that line to other circuitry.
In addition to system cost and size, click and pop perfor-
mance is affected by the value of the input coupling capaci-
tor, Ci. A high value input coupling capacitor requires more
charge to reach its quiescent DC voltage (nominally 1/2
VDD). This charge comes from the output via the feedback
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Best performance is usually obtained when the converter is
operated in “continuous” mode at the load current range of
interest, typically giving better load regulation and less out-
put ripple. Continuous operation is defined as not allowing
the inductor current to drop to zero during the cycle. It should
be noted that all boost converters shift over to discontinuous
operation as the output load is reduced far enough, but a
larger inductor stays “continuous” over a wider load current
range.
Application Information (Continued)
SETTING THE OUTPUT VOLTAGE (V1) OF BOOST
CONVERTER
The output voltage is set using the external resistors R1 and
R2 (see Figure 1). A value of approximately 13.3kΩ is rec-
ommended for R2 to establish a divider current of approxi-
mately 92µA. R1 is calculated using the formula:
To better understand these trade-offs, a typical application
circuit (5V to 12V boost with a 10µH inductor) will be ana-
lyzed. We will assume:
R1 = R2 X (V2/1.23 − 1)
(5)
FEED-FORWARD COMPENSATION FOR BOOST
CONVERTER
VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V
Although the LM4960’s internal Boost converter is internally
compensated, the external feed-forward capacitor Cf is re-
quired for stability (see Figure 1). Adding this capacitor puts
a zero in the loop response of the converter. The recom-
mended frequency for the zero fz should be approximately
6kHz. Cf1 can be calculated using the formula:
Since the frequency is 1.6MHz (nominal), the period is ap-
proximately 0.625µs. The duty cycle will be 62.5%, which
means the ON-time of the switch is 0.390µs. It should be
noted that when the switch is ON, the voltage across the
inductor is approximately 4.5V. Using the equation:
Cf1 = 1 / (2 X R1 X fz)
(6)
V = L (di/dt)
We can then calculate the di/dt rate of the inductor which is
found to be 0.45 A/µs during the ON-time. Using these facts,
we can then show what the inductor current will look like
during operation:
SELECTING DIODES
The external diode used in Figure 1 should be a Schottky
diode. A 20V diode such as the MBR0520 is recommended.
The MBR05XX series of diodes are designed to handle a
maximum average current of 0.5A. For applications exceed-
ing 0.5A average but less than 1A, a Microsemi UPS5817
can be used.
DUTY CYCLE
The maximum duty cycle of the boost converter determines
the maximum boost ratio of output-to-input voltage that the
converter can attain in continuous mode of operation. The
duty cycle for a given boost application is defined as:
20076583
Duty Cycle = VOUT + VDIODE - VIN / VOUT + VDIODE - VSW
FIGURE 2. 10µH Inductor Current
5V - 12V Boost (LM4960)
This applies for continuous mode operation.
INDUCTANCE VALUE
During the 0.390µs ON-time, the inductor current ramps up
0.176A and ramps down an equal amount during the OFF-
time. This is defined as the inductor “ripple current”. It can
also be seen that if the load current drops to about 33mA,
the inductor current will begin touching the zero axis which
means it will be in discontinuous mode. A similar analysis
can be performed on any boost converter, to make sure the
ripple current is reasonable and continuous operation will be
maintained at the typical load current values.
The first question we are usually asked is: “How small can I
make the inductor.” (because they are the largest sized
component and usually the most costly). The answer is not
simple and involves trade-offs in performance. Larger induc-
tors mean less inductor ripple current, which typically means
less output voltage ripple (for a given size of output capaci-
tor). Larger inductors also mean more load power can be
delivered because the energy stored during each switching
cycle is:
MAXIMUM SWITCH CURRENT
The maximum FET switch current available before the cur-
rent limiter cuts in is dependent on duty cycle of the appli-
cation. This is illustrated in a graph in the typical perfor-
mance characterization section which shows typical values
of switch current as a function of effective (actual) duty cycle.
E = L/2 X (lp)2
Where “lp” is the peak inductor current. An important point to
observe is that the LM4960 will limit its switch current based
on peak current. This means that since lp(max) is fixed,
increasing L will increase the maximum amount of power
available to the load. Conversely, using too little inductance
may limit the amount of load current which can be drawn
from the output.
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10
Some additional guidelines to be observed:
Application Information (Continued)
CALCULATING OUTPUT CURRENT OF BOOST
1. Keep the path between L1, D1, and Co extremely short.
Parasitic trace inductance in series with D1 and Co will
increase noise and ringing.
CONVERTER (IAMP
)
As shown in Figure 2 which depicts inductor current, the load
current is related to the average inductor current by the
relation:
2. The feedback components R1, R2 and Cf 1 must be kept
close to the FB pin of U1 to prevent noise injection on the FB
pin trace.
3. If internal ground planes are available (recommended)
use vias to connect directly to ground at pin 2 of U1, as well
as the negative sides of capacitors Cs1 and Co.
ILOAD = IIND(AVG) x (1 - DC)
(7)
Where "DC" is the duty cycle of the application. The switch
current can be found by:
GENERAL MIXED-SIGNAL LAYOUT
RECOMMENDATION
This section provides practical guidelines for mixed signal
PCB layout that involves various digital/analog power and
ground traces. Designers should note that these are only
"rule-of-thumb" recommendations and the actual results will
depend heavily on the final layout.
ISW = IIND(AVG) + 1/2 (IRIPPLE
)
(8)
Inductor ripple current is dependent on inductance, duty
cycle, input voltage and frequency:
Power and Ground Circuits
IRIPPLE = DC x (VIN-VSW) / (f x L)
(9)
For 2 layer mixed signal design, it is important to isolate the
digital power and ground trace paths from the analog power
and ground trace paths. Star trace routing techniques (bring-
ing individual traces back to a central point rather than daisy
chaining traces together in a serial manner) can have a
major impact on low level signal performance. Star trace
routing refers to using individual traces to feed power and
ground to each circuit or even device. This technique will
take require a greater amount of design time but will not
increase the final price of the board. The only extra parts
required may be some jumpers.
combining all terms, we can develop an expression which
allows the maximum available load current to be calculated:
ILOAD(max) = (1–DC)x(ISW(max)–DC(VIN-VSW))/fL (10)
The equation shown to calculate maximum load current
takes into account the losses in the inductor or turn-OFF
switching losses of the FET and diode.
DESIGN PARAMETERS VSW AND ISW
Single-Point Power / Ground Connection
The value of the FET "ON" voltage (referred to as VSW in
equations 7 thru 10) is dependent on load current. A good
approximation can be obtained by multiplying the "ON Re-
sistance" of the FET times the average inductor current.
The analog power traces should be connected to the digital
traces through a single point (link). A "Pi-filter" can be helpful
in minimizing high frequency noise coupling between the
analog and digital sections. It is further recommended to
place digital and analog power traces over the correspond-
ing digital and analog ground traces to minimize noise cou-
pling.
FET on resistance increases at VIN values below 5V, since
the internal N-FET has less gate voltage in this input voltage
range (see Typical Performance Characteristics curves).
Above VIN = 5V, the FET gate voltage is internally clamped
to 5V.
Placement of Digital and Analog Components
All digital components and high-speed digital signals traces
should be located as far away as possible from analog
components and circuit traces.
The maximum peak switch current the device can deliver is
dependent on duty cycle. For higher duty cycles, see Typical
Performance Characteristics curves.
Avoiding Typical Design / Layout Problems
INDUCTOR SUPPLIERS
Avoid ground loops or running digital and analog traces
parallel to each other (side-by-side) on the same PCB layer.
When traces must cross over each other do it at 90 degrees.
Running digital and analog traces at 90 degrees to each
other from the top to the bottom side as much as possible will
minimize capacitive noise coupling and crosstalk.
Recommended suppliers of inductors for the LM4960 in-
clude, but are not limited to Taiyo-Yuden, Sumida, Coilcraft,
Panasonic, TDK and Murata. When selecting an inductor,
make certain that the continuous current rating is high
enough to avoid saturation at peak currents. A suitable core
type must be used to minimize core (switching) losses, and
wire power losses must be considered when selecting the
current rating.
PCB LAYOUT GUIDELINES
High frequency boost converters require very careful layout
of components in order to get stable operation and low
noise. All components must be as close as possible to the
LM4802 device. It is recommended that a 4-layer PCB be
used so that internal ground planes are available.
11
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Physical Dimensions inches (millimeters) unless otherwise noted
LLP, Plastic, Quad
Order Number LM4960SQ
NS Package Number SQA28A
National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves
the right at any time without notice to change said circuitry and specifications.
For the most current product information visit us at www.national.com.
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