LMP7717MAX [NSC]
88 MHz, Precision, Low Noise, 1.8V CMOS Input, Decompensated Operational Amplifier; 88兆赫，高精度，低噪声， 1.8V CMOS输入，失代偿运算放大器
| 型号： | LMP7717MAX |
| 厂家： | 
National Semiconductor |
| 描述： | 88 MHz, Precision, Low Noise, 1.8V CMOS Input, Decompensated Operational Amplifier |
| 文件： | 总24页 (文件大小：1123K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
November 6, 2007
LMP7717/LMP7718
88 MHz, Precision, Low Noise, 1.8V CMOS Input,
Decompensated Operational Amplifier
General Description
Features
The LMP7717 (single) and the LMP7718 (dual) low noise,
CMOS input operational amplifiers offer a low input voltage
(Typical 5V supply, unless otherwise noted)
Input offset voltage
Input referred voltage noise
Input bias current
Gain bandwidth product
Supply voltage range
±150 µV (max)
5.8 nV/√Hz
100 fA
88 MHz
1.8V to 5.5V
■
noise density of 5.8 nV/
while consuming only 1.15 mA
■
■
■
■
■
(LMP7717) of quiescent current. The LMP7717/LMP7718 are
stable at a gain of 10 and have a gain bandwidth (GBW)
product of 88 MHz. The LMP7717/LMP7718 have a supply
voltage range of 1.8V to 5.5V and can operate from a single
supply. The LMP7717/LMP7718 each feature a rail-to-rail
output stage. Both amplifiers are part of the LMP® precision
amplifier family and are ideal for a variety of instrumentation
applications.
Supply current per channel
LMP7717
LMP7718
1.15 mA
1.30 mA
—
—
Rail-to-Rail output swing
■
@ 10 kΩ load
@ 2 kΩ load
Guaranteed 2.5V and 5.0V performance
25 mV from rail
45 mV from rail
—
—
The LMP7717 family provides optimal performance in low
voltage and low noise systems. A CMOS input stage, with
typical input bias currents in the range of a few femto-Am-
peres, and an input common mode voltage range, which
includes ground, make the LMP7717/LMP7718 ideal for low
power sensor applications where high speeds are needed.
■
■
■
Total harmonic distortion
Temperature range
0.04% @1 kHz, 600Ω
−40°C to 125°C
The LMP7717/LMP7718 are manufactured using National’s
advanced VIP50 process. The LMP7717 is offered in either a
5-Pin SOT23 or an 8-Pin SOIC package. The LMP7718 is
offered in either the 8-Pin SOIC or the 8-Pin MSOP.
Applications
ADC interface
■
■
■
■
Photodiode amplifiers
Active filters and buffers
Low noise signal processing
Medical instrumentation
■
Sensor interface applications
■
Typical Application
30010839
30010869
Input Referred Voltage Noise vs. Frequency
Photodiode Transimpedance Amplifier
LMP® is a registered trademark of National Semiconductor Corporation.
© 2007 National Semiconductor Corporation
300108
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Soldering Information
Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
ꢀInfrared or Convection (20 sec)
ꢀWave Soldering Lead Temp (10 sec)
235°C
260°C
Operating Ratings (Note 1)
Temperature Range (Note 3)
Supply Voltage (V+ – V−)
−40°C ≤ TA ≤ 125°C
0°C ≤ TA ≤ 125°C
Package Thermal Resistance (θJA (Note 3))
5-Pin SOT23
8-Pin SOIC
8-Pin MSOP
ESD Tolerance (Note 2)
−40°C to 125°C
ꢀHuman Body Model
ꢀMachine Model
VIN Differential
Supply Voltage (V+ – V−)
Input/Output Pin Voltage
Storage Temperature Range
Junction Temperature (Note 3)
2000V
200V
±0.3V
2.0V to 5.5V
1.8V to 5.5V
6.0V
V+ +0.3V, V− −0.3V
−65°C to 150°C
+150°C
180°C/W
190°C/W
236°C/W
2.5V Electrical Characteristics (Note 4)
Unless otherwise specified, all limits are guaranteed for TA = 25°C, V+ = 2.5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply at
the temperature extremes.
Symbol
Parameter
Conditions
Min
Typ
Max
Units
(Note 6) (Note 5) (Note 6)
VOS
Input Offset Voltage
±20
±180
±480
µV
TC VOS Input Offset Average Drift
(Note 7)
LMP7717
LMP7718
VCM = 1.0V
(Notes 8, 9)
−1.0
−1.8
0.05
±4
μV/°C
IB
Input Bias Current
1
25
−40°C ≤ TA ≤ 85°C
−40°C ≤ TA ≤ 125°C
pA
0.05
.006
94
1
100
IOS
Input Offset Current
VCM = 1.0V
(Note 9)
0.5
50
pA
dB
CMRR Common Mode Rejection Ratio
83
80
0V ≤ VCM ≤ 1.4V
2.0V ≤ V+ ≤ 5.5V, VCM = 0V
PSRR
Power Supply Rejection Ratio
85
80
100
98
dB
V
1.8V ≤ V+ ≤ 5.5V, VCM = 0V
85
CMVR
AVOL
Input Common-Mode Voltage
Range
−0.3
−0.3
1.5
1.5
CMRR ≥ 60 dB
CMRR ≥ 55 dB
VOUT = 0.15V to 2.2V,
RL = 2 kΩ to V+/2
Open Loop Gain
LMP7717
88
82
98
92
110
95
25
20
30
15
LMP7718
LMP7717
LMP7718
84
80
dB
VOUT = 0.15V to 2.2V,
RL = 10 kΩ to V+/2
92
88
90
86
RL = 2 kΩ to V+/2
RL = 10 kΩ to V+/2
RL = 2 kΩ to V+/2
RL = 10 kΩ to V+/2
VOUT
Output Swing High
Output Swing Low
70
77
60
66
mV from
rail
70
73
60
62
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IOUT
Output Short Circuit Current
Sourcing to V−
36
30
47
15
VIN = 200 mV (Note 10)
mA
mA
Sinking to V+
7.5
5
VIN = –200 mV (Note 10)
IS
Supply Current per Amplifier
Slew Rate
LMP7717
0.95
1.1
1.30
1.65
LMP7718 per channel
1.5
1.85
SR
AV = +10, Rising (10% to 90%)
AV = +10, Falling (90% to 10%)
AV = +10, RL = 10 kΩ
32
24
88
V/μs
GBWP Gain Bandwidth Product
MHz
en
in
Input-Referred Voltage Noise f = 1 kHz
Input-Referred Current Noise f = 1 kHz
6.2
nV/
pA/
0.01
0.01
THD+N Total Harmonic Distortion +
Noise
%
f = 1 kHz, AV = 1, RL = 600Ω
5V Electrical Characteristics (Note 4)
Unless otherwise specified, all limits are guaranteed for TA = 25°C, V+ = 5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply at
the temperature extremes.
Symbol
Parameter
Conditions
Min
Typ
Max
Units
(Note 6) (Note 5) (Note 6)
VOS
Input Offset Voltage
±10
±150
±450
µV
TC VOS Input Offset Average Drift
(Note 7)
LMP7717
LMP7718
VCM = 2.0V
(Notes 8, 9)
−1.0
−1.8
0.1
±4
μV/°C
IB
Input Bias Current
1
25
−40°C ≤ TA ≤ 85°C
−40°C ≤ TA ≤ 125°C
pA
0.1
.01
100
100
98
1
100
IOS
Input Offset Current
VCM = 2.0V
(Note 9)
0.5
50
pA
dB
CMRR Common Mode Rejection Ratio
85
80
0V ≤ VCM ≤ 3.7V
2.0V ≤ V+ ≤ 5.5V, VCM = 0V
PSRR
Power Supply Rejection Ratio
85
80
dB
V
1.8V ≤ V+ ≤ 5.5V, VCM = 0V
85
CMVR
AVOL
Input Common-Mode Voltage
Range
−0.3
−0.3
4
4
CMRR ≥ 60 dB
CMRR ≥ 55 dB
VOUT = 0.3V to 4.7V,
RL = 2 kΩ to V+/2
Open Loop Gain
LMP7717
88
82
107
90
LMP7718
LMP7717
LMP7718
84
80
dB
VOUT = 0.3V to 4.7V,
RL = 10 kΩ to V+/2
92
88
110
95
90
86
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RL = 2 kΩ to V+/2
VOUT
Output Swing High
LMP7717
LMP7718
35
45
25
42
50
25
60
70
77
80
77
RL = 10 kΩ to V+/2
RL = 2 kΩ to V+/2
60
66
mV from
rail
Output Swing Low
LMP7717
LMP7718
70
73
80
78
RL = 10 kΩ to V+/2
60
66
IOUT
Output Short Circuit Current
Sourcing to V−
46
38
VIN = 200 mV (Note 10)
mA
Sinking to V+
10.5
21
6.5
VIN = –200 mV (Note 10)
IS
Supply Current per Amplifier
Slew Rate
LMP7717
1.15
1.30
1.40
1.75
mA
LMP7718 per channel
1.70
2.05
SR
AV = +10, Rising (10% to 90%)
AV = +10, Falling (90% to 10%)
AV = +10, RL = 10 kΩ
35
28
88
5.8
V/μs
GBWP Gain Bandwidth Product
MHz
nV/
en
in
Input-Referred Voltage Noise f = 1 kHz
Input-Referred Current Noise f = 1 kHz
0.01
0.01
pA/
%
THD+N Total Harmonic Distortion +
Noise
f = 1 kHz, AV = 1, RL = 600Ω
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see the Electrical Characteristics
Tables.
Note 2: Human Body Model, applicable std. MIL-STD-883, Method 3015.7. Machine Model, applicable std. JESD22-A115-A (ESD MM std. of JEDEC)
Field-Induced Charge-Device Model, applicable std. JESD22-C101-C (ESD FICDM std. of JEDEC).
Note 3: The maximum power dissipation is a function of TJ(MAX), θJA. The maximum allowable power dissipation at any ambient temperature is
PD = (TJ(MAX) - TA)/θJA. All numbers apply for packages soldered directly onto a PC Board.
Note 4: Electrical Table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very limited self-heating
of the device such that TJ = TA. No guarantee of parametric performance is indicated in the electrical tables under conditions of internal self-heating where TJ >
TA.
Note 5: Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary over time and will
also depend on the application and configuration. The typical values are not tested and are not guaranteed on shipped production material.
Note 6: Limits are 100% production tested at 25°C. Limits over the operating temperature range are guaranteed through correlations using the statistical quality
control (SQC) method.
Note 7: Offset voltage average drift is determined by dividing the change in VOS by temperature change.
Note 8: Positive current corresponds to current flowing into the device.
Note 9: Input bias current and input offset current are guaranteed by design
Note 10: The short circuit test is a momentary test, the short circuit duration is 1.5 ms.
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4
Connection Diagrams
5-Pin SOT23 (LMP7717)
8-Pin SOIC (LMP7717)
8-Pin SOIC/MSOP (LMP7718)
30010801
Top View
30010802
30010885
Top View
Ordering Information
Package
Part Number
Package Marking
Transport Media
1k Units Tape and Reel
250 Units Tape and Reel
3k Units Tape and Reel
95 Units/Rail
NSC Drawing
LMP7717MF
LMP7717MFE
LMP7717MFX
LMP7717MA
LMP7717MAE
LMP7717MAX
LMP7718MA
LMP7718MAE
LMP7718MAX
LMP7718MM
LMP7718MME
LMP7718MMX
5-Pin SOT23
AT4A
MF05A
LMP7717MA
LMP7718MA
AP4A
250 Units Tape and Reel
2.5k Units Tape and Reel
95 Units/Rail
8-Pin SOIC
8-Pin MSOP
M08A
250 Units Tape and Reel
2.5k Units Tape and Reel
1k Units Tape and Reel
250 Units Tape and Reel
3.5k Units Tape and Reel
MUA08A
5
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Typical Performance Characteristics Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V,
VS = V+ - V−, VCM = VS/2.
TCVOS Distribution (LMP7717)
Offset Voltage Distribution
Offset Voltage Distribution
VOS vs. VCM
30010890
30010891
TCVOS Distribution (LMP7717)
30010893
30010892
Supply Current vs. Supply Voltage (LMP7717)
30010805
30010809
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6
VOS vs. VCM
VOS vs. VCM
30010811
30010851
VOS vs. Supply Voltage
Slew Rate vs. Supply Voltage
30010852
30010812
Input Bias Current vs. VCM
Input Bias Current vs. VCM
30010887
30010862
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Sourcing Current vs. Supply Voltage
Sinking Current vs. Supply Voltage
30010820
30010819
Sourcing Current vs. Output Voltage
Sinking Current vs. Output Voltage
30010850
30010854
Positive Output Swing vs. Supply Voltage
Negative Output Swing vs. Supply Voltage
30010817
30010815
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Positive Output Swing vs. Supply Voltage
Negative Output Swing vs. Supply Voltage
30010816
30010814
Positive Output Swing vs. Supply Voltage
Negative Output Swing vs. Supply Voltage
30010813
30010818
Input Referred Voltage Noise vs. Frequency
Overshoot and Undershoot vs. CLOAD
30010839
30010830
9
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THD+N vs. Frequency
THD+N vs. Frequency
30010804
30010826
THD+N vs. Peak-to-Peak Output Voltage (VOUT
)
THD+N vs. Peak-to-Peak Output Voltage (VOUT)
30010874
30010875
Open Loop Gain and Phase
Closed Loop Output Impedance vs. Frequency
30010832
30010806
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Crosstalk Rejection
Small Signal Transient Response, AV = +10
30010853
30010880
Large Signal Transient Response, AV = +10
Small Signal Transient Response, AV = +10
30010855
30010857
Large Signal Transient Response, AV = +10
PSRR vs. Frequency
30010863
30010870
11
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CMRR vs. Frequency
Input Common Mode Capacitance vs. VCM
30010856
30010876
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12
The LMP7717/LMP7718 require a gain of ±10 to be stable.
However, with an external compensation network (a simple
RC network) these parts can be stable with gains of ±1 and
still maintain the higher slew rate. Looking at the Bode plots
for the LMP7717 and its closest equivalent unity gain stable
op amp, the LMP7715, one can clearly see the increased
bandwidth of the LMP7717. Both plots are taken with a par-
allel combination of 20 pF and 10 kΩ for the output load.
Application Information
ADVANTAGES OF THE LMP7717/LMP7718
Wide Bandwidth at Low Supply Current
The LMP7717/LMP7718 are high performance op amps that
provide a GBW of 88 MHz with a gain of 10 while drawing a
low supply current of 1.15 mA. This makes them ideal for pro-
viding wideband amplification in data acquisition applications.
With the proper external compensation the LMP7717 can be
operated at gains of ±1 and still maintain much faster slew
rates than comparable unity gain stable amplifiers. The in-
crease in bandwidth and slew rate is obtained without any
additional power consumption over the LMP7715.
Low Input Referred Noise and Low Input Bias Current
The LMP7717/LMP7718 have a very low input referred volt-
age noise density (5.8 nV/
ensures a small input bias current (100 fA) and low input re-
ferred current noise (0.01 pA/ ). This is very helpful in
at 1 kHz). A CMOS input stage
maintaining signal integrity, and makes the LMP7717/
LMP7718 ideal for audio and sensor based applications.
Low Supply Voltage
The LMP7717 and the LMP7718 have performance guaran-
teed at 2.5V and 5V supply. These parts are guaranteed to
be operational at all supply voltages between 2.0V and 5.5V,
for ambient temperatures ranging from −40°C to 125°C, thus
utilizing the entire battery lifetime. The LMP7717/LMP7718
are also guaranteed to be operational at 1.8V supply voltage,
for temperatures between 0°C and 125°C optimizing their us-
age in low-voltage applications.
30010822
FIGURE 1. LMP7717 AVOL vs. Frequency
RRO and Ground Sensing
Rail-to-Rail output swing provides the maximum possible dy-
namic range. This is particularly important when operating at
low supply voltages. An innovative positive feedback scheme
is used to boost the current drive capability of the output
stage. This allows the LMP7717/LMP7718 to source more
than 40 mA of current at 1.8V supply. This also limits the per-
formance of the these parts as comparators, and hence the
usage of the LMP7717 and the LMP7718 in an open-loop
configuration is not recommended. The input common-mode
range includes the negative supply rail which allows direct
sensing at ground in single supply operation.
Small Size
The small footprints of the LMP7717 packages and the
LMP7718 packages save space on printed circuit boards, and
enable the design of smaller electronic products, such as cel-
lular phones, pagers, or other portable systems. Long traces
between the signal source and the op amp make the signal
path more susceptible to noise pick up.
30010823
FIGURE 2. LMP7715 AVOL vs. Frequency
Figure 1 shows the much larger 88 MHz bandwidth of the
LMP7717 as compared to the 17 MHz bandwidth of the
LMP7715 shown in Figure 2. The decompensated LMP7717
has five times the bandwidth of the LMP7715.
The physically smaller LMP7717 or LMP7718 packages allow
the op amp to be placed closer to the signal source, thus re-
ducing noise pickup and maintaining signal integrity.
What is a Decompensated Op Amp?
USING THE DECOMPENSATED LMP7717
Advantages of Decompensated Op Amp
The differences between the unity gain stable op amp and the
decompensated op amp are shown in Figure 3. This Bode plot
assumes an ideal two pole system. The dominant pole of the
decompensated op amp is at a higher frequency, f1, as com-
pared to the unity gain stable op amp which is at fd as shown
in Figure 3. This is done in order to increase the speed capa-
bility of the op amp while maintaining the same power dissi-
pation of the unity gain stable op amp. The LMP7717/
LMP7718 have a dominant pole at 8.6 Hz. The unity gain sta-
ble LMP7715/LMP7716 have their dominant pole at 1.6 Hz.
A unity gain stable op amp, which is fully compensated, is
designed to operate with good stability down to gains of ±1.
The large amount of compensation does provide an op amp
that is relatively easy to use; however, a decompensated op
amp is designed to maximize the bandwidth and slew rate
without any additional power consumption. This can be very
advantageous.
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30010825
FIGURE 4. LMP7717 with Lead-Lag Compensation for
Inverting Configuration
30010824
To cover how to calculate the compensation network values
it is necessary to introduce the term called the feedback factor
or F. The feedback factor F is the feedback voltage VA-VB
across the op amp input terminals relative to the op amp out-
FIGURE 3. Open Loop Gain for Unity Gain Stable Op Amp
and Decompensated Op Amp
put voltage VOUT
.
Having a higher frequency for the dominate pole will result in:
1. The DC open loop gain (AVOL) extending to a higher
frequency.
2. A wider closed loop bandwidth.
3. Better slew rate due to reduced compensation
capacitance within the op amp.
From feedback theory the classic form of the feedback equa-
tion for op amps is:
The second open loop pole (f2) for the LMP7717/LMP7718
occurs at 45 MHz. The unity gain (fu’) occurs after the second
pole at 51 MHz. An ideal two pole system would give a phase
margin of 45° at the location of the second pole. The
LMP7717/LMP7718 have parasitic poles close to the second
pole, giving a phase margin closer to 0°. Therefore it is nec-
essary to operate the LMP7717/LMP7718 at a closed loop
gain of 10 or higher, or to add external compensation in order
to assure stability.
A is the open loop gain of the amplifier and AF is the loop gain.
Both are highly important in analyzing op amps. Normally AF
>>1 and so the above equation reduces to:
For the LMP7715, the gain bandwidth product occurs at 17
MHz. The curve is constant from fd to fu which occurs before
the second pole.
For the LMP7717/LMP7718 the GBW = 88 MHz and is con-
stant between f1 and f2. The second pole at f2 occurs before
AVOL =1. Therefore fu’ occurs at 51 MHz, well before the GBW
frequency of 88 MHz. For decompensated op amps the unity
gain frequency and the GBW are no longer equal. Gmin is the
minimum gain for stability and for the LMP7717/LMP7718 this
is a gain of 10 or 20 dB.
Deriving the equations for the lead-lag compensation is be-
yond the scope of this datasheet. The derivation is based on
the feedback equations that have just been covered. The in-
verse of feedback factor for the circuit in Figure 4 is:
Input Lead-Lag Compensation
(1)
The recommended technique which allows the user to com-
pensate the LMP7717/LMP7718 for stable operation at any
gain is lead-lag compensation. The compensation compo-
nents added to the circuit allow the user to shape the feedback
function to make sure there is sufficient phase margin when
the loop gain is as low as 0 dB and still maintain the advan-
tages over the unity gain op amp. Figure 4 shows the lead-
lag configuration. Only RC and C are added for the necessary
compensation.
where 1/F's pole is located at
(2)
1/F's zero is located at
(3)
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(4)
The circuit gain for Figure 4 at low frequencies is −RF/RIN, but
F, the feedback factor is not equal to the circuit gain. The
feedback factor is derived from feedback theory and is the
same for both inverting and non-inverting configurations. Yes,
the feedback factor at low frequencies is equal to the gain for
the non-inverting configuration.
(5)
From this formula, we can see that
•
1/F's zero is located at a lower frequency compared with
1/F's pole.
•
•
1/F's value at low frequency is 1 + RF/RIN.
This method creates one additional pole and one
additional zero.
30010848
FIGURE 5. LMP7717/LMP7718 Simplified Bode Plot
•
This pole-zero pair will serve two purposes:
To raise the 1/F value at higher frequencies prior to its
intercept with A, the open loop gain curve, in order to
meet the Gmin = 10 requirement. For the LMP7717
some overcompensation will be necessary for good
stability.
—
To obtain stable operation with gains under 10 V/V the open
loop gain margin must be reduced at high frequencies to
where there is a 45° phase margin when the gain margin of
the circuit with the external compensation is 0 dB. The pole
and zero in F, the feedback factor, control the gain margin at
the higher frequencies. The distance between F and AVOL is
the gain margin; therefore, the unity gain point (0 dB) is where
F crosses the AVOL curve.
To achieve the previous purpose above with no
additional loop phase delay.
—
Please note the constraint 1/F ≥ Gmin needs to be satisfied
only in the vicinity where the open loop gain A and 1/F inter-
sect; 1/F can be shaped elsewhere as needed. The 1/F pole
must occur before the intersection with the open loop gain A.
For the example being used RIN = RF for a gain of −1. There-
fore F = 6 dB at low frequencies. At the higher frequencies
the minimum value for F is 18 dB for 45° phase margin. From
Equation 5 we have the following relationship:
In order to have adequate phase margin, it is desirable to fol-
low these two rules:
Rule 1
1/F and the open loop gain A should intersect at the
frequency where there is a minimum of 45° of phase
margin. When over-compensation is required the in-
tersection point of A and 1/F is set at a frequency
where the phase margin is above 45°, therefore in-
creasing the stability of the circuit.
Now set RF = RIN = R. With these values and solving for RC
we have RC = R/5.9. Note that the value of C does not affect
the ratio between the resistors. Once the value of the resistors
is set, then the position of the pole in F must be set. A 2 kΩ
resistor is used for RF and RIN in this design. Therefore the
value for RC is set at 330Ω, the closest standard value for 2
kΩ/5.9.
Rule 2
1/F’s pole should be set at least one decade below
the intersection with the open loop gain A in order to
take advantage of the full 90° of phase lead brought
by 1/F’s pole which is F’s zero. This ensures that the
effect of the zero is fully neutralized when the 1/F and
A plots intersect each other.
Rewriting Equation 2 to solve for the minimum capacitor value
gives the following equation:
Calculating Lead-Lag Compensation for LMP7717
C = 1/(2πfpRC)
Figure 5 is the same plot as Figure 1, but the AVOL and phase
curves have been redrawn as smooth lines to more readily
show the concepts covered, and to clearly show the key pa-
rameters used in the calculations for lead-lag compensation.
The feedback factor curve, F, intersects the AVOL curve at
about 12 MHz. Therefore the pole of F should not be any
larger than 1.2 MHz. Using this value and RC = 330Ω the
minimum value for C is 390 pF. Figure 6 shows that there is
too much overshoot, but the part is stable. Increasing C to 2.2
nF did not improve the ringing, as shown in Figure 7.
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30010803
30010810
FIGURE 6. First Try at Compensation, Gain = −1
FIGURE 9. RC = 240Ω and C = 2.2 nF, Gain = −1
To summarize, the following steps were taken to compensate
the LMP7717 for a gain of −1:
1. Values for Rc and C were calculated from the Bode plot
to give an expected phase margin of 45°. The values
were based on RIN = RF = 2 kΩ. These calculations gave
Rc = 330Ω and C = 390 pF.
2. To reduce the ringing C was increased to 2.2 nF which
moved the pole of F, the feedback factor, farther away
from the AVOL curve.
3. There was still too much ringing so 2 dB of over-
compensation was added to F. This was done by
decreasing RC to 240Ω.
The LMP7715 is the fully compensated part which is compa-
rable to the LMP7717. Using the LMP7715 in the same setup,
but removing the compensation network, provided the re-
sponse shown in Figure 10 .
30010807
FIGURE 7. C Increased to 2.2 nF, Gain = −1
Some over-compensation appears to be needed for the de-
sired overshoot characteristics. Instead of intersecting the
AVOL curve at 18 dB, 2 dB of over-compensation will be used,
and the AVOL curve will be intersected at 20 dB. Using Equa-
tion 5 for 20 dB, or 10 V/V, the closest standard value of RC
is 240Ω. The following two waveforms show the new resistor
value with C = 390 pF and 2.2 nF. Figure 9 shows the final
compensation and a very good response for the 1 MHz
square wave.
30010821
FIGURE 10. LMP7715 Response
For large signal response the rise and fall times are dominat-
ed by the slew rate of the op amps. Even though both parts
are quite similar the LMP7717 will give rise and fall times
about 2.5 times faster than the LMP7715. This is possible
because the LMP7717 is a decompensated op amp and even
though it is being used at a gain of −1, the speed is preserved
by using a good technique for external compensation.
30010808
FIGURE 8. RC = 240Ω and C = 390 pF, Gain = −1
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16
Non-Inverting Compensation
than the fully compensated parts. Figure 13 shows the gain =
1, or the buffer configuration, for these parts.
For the non-inverting amp the same theory applies for estab-
lishing the needed compensation. When setting the inverting
configuration for a gain of −1, F has a value of 2. For the non-
inverting configuration both F and the actual gain are the
same, making the non-inverting configuration more difficult to
compensate. Using the same circuit as shown in Figure 4, but
setting up the circuit for non-inverting operation (gain of +2)
results in similar performance as the inverting configuration
with the inputs set to half the amplitude to compensate for the
additional gain. Figure 11 below shows the results.
30010884
FIGURE 13. LMP7717 with Lead-Lag Compensation for
Non-Inverting Configuration
Figure 13 is the result of using Equation 5 and additional ex-
perimentation in the lab. RP is not part of Equation 5, but it is
necessary to introduce another pole at the input stage for
good performance at gain = +1. Equation 5 is shown below
with RIN = ∞.
Using 2 kΩ for RF and solving for RC gives RC = 2000/6.9 =
290Ω. The closest standard value for RC is 300Ω. After some
fine tuning in the lab RC = 330Ω and RP = 1.5 kΩ were chosen
as the optimum values. RP together with the input capacitance
at the non-inverting pin inserts another pole into the compen-
sation for the LMP7717. Adding this pole and slightly reducing
the compensation for 1/F (using a slightly higher resistor value
for RC) gives the optimum response for a gain of +1. Figure
14 is the response of the circuit shown in Figure 13. Figure
15 shows the response of the LMP7715 in the buffer config-
uration with no compensation and RP = RF = 0.
30010882
FIGURE 11. RC = 240Ω and C = 2.2 nF, Gain = +2
30010883
FIGURE 12. LMP7715 Response Gain = +2
The response shown in Figure 11 is close to the response
shown in Figure 9. The part is actually slightly faster in the
non-inverting configuration. Decreasing the value of RC to
around 200Ω can decrease the negative overshoot but will
have slightly longer rise and fall times. The other option is to
add a small resistor in series with the input signal. Figure 12
shows the performance of the LMP7715 with no compensa-
tion. Again the decompensated parts are almost 2.5 times
faster than the fully compensated op amp.
30010888
FIGURE 14. RC = 330Ω and C = 10 nF, Gain = +1
The most difficult op amp configuration to stabilize is the gain
of +1. With proper compensation the LMP7717/LMP7718 can
be used in this configuration and still maintain higher speeds
17
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30010861
FIGURE 16. Transimpedance Amplifier
Figure 16 is the complete schematic for a transimpedance
amplifier. Only the supply bypass capacitors are not shown.
CD represents the photodiode capacitance which is given on
its datasheet. CCM is the input common mode capacitance of
the op amp and, for the LMP7717 it is shown in the last graph
of the Typical Performance Characteristics section of this
datasheet. In Figure 16 the inverting input pin of the LMP7717
is kept at virtual ground. Even though the diode is connected
to the 2.5V line, a power supply line is AC ground, thus CD is
connected to ground.
30010889
FIGURE 15. LMP7715 Response Gain = +1
With no increase in power consumption the decompensated
op amp offers faster speed than the compensated equivalent
part . These examples used RF = 2 kΩ. This value is high
enough to be easily driven by the LMP7717/LMP7718, yet
small enough to minimize the effects from the parasitic ca-
pacitance of both the PCB and the op amp.
Figure 17 shows the schematic needed to derive F, the feed-
back factor, for a transimpedance amplifier. In this figure
CD + CCM = CIN. Therefore it is critical that the designer knows
the diode capacitance and the op amp input capacitance. The
photodiode is close to an ideal current source once its ca-
pacitance is included in the model. What kind of circuit is this?
Without CF there is only an input capacitor and a feedback
resistor. This circuit is a differentiator! Remember, differen-
tiator circuits are inherently unstable and must be compen-
sated. In this case CF compensates the circuit.
Note: When using the LMP7717/LMP7718, proper high fre-
quency PCB layout must be followed. The GBW of these parts
is 88 MHz, making the PCB layout significantly more critical
than when using the compensated counterparts which have
a GBW of 17 MHz.
TRANSIMPEDANCE AMPLIFIER
An excellent application for either the LMP7717 or the
LMP7718 is as a transimpedance amplifier. With a GBW
product of 88 MHz these parts are ideal for high speed data
transmission by light. The circuit shown on the front page of
the datasheet is the circuit used to test the
LMP7717/LMP7718 as transimpedance amplifiers. The only
change is that VB is tied to the VCC of the part, thus the direc-
tion of the diode is reversed from the circuit shown on the front
page.
Very high speed components were used in testing to check
the limits of the LMP7717/LMP7718 in a transimpedance
configuration. The photodiode part number is PIN-HR040
from OSI Optoelectronics. The diode capacitance for this part
is only about 7 pF for the 2.5V bias used (VCC to virtual
ground). The rise time for this diode is 1 nsec. A laser diode
was used for the light source. Laser diodes have on and off
times under 5 nsec. The speed of the selected optical com-
ponents allowed an accurate evaluation of the LMP7717 as
a transimpedance amplifier. Nationals evaluation board for
decompensated op amps, PN 551013271-001 A, was used
and only minor modifications were necessary and no traces
had to be cut.
30010864
FIGURE 17. Transimpedance Feedback Model
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18
Using feedback theory, F = VA/VOUT, this becomes a voltage
divider giving the following equation:
After a bit of algebraic manipulation the above equation re-
duces to:
The noise gain is 1/F. Because this is a differentiator circuit,
a zero must be inserted. The location of the zero is given by:
In the above equation the only unknown is CF. In trying to
solve this equation the fourth power of CF must be dealt with.
An excel spread sheet with this equation can be used and all
the known values entered. Then through iteration, the value
of CF when both sides are equal will be found. That is the
correct value for CF and of course the closest standard value
is used for CF.
CF has been added for stability. The addition of this part adds
a pole to the circuit. The pole is located at:
Before moving to the lab, the transfer function of the tran-
simpedance amplifier must be found and the units must be in
Ohms.
To attain maximum bandwidth and still have good stability the
pole is to be located on the open loop gain curve which is A.
If additional compensation is required one can always in-
crease the value of CF, but this will also reduce the bandwidth
of the circuit. Therefore A = 1/F, or AF = 1. For A the equation
is:
The LMP7717 was evaluated for RF = 10 kΩ and 100 kΩ,
representing a somewhat lower gain configuration and with
the 100 kΩ feedback resistor a fairly high gain configuration.
The RF = 10 kΩ is covered first. Looking at the Input Common
Mode Capacitance vs. VCM chart for CCM for the operating
point selected CCM = 15 pF. Note that for split supplies VCM
=
2.5V, CIN = 22 pF and fGBW = 88 MHz. Solving for CF the cal-
culated value is 1.75 pF, so 1.8 pF is selected for use.
Checking the frequency of the pole finds that it is at 8.8 MHz,
which is right at the minimum gain recommended for this part.
Some over compensation was necessary for stability and the
final selected value for CF is 2.7 pF. This moves the pole to
5.9 MHz. Figure 18 and Figure 19 show the rise and fall times
obtained in the lab with a 1V output swing. The laser diode
was difficult to drive due to thermal effects making the starting
and ending point of the pulse quite different, therefore the two
separate scope pictures.
The expression fGBW is the gain bandwidth product of the part.
For a unity gain stable part this is the frequency where A = 1.
For the LMP7717 fGBW = 88 MHz. Multiplying A and F results
in the following equation:
For the above equation s = jω. To find the actual amplitude of
the equation the square root of the square of the real and
imaginary parts are calculated. At the intersection of F and A,
we have:
30010894
FIGURE 18. Fall Time
19
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pole is at 2.5 MHz. Figure 20 shows the response for a 1V
output.
30010895
FIGURE 19. Rise Time
30010896
In Figure 18 the ringing and the hump during the on time is
from the laser. The higher drive levels for the laser gave ring-
ing in the light source as well as light changing from the
thermal characteristics. The hump is due to the thermal char-
acteristics.
FIGURE 20. High Gain Response
A transimpedance amplifier is an excellent application for the
LMP7717. Even with the high gain using a 100 kΩ feedback
resistor, the bandwidth is still well over 1 MHz. Other than a
little over compensation for the 10 kΩ feedback resistor con-
figuration using the LMP7717 was quite easy. Of course a
very good board layout was also used for this test.
Solving for CF using a 100 kΩ feedback resistor, the calcu-
lated value is 0.54 pF. One of the problems with more gain is
the very small value for CF. A 0.5 pF capacitor was used, its
measured value being 0.64 pF. For the 0.64 pF location the
www.national.com
20
Physical Dimensions inches (millimeters) unless otherwise noted
5-Pin SOT23
NS Package Number MF05A
8-Pin SOIC
NS Package Number M08A
21
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8-Pin MSOP
NS Package Number MUA08A
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22
23
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