NE5210 [NXP]
Transimpedance amplifier 280MHz; 跨阻放大器280MHz
| 型号： | NE5210 |
| 厂家： | 
NXP |
| 描述： | Transimpedance amplifier 280MHz |
| 文件： | 总14页 (文件大小：176K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
DESCRIPTION
PIN CONFIGURATION
The NE5210 is a 7kΩ transimpedance wide band, low noise
amplifier with differential outputs, particularly suitable for signal
recovery in fiber-optic receivers. The part is ideally suited for many
other RF applications as a general purpose gain block.
D Package
1
2
3
4
5
6
7
14 OUT (–)
GND
GND
2
2
13
12
11
10
9
GND
2
NC
OUT (+)
FEATURES
• Low noise: 3.5pA/√Hz
GND
1
I
IN
• Single 5V supply
NC
GND
1
• Large bandwidth: 280MHz
• Differential outputs
V
V
GND
1
CC1
CC2
8
GND
1
• Low input/output impedances
• High power supply rejection ratio
• High overload threshold current
• Wide dynamic range
TOP VIEW
SD00318
• Wideband gain block
• Medical and scientific instrumentation
• Sensor preamplifiers
• 7kΩ differential transresistance
• Single-ended to differential conversion
• Low noise RF amplifiers
APPLICATIONS
• Fiber-optic receivers, analog and digital
• RF signal processing
• Current-to-voltage converters
ORDERING INFORMATION
DESCRIPTION
TEMPERATURE RANGE
ORDER CODE
DWG #
14-Pin Plastic Small Outline (SO) Package
0 to +70°C
NE5210D
SOT108-1
ABSOLUTE MAXIMUM RATINGS
SYMBOL
PARAMETER
RATING
6
UNIT
V
V
CC
Power supply
T
Operating ambient temperature range
Operating junction temperature range
0 to +70
-55 to +150
-65 to +150
1.0
°C
A
T
°C
J
T
STG
Storage temperature range
°C
1
P
Power dissipation, T =25°C (still air)
W
DMAX
INMAX
A
2
I
Maximum input current
5
mA
NOTES:
1. Maximum dissipation is determined by the operating ambient temperature and the thermal resistance: θ =125°C/W.
JA
2. The use of a pull-up resistor to V for the PIN diode, is recommended.
CC
RECOMMENDED OPERATING CONDITIONS
SYMBOL
PARAMETER
RATING
4.5 to 5.5
0 to +70
0 to +90
UNIT
V
V
CC
Supply voltage
T
A
Ambient temperature range
Junction temperature range
°C
T
J
°C
1
1995 Apr 26
853-1654 15170
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
DC ELECTRICAL CHARACTERISTICS
Min and Max limits apply over operating temperature range at V =5V, unless otherwise specified. Typical data applies at V =5V and
CC
CC
T =25°C.
A
LIMITS
Typ
0.8
3.3
0
SYMBOL
PARAMETER
Input bias voltage
TEST CONDITIONS
UNIT
Min
0.6
2.8
Max
0.95
3.7
80
V
V
V
V
IN
Output bias voltage
V
±
O
Output offset voltage
Supply current
mV
mA
mA
µA
OS
I
I
I
21
3
26
32
CC
1
Output sink/source current
Input current (2% linearity)
4
OMAX
IN
Test Circuit 8, Procedure 2
Test Circuit 8, Procedure 4
±120
±160
Maximum input current
overload threshold
I
±160
±240
µA
INMAX
NOTES:
1. Test condition: output quiescent voltage variation is less than 100mV for 3mA load current.
AC ELECTRICAL CHARACTERISTICS
Typical data and Min/Max limits apply at V =5V and T =25°C.
CC
A
LIMITS
Typ
SYMBOL
PARAMETER
Transresistance
TEST CONDITIONS
UNIT
kΩ
Ω
Min
Max
DC tested, R =∞
L
R
R
R
R
4.9
7
10
T
O
T
(differential output)
Test Circuit 8, Procedure 1
Output resistance
(differential output)
DC tested
16
30
3.5
15
42
5
Transresistance
(single-ended output)
DC tested, R =∞
2.45
kΩ
Ω
L
Output resistance
(single-ended output)
DC tested
8
21
O
f
Bandwidth (-3dB)
Input resistance
Input capacitance
Test Circuit 1, T =25°C
200
280
60
MHz
Ω
3dB
A
R
C
IN
IN
7.5
pF
Transresistance power
supply sensitivity
∆R/∆V
∆R/∆T
V
=5±0.5V
9.6
0.05
3.5
20
0.1
6
%/V
%/°C
CC
Transresistance ambient
temperature sensitivity
∆T =T
A
-T
A MAX A MIN
RMS noise current spectral density
(referred to input)
f=10MHz, T =25°C
A
I
N
pA/√Hz
Test Circuit 2
T =25°C
A
Test Circuit 2
∆f=100MHz
∆f=200MHz
∆f=300MHz
∆f=100MHz
∆f=200MHz
∆f=300MHz
Integrated RMS noise current over
the bandwidth (referred to input)
37
56
71
40
66
89
1
C =0
S
I
T
nA
C =1pF
S
2
2
2
2
Power supply rejection ratio
DC tested, ∆V =0.1V
Equivalent AC test circuit 3
CC
PSRR
PSRR
PSRR
PSRR
20
20
36
36
65
23
dB
dB
dB
dB
(V
CC1
=V
CC2
)
Power supply rejection ratio
(V
DC tested, ∆V =0.1V
CC
)
Equivalent AC test circuit 4
CC1
Power supply rejection ratio
(V
DC tested, ∆V =0.1V
CC
)
Equivalent AC test circuit 5
CC2
Power supply rejection ratio (ECL
configuration)
f=0.1MHz, Test Circuit 6
2
1995 Apr 26
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
AC ELECTRICAL CHARACTERISTICS (Continued)
LIMITS
Typ
SYMBOL
PARAMETER
TEST CONDITIONS
UNIT
Min
Max
Maximum output voltage swing dif-
ferential
R =∞
L
V
OMAX
V
INMAX
2.4
3.2
V
P-P
Test Circuit 8, Procedure 3
Maximum input amplitude for
Test Circuit 7
650
mV
P-P
3
output duty cycle of 50±5%
Rise time for 50 mV
output signal
P-P
t
R
Test Circuit 7
0.8
1.2
ns
4
NOTES:
1. Package parasitic capacitance amounts to about 0.2pF
2. PSRR is output referenced and is circuit board layout dependent at higher frequencies. For best performance use RF filter in V line.
CC
3. Guaranteed by linearity and overload tests.
4. t defined as 20-80% rise time. It is guaranteed by a -3dB bandwidth test.
R
TEST CIRCUITS
SINGLE-ENDED
DIFFERENTIAL
NETWORK ANALYZER
V
V
OUT
OUT
R
R
[
R
+
2 @ S21 @ R
R
+
R + 4 @ S21 @ R
T
T
V
V
IN
IN
S-PARAMETER TEST SET
O Ť1 S22Ť *
)
O Ť1 S22Ť *
)
[ Z
33
R
+
2Z
66
O
O
1
*
S22
1 * S22
PORT 1
PORT 2
5V
V
V
CC2
CC1
0.1µF
0.1µF
Z
= 50
= 50
33
O
OUT
OUT
0.1µF
R = 1k
Z
= 50
O
IN DUT
33
R
50
L
GND
GND
1
2
Test Circuit 1
SPECTRUM ANALYZER
5V
A
= 60DB
V
V
V
CC2
CC1
0.1µF
0.1µF
Z
= 50
= 50
O
33
OUT
OUT
IN DUT
NC
33
R
L
GND
GND
1
2
Test Circuit 2
SD00319
3
1995 Apr 26
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TEST CIRCUITS (Continued)
NETWORK ANALYZER
5V
S-PARAMETER TEST SET
10µF
10µF
0.1µF
PORT 1
PORT 2
CURRENT PROBE
1mV/mA
0.1µF
16
CAL
V
V
CC1
CC2
0.1µF
0.1µF
33
33
OUT
50
TEST
100
BAL.
IN
TRANSFORMER
NH0300HB
UNBAL.
OUT
GND
GND
2
1
Test Circuit 3
NETWORK ANALYZER
5V
S-PARAMETER TEST SET
10µF
0.1µF
PORT 1
PORT 2
CURRENT PROBE
1mV/mA
10µF
10µF
0.1µF
16
CAL
5V
V
V
CC2
CC1
0.1µF
0.1µF
33
33
OUT
50
0.1µF
IN
TEST
100
BAL.
TRANSFORMER
NH0300HB
UNBAL.
OUT
GND
GND
2
1
Test Circuit 4
SD00320
4
1995 Apr 26
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TEST CIRCUITS (Continued)
NETWORK ANALYZER
5V
S-PARAMETER TEST SET
10µF
0.1µF
PORT 1
PORT 2
CURRENT PROBE
1mV/mA
10µF
10µF
0.1µF
16
CAL
5V
V
V
CC1
CC2
0.1µF
0.1µF
33
33
OUT
50
0.1µF
IN
TEST
100
BAL.
TRANSFORMER
NH0300HB
UNBAL.
OUT
GND
GND
2
1
Test Circuit 5
NETWORK ANALYZER
S-PARAMETER TEST SET
GND
PORT 1
PORT 2
CURRENT PROBE
1mV/mA
10µF
0.1µF
16
CAL
GND
GND
1
2
0.1µF
0.1µF
33
33
OUT
50
TEST
100
BAL.
IN
TRANSFORMER
NH0300HB
UNBAL.
OUT
V
V
CC1
CC2
5.2V
10µF
0.1µF
Test Circuit 6
SD00321
5
1995 Apr 26
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TEST CIRCUITS (Continued)
PULSE GEN.
V
V
CC2
CC1
0.1µF
0.1µF
33
33
OUT
OUT
A
B
Z
= 50Ω
0.1µF
IN
O
1k
DUT
OSCILLOSCOPE
= 50Ω
Z
O
50
Measurement done using
differential wave forms
GND
GND
2
1
Test Circuit 7
SD00322
6
1995 Apr 26
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TEST CIRCUITS (Continued)
Typical Differential Output Voltage
vs Current Input
5V
+
–
OUT +
V
(V)
OUT
IN
DUT
OUT –
I
(µA)
IN
GND
GND
2
1
2.00
1.60
1.20
0.80
0.40
0.00
–0.40
–0.80
–1.20
–1.60
–2.00
–400
–320
–240
–160
–80
0
80
160
240
320
400
CURRENT INPUT (µA)
NE5210 TEST CONDITIONS
Procedure 1
R
R
measured at 60µA
T
T
= (V
O1
– V )/(+60µA – (–60µA))
O2
Where: V
Measured at I = +60µA
O1
IN
V
Measured at I = –60µA
O2
IN
Procedure 2
Linearity = 1 – ABS((V
– V
OB
) / (V
O3
– V ))
O4
OA
Where: V
Measured at I = +120µA
O3
IN
V
Measured at I = –120µA
O4
IN
V
+ R @ () 120mA) ) V
OA
T
OB
OB
V
+ R @ (* 120mA) ) V
OB
= V
T
Procedure 3
Procedure 4
V
– V
OMAX
Where: V
O7
O8
Measured at I = +260µA
O7
IN
V
Measured at I = –260µA
O8
IN
I
Test Pass Conditions:
IN
V
– V
O5
> 20mV and V – V > 20mV
06 O5
O7
Where: V
Measured at I = +160µA
O5
IN
V
Measured at I = –160µA
O6
O7
O8
IN
V
Measured at I = +260µA
IN
V
Measured at I = –260µA
IN
SD00323
Test Circuit 8
7
1995 Apr 26
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TYPICAL PERFORMANCE CHARACTERISTICS
NE5210 Supply Current
vs Temperature
NE5210 Output Bias Voltage
Output Voltage
vs Input Current
vs Temperature
32
4.5
3.0
3.50
3.46
3.42
3.38
3.34
3.30
+85°C
–55°C
V
= 5.0V
+25°C
30
28
26
24
22
20
18
CC
+125°C
PIN 14
PIN 12
+125°C
+85°C
2.5
–300.0
–10
0
10 20 30 40 50 60 70 80
–10
0
10 20 30 40 50 60 70 80
0
+300.0
INPUT CURRENT (µA)
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
NE5210 Input Bias Voltage
vs Temperature
NE5210 Output Bias Voltage
vs Temperature
Differential Output Voltage
vs Input Current
4.1
900
2.0
5.5V
PIN 14
5.5V
5.5V
3.9
3.7
3.5
3.3
3.1
2.9
2.7
5.0V
5.0V
850
4.5V
4.5V
5.0V
4.5V
0
800
4.5V
750
700
5.0V
5.5V
–2.0
–300.0
–10
0
10 20 30 40 50 60 70 80
0
+300.0
–10
0
10 20 30 40 50 60 70 80
INPUT CURRENT (µA)
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
NE5210 Output Offset Voltage
vs Temperature
NE5210 Differential Output Swing
vs Temperature
Differential Output Voltage
vs Input Current
20
4.0
2.0
DC TESTED
3.8
V
= V
OUT12
– V
OUT14
OS
R
= ∞
L
0
–20
–40
3.6
5.5V
4.5V
3.4
3.2
5.0V
0
5.0V
5.5V
3.0
2.8
2.6
4.5V
–55°C
+25°C
+85°C
+125°C
–60
–80
2.4
2.2
–2.0
–300.0
–10
0
10 20 30 40 50 60 70 80
0
+300.0
–10
0
10 20 30 40 50 60 70 80
INPUT CURRENT (µA)
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
SD00324
8
1995 Apr 26
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TYPICAL PERFORMANCE CHARACTERISTICS (Continued)
NE5210 Differential Transresistance
vs Temperature
Gain vs Frequency
Gain vs Frequency
8.6
8
7
8
7
PIN 12
PIN 12
= 5V
R
= ∞
L
V
R
= 5V
= 50Ω
V
CC
5.5V
4.5V
5.5V
4.5V
CC
L
8.4
8.2
8.0
7.8
7.6
7.4
6
6
R
= 50Ω
L
5
5
4
4
5.0V
5.0V
3
3
5.5V
2
2
5.0V
4.5V
1
1
0
0
–1
–1
1
10
100
1000
1
10
100
1000
–10
0
10 20 30 40 50 60 70 80
FREQUENCY (MHz)
FREQUENCY (MHz)
AMBIENT TEMPERATURE (°C)
Gain vs Frequency
Gain vs Frequency
NE5210 Bandwidth vs Temperature
450
8
7
8
7
PIN 12
= 5V
PIN 12
PIN 14
= 5V
V
SINGLE-ENDED
V
CC
+125°C
–55°C
CC
400
6
6
5.5V
R
= Ω
L
5
–55°C
5
–55°C
350
300
250
200
4
4
5.0V
4.5V
3
3
+125°C
+85°C
25°C
85°C
25°C
2
2
1
1
0
+125°C
0
–1
–1
1
10
100
1000
1
10
100
1000
–10
0
10 20 30 40 50 60 70 80
FREQUENCY (MHz)
FREQUENCY (MHz)
AMBIENT TEMPERATURE (°C)
NE5210 Typical
Bandwidth Distribution
(70 Parts from 4 Wafer Lots)
Gain and Phase Shift
vs Frequency
Gain and Phase Shift
vs Frequency
8
7
6
5
4
3
2
1
0
180
90
8
360
270
180
90
50
40
30
20
10
0
PIN 12
PIN 12
SINGLE-ENDED
PIN 14
V
T
= 5.0V
7
6
CC
= 25°C
V
T
= 5V
CC
= 25°C
V
T
= 5V
CC
= 25°C
A
R
= 50Ω
L
A
A
5
0
4
3
–90
–180
2
1
0
0
–1
–1
1
10
100
1000
1
10
100
1000
223
255
287
319
351
383
FREQUENCY (MHz)
FREQUENCY (MHz)
FREQUENCY (MHz)
SD00325
9
1995 Apr 26
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TYPICAL PERFORMANCE CHARACTERISTICS (Continued)
NE5210 Output Resistance
vs Temperature
NE5210 Output Resistance
vs Temperature
NE5210 Output Resistance
vs Temperature
17
16
15
16
15
14
13
12
17
16
15
14
13
V
= 5.0V
PIN 12
OUTPUT REFERRED
PIN 14
OUTPUT REFERRED
CC
DC TESTED
PIN 14 R
OUT
4.5V
5.0V
4.5V
5.0V
5.5V
5.0V
5.5V
14
5.0V
PIN 12 R
OUT
13
12
–10
0
10 20 30 40 50 60 70 80
–10
0
10 20 30 40 50 60 70 80
–10
0
10 20 30 40 50 60 70 80
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
Output Resistance
vs Frequency
NE5210 Power Supply Rejection Ratio
vs Temperature
Group Delay
40
10
8
V
= V = 5.0V
CC2
CC1
∆V = ±0.1V
V
= 5V
80
70
60
50
40
30
20
10
0
CC
39
38
37
36
35
34
33
V
= 5.0V
CC
CC
DC TESTED
OUTPUT REFERRED
T
= 25°C
A
6
T
= 25°C
A
4
2
PIN 12
0
PIN 14
100 200
0.1
1
10
FREQUENCY (MHz)
0.1 20 40 60 80 100 120 140 160 180 200
FREQUENCY (MHz)
–10
0
10 20 30 40 50 60 70 80
AMBIENT TEMPERATURE (°C)
Output Step Response
V
T
= 5V
CC
= 25°C
A
20mV/Div
0
2
4
6
8
10
(ns)
12
14
16
18
20
SD00326
10
1995 Apr 26
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
THEORY OF OPERATION
Transimpedance amplifiers have been widely used as the
preamplifier in fiber-optic receivers. The NE5210 is a wide
bandwidth (typically 280MHz) transimpedance amplifier designed
primarily for input currents requiring a large dynamic range, such as
those produced by a laser diode. The maximum input current before
output stage clipping occurs at typically 240µA. The NE5210 is a
bipolar transimpedance amplifier which is current driven at the input
and generates a differential voltage signal at the outputs. The
forward transfer function is therefore a ratio of the differential output
voltage to a given input current with the dimensions of ohms. The
main feature of this amplifier is a wideband, low-noise input stage
which is desensitized to photodiode capacitance variations. When
connected to a photodiode of a few picoFarads, the frequency
response will not be degraded significantly. Except for the input
stage, the entire signal path is differential to provide improved
power-supply rejection and ease of interface to ECL type circuitry. A
block diagram of the circuit is shown in Figure 1. The input stage
(A1) employs shunt-series feedback to stabilize the current gain of
the amplifier. The transresistance of the amplifier from the current
OUTPUT +
A3
INPUT
A1
A2
R
F
A4
OUTPUT –
SD00327
Figure 1. NE5210 – Block Diagram
BANDWIDTH CALCULATIONS
The input stage, shown in Figure 3, employs shunt-series feedback
to stabilize the current gain of the amplifier. A simplified analysis can
determine the performance of the amplifier. The equivalent input
source to the emitter of Q is approximately the value of the
3
capacitance, C , in
feedback resistor, R =3.6kΩ. The gain from the second stage (A2)
IN
F
parallel with the source, I , is approximately 7.5pF, assuming that
and emitter followers (A3 and A4) is about two. Therefore, the
S
C =0 where C is the external source capacitance.
differential transresistance of the entire amplifier, R is
S
S
T
V
OUT(diff)
IIN
Since the input is driven by a current source the input must have a
RT
+
+ 2RF + 2(3.6K) + 7.2k
low input resistance. The input resistance, R , is the ratio of the
IN
incremental input voltage, V , to the corresponding input current, I
IN
IN
The single-ended transresistance of the amplifier is typically 3.6kΩ.
and can be calculated as:
The simplified schematic in Figure 2 shows how an input current is
converted to a differential output voltage. The amplifier has a single
input for current which is referenced to Ground 1. An input current
from a laser diode, for example, will be converted into a voltage by
VIN
IIN
RF
3.6K
71
RIN
+
+
+
+ 51
1 ) AVOL
More exact calculations would yield a higher value of 60Ω.
the feedback resistor R . The transistor Q1 provides most of the
F
Thus C and R will form the dominant pole of the entire amplifier;
IN
IN
open loop gain of the circuit, A
≈70. The emitter follower Q
2
VOL
1
minimizes loading on Q . The transistor Q , resistor R , and V
B1
f*3dB
+
1
4
7
2
RIN CIN
provide level shifting and interface with the Q – Q differential
15
16
pair of the second stage which is biased with an internal reference,
. The differential outputs are derived from emitter followers Q
Assuming typical values for R = 3.6kΩ, R = 60Ω, C = 7.5pF
F
IN
IN
V
B2
–
11
1
f*3dB
+
+ 354MHz
Q
Q
which are biased by constant current sources. The collectors of
12
2
7.5pF 60
– Q are bonded to an external pin, V
, in order to reduce
11
12
CC2
the feedback to the input stage. The output impedance is about 17Ω
single-ended. For ease of performance evaluation, a 33Ω resistor is
used in series with each output to match to a 50Ω test system.
V
CC1
R
V
CC2
R
R
R
1
3
12
13
Q
Q
Q
2
4
11
INPUT
+
Q
Q
12
3
Q
1
Q
Q
16
OUT–
OUT+
15
R
R
2
R
14
15
GND
1
R
+
7
PHOTODIODE
VB2
R
5
R
4
GND
2
SD00328
Figure 2. Transimpedance Amplifier
11
1995 Apr 26
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
For a given wavelength λ; (meters)
V
hc
CC
Energy of one Photon =
watt sec (Joule)
I
R3
-34
C1
R1
Where h=Planck’s Constant = 6.6 × 10 Joule sec.
c = speed of light = 3 × 10 m/sec
c / λ = optical frequency (Hz)
8
INPUT
Q2
I
B
I
Q3
IN
No. of incident photons/sec= where P=optical incident power
Q1
R2
P
hs
I
V
F
EQ3
No. of incident photons/sec =
V
IN
where P = optical incident power
P
R
F
R4
hs
No. of generated electrons/sec =
@
SD00329
where η = quantum efficiency
no. of generated electron hole paris
no. of incident photons
Figure 3. Shunt-Series Input Stage
+
The operating point of Q1, Figure 2, has been optimized for the
lowest current noise without introducing a second dominant pole in
the pass-band. All poles associated with subsequent stages have
been kept at sufficiently high enough frequencies to yield an overall
single pole response. Although wider bandwidths have been
achieved by using a cascode input stage configuration, the present
solution has the advantage of a very uniform, highly desensitized
frequency response because the Miller effect dominates over the
external photodiode and stray capacitances. For example, assuming
P
hs
I +
@
@ e Amps (Coulombs sec.)
-19
where e = electron charge = 1.6 × 10 Coulombs
@e
hs
Responsivity R =
Amp/watt
I + P @ R
a source capacitance of 1pF, input stage voltage gain of 70, R
=
IN
Assuming a data rate of 400 Mbaud (Bandwidth, B=200MHz), the
noise parameter Z may be calculated as:
60Ω then the total input capacitance, C = (1+7.5) pF which will
IN
1
lead to only a 12% bandwidth reduction.
IEQ
qB
66 @ 10*9
(1.6 @ 10*19)(200 @ 106)
Z +
+
+ 2063
NOISE
where Z is the ratio of
noise output to the peak response to a
RMS
Most of the currently installed fiber-optic systems use non-coherent
transmission and detect incident optical power. Therefore, receiver
noise performance becomes very important. The input stage
achieves a low input referred noise current (spectral density) of
3.5pA/√Hz. The transresistance configuration assures that the
external high value bias resistors often required for photodiode
biasing will not contribute to the total noise system noise. The
single hole-electron pair. Assuming 100% photodetector quantum
efficiency, half mark/half space digital transmission, 850nm
lightwave and using Gaussian approximation, the minimum required
-9
optical power to achieve 10 BER is:
hc
P
avMIN + 12 B Z + 12 2.3 @ 10*19
equivalent input
quiescent current of Q , the feedback resistor R , and the
bandwidth; however, it is not dependent upon the internal
Miller-capacitance. The measured wideband noise was 66nA
a 200MHz bandwidth.
noise current is strongly determined by the
RMS
200 @ 106 2063
1
F
+ 1139nW + * 29.4dBm
in
RMS
where h is Planck’s Constant, c is the speed of light, λ is the
wavelength. The minimum input current to the NE5210, at this input
power is:
DYNAMIC RANGE CALCULATIONS
The electrical dynamic range can be defined as the ratio of
maximum input current to the peak noise current:
IavMIN + qP
avMIN hc
1139 @ 10*9 @ 1.6 @ 10*19
+
2.3 @ 10*19
Electrical dynamic range, D , in a 200MHz bandwidth assuming
E
I
= 240µA and a wideband noise of I =66nA
for an
= 792nA
INMAX
EQ
RMS
external source capacitance of C = 1pF.
S
Choosing the maximum peak overload current of I
maximum mean optical power is:
=240µA, the
avMAX
(Max. input current) (PK)
DE + 20log
(Peak noise current) (RMS) @ 2
hcIavMAX
q
2.3 @ 10*19
1.6 @ 10*19
PavMAX
+
+
240 @ 10*6
(240 @ 10*6
( 2 66 10*9
)
+ 20log
+ 68dB
)
Thus the optical dynamic range, D is:
O
In order to calculate the optical dynamic range the incident optical
power must be considered.
D
= P
- P
= -4.6 -(-29.4) = 24.8dB.
O
avMAX
avMIN
12
1995 Apr 26
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
This represents the maximum limit attainable with the NE5210
operating at 200MHz bandwidth, with a half mark/half space digital
transmission at 850nm wavelength.
quiescent values of 3.3V (for a 5V supply), then the circuit may be
oscillating. Input pin layout necessitates that the photodiode be
physically very close to the input and Ground 1. Connecting Pins 3
and 5 to Ground 1 will tend to shield the input but it will also tend to
increase the capacitance on the input and slightly reduce the
bandwidth.
APPLICATION INFORMATION
Package parasitics, particularly ground lead inductances and
parasitic capacitances, can significantly degrade the frequency
response. Since the NE5210 has differential outputs which can feed
back signals to the input by parasitic package or board layout
capacitances, both peaking and attenuating type frequency
response shaping is possible. Constructing the board layout so that
Ground 1 and Ground 2 have very low impedance paths has
produced the best results. This was accomplished by adding a
ground-plane stripe underneath the device connecting Ground 1,
Pins 8–11, and Ground 2, Pins 1 and 2 on opposite ends of the
SO14 package. This ground-plane stripe also provides isolation
As with any high-frequency device, some precautions must be
observed in order to enjoy reliable performance. The first of these is
the use of a well-regulated power supply. The supply must be
capable of providing varying amounts of current without significantly
changing the voltage level. Proper supply bypassing requires that a
good quality 0.1µF high-frequency capacitor be inserted between
V
CC1
and V
, preferably a chip capacitor, as close to the package
CC2
pins as possible. Also, the parallel combination of 0.1µF capacitors
with 10µF tantalum capacitors from each supply, V and V , to
CC1
CC2
the ground plane should provide adequate decoupling. Some
applications may require an RF choke in series with the power
supply line. Separate analog and digital ground leads must be
maintained and printed circuit board ground plane should be
employed whenever possible.
between the output return currents flowing to either V
or Ground
CC2
2 and the input photodiode currents to flowing to Ground 1. Without
this ground-plane stripe and with large lead inductances on the
board, the part may be unstable and oscillate near 800MHz. The
easiest way to realize that the part is not functioning normally is to
measure the DC voltages at the outputs. If they are not close to their
Figure 4 depicts a 50Mb/s TTL fiber-optic receiver using the BPF31,
850nm LED, the NE5210 and the NE5214 post amplifier.
+V
CC
GND
47µF
C1
C2
.01µF
L1
10µH
C5
1.0µF
R1
100
D1
LED
R2
220
C7
LED
C
GND
1
20
19
V
V
IN
IN
8
9
7
6
CC
1B
1A
100pF
C8
C9
C3
10µF
.01µF
C4
.01µF
2
PKDET
GND
GND
GND
CC
NC
100pF
THRESH
3
18
10
11
5
4
C
C6
AZP
AZN
GND
A
I
IN
4
5
17
16
C
0.1µF
BPF31
OPTICAL
INPUT
R3
47k
FLAG
JAM
NC
OUT
OUT
12
13
3
2
1B
L2
10µH
6
7
15
14
GND
GND
GND
OUT
IN
8B
V
CCD
OUT
14
1
1A
C11
C10
V
CCA
8
13
12
10µF
.01µF
IN
R
8A
GND
D
9
HYST
R
TTL
10
11
C12
10µF
PKDET
OUT
L3
10µH
C13
.01µF
R4
4k
V
(TTL)
OUT
NOTE:
The NE5210/NE5217 combination can operate at data rates in excess of 100Mb/s NRZ
The capacitor C7 decreases the NE5210 bandwidth to improve overall S/N ratio in the DC–50MHz band, but does create extra high frequency noise
on the NE5210 V pin(s).
CC
SD00330
Figure 4. A 50Mb/s Fiber Optic Receiver
13
1995 Apr 26
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
1
14
OUT (–)
GND 2
2
13
GND 2
GND 2
12
3
OUT (+)
NC
GND 1
11
4
INPUT
NC
10
GND 1
5
GND 1
VCC1
9
6
ECN No.: 06027
1992 Mar 13
GND 1
7
8
VCC 2
SD00488
Figure 5. NE5210 Bonding Diagram
carriers, it is impossible to guarantee 100% functionality through this
process. There is no post waffle pack testing performed on
individual die.
Die Sales Disclaimer
Due to the limitations in testing high frequency and other parameters
at the die level, and the fact that die electrical characteristics may
shift after packaging, die electrical parameters are not specified and
die are not guaranteed to meet electrical characteristics (including
temperature range) as noted in this data sheet which is intended
only to specify electrical characteristics for a packaged device.
Since Philips Semiconductors has no control of third party
procedures in the handling or packaging of die, Philips
Semiconductors assumes no liability for device functionality or
performance of the die or systems on any die sales.
All die are 100% functional with various parametrics tested at the
wafer level, at room temperature only (25°C), and are guaranteed to
be 100% functional as a result of electrical testing to the point of
wafer sawing only. Although the most modern processes are
utilized for wafer sawing and die pick and place into waffle pack
Although Philips Semiconductors typically realizes a yield of 85%
after assembling die into their respective packages, with care
customers should achieve a similar yield. However, for the reasons
stated above, Philips Semiconductors cannot guarantee this or any
other yield on any die sales.
14
1995 Apr 26
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