AD7450AR-REEL [ROCHESTER]
ADC, Successive Approximation, 12-Bit, 1 Func, 1 Channel, Serial Access, PDSO8, SOIC-8;
| 型号： | AD7450AR-REEL |
| 厂家： | 
Rochester Electronics |
| 描述： | ADC, Successive Approximation, 12-Bit, 1 Func, 1 Channel, Serial Access, PDSO8, SOIC-8 光电二极管 |
| 文件： | 总21页 (文件大小：1022K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Differential Input, 1 MSPS
a
12-Bit ADC in ꢀSOIC-8 and SO-8
AD7450
FEATURES
FUNCTIONAL BLOCK DIAGRAM
Fast Throughput Rate: 1 MSPS
Specified for VDD of 3 V and 5 V
Low Power at Max Throughput Rate:
3.75 mW Max at 833 kSPS with 3 V Supplies
9 mW Max at 1 MSPS with 5 V Supplies
Fully Differential Analog Input
Wide Input Bandwidth:
V
DD
V
IN+
12-BIT SUCCESSIVE
APPROXIMATION
ADC
T/H
V
IN–
70 dB SINAD at 300 kHz Input Frequency
Flexible Power/Serial Clock Speed Management
No Pipeline Delays
V
REF
High-Speed Serial Interface—SPITM/QSPITM
MICROWIRETM/DSP Compatible
Power-Down Mode: 1 ꢀA Max
8-Lead ꢀSOIC and SOIC Packages
SCLK
SDATA
CS
AD7450
CONTROL
LOGIC
APPLICATIONS
Transducer Interface
Battery-Powered Systems
Data Acquisition Systems
Portable Instrumentation
Motor Control
GND
Communications
GENERAL DESCRIPTION
The AD7450 is a 12-bit, high-speed, low power, successive
approximation (SAR) analog-to-digital converter that features a
fully differential analog input. It operates from a single 3 V or 5 V
power supply and features throughput rates up to 833 kSPS or
1 MSPS, respectively.
The AD7450 uses advanced design techniques to achieve low
power dissipation at high throughput rates.
PRODUCT HIGHLIGHTS
1. Operation with either 3 V or 5 V power supplies.
This part contains a low noise, wide bandwidth, differential track
and-hold amplifier (T/H) that can handle input frequencies in
excess of 1 MHz with the –3 dB point typically being 20 MHz.
-
2. High throughput with low power consumption. With a 3 V
supply, the AD7450 offers 3.75 mW max power consumption
for 833 kSPS throughput.
The reference voltage for the AD7450 is applied externally to the
VREF pin and can be varied from 100 mV to 3.5 V, depending
on the power supply and what suits the application. The value of
the reference voltage determines the common-mode voltage
range of the part. With this truly differential input structure and
variable reference input, the user can select a variety of input
ranges and bias points.
3. Fully differential analog input.
4. Flexible power/serial clock speed management. The conversion
rate is determined by the serial clock, allowing the power
to be reduced as the conversion time is reduced through
the serial clock speed increase. This part also features a
shutdown mode to maximize power efficiency at lower
throughput rates.
The conversion and data acquisition processes are controlled
using CS and the serial clock, allowing the device to interface
with microprocessors or DSPs. The input signals are sampled
on the falling edge of CS, and the conversion is also initiated at
this point.
5. Variable voltage reference input.
6. No pipeline delay.
7. Accurate control of the sampling instant via a CS input and
once-off conversion control.
The SAR architecture of this part ensures that there are no
pipeline delays.
8. ENOB > 8 bits typically with 100 mV reference.
SPI and QSPI are trademarks of Motorola, Inc.
MICROWIRE is a trademark of National Semiconductor Corporation.
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, norforanyinfringementsofpatentsorotherrightsofthirdpartiesthat
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
Fax: 781/326-8703
www.analog.com
© Analog Devices, Inc., 2002
1
AD7450–SPECIFICATIONS
(VDD = 2.7 V to 3.3 V, fSCLK = 15 MHz, fS = 833 kSPS, VREF = 1.25 V, FIN = 200 kHz;
VDD = 4.75 V to 5.25 V, fSCLK = 18 MHz, fS = 1 MSPS, VREF = 2.5 V, FIN = 300 kHz; VCM2 = VREF; TA = TMIN to TMAX, unless otherwise noted.)
Parameter
Conditions/Comments
A Version
B Version
Unit
DYNAMIC PERFORMANCE
Signal-to-(Noise + Distortion) Ratio
(SINAD)3
VDD = 5 V
VDD = 3 V
VDD = 5 V, –80 dB typ
70
68
–75
–73
–75
–73
70
68
–75
–73
–75
–73
dB min
dB min
dB max
dB max
dB max
dB max
Total Harmonic Distortion (THD)3
V
DD = 3 V, –78 dB typ
Peak Harmonic or Spurious Noise3
VDD = 5 V, –82 dB typ
VDD = 3 V, –80 dB typ
Intermodulation Distortion (IMD)3
Second Order Terms
Third Order Terms
–85
–85
10
50
20
–85
–85
10
50
20
dB typ
dB typ
ns typ
ps typ
MHz typ
MHz typ
Aperture Delay3
Aperture Jitter3
Full Power Bandwidth3
@ –3 dB
@ –0.1 dB
2.5
2.5
Power Supply Rejection Ratio
(PSRR)3, 4
–87
–87
dB typ
DC ACCURACY
Resolution
12
2
12
1
Bits
LSB max
Integral Nonlinearity (INL)3
Differential Nonlinearity (DNL)3
Guaranteed No Missed
Codes to 12 Bits
VDD = 5 V
–1/+2
1
3
6
3
6
3
6
LSB max
LSB max
LSB max
LSB max
LSB max
LSB max
LSB max
Zero Code Error3
3
6
3
6
3
6
V
DD = 3 V
Positive Gain Error3
Negative Gain Error3
VDD = 5 V
VDD = 3 V
VDD = 5 V
VDD = 3 V
ANALOG INPUT
Full-Scale Input Span
Absolute Input Voltage
VIN+
5
2 ϫ VREF
VIN+ – VIN–
VIN+ – VIN–
V
VCM2 = VREF
VCM2 = VREF
VCM VREF/2
VCM VREF/2
VCM VREF/2
VCM VREF/2
V
V
VIN–
DC Leakage Current
Input Capacitance
1
20
6
1
20
6
µA max
pF typ
pF typ
When in Track
When in Hold
REFERENCE INPUT
VREF Input Voltage
5 V supply ( 1% tolerance for
specified performance)
2.56
2.56
V
3 V supply ( 1% tolerance for
specified performance)
1.257
1
15
1.257
1
15
V
DC Leakage Current
VREF Input Capacitance
µA max
pF typ
LOGIC INPUTS
Input High Voltage, VINH
Input Low Voltage, VINL
Input Current, IIN
2.4
0.8
1
2.4
0.8
1
V min
V max
µA max
pF max
Typically 10 nA, VIN = 0 V or VDD
8
Input Capacitance, CIN
10
10
LOGIC OUTPUTS
Output High Voltage, VOH
VDD = 5 V, ISOURCE = 200 µA
VDD = 3 V, ISOURCE = 200 µA
ISINK = 200 µA
2.8
2.4
0.4
1
10
Two’s
2.8
2.4
0.4
1
10
Two’s
V min
V min
V max
µA max
pF max
Output Low Voltage, VOL
Floating-State Leakage Current
Floating-State Output Capacitance8
Output Coding
Complement
Complement
–2–
REV. 0
AD7450
Parameter
Conditions/Comments
A Version
B Version
Unit
CONVERSION RATE
Conversion Time
888 ns with an 18 MHz SCLK
1.07 µs with a 15 MHz SCLK
Sine Wave Input
16
16
SCLK Cycles
ns max
Track-and-Hold
200
200
Acquisition Time3, 8
Throughput Rate9
VDD = 5 V
VDD = 3 V
1
833
1
833
MSPS max
kSPS max
POWER REQUIREMENTS
VDD
IDD
Range: 3 V 10%; 5 V 5%
3/5
3/5
V min/max
10, 11
Normal Mode (Static)
Normal Mode (Operational)
VDD = 3 V/5 V SCLK; ON or OFF
VDD = 5 V; fSAMPLE = 1 MSPS
VDD = 3 V; fSAMPLE = 833 kSPS
SCLK ON or OFF
0.5
1.8
1.25
1
0.5
1.8
1.25
1
mA typ
mA max
mA max
µA max
Full Power-Down Mode
Power Dissipation
Normal Mode (Operational)
VDD = 5 V; fSAMPLE = 1 MSPS;
1.38 mW typ for 100 KSPS10
VDD = 3 V; fSAMPLE = 833 kSPS;
0.53 mW typ for 100 KSPS10
VDD = 5 V; SCLK ON or OFF
VDD = 3 V; SCLK ON or OFF
9
9
mW max
mW max
3.75
3.75
Full Power-Down Mode
5
3
5
3
µW max
µW max
NOTES
1Temperature range is as follows: A and B Versions: –40°C to +85°C.
2Common-mode voltage. The input signal can be centered on any choice of dc common-mode voltage as long as this value is in the range specified in Figures 8 and 9.
3See Terminology section.
4A 200 mV p-p sine wave, varying in frequency from 1 kHz to 200 kHz is coupled onto VDD. A 2.2 nF capacitor is used to decouple VDD to GND.
5If the input spans of VIN+ and VIN– are both VREF, and they are 180° out of phase, the differential voltage is 2 ϫ VREF
.
6The AD7450 is functional with a reference input from 100 mV and for VDD = 5 V, the reference can range up to 3.5 V (see References section).
7The AD7450 is functional with a reference input from 100 mV and for VDD = 3 V, the reference can range up to 2.2 V (see References section).
8Sample tested @ 25°C to ensure compliance.
9See Serial Interface section.
10See Power Versus Throughput Rate section.
11Measured with a midscale dc input.
REV. 0
–3–
AD7450
TIMING SPECIFICATIONS1, 2
(VDD = 2.7 V to 3.3 V, fSCLK = 15 MHz, fS = 833 kSPS, VREF = 1.25 V; VDD = 4.75 V to 5.25 V,
fSCLK = 18 MHz, fS = 1 MSPS, VREF = 2.5 V; VCM3 = VREF; TA = TMIN to TMAX, unless otherwise noted.)
Limit at TMIN, TMAX
Parameter
3 V
5 V
Unit
Description
4
fSCLK
50
50
kHz min
15
18
MHz max
tCONVERT
tQUIET
16 ϫ tSCLK
1.07
25
16 ϫ tSCLK
0.88
25
tSCLK = 1/fSCLK
SCLK = 15 MHz, 18 MHz
Minimum Quiet Time between the End of a Serial Read and the Next
Falling Edge of CS
µs max
ns min
t1
10
10
20
40
0.4 tSCLK
0.4 tSCLK
10
10
35
1
10
10
20
40
0.4 tSCLK
0.4 tSCLK
10
10
35
1
ns min
ns min
ns max
ns max
ns min
ns min
ns min
ns min
ns max
µs max
Minimum CS Pulsewidth
t25
t35
t4
t5
t6
CS Falling Edge to SCLK Falling Edge Setup Time
Delay from CS Falling Edge until SDATA Three-State Disabled
Data Access Time after SCLK Falling Edge
SCLK High Pulsewidth
SCLK Low Pulsewidth
t76
SCLK Edge to Data Valid Hold Time
SCLK Falling Edge to SDATA Three-State Enabled
SCLK Falling Edge to SDATA Three-State Enabled
Power-Up Time from Full Power-Down
t8
7
tPOWER-UP
NOTES
1Sample tested at 25°C to ensure compliance. All input signals are specified with tr = tf = 5 ns (10% to 90% of VDD) and timed from a voltage level of 1.6 V.
2See Figure 1 and the Serial Interface section.
3Common-mode voltage.
4Mark/space ratio for the SCLK input is 40/60 to 60/40.
5Measured with the load circuit of Figure 2 and defined as the time required for the output to cross 0.8 V or 2.4 V with VDD = 5 V, and the time for an output to cross
0.4 V or 2.0 V for VDD = 3 V.
6t8 is derived from the measured time taken by the data outputs to change 0.5 V when loaded with the circuit of Figure 2. The measured number is then extrapolated
back to remove the effects of charging or discharging the 50 pF capacitor. This means that the time, t8, quoted in the timing characteristics is the true bus relinquish
time of the part and is independent of the bus loading.
7See Power-Up Time section.
Specifications subject to change without notice.
t1
CS
tCONVERT
t2
t5
SCLK
1
2
3
4
5
13
14
t6
15
16
t7
t8
tQUIET
t3
t4
DB0
SDATA
0
0
0
0
DB11
DB10
DB2
DB1
THREE-STATE
4 LEADING ZEROS
Figure 1. Serial Interface Timing Diagram
–4–
REV. 0
AD7450
ABSOLUTE MAXIMUM RATINGS1
Lead Temperature, Soldering
Vapor Phase (60 secs) . . . . . . . . . . . . . . . . . . . . . . . . 215oC
Infrared (15 secs) . . . . . . . . . . . . . . . . . . . . . . . . . . . . 220oC
ESD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.5 kV
(TA = 25°C, unless otherwise noted.)
VDD to GND . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V
VIN+ to GND . . . . . . . . . . . . . . . . . . . . –0.3 V to VDD + 0.3 V
VIN– to GND . . . . . . . . . . . . . . . . . . . . –0.3 V to VDD + 0.3 V
Digital Input Voltage to GND . . . . . . . . –0.3 V to VDD + 0.3 V
Digital Output Voltage to GND . . . . . –0.3 V to VDD + 0.3 V
VREF to GND . . . . . . . . . . . . . . . . . . . . –0.3 V to VDD + 0.3 V
NOTES
1Stresses above those listed under the Absolute Maximum Ratings may cause
permanent damage to the device. This is a stress rating only and functional
operation of the device at these or any conditions above those listed in the
operational sections of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
2Transient currents of up to 100 mA will not cause SCR latch up.
Input Current to Any Pin Except Supplies2 . . . . . . .
Operating Temperature Range
10 mA
Commercial (A and B Version) . . . . . . . . . –40oC to +85oC
Storage Temperature Range . . . . . . . . . . . . –65oC to +150oC
Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . 150oC
SOIC, µSOIC Package, Power Dissipation . . . . . . . . 450 mW
I
200ꢀA
OL
TO
OUTPUT
PIN
JA Thermal Impedance . . . . . . . . . . . . . . . . 157°C/W (SOIC)
. . . . . . . . . . . . . . . . . . . . . . . . . . . 205.9°C/W (µSOIC)
JC Thermal Impedance . . . . . . . . . . . . . . . . . 56°C/W (SOIC)
. . . . . . . . . . . . . . . . . . . . . . . . . . . 43.74°C/W (µSOIC)
1.6V
C
L
50pF
200ꢀA
I
OH
Figure 2. Load Circuit for Digital Output Timing
Specifications
ORDERING GUIDE
Temperature
Range
Linearity
Package
Option2
Branding
Information
Model
Error (LSB)1
AD7450AR
AD7450ARM
AD7450BR
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
Evaluation Board
Controller Board
2 LSB
2 LSB
1 LSB
1 LSB
SO-8
RM-8
SO-8
RM-8
AD7450AR
CPA
AD7450BR
CPB
AD7450BRM
EVAL-AD7450CB3
EVAL-CONTROL BRD24
NOTES
1Linearity error here refers to integral nonlinearity error.
2SO = SOIC; RM = µSOIC.
3This can be used as a standalone evaluation board or in conjunction with the Evaluation Board Controller for evaluation/demonstration purposes.
4Evaluation Board Controller. This board is a complete unit allowing a PC to control and communicate with all Analog Devices evaluation boards
ending in the CB designators. To order a complete evaluation kit, you will need to order the ADC evaluation board, i.e.. EVAL-AD7450CB, the
EVAL-CONTROL BRD2, and a 12 V ac transformer. See the AD7450 evaluation board technical note for more details.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although the
AD7450 features proprietary ESD protection circuitry, permanent damage may occur on devices
subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are recommended to
avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
REV. 0
–5–
AD7450
PIN CONFIGURATION
1
2
3
4
8
7
6
5
V
V
DD
REF
AD7450
SCLK
SDATA
CS
V
IN+
TOP VIEW
V
(Not to Scale)
IN–
GND
PIN FUNCTION DESCRIPTION
Pin Number
Mnemonic
Function
1
VREF
Reference Input for the AD7450. An external reference must be applied to this input. For a
5 V power supply, the reference is 2.5 V ( 1%), and for a 3 V power supply, the reference is
1.25 V ( 1%) for specified performance. This pin should be decoupled to GND with a
capacitor of at least 0.1 µF. See the References section for more details.
2
3
4
VIN+
VIN–
Positive Terminal for Differential Analog Input
Negative Terminal for Differential Analog Input
GND
Analog Ground. Ground reference point for all circuitry on the AD7450. All analog input
signals and any external reference signal should be referred to this GND voltage.
5
6
CS
Chip Select. Active low logic input. This input provides the dual function of initiating a
conversion on the AD7450 and framing the serial data transfer.
SDATA
Serial Data. Logic output. The conversion result from the AD7450 is provided on this
output as a serial data stream. The bits are clocked out on the falling edge of the SCLK
input. The data stream consists of four leading zeros followed by the 12 bits of conversion
data that is provided MSB first. The output coding is two’s complement.
7
8
SCLK
VDD
Serial Clock. Logic input. SCLK provides the serial clock for accessing data from the part.
This clock input is also used as the clock source for the AD7450’s conversion process.
Power Supply Input. VDD is 3 V ( 10%) or 5 V ( 5%). This supply should be decoupled to
GND with a 0.1 µF capacitor and a 10 µF tantalum capacitor.
–6–
REV. 0
AD7450
TERMINOLOGY
Aperture Jitter
Signal-to-(Noise + Distortion) Ratio
This is the sample-to-sample variation in the effective point in
time at which the actual sample is taken.
This is the measured ratio of signal-to-(noise + distortion) at
the output of the ADC. The signal is the rms amplitude of the
fundamental. Noise is the sum of all nonfundamental signals up
to half the sampling frequency (fS/2), excluding dc. The ratio is
dependent on the number of quantization levels in the digitiza-
tion process; the more levels, the smaller the quantization noise.
The theoretical signal-to-(noise + distortion) ratio for an ideal
N-bit converter with a sine wave input is given by:
Full Power Bandwidth
The full power bandwidth of an ADC is that input frequency at
which the amplitude of the reconstructed fundamental is reduced
by 0.1 dB or 3 dB for a full-scale input.
Common-Mode Rejection Ratio (CMRR)
The common-mode rejection ratio is defined as the ratio of the
power in the ADC output at full-scale frequency, f, to the power
of a 200 mV p-p sine wave applied to the common-mode volt-
age of VIN+ and VIN– of frequency fs:
Signal–to–(Noise + Distortion) = (6.02 N + 1.76) dB
Thus, for a 12-bit converter, this is 74 dB.
CMRR(dB) = 10 log (Pf/Pfs)
Total Harmonic Distortion
Total harmonic distortion (THD) is the ratio of the rms sum of
harmonics to the fundamental. For the AD7450, it is defined as:
Pf is the power at the frequency f in the ADC output; Pfs is the
power at frequency fs in the ADC output.
2
2
2
2
2
Integral Nonlinearity (INL)
This is the maximum deviation from a straight line passing
through the endpoints of the ADC transfer function.
V2 +V3 +V4 +V5 +V6
THD(dB) = 20 log
V1
where V1 is the rms amplitude of the fundamental and V2, V3,
V4, V5, and V6 are the rms amplitudes of the second to the sixth
harmonics.
Differential Nonlinearity (DNL)
This is the difference between the measured and the ideal 1 LSB
change between any two adjacent codes in the ADC.
Peak Harmonic or Spurious Noise
Zero Code Error
This is the deviation of the midscale code transition (111...111
to 000...000) from the ideal VIN+ – VIN– (i.e., 0 LSB).
Peak harmonic or spurious noise is defined as the ratio of the
rms value of the next largest component in the ADC output
spectrum (up to fS/2 and excluding dc) to the rms value of the
fundamental. Normally, the value of this specification is deter-
mined by the largest harmonic in the spectrum, but for ADCs
where the harmonics are buried in the noise floor, it will be a
noise peak.
Positive Gain Error
This is the deviation of the last code transition (011...110 to
011...111) from the ideal VIN+ – VIN– (i.e., +VREF – 1 LSB),
after the zero code error has been adjusted out.
Negative Gain Error
Intermodulation Distortion
This is the deviation of the first code transition (100...000 to
100...001) from the ideal VIN+ – VIN– (i.e., –VREF + 1 LSB), after
the zero code error has been adjusted out.
With inputs consisting of sine waves at two frequencies, fa and fb,
any active device with nonlinearities will create distortion
products at sum and difference frequencies of mfa nfb where
m and n = 0, 1, 2, or 3. Intermodulation distortion terms are those
for which neither m nor n are equal to zero. For example, the
second order terms include (fa + fb) and (fa – fb), while the third
order terms include (2fa + fb), (2fa – fb), (fa + 2fb), and (fa –2fb).
Track and Hold Acquisition Time
The track and hold acquisition time is the minimum time re-
quired for the track and hold amplifier to remain in track mode
for its output to reach and settle to within 0.5 LSB of the ap-
plied input signal.
The AD7450 is tested using the CCIF standard, where two
input frequencies near the top end of the input bandwidth are
used. In this case, the second order terms are usually distanced
in frequency from the original sine waves, while the third order
terms are usually at a frequency close to the input frequencies.
As a result, the second and third order terms are specified
separately. The calculation of the intermodulation distortion is
as per the THD specification, where it is the ratio of the rms sum
of the individual distortion products to the rms amplitude of the
sum of the fundamentals expressed in dBs.
Power Supply Rejection Ratio (PSRR)
The power supply rejection ratio is defined as the ratio of the
power in the ADC output at full-scale frequency, f, to the power
of a 200 mV p-p sine wave applied to the ADC VDD supply of
frequency fS.
PSRR (dB) = 10 log (Pf/Pfs)
Pf is the power at frequency f in the ADC output; Pfs is the
power at frequency fs in the ADC output.
Aperture Delay
This is the amount of time from the leading edge of the sampling
clock until the ADC actually takes the sample.
REV. 0
–7–
AD7450–Typical Performance Characteristics
(Default Conditions: TA = 25ꢁC)
0
0
–63
–65
–67
–69
–71
–73
–75
8192 POINT FFT
fSAMPLE = 833kSPS
fIN = 300kHz
8192 POINT FFT
fSAMPLE = 1MSPS
fIN = 300kHz
–20
–20
SINAD = 70.2dB
SINAD = 71.7dB
THD = –82dB
THD = –82.8dB
–40
–40
PK NOISE = –87.1dB
PK NOISE = –85.3dB
V
= 2.7V
DD
–60
–60
V
= 3.3V
DD
–80
–80
–100
–120
–100
–120
V
= 4.75V
V
= 5.25V
DD
DD
0
50
100 150 200 250 300 350
FREQUENCY – kHz
0
50 100 150 200 250 300 350 400 450 500
FREQUENCY – kHz
10
100
INPUT FREQUENCY – kHz
1000
TPC 1. Dynamic Performance at
1 MSPS with VDD = 5 V
TPC 3. SINAD vs. Analog Frequency
for Various Supply Voltages
TPC 2. Dynamic Performance at
833 kSPS with VDD = 3 V
1.0
0.8
1.0
0.8
1.0
0.8
0.6
0.6
0.6
0.4
0.4
0.4
0.2
0.2
0.2
0
0
0
–0.2
–0.4
–0.6
–0.8
–1.0
–0.2
–0.4
–0.6
–0.8
–1.0
–0.2
–0.4
–0.6
–0.8
–1.0
0
1024
2048
3072
4096
0
1024
2048
3072
4096
0
1024
2048
CODE
3072
4096
CODE
CODE
TPC 4. Typical Differential
Nonlinearity (DNL) VDD = 5 V
TPC 5. Typical Differential
Nonlinearity (DNL) VDD = 3 V
TPC 6. Typical Integral
Nonlinearity (INL) VDD = 5 V
1.5
1.0
1.0
0.8
1.0
0.6
POSITIVE DNL
NEGATIVE DNL
0.4
POSITIVE DNL
0.5
0
0.5
0.2
0
0
–0.2
–0.4
–0.6
–0.8
–1.0
–0.5
–1.0
–0.5
–1.0
NEGATIVE DNL
0
1024
2048
CODE
3072
4096
0
0.6
1.2
V
1.8
2.4
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
V
REF
REF
TPC 7. Typical Integral
Nonlinearity (INL) VDD = 3 V
TPC 9. Change in DNL vs. Reference
Voltage VDD = 3.3 V*
TPC 8. Change in DNL vs. Reference
Voltage VDD = 5 V
–8–
REV. 0
AD7450
1.5
1.0
1
0
V
= 5V
DD
2.0
1.5
fS = 1MSPS
–1
–2
–3
–4
–5
–6
–7
–8
–9
V
= 3.3V
POSITIVE INL
DD
fS = 833kSPS
0.5
1.0
POSITIVE INL
0.5
0
0
–0.5
–1.0
–1.5
–0.5
–1.0
–1.5
NEGATIVE INL
NEGATIVE INL
1.8
0.25 0.75 1.25 1.75 2.25 2.75 3.25 3.50
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
0
0.6
1.2
V
2.4
V
REF
V
REF
REF
TPC 12. Change in Zero-Code Error vs.
Reference Voltage VDD = 5 V and 3.3 V*
TPC 10. Change in INL vs. Reference
Voltage VDD = 5 V
TPC 11. Change in INL vs. Reference
Voltage VDD = 3.3 V*
12
11
10,000
10,000
10,000
9,839
9,000
9,000
CODES
CODES
V
= 5V
= 1MSPS
8,000
7,000
6,000
5,000
4,000
V
ꢂ =V
–
IN
8,000
7,000
6,000
5,000
4,000
V
ꢂ =V –
IN
DD
IN
IN
f
10,000 CONVERSIONS
= 1MSPS
10,000 CONVERSIONS
S
10
9
f
f
= 833kSPS
S
S
8
V
= 3.3V
= 833kSPS
DD
3,000
2,000
1,000
0
3,000
2,000
1,000
0
f
S
7
90
71
CODES
CODES
6
0
0.5
1.0
1.5
2.0
REF
2.5
3.0
3.5
2044
2045
2046
2047
2048
2049
2044
2045
2046
2047
2048
2049
V
CODE
CODE
TPC 15. Change in ENOB vs. Refer-
ence Voltage VDD = 5 V and 3.3 V*
TPC 13. Histogram of the Output
Codes with a DC Input for VDD = 5 V
TPC 14. Histogram of the Output
Codes with a DC Input for VDD = 3 V
90
V
= 5V
DD
80
70
60
50
40
30
20
10
0
V
= 3V
DD
10
100
1,000
10,000
FREQUENCY – kHz
TPC 16. CMRR vs. Input Frequency
for VDD = 5 V and 3 V
*See References section.
REV. 0
–9–
AD7450
CIRCUIT INFORMATION
CAPACITIVE
DAC
The AD7450 is a fast, low power, single-supply, 12-bit successive
approximation analog-to-digital converter (ADC). It can operate
with a 5 V and 3 V power supply and is capable of throughput
rates up to 1 MSPS and 833 kSPS when supplied with an
18 MHz or 15 MHz clock, respectively. This part requires an
external reference to be applied to the VREF pin, with the value
of the reference chosen depending on the power supply and
what suits the application.
COMPARATOR
+
B
C
C
S
V
IN+
A
A
CONTROL
LOGIC
SW1
SW2
SW3
V
–
IN–
B
S
CAPACITIVE
DAC
When operated with a 5 V supply, the maximum reference that
can be applied to the part is 3.5 V, and when operated with a 3 V
supply, the maximum reference that can be applied to the part
is 2.2 V. (See the References section.)
Figure 4. ADC Conversion Phase
ADC TRANSFER FUNCTION
The AD7450 has an on-chip differential track-and-hold amplifier,
a successive approximation (SAR) ADC, and a serial interface that
is housed in either an 8-lead SOIC or µSOIC package. The serial
clock input accesses data from the part and also provides the
clock source for the successive approximation ADC. The AD7450
features a power-down option for reduced power consumption
between conversions. The power-down feature is implemented
across the standard serial interface as described in the Modes of
Operation section.
The output coding for the AD7450 is two’s complement. The
designed code transitions occur at successive LSB values (i.e.,
1 LSB, 2 LSB, and so on), and the LSB size is 2 ϫ VREF / 4096.
The ideal transfer characteristic of the AD7450 is shown in Figure 5.
1LSB = 2 ꢃ V
/4096
REF
011...111
011...110
CONVERTER OPERATION
000...001
000...000
111...111
The AD7450 is a successive approximation ADC based on two
capacitive DACs. Figures 3 and 4 show simplified schematics of
the ADC in acquisition and conversion phase, respectively. The
ADC is comprised of control logic, a SAR, and two capacitive
DACs. In Figure 3 (the acquisition phase), SW3 is closed and
SW1 and SW2 are in Position A, the comparator is held in a
balanced condition, and the sampling capacitor arrays acquire
the differential signal on the input.
100...010
100...001
100...000
–V
REF
+ 1LSB
0LSB
+V – 1LSB
REF
ANALOG INPUT
(V
–V
)
IN+ IN–
CAPACITIVE
DAC
Figure 5. Ideal Transfer Characteristics
TYPICAL CONNECTION DIAGRAM
COMPARATOR
B
C
C
S
Figure 6 shows a typical connection diagram for the AD7450
for both 5 V and 3 V supplies. In this setup, the GND pin is
connected to the analog ground plane of the system. The VREF
pin is connected to either a 2.5 V or a 1.25 V decoupled reference
source, depending on the power supply, to set up the analog
input range. The common-mode voltage has to be set up exter-
nally and is the value that the two inputs are centered on. For
more details on driving the differential inputs and setting up the
common mode, see the Driving Differential Inputs section.
The conversion result for the ADC is output in a 16-bit word
consisting of four leading zeros followed by the MSB of the
12-bit result. For applications where power consumption is of
concern, the power-down mode should be used between
conversions, or bursts of several conversions, to improve power
performance. See Modes of Operation section.
V
+
–
IN+
A
A
CONTROL
LOGIC
SW1
SW2
SW3
V
IN–
B
S
CAPACITIVE
DAC
Figure 3. ADC Acquisition Phase
When the ADC starts a conversion (Figure 4), SW3 will open and
SW1 and SW2 will move to Position B, causing the comparator to
become unbalanced. Both inputs are disconnected once the con-
version begins. The control logic and the charge redistribution
DACs are used to add and subtract fixed amounts of charge
from the sampling capacitor arrays to bring the comparator back
into a balanced condition. When the comparator is rebalanced,
the conversion is complete. The control logic generates the ADC’s
output code. The output impedances of the sources driving the
VIN+ and VIN– pins must be matched; otherwise, the two inputs
will have different settling times, resulting in errors.
–10–
REV. 0
AD7450
3V/5V
Figures 8 and 9 show how the common-mode range typically
varies with VREF for both a 5 V and a 3 V power supply. The
common mode must be in this range to guarantee the
functionality of the AD7450.
SUPPLY
0.1ꢀF
10ꢀF
SERIAL
INTERFACE
V
DD
For ease of use, the common mode can be set up to be equal to
V
REF, resulting in the differential signal being VREF centered
V
p-p
p-p
V
CM*
CM*
SCLK
REF
IN+
on VREF. When a conversion takes place, the common mode is
rejected resulting in a virtually noise free signal of amplitude
–VREF to +VREF corresponding to the digital codes of 0 to 4095.
AD7450
ꢀC/ꢀP
SDATA
CS
V
V
IN–
REF
5.0
4.5
4.0
GND
V
REF
1.25V/2.5V
3.5
V
REF
3.25V
0.1ꢀF
3.0
COMMON-MODE RANGE
2.5
2.0
1.5
1.0
0.5
0
*CM = COMMON-MODE VOLTAGE
1.75V
Figure 6. Typical Connection Diagram
THE ANALOG INPUT
The analog input of the AD7450 is fully differential. Differential
signals have a number of benefits over single-ended signals,
including noise immunity based on the device’s common-mode
rejection, improvements in distortion performance, doubling of
the device’s available dynamic range, and flexibility in input ranges
and bias points.
0.25
0.75
1.25
1.75
V
2.25
2.75
3.25 3.50
REF
Figure 8. Input Common-Mode Range vs. VREF
(VDD = 5 V and VREF (Max) = 3.5 V)
Figure 7 defines the fully differential analog input of the AD7450.
3.0
2.5
V
p-p
p-p
REF
V
IN+
AD7450
2.0
1.5
1.0
0.5
2V
1V
V
V
IN–
REF
COMMON-MODE
VOLTAGE
COMMON-MODE RANGE
Figure 7. Differential Input Definition
The amplitude of the differential signal is the difference between
the signals applied to the VIN+ and VIN– pins (i.e., VIN+ – VIN–).
VIN+ and VIN– are simultaneously driven by two signals each of
amplitude VREF that are 180° out of phase. The amplitude of
the differential signal is therefore –VREF to +VREF p-p
(i.e., 2 ϫ VREF). This is regardless of the common mode (CM).
The common mode is the average of the two signals, i.e.,
(VIN+ + VIN–)/2, and is therefore the voltage that the two inputs
are centered on. This results in the span of each input being
CM VREF/2. This voltage has to be set up externally and its
range varies with VREF. As the value of VREF increases, the com-
mon-mode range decreases. When driving the inputs with an
amplifier, the actual common-mode range will be determined
by the amplifier’s output voltage swing.
0
0.25
0.50
0.75
1.00
1.25
1.50
1.75
2.00 2.20
V
REF
Figure 9. Input Common-Mode Range vs. VREF (VDD = 3 V
and VREF (Max) = 2.2 V)
Figure 10 shows examples of the inputs to VIN+ and VIN– for
different values of VREF for VDD = 5 V. It also gives the maxi-
mum and minimum common-mode voltages for each reference
value according to Figure 8.
REV. 0
–11–
AD7450
REFERENCE = 1.25V
total harmonic distortion (THD) that can be tolerated. The THD
will increase as the source impedance increases and the perfor-
mance will degrade. Figure 12 shows a graph of the THD versus
the analog input signal frequency for different source impedances.
V
–
IN
1.25V p-p
COMMON-MODE (CM)
CM
= 0.625V
MIN
CM
MAX
= 4.42V
V
ꢂ
IN
REFERENCE = 2.5V
–70
V
–
IN
T
= 25ꢁC
A
V
= 3V
DD
2.5V p-p
COMMON-MODE (CM)
R
= 1kꢄ
IN
–72
–74
–76
–78
–80
–82
CM
= 1.25V
MIN
CM
= 3.75V
MAX
V
ꢂ
IN
Figure 10. Examples of the Analog Inputs to VIN+
and VIN– for Different Values of VREF for VDD = 5 V
V
R
= 3V
= 100ꢄ
DD
IN
Analog Input Structure
Figure 11 shows the equivalent circuit of the analog input struc-
ture of the AD7450. The four diodes provide ESD protection
for the analog inputs. Care must be taken to ensure that the
analog input signals never exceed the supply rails by more than
300 mV. This will cause these diodes to become forward biased
and start conducting into the substrate. These diodes can conduct
up to 10 mA without causing irreversible damage to the part.
V
= 5V
DD
V
R
= 5V
= 100ꢄ
DD
R
= 1kꢄ
IN
IN
10
100
1000
INPUT FREQUENCY – kHz
The capacitors, C1, in Figure 11 are typically 4 pF and can prima-
rily be attributed to pin capacitance. The resistors are lumped
components made up of the ON resistance of the switches. The
value of these resistors is typically about 100 Ω. The capacitors,
C2, are the ADC’s sampling capacitors and have a capacitance
of 16 pF typically.
Figure 12. THD vs. Analog Input Frequency for
Various Source Impedances for VDD = 5 V and 3 V
Figure 13 shows a graph of the THD versus the analog input
frequency for VDD of 5 V 5% and 3 V 10%, while sampling
at 1 MSPS and 833 kSPS with a SCLK of 18 MHz and
15 MHz, respectively. In this case, the source impedance is 10 Ω.
For ac applications, removing high-frequency components from
the analog input signal is recommended by the use of an RC
low-pass filter on the relevant analog input pins. In applications
where harmonic distortion and signal-to-noise ratio are critical,
the analog input should be driven from a low impedance source.
Large source impedances will significantly affect the ac perfor-
mance of the ADC. This may necessitate the use of an input
buffer amplifier. The choice of the op amp will be a function of
the particular application.
–60
T
= 25ꢁC
A
–65
–70
–75
–80
–85
–90
–95
V
= 2.7V
DD
V
= 3.3V
DD
V
DD
V
= 4.75V
V
= 5.25V
100
DD
DD
D
C2
R1
V
IN+
C1
D
10
1000
INPUT FREQUENCY – kHz
V
DD
Figure 13. THD vs. Analog Input Frequency for 3 V
10% and 5 V 5% Supply Voltages
D
D
C2
R1
V
DRIVING DIFFERENTIAL INPUTS
IN–
Differential operation requires that VIN+ and VIN– be simulta-
neously driven with two equal signals that are 180o out of phase.
The common mode must be set up externally and has a range
that is determined by VREF, the power supply, and the particular
amplifier used to drive the analog inputs (see Figures 8 and 9).
Differential modes of operation with either an ac or dc input
provide the best THD performance over a wide frequency range.
Since not all applications have a signal preconditioned for
differential operation, there is often a need to perform single-
ended-to-differential conversion.
C1
Figure 11. Equivalent Analog Input Circuit
Conversion Phase—Switches Open
Track Phase—Switches Closed
When no amplifier is used to drive the analog input, the source
impedance should be limited to values lower than 1 kΩ. The
maximum source impedance will depend on the amount of
–12–
REV. 0
AD7450
3.75V
2.5V
Rf1
1.25V
Rs*
Rs*
Rg1
V
V
V
IN+
C
*
*
+2.5V
OCM
AD8138
Rf2
AD7450
Rg2
GND
51R
V
IN–
REF
–2.5V
C
3.75V
2.5V
1.25V
*MOUNT AS CLOSE TO THE AD7450
AS POSSIBLE AND ENSURE HIGH
PRECISION Rs AND Cs ARE USED
EXTERNAL
(2.5V)
V
REF
Rs – 50R; C – 1nF;
Rg1 = Rf1 = Rf2 = 499R; Rg2 = 523R
Figure 14. Using the AD8138 as a Single-Ended-to-Differential Amplifier
Differential Amplifier
The voltage applied to Point A sets up the common-mode voltage.
In both diagrams, it is connected in some way to the reference,
but any value in the common-mode range can be input here to
set up the common mode. Examples of suitable dual op amps
that could be used in this configuration to provide differential
drive to the AD7450 are the AD8042, AD8056, and AD8022.
An ideal method of applying differential drive to the AD7450 is to
use a differential amplifier, such as the AD8138. This part can be
used as a single-ended-to-differential amplifier or as a differential-
to-differential amplifier. In both cases, the analog input needs to
be bipolar. It also provides common-mode level shifting and buffer-
ing of the bipolar input signal. Figure 14 shows how the AD8138
can be used as a single-ended-to-differential amplifier. The positive
and negative outputs of the AD8138 are connected to the respective
inputs on the ADC via a pair of series resistors to minimize the
effects of switched capacitance on the front end of the ADC.
The RC low-pass filter on each analog input is recommended in
ac applications to remove the high-frequency components of the
analog input. The architecture of the AD8138 results in outputs
that are highly balanced over a wide frequency range without
requiring tightly matched external components.
Care must be taken when choosing the op amp, since the selec-
tion will depend on the required power supply and the system
performance objectives. The driver circuits in Figure 15a and
Figure 15b are optimized for dc coupling applications requiring
optimum distortion performance.
The differential op amp driver circuit in Figure 15a is configured
to convert and level shift a single-ended, ground referenced
(bipolar) signal to a differential signal centered at the VREF level
of the ADC.
If the analog input source being used has zero impedance then all
four resistors (Rg1, Rg2, Rf1, and Rf2) should be the same. If the
source has a 50 Ω impedance and a 50 Ω termination, for example,
the value of Rg2 should be increased by 25 Ω to balance this paral-
lel impedance on the input and thus ensure that both the positive
and negative analog inputs have the same gain (see Figure 14).
The outputs of the amplifier are perfectly matched, balanced
differential outputs of identical amplitude and exactly 180o out
of phase.
220ꢄ
2 ꢃ V
p-p
REF
V+
390ꢄ
220ꢄ
V
DD
GND
27ꢄ
V–
V
IN+
220ꢄ
220ꢄ
V+
AD7450
V
IN–
V
REF
The AD8138 is specified with 3 V, 5 V, and 5 V power supplies,
but the best results are obtained when it is supplied by 5 V.
A lower cost device that could also be used in this configuration
with slight differences in characteristics to the AD8138, but with
similar performance and operation, is the AD8132.
0.1ꢀF
27ꢄ
A
V–
10kꢄ
20kꢄ
EXTERNAL
V
REF
Op Amp Pair
An op amp pair can be used to directly couple a differential
signal to the AD7450. The circuit configurations shown in
Figures 15a and 15b show how a dual op amp can be used to
convert a single-ended signal into a differential signal for both a
bipolar and a unipolar input signal, respectively.
Figure 15a. Dual Op Amp Circuit to Convert a
Single-Ended Bipolar Input into a Differential Input
REV. 0
–13–
AD7450
The circuit configuration shown in Figure 15b converts a unipolar,
single-ended signal into a differential signal.
Example 1:
VIN max =VDD + 0.3
VIN max =VREF +VREF
If VDD = 5V
2
220ꢄ
2 ꢃ V
p-p
REF
V+
390ꢄ
V
DD
27ꢄ
VREF
GND
ThenVIN max = 5.3V
Therefore 3 ×VREF 2 = 5.3V
V–
V
IN+
VREF max = 3.5V
220ꢄ
220ꢄ
V+
AD7450
V
IN–
V
REF
Therefore, when operating at VDD = 5 V, the value of VREF can
range from 100 mV to a maximum value of 3.5 V. When VDD
4.75 V, VREF max = 3.37 V.
=
0.1ꢀF
27ꢄ
A
Example 2:
V–
VIN max =VDD + 0.3
10kꢄ
VIN max =VREF +VREF
If VDD = 3.3V
2
EXTERNAL
V
REF
ThenVIN max = 3.6V
Figure 15b. Dual Op Amp Circuit to Convert a
Single-Ended Unipolar Input into a Differential Input
Therefore 3 ×VREF 2 = 3.6V
VREF max = 2.4V
RF Transformer
In systems that do not need to be dc-coupled, an RF transformer
with a center tap offers a good solution for generating differential
inputs. Figure 16 shows how a transformer is used for single-
ended-to-differential conversion. It provides the benefits of
operating the ADC in the differential mode without contributing
additional noise and distortion. An RF transformer also has the
benefit of providing electrical isolation between the signal source
and the ADC. A transformer can be used for most ac applications.
The center tap is used to shift the differential signal to the
common-mode level required. In this case, it is connected to the
reference so the common-mode level is the value of the reference.
Therefore, when operating at VDD = 3.3 V, the value of VREF
can range from 100 mV to a maximum value of 2.4 V. When
VDD = 2.7 V, VREF max = 2 V.
These examples show that the maximum reference applied to
the AD7450 is directly dependant on the value of VDD
.
The performance of the part at different reference values is shown
in TPC 8 to TPC 12 and in TPC 15. The value of the reference
sets the analog input span and the common-mode voltage range.
Errors in the reference source will result in gain errors in the
AD7450 transfer function and will add to specified full-scale errors
on the part. A capacitor of 0.1 µF should be used to decouple
the VREF pin to GND. Table I lists examples of suitable voltage
references to be used that are available from Analog Devices, and
Figure 17 shows a typical connection diagram for the VREF pin.
3.75V
2.5V
1.25V
R
R
R
V
IN+
Table I. Examples of Suitable Voltage References
C
AD7450
V
V
IN–
REF
Output
Initial
Operating
3.75V
2.5V
Reference Voltage Accuracy (% Max) Current (ꢀA)
1.25V
AD589
AD1580
REF192
REF43
AD780
1.235
1.225
2.5
2.5
2.5
1.2–2.8
50
50
45
600
1000
0.08–0.8
0.08–0.4
0.06–0.1
0.04–0.2
EXTERNAL
(2.5V)
V
REF
Figure 16. Using an RF Transformer to Generate
Differential Inputs
V
DD
REFERENCES SECTION
AD780
AD7450*
An external reference source is required to supply the reference to the
AD7450. This reference input can range from 100 mV to 3.5 V. With
a 5 V power supply, the specified reference is 2.5 V and the maximum
reference is 3.5 V. With a 3.3 V power supply, the specified refer-
ence is 1.25 V and the maximum reference is 2.4 V. In both cases,
the reference is functional from 100 mV. It is important to ensure
that, when choosing the reference value for a particular application,
the maximum analog input range (VIN max) is never greater than
VDD + 0.3 V to comply with the maximum ratings of the part. The
following two examples calculate the maximum VREF input that can be
used when operating the AD7450 at VDD of 5 V and 3.3 V, respectively.
NC
O/P SEL
8
NC
NC
1
2
V
REF
V
7
6
V
DD
IN
2.5V
V
3
4
TEMP
GND
OUT
0.1ꢀF
0.1ꢀF
10nF
0.1ꢀF
NC
TRIM
5
NC = NO CONNECT
*ADDITIONAL PINS OMITTED FOR CLARITY
Figure 17. Typical VREF Connection Diagram for VDD = 5 V
REV. 0
–14–
AD7450
SINGLE-ENDED OPERATION
is valid on the 16th falling edge, having been clocked out on the
previous (15th) falling edge. Once the conversion is complete
and the data has been accessed after the 16 clock cycles, it is
important to ensure that before the next conversion is initiated,
enough time is left to meet the acquisition and quiet time speci-
fications (see timing examples). To achieve 1 MSPS with an 18
MHz clock for VDD = 5 V, an 18 clock burst will perform the
conversion and leave enough time before the next conversion for
the acquisition and quiet time. This is the same for achieving
833 kSPS with a 15 MHz clock for VDD = 3 V.
When supplied with a 5 V power supply, the AD7450 can handle
a single-ended input. The design of this part is optimized for
differential operation, so with a single-ended input, performance
will degrade. Linearity will typically degrade by 0.2 LSBs, zero
code and full-scale errors will typically degrade by 2 LSBs, and
ac performance is not guaranteed.
To operate the AD7450 in single-ended mode, the VIN+ input is
coupled to the signal source, while the VIN– input is biased to the
appropriate voltage corresponding to the midscale code transi-
tion. This voltage is the common mode, which is a fixed dc
voltage (usually the reference). The VIN+ input swings around
this value and should have voltage span of 2 ϫ VREF to make use
of the full dynamic range of the part. Therefore, the input signal
will have peak-to-peak values of common mode VREF. If the
analog input is unipolar then an op amp in a noninverting unity
gain configuration can be used to drive the VIN+ pin. Because
the ADC operates from a single supply, it is necessary to level
shift ground based bipolar signals to comply with the input
requirements. An op amp can be configured to rescale and level
shift the ground based bipolar signal so it is compatible with the
selected input range of the AD7450 (see Figure 18).
In applications with a slower SCLK, it may be possible to read
in data on each SCLK rising edge, i.e., the first rising edge of
SCLK after the CS falling edge would have the leading zero
provided and the 15th SCLK edge would have DB0 provided.
Timing Example 1
Having fSCLK = 18 MHz and a throughput rate of 1 MSPS gives
a cycle time of:
1 Throughput =11,000,000 =1µs
A cycle consists of:
t + 12.5 1 f
+ t
= 1µs
(
)
2
SCLK
ACQ
Therefore, if t2 = 10 ns then:
5V
R
2.5V
0V
10 ns + 12.5 1 18 MHz + t
= 1µs
(
)
+2.5V
ACQ
R
R
V
+
–
0V
IN
tACQ = 296 ns
–2.5V
V
IN+
AD7450
R
This 296 ns satisfies the requirement of 200 ns for tACQ. From
Figure 20, tACQ is comprised of:
V
V
IN–
REF
2.5 1 f
+ t + t
(
)
SCLK
8
QUIET
EXTERNAL
(2.5V)
0.1ꢀF
V
REF
where t8 = 35 ns. This allows a value of 122 ns for tQUIET, satis-
fying the minimum requirement of 25 ns.
Figure 18. Applying a Bipolar Single-Ended
Input to the AD7450
Timing Example 2
Having fSCLK = 5 MHz and a throughput rate of 315 kSPS gives
a cycle time of:
SERIAL INTERFACE
Figure 19 shows a detailed timing diagram for the serial interface
of the AD7450. The serial clock provides the conversion clock
and also controls the transfer of data from the AD7450 during
conversion. CS initiates the conversion process and frames the
data transfer. The falling edge of CS puts the track-and-hold into
hold mode and takes the bus out of three-state. The analog input
is sampled and the conversion initiated at this point. The
conversion will require 16 SCLK cycles to complete.
1 Throughput = 1 315,000 = 3.174 µs
A cycle consists of:
t + 12.5 1 f
+ t
= 3.174 µs
(
)
2
SCLK
ACQ
Therefore if t2 is 10 ns then:
10 ns + 12.5 1 5 MHz + t
= 3.174 µs
(
)
ACQ
Once 13 SCLK falling edges have occurred, the track-and-hold
will go back into track on the next SCLK rising edge as shown
at Point B in Figure 19. On the 16th SCLK falling edge, the
SDATA line will go back into three-state.
tACQ = 664 ns
This 664 ns satisfies the requirement of 200 ns for tACQ. From
Figure 20, tACQ is comprised of:
If the rising edge of CS occurs before 16 SCLKs have elapsed,
the conversion will be terminated, and the SDATA line will go
back into three-state. Sixteen serial clock cycles are required to
perform a conversion and to access data from the AD7450. CS
going low provides the first leading zero to be read in by the
microcontroller or DSP. The remaining data is then clocked out
on the subsequent SCLK falling edges beginning with the second
leading zero. Thus, the first falling clock edge on the serial clock
provides the second leading zero. The final bit in the data transfer
2.5 1 f
+ t + t
(
)
SCLK
8
QUIET
where t8 = 35 ns. This allows a value of 129 ns for tQUIET, satis-
fying the minimum requirement of 25 ns.
As in this example and with other slower clock values, the signal
may already be acquired before the conversion is complete, but it
is still necessary to leave 25 ns minimum tQUIET between conver-
sions. In Timing Example 2, the signal should be fully acquired
at approximately Point C in Figure 20.
REV. 0
–15–
AD7450
t1
CS
tCONVERT
B
t2
t5
SCLK
1
2
3
4
5
13
14
t6
15
16
t7
t8
tQUIET
t4
t3
0
SDATA
0
0
0
DB11
DB10
DB2
DB1
DB0
THREE-STATE
4 LEADING ZEROS
Figure 19. Serial Interface Timing Diagram
CS
tCONVERT
t2
t5
10ns
B
C
SCLK
1
2
3
4
5
13
14
t6
15
16
t8
tQUIET
tACQ
12.5(1/f
)
SCLK
1/THROUGHPUT
Figure 20. Serial Interface Timing Example
MODES OF OPERATION
Sixteen serial clock cycles are required to complete the conver-
sion and access the complete conversion result. CS may idle
high until the next conversion or idle low until sometime prior
to the next conversion. Once a data transfer is complete, i.e.,
when SDATA has returned to three-state, another conversion
can be initiated after the quiet time, tQUIET, has elapsed by again
bringing CS low.
The mode of operation of the AD7450 is selected by controlling
the logic state of the CS signal during a conversion. There are
two possible modes of operation, normal mode and power-down
mode. The point at which CS is pulled high after the conversion
has been initiated will determine whether or not the AD7450 will
enter the power-down mode. Similarly, if already in power-down,
CS controls whether the device will return to normal operation or
remain in power-down. These modes of operation are designed
to provide flexible power management options. These options
can be chosen to optimize the power dissipation/throughput rate
ratio for differing application requirements.
CS
10
16
1
SCLK
Normal Mode
This mode is intended for the fastest throughput rate perfor-
mance. The user does not have to worry about any power-up
times since the AD7450 is kept fully powered up. Figure 21
shows the general diagram of the operation of the AD7450 in
this mode. The conversion is initiated on the falling edge of CS
as described in the Serial Interface section. To ensure the part
remains fully powered up, CS must remain low until at least 10
SCLK falling edges have elapsed after the falling edge of CS.
SDATA
4 LEADING ZEROS AND CONVERSION RESULT
Figure 21. Normal Mode Operation
Power-Down Mode
This mode is intended for use in applications where slower
throughput rates are required; either the ADC is powered down
between each conversion or a series of conversions may be
performed at a high throughput rate, during which the ADC is
powered down for a relatively long duration between these bursts of
several conversions. When the AD7450 is in the power-down
mode, all analog circuitry is powered down. To enter power-down
mode, the conversion process must be interrupted by bringing CS
high anywhere after the second falling edge of SCLK and before
the 10th falling edge of SCLK as shown in Figure 22.
If CS is brought high any time after the 10th SCLK falling edge,
but before the 16th SCLK falling edge, the part will remain
powered up, but the conversion will be terminated and SDATA
will go back into three-state.
–16–
REV. 0
AD7450
track-and-hold, which was in hold mode while the part was
powered down, returns to track mode after the first SCLK
edge the part receives after the falling edge of CS. This is shown
as Point A in Figure 23.
CS
1
2
10
SCLK
Although at any SCLK frequency one dummy cycle is sufficient
to power the device up and acquire VIN, it does not necessarily
mean that a full dummy cycle of 16 SCLKs must always elapse
to power up the device and acquire VIN fully; 1 µs will be
sufficient to power the device up and acquire the input signal.
THREE-STATE
SDATA
Figure 22. Entering Power-Down Mode
Once CS has been brought high in this window of SCLKs, the
part will enter power-down, the conversion that was initiated by
the falling edge of CS will be terminated, and SDATA will go
back into three-state. The time from the rising edge of CS to
SDATA three-state enabled will never be greater than t8 (see
Timing Specifications). If CS is brought high before the second
SCLK falling edge, the part will remain in normal mode and will
not power down. This will avoid accidental power-down due to
glitches on the CS line.
For example, if a 5 MHz SCLK frequency was applied to the ADC,
the cycle time would be 3.2 µs (i.e., 1/(5 MHz) ϫ 16). In one
dummy cycle, 3.2 µs, the part would be powered up and VIN
acquired fully. However, after 1 µs with a 5 MHz SCLK, only
5 SCLK cycles would have elapsed. At this stage, the ADC would
be fully powered up and the signal acquired. So, in this case, the
CS can be brought high after the 10th SCLK falling edge and
brought low again after a time, tQUIET, to initiate the conversion.
When power supplies are first applied to the AD7450, the ADC
may either power up in the power-down mode or normal mode.
Because of this, it is best to allow a dummy cycle to elapse to
ensure the part is fully powered up before attempting a valid
conversion. Likewise, if the user wishes the part to power up in
power-down mode, then the dummy cycle may be used to ensure
the device is in power-down by executing a cycle such as that
shown in Figure 22.
To exit this mode of operation and power the AD7450 up again,
a dummy conversion is performed. On the falling edge of CS, the
device will begin to power up and continue to power up as long
as CS is held low until after the falling edge of the 10th SCLK. The
device will be fully powered up after 1 µs has elapsed and, as
shown in Figure 23, valid data will result from the next conversion.
If CS is brought high before the 10th falling edge of SCLK, the
AD7450 will again go back into power-down. This avoids
accidental power-up due to glitches on the CS line or an
inadvertent burst of eight SCLK cycles while CS is low. So although
the device may begin to power up on the falling edge of CS, it will
again power down on the rising edge of CS as long as it occurs
before the 10th SCLK falling edge.
Once supplies are applied to the AD7450, the power-up time is
the same as that when powering up from the power-down mode.
It takes approximately 1 µs to power up fully if the part powers
up in normal mode. It is not necessary to wait 1 µs before
executing a dummy cycle to ensure the desired mode of operation.
Instead, the dummy cycle can occur directly after power is
supplied to the ADC. If the first valid conversion is then performed
directly after the dummy conversion, care must be taken to ensure
that adequate acquisition time has been allowed.
Power-Up Time
The power-up time of the AD7450 is typically 1 µs, which means
that with any frequency of SCLK up to 18 MHz, one dummy cycle
will always be sufficient to allow the device to power up. Once
the dummy cycle is complete, the ADC will be fully powered up
and the input signal will be acquired properly. The quiet time,
tQUIET, must still be allowed from the point at which the bus
goes back into three-state after the dummy conversion to the
next falling edge of CS.
As mentioned earlier, when powering up from the power-down
mode, the part will return to track upon the first SCLK edge
applied after the falling edge of CS. However, when the ADC
powers up initially after supplies are applied, the track-and-hold
will already be in track. This means if (assuming one has the
facility to monitor the ADC supply current) the ADC powers
up in the desired mode of operation, and thus a dummy cycle is
not required to change the mode, then a dummy cycle is not
required to place the track-and-hold into track.
When running at the maximum throughput rate of 1 MSPS,
the AD7450 will power up and acquire a signal within 0.5 LSB
in one dummy cycle, i.e., 1 µs. When powering up from the
power-down mode with a dummy cycle, as in Figure 23, the
tPOWER-UP
THE PART IS FULLY POWERED
THE PART BEGINS
TO POWER UP
UPWITHV FULLY ACQUIRED
IN
CS
10
16
1
1
10
16
SCLK
A
SDATA
INVALID DATA
VALID DATA
Figure 23. Exiting Power-Down Mode
REV. 0
–17–
AD7450
POWER VERSUS THROUGHPUT RATE
AD7450 to ADSP-21xx
By using the power-down mode on the AD7450 when not
converting, the average power consumption of the ADC decreases
at lower throughput rates. Figure 24 shows how, as the throughput
rate is reduced, the device remains in its power-down state longer,
and the average power consumption reduces accordingly. It shows
this for both 5 V and 3 V power supplies.
The ADSP-21xx DSPs are interfaced directly to the AD7450
without any glue logic required.
The SPORT control register should be set up as follows:
TFSW = RFSW = 1, Alternate Framing
INVRFS = INVTFS = 1, Active Low Frame Signal
DTYPE = 00, Right Justify Data
SLEN = 1111, 16-Bit Data-Words
ISCLK = 1, Internal Serial Clock
TFSR = RFSR = 1, Frame Every Word
IRFS = 0
For example, if the AD7450 is operated in continuous sampling
mode with a throughput rate of 100 kSPS and an SCLK of 18 MHz,
and the device is placed in the power-down mode between
conversions, then the power consumption is calculated as follows:
Power dissipation during normal operation = 9 mW max for
VDD = 5 V.
ITFS = 1
To implement the power-down mode, SLEN should be set to
1001 to issue an 8-bit SCLK burst.
If the power-up time is one dummy cycle, i.e., 1 µs, and the
remaining conversion time is another cycle, i.e., 1 µs, then the
AD7450 can be said to dissipate 9 mW for 2 µs* during each
conversion cycle.
The connection diagram is shown in Figure 25. The ADSP-21xx
has the TFS and RFS of the SPORT tied together, with TFS
set as an output and RFS set as an input. The DSP operates in
alternate framing mode and the SPORT control register is set
up as described. The frame synchronization signal generated on
the TFS is tied to CS and, as with all signal processing applica-
tions, equidistant sampling is necessary. However, in this example,
the timer interrupt is used to control the sampling rate of the
ADC and, under certain conditions, equidistant sampling
may not be achieved.
If the throughput rate = 100 kSPS, then the cycle time = 10 µs,
and the average power dissipated during each cycle is:
(2/10) ϫ 9 mW = 1.8 mW
For the same scenario, if VDD = 3 V, the power dissipation
during normal operation is 3.75 mW max.
The AD7450 can now be said to dissipate 3.75 mW for 2 µs*
during each conversion cycle.
The average power dissipated during each cycle with a throughput
rate of 100 kSPS is therefore:
ADSP-21xx*
SCLK
AD7450*
SCLK
(2/10) ϫ 3.75 mW = 0.75 mW
DR
SDATA
This is how the power numbers in Figure 24 are calculated.
CS
RFS
TFS
For throughput rates above 320 kSPS, it is recommended that the
serial clock frequency is reduced for optimum power performance.
100
*ADDITIONAL PINS OMITTED FOR CLARITY
V
= 5V
DD
SCLK = 18MHz
Figure 25. Interfacing to the ADSP-21xx
10
1
The timer registers are loaded with a value that provides an
interrupt at the required sample interval. When an interrupt is
received, a value is transmitted with TFS/DT (ADC control word).
The TFS is used to control the RFS and hence the reading of
data. The frequency of the serial clock is set in the SCLKDIV
register. When the instruction to transmit with TFS is given,
(i.e., AX0 = TX0), the state of the SCLK is checked. The DSP
will wait until the SCLK has gone High, Low, and High before
transmission will start. If the timer and SCLK values are chosen
such that the instruction to transmit occurs on or near the rising
edge of SCLK, then the data may be transmitted, or it may wait
until the next clock edge.
V
= 3V
DD
SCLK = 15MHz
0.1
0.01
0
50
100
150
200
250
300
350
THROUGHPUT – kSPS
Figure 24. Power vs. Throughput Rate for
Power-Down Mode
For example, the ADSP-2111 has a master clock frequency of
16 MHz. If the SCLKDIV register is loaded with the value 3,
then a SCLK of 2 MHz is obtained and eight master clock
periods will elapse for every 1 SCLK period. If the timer regis-
ters are loaded with the value 803, then 100.5 SCLKs will occur
between interrupts and subsequently between transmit instruc-
tions. This situation will result in nonequidistant sampling as
the transmit instruction is occurring on a SCLK edge. If the
number of SCLKs between interrupts is a whole integer figure of
N, then equidistant sampling will be implemented by the DSP.
MICROPROCESSOR AND DSP INTERFACING
The serial interface on the AD7450 allows the part to be directly
connected to a range of different microprocessors. This section
explains how to interface the AD7450 with some of the more
common microcontroller and DSP serial interface protocols.
*This figure assumes a very small time to enter power-down mode. This will
increase as the burst of clocks used to enter the power-down mode is increased.
–18–
REV. 0
AD7450
AD7450 to TMS320C5x/C54x
AD7450 to DSP56xxx
The serial interface on the TMS320C5x/C54x uses a continuous
serial clock and frame synchronization signals to synchronize the
data transfer operations with peripheral devices, such as the
AD7450. The CS input allows easy interfacing between the
TMS320C5x/C54x and the AD7450 with no glue logic required.
The serial port of the TMS320C5x/C54x is set up to operate in
burst mode with internal CLKX (Tx serial clock) and FSX (Tx
frame sync). The serial port control register (SPC) must have
the following setup: FO = 0, FSM = 1, MCM = 1, and
TXM = 1. The format bit, FO, may be set to 1 to set the word
length to 8 bits in order to implement the power-down mode on
the AD7450. The connection diagram is shown in Figure 26. For
signal processing applications, it is imperative that the frame
synchronization signal from the TMS320C5x/C54x provide equi-
distant sampling.
The connection diagram in Figure 28 shows how the AD7450 can
be connected to the SSI (synchronous serial interface) of the
DSP56xxx family of DSPs from Motorola. The SSI is operated
in synchronous mode (SYN bit in CRB = 1) with internally
generated 1-bit clock period frame sync for both Tx and Rx
(Bits FSL1 = 1 and FSL0 = 0 in CRB). Set the word length to
16 by setting Bits WL1 = 1 and WL0 = 0 in CRA. To imple-
ment the power-down mode on the AD7450, the word length
can be changed to 8 bits by setting its WL1 = 0 and WL0 = 0 in
CRA. It should be noted that for signal processing applica-
tions, it is imperative that the frame synchronization signal
from the DSP56xxx will provide equidistant sampling.
DSP56xxx*
AD7450*
SCLK
SCLK
TMS320C5x/C54x*
CLKX
AD7450*
SDATA
SRD
SR2
SCLK
CLKR
DR
CS
SDATA
CS
FSX
FSR
*ADDITIONAL PINS OMITTED FOR CLARITY
Figure 28. Interfacing to the DSP56xxx
*ADDITIONAL PINS OMITTED FOR CLARITY
APPLICATION HINTS
Figure 26. Interfacing to the TMS320C5x/C54x
AD7450 to MC68HC16
Grounding and Layout
The printed circuit board that houses the AD7450 should be
designed so that the analog and digital sections are separated
and confined to certain areas of the board. This facilitates the
use of ground planes that can be easily separated. A minimum
etch technique is generally best for ground planes since it gives
the best shielding. Digital and analog ground planes should be
joined in only one place, and the connection should be a star
ground point established as close to the GND pin on the AD7450
as possible. Avoid running digital lines under the device, as this
will couple noise onto the die. The analog ground plane should
be allowed to run under the AD7450 to avoid noise coupling.
The power supply lines to the AD7450 should use as large a trace
as possible to provide low impedance paths and reduce the effects
of glitches on the power supply line.
The serial peripheral interface (SPI) on the MC68HC16 is configured
for master mode (MSTR) = 1, clock polarity bit (CPOL) = 1,
and clock phase bit (CPHA) = 0. The SPI is configured by
writing to the SPI control register (SPCR)—see the 68HC16 user
manual. The serial transfer will take place as a 16-bit operation
when the SIZE bit in the SPCR register is set to SIZE = 1.
To implement the power-down modes with an 8-bit transfer set
SIZE = 0. A connection diagram is shown in Figure 27.
MC68HC16*
AD7450*
SCLK/PMC2
SCLK
MISO/PMC0
SS/PMC3
SDATA
Fast switching signals, such as clocks, should be shielded with
digital ground to avoid radiating noise to other sections of the
board, and clock signals should never run near the analog
inputs. Avoid crossover of digital and analog signals. Traces on
opposite sides of the board should run at right angles to each
other. This reduces the effects of feedthrough through the
board. A microstrip technique is by far the best but is not
always possible with a double-sided board.
CS
*ADDITIONAL PINS OMITTED FOR CLARITY
Figure 27. Interfacing to the MC68HC16
In this technique, the component side of the board is dedicated
to ground planes, while signals are placed on the solder side.
Good decoupling is also important. All analog supplies should
be decoupled with 10 µF tantalum capacitors in parallel with
0.1 µF capacitors to GND. To achieve the best from these
decoupling components, they must be placed as close as possible
to the device.
REV. 0
–19–
AD7450
EVALUATING THE AD7450 PERFORMANCE
other Analog Devices evaluation boards ending with the CB
designator, to demonstrate/evaluate the ac and dc performance
of the AD7450.
The evaluation board package includes a fully assembled and
tested evaluation board, documentation, and software for
controlling the board from a PC via the Evaluation Board
Controller. The Evaluation Board Controller can be used in
conjunction with the AD7450 evaluation board, as well as many
The software allows the user to perform ac (fast Fourier
Transform) and dc (Histogram of codes) tests on the AD7450.
See the evaluation board technical note for more information.
OUTLINE DIMENSIONS
Dimensions shown in millimeters and (inches)
8-Lead SOIC
(R-8)
5.00 (0.1969)
4.80 (0.1890)
8
1
5
4
6.20 (0.2441)
5.80 (0.2283)
4.00 (0.1575)
3.80 (0.1496)
PIN 1
0.50 (0.0197)
0.25 (0.0098)
1.27 (0.0500)
BSC
ꢃ 45ꢁ
COPLANARITY
0.25 (0.0098)
0.10 (0.0039)
1.75 (0.0689)
1.35 (0.0531)
8ꢁ
0ꢁ
1.27 (0.0500)
0.41 (0.0161)
0.49 (0.0193)
0.35 (0.0138)
0.25 (0.0098)
0.19 (0.0075)
SEATING
PLANE
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
Dimensions shown in inches and (mm)
8-Lead ꢀSOIC
(RM-8)
0.122 (3.10)
0.114 (2.90)
8
5
4
0.122 (3.10)
0.114 (2.90)
0.199 (5.05)
0.187 (4.75)
1
PIN 1
0.0256 (0.65) BSC
0.120 (3.05)
0.112 (2.84)
0.120 (3.05)
0.112 (2.84)
0.043 (1.09)
0.037 (0.94)
0.006 (0.15)
0.002 (0.05)
33ꢁ
0.018 (0.46)
0.008 (0.20)
27ꢁ
0.028 (0.71)
0.016 (0.41)
0.011 (0.28)
0.003 (0.08)
SEATING
PLANE
–20–
REV. 0
相关型号：
AD7450AR-REEL7
ADC, Successive Approximation, 12-Bit, 1 Func, 1 Channel, Serial Access, PDSO8, SOIC-8
ROCHESTER
AD7450ARM-REEL
ADC, Successive Approximation, 12-Bit, 1 Func, 1 Channel, Serial Access, PDSO8, MICRO, SOIC-8
ROCHESTER
AD7450ARM-REEL7
ADC, Successive Approximation, 12-Bit, 1 Func, 1 Channel, Serial Access, PDSO8, MICRO, SOIC-8
ROCHESTER
AD7450BR
ADC, Successive Approximation, 12-Bit, 1 Func, 1 Channel, Serial Access, PDSO8, SOIC-8
ROCHESTER
©2020 ICPDF网 联系我们和版权申明