ISL6564ACRZ-T [ROCHESTER]
SWITCHING CONTROLLER, 1500 kHz SWITCHING FREQ-MAX, PQCC40, 6 X 6 MM, ROHS COMPLIANT, PLASTIC, MO-220, QFN-40;型号: | ISL6564ACRZ-T |
厂家: | Rochester Electronics |
描述: | SWITCHING CONTROLLER, 1500 kHz SWITCHING FREQ-MAX, PQCC40, 6 X 6 MM, ROHS COMPLIANT, PLASTIC, MO-220, QFN-40 开关 |
文件: | 总29页 (文件大小:1349K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
ISL6564A
®
Data Sheet
March 20, 2007
FN6285.1
Multiphase PWM Controller with Linear
6-Bit DAC Capable of Precision r
DCR Differential Current Sensing
Features
or
DS(ON)
• Precision Multiphase Core Voltage Regulation
- Differential Remote Voltage Sensing
- ±0.5% System Accuracy
The ISL6564A is a Multiphase PWM controller which controls
microprocessor core voltage regulation by driving up to
4 synchronous-rectified buck channels. It features a high
bandwidth control loop to provide optimal response to the load
transients. With switching frequency up to 1.5MHz per phase,
the ISL6564A based voltage regulator requires minimum
components and PCB area in DC/DC converter application.
- Adjustable Reference-Voltage Offset
• Precision r
DS(ON)
or DCR Current Sensing
- Accurate Load-Line Programming
- Accurate Channel-Current Balancing
- Differential Current Sense
- Low-Cost, Lossless Current Sensing
The ISL6564A senses current by utilizing patented
• Internal Shunt Regulator for 5V or 12V Biasing
techniques to measure the voltage across the on resistance,
r
, of the lower MOSFETs or DCR of the output
DS(ON)
• Microprocessor Voltage Identification Input
- Self Clocked Dynamic VID™ Control Technology
- 6-Bit VID Input
inductor during their conduction intervals. Current sensing
provides the needed signals for precision droop, channel-
current balancing, and overcurrent protection.
- 0.525V to 1.300V in 12.5mV Steps
A unity gain, differential amplifier is provided for remote
voltage sensing. Any potential difference between remote
and local grounds can be completely eliminated using the
remote-sense amplifier. Eliminating ground differences
improves regulation and protection accuracy. The threshold-
sensitive enable input is available to accurately coordinate
the start up of the ISL6564A with any other voltage rail.
Dynamic-VID™ technology allows seamless on-the-fly VID
changes. The offset pin allows accurate voltage offset
settings that are independent of VID setting. The ISL6564A
uses a 5V bias and has a built-in shunt regulator to allow
12V bias using only a small external limiting resistor.
• Threshold-Sensitive Enable Function for Power
Sequencing Control
• Overcurrent Protection
• Overvoltage Protection
- No Additional External Components Needed
- OVP Pin to Drive Crowbar Device
• 1, 2, 3, or 4 Phase Operation
• Up to 1.5MHz Per Phase Operation (>6MHz Ripple)
• QFN Package
- Compliant to JEDEC PUB95 MO-220 QFN - Quad Flat
No Leads - Product Outline
Ordering Information
- QFN Near Chip Scale Package Footprint; Improves
PCB Efficiency, Thinner in Profile
PART
TEMP.
(°C)
PKG.
PART NUMBER MARKING
PACKAGE DWG. #
• Pb-Free Plus Anneal Available (RoHS Compliant)
ISL6564ACRZ* ISL6564 ACRZ 0 to +70 40 Ld 6x6 QFN L40.6x6
(Note) (Pb-free)
ISL6564AIRZ* ISL6564 AIRZ -40 to +85 40 Ld 6x6 QFN L40.6x6
(Note)
(Pb-free)
*Add “-T” suffix for tape and reel.
NOTE: Intersil Pb-free plus anneal products employ special Pb-free
material sets; molding compounds/die attach materials and 100%
matte tin plate termination finish, which are RoHS compliant and
compatible with both SnPb and Pb-free soldering operations. Intersil
Pb-free products are MSL classified at Pb-free peak reflow
temperatures that meet or exceed the Pb-free requirements of
IPC/JEDEC J STD-020.
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2006-2007. All Rights Reserved. Dynamic VID™ is a trademark of Intersil Americas Inc.
All other trademarks mentioned are the property of their respective owners.
1
ISL6564A
Pinout
ISL6564A
(40 LD QFN)
TOP VIEW
40 39 38 37 36 35 34 33 32 31
VID5
VID4
VID3
VID2
VID1
VID0
1
2
3
4
5
6
7
8
9
30 ISEN4+
29 ISEN4-
28 ISEN2-
27
26
25
24
23
22
ISEN2+
PWM2
PWM1
ISEN1+
ISEN1-
ISEN3-
GND
GND
OFS
IOUT
DAC 10
21 ISEN3+
11 12 13 14 15 16 17 18 19 20
FN6285.1
March 20, 2007
2
ISL6564A
ISL6564A Block Diagram
OVP
PGOOD
ENLL
VCC
VDIFF
DRVEN
RGND
1.29V
OVP
R
S
POWER-ON
x1
LATCH
RESET (POR)
EN
VSEN
Q
THREE-STATE
SOFT-START
AND
FAULT LOGIC
CLOCK AND
SAWTOOTH
GENERATOR
OVP
FS
∑
+200mV
PWM1
PWM
∑
PWM2
OFS
OFFSET
PWM
REF
DAC
∑
PWM3
PWM4
PWM
PWM
VID5
VID4
∑
DYNAMIC
VID
VID3
VID2
VID1
VID0
D/A
E/A
CHANNEL
CURRENT
BALANCE
CHANNEL
DETECT
COMP
FB
ISEN1+
ISEN1-
I_TRIP
OC
SAMPLE
&
HOLD
IDROOP
ISEN2+
ISEN2-
CHANNEL
∑
CURRENT
SENSE
ISEN3+
ISEN3-
ISEN4+
ISEN4-
I_TOT
IOUT
GND
FN6285.1
March 20, 2007
3
ISL6564A
Typical Application for Voltage Regulation without Droop Using r
Sensing
DS(ON)
+12V
VIN
VCC
BOOT
UGATE
PVCC
PHASE
ISL6612
DRIVER
PWM
LGATE
GND
+5V
FB
REF
DAC
COMP
+12V
IDROOP
VIN
BOOT
VCC
VDIFF
VSEN
VCC
EN
RGND
ENLL
VIDPGOOD
UGATE
PHASE
PVCC
PGOOD
OVP
ISL6564A
ISL6612
DRIVER
ISEN1+
ISEN1-
VID5
PWM
LGATE
GND
VID4
VID3
PWM1
PWM2
ISEN2+
ISEN2-
VID2
VID1
PWM3
ISEN3+
ISEN3-
+12V
VIN
VID0
µP
LOAD
VCC
DRVEN
OFS
FS
PWM4
ISEN4+
ISEN4-
BOOT
UGATE
PVCC
PWM
IOUT
GND
PHASE
ISL6612
DRIVER
LGATE
GND
NTC
NETWORK
+12V
VOLTAGE PROPOTIONAL
TO LOAD CURRENT
VCC
VIN
BOOT
UGATE
PHASE
PVCC
PWM
ISL6612
DRIVER
LGATE
GND
FN6285.1
March 20, 2007
4
ISL6564A
Typical Application for Voltage Regulation without Droop Using DCR Sensing
+12V
VIN
VCC
BOOT
UGATE
PVCC
PHASE
ISL6612
DRIVER
PWM
LGATE
GND
+5V
FB
REF
DAC
COMP
+12V
IDROOP
VIN
VIN
VIN
VCC
BOOT
VDIFF
VSEN
VCC
EN
RGND
ENLL
UGATE
PHASE
VIDPGOOD
PVCC
PGOOD
OVP
ISL6564A
ISL6612
DRIVER
ISEN1+
ISEN1-
VID5
PWM
LGATE
GND
VID4
VID3
PWM1
PWM2
ISEN2+
ISEN2-
VID2
VID1
PWM3
ISEN3+
ISEN3-
+12V
VID0
µP
LOAD
VCC
DRVEN
OFS
FS
PWM4
ISEN4+
ISEN4-
BOOT
UGATE
PVCC
PWM
GND
IOUT
PHASE
ISL6612
DRIVER
LGATE
GND
NTC
NETWORK
VOLTAGE PROPOTIONAL
TO LOAD CURRENT
+12V
VCC
BOOT
UGATE
PHASE
PVCC
PWM
ISL6612
DRIVER
LGATE
GND
FN6285.1
March 20, 2007
5
ISL6564A
Typical Application for Load Line Regulation Using r
Sensing and External NTC
DS(ON)
+12V
VCC
VIN
BOOT
UGATE
PHASE
PVCC
ISL6612
DRIVER
PWM
LGATE
GND
NTC
THERMISTOR
+5V
FB
COMP
REF
DAC
+12V
IDROOP
VIN
BOOT
VCC
VDIFF
VSEN
RGND
ENLL
VCC
VIDPGOOD
PGOOD
OVP
UGATE
PHASE
PVCC
PWM
ISL6564A
ISL6612
DRIVER
ISEN1+
ISEN1-
VID4
VID3
VID2
VID1
VID0
LGATE
GND
PWM1
PWM2
ISEN2+
ISEN2-
+12V
PWM3
ISEN3+
ISEN3-
VIN
VID12.5
DRVEN
µP
LOAD
VCC
OFS
FS
PWM4
ISEN4+
ISEN4-
BOOT
UGATE
PVCC
PWM
PHASE
ISL6612
DRIVER
GND
IOUT
EN
+12V
LGATE
GND
+12V
NTC
NETWORK
VCC
VIN
BOOT
VOLTAGE PROPOTIONAL
TO LOAD CURRENT
UGATE
PHASE
PVCC
PWM
ISL6612
DRIVER
LGATE
GND
FN6285.1
March 20, 2007
6
ISL6564A
Typical Application for Load Line Regulation Using DCR Sensing and External NTC
+12V
VIN
VCC
BOOT
UGATE
PVCC
PHASE
ISL6612
DRIVER
PWM
LGATE
GND
NTC
THERMISTOR
+5V
FB
COMP
REF
DAC
+12V
IDROOP
VIN
VIN
VIN
VCC
BOOT
VDIFF
VSEN
RGND
ENLL
VCC
VIDPGOOD
UGATE
PHASE
PVCC
PWM
PGOOD
OVP
ISL6564A
ISL6612
DRIVER
ISEN1+
ISEN1-
VID4
LGATE
GND
VID3
VID2
PWM1
PWM2
ISEN2+
ISEN2-
VID1
VID0
+12V
PWM3
ISEN3+
ISEN3-
VID12.5
DRVEN
µP
LOAD
VCC
OFS
FS
PWM4
ISEN4+
ISEN4-
BOOT
UGATE
PVCC
PWM
PHASE
ISL6612
DRIVER
IOUT
GND EN
+12V
LGATE
GND
NTC
NETWORK
+12V
VCC
BOOT
VOLTAGE PROPOTIONAL
TO LOAD CURRENT
UGATE
PHASE
PVCC
PWM
ISL6612
DRIVER
LGATE
GND
FN6285.1
March 20, 2007
7
ISL6564A
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5.5V
Input and Output Voltage (except OVP). . GND -0.3V to V + 0.3V
Thermal Resistance
θ
(°C/W)
32
θ
(°C/W)
3.5
JA
JC
CC
QFN Package (Notes 1, 2). . . . . . . . . .
OVP Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+15V
ESD (Human body model). . . . . . . . . . . . . . . . . . . . . . . . . . . . .>4kV
ESD (Machine model) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .>300V
ESD (Charged device model) . . . . . . . . . . . . . . . . . . . . . . . . . .>2kV
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . +150°C
Maximum Storage Temperature Range. . . . . . . . . - 65°C to +150°C
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . +300°C
Operating Conditions
Supply Voltage, VCC (5V bias mode, Note 3) . . . . . . . . . . +5V ±5%
Ambient Temperature (ISL6564ACRZ) . . . . . . . . . . . . 0°C to +70°C
Ambient Temperature (ISL6564AIRZ) . . . . . . . . . . . .-40°C to +85°C
CAUTION: Stress above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational section of this specification is not implied.
NOTES:
1. θ is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
JA
Tech Brief TB379.
2. For θ , the “case temp” location is the center of the exposed metal pad on the package underside.
JC
Electrical Specifications Operating Conditions: VCC = 5V or ICC < 25mA (Note 3). Unless Otherwise Specified.
PARAMETER
VCC SUPPLY CURRENT
Nominal Supply
TEST CONDITIONS
MIN
TYP MAX UNITS
VCC = 5VDC; EN = 5VDC; R = 100kΩ,
-
-
14
10
18
14
mA
mA
T
ISEN1 = ISEN2 = ISEN3 = ISEN4 = -70μA
Shutdown Supply
VCC = 5VDC; EN = 0VDC; R = 100kΩ
T
POWER-ON RESET AND ENABLE
POR Threshold
VCC rising
VCC falling
EN rising
4.20
3.60
1.26
110
1.12
-
4.31
3.80
1.29
125
1.16
-
4.50
4.00
1.32
135
1.20
0.4
-
V
V
ENABLE Threshold
V
Hysteresis
Fault reset
mV
V
ENLL Input Logic Low Level
ENLL input Logic High Level
ENLL Leakage Current
V
0.8
-
-
V
ENLL = 5V
(Note 4)
-
1
μA
REFERENCE VOLTAGE AND DAC
System Accuracy (VID = 1.V-1.3V)
-0.5
-0.9
-55
-
-
0.5
0.9
-35
0.4
-
%VID
%VID
μA
V
System Accuracy (VID = 0.525V-0.9875V) (Note 4)
VID Pull-Up
-
-45
VID Input Low Level
-
VID Input High Level
0.8
-200
1.0
-50
-
V
DAC Source/Sink Current
VID Input Voltage when Floated
REF Source/Sink Current
PIN-ADJUSTABLE OFFSET
Voltage at OFS Pin
VID = 010100
-
1.15
-
200
1.30
50
μA
V
μA
Offset resistor connected to ground
388
2.91
-
400
3.0
-
412
3.09
50
mV
V
VCC = 5.000V, offset resistor connected to VCC
Maximum OFS Source and Sink Current
μA
FN6285.1
March 20, 2007
8
ISL6564A
Electrical Specifications Operating Conditions: VCC = 5V or ICC < 25mA (Note 3). Unless Otherwise Specified. (Continued)
PARAMETER
TEST CONDITIONS
MIN
TYP MAX UNITS
OSCILLATOR
Accuracy
R
= 100kΩ
260
300
-
345
kHz
MHz
V
T
Adjustment Range
0.08
1.5
Sawtooth Amplitude
-
-
2
-
-
Max Duty Cycle
66.7
%
ERROR AMPLIFIER
Open-Loop Gain
R
C
= 10kΩ to ground
-
-
80
18
4.3
-
-
dB
MHz
V
L
L
Open-Loop Bandwidth
Maximum Output Voltage
Output High Voltage @ 2mA
Output Low Voltage @ 2mA
REMOTE-SENSE AMPLIFIER
Bandwidth
= 100pF, R = 10kΩ to ground
-
L
4.0
3.7
-
-
-
V
-
1.35
V
-
20
-
MHz
V
Output Voltage @ 1mA Load
PWM OUTPUT
VSEN - RGND = 2.5V
2.48
2.50
2.52
PWM Output Voltage LOW
PWM Output Voltage HIGH
DRIVER ENABLE OUTPUT
DRVEN Output Voltage LOW
DRVEN Output Voltage HIGH
SENSE CURRENT OUTPUT
Sensed Current Accuracy
Overcurrent Trip Level
Maximum Voltage at IDROOP and IOUT
Iload = ±500μA
Iload = ±500μA
-
-
-
0.5
-
V
V
4.0
Iload = ±1mA
Iload = ±1mA
-
-
-
0.3
-
V
V
4.0
ISEN1 = ISEN2 = ISEN3 = ISEN4 = 80μA
75
95
85
93
125
2
μA
μA
V
110
VCC = 4.5V (Note 5)
POWER GOOD AND PROTECTION MONITORS
PGOOD Low Voltage
I
= 4mA
-
70
180
1.40
1.7
-
-
0.3
80
V
%VID
mV
V
PGOOD
Undervoltage Offset From VID
Overvoltage Threshold
VSEN falling
75
Voltage above VID, after soft-start (Note 6)
Before enable
200
1.45
1.8
0.6
1.5
3.6
230
1.50
1.87
-
VCC < POR threshold
V
Overvoltage Reset Voltage
VCC ≥ POR threshold, VSEN falling
VCC < POR threshold
V
-
-
V
OVP Drive Voltage
NOTES:
I
= -10mA, VCC = 5V
3.0
5.0
V
OVP
3. When using the internal shunt regulator, VCC is clamped to 6.2V (max). Current must be limited to 25mA or less.
4. These parts are designed and adjusted for accuracy with all errors in the voltage loop included.
5. Guaranteed by design.
6. During soft-start, VDAC rises from 0 to VID. The overvoltage trip level is the higher of 1.5V and VDAC + 0.2V.
FN6285.1
March 20, 2007
9
ISL6564A
FB and COMP
Functional Pin Description
Inverting input and output of the error amplifier, respectively.
FB is connected to VDIFF through a resistor. A negative
current, proportional to output current is present on the FB
pin. A properly sized resistor between VDIFF and FB sets
the load line (droop). The droop scale factor is set by the
VCC
Supplies all the power necessary to operate the chip. The
controller starts to operate when the voltage on this pin
exceeds the rising POR threshold and shuts down when the
voltage on this pin drops below the falling POR threshold.
Connect this pin directly to a +5V supply or through a series
300Ω resistor to a +12V supply.
ratio of the ISEN resistors and the lower MOSFET r
.
DS(ON)
COMP is tied back to FB through an external R-C network
with no DC connection to compensate the regulator.
GND
DAC and REF
Bias and reference ground for the IC.
The DAC output pin is the output of the precision internal
DAC reference. The REF input pin is the positive input of the
Error Amp. In typical applications, a 1kΩ, 1% resistor is used
between DAC and REF to generate a precise offset voltage.
This voltage is proportional to the offset current determined
by the offset resistor from OFS to ground or VCC. A
EN
This pin is a threshold-sensitive enable input for the
controller. Connecting the 12V supply to EN through an
appropriate resistor divider provides a means to synchronize
power-up of the controller and the MOSFET driver ICs.
When EN is driven above 1.29V, the ISL6564A is active
depending on status of ENLL, the internal POR, and pending
fault states. Driving EN below 1.16V will clear all fault states
and prime the ISL6564A to soft-start when re-enabled.
capacitor is used between REF and ground to smooth the
voltage transition during Dynamic VID™ operations.
PWM1, PWM2, PWM3, PWM4
Pulse-width modulation outputs. Connect these pins to the
PWM input pins of the Intersil driver IC. The number of
active channels is determined by the state of PWM3 and
PWM4. Leave PWM4 unconnected and tie PWM3 to VCC to
configure for 2-phase operation. Tie PWM4 to VCC to
configure for 3-phase operation. Tie both PWM4 and PWM3
to high for 1-phase operation.
ENLL
This pin is a logic-level enable input for the controller. When
asserted to a logic high, the ISL6564A is active depending
on status of EN, the internal POR, VID inputs and pending
fault states. Deasserting ENLL will clear all fault states and
prime the ISL6564A to soft-start when re-enabled.
ISEN1+, ISEN1-; ISEN2+, ISEN2-; ISEN3+, ISEN3-;
ISEN4+, ISEN4-
When floating, ENLL pin will be pulled to high internally with
a typical voltage as 1.15V.
The ISEN+ and ISEN- pins are current sense inputs to
individual differential amplifiers. The sensed current is used
as a reference for channel balancing, protection, and
regulation. Inactive channels should have their respective
sense inputs left open (for example, for 3-phase operation
open ISEN4+).
FS
A resistor, RT, placed from FS to ground will set the
switching frequency. There is an inverse relationship
between the value of the resistor and the switching
frequency. See Figure 20 and Equation 30.
VID5, VID4, VID3, VID2, VID1, and VID0
For DCR sensing, connect each ISEN- pin to the node
between the RC sense elements. Tie the ISEN+ pin to the
These are the inputs to the internal DAC that provides the
reference voltage for output regulation. Connect these pins
either to open-drain outputs with or without external pull-up
resistors or to active-pull-up outputs. VID5-VID0 have 45µA
internal pull-up current sources that diminish to zero as the
voltage rises above the logic-high level. These inputs can be
pulled up as high as VCC plus 0.3V.
other end of the sense capacitor through a resistor, R
.
ISEN
The voltage across the sense capacitor is proportional to the
inductor current. The sensed current is proportional to the
output current, and scaled by the DCR of the inductor,
divided by R
.
ISEN
When configured for r
current sensing, the ISEN1-,
DS(ON)
ISEN2-, ISEN3-, and ISEN4- pins are grounded at the lower
MOSFET sources. The ISEN1+, ISEN2+, ISEN3+, and
ISEN4+ pins are then held at a virtual ground, such that a
resistor connected between them, and the drain terminal of
the associated lower MOSFET, will carry a current
VDIFF, VSEN, and RGND
VSEN and RGND form the precision differential remote-
sense amplifier. This amplifier converts the differential
voltage of the remote output to a single-ended voltage
referenced to local ground. VDIFF is the amplifier’s output
and the input to the regulation and protection circuitry.
Connect VSEN and RGND to the sense pins of the remote
load.
proportional to the current flowing through that channel. The
current is determined by the negative voltage developed
across the lower MOSFET’s r
, which is the channel
DS(ON)
current scaled by r
DS(ON)
.
FN6285.1
March 20, 2007
10
ISL6564A
PGOOD
PGOOD is used as an indication of the end of soft-start per
the microprocessor specification. It is an open-drain logic
output that is low impedance until the soft-start is completed.
It will be pulled low again once the undervoltage point is
reached.
IL1 + IL2 + IL3, 7A/DIV
IL3, 7A/DIV
PWM3, 5V/DIV
OFS
IL2, 7A/DIV
The OFS pin provides a means to program a DC offset
current for generating a DC offset voltage at the REF input.
The offset current is generated via an external resistor and
precision internal voltage references. The polarity of the
offset is selected by connecting the resistor to GND or VCC.
For no offset, the OFS pin should be left unterminated.
PWM2, 5V/DIV
IL1, 7A/DIV
PWM1, 5V/DIV
OVP
1µs/DIV
Overvoltage protection pin. This pin pulls to VCC and is
latched when an overvoltage condition is detected. Connect
FIGURE 1. PWM AND INDUCTOR-CURRENT WAVEFORMS
FOR 3-PHASE CONVERTER
this pin to the gate of an SCR or MOSFET tied from V or
IN
Interleaving
V
to ground to prevent damage to the load. This pin may
OUT
be pulled above VCC as high as 15V to ground with an
external resistor. However, it is only capable of pulling low
when VCC is above 2V.
The switching of each channel in a multiphase converter is
timed to be symmetrically out of phase with each of the other
channels. In a 3-phase converter, each channel switches 1/3
cycle after the previous channel and 1/3 cycle before the
following channel. As a result, the three-phase converter has
a combined ripple frequency three times greater than the
ripple frequency of any one phase. In addition, the peak-to-
peak amplitude of the combined inductor currents is reduced
in proportion to the number of phases (Equations 1 and 2).
Increased ripple frequency and lower ripple amplitude mean
that the designer can use less per-channel inductance and
lower total output capacitance for any performance
specification.
DRVEN
Driver enable pin. This pin can be used to enable the drivers
which have enable pins such as ISL6605 or ISL6608. If
ISL6564A is used with Intersil ISL6612 drivers, it’s not
necessary to use this pin.
IDROOP and IOUT
IDROOP and IOUT are the output pins of sensed average
channel current which is proportional to load current. They
are designed for flexible application purposes.
Figure 1 illustrates the multiplicative effect on output ripple
frequency. The three channel currents (IL1, IL2, and IL3)
combine to form the AC ripple current and the DC load
current. The ripple component has three times the ripple
frequency of each individual channel current. Each PWM
pulse is terminated 1/3 of a cycle after the PWM pulse of the
previous phase. The peak-to-peak current for each phase is
about 7A, and the DC components of the inductor currents
combine to feed the load.
In the application which does not require loadline, leave
IDROOP pin open. In the application which requires load
line, connect IDROOP pin to FB so that the sensed average
current will flow through the resistor between FB and VDIFF
to create a voltage drop which is proportional to load current.
IOUT is typically used for load current indication.
Operation
Multiphase Power Conversion
To understand the reduction of ripple current amplitude in the
multiphase circuit, examine the equation representing an
individual channel’s peak-to-peak inductor current.
Microprocessor load current profiles have changed to the
point that the advantages of multiphase power conversion
are impossible to ignore. The technical challenges
associated with producing a single-phase converter which is
both cost-effective and thermally viable have forced a
change to the cost-saving approach of multiphase. The
ISL6564A controller helps reduce the complexity of
implementation by integrating vital functions and requiring
minimal output components. The block diagrams on pages
3, 4, 5, 6, and 7 provide top level views of multiphase power
conversion using the ISL6564A controller.
(V – V
) V
OUT
IN
OUT
(EQ. 1)
I
= -----------------------------------------------------
PP
Lf
V
S
IN
In Equation 1, V and V
IN
are the input and output
OUT
voltages respectively, L is the single-channel inductor value,
and f is the switching frequency.
S
FN6285.1
March 20, 2007
11
ISL6564A
PWM Operation
INPUT-CAPACITOR CURRENT, 10A/DIV
The timing of each converter leg is set by the number of
active channels. The default channel setting for the
ISL6564A is four. One switching cycle is defined as the time
between PWM1 pulse termination signals. The pulse
termination signal is an internally generated clock signal
which triggers the falling edge of PWM1. The cycle time of
the pulse termination signal is the inverse of the switching
frequency set by the resistor between the FS pin and
ground. Each cycle begins when the clock signal commands
the channel 1 PWM output to go low. The PWM1 transition
signals the channel-1 MOSFET driver to turn off the
channel 1 upper MOSFET and turn on the channel 1
synchronous MOSFET. In the default channel configuration,
the PWM2 pulse terminates 1/4 of a cycle after PWM1. The
PWM3 output follows another 1/4 of a cycle after PWM2.
PWM4 terminates another 1/4 of a cycle after PWM3.
CHANNEL 3
INPUT CURRENT
10A/DIV
CHANNEL 2
INPUT CURRENT
10A/DIV
CHANNEL 1
INPUT CURRENT
10A/DIV
1µs/DIV
FIGURE 2. CHANNEL INPUT CURRENTS AND INPUT-
CAPACITOR RMS CURRENT FOR 3-PHASE
CONVERTER
If PWM3 is connected to VCC, two channel operation is
selected and the PWM2 pulse terminates 1/2 of a cycle later.
Connecting PWM4 to VCC selects three channel operation
and the pulse-termination times are spaced in 1/3 cycle
increments. Connecting both PWM3 and PWM4 to VCC
selects single-channel operation.
The output capacitors conduct the ripple component of the
inductor current. In the case of multiphase converters, the
capacitor current is the sum of the ripple currents from each
of the individual channels. Compare Equation 1 to the
expression for the peak-to-peak current after the summation
of N symmetrically phase-shifted inductor currents in
Equation 2. Peak-to-peak ripple current decreases by an
amount proportional to the number of channels. Output-
voltage ripple is a function of capacitance, capacitor
equivalent series resistance (ESR), and inductor ripple
current. Reducing the inductor ripple current allows the
designer to use fewer or less costly output capacitors.
Once a PWM signal transitions low, it is held low for a
minimum of 1/3 cycle. This forced off time is required to
ensure an accurate current sample. Current sensing is
described in the next section. After the forced off time
expires, the PWM output is enabled. The PWM output state
is driven by the position of the error amplifier output signal,
V
, minus the current correction signal relative to the
COMP
(V – N V
) V
OUT
IN
OUT
sawtooth ramp as illustrated in Figure 7. When the modified
voltage crosses the sawtooth ramp, the PWM output
I
= -----------------------------------------------------------
(EQ. 2)
C, PP
Lf
V
S
IN
V
COMP
transitions high. The MOSFET driver detects the change in
state of the PWM signal and turns off the synchronous
MOSFET and turns on the upper MOSFET. The PWM signal
will remain high until the pulse termination signal marks the
beginning of the next cycle by triggering the PWM signal low.
Another benefit of interleaving is to reduce input ripple
current. Input capacitance is determined in part by the
maximum input ripple current. Multiphase topologies can
improve overall system cost and size by lowering input ripple
current and allowing the designer to reduce the cost of input
capacitance. The example in Figure 2 illustrates input
currents from a three-phase converter combining to reduce
the total input ripple current.
Current Sampling
During the forced off-time following a PWM transition low,
the associated channel current sense amplifier uses the
ISEN inputs to reproduce a signal proportional to the
The converter depicted in Figure 2 delivers 36A to a 1.5V load
from a 12V input. The RMS input capacitor current is 5.9A.
Compare this to a single-phase converter also stepping down
12V to 1.5V at 36A. The single-phase converter has 11.9A
RMS input capacitor current. The single-phase converter
must use an input capacitor bank with twice the RMS current
capacity as the equivalent three-phase converter.
inductor current, I . No matter the current sense method, the
L
sense current, I
SEN
, is simply a scaled version of the
inductor current. Coincident with the falling edge of the PWM
signal, the sample and hold circuitry samples I , as
SEN
illustrated in Figure 3. The sample window hold time, t
,
HOLD
is fixed and equal to 1/3 of the switching period, t
t
.
SW
1
SW
Figures 21, 22 and 23 in the section entitled Input Capacitor
Selection can be used to determine the input-capacitor RMS
current based on load current, duty cycle, and the number of
channels. They are provided as aids in determining the
optimal input capacitor solution. Figure 24 shows the single
phase input-capacitor RMS current for comparison.
----------
-----------------
t
=
=
(EQ. 3)
HOLD
3
3 ⋅ f
SW
Therefore, the sample current, I , is proportional to the
n
output current and held for one switching cycle. The sample
current is used for current balance, load-line regulation, and
overcurrent protection.
FN6285.1
March 20, 2007
12
ISL6564A
V
IN
I
(s)
L
I
L
L
DCR
V
OUT
ISL6207
INDUCTOR
-
C
OUT
PWM
V
L
-
(s)
V
C
R
I
C
SEN
PWM(n)
ISL6564A INTERNAL CIRCUIT
t
HOLD
R
ISEN(n)
(PTC)
SAMPLE CURRENT, I
n
I
n
SWITCHING PERIOD
SAMPLE
&
HOLD
ISEN-(n)
TIME
+
-
FIGURE 3. SAMPLE AND HOLD TIMING
ISEN+(n)
Current Sensing
The ISL6564A supports inductor DCR sensing, MOSFET
sensing, or resistive sensing techniques. The
DCR
I
-----------------
= I
SEN
L
R
ISEN
r
DS(ON)
internal circuitry, shown in Figures 4, 5, and 6, represents
channel n of an N-channel converter. This circuitry is
repeated for each channel in the converter, but may not be
active depending on the status of the PWM3 and PWM4
pins, as described in the PWM Operation section.
FIGURE 4. DCR SENSING CONFIGURATION
The capacitor voltage V , is then replicated across the
C
sense resistor R
. The current through the sense resistor
ISEN
is proportional to the inductor current. Equation 6 shows the
proportion between the channel current and the sensed
INDUCTOR DCR SENSING
current I
, is driven by the value of the sense resistor
An inductor’s winding is characteristic of a distributed
resistance as measured by the DCR (Direct Current
Resistance) parameter. Consider the inductor DCR as a
separate lumped quantity, as shown in Figure 4. The
SEN
chosen and the DCR of the inductor.
DCR
-----------------
I
= I
⋅
(EQ. 6)
SEN
L
R
ISEN
channel current I , flowing through the inductor, will also
L
pass through the DCR. Equation 4 shows the s-domain
DCR varies with temperature, so a Positive Temperature
Coefficient (PTC) resistor should be selected for the sense
equivalent voltage across the inductor V .
L
resistor R
.
V
= I ⋅ (s ⋅ L + DCR)
L
(EQ. 4)
ISEN
L
RESISTIVE SENSING
A simple R-C network across the inductor extracts the DCR
voltage, as shown in Figure 4.
If DCR sensing is not utilized, independent current-sense
resistors in series with each output inductor can serve as the
sense element (see Figure 5). This technique is more
accurate, but reduces overall converter efficiency due to the
addition of a lossy element directly in the output path.
The voltage on the capacitor V , can be shown to be
C
proportional to the channel current I , see Equation 5.
L
L
⎛
⎝
⎞
-------------
s ⋅
+ 1 ⋅ (DCR ⋅ I )
L
⎠
(EQ. 5)
DCR
V
= --------------------------------------------------------------------
C
(s ⋅ RC + 1)
If the R-C network components are selected such that the
RC time constant matches the inductor L/DCR time
constant, then V is equal to the voltage drop across the
C
DCR.
FN6285.1
March 20, 2007
13
ISL6564A
Channel-Current Balance
I
L
L
The sampled currents I , from each active channel are
n
R
V
OUT
SENSE
summed together and divided by the number of active
C
OUT
channels. The resulting cycle average current I
, provides
AVG
a measure of the total load current demand on the converter
during each switching cycle. Channel current balance is
achieved by comparing the sampled current of each channel
to the cycle average current, and making an appropriate
adjustment to each channel pulse width based on the error.
Intersil’s patented current-balance method is illustrated in
Figure 7, with error correction for channel 1 represented. In
the figure, the cycle average current combines with the
ISL6564A INTERNAL CIRCUIT
R
ISEN(n)
I
n
SAMPLE
&
HOLD
ISEN-(n)
ISEN+(n)
+
-
channel 1 sample, I , to create an error signal I . The
1
ER
R
filtered error signal modifies the pulse width commanded by
SENSE
I
= I --------------------------
SEN
L
R
V
to correct any unbalance and force I toward zero.
ISEN
COMP
ER
The same method for error signal correction is applied to
each active channel.
FIGURE 5. SENSE RESISTOR IN SERIES WITH INDUCTORS
MOSFET r
DS(ON)
SENSING
The controller can also sense the channel load current by
sampling the voltage across the lower MOSFET r
+
V
COMP
+
-
DS(ON)
PWM1
-
(see Figure 6). The amplifier is ground-reference by
connecting the ISEN- input to the source of the lower
MOSFET. ISEN+ connects to the PHASE node through a
SAWTOOTH SIGNAL
FILTER f(jω)
I
I
I
*
*
4
3
2
I
resistor R
. The voltage across R is equivalent to
ER
ISEN ISEN
I
AVG
the voltage drop across the r
of the lower MOSFET
÷ N
DS(ON)
while it is conducting. The resulting current into the ISEN+
Σ
-
+
pin is proportional to the channel current I . The ISEN
L
current is then sampled and held after sufficient settling time.
I
1
The sampled current I , is used for channel-current balance,
n
NOTE: *Channels 3 and 4 are optional.
load-line regulation, and overcurrent protection. From
Figure 6, Equation 7 for I
is derived.
SEN
FIGURE 7. CHANNEL 1 PWM FUNCTION AND CURRENT-
BALANCE ADJUSTMENT
V
IN
r
DS(ON)
I
= I -------------------------
SEN
L
R
ISEN
Channel current balance is essential in realizing the thermal
advantage of multiphase operation. The heat generated in
down converting is dissipated over multiple devices and a
greater area. The designer avoids the complexity of driving
multiple parallel MOSFETs, and the expense of using heat
sinks and nonstandard magnetic materials.
I
n
I
L
SAMPLE
&
HOLD
ISEN+(n)
R
ISEN
-
(PTC)
I
r
L
DS(ON)
-
Voltage Regulation
ISEN-(n)
+
+
The integrating compensation network shown in Figure 8
assures that the steady-state error in the output voltage is
limited only to the error in the reference voltage (output of
the DAC) and offset errors in the OFS current source,
remote-sense and error amplifiers. Intersil specifies the
guaranteed tolerance of the ISL6564A to include the
combined tolerances of each of these elements.
N-CHANNEL
MOSFETs
ISL6564A INTERNAL CIRCUIT EXTERNAL CIRCUIT
FIGURE 6. MOSFET r
CURRENT-SENSING CIRCUIT
DS(ON)
r
DS(ON)
(EQ. 7)
I
= I ----------------------
The output of the error amplifier, V
, is compared to the
COMP
SEN
L
R
ISEN
sawtooth waveform to generate the PWM signals. The PWM
signals control the timing of the Intersil MOSFET drivers and
regulate the converter output to the specified reference
voltage. The internal and external circuitry which control
voltage regulation is illustrated in Figure 8.
where I is the channel current. Since MOSFET r
DS(ON)
increases with temperature, a PTC resistor should be
L
chosen for R
to compensate for this change.
ISEN
FN6285.1
March 20, 2007
14
ISL6564A
TABLE 1. VOLTAGE IDENTIFICATION (VID) CODES
EXTERNAL CIRCUIT
ISL6564A INTERNAL CIRCUIT
R
C
VID5
VID4 VID3
VID2
VID1
VID0
VDAC
C
C
COMP
DAC
400
mV
200
mV
100
mV
50
mv
25
mV
12.5
mV
R
REF
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
OFF
REF
C
REF
1.3000V
1.2875V
1.2750V
1.2625V
1.2500V
1.2375V
1.2250V
1.2125V
1.2000V
1.1875V
1.1750V
1.1625V
1.1500V
1.1375V
1.1250v
1.1125V
1.1000V
1.0875V
1.0750V
1.0625V
1.0500V
1.0375V
1.0250V
1.0125V
1.0000V
0.9875V
0.9750V
0.9625V
0.9500V
0.9375V
0.9250V
0.9125V
0.9000V
0.8875V
0.8750V
+
-
V
FB
COMP
ERROR AMPLIFIER
I
IDROOP
VDIFF
+
V
-
AVG
R
FB
DROOP
VSEN
RGND
V
V
+
OUT
+
-
-
OUT
DIFFERENTIAL
REMOTE-SENSE
AMPLIFIER
FIGURE 8. OUTPUT VOLTAGE AND LOAD-LINE
REGULATION WITH OFFSET ADJUSTMENT
The ISL6564A incorporates an internal differential remote-
sense amplifier in the feedback path. The amplifier removes
the voltage error encountered when measuring the output
voltage relative to the local controller ground reference point
resulting in a more accurate means of sensing output
voltage. Connect the microprocessor sense pins to the
non-inverting input, VSEN, and inverting input, RGND, of the
remote-sense amplifier. The remote-sense output, V
connected to the inverting input of the error amplifier through
an external resistor.
, is
DIFF
A digital to analog converter (DAC) generates a reference
voltage based on the state of logic signals at pins VID4
through VID12.5. The DAC decodes the a 6-bit logic signal
(VID) into one of the discrete voltages shown in Table 1.
Each VID input offers a 45μA pull-up to an internal 2.5V
source for use with open-drain outputs. The pull-up current
diminishes to zero above the logic threshold to protect
voltage-sensitive output devices. External pull-up resistors
can augment the pull-up current sources if case leakage into
the driving device is greater than 45μA.
Load-Line Regulation
Some microprocessor manufacturers require a precisely-
controlled output resistance. This dependence of output
voltage on load current is often termed “droop” or “load line”
regulation. By adding a well controlled output impedance,
the output voltage can effectively be level shifted in a
direction which works to achieve the load-line regulation
required by these manufacturers.
FN6285.1
March 20, 2007
15
ISL6564A
TABLE 1. VOLTAGE IDENTIFICATION (VID) CODES (Continued)
down so that a larger positive spike can be sustained without
crossing the upper specification limit.
VID5
VID4 VID3
VID2
VID1
VID0
VDAC
As shown in Figure 8, a current proportional to the average
400
mV
200
mV
100
mV
50
mv
25
mV
12.5
mV
current in all active channels, I
, flows from FB through a
AVG
load-line regulation resistor, R . The resulting voltage drop
across R is proportional to the output current, effectively
FB
creating an output voltage droop with a steady-state value
defined as
FB
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
0.8625V
0.8500V
0.8375V
0.8250V
0.8125V
0.8000V
0.7875V
0.7750V
0.7625V
0.7500V
0.7375V
0.7250V
0.7125V
0.7000V
0.6875V
0.6750V
0.6625V
0.6500V
0.6375V
0.6250V
0.6125V
0.6000V
0.5875V
0.5750V
0.5625V
0.5500V
0.5375V
0.525V
V
= I
R
AVG FB
(EQ. 8)
DROOP
The regulated output voltage is reduced by the droop voltage
. The output voltage as a function of load current is
V
DROOP
derived by combining Equation 8 with the appropriate
sample current expression defined by the current sense
method employed.
I
R
X
R
ISEN
⎛
⎞
⎟
⎠
OUT
4
V
= V
– V
– ------------- ----------------- R
⎜
OFFSET FB
(EQ. 9)
OUT
REF
⎝
Where V
is the reference voltage, V
is the
REF
programmed offset voltage, I
OFS
is the total output current
OUT
is the sense resistor in the ISEN line,
of the converter, R
ISEN
and R is the feedback resistor. R has a value of DCR,
FB
, or R
X
r
depending on the sensing method.
SENSE
DS(ON)
Output-Voltage Offset Programming
The ISL6564A allows the designer to accurately adjust the
offset voltage. When a resistor, R
, is connected between
OFS
OFS to VCC, the voltage across it is regulated to 2.0V. This
causes a proportional current (I ) to flow into OFS. If
OFS
is connected to ground, the voltage across it is
R
OFS
regulated to 0.5V, and I
flows out of OFS. A resistor
OFS
between DAC and REF, R
, is selected so that the
REF
product (I
x R
) is equal to the desired offset voltage.
OFS
OFS
These functions are shown in Figure 9.
As it may be noticed in Figure 9, the OFSOUT pin must be
connected to the REF pin for this current injection to function
in ISL6564A. The current flow through R
creates an
REF
offset at the REF pin, which is ultimately duplicated at the
output of the regulator.
Once the desired output offset voltage has been determined,
use the following formulas to set R
:
OFS
For Positive Offset (connect R
to VCC):
OFS
In other cases, the designer may determine that a more
cost-effective solution can be achieved by adding droop.
Droop can help to reduce the output-voltage spike that
results from fast load-current demand changes.
2 × R
REF
(EQ. 10)
(EQ. 11)
--------------------------
OFFSET
R
=
OFS
V
For Negative Offset (connect R
to GND):
OFS
The magnitude of the spike is dictated by the ESR and ESL
of the output capacitors selected. By positioning the no-load
voltage level near the upper specification limit, a larger
negative spike can be sustained without crossing the lower
limit. By adding a well controlled output impedance, the
output voltage under load can effectively be level shifted
0.5 × R
REF
-----------------------------
OFFSET
R
=
OFS
V
FN6285.1
March 20, 2007
16
ISL6564A
Assuming the microprocessor controls the VID change at 1
bit every T , the relationship between the time constant of
FB
VID
R
and C
network and T
VID
is given by Equation 12.
REF
REF
C
R
= k T
(EQ. 12)
REF REF
VID
DAC
DYNAMIC
VID D/A
R
REF
Where, T
= 4μs, k is the number of the internal VID
VID
change cycle. If Typically R
E/A
is selected to be 1kΩ, the
REF
allowable delay time for VR to respond to new VID code is 5
REF
VID change cycles (totally 20μs), the value of C
should
REF
be 22nF based on Equation 12.
Operation Initialization
VCC
OR
GND
Prior to converter initialization, proper conditions must exist
on the enable inputs and VCC. When the conditions are met,
the controller begins soft-start. Once the output voltage is
within the proper window of operation, PGOOD asserts
logic 1.
-
R
2.0V
OFS
+
+
-
0.5V
Enable and Disable
OFS
ISL6564A
While in shutdown mode, the PWM outputs are held in a
high-impedance state to assure the drivers remain off. The
following input conditions must be met before the ISL6564A
is released from shutdown mode.
GND
VCC
FIGURE 9. OUTPUT VOLTAGE OFFSET PROGRAMMING
WITH ISL6564A
ISL6564A INTERNAL CIRCUIT
EXTERNAL CIRCUIT
Dynamic VID
Modern microprocessors need to make changes to their
core voltage as part of normal operation. They direct the
core-voltage regulator to do this by making changes to the
VID inputs during regulator operation. The power
management solution is required to monitor the DAC inputs
and respond to on-the-fly VID changes in a controlled
manner. Supervising the safe output voltage transition within
the DAC range of the processor without discontinuity or
disruption is a necessary function of the core-voltage
regulator.
+12V
VCC
10.7kΩ
ENABLE
COMPARATOR
POR
CIRCUIT
EN
+
-
1.40kΩ
1.23V
The ISL6564A checks the VID inputs at the three edges of
16MHz clock. If the VID code is found to have changed, the
controller waits half of a complete cycle before executing a
12.5mV change. If during the half-cycle wait period, the
difference between DAC level and the new VID code
changes sign, no change is made. If the VID code is more
than 1 bit higher or lower than the DAC (not recommended),
the controller will execute step-up and step down VID
change at a speed of 12.5mV every 4μs until VID and DAC
are equal.
ENLL
SOFT-START
AND
FAULT LOGIC
FIGURE 10. POWER SEQUENCING USING THRESHOLD-
SENSITIVE ENABLE (EN) FUNCTION
1. The bias voltage applied at VCC must reach the internal
power-on reset (POR) rising threshold. Once this
threshold is reached, proper operation of all aspects of
the ISL6564A is guaranteed. Hysteresis between the
rising and falling thresholds assure that once enabled,
the ISL6564A will not inadvertently turn off unless the
bias voltage drops substantially (see Electrical
Specifications).
In order to ensure the smooth transition of output voltage
during VID change, a VID step change smoothing network
composed of R
and C
is required for an ISL6564A
based voltage regulator. The selection of R is based on
REF
REF
REF
the desired offset as detailed above in Output-Voltage Offset
Programming. The selection of C is based on the time
REF
duration for 1 bit VID change and the allowable delay time.
FN6285.1
March 20, 2007
17
ISL6564A
2. The ISL6564A features an enable input (EN) for power
sequencing between the controller bias voltage and
another voltage rail. The enable comparator holds the
ISL6564A in shutdown until the voltage at EN rises above
1.29V. The enable comparator has about 125mV of
hysteresis to prevent bounce. It is important that the
driver ICs reach their POR level before the ISL6564A
becomes enabled. The schematic in Figure 10
demonstrates sequencing the ISL6564A with the
ISL66Xx family of Intersil MOSFET drivers, which require
12V bias.
VOUT, 500mV/DIV
EN, 5V/DIV
3. The voltage on ENLL must be logic high to enable the
controller. This pin is typically connected to the
VID_PGOOD.
4. The VID code must not be 111111. This code signals the
controller that no load is present. The controller will enter
shut-down mode after receiving this code and will
execute soft-start upon receiving any other code. This
code can be used to enable or disable the controller but
it is not recommended. After receiving this code, the
controller executes a 2-cycle delay before changing the
overvoltage trip level to the shut-down level and disabling
PWM. Overvoltage shutdown can not be reset using this
code.
500µs/DIV
FIGURE 11. SOFT-START WAVEFORMS WITH AN UN-BIASED
OUTPUT
Fault Monitoring and Protection
The ISL6564A actively monitors output voltage and current
to detect fault conditions. Fault monitors trigger protective
measures to prevent damage to a microprocessor load. One
common power good indicator is provided for linking to
external system monitors. The schematic in Figure 12
outlines the interaction between the fault monitors and the
power good signal.
To enable the controller, VCC must be greater than the POR
threshold; the voltage on EN must be greater than 1.29V;
For ISL6564A, ENLL must be logic high; and VID cannot be
equal to 111111. When each of these conditions is true, the
controller immediately begins the soft-start sequence.
Power Good Signal
Soft-Start
The power good pin (PGOOD) is an open-drain logic output
indication that the converter is operating after soft-start.
PGOOD pulls low during shutdown and releases high after a
successful soft-start. PGOOD will only transition low when
an undervoltage condition is detected or the controller is
disabled by a reset from EN, ENLL, POR, or one of the
no-CPU VID codes. After an undervoltage event, PGOOD
will return high unless the controller has been disabled.
PGOOD does not automatically transition low upon
detection of an overvoltage condition.
During soft-start, the DAC voltage ramps linearly from zero
to the programmed VID level as shown in Figure 11. The
PWM signals remain in the high-impedance state until the
controller detects that the ramping DAC level has reached
the pre-bias output-voltage level. This protects the system
against the large, negative inductor currents that would
otherwise occur when starting with a pre-existing charge on
the output as the controller attempted to regulate to zero
volts at the beginning of the soft-start cycle. The soft-start
time, t , begins with a delay period equal to 64 switching
SS
cycles followed by a linear ramp with a fixed rate at a speed
of 12.5mV/32µs.
t
= (2560)VID
(EQ. 13)
SS
Equation 13 can be used to calculate the soft-start time. For
example, when VID is set to 1.2V, the soft-start time will be
3.072ms.
A 100mV offset exists on the remote-sense amplifier at the
beginning of soft-start and ramps to zero during the first 640
cycles of soft-start (704 cycles following enable). This
prevents the large inrush current that would otherwise occur
should the output voltage start out with a slight negative
bias.
FN6285.1
March 20, 2007
18
ISL6564A
reoccurs, the ISL6564A will again command the lower
MOSFETs to turn on. The ISL6564A will continue to protect
the load in this fashion as long as the overvoltage condition
recurs.
PGOOD
110µA
-
Simultaneous to the protective action of the PWM outputs,
the OVP pin pulls to VCC delivering up to 100mA to the gate
of a crowbar MOSFET or SCR placed either on the input rail
or the output rail. Turning on the MOSFET or SCR collapses
the power rail and causes a fuse placed further up stream to
blow. The fuse must be sized such that the MOSFET or SCR
will not overheat before the fuse blows. The OVP pin is
tolerant to 12V (see Absolute Maximum Ratings), so an
external resistor pull-up can be used to augment the driving
capability. If using a pull-up resistor in conjunction with the
internal overvoltage protection function, care must be taken
to avoid nuisance trips that could occur when VCC is below
2V. In that case, the controller is incapable of holding OVP
low.
UV
OC
+
I
1
REPEAT FOR
EACH CHANNEL
75%
110µA
-
SOFT-START, FAULT
DAC
OC
REFERENCE
AND CONTROL LOGIC
+
I
AVG
VDIFF
+
OVP
OV
-
VID + 0.2V
Once an overvoltage condition is detected, normal PWM
operation ceases until the ISL6564A is reset. Cycling the
voltage on EN or ENLL or VCC below the POR-falling
threshold will reset the controller. Cycling the VID codes will
not reset the controller.
FIGURE 12. POWER GOOD AND PROTECTION CIRCUITRY
Undervoltage Detection
The undervoltage threshold is set at 75% of the VID code.
When the output voltage at VSEN is below the undervoltage
threshold, PGOOD gets pulled low.
Overcurrent Protection
ISL6564A has two levels of overcurrent protection. Each
phase is protected from a sustained overcurrent condition on
a delayed basis, while the combined phase currents are
protected on an instantaneous basis.
Overvoltage Protection
When VCC is above 1.4V, but otherwise not valid as defined
under Power-on Reset in Electrical Specifications, the
overvoltage trip circuit is active using auxiliary circuitry. In
this state, an overvoltage trip occurs if the voltage at VSEN
exceeds 1.8V.
In instantaneous protection mode, the ISL6564A takes
advantage of the proportionality between the load current
and the average current, I
, to detect an overcurrent
AVG
condition. See the Channel-Current Balance section for
more detail on how the average current is measured. The
average current is continually compared with a constant
110μA reference current as shown in Figure 12. Once the
average current exceeds the reference current, a
comparator triggers the converter to shutdown.
With valid VCC, the overvoltage circuit is sensitive to the
voltage at VDIFF. In this state, the trip level is 1.7V prior to
valid enable conditions being met as described in Enable
and Disable. The only exception to this is when the IC has
been disabled by an overvoltage trip. In that case the
overvoltage trip point is VID plus 200mV. During soft-start,
the overvoltage trip level is the higher of 1.5V or VID plus
200mV. Upon successful soft-start, the overvoltage trip level
is 200mV above VID. Two actions are taken by the
ISL6564A to protect the microprocessor load when an
overvoltage condition occurs.
In individual overcurrent protection mode, the ISL6564A
continuously compares the current of each channel with the
same 110μA reference current. If any channel current
exceeds the reference current continuously for eight
consecutive cycles, the comparator triggers the converter to
shutdown.
At the inception of an overvoltage event, all PWM outputs
are commanded low instantly (less than 20ns) until the
voltage at VSEN falls below 0.6V with valid VCC or 1.5V
otherwise. This causes the Intersil drivers to turn on the
lower MOSFETs and pull the output voltage below a level
that might cause damage to the load. The PWM outputs
remain low until VDIFF falls to the programmed DAC level
when they enter a high-impedance state. The Intersil drivers
respond to the high-impedance input by turning off both
upper and lower MOSFETs. If the overvoltage condition
FN6285.1
March 20, 2007
19
ISL6564A
In normal operation, DRVEN remains low until ISL6564A
begins soft-start ramp and then changes to high (Figure 14).
When an overcurrent event occurs, DRVEN is pulled to low
instantly (less than 20ns) to disable the driver so that both
upper and lower FETs be turned off (Figure 15). During an
overvoltage condition, DRVEN remains high to allow the
driver turn on the lower FETs based on the PWM input to
discharge the energy stored in the output inductor. Once the
Output voltage is reduced to 0.6V, DRVEN is pulled to low as
shown in Figure 16.
OUTPUT CURRENT, 50A/DIV
0A
OUTPUT VOLTAGE,
500mV/DIV
DRVEN, 5V/DIV
0V
0V
2ms/DIV
OUTPUT CURRENT, 50A/DIV
FIGURE 13. OVERCURRENT BEHAVIOR IN HICCUP MODE,
= 500kHz
F
SW
At the beginning of overcurrent shutdown, the controller
places all PWM signals in a high-impedance state within
20ns commanding the Intersil MOSFET driver ICs to turn off
both upper and lower MOSFETs. The system remains in this
state a period of 4096 switching cycles. If the controller is still
enabled at the end of this wait period, it will attempt a
soft-start. If the fault remains, the trip-retry cycles will
continue indefinitely (as shown in Figure 13) until either
controller is disabled or the fault is cleared. Note that the
energy delivered during trip-retry cycling is much less than
during full-load operation, so there, there is no thermal
hazard during this kind of operation.
0A
OUTPUT VOLTAGE,
500mV/DIV
0V
2ms/DIV
FIGURE 15. DRVEN DURING OVERCURRENT OPERATION
DRVEN, 5V/DIV
DRVEN, 5V/DIV
OUTPUT VOLTAGE,
500mV/DIV
VOUT, 1V/DIV
EN, 5V/DIV
2ms/DIV
FIGURE 16. DRVEN DURING OVERCURRENT OPERATION
500µs/DIV
There’s no need to use DRVEN when ISL6564A is used to
work with Intersil’s drivers such as ISL6612 and ISL6605.
FIGURE 14. DRVEN WAVEFORM AT START-UP
Current Sense Output
Driver Enable Output
The ISL6564A has 2 current sense output pins IDROOP and
IOUT. They are identical. In typical application, IDROOP pin
is connected to FB pin for the application where load line is
required. IOUT pin was designed for load current
The ISL6564A has a driver enable output pin DRVEN. The
DRVEN is designed for the application where ISL6564A
needs to work with drivers that can not recognize three-state
PWM input.
measurement. As shown in typical application schematics
FN6285.1
March 20, 2007
20
ISL6564A
on pages 4 to 7, load current information can be obtained by
current range. If through-hole MOSFETs and inductors can
be used, higher per-phase currents are possible. In cases
where board space is the limiting constraint, current can be
pushed as high as 40A per phase, but these designs require
heat sinks and forced air to cool the MOSFETs, inductors
and heat-dissipating surfaces.
measuring the voltage between IOUT to ground when a NTC
network from IOUT pin to the ground is placed. The output
current at IOUT pin is proportional to load current as shown
in Figure 17.
MOSFETs
The choice of MOSFETs depends on the current each
MOSFET will be required to conduct; the switching
frequency; the capability of the MOSFETs to dissipate heat;
and the availability and nature of heat sinking and air flow.
V_IOUT, 200mV/DIV
LOWER MOSFET POWER CALCULATION
The calculation for heat dissipated in the lower MOSFET is
simple, since virtually all of the heat loss in the lower
MOSFET is due to current conducted through the channel
resistance (r
). In Equation 14, I is the maximum
DS(ON)
M
continuous output current; I is the peak-to-peak inductor
PP
current (see Equation 1); d is the duty cycle (V
/V ); and
OUT IN
L is the per-channel inductance.
0A
50A
100A
2
2
I
(1 – d)
⎛
⎜
⎝
⎞
⎟
⎠
I
L, PP
(EQ. 14)
M
FIGURE 17. VOLTAGE AT IOUT PIN WITH A NTC NETWORK
PLACED BETWEEN IOUT TO GROUND WHEN
LOAD CURRENT CHANGES
P
= r
(1 – d) + --------------------------------
-----
LOW, 1
DS(ON)
12
N
An additional term can be added to the lower-MOSFET loss
equation to account for additional loss accrued during the
dead time when inductor current is flowing through the
lower-MOSFET body diode. This term is dependent on the
When selecting the equivalent resistor network components
values, it is important to ensure the voltage at IOUT pin not
exceed 2V.
diode forward voltage at I , V
; the switching
M
D(ON)
frequency, f ; and the length of dead times, t and t , at
the beginning and the end of the lower-MOSFET conduction
interval respectively.
When ISL6564A is operated at single phase mode (both
PWM3 and PWM4 connected to VCC and PWM2
unconnected). The output current at IOUT and IDROOP is
half of the sensed phase current.
S
d1 d2
⎛
⎜
⎝
⎞
⎟
⎠
I
I
M
N
I
I
(EQ. 15)
⎛
⎝
⎞
⎠
M
PP
2
PP
2
P
= V
f
D(ON) S
t
t
d2
+
--------
----- –
----- + --------
LOW, 2
d1
N
General Design Guide
This design guide is intended to provide a high-level
explanation of the steps necessary to create a multiphase
power converter. It is assumed that the reader is familiar with
many of the basic skills and techniques referenced below. In
addition to this guide, Intersil provides complete reference
designs that include schematics, bills of materials, and
example board layouts for all common microprocessor
applications.
Thus the total maximum power dissipated in each lower
MOSFET is approximated by the summation of P
and
LOW,1
P
.
LOW,2
UPPER MOSFET POWER CALCULATION
In addition to r losses, a large portion of the upper-
DS(ON)
MOSFET losses are due to currents conducted across the
input voltage (V ) during switching. Since a substantially
IN
higher portion of the upper-MOSFET losses are dependent
on switching frequency, the power calculation is more
complex. Upper MOSFET losses can be divided into
separate components involving the upper-MOSFET
switching times; the lower-MOSFET body-diode reverse-
Power Stages
The first step in designing a multiphase converter is to
determine the number of phases. This determination
depends heavily on the cost analysis which in turn depends
on system constraints that differ from one design to the next.
Principally, the designer will be concerned with whether
components can be mounted on both sides of the circuit
board; whether through-hole components are permitted; and
the total board space available for power-supply circuitry.
Generally speaking, the most economical solutions are
those in which each phase handles between 15 and 20A. All
surface-mount designs will tend toward the lower end of this
recovery charge, Q ; and the upper MOSFET r
rr
DS(ON)
conduction loss.
When the upper MOSFET turns off, the lower MOSFET does
not conduct any portion of the inductor current until the
voltage at the phase node falls below ground. Once the
lower MOSFET begins conducting, the current in the upper
MOSFET falls to zero as the current in the lower MOSFET
FN6285.1
March 20, 2007
21
ISL6564A
ramps up to assume the full inductor current. In Equation 16,
the required time for this commutation is t and the
decrease in temperature rise in order to cause proportionally
less current to flow in the hotter phase.
1
approximated associated power loss is P
.
ΔT
UP,1
2
(EQ. 21)
R
= R
----------
ISEN,2
ISEN
ΔT
t
I
I
⎛
⎜
⎝
⎞
⎟
⎠
1
1
M
PP
⎛
⎝
⎞
(EQ. 16)
P
≈ V
f
S
----
----- + --------
UP,1
IN
⎠
2
N
2
In Equation 21, make sure that ΔT is the desired temperature
2
rise above the ambient temperature, and ΔT is the measured
temperature rise above the ambient temperature. While a
single adjustment according to Equation 21 is usually
1
At turn on, the upper MOSFET begins to conduct and this
transition occurs over a time t . In Equation 17, the
2
approximate power loss is P
.
UP,2
sufficient, it may occasionally be necessary to adjust R
two or more times to achieve optimal thermal balance
between all channels.
ISEN
I
t
⎞
2
2
⎠
⎛I
⎞ ⎛
PP
2
M
(EQ. 17)
P
≈ V
f
--------⎟ ⎜ ---- ⎟
S
⎜
⎝
----- –
UP,2
IN
N
⎠ ⎝
Load-Line Regulation Resistor
A third component involves the lower MOSFET’s reverse-
The load-line regulation resistor is labeled R in Figure 8.
FB
Its value depends on the desired full-load droop voltage
recovery charge, Q . Since the inductor current has fully
rr
commutated to the upper MOSFET before the lower-
(V
in Figure 8). If Equation 20 is used to select each
DROOP
MOSFET’s body diode can draw all of Q , it is conducted
rr
ISEN resistor, the load-line regulation resistor is as shown in
Equation 22.
through the upper MOSFET across VIN. The power
dissipated as a result is P
and is approximately
UP,3
V
DROOP
R
= ------------------------
–6
P
= V
Q f
IN rr S
(EQ. 22)
(EQ. 18)
FB
UP,3
70 ×10
Finally, the resistive part of the upper MOSFET’s is given in
Equation 19 as P
If one or more of the ISEN resistors is adjusted for thermal
balance, as in Equation 21, the load-line regulation resistor
.
UP,4
should be selected according to Equation 23 where I is the
The total power dissipated by the upper MOSFET at full load
can now be approximated as the summation of the results
from Equations 16, 17, 18 and 19. Since the power
equations depend on MOSFET parameters, choosing the
correct MOSFETs can be an iterative process involving
repetitive solutions to the loss equations for different
MOSFETs and different switching frequencies.
FL
full-load operating current and R
connected to the n ISEN pin.
is the ISEN resistor
(EQ. 23)
ISEN(n)
th
V
DROOP
r
R
= --------------------------------
R
ISEN(n)
∑
FB
I
FL DS(ON)
n
Compensation
2
2
⎛
⎜
⎝
⎞
⎟
⎠
I
PP
I
The two opposing goals of compensating the voltage
regulator are stability and speed. Depending on whether the
regulator employs the optional load-line regulation as
described in Load-Line Regulation, there are two distinct
methods for achieving these goals.
M
(EQ. 19)
P
≈ r
d +
---------
12
-----
UP,4
DS(ON)
N
Current Sensing Resistor
The resistors connected between these pins and the
respective phase nodes determine the gains in the load-line
regulation loop and the channel-current balance loop as well
as setting the overcurrent trip point. Select values for these
COMPENSATING LOAD-LINE REGULATED
CONVERTER
The load-line regulated converter behaves in a similar
manner to a peak-current mode controller because the two
poles at the output-filter L-C resonant frequency split with
the introduction of current information into the control loop.
The final location of these poles is determined by the system
function, the gain of the current signal, and the value of the
resistors based on the room temperature r
of the
DS(ON)
lower MOSFETs, DCR of inductor or additional resistor; the
full-load operating current, I ; and the number of phases, N
FL
using Equation 20.
R
I
FL
N
X
(EQ. 20)
R
= ----------------------- -------
–
ISEN
6
70 ×10
compensation components, R and C .
C
C
Since the system poles and zero are affected by the values
of the components that are meant to compensate them, the
solution to the system equation becomes fairly complicated.
Fortunately there is a simple approximation that comes very
close to an optimal solution. Treating the system as though it
were a voltage-mode regulator by compensating the L-C
poles and the ESR zero of the voltage-mode approximation
yields a solution that is always stable with very close to ideal
transient performance.
In certain circumstances, it may be necessary to adjust the
value of one or more ISEN resistor. When the components of
one or more channels are inhibited from effectively
dissipating their heat so that the affected channels run hotter
than desired, choose new, smaller values of RISEN for the
affected phases (see the section entitled Channel-Current
Balance). Choose R
ISEN,2
in proportion to the desired
FN6285.1
March 20, 2007
22
ISL6564A
C
C
(OPTIONAL)
2
2
C
C
C
C
R
R
C
C
COMP
FB
COMP
FB
IDROOP
VDIFF
C
1
+
IDROOP
VDIFF
R
R
R
FB
FB
V
1
DROOP
-
FIGURE 18. COMPENSATION CONFIGURATION FOR
LOAD-LINE REGULATED ISL6564A CIRCUIT
FIGURE 19. COMPENSATION CIRCUIT FOR ISL6564A BASED
CONVERTER WITHOUT LOAD-LINE
REGULATION
The feedback resistor, R , has already been chosen as
FB
the bulk output-filter capacitance; and V is the peak-to-
PP
outlined in Load-Line Regulation Resistor. Select a target
peak sawtooth signal amplitude as described in Figure 7 and
bandwidth for the compensated system, f . The target
Electrical Specifications.
0
bandwidth must be large enough to assure adequate
transient performance, but smaller than 1/3 of the
per-channel switching frequency. The values of the
compensation components depend on the relationships of f
to the L-C pole frequency and the ESR zero frequency. For
each of the three cases which follow, there is a separate set
of equations for the compensation components.
The optional capacitor C , is sometimes needed to bypass
2
noise away from the PWM comparator (see Figure 18). Keep
a position available for C , and be prepared to install a high-
2
0
frequency capacitor of between 22pF and 150pF in case any
leading-edge jitter problem is noted.
Once selected, the compensation values in Equation 24
assure a stable converter with reasonable transient
performance. In most cases, transient performance can be
1
------------------- > f
Case 1:
0
2π LC
improved by making adjustments to R . Slowly increase the
C
2πf V
LC
pp
0
R
C
= R -----------------------------------
value of R while observing the transient performance on an
C
C
C
FB
0.75V
IN
oscilloscope until no further improvement is noted. Normally,
0.75V
IN
C
will not need adjustment. Keep the value of C from
C
C
= ------------------------------------
2πV
R
f
PP FB 0
Equation 24 unless some performance issue is noted.
COMPENSATION WITHOUT LOAD-LINE REGULATION
1
1
-------------------
≤ f < -----------------------------
0
2πC(ESR)
Case 2:
2π LC
The non load-line regulated converter is accurately modeled
as a voltage-mode regulator with two poles at the L-C
resonant frequency and a zero at the ESR frequency. A
type III controller, as shown in Figure 19, provides the
necessary compensation.
2
2
V
(2π)
f
LC
0
PP
R
C
= R --------------------------------------------
(EQ. 24)
C
C
FB
0.75 V
IN
0.75V
IN
= ------------------------------------------------------------
2
2
(2π)
f
V
R
LC
0
PP FB
The first step is to choose the desired bandwidth, f , of the
0
compensated system. Choose a frequency high enough to
assure adequate transient performance but not higher than
1/3 of the switching frequency. The type-III compensator has
1
Case 3:
f > -----------------------------
0
2πC(ESR)
an extra high-frequency pole, f . This pole can be used for
HF
2π f V
L
pp
0
R
C
= R
-----------------------------------------
FB
added noise rejection or to assure adequate attenuation at
the error-amplifier high-order pole and zero frequencies. A
good general rule is to choose f = 10f , but it can be
C
C
0.75 V (ESR)
IN
0.75V (ESR)
C
IN
HF
0
= -------------------------------------------------
2πV
R
f
L
higher if desired. Choosing f to be lower than 10f can
HF
0
PP FB 0
cause problems with too much phase shift below the system
bandwidth.
In Equation 24, L is the per-channel filter inductance divided
by the number of active channels; C is the sum total of all
output capacitors; ESR is the equivalent-series resistance of
In the solutions to the compensation equations, there is a
single degree of freedom. For the solutions presented in
Equation 24, R is selected arbitrarily. The remaining
FB
FN6285.1
March 20, 2007
23
ISL6564A
compensation components are then selected according to
Equation 25.
voltage deviation is less than the allowable maximum.
Neglecting the contribution of inductor current and regulator
response, the output voltage initially deviates by an amount:
C(ESR)
LC – C(ESR)
R
= R ----------------------------------------
FB
1
1
di
dt
(EQ. 26)
ΔV ≈ (ESL) ---- + (ESR) ΔI
LC – C(ESR)
C
= ----------------------------------------
R
FB
The filter capacitor must have sufficiently low ESL and ESR
so that ΔV < ΔV
.
MAX
0.75V
IN
C
R
= ------------------------------------------------------------------
2
2
(2π) f f
LCR
V
Most capacitor solutions rely on a mixture of high-frequency
capacitors with relatively low capacitance in combination
with bulk capacitors having high capacitance but limited
high-frequency performance. Minimizing the ESL of the
high-frequency capacitors allows them to support the output
voltage as the current increases. Minimizing the ESR of the
bulk capacitors allows them to supply the increased current
with less output voltage deviation.
0 HF
FB PP
2
⎛
⎝
⎞
f f LCR
V
2π
0 HF
FB
PP
⎠
= --------------------------------------------------------------------
C
⎛
⎞
LC–1
2πf
0.75 V
⎝
HF
⎠
IN
⎛
⎞
LC–1
0.75V
2πf
IN
⎝
HF
⎠
C
= ------------------------------------------------------------------
(EQ. 25)
C
2
The ESR of the bulk capacitors also creates the majority of
the output-voltage ripple. As the bulk capacitors sink and
source the inductor ac ripple current (see Interleaving and
Equation 2), a voltage develops across the bulk-capacitor
(2π) f f
LCR
V
FB PP
0 HF
In Equations 25, L is the per-channel filter inductance
divided by the number of active channels; C is the sum total
of all output capacitors; ESR is the equivalent-series
ESR equal to I
(ESR). Thus, once the output capacitors
C,PP
are selected, the maximum allowable ripple voltage,
resistance of the bulk output-filter capacitance; and V is
the peak-to-peak sawtooth signal amplitude as described in
Figure 7 and Electrical Specifications.
PP
V
, determines the lower limit on the inductance.
PP(MAX)
⎛
⎝
⎞
V
V
– N V
IN
OUT
OUT
⎠
(EQ. 27)
L
(ESR)
≥
-----------------------------------------------------------
f V
V
IN PP(MAX)
S
Output Filter Design
The output inductors and the output capacitor bank together
to form a low-pass filter responsible for smoothing the
pulsating voltage at the phase nodes. The output filter also
must provide the transient energy until the regulator can
respond. Because it has a low bandwidth compared to the
switching frequency, the output filter necessarily limits the
system transient response. The output capacitor must
supply or sink load current while the current in the output
inductors increases or decreases to meet the demand.
Since the capacitors are supplying a decreasing portion of
the load current while the regulator recovers from the
transient, the capacitor voltage becomes slightly depleted.
The output inductors must be capable of assuming the entire
load current before the output voltage decreases more than
ΔV
. This places an upper limit on inductance.
MAX
Equation 28 gives the upper limit on L for the cases when
the trailing edge of the current transient causes a greater
output-voltage deviation than the leading edge. Equation 29
addresses the leading edge. Normally, the trailing edge
dictates the selection of L because duty cycles are usually
less than 50%. Nevertheless, both inequalities should be
evaluated, and L should be selected based on the lower of
the two results. In each equation, L is the per-channel
inductance, C is the total output capacitance, and N is the
number of active channels.
In high-speed converters, the output capacitor bank is
usually the most costly (and often the largest) part of the
circuit. Output filter design begins with minimizing the cost of
this part of the circuit. The critical load parameters in
choosing the output capacitors are the maximum size of the
load step, ΔI; the load-current slew rate, di/dt; and the
maximum allowable output-voltage deviation under transient
loading, ΔV
. Capacitors are characterized according to
MAX
their capacitance, ESR, and ESL (equivalent series
inductance).
2NCV
O
(EQ. 28)
(EQ. 29)
L ≤ --------------------- ΔV
– ΔI(ESR)
MAX
2
(
)
ΔI
At the beginning of the load transient, the output capacitors
supply all of the transient current. The output voltage will
initially deviate by an amount approximated by the voltage
drop across the ESL. As the load current increases, the
voltage drop across the ESR increases linearly until the load
current reaches its final value. The capacitors selected must
have sufficiently low ESL and ESR so that the total output-
(
)
)
1.25 NC
⎛
⎝
⎞
– V
O
L ≤ ------------------------- ΔV
– ΔI(ESR)
V
MAX
IN
2
⎠
(
ΔI
FN6285.1
March 20, 2007
24
ISL6564A
Input Supply Voltage Selection
0.3
0.2
0.1
0
The VCC input of the ISL6564A can be connected either
directly to a +5V supply or through a current limiting resistor
to a +12V supply. An integrated 5.8V shunt regulator
maintains the voltage on the VCC pin when a +12V supply is
used. A 300Ω resistor is suggested for limiting the current
into the VCC pin to a worst-case maximum of approximately
25mA.
Switching Frequency
There are a number of variables to consider when choosing
the switching frequency, as there are considerable effects on
the upper-MOSFET loss calculation. These effects are
outlined in MOSFETs, and they establish the upper limit for
the switching frequency. The lower limit is established by the
requirement for fast transient response and small output-
voltage ripple as outlined in Output Filter Design. Choose the
lowest switching frequency that allows the regulator to meet
the transient-response requirements.
I
I
I
= 0
L,PP
L,PP
L,PP
= 0.5 I
O
= 0.75 I
O
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (V /V
)
IN
O
FIGURE 21. NORMALIZED INPUT-CAPACITOR RMS CURRENT
vs DUTY CYCLE FOR 2-PHASE CONVERTER
0.3
Switching frequency is determined by the selection of the
I
I
= 0
I
I
= 0.5 I
O
L,PP
L,PP
L,PP
frequency-setting resistor, R (see the figures labeled
T
= 0.25 I
= 0.75 I
O
L,PP
O
Typical Application on pages 4, 5, 6 and 7). Figure 20 and
Equation 30 are provided to assist in selecting the correct
value for R .
T
0.2
0.1
0
1000
100
10
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (V
V
)
O/ IN
FIGURE 22. NORMALIZED INPUT-CAPACITOR RMS CURRENT
vs DUTY CYCLE FOR 3-PHASE CONVERTER
For a two phase design, use Figure 21 to determine the
input-capacitor RMS current requirement given the duty
10
100
1000
10000
SWITCHING FREQUENCY (kHz)
cycle, maximum sustained output current (I ), and the ratio
O
of the per-phase peak-to-peak inductor current (I
) to I .
L,PP
O
FIGURE 20. R vs SWITCHING FREQUENCY
T
Select a bulk capacitor with a ripple current rating which will
minimize the total number of input capacitors required to
support the RMS current calculated. The voltage rating of
the capacitors should also be at least 1.25 times greater
than the maximum input voltage.
[
]
10.886 – 1.0792log(f
)
(EQ. 30)
S
R
= 10
T
Input Capacitor Selection
Figures 22 and 23 provide the same input RMS current
information for three and four phase designs respectively.
Use the same approach to selecting the bulk capacitor type
and number as described above.
The input capacitors are responsible for sourcing the ac
component of the input current flowing into the upper
MOSFETs. Their RMS current capacity must be sufficient to
handle the ac component of the current drawn by the upper
MOSFETs which is related to duty cycle and the number of
active phases.
Low capacitance, high-frequency ceramic capacitors are
needed in addition to the bulk capacitors to suppress leading
and falling edge voltage spikes. The result from the high
current slew rates produced by the upper MOSFETs turn on
FN6285.1
March 20, 2007
25
ISL6564A
and off. Select low ESL ceramic capacitors and place one as
close as possible to each upper MOSFET drain to minimize
board parasitic impedances and maximize suppression.
Layout Considerations
The following layout strategies are intended to minimize the
impact of board parasitic impedances on converter
performance and to optimize the heat-dissipating capabilities
of the printed-circuit board. These sections highlight some
important practices which should not be overlooked during the
layout process.
0.3
I
I
= 0
= 0.25 I
I
I
= 0.5 I
O
L,PP
L,PP
L,PP
L,PP
= 0.75 I
O
O
Component Placement
0.2
0.1
0
Within the allotted implementation area, orient the switching
components first. The switching components are the most
critical because they carry large amounts of energy and tend
to generate high levels of noise. Switching component
placement should take into account power dissipation. Align
the output inductors and MOSFETs such that space between
the components is minimized while creating the PHASE
plane. Place the Intersil MOSFET driver IC as close as
possible to the MOSFETs they control to reduce the parasitic
impedances due to trace length between critical driver input
and output signals. If possible, duplicate the same
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (V
V
)
O/ IN
FIGURE 23. NORMALIZED INPUT-CAPACITOR RMS CURRENT
vs DUTY CYCLE FOR 4-PHASE CONVERTER
placement of these components for each phase.
Next, place the input and output capacitors. Position one
high-frequency ceramic input capacitor next to each upper
MOSFET drain. Place the bulk input capacitors as close to
the upper MOSFET drains as dictated by the component
size and dimensions. Long distances between input
capacitors and MOSFET drains result in too much trace
inductance and a reduction in capacitor performance. Locate
the output capacitors between the inductors and the load,
while keeping them in close proximity to the microprocessor
socket.
MULTIPHASE RMS IMPROVEMENT
Figure 24 is provided as a reference to demonstrate the
dramatic reductions in input-capacitor RMS current upon the
implementation of the multiphase topology. For example,
compare the input RMS current requirements of a two-phase
converter versus that of a single phase. Assume both
converters have a duty cycle of 0.25, maximum sustained
output current of 40A, and a ratio of I
to I of 0.5. The
L,PP
O
single phase converter would require 17.3 Arms current
capacity while the two-phase converter would only require
10.9 Arms. The advantages become even more pronounced
when output current is increased and additional phases are
added to keep the component cost down relative to the
single phase approach.
The ISL6564A can be placed off to one side or centered
relative to the individual phase switching components.
Routing of sense lines and PWM signals will guide final
placement. Critical small signal components to place close
to the controller include the ISEN resistors, R resistor,
T
feedback resistor, and compensation components.
0.6
0.4
0.2
Bypass capacitors for the ISL6564A and ISL66XX driver
bias supplies must be placed next to their respective pins.
Trace parasitic impedances will reduce their effectiveness.
Plane Allocation and Routing
Dedicate one solid layer, usually a middle layer, for a ground
plane. Make all critical component ground connections with
vias to this plane. Dedicate one additional layer for power
planes; breaking the plane up into smaller islands of
I
I
I
= 0
= 0.5 I
= 0.75 I
L,PP
L,PP
L,PP
common voltage. Use the remaining layers for signal wiring.
O
O
Route phase planes of copper filled polygons on the top and
bottom once the switching component placement is set. Size
the trace width between the driver gate pins and the
MOSFET gates to carry 4A of current. When routing
components in the switching path, use short wide traces to
reduce the associated parasitic impedances.
0
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (V
V
)
O/ IN
FIGURE 24. NORMALIZED INPUT-CAPACITOR RMS
CURRENT vs DUTY CYCLE FOR SINGLE-PHASE
CONVERTER
FN6285.1
March 20, 2007
26
ISL6564A
Package Outline Drawing
L40.6x6
40 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 3, 10/06
4X
4.5
6.00
0.50
36X
A
6
B
31
40
PIN #1 INDEX AREA
6
30
1
PIN 1
INDEX AREA
4 . 10 ± 0 . 15
21
10
(4X)
0.15
11
20
0.10 M C A B
TOP VIEW
40X 0 . 4 ± 0 . 1
4
0 . 23 +0 . 07 / -0 . 05
BOTTOM VIEW
SEE DETAIL "X"
C
0.10
C
0 . 90 ± 0 . 1
BASE PLANE
( 5 . 8 TYP )
(
SEATING PLANE
0.08 C
SIDE VIEW
4 . 10 )
( 36X 0 . 5 )
5
C
0 . 2 REF
( 40X 0 . 23 )
0 . 00 MIN.
0 . 05 MAX.
( 40X 0 . 6 )
DETAIL "X"
TYPICAL RECOMMENDED LAND PATTERN
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3.
Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
Tiebar shown (if present) is a non-functional feature.
5.
6.
The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 indentifier may be
either a mold or mark feature.
FN6285.1
March 20, 2007
27
ISL6564A
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
FN6285.1
March 20, 2007
28
相关型号:
ISL6564AIRZ
Multiphase PWM Controller with Linear 6-Bit DAC Capable of Precision rDS(ON) or DCR Differential Current Sensing
INTERSIL
ISL6564AIRZ
SWITCHING CONTROLLER, 1500kHz SWITCHING FREQ-MAX, PQCC40, 6 X 6 MM, ROHS COMPLIANT, PLASTIC, MO-220, QFN-40
RENESAS
ISL6564AIRZ
SWITCHING CONTROLLER, 1500 kHz SWITCHING FREQ-MAX, PQCC40, 6 X 6 MM, ROHS COMPLIANT, PLASTIC, MO-220, QFN-40
ROCHESTER
ISL6564CR
Multi-Phase PWM Controller with Linear 6-bit DAC Capable of Precision rDS(ON) or DCR Differential Current Sensing
INTERSIL
ISL6564CR-T
Multi-Phase PWM Controller with Linear 6-bit DAC Capable of Precision rDS(ON) or DCR Differential Current Sensing
INTERSIL
ISL6564CRZ
Multi-Phase PWM Controller with Linear 6-bit DAC Capable of Precision rDS(ON) or DCR Differential Current Sensing
INTERSIL
ISL6564CRZ
SWITCHING CONTROLLER, 1500kHz SWITCHING FREQ-MAX, PQCC40, 6 X 6 MM, LEAD FREE, PLASTIC, MO-220-VJJD, QFN-40
RENESAS
ISL6564CRZ-T
Multi-Phase PWM Controller with Linear 6-bit DAC Capable of Precision rDS(ON) or DCR Differential Current Sensing
INTERSIL
ISL6564CRZ-T
SWITCHING CONTROLLER, 1500kHz SWITCHING FREQ-MAX, PQCC40, 6 X 6 MM, LEAD FREE, PLASTIC, MO-220-VJJD, QFN-40
RENESAS
ISL6564IR
Multi-Phase PWM Controller with Linear 6-bit DAC Capable of Precision rDS(ON) or DCR Differential Current Sensing
INTERSIL
ISL6564IR-T
Multi-Phase PWM Controller with Linear 6-bit DAC Capable of Precision rDS(ON) or DCR Differential Current Sensing
INTERSIL
ISL6564IRZ
Multi-Phase PWM Controller with Linear 6-bit DAC Capable of Precision rDS(ON) or DCR Differential Current Sensing
INTERSIL
©2020 ICPDF网 联系我们和版权申明