LED2001PUR_技术文档

LED2001

4 A monolithic step-down current source with synchronous

rectification

Datasheet

-

production data

Applications

• High brightness LED driving

• Halogen bulb replacement

• General lighting

• Signage

HSOP8

VFQFPN8 4x4

Description

Features

The LED2001 is an 850 kHz fixed switching

frequency monolithic step-down DC-DC converter

designed to operate as precise constant current

source with an adjustable current capability up to

4 A DC. The embedded PWM dimming circuitry

features LED brightness control. The regulated

output current is set connecting a sensing resistor

to the feedback pin. The embedded synchronous

rectification and the 100 mV typical R_SENSE

voltage drop enhance the efficiency performance.

The size of the overall application is minimized

thanks to the high switching frequency and

ceramic output capacitor compatibility. The device

is fully protected against thermal overheating,

overcurrent and output short-circuit. The

• 3.0 V to 18 V operating input voltage range

• 850 kHz fixed switching frequency

• 100 mV typ. current sense voltage drop

• PWM dimming

•

7% output current accuracy

• Synchronous rectification

• 95 m HS / 69 m LS typical R_DS(on)

Ω

• Peak current mode architecture

• Embedded compensation network

• Internal current limiting

• Ceramic output capacitor compliant

• Thermal shutdown

LED2001 is available in VFQFPN 4 mm x 4 mm

8-lead package, and HSOP8.

Figure 1. Typical application circuit

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May 2013

DocID024346 Rev 1

1/42

This is information on a product in full production.

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42

Contents

LED2001

Contents

1

Pin settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

1.1

1.2

Pin connection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

2

3

4

5

Maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

Functional description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9

5.1

5.2

5.3

5.4

5.5

Power supply and voltage reference . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

Voltage monitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

Soft-start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

Error amplifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

Thermal shutdown . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .11

6

Application notes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

6.1

6.2

6.3

6.4

6.5

6.6

Closing the loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

(s) control to output transfer function . . . . . . . . . . . . . . . . . . . . . . . . 12

G

CO

Error amplifier compensation network . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

LED small signal model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15

Total loop gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17

Dimming operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

6.6.1

Dimming frequency vs. dimming depth . . . . . . . . . . . . . . . . . . . . . . . . . 20

6.7

eDesign studio software . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

7

Application information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

7.1

Component selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

7.1.1

7.1.2

7.1.3

Sensing resistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

Inductor and output capacitor selection . . . . . . . . . . . . . . . . . . . . . . . . . 22

Input capacitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

7.2

Layout considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26

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Contents

7.3

7.4

7.5

Thermal considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

Short-circuit protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

Application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

8

Typical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

Ordering information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36

Package mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41

9

10

11

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List of tables

LED2001

List of tables

Table 1.

Table 2.

Table 3.

Table 4.

Table 5.

Table 6.

Table 7.

Table 8.

Table 9.

Table 10.

Table 11.

Table 12.

Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

Absolute maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

Thermal data. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

Uncompensated error amplifier characteristics. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

Inductor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

List of ceramic capacitors for the LED2001 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

Component list . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31

Ordering information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36

VFQFPN8 (4x4x1.08 mm) mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

HSOP8 mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39

Document revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41

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List of figures

Figure 1.

Figure 2.

Figure 3.

Figure 4.

Figure 5.

Figure 6.

Figure 7.

Figure 8.

Figure 9.

Typical application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

Pin connection (top view) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

LED2001 block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9

Internal circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

Block diagram of the loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

Transconductance embedded error amplifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14

Equivalent series resistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

Load equivalent circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

Module plot . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

Figure 10. Phase plot. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

Figure 11. Dimming operation example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19

Figure 12. LED current falling edge operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

Figure 13. Dimming signal . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

Figure 14. eDesign studio screenshot . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

Figure 15. Equivalent circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

Figure 16. Layout example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26

Figure 17. Switching losses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

Figure 18. Constant current protection triggering Hiccup mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

Figure 19. Demonstration board application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

Figure 20. PCB layout (component side) DFN package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31

Figure 21. PCB layout (bottom side) DFN package. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32

Figure 22. PCB layout (component side) HSOP8 package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32

Figure 23. PCB layout (bottom side) HSOP8 package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33

Figure 24. Soft-start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

Figure 25. Load regulation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

Figure 26. Dimming operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

Figure 27. LED current rising edge . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

Figure 28. LED current falling edge . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

Figure 29. Hiccup current protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

Figure 30. OCP blanking time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35

Figure 31. Thermal shutdown protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35

Figure 32. VFQFPN8 (4x4x1.08 mm) package dimensions. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

Figure 33. HSOP8 package dimensions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40

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Pin settings

LED2001

1

Pin settings

1.1

Pin connection

Figure 2. Pin connection (top view)

1

8

VINA

PGND

9

2

3

6

DIM

FB

SW

DIM

VINSW

7

5

NC

4

NC

GND

VFQFPN8 4x4

HSOP8

AM12893v1

1.2

Pin description

Table 1. Pin description

Package/pin

Type

Description

VFQFPN8

HS0P8

4x4

1

VIN_A

DIM

Analog circuitry power supply connection

Dimming control input. Logic low prevents the switching

activity, logic high enables it. A square wave on this pin

implements LED current PWM dimming. Connect to VIN_Aif

not used (see Section 6.6)

2

Feedback input. Connect a proper sensing resistor to set

the LED current

3

FB

4

5

4

-

AGND

NC

Analog circuitry ground connection

Not connected

6

VIN_SW

SW

Power input voltage

Regulator switching pin

Power ground

7

8

PGND

ep

Exposed pad Connect the exposed pad to AGND

6/42

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LED2001

Maximum ratings

2

Maximum ratings

Table 2. Absolute maximum ratings

Parameter

Symbol

Value

Unit

V_INSW

V_INA

V_DIM

V_SW

V_PG

V_FB

Power input voltage

-0.3 to 20

-0.3 to V_INA

-1 to V_IN

-0.3 to V_IN

-0.3 to 2.5

-1 to +1

Input voltage

Dimming voltage

V

Output switching voltage

Power Good

Feedback voltage

I_FB

FB current

mA

W

P_TOT

T_OP

Power dissipation at T_A< 60 °C

Operating junction temperature range

Storage temperature range

2

-40 to 150

-55 to 150

°C

T_stg

3

Thermal data

Table 3. Thermal data

Parameter

Symbol

Value

Unit

°C/W

VFQFPN8 4x4

Maximum thermal resistance

junction-ambient ⁽¹⁾

R_thJA

40

HSOP8

1. Package mounted on demonstration board.

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Electrical characteristics

LED2001

4

Electrical characteristics

T_J=25 °C, V_CC=12 V, unless otherwise specified.

Table 4. Electrical characteristics

Value

Typ.

Symbol

Parameter

Test conditions

Unit

Min.

Max.

Operating input voltage

range

(1)

3

18

V_IN

V

Device ON level

Device OFF level

2.6

2.4

90

2.75

2.55

97

2.9

2.7

Tj=25 °C

104

110

600

V_FB

I_FB

Feedback voltage

mV

Tj=125 °C ⁽¹⁾

90

100

(1)

V_FBpin bias current

nA

High-side switch on-

resistance

R_DSON-P

I_SW=750 mA

95

mΩ

Low-side switch on-

resistance

R_DSON-N

I_SW=750 mA

69

mΩ

(2)

I_LIM

Maximum limiting current

5.6

A

Oscillator

F_SW

D

Switching frequency

Duty cycle

0.7

0

0.85

1.5

1

MHz

%

(2)

100

DC characteristics

I_Q

Quiescent current

2.5

0.4

mA

Dimming

Switching activity

1.2

V_DIM

DIM threshold voltage

DIM current

V

Switching activity

prevented

I_DIM

2

1

μA

Soft-start

T_SS

Soft-start duration

ms

°C

Protection

Thermal shutdown

Hystereris

150

15

T_SHDN

1. Specifications referred to T from -40 to +125 °C. Specifications in the -40 to +125 °C temperature range

J

are assured by design, characterization and statistical correlation.

2. Guaranteed by design.

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LED2001

Functional description

5

Functional description

The LED2001 is based on a “peak current mode” architecture with fixed frequency control.

As a consequence, the intersection between the error amplifier output and the sensed

inductor current generates the control signal to drive the power switch.

The main internal blocks shown in the block diagram in Figure 3 are:

•

high-side and low-side embedded power element for synchronous rectification;

a fully integrated sawtooth oscillator with a typical frequency of 850 kHz;

a transconductance error amplifier;

an high-side current sense amplifier to track the inductor current;

a pulse width modulator (PWM) comparator and the circuitry necessary to drive the

internal power element;

•

the soft-start circuitry to decrease the inrush current at power-up;

the current limitation circuit based on the pulse-by-pulse current protection with

frequency divider;

•

the dimming circuitry for output current PWM;

the thermal protection function circuitry.

Figure 3. LED2001 block diagram

VI NA

VI N SW

OCP

REF

OSC

I2V

I_ SENSE

RSENSE

COMP

REGULATOR

UVLO

_p

Vdrv

OCP

DRIVER

MOSFET

Vsum

Vc

CONTROL

LOGIC

_n

Vdrv

PWM

SW

OTP

DMD

E/A

DIMMING

DRIVER

SOFT-START

0.1V

FB

DIM

GNDA

GNDP

AM12894v1

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Functional description

LED2001

5.1

Power supply and voltage reference

The internal regulator circuit consists of a startup circuit, an internal voltage pre-regulator,

the BandGap voltage reference and the bias block that provides current to all the blocks.

The starter supplies the startup current to the entire device when the input voltage goes high

and the device is enabled. The pre-regulator block supplies the BandGap cell with a pre-

regulated voltage that has a very low supply voltage noise sensitivity.

5.2

Voltage monitor

An internal block continuously senses the V_CC, V_refand V_bg. If the monitored voltages are

good, the regulator begins operating. There is also a hysteresis on the V_CC(UVLO).

Figure 4. Internal circuit

Vcc

PREREGULATOR

VREG

STARTER

BANDGAP

IC BIAS

AM12895v1

D00IN126

VREF

5.3

Soft-start

The startup phase is implemented ramping the reference of the embedded error amplifier in

1 ms typ. time. It minimizes the inrush current and decreases the stress of the power

components at power-up.

During normal operation a new soft-start cycle takes place in case of:

•

thermal shutdown event;

UVLO event.

The soft-start is disabled when DIM input goes high in order to maximize the dimming

performance.

5.4

Error amplifier

The voltage error amplifier is the core of the loop regulation. It is a transconductance

operational amplifier whose non-inverting input is connected to the internal voltage

reference (100 mV), while the inverting input (FB) is connected to the output current sensing

resistor.

The error amplifier is internally compensated to minimize the size of the final application.

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LED2001

Functional description

Table 5. Uncompensated error amplifier characteristics

Description

Value

Transconductance

250 µS

96 dB

Low frequency gain

C_C

R_C

195 pF

70 KΩ

The error amplifier output is compared with the inductor current sense information to

perform PWM control.

5.5

Thermal shutdown

The shutdown block generates a signal that disables the power stage if the temperature of

the chip goes higher than a fixed internal threshold (150 ± 10 °C typical). The sensing

element of the chip is close to the PDMOS area, ensuring fast and accurate temperature

detection. A 15 °C typical hysteresis prevents the device from turning ON and OFF

continuously during the protection operation.

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Application notes

LED2001

6

Application notes

6.1

Closing the loop

Figure 5. Block diagram of the loop

G_CO(s)

V_IN

PWM control

Currentsense

HS

switch

L

V_OUT

LC filter

LS

switch

C_OUT

-

error

FB

-

amplifier

V_CONTROL

+

PWM

V_REF

+

comparator

R_S

R_C

C_C

compensation

network

α

LED

A_O(s)

6.2

G (s) control to output transfer function

CO

The accurate control to output transfer function for a buck peak current mode converter can

be written as:

Equation 1

s

ω_z









1 + ------

R₀

1

G_CO(s) = ------ ⋅ ---------------------------------------------------------------------------------------- ⋅ --------------------- ⋅ F_H(s)

R_i

R₀⋅ T_SW

s







1 + ------

1 + ---------------------- ⋅ [m_C⋅ (1 – D) – 0.5]



ω_p

L

where R₀represents the load resistance, R_ithe equivalent sensing resistor of the current

sense circuitry, ω_pthe single pole introduced by the LC filter and ω_zthe zero given by the

ESR of the output capacitor.

F_H(s) accounts for the sampling effect performed by the PWM comparator on the output of

the error amplifier that introduces a double pole at one half of the switching frequency.

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Application notes

Equation 2

1

ω_Z= -------------------------------

ESR ⋅ C_OUT

Equation 3

m_C⋅ (1 – D) – 0.5

L ⋅ C_OUT⋅ f_SW

1

ω_P= ------------------------------------- + ---------------------------------------------

R_LOAD⋅ C_OUT

where:

Equation 4

S_e



m_C= 1 + ------





S_n

S_e= V_pp⋅ f_SW

V

_IN– V_OUT

S_n= ----------------------------- ⋅ R_i

L

S_nrepresents the slope of the sensed inductor current, S_ethe slope of the external ramp

(V_PPpeak-to-peak amplitude) that implements the slope compensation to avoid sub-

harmonic oscillations at duty cycle over 50%.

The sampling effect contribution F_H(s) is:

Equation 5

1

F_H(s) = ------------------------------------------

s

s²

1 + ------------------ + ------

ω²_n

ω_n⋅ Q_P

where:

Equation 6

ω_n= π ⋅ f_SW

and

Equation 7

1

Q_P= ----------------------------------------------------------

π ⋅ [m_C⋅ (1 – D) – 0.5]

6.3

Error amplifier compensation network

The LED2001 embeds (see Figure 6) the error amplifier and a pre-defined compensation

network which is effective in stabilizing the system in most application conditions.

DocID024346 Rev 1

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Application notes

LED2001

Figure 6. Transconductance embedded error amplifier

E/A

+

COMP

FB

-

R_C

C_C

C_P

V⁺

R_C

R₀

C₀

dV

C_P

G_mdV

C_C

AM12897v1

R_Cand C_Cintroduce a pole and a zero in the open loop gain. C_Pdoes not significantly affect

system stability but it is useful to reduce the noise at the output of the error amplifier.

The transfer function of the error amplifier and its compensation network is:

Equation 8

A_V0⋅ (1 + s ⋅ R_c⋅ C_c)

A₀(s) = -----------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------

s²⋅ R₀⋅ (C₀+ C_p) ⋅ R_c⋅ C_c+ s ⋅ (R₀⋅ C_c+ R₀⋅ (C₀+ C_p) + R_c⋅ C_c) + 1

where A_vo= G_m· R_o.

The poles of this transfer function are (if C_C>> C₀+C_P):

Equation 9

1

f_{P LF}= ---------------------------------

2 ⋅ π ⋅ R₀⋅ C_c

Equation 10

1

f_{P HF}= ----------------------------------------------------

2 ⋅ π ⋅ R_c⋅ (C₀+ C_p)

whereas the zero is defined as:

Equation 11

1

F_Z= ---------------------------------

2 ⋅ π ⋅ R_c⋅ C_c

14/42

DocID024346 Rev 1

LED2001

Application notes

The embedded compensation network is R_C=70 K, C_C=195 pF while C_Pand C_Ocan be

considered as negligible. The error amplifier output resistance is 240 MΩ, so the relevant

singularities are:

Equation 12

f_Z= 11, 6 kHz

f_{P LF}= 3, 4 Hz

6.4

LED small signal model

Once the system reaches the working condition, the LEDs composing the row are biased

and their equivalent circuit can be considered as a resistor for frequencies << 1 MHz.

The LED manufacturer typically provides the equivalent dynamic resistance of the LED

biased at different DC currents. This parameter is required to study the behavior of the

system in the small signal analysis.

For instance, the equivalent dynamic resistance of the Luxeon III Star from Lumiled

measured with different biasing current level is reported below:

Equation 13

1.3Ω

0.9Ω

I

_LED= 350mA

_LED= 700mA







r_LED

If the LED datasheet does not report the equivalent resistor value, it can be simply derived

as the tangent to the diode I-V characteristic in the present working point (see Figure 7).

DocID024346 Rev 1

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Application notes

LED2001

Figure 7. Equivalent series resistor

[A]

1

working point

0.1

2

[V]

1

3

4

AM12898v1

Figure 8 shows the equivalent circuit of the LED constant current generator.

Figure 8. Load equivalent circuit

L

Dled1

D

Dled2

Rs

VIN

COUT

L

Rd1

Rd2

D1

VIN

COUT

Rs

AM12899v1

16/42

DocID024346 Rev 1

LED2001

Application notes

As a consequence, the LED equivalent circuit gives the α_LED(s) term correlating the output

voltage with the high impedance FB input:

Equation 14

R_SENSE

α_LED(n_LED)= ---------------------------------------------------------

n_LED⋅ r_LED+ R_SENSE

6.5

Total loop gain

In summary, the open loop gain can be expressed as:

Equation 15

G(s) = G_CO(s) ⋅ A₀(s) ⋅ α_LED(n_LED

)

Example 1

Design specification:

V_IN=12 V, V_{FW_LED}=3.5 V, n_LED= 2, r_LED= 1.1 Ω, I_LED= 4 A, I_{LED RIPPLE}= 2%

The inductor and capacitor value are dimensioned in order to meet the I_{LED RIPPLE}

specification (see Section 7.1.2 for output capacitor and inductor selection guidelines):

L=2.2 μH, C_OUT=2.2 μF MLCC (negligible ESR)

Accordingly, with Section 7.1.1 the sensing resistor value is:

Equation 16

100 mV

R_S= -------------------- ≅ 25 mΩ

4A

Equation 17

R_SENSE

25 mΩ

α_LED(n_LED)= --------------------------------------------------------- = --------------------------------------------- = 0.011

n_LED⋅ r_LED+ R_SENSE

2 ⋅ 1.1Ω + 25 mΩ

The gain and phase margin Bode diagrams are plotted respectively in Figure 9 and

Figure 10.

DocID024346 Rev 1

17/42

Application notes

LED2001

Figure 9. Module plot

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The cut-off frequency and the phase margin are:

Equation 18

f_C= 14 kHz

pm = 120°

6.6

Dimming operation

The dimming input disables the switching activity, masking the PWM comparator output.

The inductor current dynamic when dimming input goes high depends on the designed

system response. The best dimming performance is obtained maximizing the bandwidth

and phase margin, when it is possible.

As a general rule, the output capacitor minimization improves the dimming performance.

18/42

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LED2001

Application notes

Figure 11. Dimming operation example

AM12902v1

In fact, when dimming enables the switching activity, a small capacitor value is fast charged

with low inductor value. As a consequence, the LEDs current rising edge time is improved

and the inductor current oscillation reduced. An oversized output capacitor value requires

extra current for fast charge so generating certain inductor current oscillations

The switching activity is prevented as soon as the dimming signal goes low. Nevertheless,

the LED current drops to zero only when the voltage stored in the output capacitor goes

below a minimum voltage determined by the selected LEDs. As a consequence, a big

capacitor value makes the LED current falling time worse than a smaller one.

The LED2001 embeds dedicated circuitry to improve LED current falling time.

As soon as the dimming input goes low, the low-side is kept enabled to discharge C_OUTuntil

the LED current drops to 60% of the nominal current. A negative current limitation (-1 A

typical) protects the device during this operation (see Figure 12).

DocID024346 Rev 1

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Application notes

LED2001

Figure 12. LED current falling edge operation

AM12903v1

6.6.1

Dimming frequency vs. dimming depth

As seen in Section 6.6, the LEDs current rising and falling edge time mainly depends on the

system bandwidth (T_RISE) and the selected output capacitor value (T_RISEand T_FALL).

The dimming performance depends on the minimum current pulse shape specification of

the final application. The ideal minimum current pulse has rectangular shape, however, it

degenerates into a trapezoid or, at worst, into a triangle, depending on the ratio (T_RISE

_FALL)/ T_DIM

+

T

.

Equation 19

rectangle

trapezoid

triangle

T

+ T

T

+ T

T

+ T

→

RISE

FALL

RISE

FALL

RISE

FALL

-------------------------------------------- « 1

-------------------------------------------- < 1

-------------------------------------------- = 1

T

DIM

The small signal response in Figure 11 and Figure 12 is considered as an example.

Equation 20

T

_RISE≅ 20μs







T

_FALL≅ 5μs

Assuming the minimum current pulse shape specification as:

Equation 21:

T

_RISE+ T_FALL= 0.5 ⋅ T_{MIN_PULSE}= 0.5 ⋅ D_MIN⋅ T_DIMMING

it is possible to calculate the maximum dimming depth given the dimming frequency or vice

versa.

20/42

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LED2001

Application notes

Figure 13. Dimming signal

AM12904v1

For example, assuming a 1 kHz dimming frequency the maximum dimming depth is 5% or,

given a 2% dimming depth, it follows a 200 Hz maximum f_DIM

.

The LED2001 dimming performance is strictly dependent on the system small signal

response. As a consequence, an optimized compensation (good phase margin and

bandwidth maximized) and minimized C_OUTvalue are crucial for the best performance.

6.7

eDesign studio software

The LED2001 is supported by the eDesign software which can be viewed online at

www.st.com.

Figure 14. eDesign studio screenshot

The software easily supports the component sizing according to the technical information

given in this datasheet (see Section 6 and Section 7).

The end user is requested to fill in the requested information such as the input voltage

range, the selected LED parameters and the number of LEDs composing the row.

DocID024346 Rev 1

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Application information

LED2001

The software calculates external components according to the internal database. It is also

possible to define new components and ask the software to use them.

Bode plots, estimated efficiency and thermal performance are provided.

Finally, the user can save the design and print all the information including the bill of material

of the board.

7

Application information

7.1

Component selection

7.1.1

Sensing resistor

In closed loop operation the LED2001 feedback pin voltage is 100 mV, so the sensing

resistor calculation is expressed as:

Equation 22

100 mV

R_S= --------------------

I_LED

Since the main loop (see Section 6.1) regulates the sensing resistor voltage drop, the

average current is regulated into the LEDs. The integration period is at minimum 5*T_SW

since the system bandwidth can be dimensioned up to f_SW/5 at maximum.

The system performs the output current regulation over a period which is at least five times

longer than the switching frequency. The output current regulation neglects the ripple

current contribution and its reliance on external parameters like input voltage and output

voltage variations (line transient and LED forward voltage spread). This performance can

not be achieved with simpler regulation loops such as a hysteretic control.

For the same reason, the switching frequency is constant over the application conditions,

which helps to tune the EMI filtering and to guarantee the maximum LED current ripple

specification in the application range. This performance can not be achieved using constant

ON/OFF-time architecture.

7.1.2

Inductor and output capacitor selection

The output capacitor filters the inductor current ripple that, given the application condition,

depends on the inductor value. As a consequence, the LED current ripple, that is the main

specification for a switching current source, depends on the inductor and output capacitor

selection.

22/42

DocID024346 Rev 1

LED2001

Application information

Figure 15. Equivalent circuit

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The LED ripple current can be calculated as the inductor ripple current ratio flowing into the

output impedance using the Laplace transform (see Figure 11):

Equation 23

8

π²

----- ⋅ ΔI_L⋅ (1 + s ⋅ ESR ⋅ C_OUT

)

ΔI_RIPPLE(s) = ----------------------------------------------------------------------------------------------------------

1 + s ⋅ (R_S+ ESR + n_LED⋅ R_LED) ⋅ C_OUT

where the term 8/

π

²represents the main harmonic of the inductor current ripple (which has

a triangular shape) and

Equation 24

V_OUT

ΔI_Lis the inductor current ripple.

n_LED⋅ V_{FW_LED}+ 100mV

ΔI_L= -------------- ⋅ T_OFF= ----------------------------------------------------------------- ⋅ T_OFF

L

so L value can be calculated as:

Equation 25

n_LED⋅ V_{FW_LED}+ 100mV





L = ----------------------------------------------------------------- ⋅ T_OFF= ----------------------------------------------------------------- ⋅ 1 – -----------------------------------------------------------------





ΔI_L

V_IN

where T_OFFis the OFF-time of the embedded high switch, given by 1-D.

As a consequence, the lower the inductor value (so the higher the current ripple), the higher

the C_OUTvalue would be to meet the specification.

A general rule to dimension L value is:

Equation 26

ΔI_L

----------- ≤ 0.5

I_LED

Finally, the required output capacitor value can be calculated equalizing the LED current

ripple specification with the module of the Fourier transformer (see Equation 23) calculated

at f_SWfrequency.

DocID024346 Rev 1

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Application information

Equation 27

LED2001

ΔI_RIPPLE(s=j ⋅ ω) = ΔI_{RIPPLE_SPEC}

Example (see Section Example 1):

V_IN=12 V, I_LED=700 mA, _ILED/I_LED=2%, V_{FW_LED}=3.5 V, n_LED=2.

Δ

A lower inductor value maximizes the inductor current slew rate for better dimming

performance. Equation 26 becomes:

Equation 28

ΔI_L

----------- = 0.5

I_LED

which is satisfied selecting a10 μH inductor value.

The output capacitor value must be dimensioned according to Equation 27.

Finally, given the selected inductor value, a 2.2 μF ceramic capacitor value keeps the LED

current ripple ratio lower than the 2% of the nominal current. An output ceramic capacitor

type (negligible ESR) is suggested to minimize the ripple contribution given a fixed capacitor

value.

Table 6. Inductor selection

Manufacturer

Series

Inductor value (µH)

Saturation current (A)

WE-HCI 7040

WE-HCI 7050

XPL 7030

1 to 4.7

4.9 to 10

2.2 to 10

20 to 7

20 to 4.0

29 to 7.2

Wurth Elektronik

Coilcraft

7.1.3

Input capacitor

The input capacitor must be able to support the maximum input operating voltage and the

maximum RMS input current.

Since step-down converters draw current from the input in pulses, the input current is

squared and the height of each pulse is equal to the output current. The input capacitor

must absorb all this switching current, whose RMS value can be up to the load current

divided by two (worst case, with duty cycle of 50%). For this reason, the quality of these

capacitors must be very high to minimize the power dissipation generated by the internal

ESR, thereby improving system reliability and efficiency. The critical parameter is usually the

RMS current rating, which must be higher than the RMS current flowing through the

capacitor. The maximum RMS input current (flowing through the input capacitor) is:

Equation 29

2 ⋅ D²D²

I_RMS= I_O

⋅

D – -------------- + ------

η²

η

where η is the expected system efficiency, D is the duty cycle and I_Ois the output DC

current. Considering η = 1 this function reaches its maximum value at D = 0.5 and the

24/42

DocID024346 Rev 1

LED2001

Application information

equivalent RMS current is equal to I_Odivided by 2. The maximum and minimum duty cycles

are:

Equation 30

V

_OUT+ V_F

D_MAX= ------------------------------------

_INMIN– V_SW

V

and

Equation 31

V

_OUT+ V_F

D_MIN= -------------------------------------

V

_INMAX– V_SW

where V_Fis the free-wheeling diode forward voltage and V_SWthe voltage drop across the

internal PDMOS. Considering the range D_MINto D_MAX, it is possible to determine the max.

I_RMSgoing through the input capacitor. Capacitors that can be considered are:

–

Electrolytic capacitors:

these are widely used due to their low price and their availability in a wide range of

RMS current ratings.

The only drawback is that, considering ripple current rating requirements, they are

physically larger than other capacitors.

–

Ceramic capacitors:

if available for the required value and voltage rating, these capacitors usually have a

higher RMS current rating for a given physical dimension (due to very low ESR).

The drawback is the considerably high cost.

–

Tantalum capacitors:

small tantalum capacitors with very low ESR are becoming more widely available.

However, they can occasionally burn if subjected to very high current during charge.

Therefore, it is suggested to avoid this type of capacitor for the input filter of the device

as they may be stressed by a high surge current when connected to the power supply.

Table 7. List of ceramic capacitors for the LED2001

Manufacturer

Series

Capacitor value (µC)

Rated voltage (V)

Taiyo yuden

Murata

UMK325BJ106MM-T

10

50

GRM42-2 X7R 475K 50

4.7

If the selected capacitor is ceramic (so neglecting the ESR contribution), the input voltage

ripple can be calculated as:

Equation 32

I_O

D

η





V_{IN PP}= ----------------------- ⋅ 1 – --- ⋅ D + --- ⋅ (1 – D)





C_IN⋅ f_SW

η

DocID024346 Rev 1

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Application information

LED2001

7.2

Layout considerations

The layout of switching DC-DC converters is very important to minimize noise and

interference. Power-generating portions of the layout are the main cause of noise and so

high switching current loop areas should be kept as small as possible and lead lengths as

short as possible.

High impedance paths (in particular the feedback connections) are susceptible to

interference, so they should be as far as possible from the high current paths. A layout

example is provided in Figure 16.

The input and output loops are minimized to avoid radiation and high frequency resonance

problems. The feedback pin to the sensing resistor path must be designed as short as

possible to avoid pick-up noise. Another important issue is the ground plane of the board.

As the package has an exposed pad, it is very important to connect it to an extended ground

plane in order to reduce the thermal resistance junction-to-ambient.

To increase the design noise immunity, different signal and power ground should be

implemented in the layout (see Section 7.5: Application circuit). The signal ground serves

the small signal components, the device analog ground pin, the exposed pad and a small

filtering capacitor connected to the VCC pin. The power ground serves the device ground

pin and the input filter. The different grounds are connected underneath the output capacitor.

Neglecting the current ripple contribution, the current flowing through this component is

constant during the switching activity and so this is the cleanest ground point of the buck

application circuit.

Figure 16. Layout example

26/42

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LED2001

Application information

7.3

Thermal considerations

The dissipated power of the device is tied to three different sources:

Conduction losses due to the R_DSON, which are equal to:

•

Equation 33

P_ON= R_{RDSON_HS}⋅ (I_OUT)²⋅ D

P_OFF= R_{RDSON_LS}⋅ (I_OUT)²⋅ (1 – D)

where D is the duty cycle of the application. Note that the duty cycle is theoretically given by

the ratio between V_OUT(n_LEDV_LED+ 100 mV) and V_IN, but in practice it is substantially

∗

higher than this value to compensate for the losses in the overall application. For this

reason, the conduction losses related to the R_DSONincrease compared to an ideal case.

•

Switching losses due to turn-ON and turn-OFF. These are derived using the following

equation:

Equation 34

(T_RISE+ T_FALL

)

P_SW= V_IN⋅ I_OUT⋅ ---------------------------------------- ⋅ F_SW= V_IN⋅ I_OUT⋅ T_{SW_EQ}⋅ F_SW

2

where T_RISEand T_FALLrepresent the switching times of the power element that cause the

switching losses when driving an inductive load (see Figure 17). T_SWis the equivalent

switching time.

Figure 17. Switching losses

AM12908v1

•

Quiescent current losses.

Equation 35

P_Q= V_IN⋅ I_Q

Example (see Section Example 1):

DocID024346 Rev 1

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Application information

LED2001

V_IN=12 V, V_{FW_LED}=3.5 V, n_LED=2, I_LED=700 mA

The typical output voltage is:

Equation 36

V_OUT= n_LED⋅ V_{FW_LED}+ V_FB= 7.1V

R

_{DSON_HS}has a typical value of 95 mΩ and R_{DSON_LS}is 69 mΩ @ 25 °C.

For the calculation we can estimate R_{DSON_HS}= 140 mΩ and R_{DSON_LS}= 100 mΩ as a

consequence of Tj increase during the operation.

T

_{SW_EQ}is approximately 12 ns.

I_Qhas a typical value of 1.5 mA @ V_IN= 12 V.

The overall losses are:

Equation 37

P_TOT= R_{DSON_HS}⋅ (I_OUT)²⋅ D + R_{DSON_LS}⋅ (I_OUT)²⋅ (1 – D) + V_IN⋅ I_OUT⋅ f_SW⋅ T_SW+ V_IN⋅ I_Q

Equation 38

P_TOT= 0.14 ⋅ 0.7²⋅ 0.6 + 0.1 ⋅ 0.7²⋅ 0.4 + 12 ⋅ 0.7 ⋅ 12 ⋅ 10^–9⋅ 850 ⋅ 10³+ 12 ⋅ 1.5 ⋅ 10^–3≅ 205mW

The junction temperature of the device is:

Equation 39

T_J= T_A+ Rth_{J – A}⋅ P_TOT

where T_Ais the ambient temperature and Rth_J-Ais the thermal resistance junction-to-

ambient. The junction-to-ambient (Rth_J-A) thermal resistance of the device assembled in the

HSO8 package and mounted on the board is about 40 °C/W.

Assuming the ambient temperature is around 40 °C, the estimated junction temperature is:

T_J= 60 + 0.205 ⋅ 40 ≅ 68°C

7.4

Short-circuit protection

In overcurrent protection mode, when the peak current reaches the current limit threshold,

the device disables the power element and it is able to reduce the conduction time down to

the minimum value (approximately 100 nsec typical) to keep the inductor current limited.

This is the pulse-by-pulse current limitation to implement the constant current protection

feature.

In overcurrent condition, the duty cycle is strongly reduced and, in most applications, this is

enough to limit the switch current to the current threshold.

28/42

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LED2001

Application information

The inductor current ripple during ON and OFF phases can be written as:

ON phase

•

Equation 40

V

_IN– V_OUT– (DCR_L+ R_{DSON HS}) ⋅ I

ΔI_{L TON}= ----------------------------------------------------------------------------------------------- (T_ON

)

L

•

OFF phase

Equation 41

–(V_OUT+ (DCR_L+ R_{DSON LS}) ⋅ I)

ΔI_{L TON}= ---------------------------------------------------------------------------------------(T_OFF

)

L

where DCR_Lis the series resistance of the inductor.

The pulse-by-pulse current limitation is effective to implement constant current protection

when:

Equation 42

ΔI_{L TON}= ΔI_{L TOFF}

From Equation 40 and Equation 41 it can be seen that the implementation of the constant

current protection becomes more critical the lower the V_OUTand the higher the V_IN.

In fact, in short-circuit condition the voltage applied to the inductor during the OFF-time

becomes equal to the voltage drop across parasitic components (typically the DCR of the

inductor and the R_DSONof the low-side switch) since VOUT is negligible, while during T_ON

the voltage applied at the inductor is maximized and is approximately equal to V_IN.

In general, the worst case scenario is heavy short-circuit at the output with maximum input

voltage. Equation 40 and Equation 41 in overcurrent conditions can be simplified to:

Equation 43

V

_IN–(DCR_L+ R_{DSON HS}) ⋅ I

V_IN

ΔI_{L TON}= ------------------------------------------------------------------------ (T_{ON MIN}) ≅ --------(90ns)

L

considering T_ONwhich has already been reduced to its minimum.

Equation 44

–(DCR_L+ R_{DSON LS}) ⋅ I

ΔI_{L TOFF}= -------------------------------------------------------------(T_SW– 90ns) ≅ -------------------------------------------------------------(1.18μs)

L

where T_SW=1/f_SWand considering the nominal f_SW

.

At higher input voltage I_{L TON}may be higher than

escalate. As a consequence, the system typically meets Equation 42 at a current level

higher than the nominal value thanks to the increased voltage drop across stray

Δ

I_{L TOFF}and so the inductor current can

components. In most of the application conditions the pulse-by-pulse current limitation is

effective to limit the inductor current. Whenever the current escalates, a second level current

protection called “Hiccup mode” is enabled. Hiccup protection offers an additional protection

against heavy short-circuit conditions at very high input voltage even considering the spread

DocID024346 Rev 1

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Application information

LED2001

of the minimum conduction time of the power element. If the hiccup current level (6.2 A

typical) is triggered, the switching activity is prevented for 12 cycles.

Figure 18 shows the operation of the constant current protection when a short-circuit is

applied at the output at the maximum input voltage.

Figure 18. Constant current protection triggering Hiccup mode

During pulse skipping, high side OFF, low side keeps ON till 26clks finish (13clks for

LED2001) or Ipk decreases to be zero value.

7.5

Application circuit

Figure 19. Demonstration board application circuit

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30/42

DocID024346 Rev 1

LED2001

Application information

Description Manufacturer

Table 8. Component list

Reference

Part number

1 μF 25 V

(size 0805)

C1

22 μF 25 V

(size 1206)

C2

C3

GRM31CR61E226KE15L

GRM21BR71E475KA73L

Murata

4.7 μF 25 V

(size 0805)

4.7 KΩ 5%

(size 0603)

R1

R2

Rs

Not mounted

0.15 Ω 1%

(size 1206)

ERJ14BSFR15U

Panasonic

3.3 μH

I

_SAT= 8.4 A

L1

XAL6030-332MEB

(20% drop) I_RMS= 7.3 A

Coilcraft

(40 °C rise)

(size 6.36 x 6.56 x 6.1 mm)

Figure 20. PCB layout (component side) DFN package

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Application information

LED2001

Figure 21. PCB layout (bottom side) DFN package

Figure 22. PCB layout (component side) HSOP8 package

It is strongly recommended that the input capacitors are to be put as close as possible to the

pins, see C1 and C2.

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Application information

Figure 23. PCB layout (bottom side) HSOP8 package

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Typical characteristics

LED2001

8

Typical characteristics

Figure 24. Soft-start

Figure 25. Load regulation

Vin 12V

Vled 7V

AM12914v1

AM12913v1

Figure 26. Dimming operation

Figure 27. LED current rising edge

a

AM12915v1

AM12916v1

Figure 28. LED current falling edge

Figure 29. Hiccup current protection

To maximize the dimming

performance the embedded LS

discharges C _OUTwhen DIM goes low.

(DIM = 0 && V _FB> 60mV):

the low side is enabled

as long as I > -1A

L

(implements negative

current limitation)

AM12918v1

AM12917v1

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Typical characteristics

Figure 30. OCP blanking time

Figure 31. Thermal shutdown protection

130 ns typ.

AM12920v1

AM12919v1

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Ordering information

LED2001

9

Ordering information

Table 9. Ordering information

Package

Order code

Packaging

LED2001PUR

LED2001PHR

VFQFPN 4x4 8L

HSOP8

Tape and reel

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Package mechanical data

10

Package mechanical data

In order to meet environmental requirements, ST offers these devices in different grades of

ECOPACK^®packages, depending on their level of environmental compliance. ECOPACK^®

specifications, grade definitions and product status are available at: www.st.com.

ECOPACK^®is an ST trademark.

Table 10. VFQFPN8 (4x4x1.08 mm) mechanical data

mm

Dim.

Min.

Typ.

Max.

A

0.80

0.90

0.02

0.20

1.00

0.05

A1

A3

b

D

0.23

3.90

2.82

3.90

2.05

0.30

4.00

3.00

4.00

2.20

0.80

0.50

0.38

4.10

3.23

4.10

2.30

D2

E

E2

e

L

0.40

0.60

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Package mechanical data

LED2001

Figure 32. VFQFPN8 (4x4x1.08 mm) package dimensions

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Package mechanical data

Table 11. HSOP8 mechanical data

mm

Dim

Min.

Typ.

Max.

A

A1

A2

b

1.70

0.00

1.25

0.31

0.17

4.80

5.80

3.80

0.150

0.51

0.25

5.00

6.20

4.00

c

D

4.90

6.00

3.90

1.27

E

E1

e

h

0.25

0.40

0.00

0.50

1.27

8.00

0.10

L

k

ccc

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Package mechanical data

LED2001

Figure 33. HSOP8 package dimensions

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(ꢀ ꢀꢉꢀꢁꢋPPꢋ7\Sꢉ

$0ꢂꢂꢆꢊꢊYꢂ

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Revision history

11

Revision history

Table 12. Document revision history

Changes

Date

Revision

20-May-2013

1

Initial release.

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LED2001

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right to make changes, corrections, modifications or improvements, to this document, and the products and services described herein at any

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型号：	LED2001PUR
厂家：	ST
描述：	4 A monolithic step-down current source with synchronous rectification
文件：	总42页 (文件大小：2433K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

LED2001PUR [STMICROELECTRONICS]

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