C4532X7R2A225M [TI]
LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs; LM3402 / LM3402HV 0.5A恒流降压稳压器用于驱动高功率LED
| 型号： | C4532X7R2A225M |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs |
| 文件： | 总30页 (文件大小：570K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LM3402,LM3402HV
LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High
Power LEDs
Literature Number: SNVS450D
February 5, 2010
LM3402/LM3402HV
0.5A Constant Current Buck Regulator for Driving High
Power LEDs
General Description
Features
The LM3402/02HV are monolithic switching regulators de-
signed to deliver constant currents to high power LEDs. Ideal
for automotive, industrial, and general lighting applications,
they contain a high-side N-channel MOSFET switch with a
current limit of 735 mA (typical) for step-down (Buck) regula-
tors. Hysteretic control with controlled on-time coupled with
an external resistor allow the converter output voltage to ad-
just as needed to deliver a constant current to series and
series - parallel connected arrays of LEDs of varying number
and type, LED dimming by pulse width modulation (PWM),
broken/open LED protection, low-power shutdown and ther-
mal shutdown complete the feature set.
Integrated 0.5A N-channel MOSFET
■
■
■
■
■
■
■
■
VIN Range from 6V to 42V (LM3402)
VIN Range from 6V to 75V (LM3402HV)
500 mA Output Current Over Temperature
Cycle-by-Cycle Current Limit
No Control Loop Compensation Required
Separate PWM Dimming and Low Power Shutdown
Supports all-ceramic output capacitors and capacitor-less
outputs
Thermal shutdown protection
■
■
MSOP-8, PSOP-8 Packages
Applications
LED Driver
■
■
■
■
Constant Current Source
Automotive Lighting
General Illumination
Industrial Lighting
■
Typical Application
20192101
© 2010 National Semiconductor Corporation
201921
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Connection Diagrams
20192145
8-Lead Plastic PSOP-8 Package
NS Package Number MRA08B
20192102
8-Lead Plastic MSOP-8 Package
NS Package Number MUA08A
Ordering Information
Order Number
LM3402MM
Package Type
NSC Package Drawing
Supplied As
1000 units on tape and reel
3500 units on tape and reel
1000 units on tape and reel
3500 units on tape and reel
95 units in anti-static rails
2500 units on tape and reel
95 units in anti-static rails
2500 units on tape and reel
LM3402MMX
LM3402HVMM
LM3402HVMMX
LM3402MR
MSOP-8
MUA08A
LM3402MRX
LM3402HVMR
LM3402HVMRX
PSOP-8
MRA08B
Pin Descriptions
Pin(s)
Name
SW
Description
Application Information
Connect this pin to the output inductor and Schottky diode.
1
2
3
Switch pin
BOOT
DIM
MOSFET drive bootstrap pin Connect a 10 nF ceramic capacitor from this pin to SW.
Input for PWM dimming
Connect a logic-level PWM signal to this pin to enable/disable the
power FET and reduce the average light output of the LED array.
4
5
GND
CS
Ground pin
Connect this pin to system ground.
Current sense feedback pin
Set the current through the LED array by connecting a resistor from
this pin to ground.
6
7
RON
VCC
VIN
On-time control pin
A resistor connected from this pin to VIN sets the regulator controlled
on-time.
Output of the internal 7V linear Bypass this pin to ground with a minimum 0.1 µF ceramic capacitor
regulator
with X5R or X7R dielectric.
8
Input voltage pin
Nominal operating input range is 6V to 42V (LM3402) or 6V to 75V
(LM3402HV).
DAP
GND
Thermal Pad
Connect to ground. Place 4 to 6 vias from DAP to bottom layer ground
plane.
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Absolute Maximum Ratings
Operating Ratings
(LM3402) (Note 1)
(LM3402) (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
VIN
6V to 42V
−40°C to +125°C
Junction Temperature Range
Thermal Resistance θJA (MSOP-8 Package)
(Note 3)
VIN to GND
BOOT to GND
SW to GND
-0.3V to 45V
-0.3V to 59V
-1.5V
200°C/W
50°C/W
Thermal Resistance θJA (PSOP-8 Package)
(Note 5)
BOOT to VCC
BOOT to SW
VCC to GND
DIM to GND
CS to GND
-0.3V to 45V
-0.3V to 14V
-0.3V to 14V
-0.3V to 7V
-0.3V to 7V
-0.3V to 7V
150°C
RON to GND
Junction Temperature
Storage Temp. Range
ESD Rating (Note 2)
Soldering Information
-65°C to 125°C
2kV
Lead Temperature (Soldering,
10sec)
Infrared/Convection Reflow (15sec)
260°C
235°C
3
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Storage Temp. Range
ESD Rating (Note 2)
Soldering Information
Lead Temperature (Soldering,
10sec)
-65°C to 125°C
2kV
Absolute Maximum Ratings
(LM3402HV) (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
260°C
235°C
Infrared/Convection Reflow (15sec)
VIN to GND
-0.3V to 76V
-0.3V to 90V
-1.5V
Operating Ratings
BOOT to GND
SW to GND
(LM3402HV) (Note 1)
BOOT to VCC
BOOT to SW
VCC to GND
DIM to GND
CS to GND
RON to GND
Junction Temperature
-0.3V to 76V
-0.3V to 14V
-0.3V to 14V
-0.3V to 7V
-0.3V to 7V
-0.3V to 7V
150°C
VIN
6V to 75V
−40°C to +125°C
Junction Temperature Range
Thermal Resistance θJA (MSOP-8 Package)
(Note 3)
200°C/W
50°C/W
Thermal Resistance θJA (PSOP-8 Package)
(Note 5)
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Electrical Characteristics VIN = 24V unless otherwise indicated. Typicals and limits appearing in plain type apply
for TA = TJ = +25°C. (Note 4) Limits appearing in boldface type apply over full Operating Temperature Range. Datasheet min/max
specification limits are guaranteed by design, test, or statistical analysis.
LM3402
Symbol
Parameter
Conditions
Min
Typ
Max
Units
SYSTEM PARAMETERS
tON-1
tON-2
On-time 1
On-time 2
2.1
2.75
650
3.4
µs
ns
VIN = 10V, RON = 200 kΩ
VIN = 40V, RON = 200 kΩ
490
810
LM3402HV
Symbol
Parameter
Conditions
Min
Typ
Max
Units
SYSTEM PARAMETERS
tON-1 On-time 1
tON-2 On-time 2
2.1
2.75
380
3.4
µs
ns
VIN = 10V, RON = 200 kΩ
VIN = 70V, RON = 200 kΩ
290
470
LM3402/LM3402HV
Symbol
Parameter
Conditions
Min
194
Typ
Max
206
Units
REGULATION AND OVER-VOLTAGE COMPARATORS
VREF-REG
VREF-0V
CS Regulation Threshold
CS Over-voltage Threshold
CS Bias Current
CS Decreasing, SW turns on
CS Increasing, SW turns off
CS = 0V
200
300
0.1
mV
mV
µA
ICS
SHUTDOWN
VSD-TH
Shutdown Threshold
Shutdown Hysteresis
RON / SD Increasing
RON / SD Decreasing
0.3
6.6
0.7
40
1.05
7.4
V
VSD-HYS
OFF TIMER
tOFF-MIN
mV
Minimum Off-time
CS = 0V
300
ns
INTERNAL REGULATOR
VCC-REG VCC Regulated Output
VIN-DO
7
V
mV
V
VIN - VCC Dropout
ICC = 5 mA, 6.0V < VIN < 8.0V
VIN Increasing
VIN Decreasing
VIN = 6V
300
8.8
225
55
VCC-BP-TH
VCC-BP-HYS
VCC-Z-6
VCC Bypass Threshold
VCC Bypass Hysteresis
mV
Ω
VCC Output Impedance
(0 mA < ICC < 5 mA)
VCC-Z-8
VIN = 8V
50
VCC-Z-24
VCC-LIM
VIN = 24V
0.4
16
VCC Current Limit (Note 3)
VIN = 24V, VCC = 0V
VCC Increasing
mA
V
VCC-UV-TH
VCC Under-voltage Lock-out
Threshold
5.25
VCC-UV-HYS
VCC-UV-DLY
VCC Under-voltage Lock-out
Hysteresis
VCC Decreasing
150
3
mV
µs
VCC Under-voltage Lock-out
Filter Delay
100 mV Overdrive
IIN-OP
IIN Operating Current
Non-switching, CS = 0V
RON / SD = 0V
600
90
900
180
µA
µA
IIN-SD
IIN Shutdown Current
CURRENT LIMIT
ILIM
Current Limit Threshold
530
735
940
mA
5
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Symbol
Parameter
Conditions
DIM Increasing
Min
2.2
Typ
Max
0.8
Units
DIM COMPARATOR
VIH
Logic High
Logic Low
V
V
VIL
DIM Decreasing
DIM = 1.5V
IDIM-PU
DIM Pull-up Current
75
µA
N-MOSFET AND DRIVER
RDS-ON
Buck Switch On Resistance
ISW = 200mA, BOOT-SW = 6.3V
0.7
3
1.5
4
Ω
V
VDR-UVLO
BOOT Under-voltage Lock-out BOOT–SW Increasing
1.7
Threshold
VDR-HYS
BOOT Under-voltage Lock-out BOOT–SW Decreasing
Hysteresis
400
mV
THERMAL SHUTDOWN
TSD
Thermal Shutdown Threshold
165
25
°C
°C
TSD-HYS
Thermal Shutdown Hysteresis
Junction to Ambient
THERMAL RESISTANCE
MSOP-8 Package
PSOP-8 Package
200
50
°C/W
θJA
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see Electrical Characteristics.
Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.
Note 3: VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading.
Note 4: Typical specifications represent the most likely parametric norm at 25°C operation.
Note 5: θJA of 50°C/W with DAP soldered to a minimum of 2 square inches of 1 oz. copper on the top or bottom PCB layer.
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Typical Performance Characteristics
VREF vs Temperature (VIN = 24V)
VREF vs VIN, LM3402 (TA = 25°C)
20192129
20192130
VREF vs VIN, LM3402HV (TA = 25°C)
Current Limit vs Temperature (VIN = 24V)
20192131
20192132
Current Limit vs VIN, LM3402 (TA = 25°C)
Current Limit vs VIN, LM3402HV (TA = 25°C)
20192133
20192134
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TON vs VIN,
RON = 100 kΩ (TA = 25°C)
TON vs VIN,
(TA = 25°C)
20192136
20192135
20192137
20192138
TON vs VIN,
(TA = 25°C)
TON vs RON, LM3402
(TA = 25°C)
20192144
TON vs RON, LM3402HV
(TA = 25°C)
VCC vs VIN
(TA = 25°C)
20192139
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VO-MAX vs fSW, LM3402
(TA = 25°C)
VO-MIN vs fSW, LM3402
(TA = 25°C)
20192140
20192141
VO-MAX vs fSW, LM3402HV
(TA = 25°C)
VO-MIN vs fSW, LM3402HV
(TA = 25°C)
20192142
20192143
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Block Diagram
20192103
created as the LED current flows through the current setting
resistor, RSNS, to ground. VSNS is fed back to the CS pin,
where it is compared against a 200 mV reference, VREF. The
on-comparator turns on the power MOSFET when VSNS falls
below VREF. The power MOSFET conducts for a controlled
on-time, tON, set by an external resistor, RON, and by the input
voltage, VIN. On-time is governed by the following equation:
Application Information
THEORY OF OPERATION
The LM3402 and LM3402HV are buck regulators with a wide
input voltage range, low voltage reference, and a fast output
enable/disable function. These features combine to make
them ideal for use as a constant current source for LEDs with
forward currents as high as 500 mA. The controlled on-time
(COT) architecture is a combination of hysteretic mode con-
trol and a one-shot on-timer that varies inversely with input
voltage. Hysteretic operation eliminates the need for small-
signal control loop compensation. When the converter runs in
continuous conduction mode (CCM) the controlled on-time
maintains a constant switching frequency over the range of
input voltage. Fast transient response, PWM dimming, a low
power shutdown mode, and simple output overvoltage pro-
tection round out the functions of the LM3402/02HV.
At the conclusion of tON the power MOSFET turns off for a
minimum off-time, tOFF-MIN, of 300 ns. Once tOFF-MIN is com-
plete the CS comparator compares VSNS and VREF again,
waiting to begin the next cycle.
CONTROLLED ON-TIME OVERVIEW
Figure 1 shows the feedback system used to control the cur-
rent through an array of LEDs. A voltage signal, VSNS, is
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20192105
FIGURE 1. Comparator and One-Shot
The LM3402/02HV regulators should be operated in contin-
uous conduction mode (CCM), where inductor current stays
positive throughout the switching cycle. During steady-state
operationin the CCM, the converter maintains a constant
switching frequency, which can be selected using the follow-
ing equation:
The maximum number of LEDs, nMAX, that can be placed in
a single series string is governed by VO(MAX) and the maxi-
mum forward voltage of the LEDs used, VF(MAX), using the
expression:
VF = forward voltage of each LED, n = number of LEDs in
series
At low switching frequency the maximum duty cycle and out-
put voltage are higher, allowing the LM3402/02HV to regulate
output voltages that are nearly equal to input voltage. The
following equation relates switching frequency to maximum
output voltage.
AVERAGE LED CURRENT ACCURACY
The COT architecture regulates the valley of ΔVSNS, the AC
portion of VSNS. To determine the average LED current (which
is also the average inductor current) the valley inductor cur-
rent is calculated using the following expression:
In this equation tSNS represents the propagation delay of the
CS comparator, and is approximately 220 ns. The average
inductor/LED current is equal to IL-MIN plus one-half of the in-
ductor current ripple, ΔiL:
MINIMUM OUTPUT VOLTAGE
The minimum recommended on-time for the LM3402/02HV is
300 ns. This lower limit for tON determines the minimum duty
cycle and output voltage that can be regulated based on input
voltage and switching frequency. The relationship is deter-
mined by the following equation:
IF = IL = IL-MIN + ΔiL / 2
Detailed information for the calculation of ΔiL is given in the
Design Considerations section.
MAXIMUM OUTPUT VOLTAGE
The 300 ns minimum off-time limits on the maximum duty cy-
cle of the converter, DMAX, and in turn ,the maximum output
voltage VO(MAX) is determined by the following equations:
11
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HIGH VOLTAGE BIAS REGULATOR
current while the MOSFET is on) exceeds 735 mA (typical).
The power MOSFET is disabled for a cool-down time that is
10x the steady-state on-time. At the conclusion of this cool-
down time the system re-starts. If the current limit condition
persists the cycle of cool-down time and restarting will con-
tinue, creating a low-power hiccup mode, minimizing thermal
stress on the LM3402/02HV and the external circuit compo-
nents.
The LM3402/02HV contains an internal linear regulator with
a 7V output, connected between the VIN and the VCC pins.
The VCC pin should be bypassed to the GND pin with a 0.1
µF ceramic capacitor connected as close as possible to the
pins of the IC. VCC tracks VIN until VIN reaches 8.8V (typical)
and then regulates at 7V as VIN increases. Operation begins
when VCC crosses 5.25V.
OVER-VOLTAGE/OVER-CURRENT COMPARATOR
INTERNAL MOSFET AND DRIVER
The CS pin includes an output over-voltage/over-current
comparator that will disable the power MOSFET whenever
VSNS exceeds 300 mV. This threshold provides a hard limit
for the output current. Output current overshoot is limited to
300 mV / RSNS by this comparator during transients.
The LM3402/02HV features an internal power MOSFET as
well as a floating driver connected from the SW pin to the
BOOT pin. Both rise time and fall time are 20 ns each (typical)
and the approximate gate charge is 3 nC. The high-side rail
for the driver circuitry uses a bootstrap circuit consisting of an
internal high-voltage diode and an external 10 nF capacitor,
CB. VCC charges CB through the internal diode while the power
MOSFET is off. When the MOSFET turns on, the internal
diode reverse biases. This creates a floating supply equal to
the VCC voltage minus the diode drop to drive the MOSFET
when its source voltage is equal to VIN.
The OVP/OCP comparator can also be used to prevent the
output voltage from rising to VO(MAX) in the event of an output
open-circuit. This is the most common failure mode for LEDs,
due to breaking of the bond wires. In a current regulator an
output open circuit causes VSNS to fall to zero, commanding
maximum duty cycle. Figure 2 shows a method using a zener
diode, Z1, and zener limiting resistor, RZ, to limit output volt-
age to the reverse breakdown voltage of Z1 plus 200 mV. The
zener diode reverse breakdown voltage, VZ, must be greater
than the maximum combined VF of all LEDs in the array. The
maximum recommended value for RZ is 1 kΩ.
As discussed in the Maximum Output Voltage section, there
is a limit to how high VO can rise during an output open-circuit
that is always less than VIN. If no output capacitor is used, the
output stage of the LM3402/02HV is capable of withstanding
VO(MAX) indefinitely, however the voltage at the output end of
the inductor will oscillate and can go above VIN or below 0V.
A small (typically 10 nF) capacitor across the LED array
dampens this oscillation. For circuits that use an output ca-
pacitor, the system can still withstand VO(MAX) indefinitely as
long as CO is rated to handle VIN. The high current paths are
blocked in output open-circuit and the risk of thermal stress is
minimal, hence the user may opt to allow the output voltage
to rise in the case of an open-circuit LED failure.
FAST SHUTDOWN FOR PWM DIMMING
The DIM pin of the LM3402/02HV is a TTL logic compatible
input for low frequency PWM dimming of the LED. A logic low
(below 0.8V) at DIM will disable the internal MOSFET and
shut off the current flow to the LED array. While the DIM pin
is in a logic low state the support circuitry (driver, bandgap,
VCC) remains active in order to minimize the time needed to
turn the LED array back on when the DIM pin sees a logic
high (above 2.2V). A 75 µA (typical) pull-up current ensures
that the LM3402/02HV is on when DIM pin is open circuited,
eliminating the need for a pull-up resistor. Dimming frequen-
cy, fDIM, and duty cycle, DDIM, are limited by the LED current
rise time and fall time and the delay from activation of the DIM
pin to the response of the internal power MOSFET. In general,
fDIM should be at least one order of magnitude lower than the
steady state switching frequency in order to prevent aliasing.
PEAK CURRENT LIMIT
The current limit comparator of the LM3402/02HV will engage
whenever the power MOSFET current (equal to the inductor
20192112
FIGURE 2. Output Open Circuit Protection
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LOW POWER SHUTDOWN
long as the logic low voltage is below the over temperature
minimum threshold of 0.3V. Noise filter circuitry on the RON
pin can cause a few pulses with a longer on-time than normal
after RON is grounded or released. In these cases the OVP/
OCP comparator will ensure that the peak inductor or LED
The LM3402/02HV can be switched to a low power state (IIN-
= 90 µA) by grounding the RON pin with a signal-level
SD
MOSFET as shown in Figure 3. Low power MOSFETs like the
2N7000, 2N3904, or equivalent are recommended devices
for putting the LM3402/02HV into low power shutdown. Logic
gates can also be used to shut down the LM3402/02HV as
current does not exceed 300 mV / RSNS
.
20192113
FIGURE 3. Low Power Shutdown
THERMAL SHUTDOWN
ceeded. The threshold for thermal shutdown is 165°C with a
25°C hysteresis (both values typical). During thermal shut-
down the MOSFET and driver are disabled.
Internal thermal shutdown circuitry is provided to protect the
IC in the event that the maximum junction temperature is ex-
13
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ered, making the magnetics smaller and less expensive.
Alternatively, the circuit could be run at lower frequency but
keep the same inductor value, improving the efficiency and
expanding the range of output voltage that can be regulated.
Both the peak current limit and the OVP/OCP comparator still
monitor peak inductor current, placing a limit on how large
ΔiL can be even if ΔiF is made very small. A parallel output
capacitor is also useful in applications where the inductor or
input voltage tolerance is poor. Adding a capacitor that re-
duces ΔiF to well below the target provides headroom for
changes in inductance or VIN that might otherwise push the
peak LED ripple current too high.
Design Considerations
SWITCHING FREQUENCY
Switching frequency is selected based on the tradeoffs be-
tween efficiency (better at low frequency), solution size/cost
(smaller at high frequency), and the range of output voltage
that can be regulated (wider at lower frequency.) Many appli-
cations place limits on switching frequency due to EMI sen-
sitivity. The on-time of the LM3402/02HV can be programmed
for switching frequencies ranging from the 10’s of kHz to over
1 MHz. The maximum switching frequency is limited only by
the minimum on-time requirement.
Figure 4 shows the equivalent impedances presented to the
inductor current ripple when an output capacitor, CO, and its
equivalent series resistance (ESR) are placed in parallel with
the LED array. The entire inductor ripple current flows through
RSNS to provide the required 25 mV of ripple voltage for proper
operation of the CS comparator.
LED RIPPLE CURRENT
Selection of the ripple current, ΔiF, through the LED array is
analogous to the selection of output ripple voltage in a stan-
dard voltage regulator. Where the output ripple in a voltage
regulator is commonly ±1% to ±5% of the DC output voltage,
LED manufacturers generally recommend values for ΔiF
ranging from ±5% to ±20% of IF. Higher LED ripple current
allows the use of smaller inductors, smaller output capacitors,
or no output capacitors at all. The advantages of higher ripple
current are reduction in the solution size and cost. Lower rip-
ple current requires more output inductance, higher switching
frequency, or additional output capacitance. The advantages
of lower ripple current are a reduction in heating in the LED
itself and greater range of the average LED current before the
current limit of the LED or the driving circuitry is reached.
BUCK CONVERTERS WITHOUT OUTPUT CAPACITORS
The buck converter is unique among non-isolated topologies
because of the direct connection of the inductor to the load
during the entire switching cycle. By definition an inductor will
control the rate of change of current that flows through it, and
this control over current ripple forms the basis for component
selection in both voltage regulators and current regulators. A
current regulator such as the LED driver for which the
LM3402/02HV was designed focuses on the control of the
current through the load, not the voltage across it. A constant
current regulator is free of load current transients, and has no
need of output capacitance to supply the load and maintain
output voltage. Referring to the Typical Application circuit on
the front page of this datasheet, the inductor and LED can
form a single series chain, sharing the same current. When
no output capacitor is used, the same equations that govern
inductor ripple current, ΔiL, also apply to the LED ripple cur-
rent, ΔiF. For a controlled on-time converter such as
LM3402/02HV the ripple current is described by the following
expression:
20192115
FIGURE 4. LED and CO Ripple Current
To calculate the respective ripple currents the LED array is
represented as a dynamic resistance, rD. LED dynamic resis-
tance is not always specified on the manufacturer’s
datasheet, but it can be calculated as the inverse slope of the
LED’s VF vs. IF curve. Note that dividing VF by IF will give an
incorrect value that is 5x to 10x too high. Total dynamic re-
sistance for a string of n LEDs connected in series can be
calculated as the rD of one device multiplied by n. Inductor
ripple current is still calculated with the expression from Buck
Regulators without Output Capacitors. The following equa-
tions can then be used to estimate ΔiF when using a parallel
capacitor:
A minimum ripple voltage of 25 mV is recommended at the
CS pin to provide good signal-to-noise ratio (SNR). The CS
pin ripple voltage, ΔVSNS, is described by the following:
The calculation for ZC assumes that the shape of the inductor
ripple current is approximately sinusoidal.
ΔVSNS = ΔiF x RSNS
Small values of CO that do not significantly reduce ΔiF can
also be used to control EMI generated by the switching action
of the LM3402/02HV. EMI reduction becomes more important
as the length of the connections between the LED and the
rest of the circuit increase.
BUCK CONVERTERS WITH OUTPUT CAPACITORS
A capacitor placed in parallel with the LED or array of LEDs
can be used to reduce the LED current ripple while keeping
the same average current through both the inductor and the
LED array. This technique is demonstrated in Design Exam-
ple 1. With this topology the output inductance can be low-
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INPUT CAPACITORS
dependent on the selection of D1 at low duty cycles, where
the recirculating diode carries the load current for an increas-
ing percentage of the time. This power dissipation can be
calculated by checking the typical diode forward voltage, VD,
from the I-V curve on the product datasheet and then multi-
plying it by ID. Diode datasheets will also provide a typical
junction-to-ambient thermal resistance, θJA, which can be
used to estimate the operating die temperature of the Schot-
tky. Multiplying the power dissipation (PD = ID x VD) by θJA
gives the temperature rise. The diode case size can then be
selected to maintain the Schottky diode temperature below
the operational maximum.
Input capacitors at the VIN pin of the LM3402/02HV are se-
lected using requirements for minimum capacitance and rms
ripple current. The input capacitors supply pulses of current
approximately equal to IF while the power MOSFET is on, and
are charged up by the input voltage while the power MOSFET
is off. Switching converters such as the LM3402/02HV have
a negative input impedance due to the decrease in input cur-
rent as input voltage increases. This inverse proportionality of
input current to input voltage can cause oscillations (some-
times called ‘power supply interaction’) if the magnitude of the
negative input impedance is greater the the input filter
impedance. Minimum capacitance can be selected by com-
paring the input impedance to the converter’s negative resis-
tance; however this requires accurate calculation of the input
voltage source inductance and resistance, quantities which
can be difficult to determine. An alternative method to select
the minimum input capacitance, CIN(MIN), is to select the max-
imum voltage ripple which can be tolerated. This value,ΔvIN
(MAX), is equal to the change in voltage across CIN during the
converter on-time, when CIN supplies the load current. CIN
(MIN) can be selected with the following:
LED CURRENT DURING DIM MODE
The LM3402 contains high speed MOSFET gate drive cir-
cuitry that switches the main internal power MOSFET be-
tween “on” and “off” states. This circuitry uses current derived
from the VCC regulator to charge the MOSFET during turn-
on, then dumps current from the MOSFET gate to the source
(the SW pin) during turn-off. As shown in the block diagram,
the MOSFET drive circuitry contains a gate drive under-volt-
age lockout (UVLO) circuit that ensures the MOSFET remains
off when there is inadequate VCC voltage for proper operation
of the driver. This watchdog circuitry is always running in-
cluding during DIM and shutdown modes, and supplies a
small amount of current from VCC to SW. Because the SW
pin is connected directly to the LEDs through the buck induc-
tor, this current returns to ground through the LEDs. The
amount of current sourced is a function of the SW voltage, as
shown in Figure 5.
A good starting point for selection of CIN is to use an input
voltage ripple of 5% to 10% of VIN. A minimum input capaci-
tance of 2x the CIN(MIN) value is recommended for all
LM3402/02HV circuits. To determine the rms current rating,
the following formula can be used:
Ceramic capacitors are the best choice for the input to the
LM3402/02HV due to their high ripple current rating, low ESR,
low cost, and small size compared to other types. When se-
lecting a ceramic capacitor, special attention must be paid to
the operating conditions of the application. Ceramic capaci-
tors can lose one-half or more of their capacitance at their
rated DC voltage bias and also lose capacitance with ex-
tremes in temperature. A DC voltage rating equal to twice the
expected maximum input voltage is recommended. In addi-
tion, the minimum quality dielectric which is suitable for
switching power supply inputs is X5R, while X7R or better is
preferred.
20192157
RECIRCULATING DIODE
The LM3402/02HV is a non-synchronous buck regulator that
requires a recirculating diode D1 (see the Typical Application
circuit) to carrying the inductor current during the MOSFET
off-time. The most efficient choice for D1 is a Schottky diode
due to low forward drop and near-zero reverse recovery time.
D1 must be rated to handle the maximum input voltage plus
any switching node ringing when the MOSFET is on. In prac-
tice all switching converters have some ringing at the switch-
ing node due to the diode parasitic capacitance and the lead
inductance. D1 must also be rated to handle the average cur-
rent, ID, calculated as:
FIGURE 5. LED Current From SW Pin
Though most power LEDs are designed to run at several
hundred milliamps, some can be seen to glow with a faint light
at extremely low current levels, as low as a couple microamps
in some instances. In lab testing, the forward voltage was
found to be approximately 2V for LEDs that exhibited visible
light at these low current levels. For LEDs that did not show
light emission at very low current levels, the forward voltage
was found to be around 900mV. It is important to remember
that the forward voltage is also temperature dependent, de-
creasing at higher temperatures. Consequently, with a maxi-
mum Vcc voltage of 7.4V, current will be observed in the LEDs
if the total stack voltage is less than about 6V at a forward
current of several microamps. No current is observed if the
stack voltage is above 6V, as shown in Figure 5. The need for
ID = (1 – D) x IF
This calculation should be done at the maximum expected
input voltage. The overall converter efficiency becomes more
15
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absolute darkness during DIM mode is also application de-
pendent. It will not affect regular PWM dimming operation.
represented as a short from the pin to ground as the extreme
localized heat of the ESD / EOS event causes the aluminum
metal on the chip to melt, causing the short. This situation is
common to all integrated circuits and not just unique to the
LM340X device.
The fix for this issue is extremely simple. Place a resistor from
the SW pin to ground according to the chart below.
Number of LEDs
Resistor Value (kΩ)
CS PIN PROTECTION
1
2
20
50
When hot swapping in a load (e.g. test points, load boards,
LED stack), any residual charge on the load will be immedi-
ately transferred through the output capacitor to the CS pin,
which is then damaged as shown in Figure 6 below. The EOS
event due to the residual charge from the load is represented
3
90
4
150
200
300
5
as VTRANSIENT
.
>5
From measurements, we know that the 8V ESD structure on
the CS pin can typically withstand 25mA of direct current
(DC). Adding a 1kΩ resistor in series with the CS pin, shown
in Figure 7, results in the majority of the transient energy to
pass through the discrete sense resistor rather than the de-
vice. The series resistor limits the peak current that can flow
during a transient event, thus protecting the CS pin. With the
1kΩ resistor shown, a 33V, 49A transient on the LED return
connector terminal could be absorbed as calculated by:
The luminaire designer should ensure that the suggested re-
sistor is effective in eliminating the off-state light output. A
combination of calculations based on LED manufacturer data
and lab measurements over temperature will ensure the best
design.
Transient Protection
Considerations
Considerations need to be made when external sources,
loads or connections are made to the switching converter cir-
cuit due to the possibility of Electrostatic Discharge (ESD) or
Electric Over Stress (EOS) events occurring and damaging
the integrated circuit (IC) device. All IC device pins contain
zener based clamping structures that are meant to clamp
ESD. ESD events are very low energy events, typically less
than 5µJ (microjoules). Any event that transfers more energy
than this may damage the ESD structure. Damage is typically
V = 25mA * 1kΩ + 8V = 33V
I = 33V / 0.67Ω = 49A
This is an extremely high energy event, so the protection
measures previously described should be adequate to solve
this issue.
20192158
FIGURE 6. CS Pin, Transient Path
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20192159
FIGURE 7. CS Pin, Transient Path with Protection
Adding a resistor in series with the CS pin causes the ob-
served output LED current to shift very slightly. The reason
for this is twofold: (1) the CS pin has about 20pF of inherent
capacitance inside it which causes a slight delay (20ns for a
1kΩ series resistor), and (2) the comparator that is watching
the voltage at the CS pin uses a pnp bipolar transistor at its
input. The base current of this pnp transistor is approximately
100nA which will cause a 0.1mV change in the 200mV thresh-
old. These are both very minor changes and are well under-
stood. The shift in current can either be neglected or taken
into consideration by changing the current sense resistance
slightly.
on the CS pin requires additional consideration. As shown in
Figure 8, adding a zener diode from the output to the CS pin
(with the series resistor) for output overvoltage protection will
now again allow the transient energy to be passed through
the CS pin’s ESD structure thereby damaging it.
Adding an additional series resistor to the CS pin as shown
in Figure 9 will result in the majority of the transient energy to
pass through the sense resistor thereby protecting the
LM340X device.
CS PIN PROTECTION WITH OVP
When designing output overvoltage protection into the switch-
ing converter circuit using a zener diode, transient protection
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20192160
FIGURE 8. CS Pin with OVP, Transient Path
20192161
FIGURE 9. CS Pin with OVP, Transient Path with Protection
VIN PIN PROTECTION
switching converter circuit, damage to the VIN pin can still
occur.
The VIN pin also has an ESD structure from the pin to GND
with a breakdown voltage of approximately 80V. Any transient
that exceeds this voltage may damage the device. Although
transient absorption is usually present at the front end of a
When VIN is hot swapped in, the current that rushes in to
charge CIN up to the VIN value also charges (energizes) the
circuit board trace inductance as shown in Figure 10. The ex-
cited trace inductance then resonates with the input capaci-
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18
tance (similar to an under-damped LC tank circuit) and
causes voltages at the VIN pin to rise well in excess of both
VIN and the voltage at the module input connector as clamped
by the input TVS. If the resonating voltage at the VIN pin ex-
ceeds the 80V breakdown voltage of the ESD structure, the
ESD structure will activate and then “snap-back” to a lower
voltage due to its inherent design. If this lower snap-back
voltage is less than the applied nominal VIN voltage, then sig-
nificant current will flow through the ESD structure resulting
in the IC being damaged.
An additional TVS or small zener diode should be placed as
close as possible to the VIN pins of each IC on the board, in
parallel with the input capacitor as shown in Figure 11. A mi-
nor amount of series resistance in the input line would also
help, but would lower overall conversion efficiency. For this
reason, NTC resistors are often used as inrush limiters in-
stead.
20192162
FIGURE 10. VIN Pin with Typical Input Protection
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20192163
FIGURE 11. VIN Pin with Additional Input Protection
GENERAL COMMENTS REGARDING OTHER PINS
A regulated DC voltage input of 24V ±10% will power a single
1W white LED at a forward current of 350 mA ±5%. The typical
forward voltage of a 1W InGaN LED is 3.5V, hence the esti-
mated average output voltage will be 3.7V. The objective of
this application is to place the complete current regulator and
LED in the compact space formerly occupied by an MR16
halogen light bulb. (The LED will be on a separate metal-core
PCB.) Switching frequency will be as fast as the 300 ns tON
limit allows, with the emphasis on space savings over effi-
ciency. Efficiency cannot be ignored, however, as the con-
fined space with little air-flow requires a maximum tempera-
ture rise of 40°C in each circuit component. A complete bill of
materials can be found in Table 1 at the end of this datasheet.
Any pin that goes “off-board” through a connector should have
series resistance of at least 1kΩ to 10kΩ in series with it to
protect it from ESD or other transients. These series resistors
limit the peak current that can flow (or cause a voltage drop)
during a transient event, thus protecting the pin and the de-
vice. Pins that are not used should not be left floating. They
should instead be tied to GND or to an appropriate voltage
through resistance.
Design Example 1: LM3402
The first example circuit will guide the user through compo-
nent selection for an architectural accent lighting application.
20192119
FIGURE 12. Schematic for Design Example 1
RON and tON
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20
ΔiL(MIN) = [(26.4 – 3.7) x 300 x 10-9] / 39.6 x 10-6
To select RON the expression relating tON to input voltage from
the Controlled On-time Overview section can be re-written as:
= 172 mAP-P
ΔiL(MAX) = [(26.4 – 3.7) x 300 x 10-9] / 26.4 x 10-6
= 258 mAP-P
Minimum on-time occurs at the maximum VIN, which is 24V x
110% = 26.4V. RON is therefore calculated as:
The third specification for an inductor is the peak current rat-
ing, normally given as the point at which the inductance drops
off by a given percentage due to saturation of the core. The
worst-case peak current occurs at maximum input voltage
and at minimum inductance, and can be determined with the
equation from the Design Considerations section:
RON = (300 x 10-9 x 26.4) / 1.34 x 10-10 = 59105 Ω
The closest 1% tolerance resistor is 59.0 kΩ. The switching
frequency of the circuit can then be found using the equation
relating RON to fSW
:
fSW = 3.7 / (59000 x 1.34 x 10-10) = 468 kHz
IL(PEAK) = 0.35 + 0.258 / 2 = 479 mA
USING AN OUTPUT CAPACITOR
The inductor will be the largest component used in this design.
Because the application does not require any PWM dimming,
an output capacitor can be used to greatly reduce the induc-
tance needed without worry of slowing the potential PWM
dimming frequency. The total solution size will be reduced by
using an output capacitor and small inductor as opposed to
one large inductor.
For this example the peak current rating of the inductor should
be greater than 479 mA. In the case of a short circuit across
the LED array, the LM3402 will continue to deliver rated cur-
rent through the short but will reduce the output voltage to
equal the CS pin voltage of 200 mV. Worst-case peak current
in this condition is equal to:
ΔiL(LED-SHORT) = [(26.4 – 0.2) x 300 x 10-9] / 26.4 x 10-6
= 298 mAP-P
OUTPUT INDUCTOR
Knowing that an output capacitor will be used, the inductor
can be selected for a larger current ripple. The desired max-
IL(PEAK) = 0.35 + 0.149 = 499 mA
imum value for ΔiL is ±30%, or 0.6 x 350 mA = 210 mAP-P
.
In the case of a short at the switch node, the output, or from
the CS pin to ground the short circuit current limit will engage
at a typical peak current of 735 mA. In order to prevent in-
ductor saturation during these short circuits the inductor’s
peak current rating must be above 735 mA. The device se-
lected is an off-the-shelf inductor rated 33 µH ±20% with a
DCR of 96 mΩ and a peak current rating of 0.82A. The phys-
ical dimensions of this inductor are 7.0 x 7.0 x 4.5 mm.
Minimum inductance is selected at the maximum input volt-
age. Re-arranging the equation for current ripple selection
yields the following:
LMIN = [(26.4 – 3.7) x 300 x 10-9] / (0.6 x 0.35) = 32.4 µH
RSNS
The current sensing resistor value can be determined by re-
arranging the expression for average LED current from the
LED Current Accuracy section:
The closest standard inductor value is 33 µH. Off-the-shelf
inductors rated at 33 µH are available from many magnetics
manufacturers.
Inductor datasheets should contain three specifications which
are used to select the inductor. The first of these is the aver-
age current rating, which for a buck regulator is equal to the
average load current, or IF. The average current rating is given
by a specified temperature rise in the inductor, normally 40°
C. For this example, the average current rating should be
greater than 350 mA to ensure that heat from the inductor
does not reduce the lifetime of the LED or cause the LM3402
to enter thermal shutdown.
RSNS = 0.74Ω, tSNS = 220 ns
Sub-1Ω resistors are available in both 1% and 5% tolerance.
A 1%, 0.75Ω resistor will give the best accuracy of the aver-
age LED current. To determine the resistor size the power
dissipation can be calculated as:
The second specification is the tolerance of the inductance
itself, typically ±10% to ±30% of the rated inductance. In this
example an inductor with a tolerance of ±20% will be used.
With this tolerance the typical, minimum, and maximum in-
ductor current ripples can be calculated:
PSNS = (IF)2 x RSNS
PSNS = 0.352 x 0.75 = 92 mW
ΔiL(TYP) = [(26.4 – 3.7) x 300 x 10-9] / 33 x 10-6
= 206 mAP-P
Standard 0805 size resistors are rated to 125 mW and will be
suitable for this application.
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To select the proper output capacitor the equation from Buck
Regulators with Output Capacitors is re-arranged to yield the
following:
Schottky diodes are available at forward current ratings of
0.5A, however the current rating often assumes a 25°C am-
bient temperature and does not take into account the appli-
cation restrictions on temperature rise. A diode rated for
higher current may be needed to keep the temperature rise
below 40°C.To determine the proper case size, the dissipa-
tion and temperature rise in D1 can be calculated as shown
in the Design Considerations section. VD for a small case size
such as SOD-123 in a 40V, 0.5A Schottky diode at 350 mA is
approximately 0.4V and the θJA is 206°C/W. Power dissipa-
tion and temperature rise can be calculated as:
The target tolerance for LED ripple current is ±5% or 10%P-
P = 35 mAP-P, and the LED datasheet gives a typical value for
rD of 1.0Ω at 350 mA. The required capacitor impedance to
reduce the worst-case inductor ripple current of 258 mAP-P is
therefore:
PD = 0.298 x 0.4 = 119 mW
TRISE = 0.119 x 206 = 24.5°C
ZC = [0.035 / (0.258 - 0.035] x 1.0 = 0.157Ω
According to these calculations the SOD-123 diode will meet
the requirements. Heating and dissipation are among the fac-
tors most difficult to predict in converter design. If possible, a
footprint should be used that is capable of accepting both
SOD-123 and a larger case size, such as SMA. A larger diode
with a higher forward current rating will generally have a lower
forward voltage, reducing dissipation, as well as having a
lower θJA, reducing temperature rise.
A ceramic capacitor will be used and the required capacitance
is selected based on the impedance at 468 kHz:
CO = 1/(2 x π x 0.157 x 4.68 x 105) = 2.18 µF
This calculation assumes that impedance due to the equiva-
lent series resistance (ESR) and equivalent series inductance
(ESL) of CO is negligible. The closest 10% tolerance capacitor
value is 2.2 µF. The capacitor used should be rated to 10V or
more and have an X7R dielectric. Several manufacturers pro-
duce ceramic capacitors with these specifications in the 0805
case size. A typical value for ESR of 1 mΩ can be read from
the curve of impedance vs. frequency in the product
datasheet.
CB and CF
The bootstrap capacitor CB should always be a 10 nF ceramic
capacitor with X7R dielectric. A 25V rating is appropriate for
all application circuits. The linear regulator filter capacitor CF
should always be a 100 nF ceramic capacitor, also with X7R
dielectric and a 25V rating.
INPUT CAPACITOR
EFFICIENCY
Following the calculations from the Input Capacitor section,
ΔvIN(MAX) will be 1%P-P = 240 mV. The minimum required ca-
pacitance is:
To estimate the electrical efficiency of this example the power
dissipation in each current carrying element can be calculated
and summed. This term should not be confused with the op-
tical efficacy of the circuit, which depends upon the LEDs
themselves.
CIN(MIN) = (0.35 x 300 x 10-9) / 0.24 = 438 nF
Total output power, PO, is calculated as:
In expectation that more capacitance will be needed to pre-
vent power supply interaction a 1.0 µF ceramic capacitor
rated to 50V with X7R dielectric in a 1206 case size will be
used. From the Design Considerations section, input rms cur-
rent is:
PO = IF x VO = 0.35 x 3.7 = 1.295W
Conduction loss, PC, in the internal MOSFET:
PC = (IF2 x RDSON) x D = (0.352 x 1.5) x 0.154 = 28 mW
IIN-RMS = 0.35 x Sqrt(0.154 x 0.846) = 126 mA
Gate charging and VCC loss, PG, in the gate drive and linear
regulator:
Ripple current ratings for 1206 size ceramic capacitors are
typically higher than 1A, more than enough for this design.
RECIRCULATING DIODE
PG = (IIN-OP + fSW x QG) x VIN
The first parameter for D1 which must be determined is the
reverse voltage rating. Schottky diodes are available at re-
verse ratings of 30V and 40V, often in the same package, with
the same forward current rating. To account for ringing a 40V
Schottky will be used.
PG = (600 x 10-6 + 468000 x 3 x 10-9) x 24 = 48 mW
Switching loss, PS, in the internal MOSFET:
PS = 0.5 x VIN x IF x (tR + tF) x fSW
The next parameters to be determined are the forward current
rating and case size. In this example the low duty cycle (D =
3.7 / 24 = 15%) requires the recirculating diode D1 to carry
the load current much longer than the internal power MOS-
FET of the LM3402. The estimated average diode current is:
PS = 0.5 x 24 x 0.35 x (40 x 10-9) x 468000 = 78 mW
AC rms current loss, PCIN, in the input capacitor:
PCIN = IIN(rms)2 x ESR = (0.126)2 x 0.006 = 0.1 mW (negligible)
ID = 0.35 x 0.85 = 298 mA
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The closest 1% tolerance resistor is 1.21 MΩ. The switching
frequency and on-time of the circuit can then be found using
DCR loss, PL, in the inductor
PL = IF2 x DCR = 0.352 x 0.096 = 11.8 mW
the equations relating RON and tON to fSW
:
fSW = 49.2 / (1210000 x 1.34 x 10-10) = 303 kHz
Recirculating diode loss, PD = 119 mW
tON = (1.34 x 10-10 x 1210000) / 60 = 2.7 µs
Current Sense Resistor Loss, PSNS = 92 mW
Electrical efficiency, η = PO / (PO + Sum of all loss terms) =
USING AN OUTPUT CAPACITOR
1.295 / (1.295 + 0.377) = 77%
This application is dominated by the need for fast PWM dim-
ming, requiring a circuit without any output capacitance.
DIE TEMPERATURE
TLM3402 = (PC + PG + PS) x θJA
TLM3402 = (0.028 + 0.05 + 0.078) x 200 = 31°C
OUTPUT INDUCTOR
In this example the ripple current through the LED array and
the inductor are equal. Inductance is selected to give the
smallest ripple current possible while still providing enough
ΔvSNS signal for the CS comparator to operate correctly. De-
signing to a desired ΔvSNS of 25 mV and assuming that the
average inductor current will equal the desired average LED
current of 350 mA yields the target current ripple in the in-
ductor and LEDs:
Design Example 2: LM3402HV
The second example application is an RGB backlight for a flat
screen monitor. A separate boost regulator provides a 60V
±5% DC input rail that feeds three LM3402HV current regu-
lators to drive one series array each of red, green, and blue
1W LEDs. The target for average LED current is 350 mA ±5%
in each string. The monitor will adjust the color temperature
dynamically, requiring fast PWM dimming of each string with
external, parallel MOSFETs. 1W green and blue InGaN LEDs
have a typical forward voltage of 3.5V, however red LEDs use
AlInGaP technology with a typical forward voltage of 2.9V. In
order to match color properly the design requires 14 green
LEDs, twice as many as needed for the red and blue LEDs.
This example will follow the design for the green LED array,
providing the necessary information to repeat the exercise for
the blue and red LED arrays. The circuit schematic for Design
Example 2 is the same as the Typical Application on the front
page. The bill of materials (green array only) can be found in
Table 2 at the end of this datasheet.
ΔiF = ΔiL = ΔvSNS / RSNS, RSNS = VSNS / IF
ΔiF = 0.025 / 0.57 = 43.8 mA
With the target ripple current determined the inductance can
be chosen:
LMIN = [(60 – 49.2) x 2.7 x 10-6] / (0.044) = 663 µH
OUTPUT VOLTAGE
Green Array: VO(G) = 14 x 3.5 + 0.2 = 49.2V
Blue Array: VO(B) = 7 x 3.5 + 0.2 = 24.7V
Red Array: VO(R) = 7 x 2.9 + 0.2 = 20.5V
The closest standard inductor value is 680 µH. As with the
previous example, the average current rating should be
greater than 350 mA. Separation between the LM3402HV
drivers and the LED arrays mean that heat from the inductor
will not threaten the lifetime of the LEDs, but an overheated
inductor could still cause the LM3402HV to enter thermal
shutdown.
RON and tON
A compromise in switching frequency is needed in this appli-
cation to balance the requirements of magnetics size and
efficiency. The high duty cycle translates into large conduc-
tion losses and high temperature rise in the IC. For best
response to a PWM dimming signal this circuit will not use an
output capacitor; hence a moderate switching frequency of
300 kHz will keep the inductance from becoming so large that
a custom-wound inductor is needed. This design will use only
surface mount components, and the selection of off-the-shelf
SMT inductors for switching regulators is poor at 1000 µH and
above. RON is selected from the equation for switching fre-
quency as follows:
The inductance itself of the standard part chosen is ±20%.
With this tolerance the typical, minimum, and maximum in-
ductor current ripples can be calculated:
ΔiF(TYP) = [(60 - 49.2) x 2.7 x 10-6] / 680 x 10-6
= 43 mAP-P
ΔiF(MIN) = [(60 - 49.2) x 2.7 x 10-6] / 816 x 10-6
= 36 mAP-P
ΔiF(MAX) = [(60 - 49.2) x 2.7 x 10-6] / 544 x 10-6
= 54 mAP-P
The peak LED/inductor current is then estimated:
RON = 49.2 / (1.34 x 10-10 x 3 x 105) = 1224 kΩ
IL(PEAK) = IL + [ΔiL(MAX)] / 2
23
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Selecting a 100V rated diode provides a large safety margin
for the ringing of the switch node and also makes cross-ref-
erencing of diodes from different vendors easier.
IL(PEAK) = 0.35 + 0.027 = 377 mA
In the case of a short circuit across the LED array, the
LM3402HV will continue to deliver rated current through the
short but will reduce the output voltage to equal the CS pin
voltage of 200 mV. Worst-case peak current in this condition
would be equal to:
The next parameters to be determined are the forward current
rating and case size. In this example the high duty cycle (D =
49.2 / 60 = 82%) places less thermals stress on D1 and more
on the internal power MOSFET of the LM3402. The estimated
average diode current is:
ΔiF(LED-SHORT) = [(63 – 0.2) x 2.7 x 10-6] / 544 x 10-6
= 314 mAP-P
ID = 0.361 x 0.18 = 65 mA
IF(PEAK) = 0.35 + 0.156 = 506 mA
A Schottky with a forward current rating of 0.5A would be ad-
equate, however at 100V the majority of diodes have a mini-
mum forward current rating of 1A. To determine the proper
case size, the dissipation and temperature rise in D1 can be
calculated as shown in the Design Considerations section.
VD for a small case size such as SOD-123F in a 100V, 1A
Schottky diode at 350 mA is approximately 0.65V and the
In the case of a short at the switch node, the output, or from
the CS pin to ground the short circuit current limit will engage
at a typical peak current of 735 mA. In order to prevent in-
ductor saturation during these fault conditions the inductor’s
peak current rating must be above 735 mA. A 680 µH off-the
shelf inductor rated to 1.2A (peak) and 0.72A (average) with
a DCR of 1.1Ω will be used for the green LED array.
θ
JA is 88°C/W. Power dissipation and temperature rise can be
calculated as:
RSNS
A preliminary value for RSNS was determined in selecting
ΔiL. This value should be re-evaluated based on the calcula-
tions for ΔiF:
PD = 0.065 x 0.65 = 42 mW
TRISE = 0.042 x 88 = 4°C
CB AND CF
The bootstrap capacitor CB should always be a 10 nF ceramic
capacitor with X7R dielectric. A 25V rating is appropriate for
all application circuits. The linear regulator filter capacitor CF
should always be a 100 nF ceramic capacitor, also with X7R
dielectric and a 25V rating.
Sub-1Ω resistors are available in both 1% and 5% tolerance.
A 1%, 0.56Ω device is the closest value, and a 0.125W, 0805
size device will handle the power dissipation of 69 mW. With
the resistance selected, the average value of LED current is
re-calculated to ensure that current is within the ±5% toler-
ance requirement. From the expression for LED current ac-
curacy:
EFFICIENCY
To estimate the electrical efficiency of this example the power
dissipation in each current carrying element can be calculated
and summed. Electrical efficiency, η, should not be confused
with the optical efficacy of the circuit, which depends upon the
LEDs themselves.
Total output power, PO, is calculated as:
IF = 0.19 / 0.56 + 0.043 / 2 = 361 mA, 3% above 350 mA
INPUT CAPACITOR
PO = IF x VO = 0.361 x 49.2 = 17.76W
Following the calculations from the Input Capacitor section,
ΔvIN(MAX) will be 1%P-P = 600 mV. The minimum required ca-
pacitance is:
Conduction loss, PC, in the internal MOSFET:
PC = (IF2 x RDSON) x D = (0.3612 x 1.5) x 0.82 = 160 mW
CIN(MIN) = (0.35 x 2.7 x 10-6) / 0.6 = 1.6 µF
Gate charging and VCC loss, PG, in the gate drive and linear
regulator:
In expectation that more capacitance will be needed to pre-
vent power supply interaction a 2.2 µF ceramic capacitor
rated to 100V with X7R dielectric in an 1812 case size will be
used. From the Design Considerations section, input rms cur-
rent is:
PG = (IIN-OP + fSW x QG) x VIN
PG = (600 x 10-6 + 3 x 105 x 3 x 10-9) x 60 = 90 mW
IIN-RMS = 0.35 x Sqrt(0.82 x 0.18) = 134 mA
Switching loss, PS, in the internal MOSFET:
Ripple current ratings for 1812 size ceramic capacitors are
typically higher than 2A, more than enough for this design.
PS = 0.5 x VIN x IF x (tR + tF) x fSW
PS = 0.5 x 60 x 0.361 x 40 x 10-9 x 3 x 105 = 130 mW
RECIRCULATING DIODE
AC rms current loss, PCIN, in the input capacitor:
The input voltage of 60V ±5% requires Schottky diodes with
a reverse voltage rating greater than 60V. Some manufactur-
ers provide Schottky diodes with ratings of 70, 80 or 90V;
however the next highest standard voltage rating is 100V.
PCIN = IIN(rms)2 x ESR = (0.134)2 x 0.006 = 0.1 mW (negligible)
www.national.com
24
DCR loss, PL, in the inductor
PL = IF2 x DCR = 0.352 x 1.1 = 135 mW
The following guidelines will help the user design a circuit with
maximum rejection of outside EMI and minimum generation
of unwanted EMI.
COMPACT LAYOUT
Parasitic inductance can be reduced by keeping the power
path components close together and keeping the area of the
loops that high currents travel small. Short, thick traces or
copper pours (shapes) are best. In particular, the switch node
(where L1, D1, and the SW pin connect) should be just large
enough to connect all three components without excessive
heating from the current it carries. The LM3402/02HV oper-
ates in two distinct cycles whose high current paths are shown
in Figure 6:
Recirculating diode loss, PD = 42 mW
Current Sense Resistor Loss, PSNS = 69 mW
Electrical efficiency, η = PO / (PO + Sum of all loss terms) =
17.76 / (17.76 + 0.62) = 96%
Temperature Rise in the LM3402HV IC is calculated as:
TLM3402 = (PC + PG + PS) x θJA = (0.16 + 0.084 + 0.13) x 200
= 74.8°C
Layout Considerations
The performance of any switching converter depends as
much upon the layout of the PCB as the component selection.
20192128
FIGURE 13. Buck Converter Current Loops
The dark grey, inner loop represents the high current path
during the MOSFET on-time. The light grey, outer loop rep-
resents the high current path during the off-time.
at the pad of the input capacitor to connect the component
side shapes to the ground plane. A second pulsating current
loop that is often ignored is the gate drive loop formed by the
SW and BOOT pins and capacitor CB. To minimize this loop
at the EMI it generates, keep CB close to the SW and BOOT
pins.
GROUND PLANE AND SHAPE ROUTING
The diagram of Figure 6 is also useful for analyzing the flow
of continuous current vs. the flow of pulsating currents. The
circuit paths with current flow during both the on-time and off-
time are considered to be continuous current, while those that
carry current during the on-time or off-time only are pulsating
currents. Preference in routing should be given to the pulsat-
ing current paths, as these are the portions of the circuit most
likely to emit EMI. The ground plane of a PCB is a conductor
and return path, and it is susceptible to noise injection just as
any other circuit path. The continuous current paths on the
ground net can be routed on the system ground plane with
less risk of injecting noise into other circuits. The path be-
tween the input source and the input capacitor and the path
between the recirculating diode and the LEDs/current sense
resistor are examples of continuous current paths. In contrast,
the path between the recirculating diode and the input capac-
itor carries a large pulsating current. This path should be
routed with a short, thick shape, preferably on the component
side of the PCB. Multiple vias in parallel should be used right
CURRENT SENSING
The CS pin is a high-impedance input, and the loop created
by RSNS, RZ (if used), the CS pin and ground should be made
as small as possible to maximize noise rejection. RSNS should
therefore be placed as close as possible to the CS and GND
pins of the IC.
REMOTE LED ARRAYS
In some applications the LED or LED array can be far away
(several inches or more) from the LM3402/02HV, or on a sep-
arate PCB connected by a wiring harness. When an output
capacitor is used and the LED array is large or separated from
the rest of the converter, the output capacitor should be
placed close to the LEDs to reduce the effects of parasitic
inductance on the AC impedance of the capacitor. The current
sense resistor should remain on the same PCB, close to the
LM3402/02HV.
25
www.national.com
TABLE 1. BOM for Design Example 1
ID
U1
Part Number
LM3402
Type
LED Driver
Inductor
Size
MSOP-8
7.0x7.0 x4.5mm
SOD-123
0805
Parameters
40V, 0.5A
Qty
1
Vendor
NSC
L1
SLF7045T-330MR82
CMHSH5-4
1
TDK
33µH, 0.82A, 96mΩ
40V, 0.5A
D1
Schottky Diode
Capacitor
Capacitor
Capacitor
Capacitor
Resistor
1
Central Semi
Vishay
Vishay
TDK
Cf
VJ0805Y104KXXAT
VJ0805Y103KXXAT
C3216X7R1H105M
C2012X7R1A225M
ERJ6BQFR75V
CRCW08055902F
100nF 10%
10nF 10%
1
Cb
0805
1
Cin
Co
1206
1µF 50V
1
0805
2.2 µF 10V
0.75Ω 1%
1
TDK
Rsns
Ron
0805
1
Panasonic
Vishay
Resistor
0805
1
59.0 kΩ 1%
TABLE 2. BOM for Design Example 2
ID
U1
Part Number
LM3402HV
Type
LED Driver
Inductor
Size
MSOP-8
18.5x15.2 x7.1mm
SOD-123F
0805
Parameters
75V, 0.5A
Qty
1
Vendor
NSC
L1
DO5022P-684
1
Coilcraft
Central Semi
Vishay
680µH, 1.2A, 1.1Ω
100V, 1A
D1
CMMSH1-100
Schottky Diode
Capacitor
Capacitor
Capacitor
Resistor
1
Cf
VJ0805Y104KXXAT
VJ0805Y103KXXAT
C4532X7R2A225M
ERJ6BQFR56V
CRCW08051214F
100nF 10%
10nF 10%
1
Cb
0805
1
Vishay
Cin
Rsns
Ron
1812
2.2µF 100V
0.56Ω 1%
1
TDK
0805
1
Panasonic
Vishay
Resistor
0805
1
1.21MΩ 1%
www.national.com
26
Physical Dimensions inches (millimeters) unless otherwise noted
8-Lead MSOP Package
NS Package Number MUA08A
8-Lead PSOP Package
NS Package Number MRA08B
27
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