LM25085SDE [TI]
LM25085, LM25085-Q1 42V Constant On-Time PFET Buck Switching Controller; LM25085 ， LM25085 -Q1 42V恒定导通时间PFET降压开关控制器
| 型号： | LM25085SDE |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | LM25085, LM25085-Q1 42V Constant On-Time PFET Buck Switching Controller |
| 文件： | 总30页 (文件大小：1508K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LM25085, LM25085-Q1
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SNVS593H –OCTOBER 2008–REVISED MARCH 2013
LM25085, LM25085-Q1 42V Constant On-Time PFET Buck Switching Controller
Check for Samples: LM25085, LM25085-Q1
1
FEATURES
PACKAGE
2
•
LM25085-Q1 is an Automotive Grade product
that is AEC-Q100 Grade 1 Qualified (-40°C to
125°C Operating Junction Temperature)
•
•
•
HVSSOP-8
VSSOP-8
WSON-8 (3mm x 3mm)
•
•
Wide 4.5V to 42V Input Voltage Range
Adjustable Current Limit Using RDS(ON) or a
Current Sense Resistor
DESCRIPTION
The LM25085 is a high efficiency PFET switching
regulator controller that can be used to quickly and
easily develop a small, efficient buck regulator for a
wide range of applications. This high voltage
controller contains a PFET gate driver and a high
voltage bias regulator which operates over a wide
4.5V to 42V input range. The constant on-time
regulation principle requires no loop compensation,
simplifies circuit implementation, and results in ultra-
fast load transient response. The operating frequency
remains nearly constant with line and load variations
due to the inverse relationship between the input
voltage and the on-time. The PFET architecture
allows 100% duty cycle operation for a low dropout
voltage. Either the RDS(ON) of the PFET or an external
sense resistor can be used to sense current for over-
current detection.
•
•
•
•
Programmable Switching Frequency to 1MHz
No Loop Compensation Required
Ultra-Fast Transient Response
Nearly Constant Operating Frequency with
Line and Load Variations
•
•
•
•
•
•
Adjustable Output Voltage from 1.25V
Precision ±2% Feedback Reference
Capable of 100% Duty Cycle Operation
Internal Soft-Start Timer
Integrated High Voltage Bias Regulator
Thermal Shutdown
Typical Application, Basic Step Down Controller
4.5V to 42V
C
VCC
LM25085
Input
VIN
V
IN
VCC
C
ADJ
C
IN
ADJ
GND
R
T
R
ADJ
L1
PGATE
Q1
RT
SHUTDOWN
V
OUT
ISEN
FB
C
OUT
GND
D1
Cff
R
FB2
GND
R
FB1
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
2
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2008–2013, Texas Instruments Incorporated
LM25085, LM25085-Q1
SNVS593H –OCTOBER 2008–REVISED MARCH 2013
www.ti.com
Connection Diagram
VIN
8
7
1
ADJ
RT
Exposed Pad on Bottom
Connect to Ground
2
VCC
VIN
1
8
7
6
5
ADJ
RT
FB
3
4
6
5
PGATE
ISEN
VCC
2
3
FB
PGATE
ISEN
GND
GND
4
Exposed Pad on Bottom
Connect to Ground
Figure 1. Top View
8-Lead HVSSOP-PowerPAD
Figure 3. Top View
8-Lead WSON
See Package Number DGN0008A
See Package Number NGQ0008A
1
8
7
6
5
VIN
ADJ
RT
VCC
2
3
FB
PGATE
ISEN
GND
4
Figure 2. Top View
8-Lead VSSOP
See Package Number DGK0008A
PIN DESCRIPTIONS
Pin
No.
Name
Description
Application Information
1
2
3
ADJ
RT
Current Limit Adjust
The current limit threshold is set by an external resistor from VIN to ADJ in
conjunction with the external sense resistor or the PFET’s RDS(ON)
.
On-time control and shutdown
An external resistor from VIN to RT sets the buck switch on-time and switching
frequency. Grounding this pin shuts down the controller.
FB
Voltage Feedback from the
regulated output
Input to the regulation and over-voltage comparators. The regulation level is 1.25V.
4
5
GND
ISEN
Circuit Ground
Ground reference for all internal circuitry
Current sense input for current
limit detection.
Connect to the PFET drain when using RDS(ON) current sense. Connect to the PFET
source and the sense resistor when using a current sense resistor.
6
7
PGATE Gate Driver Output
Connect to the gate of the external PFET.
VCC
Output of the gate driver bias
Output of the negative voltage regulator (relative to VIN) that biases the PFET gate
driver. A low ESR capacitor is required from VIN to VCC, located as close as
possible to the pins.
regulator
8
VIN
EP
Input supply voltage
Exposed Pad
The operating input range is from 4.5V to 42V. A low ESR bypass capacitor must be
located as close as possible to the VIN and GND pins.
Exposed pad on the underside of the package (HVSSOP-PowerPAD-8 and WSON
only). This pad is to be soldered to the PC board ground plane to aid in heat
dissipation.
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
2
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SNVS593H –OCTOBER 2008–REVISED MARCH 2013
(1)(2)
Absolute Maximum Ratings
VIN to GND
-0.3V to 45V
ISEN to GND
-0.3V to VIN + 0.3V
-0.3V to VIN + 0.3V
-0.3V to 7V
ADJ to GND
RT, FB to GND
VIN to VCC, VIN to PGATE
-0.3V to 10V
(3)
ESD Rating
Human Body Model
2kV
Storage Temperature Range
-65°C to +150°C
(1) Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which
operation of the device is intended to be functional. For specifications and test conditions, see the Electrical Characteristics.
(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
(3) The human body model is a 100pF capacitor discharged through a 1.5kΩ resistor into each pin.
(1)
Operating Ratings
VIN Voltage
4.5V to 42V
Junction Temperature
−40°C to + 125°C
(1) Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which
operation of the device is intended to be functional. For specifications and test conditions, see the Electrical Characteristics.
Electrical Characteristics
Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature (TJ) range of -40°C
to +125°C. Minimum and Maximum limits are specified through test, design, or statistical correlation. Typical values represent
the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated the
following conditions apply: VIN = 24V, RT = 100kΩ.
Symbol
VIN Pin
Parameter
Conditions
Min
Typ
Max
Units
(1)
IIN
Operating Current
Shutdown Current
Non-Switching, FB = 1.4V
1.25
175
1.75
300
mA
µA
(1)
IQ
RT = 0V
VCC Regulator(2)
VCC(reg)
VIN - VCC
Vin = 9V, FB = 1.4V, ICC = 0mA
Vin = 9V, FB = 1.4V, ICC = 20mA
Vin = 42V, FB = 1.4V, ICC = 0mA
VCC Increasing
6.9
7.7
7.7
7.7
3.8
8.5
V
V
V
V
UVLOVcc
VCC Under-Voltage Lock-Out
Threshold
UVLOVcc Hysteresis
VCC Current Limit
VCC Decreasing
FB = 1.4V
260
40
mV
mA
VCC(CL)
PGATE Pin
VPGATE(HI)
20
PGATE High Voltage
PGATE Low Voltage
PGATE Pin = Open
PGATE Pin = Open
VIN -0.1
VIN -0.1
VIN
VCC
VIN
V
V
V
V
A
A
Ω
Ω
VPGATE(LO)
VPGATE(HI)4.5
VPGATE(LO)4.5
IPGATE
VCC+0.1
VCC+0.1
PGATE High Voltage at Vin = 4.5V PGATE Pin = Open
PGATE Low Voltage at Vin = 4.5V
Driver Output Source Current
Driver Output Sink Current
Driver Output Resistance
PGATE Pin = Open
VCC
1.75
1.5
VIN = 12V, PGATE = VIN - 3.5V
VIN = 12V, PGATE = VIN - 3.5V
Source current = 500mA
Sink current = 500mA
RPGATE
2.3
2.3
Current Limit Detection
IADJ
ADJUST Pin Current Source
Current Limit Comparator Offset
VADJ = 22.5V
32
-9
40
0
48
9
µA
VCL OFFSET
VADJ = 22.5V, VADJ - VISEN
mV
(1) Operating current and shutdown current do not include the current in the RT resistor.
(2) VCC provides self bias for the internal gate drive.
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Electrical Characteristics (continued)
Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature (TJ) range of -40°C
to +125°C. Minimum and Maximum limits are specified through test, design, or statistical correlation. Typical values represent
the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated the
following conditions apply: VIN = 24V, RT = 100kΩ.
Symbol
RT Pin
Parameter
Conditions
Min
Typ
Max
Units
RTSD
Shutdown Threshold
RT Pin Voltage Rising
0.73
50
V
RTHYS
Shutdown Threshold Hysteresis
mV
On-Time
tON – 1
tON – 2
tON - 3
tON - 4
On-Time
VIN = 4.5V, RT = 100kΩ
VIN = 24V, RT = 100kΩ
VIN = 42V, RT = 100kΩ
3.5
560
329
55
5
7.15
870
500
235
µs
ns
ns
ns
720
415
140
Minimum On-Time in Current Limit VIN = 24V, 25mV Overdrive at ISEN
(3)
Off-Time
(3)
tOFF(CL1)
tOFF(CL2)
tOFF(CL3)
tOFF(CL4)
Off-Time (Current Limit)
VIN = 12V, VFB = 0V
VIN = 12V, VFB = 1V
VIN = 24V, VFB = 0V
VIN = 24V, VFB = 1V
5.35
1.42
8.9
7.9
1.9
13
10.84
3.03
17.7
4.68
µs
µs
µs
µs
2.22
3.2
Regulation and Over-Voltage Comparators (FB Pin)
VREF
FB Regulation Threshold
FB Over-Voltage Threshold
FB Bias Current
1.225
1.4
1.25
350
10
1.275
V
VOV
Measured With Respect to VREF
mV
nA
IFB
Soft-Start Function
tSS
Soft-Start Time
2.5
4.3
ms
Thermal Shutdown
TSD
Junction Shutdown Temperature
Junction Shutdown Hysteresis
Junction Temperature Rising
170
20
°C
°C
THYS
Thermal Resistance(4)
θJA Junction to Ambient, 0 LFPM Air
VSSOP-8 package
126
46
°C/W
°C/W
(5)
Flow
HVSSOP-PowerPAD-8 package
WSON-8 package
54
θJC
Junction to Case, 0 LFPM Air Flow VSSOP-8 package
29
(5)
HVSSOP-PowerPAD-8 package
WSON-8 package
5.5
9.1
(3) The tolerance of the minimum on-time (tON-4) and the current limit off-times (tOFF(CL1) through (tOFF(CL4)) track each other over process
and temperature variations. A device which has an on-time at the high end of the range will have an off-time that is at the high end of its
range.
(4) For detailed information on soldering plastic VSSOP and WSON packages visit www.ti.com/packaging.
(5) Tested on a 4 layer JEDEC board. Four vias provided under the exposed pad. See JEDEC standards JESD51-5 and JESD51-7.
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Typical Performance Characteristics
Unless otherwise specified the following conditions apply: TJ = 25°C, VIN = 24V.
Efficiency (Circuit of Figure 28)
Input Operating Current vs. VIN
Figure 4.
Figure 5.
Shutdown Current vs. VIN
VCC vs. VIN
Figure 6.
Figure 7.
VCC vs. ICC
On-Time vs. RT and VIN
Figure 8.
Figure 9.
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Typical Performance Characteristics (continued)
Unless otherwise specified the following conditions apply: TJ = 25°C, VIN = 24V.
Off-Time vs. VIN and VFB
Voltage at the RT Pin
Figure 10.
Figure 11.
ADJ Pin Current vs. VIN
Input Operating Current vs. Temperature
Figure 12.
Figure 13.
Shutdown Current vs. Temperature
VCC vs. Temperature
Figure 14.
Figure 15.
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Typical Performance Characteristics (continued)
Unless otherwise specified the following conditions apply: TJ = 25°C, VIN = 24V.
On-Time vs. Temperature
Minimum On-Time vs. Temperature
Figure 16.
Figure 17.
Off-Time vs. Temperature
Current Limit Comparator Offset vs. Temperature
Figure 18.
Figure 19.
ADJ Pin Current vs. Temperature
PGATE Driver Output Resistance vs. Temperature
Figure 20.
Figure 21.
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Typical Performance Characteristics (continued)
Unless otherwise specified the following conditions apply: TJ = 25°C, VIN = 24V.
Feedback Reference Voltage vs. Temperature
Soft-Start Time vs. Temperature
Figure 22.
Figure 23.
RT Pin Shutdown Threshold vs. Temperature
Figure 24.
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Block Diagram
4.5V to 42V
Input
Negative Bias
Regulator
+
V
LM25085
IN
C
VCC
VCC
V
IN
C
IN
C
ADJ
+
Thermal
Shutdown
7.7V
GND
C
-
BYP
V
IN
R
-
+
T
R
ADJ
R
SEN
0.73V
VCC
UVLO
Gate
Driver
RT
ON Time
One-Shot
PGATE
Q1
SHUTDOWN
VCC
1.25V
Soft-Start
L1
C1
V
OUT
Gate Driver
Control Logic
R3
C2
ADJ
C
OUT
40 mA
+
-
D1
Q S
R
+
-
R
R
FB2
ISEN
REGULATION
COMPARATOR
1.6V
CURRENT
LIMIT
COMPARATOR
FB1
OFF Time
One-Shot
GND
-
+
OVER-VOLTAGE
COMPARATOR
V
IN
FB
Sense resistor method shown for current limit detection.
Minimum output ripple configuration shown.
FUNCTIONAL DESCRIPTION
OVERVIEW
The LM25085 is a PFET buck (step-down) DC-DC controller using the constant on-time (COT) control principle.
The input operating voltage range of the LM25085 is 4.5V to 42V. The use of a PFET in a buck regulator greatly
simplifies the gate drive requirements and allows for 100% duty cycle operation to extend the regulation range
when operating at low input voltage. However, PFET transistors typically have higher on-resistance and gate
charge when compared to similarly rated NFET transistors. Consideration of available PFETs, input voltage
range, gate drive capability of the LM25085, and thermal resistances indicate an upper limit of 10A for the load
current for LM25085 applications. Constant on-time control is implemented using an on-time one-shot that is
triggered by the feedback signal. During the off-time, when the PFET (Q1) is off, the load current is supplied by
the inductor and the output capacitor. As the output voltage falls, the voltage at the feedback comparator input
(FB) falls below the regulation threshold. When this occurs Q1 is turned on for the one-shot period which is
determined by the input voltage (VIN) and the RT resistor. During the on-time the increasing inductor current
increases the voltage at FB above the feedback comparator threshold. For a buck regulator the basic relationship
between the on-time, off-time, input voltage and output voltage is:
VOUT
VIN
tON
=
= tON x FS
Duty Cycle =
tON + tOFF
where
•
Fs is the switching frequency
(1)
Equation 1 is valid only in continuous conduction mode (inductor current does not reach zero). Since the
LM25085 controls the on-time inversely proportional to VIN, the switching frequency remains relatively constant
as VIN is varied. If the input voltage falls to a level that is equal to or less than the regulated output voltage Q1 is
held on continuously (100% duty cycle) and VOUT is approximately equal to VIN.
The COT control scheme, with the feedback signal applied to a comparator rather than an error amplifier,
requires no loop compensation, resulting in very fast load transient response.
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The LM25085 is available in both an 8 pin HVSSOP-PowerPAD package and an 8 pin WSON package with an
exposed pad to aid in heat dissipation. An 8 pin VSSOP package without an exposed pad is also available.
REGULATION CONTROL CIRCUIT
The LM25085 buck DC-DC controller employs a control scheme based on a comparator and a one-shot on-
timer, with the output voltage feedback compared to an internal reference voltage (1.25V). When the FB pin
voltage falls below the feedback reference, Q1 is switched on for a time period determined by the input voltage
and a programming resistor (RT). Following the on-time Q1 remains off until the FB voltage falls below the
reference. Q1 is then switched on for another on-time period. The output voltage is set by the feedback resistors
(RFB1, RFB2 in Block Diagram). The regulated output voltage is calculated as follows:
VOUT = 1.25V x (RFB2+ RFB1)/ RFB1
(2)
The feedback voltage supplied to the FB pin is applied to a comparator rather than a linear amplifier. For proper
operation sufficient ripple amplitude is necessary at the FB pin to switch the comparator at regular intervals with
minimum delay and noise susceptibility. This ripple is normally obtained from the output voltage ripple attenuated
through the feedback resistors. The output voltage ripple is a result of the inductor’s ripple current passing
through the output capacitor’s ESR, or through a resistor in series with the output capacitor. Multiple methods are
available to ensure sufficient ripple is supplied to the FB pin, and three different configurations are discussed in
Applications Information.
When in regulation, the LM25085 operates in continuous conduction mode at medium to heavy load currents and
discontinuous conduction mode at light load currents. In continuous conduction mode the inductor’s current is
always greater than zero, and the operating frequency remains relatively constant with load and line variations.
The minimum load current for continuous conduction mode is one-half the inductor’s ripple current amplitude. In
discontinuous conduction mode, where the inductor’s current reaches zero during the off-time, the operating
frequency is lower than in continuous conduction mode and varies with load current. Conversion efficiency is
maintained at light loads since the switching losses are reduced with the reduction in load and frequency.
If the voltage at the FB pin exceeds 1.6V due to a transient overshoot or excessive ripple at VOUT the internal
over-voltage comparator immediately switches off Q1. The next on-time period starts when the voltage at FB falls
below the feedback reference voltage.
ON-TIME TIMER
The on-time of the PFET gate drive output (PGATE pin) is determined by the resistor (RT) and the input voltage
(VIN), and is calculated from:
1.45 x 10-7 x (RT + 1.4)
tON
=
+ 50 ns
(VIN - 1.56V + RT/3167)
where
•
RT is in kΩ
(3)
The minimum on-time, which occurs at maximum VIN, should not be set less than 150ns (see CURRENT
LIMITING). The buck regulator effective on-time, measured at the SW node (junction of Q1, L1, and D1) is
typically longer than that calculated in Equation 3 due to the asymmetric delay of the PFET. The on-time
difference caused by the PFET switching delay can be estimated as the difference of the turn-off and turn-on
delays listed in the PFET data sheet. Measuring the difference between the on-time at the PGATE pin versus the
SW node in the actual application circuit is also recommended.
In continuous conduction mode, the inverse relationship of tON with VIN results in a nearly constant switching
frequency as VIN is varied. The operating frequency can be calculated from:
VOUT x (VIN - 1.56V + RT/3167)
FS =
VIN x [(1.45 x 10-7 x (RT + 1.4)) + (tD x (VIN - 1.56V + RT/3167))]
where
•
•
RT is in kΩ
tD is equal to 50ns plus the PFET’s delay difference
(4)
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To set a specific continuous conduction mode switching frequency (FS), the RT resistor is determined from the
following:
where
•
RT is in kΩ
(5)
A simplified version of Equation 5 at VIN = 12V, and tD = 100ns, is:
VOUT x 6 x 106
- 8.6
RT =
FS
For VIN = 42V and tD = 100ns, the simplified equation is:
SHUTDOWN
The LM25085 can be shutdown by grounding the RT pin (see Figure 25). In this mode the PFET is held off, and
the VCC regulator is disabled. The internal operating current is reduced to the value shown in Figure 6. The
shutdown threshold at the RT pin is ≊0.73V, with ≊50mV of hysteresis. Releasing the pin enables normal
operation. The RT pin must not be forced high during normal operation.
VIN
Input
Voltage
R
LM25085
T
RT
STOP
RUN
Figure 25. Shutdown Implementation
CURRENT LIMITING
The LM25085 current limiting operates by sensing the voltage across either the RDS(ON) of Q1, or a sense
resistor, during the on-time and comparing it to the voltage across the resistor RADJ (see Figure 26). The current
limit function is much more accurate and stable over temperature when a sense resistor is used. The RDS(ON) of a
MOSFET has a wide process variation and a large temperature coefficient.
If the voltage across RDS(ON) of Q1, or the sense resistor, is greater than the voltage across RADJ, the current limit
comparator switches to turn off Q1. Current sensing is disabled for a blanking time of ≊100ns at the beginning of
the on-time to prevent false triggering of the current limit comparator due to leading edge current spikes.
Because of the blanking time and the turn-on and turn-off delays created by the PFET, the on-time at the PGATE
pin should not be set less than 150ns. An on-time shorter than that may prevent the current limit detection circuit
from properly detecting an over-current condition. The duration of the subsequent forced off-time is a function of
the input voltage and the voltage at the FB pin, as shown in Figure 10. The longer-than-normal forced off-time
allows the inductor current to decrease to a low level before the next on-time. This cycle-by-cycle monitoring,
followed by a forced off-time, provides effective protection from output load faults over a wide range of operating
conditions.
The voltage across the RADJ resistor is set by an internal 40µA current sink at the ADJ pin. When using Q1’s
RDS(ON) for sensing, the current at which the current limit comparator switches is calculated from:
ICL = 40µA x RADJ/RDS(ON)
(6)
When using a sense resistor (RSEN) the threshold of the current limit comparator is calculated from:
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ICL = 40µA x RADJ/RSEN
(7)
When using Equation 6 or Equation 7, the tolerances for the ADJ pin current sink and the offset of the current
limit comparator should be included to ensure the resulting minimum current limit is not less than the required
maximum switch current. Simultaneously increasing the values of RADJ and RSEN decreases the effects of the
current limit comparator offset, but at the expense of higher power dissipation. When using a sense resistor, the
RSEN resistor value should be chosen within the practical limitations of power dissipation and physical size. For
example, for a 10A current limit, setting RSEN = 0.005Ω results in a power dissipation as high as 0.5W. Current
sense connections to the RSEN resistor, or to Q1, must be Kelvin connections to ensure accuracy.
The CADJ capacitor filters noise from the ADJ pin, and helps prevent unintended switching of the current limit
comparator due to input voltage transients. The recommended value for CADJ is 1000pF.
CURRENT LIMIT OFF-TIME
When the current through Q1 exceeds the current limit threshold, the LM25085 forces an off-time longer than the
normal off-time defined by Equation 1. See Figure 10 or calculate the current limit off-time from the following
equation:
where
•
•
VIN is the input voltage
VFB is the voltage at the FB pin at the time current limit was detected
(8)
This feature is necessary to allow the inductor current to decrease sufficiently to offset the current increase which
occurred during the on-time. During the on-time, the inductor current increases an amount equal to:
(VIN - VOUT) x tON
DI =
L
(9)
During the off-time the inductor current decreases due to the reverse voltage applied across the inductor by the
output voltage, the freewheeling diode’s forward voltage (VFD), and the voltage drop due to the inductor’s series
resistance (VESR). The current decrease is equal to:
(VOUT + VFD + VESR) x tOFF
DI =
L
(10)
The on-time in Equation 9 is shorter than the normal on-time since the PFET is shut off when the current limit
threshold is crossed. If the off-time is not long enough, such that the current decrease (Equation 10) is less than
the current increase (Equation 9), the current levels are higher at the start of the next on-time. This results in a
further decrease in on-time, since the current limit threshold is crossed sooner. A balance is reached when the
current changes in Equation 9 and Equation 10 are equal. The worst case situation is that of a direct short circuit
at the output terminals, where VOUT = 0V, as that results in the largest current increase during the on-time, and
the smallest decrease during the off-time. The sum of the diode’s forward voltage and the inductor’s ESR voltage
must be sufficient to ensure current runaway does not occur. Using Equation 9 and Equation 10, this requirement
can be stated as:
VIN x tON
VFD + VESR
í
tOFF
(11)
For tON in Equation 11 use the minimum on-time at the SW node. To determine this time period add the
“Minimum on-time in current limit” specified in Electrical Characteristics (tON-4) to the difference of the turn-off
and turn-on delays of the PFET. For tOFF use the value in Figure 10 or use Equation 8, where VFB is equal to
zero volts. When using the minimum or maximum limits of those specifications to determine worst case
situations, the tolerance of the minimum on-time (tON-4) and the current limit off-times (tOFF(CL1) through tOFF(CL4)
)
track each other over the process and temperature variations. A device which has an on-time at the high end of
the range will have an off-time that is at the high end of its range.
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V
IN
V
IN
LM25085
LM25085
R
ADJ
R
ADJ
ADJ
ADJ
40 mA
40 mA
CURRENT LIMIT
COMPARATOR
CURRENT LIMIT
COMPARATOR
R
SEN
C
ADJ
C
ADJ
+
-
+
-
ISEN
ISEN
VIN
VIN
GATE
GATE
DRIVER
DRIVER
L1
Q1
Q1
L1
PGATE
PGATE
VCC
VCC
D1
D1
USING Q1 R
DS(ON)
USING SENSE RESISTOR R
SEN
Figure 26. Current Limit Sensing
VCC REGULATOR
The VCC regulator provides a regulated voltage between the VIN and the VCC pins to provide the bias and gate
current for the PFET gate driver. The 0.47µF capacitor at the VCC pin must be a low ESR capacitor, preferably
ceramic as it provides the high surge current for the PFET’s gate at each turn-on. The capacitor must be located
as close as possible to the VIN and VCC pins to minimize inductance in the PC board traces.
Referring to Figure 7, the voltage across the VCC regulator (VIN – VCC) is equal to VIN until VIN reaches
approximately 8.5V. At higher values of VIN, the voltage at the VCC pin is regulated at approximately 7.7V below
VIN. If VIN drops below about 8V due to voltage transients, the VCC pin can be pulled down below GND. To
prevent the negative VCC voltage from disturbing the internal circuit and causing abnormal operation, a Schottky
diode is recommended between VCC pin and GND pin. The VCC regulator has a maximum current capability of
at least 20mA. The regulator is disabled when the LM25085 is shutdown using the RT pin, or when the thermal
shutdown is activated.
PGATE DRIVER OUTPUT
The PGATE pin output swings between VIN (Q1 off) and the VCC pin voltage (Q1 on). The rise and fall times
depend on the PFET gate capacitance and the source and sink currents provided by the internal gate driver. See
Electrical Characteristics for the current capability of the driver.
P-CHANNEL MOSFET SELECTION
The PFET must be rated for the maximum input voltage, with some margin above that to allow for transients and
ringing which can occur on the supply line and the switching node. The gate-to-source voltage (VGS) normally
provided to the PFET is 7.7V for VIN greater than 8.5V. However, if the circuit is to be operated at lower values
of VIN, the selected PFET must be able to fully turn-on with a VGS voltage equal to VIN. The minimum input
operating voltage for the LM25085 is 4.5V.
Similar to NFETs, the case or exposed thermal pad for a PFET is electrically connected to the drain terminal.
When designing a PFET buck regulator the drain terminal is connected to the switching node. This situation
requires a trade-off between thermal and EMI performance since increasing the PC board area of the switching
node to aid the PFET power dissipation also increases radiated noise, possibly disrupting the circuit operation.
Typically the switching node area is kept to a reasonable minimum and the PFET peak current is derated to stay
within the recommended temperature rating of the PFET. The RDS(ON) of the PFET determines a portion of the
power dissipation in the PFET. However, PFETs with very low RDS(ON) usually have large values of gate charge.
A PFET with a higher gate charge has a corresponding slower switching speed, leading to higher switching
losses and affecting the PFET power dissipation.
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If the PFET RDS(ON) is used for current limit detection, note that it typically has a positive temperature coefficient.
At 100°C the RDS(ON) may be as much as 50% higher than the value at 25°C which could result in incorrect
current limiting if not accounted for when determining the value of the RADJ resistor. The PFET Total Gate
Charge determines most of the power dissipation in the LM25085 due to the repetitive charge and discharge of
the PFET’s gate capacitance by the gate driver (powered from the VCC regulator). The LM25085’s internal
power dissipation can be calculated from the following:
PDISS = VIN x ((QG x FS) + IIN)
where
•
•
•
QG is the PFET Total Gate Charge obtained from its datasheet
FS is the switching frequency
IIN is the LM25085's operating current obtained from Figure 5
(12)
Using the Thermal Resistance specifications in Electrical Characteristics, the approximate junction temperature
can be determined. If the calculated junction temperature is near the maximum operating temperature of 125°C,
either the switching frequency must be reduced, or a PFET with a smaller Total Gate Charge must be used.
SOFT-START
The internal soft-start feature of the LM25085 allows the regulator to gradually reach a steady state operating
point at power up, thereby reducing startup stresses and current surges. Upon turn-on, when VCC reaches its
under-voltage lockout threshold, the internal soft-start circuit ramps the feedback reference voltage from 0V to
1.25V, causing VOUT to ramp up in a proportional manner. The soft-start ramp time is typically 2.5ms.
In addition to controlling the initial power up cycle, the soft-start circuit also activates when the LM25085 is
enabled by releasing the RT pin, and when the circuit is shutdown and restarted by the internal Thermal
Shutdown circuit.
If the voltage at FB is below the regulation threshold value due to an over-current condition or a short circuit at
VOUT, the internal reference voltage provided by the soft-start circuit to the regulation comparator is reduced
along with FB. When the over-current or short circuit condition is removed, VOUT returns to the regulated value at
a rate determined by the soft-start ramp. This feature helps prevent the output voltage from overshooting
following an overload event.
THERMAL SHUTDOWN
The LM25085 should be operated such that the junction temperature does not exceed 125°C. If the junction
temperature increases above that, an internal Thermal Shutdown circuit activates at 170°C (typical) to disable
the VCC regulator and the gate driver, and discharge the soft-start capacitor. This feature helps prevent
catastrophic failures from accidental device overheating. When the junction temperature falls below 150°C
(typical hysteresis = 20°C), the gate driver is enabled, the soft-start circuit is released, and normal operation
resumes.
Applications Information
EXTERNAL COMPONENTS
The procedure for calculating the external components is illustrated with the following design example. Referring
to Block Diagram, the circuit is to be configured for the following specifications:
•
•
•
•
•
•
•
•
VOUT = 5V
VIN = 7V to 42V, 12V Nominal
Maximum load current (IOUT(max)) = 5A
Minimum load current (IOUT(min)) = 600mA (for continuous conduction mode)
Switching Frequency (FSW) = 300kHz
Maximum allowable output ripple (VOS) = 5mVp-p
Selected PFET: Vishay Si7465
RFB1 and RFB2: These resistors set the output voltage. The ratio of these resistors is calculated from:
RFB2/RFB1 = (VOUT/1.25V) - 1
For this example, RFB2 / RFB1 = 3. Typically, RFB1 and RFB2 should be chosen from standard value resistors in
the range of 1kΩ to 20kΩ which satisfy the above ratio. For this example, RFB2 = 10kΩ, and RFB1 = 3.4kΩ.
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•
RT, PFET: Before selecting the RT resistor, the PFET must be selected as its turn-on and turn-off delays
affect the calculated value of RT. For the Vishay Si7465 PFET, the difference of its typical turn-off and turn-on
delays is 57ns. Using Equation 5 at nominal input voltage, RT calculates to be:
(50 ns + 57 ns) x (12 - 1.56V)
1.45 x 10-7
5 x (12 - 1.56V)
1.45 x 10-7 x 12 x 300 kHz
- 1.4 = 90.9
-
RT =
A standard value 90.9kΩ resistor is selected. Using Equation 3 the minimum on-time at the PGATE pin, which
occurs at maximum input voltage (42V), is calculated to be 381ns. This minimum one-shot period is
sufficiently longer than the minimum recommended value of 150ns. The minimum on-time at the SW node is
longer due to the delay added by the PFET (57ns). Therefore the minimum SW node on-time is 438ns at
42V. At the SW node the maximum on-time is calculated to be 2.55µs at 7V.
•
L1: The main parameter controlled by the inductor value is the current ripple amplitude (IOR). See Figure 27.
The minimum load current for continuous conduction mode is used to determine the maximum allowable
ripple such that the inductor current’s lower peak does not fall below 0mA. Continuous conduction mode
operation at minimum load current is not a requirement of the LM25085, but serves as a guideline for
selecting L1. For this example, the maximum ripple current is:
IOR(max) = 2 x IOUT(min) = 1.2 Amp
(13)
If an application’s minimum load current is zero, a good initial estimate for the maximum ripple current
(IOR(max)) is 20% of the maximum load current. The ripple calculated in Equation 13 is then used in the
following equation to calculate L1:
tON(min) x (VIN(max) - VOUT
)
L1 =
= 13.5 mH
IOR(max)
(14)
A standard value 15µH inductor is selected. Using this inductance value, the maximum ripple current
amplitude, which occurs at maximum input voltage, calculates to 1.08Ap-p. The peak current (IPK) at
maximum load current is 5.54A. However, the current rating of the selected inductor must be based on the
maximum current limit value calculated below.
I
PK
I
OUT
I
OR
1/F
S
Figure 27. Inductor Current Waveform
•
RSEN, RADJ: To achieve good current limit accuracy and avoid over designing the power stage components,
the sense resistor method is used for current limiting in this example. A standard value 10mΩ resistor is
selected for RSEN, resulting in a 50mV drop at maximum load current, and a maximum 0.25W power
dissipation in the resistor. Since the LM25085 uses peak current detection, the minimum value for the current
limit threshold must be equal to the maximum load current (5A) plus half the maximum ripple amplitude
calculated above:
ICL(min) = 5A + 1.08A/2 = 5.54A
At this current level the voltage across RSEN is 55.4mV. Adding the current limit comparator offset of 9mV
(max) increases the required current limit threshold to 6.44A. Using Equation 7 with the minimum value for
the ADJ pin current (32µA), the required RADJ resistor calculates to:
6.44A x 0.01W
= 2.01 kW
RADJ
=
32 mA
A standard value 2.1kΩ resistor is selected. The nominal current limit threshold calculates to:
(2.1 kW x 40 mA)
= 8.4A
ICL(nom)
=
0.01W
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Using the tolerances for the ADJ pin current and the current limit comparator offset, the maximum current limit
threshold calculates to:
(2.1 kW x 48 mA) + 9 mV
= 11A
ICL(max)
=
0.01W
The minimum current limit thresholds calculate to:
(2.1 kW x 32 mA) - 9 mV
= 5.82A
ICL(min)
=
0.01W
The load current in each case is equal to the current limit threshold minus half the current ripple amplitude.
The recommended value of 1000pF for CADJ is used in this example.
•
COUT: Since the maximum allowed output ripple voltage is very low in this example (5mVp-p), the minimum
ripple configuration (R3, C1, and C2 in the Block Diagram) must be used. The resulting ripple at VOUT is then
due to the inductor’s ripple current passing through COUT. This capacitor’s value can be selected based on the
maximum allowable ripple voltage at VOUT, or based on transient response requirements. The following
calculation, based on ripple voltage, provides a first order result for the value of COUT
:
IOR(max)
COUT
=
8 x FS x VRIPPLE
where IOR(max) is the maximum ripple current calculated above, and VRIPPLE is the allowable ripple at VOUT
.
1.08A
8 x 300 kHz x 0.005V
COUT
=
= 90 mF
•
R3, C1, C2: The minimum ripple configuration uses these three components to generate the ripple voltage
required at the FB pin since there is insufficient ripple at VOUT. A minimum of 25mVp-p must be applied to the
FB pin to obtain stable constant frequency operation. R3 and C1 are selected to generate a sawtooth
waveform at their junction, and that waveform is AC coupled to the FB pin via C2. The values of the three
components are determined using the following procedure:
A 100µF capacitor is selected. Typically the ripple amplitude will be higher than the calculations indicate due
to the capacitor’s ESR.
Calculate VA = VOUT - (VSW x (1 – (VOUT/VIN(min))))
where VSW is the absolute value of the voltage at the SW node during the off-time, typically 0.5V to 1V
depending on the diode D1. Using a typical value of 0.65V, VA calculates to 4.81V. VA is the nominal DC
voltage at the R3/C1 junction, and is used in the next equation:
(VIN(min) - VA) x tON
R3 x C1 =
DV
where tON is the maximum on-time (at minimum input voltage, and ΔV is the desired ripple amplitude at the
R3/C1 junction, typically 25mVp-p.
(7V - 4.81V) x 2.55 ms
= 2.23 x 10-4
R3 x C1 =
0.025V
R3 and C1 are then selected from standard value components to produce the product calculated above.
Typical values for C1 are 3000pF to 10,000pF, and R3 is typically from 10kΩ to 300kΩ. C2 is then chosen
large compared to C1, typically 0.1µF. For this example, 3300pF is chosen for C1, requiring R3 to be 67.7kΩ.
A standard value 66.5kΩ resistor is selected.
•
CIN, CBYP: These capacitors limit the voltage ripple at VIN by supplying most of the switch current during the
on-time. At maximum load current, when Q1 is switched on, the current through Q1 suddenly increases to the
lower peak of the inductor’s ripple current, then ramps up to the upper peak, and then drops to zero at turn-
off. The average current during the on-time is the load current. For a worst case calculation, these capacitors
must supply this average load current during the maximum on-time, while limiting the voltage drop at VIN. For
this example, 0.5V is selected as the maximum allowable droop at VIN. Their minimum value is calculated
from:
IOUT(max) x tON(max)
5A x 2.55 ms
CIN + CBYP
=
=
= 25.5 mF
DV
0.5V
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A 33µF electrolytic capacitor is selected for CIN, and a 1µF ceramic capacitor is selected for CBYP. Due to the
ESR of CIN, the ripple at VIN will likely be higher than the calculation indicates, and therefore it may be
desirable to increase CIN to 47µF or 68µF. CBYP must be located as close as possible to the VIN and GND
pins of the LM25085. The voltage rating for both capacitors must be at least 42V. The RMS ripple current
rating for the input capacitors must also be considered. A good approximation for the required ripple current
rating is IRMS > IOUT/2.
•
D1: A Schottky diode is recommended. Ultra-fast recovery diodes are not recommended as the high speed
transitions at the SW pin may affect the regulator’s operation due to the diode’s reverse recovery transients.
The diode must be rated for the maximum input voltage, and the worst case current limit level. The average
power dissipation in the diode is calculated from:
PD1 = VF x IOUT x (1-D)
where VF is the diode’s forward voltage drop, and D is the on-time duty cycle. Using Equation 1, the minimum
duty cycle occurs at maximum input voltage, and is calculated to be ≊11.9% in this example. The diode power
dissipation calculates to be:
PD1 = 0.65V x 5A x (1- 0.119) = 2.86W
•
•
CVCC: The capacitor at the VCC pin (from VIN to VCC) provides not only noise filtering and stability for the
VCC regulator, but also provides the surge current for the PFET gate drive. The typical recommended value
for CVCC is 0.47µF. A good quality, low ESR, ceramic capacitor is recommended. CVCC must be located as
close as possible to the VIN and VCC pins. If the selected PFET has a Total Gate Charge specification of
100nC or larger, or if the circuit is required to operate at input voltages below 7V, a larger capacitor may be
required. The maximum recommended value for CVCC is 1µF.
IC Power Dissipation: The maximum power dissipated in the LM25085 package is calculated using
Equation 12 at the maximum input voltage. The Total Gate Charge for the Si7465 PFET is specified to be
40nC (max) in its data sheet. Therefore the total power dissipation within the LM25085 is calculated to be:
PDISS = 42V x ((40nC x 300kHz) + 1.3mA) = 559mW
Using an HVSSOP-PowerPAD-8 package with a θJA of 46°C/W produces a temperature rise of 26°C from
junction to ambient.
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Final Design Example Circuit
The final circuit is shown in Figure 28, and its performance is presented in Figure 29 through Figure 32.
C
VCC
7V to 42V
Input
LM25085
0.47 mF
VCC
ADJ
C
VIN
RT
V
ADJ
IN
1000 pF
C
BYP
C
IN
1 mF
33 mF
R
ADJ
R
SEN
GND
R
T
2.1 kW
0.01W
90.9 kW
ISEN
L1 15 mH
PGATE
V
OUT
SHUTDOWN
Q1
5V
C
OUT
R
R3
66.5 kW
C2
FB2
C1
3300 pF
100 mF
10 kW
D1
GND
0.1 mF
GND
R
FB1
FB
3.4 kW
Figure 28. Example Circuit
Figure 29. Efficiency vs. Load Current and VIN
(Circuit of Figure 28)
Figure 30. Frequency vs. VIN (Circuit of Figure 28)
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Figure 31. Current Limit vs. VIN (Circuit of
Figure 32. LM25085 Power Dissipation (Circuit of
Figure 28)
Figure 28)
Alternate Output Ripple Configurations
The minimum ripple configuration, using C1, C2 and R3, used in the example circuit, Figure 28, results in a low
ripple amplitude at VOUT determined mainly by the characteristics of the output capacitor and the ripple current in
L1. This configuration allows multiple ceramic capacitors to be used for VOUT if the output voltage is provided to
several places on the PC board. However, if a slightly higher level of ripple at VOUT is acceptable in the
application, and distributed capacitance is not used, the ripple required for the FB comparator pin can be
generated with fewer external components using the circuits shown below.
Reduced ripple configuration: In Figure 33, R3, C1 and C2 are removed (compared to Figure 28). A low value
resistor (R4) is added in series with COUT, and a capacitor (Cff) is added across RFB2. Ripple is generated at
VOUT by the inductor’s ripple current flowing through R4, and that ripple voltage is passed to the FB pin via Cff.
The ripple at VOUT can be set as low as 25mVp-p since it is not attenuated by RFB2 and RFB1. The minimum value
for R4 is calculated from:
25 mV
IOR(min)
R4 =
where IOR(min) is the minimum ripple current, which occurs at minimum input voltage. The minimum value for Cff
is determined from:
3 x tON(max)
Cff =
(RFB1//RFB2
)
where tON(max) is the maximum on-time, which occurs at minimum VIN. The next larger standard value capacitor
should be used for Cff.
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LM25085
L1
PGATE
Q1
V
OUT
D1
Cff
R4
R
FB2
FB
C
OUT
GND
R
FB1
GND
Figure 33. Reduced Ripple Configuration
Lowest cost configuration: This configuration, shown in Figure 34, is the same as Figure 33 except Cff is
removed. Since the ripple voltage at VOUT is attenuated by RFB2 and RFB1, the minimum ripple required at VOUT is
equal to:
VRIP(min) = 25mV x (RFB2 + RFB1)/RFB1
The minimum value for R4 is calculated from:
VRIP(min)
R4 =
IOR(min)
where IOR(min) is the minimum ripple current, which occurs at minimum input voltage.
LM25085
L1
PGATE
Q1
V
OUT
D1
R4
R
R
FB2
FB
C
OUT
GND
FB1
GND
Figure 34. Lowest Cost Ripple Generating Configuration
PC Board Layout
In most applications, the heat sink pad or tab of Q1 is connected to the switch node, i.e. the junction of Q1, L1
and D1. While it is common to extend the PC board pad from under these devices to aid in heat dissipation, the
pad size should be limited to minimize EMI radiation from this switching node. If the PC board layout allows, a
similarly sized copper pad can be placed on the underside of the PC board, and connected with as many vias as
possible to aid in heat dissipation.
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The voltage regulation, over-voltage, and current limit comparators are very fast and can respond to short
duration noise pulses. Layout considerations are therefore critical for optimum performance. The layout must be
as neat and compact as possible with all the components as close as possible to their associated pins. Two
major current loops conduct currents which switch very fast, requiring the loops to be as small as possible to
minimize conducted and radiated EMI. The first loop is that formed by CIN, Q1, L1, COUT, and back to CIN. The
second loop is that formed by D1, L1, COUT, and back to D1. The connection from the anode of D1 to the ground
end of CIN must be short and direct. CIN must be as close as possible to the VIN and GND pins, and CVCC must
be as close as possible to the VIN and VCC pins.
If the anticipated internal power dissipation of the LM25085 will produce excessive junction temperatures during
normal operation, a package option with an exposed pad must be used (HVSSOP-PowerPAD-8 or WSON-8).
Effective use of the PC board ground plane can help dissipate heat. Additionally, the use of wide PC board
traces, where possible, helps conduct heat away from the IC. Judicious positioning of the PC board within the
end product, along with the use of any available air flow (forced or natural convection) also helps reduce the
junction temperature.
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REVISION HISTORY
Changes from Revision G (March 2013) to Revision H
Page
•
Changed layout of National Data Sheet to TI format .......................................................................................................... 21
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PACKAGE OPTION ADDENDUM
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19-Jul-2013
PACKAGING INFORMATION
Orderable Device
LM25085MM/NOPB
LM25085MME/NOPB
LM25085MMX/NOPB
LM25085MY/NOPB
LM25085MYE/NOPB
LM25085MYX/NOPB
LM25085QMY/NOPB
LM25085QMYE/NOPB
LM25085QMYX/NOPB
LM25085SD/NOPB
LM25085SDE/NOPB
LM25085SDX/NOPB
Status Package Type Package Pins Package
Eco Plan Lead/Ball Finish
MSL Peak Temp
Op Temp (°C)
-40 to 125
-40 to 125
-40 to 125
-40 to 125
-40 to 125
-40 to 125
-40 to 125
-40 to 125
-40 to 125
-40 to 125
-40 to 125
-40 to 125
Device Marking
Samples
Drawing
Qty
(1)
(2)
(3)
(4/5)
ACTIVE
VSSOP
VSSOP
VSSOP
DGK
8
8
8
8
8
8
8
8
8
8
8
8
1000
Green (RoHS
& no Sb/Br)
CU SN
CU SN
CU SN
CU SN
CU SN
CU SN
CU SN
CU SN
CU SN
Call TI
Call TI
Call TI
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
SVZB
SVZB
SVZB
SVYB
SVYB
SVYB
SYLB
SYLB
SYLB
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
DGK
DGK
DGN
DGN
DGN
DGN
DGN
DGN
NGQ
NGQ
NGQ
250
3500
1000
250
Green (RoHS
& no Sb/Br)
Green (RoHS
& no Sb/Br)
MSOP-
PowerPAD
Green (RoHS
& no Sb/Br)
MSOP-
PowerPAD
Green (RoHS
& no Sb/Br)
MSOP-
PowerPAD
3500
1000
250
Green (RoHS
& no Sb/Br)
MSOP-
PowerPAD
Green (RoHS
& no Sb/Br)
MSOP-
PowerPAD
Green (RoHS
& no Sb/Br)
MSOP-
PowerPAD
3500
1000
250
Green (RoHS
& no Sb/Br)
WSON
WSON
WSON
Green (RoHS
& no Sb/Br)
L246B
L246B
L246B
Green (RoHS
& no Sb/Br)
4500
Green (RoHS
& no Sb/Br)
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
19-Jul-2013
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
OTHER QUALIFIED VERSIONS OF LM25085, LM25085-Q1 :
Catalog: LM25085
•
Automotive: LM25085-Q1
•
NOTE: Qualified Version Definitions:
Catalog - TI's standard catalog product
•
Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects
•
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
11-Oct-2013
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
B0
K0
P1
W
Pin1
Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant
(mm) W1 (mm)
LM25085MM/NOPB
LM25085MME/NOPB
LM25085MMX/NOPB
LM25085MY/NOPB
VSSOP
VSSOP
VSSOP
DGK
DGK
DGK
DGN
8
8
8
8
1000
250
178.0
178.0
330.0
178.0
12.4
12.4
12.4
12.4
5.3
5.3
5.3
5.3
3.4
3.4
3.4
3.4
1.4
1.4
1.4
1.4
8.0
8.0
8.0
8.0
12.0
12.0
12.0
12.0
Q1
Q1
Q1
Q1
3500
1000
MSOP-
Power
PAD
LM25085MYE/NOPB
LM25085MYX/NOPB
LM25085QMY/NOPB
LM25085QMYE/NOPB
LM25085QMYX/NOPB
LM25085SD/NOPB
MSOP-
Power
PAD
DGN
DGN
DGN
DGN
DGN
NGQ
8
8
8
8
8
8
250
3500
1000
250
178.0
330.0
178.0
178.0
330.0
178.0
12.4
12.4
12.4
12.4
12.4
12.4
5.3
5.3
5.3
5.3
5.3
3.3
3.4
3.4
3.4
3.4
3.4
3.3
1.4
1.4
1.4
1.4
1.4
1.0
8.0
8.0
8.0
8.0
8.0
8.0
12.0
12.0
12.0
12.0
12.0
12.0
Q1
Q1
Q1
Q1
Q1
Q1
MSOP-
Power
PAD
MSOP-
Power
PAD
MSOP-
Power
PAD
MSOP-
Power
PAD
3500
1000
WSON
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
11-Oct-2013
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
B0
K0
P1
W
Pin1
Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant
(mm) W1 (mm)
LM25085SDE/NOPB
LM25085SDX/NOPB
WSON
WSON
NGQ
NGQ
8
8
250
178.0
330.0
12.4
12.4
3.3
3.3
3.3
3.3
1.0
1.0
8.0
8.0
12.0
12.0
Q1
Q1
4500
*All dimensions are nominal
Device
Package Type Package Drawing Pins
SPQ
Length (mm) Width (mm) Height (mm)
LM25085MM/NOPB
LM25085MME/NOPB
LM25085MMX/NOPB
LM25085MY/NOPB
LM25085MYE/NOPB
LM25085MYX/NOPB
LM25085QMY/NOPB
VSSOP
DGK
DGK
DGK
DGN
DGN
DGN
DGN
DGN
DGN
NGQ
NGQ
NGQ
8
8
8
8
8
8
8
8
8
8
8
8
1000
250
210.0
210.0
367.0
210.0
210.0
367.0
210.0
210.0
367.0
210.0
210.0
367.0
185.0
185.0
367.0
185.0
185.0
367.0
185.0
185.0
367.0
185.0
185.0
367.0
35.0
35.0
35.0
35.0
35.0
35.0
35.0
35.0
35.0
35.0
35.0
35.0
VSSOP
VSSOP
3500
1000
250
MSOP-PowerPAD
MSOP-PowerPAD
MSOP-PowerPAD
MSOP-PowerPAD
3500
1000
250
LM25085QMYE/NOPB MSOP-PowerPAD
LM25085QMYX/NOPB MSOP-PowerPAD
3500
1000
250
LM25085SD/NOPB
LM25085SDE/NOPB
LM25085SDX/NOPB
WSON
WSON
WSON
4500
Pack Materials-Page 2
MECHANICAL DATA
DGN0008A
MUY08A (Rev A)
BOTTOM VIEW
www.ti.com
MECHANICAL DATA
NGQ0008A
SDA08A (Rev A)
www.ti.com
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Copyright © 2013, Texas Instruments Incorporated
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