LM2647LQX/NOPB中文资料_数据手册_规格书

LM2647

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SNVS210F –JUNE 2003–REVISED APRIL 2013

LM2647 Dual Synchronous Buck Regulator Controller

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1

FEATURES

DESCRIPTION

The LM2647 is an adjustable 200-500kHz dual

channel voltage-mode controlled high-speed

2

•

Input Voltage Range from 5.5V to 28V

Synchronous Dual-channel Interleaved

Switching

synchronous buck regulator controller ideally suited

for battery powered applications such as laptop and

notebook computers. The LM2647 requires only N-

channel FETs for both the upper and lower positions

of each synchronous stage. It features line

feedforward to improve the response to input

transients. At very light loads, the user can choose

between the high-efficiency Pulse-skip mode or the

constant frequency Forced-PWM mode. Lossless

current limiting without the use of external sense

resistors is made possible by sensing the voltage

drop across the bottom FET. A unique adaptive duty

cycle clamping technique is incorporated to

significantly reduce peak currents under abnormal

•

Forced-PWM or Pulse-skip Modes

Lossless Bottom-side FET Current Sensing

Adaptive Duty Cycle Clamping

High Current N-channel FET Drivers

Low Shutdown Supply Currents

Reference Voltage Accurate to Within ±1.5%

Output Voltage Adjustable down to 0.6V

Power Good flag and Chip Enable

Under-voltage Lockout

Over-voltage/Under-voltage Protection

Soft-start and Soft-shutdown

load

conditions.

The

two

independently

programmable outputs switch 180° out of phase

(interleaved switching) to reduce the input capacitor

and filter requirements. The input voltage range is

5.5V to 28V while the output voltages are adjustable

down to 0.6V.

Switching Frequency Adjustable 200kHz-

500kHz

APPLICATIONS

Standard supervisory and control features include

Soft-start, Power Good, output Under-voltage and

Over-voltage protection, Under-voltage Lockout, Soft-

shutdown and Enable.

•

Notebook Chipset Power Supplies

Low Output Voltage High-Efficiency Buck

Regulators

1

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of

Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

2

All trademarks are the property of their respective owners.

PRODUCTION DATA information is current as of publication date.

Products conform to specifications per the terms of the Texas

Instruments standard warranty. Production processing does not

necessarily include testing of all parameters.

LM2647

SNVS210F –JUNE 2003–REVISED APRIL 2013

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V

IN

5V

D2

(D1)

Q1

(Q4)

V

IN

BOOT1 (2)

HDRV1 (2)

C30

(C28)

C32

(C1)

FPWM VRON

L1

(L2)

Vo1

(Vo2)

SW1 (2)

R8 (R7)

ILIM1 (2)

C26

(C23)

C25

(C22)

R18

FPWM

EN

LDRV1 (2)

Q2

(Q5)

R17

R20

PGND1 (2)

SENSE1 (2)

POK

C31

PGOOD

V5

C17

C6 (C5)

R9 (R6)

(C14)

COMP1 (2)

C7 (C4)

V

FB1 (2)

SS1 (2)

DD

R1

R21

R23

FREQ

(R15)

(R14)

SGND

C16

(C15)

C29

R22

(R16)

R19

Figure 1. Typical Application (Channel 2 in parenthesis)

See Figure 24 for Expanded View

Connection Diagrams

Top View

28

27

26

SENSE1

FB1

1

2

3

ILIM1

SW1

COMP1

SS1

HDRV1

BOOT1

28 27 26 25 24 23 22

25

24

23

4

5

6

SS1

PGND1

LDRV

1

2

3

4

5

6

7

21

20

19

18

17

16

15

VDD

PGND1

LDRV1

V

DD

FREQ

SGND

EN

V

FREQ

SGND

EN

IN

22

21

7

8

VIN

V5

LDRV

V5

2

PGND

9

20

19

18

LDRV2

PGND2

BOOT2

PGOOD

FPWM

SS2

PGOOD

FPWM

2

BOOT

10

2

11

12

13

14

8

9 10 11 12 13 14

17

16

15

HDRV2

SW2

COMP2

FB2

ILIM2

SENSE2

Figure 2. 28-Lead TSSOP Package

See Package Number PW0028A

Figure 3. 28-Lead WQFN Package

See Package Number NJB0028A

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PIN DESCRIPTION

(All pin numbers referred to here correspond to the TSSOP package)

Pin 1, SENSE1: Output voltage sense pin for Channel 1. It is tied directly to the output rail. The SENSE pin voltage is used together with

the VIN voltage (on Pin 22) to (internally) calculate the CCM (continuous conduction mode) duty cycle. This calculation is used by the IC to

set the minimum duty cycle in the SKIP mode to 85% of the CCM value. It is also used to set the adaptive duty cycle clamp (see Pin 3). An

internal 20Ω resistor from the SENSE pin to ground discharges the output capacitor gently (Soft-shutdown) whenever Power Not Good is

signaled on Pin 9.

Pin 2, FB1: Feedback pin for Channel 1. This is the inverting input of the error amplifier. The voltage on this pin under regulation is

nominally at 0.6V. A Power Good window on this pin determines if the output voltage is within regulation limits (±13%). If the voltage (on

either channel) falls outside this window for more than 7µs, Power Not Good is signaled on the PGOOD pin (Pin 9). Output over-voltage

and under-voltage conditions are also detected by comparing the voltage on the Feedback pin with appropriate internal reference voltage

levels. If the voltage exceeds the safe window (±30%) for longer than 7µs, a fault condition is asserted. Then both the lower FETs are

latched ON and the upper FETs are latched OFF. When single channel operation is desired, the Feedback pins of both channels should be

connected together, near the IC. All other pins specific to the unused channel should be left floating (not connected to each other either).

Pin 3, COMP1: Compensation pin for Channel 1. This is also the output of the error amplifier of this channel. The voltage level on this pin is

compared with an internally generated ramp signal to set the duty cycle for normal regulation. Since the Feedback pin is the inverting input

of the same error amplifier, appropriate control loop compensation components are placed between this pin and the Feedback pin. The

COMP pin is internally pulled low during Soft-start so as to limit the duty cycle. Once Soft-start is completed, the voltage on this pin can

take up the value required to maintain output regulation. But an internal voltage clamp does not allow the pin to go much higher than the

steady-state requirement. This forms the adaptive duty cycle clamp feature which serves to limit the maximum allowable duty cycle and

peak currents under sudden overloads. But at the same time it has enough headroom to permit an adequate response to step loads within

the normal operating range.

Pin 4, SS1: Channel 1 Soft-start pin. A Soft-start capacitor is placed between this pin and ground. A typical capacitance of 0.1µF is always

recommended between this pin and ground. The IC connects an internal 1.8 kΩ resistor (R_{SS_DCHG}, see Electrical Characteristics table)

between this pin and ground to discharge any remaining charge on the Soft-start capacitor under several conditions. These conditions

include the initial power-up sequence, start-up by toggling the EN pin, and also recovery from a fault condition. The purpose is to bring

down the voltage on both the Soft-start pins to below 100mV for obtaining reset. Reset having thus been obtained, an 11µA current source

at this pin charges up the Soft-start capacitor. The voltage on this pin controls the maximum duty cycle, and this produces a gradual ramp-

up of the output voltage, thereby preventing large inrush currents into the output capacitors. The voltage on this pin finally clamps close to

5V. This pin is again connected to the internal 115µA current sink whenever a current limit event is in progress. This sink current discharges

the Soft-start capacitor and forces the duty cycle low to protect the power components. When a fault condition is asserted (See Pin 2) the

SS pin is internally connected to ground via the 1.8 kΩ resistor.

Pin 5, VDD: 5V supply rail for the control and logic sections of both channels. For normal operation to start, the voltage on this pin must be

brought above 4.5V. Subsequently, the voltage on this pin (including any ripple component) should not allowed to fall below 4V for a

duration longer than 7µs. Since this pin is also the supply rail for the internal control sections, it should be well-decoupled particularly at

high frequencies. A minimum 0.1µF-0.47µF (ceramic) capacitor should be placed on the component side very close to the IC with no

intervening vias between this capacitor and the VDD/SGND pins. If the voltage on Pin 5 falls below the lower UVLO threshold, both upper

FETs are latched OFF and lower FETs latched ON. Power Not Good is then also signaled immediately (on Pin 9). To effect recovery, the

EN pin must be taken below 0.8V and then back above 2V (with VDD held above 4.5V). Or the voltage on the VDD pin must be taken

below 1.0V and then back again above 4.5V (with EN pin held above 2V). Normal operation will then resume assuming that the fault

condition has cleared.

Pin 6, FREQ: Frequency adjust pin. The switching frequency (for both channels) is set by a resistor connected between this pin and

ground. A value of 22.1kΩ sets the frequency to 300kHz (nominal). If the resistance is increased, the switching frequency falls. An

approximate relationship is that for every 7.3kΩ increase (or decrease) in the value of the frequency adjust resistance, the time period (1/f)

increases (or decreases) by about 1µs.

Pin 7, SGND: Signal Ground pin. This is the lower rail for the control and logic sections of both channels. SGND should be connected on

the PCB to the system ground, which in turn is connected to PGND1 and PGND2. The layout is important and the recommendations in the

section LAYOUT GUIDELINES should be followed.

Pin 8, EN: IC Enable pin. When EN is taken high, both channels are enabled by means of a Soft-start power-up sequence (see Pin 4).

When EN is brought low, Power Not Good is signaled within 100ns. This causes Soft-shutdown to occur (see Pins 1 and 9). The Soft-start

capacitor is then discharged by an internal 1.8kΩ resistor (R_{SS_DCHG}, see Electrical Characteristics table). But note that when the Enable

pin is toggled, a fault condition is not asserted. Therefore in this case, the lower FETs are not latched ON, even as the output voltage ramps

down, eventually falling below the under-voltage threshold. In fact, in this situation, both the upper and the lower FETs of the two channels

are latched OFF, until the Enable pin is taken high again. If a fault shutdown has occurred, taking the Enable pin low and then high again

(toggling), resets the internal latches, and the IC will resume normal switching operation.

Pin 9, PGOOD: Power Good output pin. An open-Drain logic output that is pulled high with an external pull-up resistor, indicating that both

output voltages are within a pre-defined Power Good window. Outside this window, the pin is internally pulled low (Power Not Good

signaled) provided the output error lasts for more than 7µs. But the pin is also pulled low within 100ns of the Enable pin being taken low,

irrespective of the output voltage level. Note that PGOOD must always be high before it can respond by going low. So regulation on both

channels must be achieved first. Further, for fault monitoring to be in place, PGOOD must have been high prior to occurrence of the fault

condition. Note that since under a fault assertion, the lower FETs are always latched ON, this will not happen if regulation has not been

already been achieved first. For correct signaling on this pin under single-channel operation, see description of Pin 2.

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PIN DESCRIPTION (continued)

Pin 10, FPWM: Logic input for selecting either the Forced PWM (FPWM) Mode or Pulse-skip Mode (SKIP) for both channels (together).

When the pin is driven high, the IC operates in the FPWM mode, and when pulled low or left floating, the SKIP mode is enabled. In FPWM

mode, the lower FET of a given channel is always ON whenever the upper FET is OFF (except for a narrow shoot-through protection

deadband). This leads to continuous conduction mode of operation, which has a fixed frequency and (almost) fixed duty cycle down to very

light loads. But this does reduce efficiency at light loads. The alternative is the SKIP mode, where the lower FET remains ON only until the

voltage on the Switch pin (see Pin 27 or Pin 16) is more negative than 2.2mV (typical). So for example, for a 21mΩ FET, this translates to a

current threshold of 2.2mV/21mΩ = 0.1A. Therefore, if the (instantaneous) inductor current falls below this value, the lower FET will turn

OFF every cycle at this point (when operated in SKIP mode). This threshold is set by the zero-cross Comparator in the Figure 4. Note that if

the inductor current waveform is high enough to be always above this zero-cross threshold (V_{SW_ZERO}, see Table of Electrical

Characteristics), there will be no observable difference between FPWM and SKIP mode settings (in steady-state). SKIP mode, when it

actually occurs, is clearly a discontinuous mode of operation. However, note that in conventional discontinuous mode, the duty cycle keeps

falling (towards zero) as the load decreases. But the LM2647 does not allow the duty cycle to fall by more than 15% of its original value (at

the CCM-DCM boundary). This forces pulse-skipping, and the average frequency is effectively decreased as the load decreases. This

mode of operation improves efficiency at light loads, but the frequency is effectively no longer a constant. Note that a minimum pre-load of

0.1mA should be maintained on the output of each channel to ensure regulation in SKIP mode. The resistive divider from output to ground

used to set the output voltage could be designed to serve as part or all of this required pre-load.

Pin 11, SS2: Soft-start pin for Channel 2. See Pin 4.

Pin 12, COMP2: Soft-start pin for Channel 2. See Pin 3.

Pin 13, FB2: Feedback pin for Channel 2. See Pin 2.

Pin 14, SENSE2: Output voltage sense pin for Channel 2. See Pin 1.

Pin 15, ILIM2: Channel 2 Current Limit pin. When the bottom FET is ON, a 62µA (typical) current flows out of this pin into an external

current limit setting resistor connected to the Drain of the lower FET. This is a current source, therefore the drop across this resistor serves

to push the voltage on this pin to a more positive value. However, the Drain of the lower FET which is connected to the other side of the

same resistor is trying to go more negative as the load current increases. At some value of instantaneous current, the voltage on this pin

will transit from positive to negative. The point where it is zero is the current limiting condition and is detected by the Current Limit

Comparator in the Figure 4. When current limit condition has been detected, the next ON-pulse of the upper FET will be omitted. The lower

FET will again be monitored to determine if the current has fallen below the threshold. If it has, the next ON-pulse will be permitted. If not,

the upper FET will be turned OFF and will stay so for several cycles if necessary, until the current returns to normal. Eventually, if the

overcurrent condition persists, and the upper FET has not been turned ON, the output will clearly start to fall. Ultimately the output will fall

below the under-voltage threshold, and a fault condition will be asserted by the IC.

Pin 16, SW2: The Switching node of the buck regulator of Channel 2. Also serves as the lower rail of the floating driver of the upper FET.

Pin 17, HDRV2: Gate drive pin for the upper FET of Channel 2 (High-side drive). The top gate driver is interlocked with the bottom gate

driver to prevent shoot-through/cross-conduction.

Pin 18, BOOT2: Bootstrap pin for Channel 2. This is the upper supply rail for the floating driver of the upper FET. It is bootstrapped by

means of a ceramic capacitor connected to the channel Switching node. This capacitor is charged up by the IC to a value of about 5V as

derived from the V5 pin (Pin 21).

Pin 19, PGND2: Power Ground pin of Channel 2. This is the return path for the bottom FET gate drive. Both the PGND's are to be

connected on the PCB to the system ground and also to the Signal ground (Pin 7) in accordance with the recommended LAYOUT

GUIDELINES.

Pin 20, LDRV2: Gate drive pin for the Channel 2 bottom FET (Low-side drive). The bottom gate driver is interlocked with the top gate driver

to prevent shoot-through/cross-conduction. It is always latched high when a fault condition is asserted by the IC.

Pin 21, V5: Upper rail of the lower FET drivers of both channels. Also used to charge up the bootstrap capacitors of the upper FET drivers.

This is connected to an external 5V supply. The 5V rail may be the same as the rail used to provide power to the VDD pin (Pin 5), but the

VDD pin will then require to be well-decoupled so that it does not interact with the V5 pin. A low-pass RC filter consisting of a ceramic 0.1µF

capacitor (preferably 0.22µF) and a 10Ω resistor will suffice as shown in the Typical Applications circuit.

Pin 22, VIN: The input to both the Buck regulator power stages. It also is used by the internal ramp generator to implement the line

feedforward feature. The VIN pin is also used with the SENSE pin voltage to predict the CCM (continuous conduction mode) duty cycle and

to thereby set the minimum allowed DCM duty cycle to 85% of the CCM value (in SKIP mode, see Pin 10). This is a high input impedance

pin, drawing only about 100µA (typical) from the input rail.

Pin 23, LDRV1: LDRV pin of Channel 1. See Pin 20.

Pin 24, PGND1: PGND pin for Channel 1.See Pin 19.

Pin 25, BOOT1: Boot pin of Channel 1. See Pin 18.

Pin 26, HDRV1: HDRV pin of Channel 1. See Pin 17.

Pin 27, SW1: SW pin of Channel 1. See Pin 16.

Pin 28, ILIM1: Channel 2 Current Limit pin. See Pin 15.

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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam

during storage or handling to prevent electrostatic damage to the MOS gates.

Absolute Maximum Ratings⁽¹⁾⁽²⁾

Voltages from the indicated pins to SGND/PGND unless otherwise indicated.⁽³⁾

VIN

30V

V5

7V

VDD

7V

BOOT1, BOOT2

36V

BOOT1 to SW1, BOOT2 to SW2

SW1, SW2

7V

30V

ILIM1, ILIM2

SENSE1, SENSE2, FB1, FB2

PGOOD

7V

EN

7V

Power Dissipation (T_A= 25°C)⁽⁴⁾

Junction Temperature

ESD Rating⁽⁵⁾

1.0W

+150°C

2kV

Ambient Storage Temperature Range

Soldering Dwell Time, Temperature

-65°C to +150°C

4 sec, 260°C

10 sec, 240°C

75 sec, 219°C

Wave

Infrared

Vapor Phase

(1) Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating Ratings are conditions under which

operation of the device is ensured. For ensured performance limits and associated test conditions, see the Electrical Characteristics

table.

(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and

specifications.

(3) PGND1, PGND2 and SGND are all electrically connected together on the PCB.

(4) The maximum allowable power dissipation is calculated by using P_Dmax= (T_JMAX- T_A) /θ_JA, where T_JMAXis the maximum junction

temperature, T_Ais the ambient temperature, and θ_JAis the junction-to-ambient thermal resistance of the specified package. The 1.0W

rating of the TSSOP-28 package for example results from using 125°C, 25°C, and 97°C/W for T_JMAX, T_A, and θ_JArespectively. The

2.85W rating of the 28-pin WQFN package results from using 125°C, 25°C, and 35°C/W for T_JMAX, T_A, and θ_JArespectively. The rated

power dissipation should be derated by 10mW/°C above 25°C ambient for the TSSOP package and 29mW/°C above 25°C ambient for

the WQFN package. The θ_JAvalue above represents the worst-case condition with no heat sinking. Heat sinking will permit more power

to be dissipated at higher ambient temperatures. For detailed information on soldering plastic TSSOP and WQFN packages, refer to

http://www.ti.com/packaging/.

(5) ESD is applied by the human body model, which is a 100pF capacitor discharged through a 1.5 kΩ resistor into each pin.

Operating Ratings⁽¹⁾

VIN

5.5V to 28V

4.5V to 5.5V

VDD, V5

Junction Temperature

-5°C to +125°C

(1) Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating Ratings are conditions under which

operation of the device is ensured. For ensured performance limits and associated test conditions, see the Electrical Characteristics

table.

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Electrical Characteristics

Specifications with standard typeface are for T_J= 25°C, and those with boldface apply over full Operating Junction

Temperature range. VDD = V5 = 5V, V_SGND= V_PGND= 0V, VIN = 15V, V_EN= 3V, R_FADJ= 22.1K unless otherwise stated.⁽¹⁾

Symbol

Reference

V_{FB_REG}

Parameter

Conditions

Min⁽²⁾

Typical⁽³⁾

Max⁽²⁾

Units

FB Pin Voltage at Regualtion

(either FB Pin)

VDD = 4.5V to 5.5V,

VIN = 5.5V to 28V

591

600

0.5

20

609

mV

V_FBLine Regulation

VDD = 4.5V to 5.5V,

VIN = 5.5V to 28V

I_FB

FB Pin Current (sourcing)

V_FBat regulation

100

nA

Chip Supply

I_{Q_VIN}

VIN Quiescent Current

VIN Shutdown Current

VDD Quiescent Current

VDD Shutdown Current

V5 Normal Operating Current

V_FB1= V_FB2= 0.7V

V_EN= 0V

100

0

200

5

µA

I_{SD_VN}

I_{Q_VDD}

I_{SD_VDD}

I_{Q_V5}

V_FB1= V_FB2= 0.7V

V_EN= 0V

2.5

8

4

mA

µA

15

0.5

1.5

5

V_FB1= V_FB2= 0.7V

V_FB1= V_FB2= 0.5V

V_EN= 0V

0.3

1

mA

I_{SD_V5}

V5 Shutdown Current

0

µA

I_{Q_BOOT}

BOOT Quiescent Current

V_FB1= V_FB2= 0.7V

V_FB1= V_FB2= 0.5V

V_EN= 0V

2

5

300

1

500

5

I_{SD_BOOT}

V_UVLO

BOOT Shutdown Current

VDD UVLO Threshold

VDD UVLO Hysteresis

µA

V

VDD rising from 0V

VDD = V5 falling from V_UVLO

3.9

0.5

4.2

0.7

4.5

0.9

V

Logic

I_EN

EN Input Current

V_EN= 0 to 5V

0

µA

V

V_{EN_HI}

V_{EN_LO}

EN Input Logic High

EN Input Logic Low

FPWM Pull-down

2

1.8

1.3

200

1.8

1.3

0.8

V

V_FPWM= 2V

100

2

1000

kΩ

V

V_{FPWM_HI}

FPWM Input Logic High

FPWM Input Logic Low

0.8

V

FPWM_LO

V

Power Good

V_{PGOOD_HI}

Power Good Upper Threshold as

a Percentage of Internal

Reference

FB voltage rising above V_{FB_REG}

FB voltage falling below V_{FB_REG}

110

84

113

87

116

%

V_{PGOOD_LOW}

Power Good Lower Threshold as

a Percentage of Internal

Reference

90

Power Good Hysteresis

Power Good Delay

7

%

Δt_{PG_OK}

Δt_{PG_NOK}

Δt_SD

From both output voltages “good”

to PGOOD assertion.

10

4

20

30

10

µs

From the first output voltage “bad”

to PGOOD de-assertion

7

From Enable low to PGOOD low

0.03

0.12

0.1

0.4

PGOOD Saturation Voltage

PGOOD Leakage Current

PGOOD de-asserted (Power Not

Good) and sinking 1.5mA

V

PGOOD = 5V and asserted

0

1

µA

OV and UV Protection

Fault OVP Latch Threshold as a

Percentage of Internal Reference

FB voltage rising above V_{FB_REG}

125

130

135

%

(1) R_FADJis the frequency adjust resistor between FREQ pin and Ground.

(2) All limits are specified at room temperature (standard face type) and at temperature extremes (bold face type). All room temperature

limits are 100% production tested. All limits at temperature extremes are specified via correlation using Statistical Quality Control (SQC)

methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).

(3) Typical numbers are at 25°C and represent the most likely norm.

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Electrical Characteristics (continued)

Specifications with standard typeface are for T_J= 25°C, and those with boldface apply over full Operating Junction

Temperature range. VDD = V5 = 5V, V_SGND= V_PGND= 0V, VIN = 15V, V_EN= 3V, R_FADJ= 22.1K unless otherwise stated.⁽¹⁾

Symbol

Parameter

Conditions

Min⁽²⁾

Typical⁽³⁾

Max⁽²⁾

Units

Fault UVP Latch Threshold as a

Percentage of Internal Reference

FB voltage falling below V_{FB_REG}

65

70

75

%

Δt_FAULT

Fault Delay

From Fault detection (any output)

to Fault assertion

7

µs

Soft-start

I_{SS_CHG}

Soft-start Charging Current

V_SS= 1V

8

11

14

µA

R_{SS_DCHG}

Soft-shutdown Resistance (SS pin V_EN= 0V, V_SS= 1V

to SGND, either channel)

1800

Ω

I_{SS_DCHG}

Soft-start Discharge Current

Soft-start pin reset voltage⁽⁴⁾

In Current Limit

80

115

100

160

µA

V_{SS_RESET}

SS charged to 0.5V, EN low to

high

mV

SS to COMP Offset Voltage

V_SS= 0.5V and 1V, V_FB1= V_FB2

0V

=

600

mV

Error Amplifier

GAIN

DC Gain

70

4.45

2.25

6.5

5

dB

Voltage Slew Rate

COMP rising

COMP falling

V/µs

BW

Unity Gain Bandwidth

COMP Source Current

MHz

mA

V_FB< V_{FB_REG}

V_COMP= 0.5V

2

7

COMP Sink Current

V_FB> V_{FB_REG}

V_COMP= 0.5V

14

62

mA

µA

Current Limit and Zero-Cross

I_ILIM

I_LIMPin Current (sourcing, either

ILIM pin)

V_ILIM1= V_ILIM2= 0V

LDRV goes low

46

76

10

I_ILIMThreshold Voltage

Zero-cross Threshold (SW Pin)

-10

0

mV

V_{SW_ZERO}

-2.2

Osillator

PWM Frequency

R_FADJ= 22.1kΩ

R_FADJ= 12.4kΩ

R_FADJ= 30.9kΩ

VIN = 15V

255

300

500

200

1.6

345

kHz

V

PWM Ramp Peak-to-peak

Amplitude

VIN = 24V

2.95

0.8

PWM Ramp Valley

V

%

Frequency Change with VIN

Frequency Change with VDD

Phase Shift Between Channels

FREQ Pin Voltage vs. VIN

VIN = 5.5V to 24V

±1

VDD = 4.5V to 5.5V

±2

%

Phase from HDRV1 to HDRV2

165

180

0.105

195

deg

V/V

System

Minimum ON Time

V_FPWM= 3V

30

75

50

28

ns

%

VIN = 5.5V

60

40

22

Maxmimum Duty Cycle

VIN = 15V

VIN = 28V, VDD= 4.5V

Gate Drivers

HDRV Source Impedance

HDRV Sink Impedance

HDRV Pin Current (sourcing)=

1.2A

7

2

Ω

HDRV Pin Current (sinking) = 1A

(4) If the LM2647 starts up with a pre-charged soft start capacitor, it will first discharge the capacitor to V_{SS_RESET}and then begin the

normal Soft-start process.

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Electrical Characteristics (continued)

Specifications with standard typeface are for T_J= 25°C, and those with boldface apply over full Operating Junction

Temperature range. VDD = V5 = 5V, V_SGND= V_PGND= 0V, VIN = 15V, V_EN= 3V, R_FADJ= 22.1K unless otherwise stated.⁽¹⁾

Symbol

Parameter

Conditions

Min⁽²⁾

Typical⁽³⁾

Max⁽²⁾

Units

LDRV Source Impedance

LDRV Pin Current (sourcing) =

1.2A

7

Ω

LDRV Sink Impedance

LDRV Pin Current (sinking) = 2A

1

Ω

Cross-conduction protection delay HDRV Falling to LDRV Rising

40

70

ns

(deadtime)

LDRV Falling to HDRV Rising

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Block Diagram

SENSE1

1

V_DD

Controller 1

BOOT1

25

SOFT-SHDN1

6

2

mA

SKIP

HS_ON

CMP

HDRV1

20R

0.35V

IN

+

-

26

COMP1

3

EN

SW1

27

ILIM1

28

CURRENT

LIMIT CMP

PWM

CMP

Vref1

E.A.

-

+

-

+

PWM

LOGI

C

V_DD

CL1

V5

LS_ON

+

-

V5

-

85%Ton

21

+

ZERO

CROSS CMP

EN

IN

LDRV1

23

SENSE1

PGND1

24

FB1

2

BOOT2

18

SENSE2

14

Supervisory 1

1.3 x Vref1

0.7 x Vref1

OVP1

+

-

SOFT-

SHDN1

HDRV2

17

COMP2

12

UVP1

+

-

SW2

16

CONTROL

LOGIC

FB2

13

1.13 x Vref1

Controller2

PGOOD1

+

-

ILIM2

15

+

Supervisory 2

SS2

11

0.87 x Vref1

V_DD

+

-

V5

LDRV2

20

V_DD

11mA

PGND2

19

SS1

4

COMP1

1.8k

115mA

OVP1/UVP1

CL1

V_DD

5

RAMP/

TIMING

BIAS/

REFERENCES

V_DDUV

+

-

4.2V

V_DD

EN

FPWM

SGND

V_IN

22

FREQ

PGOOD

7

8

10

6

9

PINOUTS SHOWN

ARE FOR TSSOP-28

Figure 4. Block Diagram

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Typical Performance Characteristics

Input Voltage is 15V, 20V, 24V,28V (in order) starting from uppermost curve to lowermost curve in each of the Efficiency plots

below.

Efficiency for 5V/3.3V Outputs

Efficiency for 2.5V/3.3V Outputs

100

95

100

95

90

85

80

75

85

80

75

70

65

60

70

65

60

1

2.75

3

2.75

3

2.25 2.5

0.5 0.75

1.25 1.5 1.75

(A)

2

2.25 2.5

0.5 0.75 1 1.25 1.5 1.75

(A)

2

I

O

I

O

Figure 5.

Figure 6.

Efficiency for 1.8V/1.2V Outputs

Modulator (Plant) Gain

100

40

16

-8

5 mH

95

90

3.3 mH

85

80

75

-32

-56

10 mH

70

65

60

-80

10k

100

100k

1k

1M

10

10M

1

2.75 3

0.5 0.75

1.25 1.5 1.75 2 2.25 2.5

(A)

FREQUENCY (Hz)

I

O

Figure 7.

Figure 8.

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OPERATION DESCRIPTIONS

GENERAL

The LM2647 provides two identical synchronously switched buck regulator channels that operate 180° out of

phase. A voltage-mode control topology was selected to provide fixed-frequency PWM regulation at very low

duty cycles, in preference to current-mode control, because the latter has inherent limitations in being able to

achieve low pulse widths due to blanking time requirements. Because of a minimum pulse width of about 30ns

for the LM2647, very low duty cycles (low output, high input) are possible. The main advantage of current-mode

control is the fact that the slope of its ramp (derived from the switch current), automatically increases with

increase in input voltage. This leads to improved line rejection and fast response to line variations. In typical

voltage-mode control, the ramp is derived from the clock, not from the switch current. But by using the input

voltage together with the clock signal to generate the ramp as in the LM2647, this advantage of current-mode

control can in fact be completely replicated. The technique is called line feedforward. In addition, the LM2647

features a user-selectable Pulse-skip mode that significantly improves efficiency at light loads by reducing

switching losses, and driver consumption, both of which are proportional to switching frequency.

INPUT VOLTAGE FEEDFORWARD

The feedforward circuit of the LM2647 adjusts the slope of the internal PWM ramp in proportion to the regulator

input voltage. See Figure 9 for an illustration of how the duty cycle changes as a result of the change in the slope

of the ramp, even though the error amplifier output has not had time to react to the line disturbance. The almost

instantaneous duty cycle correction provided by the feedforward circuit significantly improves line transient

rejection.

RAMP

Error Amp O/P

VIN = Low

PWM

RAMP

Error Amp O/P

PWM

VIN = High

Figure 9. Voltage Feedforward

FORCED-PWM MODE AND PULSE-SKIP MODE

Forced-PWM mode (FPWM) leads to Continuous Conduction Mode (CCM) even at very light loads. It is one of

two user-selectable modes of operation provided by the LM2647. When FPWM is chosen (FPWM pin high), the

bottom FET will always be turned ON whenever the top FET is OFF. See Figure 10 for a typical FPWM plot.

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CH1: HDRV, CH2: LDRV, CH3: SW, CH4: I_L(0.2A/div)

Output 1V @ 0.04A, VIN = 10V, FPWM, L = 10µH, f = 300kHz

Figure 10. Normal FPWM Mode Operation at Light Loads

In a conventional converter, as the load is decreased to about 10-30% of maximum load current, DCM

(Discontinuous Conduction Mode) occurs. In this condition the inductor current falls to zero during the OFF-time,

and stays there until the start of the next switching cycle. In this mode, if the load is decreased further, the duty

cycle decreases (pinches off), and ultimately may decrease to the point where the required pulse width becomes

less than the minimum ON-time achievable by the converter (controller + FETs). Then a sort of random skipping

behavior occurs as the error amplifier struggles to maintain regulation. This is not the most desirable type of

behavior. There are two ways out of this problem.

One way is to keep the lower FET ON until the start of the next cycle (as in the LM2647 operated in FPWM

mode). This allows the inductor current to drop to zero and then actually reverse direction (negative direction

through inductor, passing from Drain to Source of lower FET, see Channel 4 in Figure 10). Now the current can

continue to flow continuously till the end of the switching cycle. This maintains CCM and so the duty cycle does

not start to pinch off as in typical DCM. Nor does it lead to the undesirable random skipping described above.

Note that the pulse width (duty cycle) for CCM is virtually constant for any load and therefore does not usually

run into the minimum ON-time restriction. But it can happen, especially when the application consists of a very

high input voltage, a low output voltage rail, and also the switching frequency is set high. Let us check the

LM2647 to rule out this remote possibility. For example, with an input of 24V, an output of 1V, the duty cycle is

1/24 = 4.2%. This leads to a required ON-time of 0.042* 3.3 = 0.14 µs at a switching frequency of 300kHz (T=3.3

µs). Since 140ns exceeds the minimum ON-time of 30ns of the LM2647, normal constant frequency CCM mode

of operation is assured in FPWM mode, at virtually any load.

The second way out of the problems of discontinuous mode is the second operating mode of the LM2647, the

Pulse-skip (SKIP) Mode. In SKIP Mode, a zero-cross detector at the SW pin turns off the bottom FET when the

inductor current decays to zero (actually at V_{SW_ZERO}, see Electrical Characteristics table). This would however

still amount to conventional DCM, with its attendant problems at extremely light loads as described earlier. The

LM2647 however avoids the random skipping behavior described earlier, and replaces it with a more defined or

formal SKIP mode. In conventional DCM, a converter would try to reduce its duty cycle from the CCM value as

the load decreases, as explained previously. So it would start with the CCM duty cycle value (at the CCM-DCM

boundary), but as the load decreases, the duty cycle would try to shrink to zero. However, in the LM2647, the

DCM duty cycle is not allowed to fall below 85% of the CCM value. So when the theoretically required DCM duty

cycle value falls below what the LM2647 is allowed to deliver (in this mode), pulse-skipping starts. It will be seen

that several of these excess pulses may be delivered, until the output capacitors charge up enough to notify the

error amplifier and cause its output to reverse. Thereafter several pulses could be skipped entirely until the

output of the error amplifier again reverses. The SKIP mode therefore leads to a reduction in the average

switching frequency. Switching losses and FET driver losses, both of which are proportional to frequency, are

significantly reduced at very light loads and efficiency is boosted. SKIP mode also reduces the circulating

currents and energy associated with the FPWM mode. See Figure 11 for a typical plot of SKIP mode at very light

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loads. Note the bunching of several fixed-width pulses followed by skipped pulses. The average frequency can

actually fall very low at very light loads. Note however that when this happens the inductor core is seeing only

very mild flux excursions, and so no significant audible noise is created. But if EMI is a particularly sensitive issue

for the particular application, the user can simply opt for the slightly less efficient, though constant frequency

FPWM mode.

CH1: HDRV, CH2: LDRV, CH3: SW, CH4: I_L(0.2A/div)

Output 1V @ 0.04A, VIN = 10V, SKIP, L = 10µH, f = 300kHz

Figure 11. Normal SKIP Mode Operation at Light Loads

The SKIP mode is enabled when the FPWM pin is held low (or left floating). Note that at higher loads, and under

steady state conditions (above CCM-DCM boundary), there will be absolutely no difference in the behavior of the

LM2647 or the associated converter waveforms based on the voltage applied on the FPWM pin. The differences

show up only at light loads.

Under startup too, since the currents are high until the output capacitors have charged up, there will be no

observable difference in the shape of the ramp-up of the output rails in either SKIP mode or FPWM mode. The

design has thus forced the startup waveforms to be identical irrespective of whether the FPWM mode or the

SKIP mode has been selected.

The designer must realize that even at zero load condition, there is circulating current when operated in FPWM

mode. This is illustrated in Figure 12. Since duty cycle is the same as for conventional CCM, from V = L* ΔI / Δt it

can be seen that ΔI (or Ipp in Figure 12) must remain constant for any load, including zero. At zero load, the

average current through the inductor is zero, so the geometric center of the sawtooth waveform (the center being

always equal to load current) is along the x-axis. At critical conduction (boundary between conventional CCM and

what should have been DCM were it not in FPWM mode), the load current is equal to Ipp/2. Note that

excessively low values of inductance will produce much higher current ripple and this will lead to higher

circulating currents and dissipation.

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CRITICAL CONDUCTION

Io = Ipp/2

Ipp/2 > Io > 0

NO LOAD

Io = 0

LDRV

HDRV

Figure 12. Inductor Current in FPWM Mode

NOTE

A common question is: can one change from FPWM to SKIP Mode ‘on the fly’? That

means that the voltage on the FPWM pin would be changed while the converter is

operating normally (with outputs in regulation). This is generally not recommended. The

designer must realize that doing so would in essence represent a fundamental change

applied to the system. The pulse widths would need to re-adjust suddenly and in the

process momentary imbalances can be created. For example, there is an observed

negative surge current passing from Drain to Source of the lower FET. It must be kept in

mind that though the LM2647 has current limiting for current passing in the ‘positive'

direction (positive with regards to the inductor, i.e. passing from Source to Drain of the

lower FET), there is no set limit for reverse currents. The amount of reverse current when

the FPWM pin is toggled ‘on the fly' can be very high. This current is determined by

several factors. One key factor is the output capacitance. Large output capacitances will

lead to higher peak reverse currents. The reverse swing will be worse for lighter loads

because of the bigger difference between the duty cycles/average frequency in the two

modes. See Figure 13 for a plot of what happened in going from SKIP to FPWM mode at

0A load (worst case). The peak reverse current was as high as 3A, lasting about 0.1ms.

The inductor could also saturate severely at this point if designed for light loads. In

general, if the designer wants to toggle the FPWM pin while the converter is operating,

both the low side FET rating and the inductor peak current rating must be closely

evaluated under this condition.

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CH1: PGOOD, CH2: Vo, CH3: LDRV, CH4: I_L(1A/div)

Output 1V @ 0A, VIN = 10V, L = 10µH, f = 300kHz

Figure 13. SKIP to FPWM 'On The Fly'

SOFT-START

The maximum output voltage of the error amplifier is limited during start-up by the voltage on the 0.1µF capacitor

connected between the SS pin and ground. When the controller is enabled (by taking EN pin high) the following

steps may occur. First the SS capacitor is discharged (if it has a pre-charge) by a 1.8 kΩ internal resistor

(R_{SS_DCHG}, see Electrical Characteristics). This ensures that reset is obtained. Note that reset is said to occur

only when the voltage on both the SS pins falls below 100mV (V_{SS_RESET}, see Electrical Characteristics table).

Then a charging current source I_{SS_CHG}of 11µA is applied at this pin to bring up the voltage of the Soft-start

capacitor voltage gradually. This causes the (maximum allowable) duty cycle to increase slowly, thereby limiting

the charging current into the output capacitor and also ensuring that the inductor does not saturate. The Soft-

start capacitor will eventually charge up close to the 5V input rail. When EN is pulled low the Soft-start capacitor

is discharged by the same 1.8 kΩ internal resistor and the controller is shutdown. Now the sequence is allowed

to repeat the next time EN is taken high.

The above Soft-start sequence is actually initiated not only whenever EN is taken high, but also under a normal

power-up or during recovery from a fault condition (more on this later).

As mentioned in the section FORCED-PWM MODE AND PULSE-SKIP MODE under startup, since the currents

are high until the output capacitors have charged up, there will be no observable difference in the shape of the

ramp-up of the output rails in either SKIP mode or FPWM mode. The design has thus forced the startup

waveforms to be identical irrespective of whether the FPWM mode or the SKIP mode has been selected.

SHUTDOWN/SOFT-SHUTDOWN

When the EN pin is driven low, the LM2647 initiates shutdown by turning OFF both upper and lower FETs

completely (this occurs irrespective of FPWM or SKIP modes). See Figure 14 for a typical shutdown plot and

note that the LDRV goes to zero (and stays there). Though not displayed, Power Good also goes low within less

than 100ns of the EN pin going low (Δt_SD, see Electrical Characteristics table). Therefore in this case, the

controller is NOT waiting for the output to actually fall out of the Power Good window before it signals Power Not

Good. Note that since there is a constant current 2A load applied at the output, the stored charge on the output

capacitor continues to be discharged into the load. From ΔV/Δt=i/C=2A/330µF it can be seen that the output

voltage (say 1V) will fall to zero in about 165µs, as will be observed.

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CH1: LDRV, CH2: Vo, CH3: SW, CH4: I_L(1A/div)

Output 1V @ 2A, VIN = 10V, FPWM/SKIP, L = 10µH, f = 300kHz, C_OUT= 330µF

Figure 14. Shutdown

But if the load is very close to zero, the only means for the output capacitor to discharge is through the resistive

divider on the feedback pin (if any) and any internal bleeder resistor present. In fact there is such an internal

bleeder resistor in the LM2647 and it performs Soft-shutdown by discharging the output capacitors gradually. Its

value is about 20Ω and it is internally connected between the SENSE pin and ground whenever the EN pin is

taken low. Note that this will be perceivable only when the external load is small, and provided a normal

shutdown is being carried out. Normal shutdown as being defined here calls for the Enable pin to be the cause of

the outputs being disabled. In a shutdown provoked by a fault, the situation is very different as will be explained

later.

POWER GOOD/NOT GOOD SIGNALING

PGOOD is an open-Drain output pin with an external pull-up resistor connected to 5V. It goes high (non-

conducting) when both the outputs are within the regulation band as determined by the Power Good window

detector stage on the feedback pin (see Figure 4). PGOOD goes low (conducting) when either of the two outputs

falls out of this window. This signal is referred to as Power Not Good here. A glitch filter of 7µs filters out noise,

and helps prevent spurious PGOOD responses. So Power Not Good is not asserted until 7µs after either of the

two outputs have fallen out of the Power Good window (see Δt_{PG_NOK}in Electrical Characteristics table). With the

feedback pin voltage rising towards regulation value, there is a 20µs delay between both the outputs being in

regulation and the signaling of Power Good (see Δt_{PG_OK}in Electrical Characteristics). Power Not Good is

signaled within 100ns of the Enable pin being pulled low (see Δt_SDin Electrical Characteristics table), irrespective

of the fact that the outputs could still be in regulation. The Soft-start capacitor is also then discharged as

explained earlier.

VIN POWER-OFF

The LM2647 has an internal comparator that also looks at VIN. If VIN falls to about 4.5V (roughly), switching

ceases. The response is slightly different under FPWM or SKIP modes, but the final result is the same. In both

cases ultimately, LDRV is latched high and so the output capacitors are discharged through the lower FETs.

Power Not Good has meanwhile already been signaled and a fault condition is asserted shortly thereafter.

In Figure 15 and Figure 16 the situation where the connection to the input DC power source is abruptly removed

is shown for two cases.

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CH1: PGOOD, CH2: VIN, CH3: LDRV, CH4: Vo

Output 1V @ 0.02A, VIN = 9.75V, FPWM, L = 10µH, f = 300kHz, C_OUT= 660µF

Figure 15. VIN Removal in FPWM Mode

In the first case (FPWM mode, Figure 15), LDRV goes high immediately, as soon as VIN falls to about 4.5V. For

the second case (SKIP mode, Figure 16), the output starts to discharge into the load resistor. Then Power Not

Good is signaled. Finally, when the output falls below the Under-voltage threshold a fault condition is asserted.

This is accompanied by LDRV latching high. The output then suddenly collapses just as it does for FPWM mode.

Note that once VIN reaches 4.5V, it does not fall quickly thereafter. The reason is that there is no applied

external voltage dragging it low (in our case as it is described), nor is there any significant consumption from the

VIN rail since the converter has stopped switching.

CH1: PGOOD, CH2: VIN, CH3: LDRV, CH4: Vo

Output 1V @ 0.02A, VIN = 9.75V, SKIP, L = 10µH, f = 300kHz, C_OUT= 660µF

Figure 16. VIN Removal in SKIP Mode

The recovery procedure from a VIN Power-off is the same as for any fault condition.

VDD POWER-OFF (UVLO)

Whenever VDD starts to fall, and drops below about 4V, LDRV goes high immediately, ‘Power Not Good’ is

signaled and in effect a fault condition (in this case an Under-voltage lockout) is asserted. Recovery from a fault

is discussed next.

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FAULT AND RECOVERY

If any output falls outside the Power Good window, the response is a ‘Power Not Good’ signal. The FET drive

signals are not affected. But under a fault condition assertion, LDRV goes high immediately turning the low side

FETs ON and discharging the output capacitors. Note that the current will then invariably slew momentarily

negative (passing from Drain to Source of lower FETs), before it settles down to zero.

A fault will be detected when either output falls below the Under-voltage threshold, or rises above the Over-

voltage threshold. From its detection to assertion, there is a 7µs delay to help prevent spurious responses.

A fault condition is also asserted during a loss of the VIN rail or the VDD rail, though not if shutdown is achieved

by use of the Enable pin.

To recover from a fault, either of the following options is available:

a) Enable pin is toggled: i.e. taken low (below 0.8V), then high again (2V to 5V). This must be done with VDD

between 4.5V to 5V and VIN within normal range (5.5V to 28V).

b) VDD is brought below 1.0V and then brought back up between 4.5V to 5V. This must be done with the Enable

pin held high (2V to 5V) and VIN within normal range (5.5V to 28V).

Recovery will initiate a Soft-start sequence (see description under section SOFT-START above).

CURRENT LIMIT AND PROTECTION

Output current limiting is achieved by sensing the negative Vds drop across the low side FET when the FET is

turned on. The Current Limit Comparator (see Figure 4) monitors the voltage at the ILIM pin with 62µA (typical

value) of current being sourced from the pin. The 62µA source flows through an external resistor connected

between ILIM and the Drain of the lower FET. The voltage drop across the ILIM resistor is compared with the

drop across the lower FET and the current limit comparator trips when the two are of the same magnitude. This

determines the threshold of current limiting. For example, if excessive inductor current causes the voltage across

the lower FET to exceed the voltage drop across the ILIM resistor, the ILIM pin will go negative (with respect to

ground) and trip the comparator. The comparator then sets a latch which prevents the top FET from turning ON

during the next PWM clock cycle. The top FET will resume switching only if the current limit comparator was not

tripped in the previous switching cycle.

The Soft-start capacitor at the SS pin is discharged with a 115µA current source when an overcurrent event is in

progress. Therefore if the overcurrent condition does not last long enough to cause a fault assertion, the Soft-

start capacitor will charge back up (by I_{SS_CHG}, see Electrical Characteristics table), without any user intervention.

The purpose of discharging the Soft-start capacitor during an overcurrent event is to eventually allow the voltage

on the SS pin to fall low enough to cause additional duty cycle limiting (over and above the protection provided

by the adaptive duty cycle clamp). Note that once the duty cycle starts pinching-off as a result of the progressive

reduction in SS pin voltage, the output voltage will certainly start collapsing (if it hasn’t done so already), and this

will hasten a fault condition assertion (an Under-voltage in this case). Thereafter, a normal fault-recovery

sequence will have to be initiated to cause the outputs to return to regulation.

There is a race condition in effect, between the current limit being reached and a fault being asserted (Under-

voltage). It could happen that if the load current was very low before the sudden overload was applied, a fault

condition could be asserted even before the current limit has been reached. See the differences between

Figure 17 and Figure 18 to see the possibilities. Also see Application Information for a deeper understanding of

current limiting discussed at a quantitative level.

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CH1: PGOOD, CH2: Vo, CH3: ILIM Pin, CH4: I_L(1A/div)

Output 1V, 0.04A to Overload, VIN = 10V, FPWM, L = 10µH, f = 300kHz, RLIM = 1k

Figure 17. Response to Severe Overload (Type A: fault threshold first)

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CH1: LDRV, CH2: PGOOD, CH3: ILIM Pin, CH4: I_L(5A/div)

Output 1V, 2A to Overload, VIN = 10V, L = 10µH, f = 300kHz, RLIM = 2k

Figure 18. Response to Severe Overload (Type B: current limit threshold first)

Application Information

CURRENT LIMIT RESISTOR

The timing scheme implemented in the LM2647 makes it possible for the IC to continue monitoring an

overcurrent condition and to respond appropriately every cycle. This is explained as follows.

Consider the LM2647 working under normal conditions, just before an overload occurs. After the end of a given

ON-pulse (say ‘ton1’), the LM2647 starts sampling the current in the low-side FET. This is the OFF-duration

called ‘toff1’ in this analysis. Therefore, if an overcurrent condition is detected during this OFF-duration ‘toff1’, the

controller will decide to omit the next ON-pulse (which would have occurred during the duration ‘ton2’). This is

done by setting an internal ‘overcurrent latch’ which will keep HDRV low. The LDRV will now not only stay high

during the present OFF-duration (‘toff1’) but during the duration of the next (omitted) ON-pulse (‘ton2’), and then

as expected also during the succeeding OFF-duration (‘toff2’). But the ‘overcurrent latch’ is reset at the very start

of the next OFF-duration ‘toff2’. Therefore if the overcurrent condition persists, it can be recognized during ‘toff2’

and a decision to skip the next ON-pulse (duration ‘ton3’) can be taken. Finally, several ON-pulses may get

skipped until the current in the lower FET falls below the current limit threshold.

Note that about 150ns after LDRV first goes high (start of low-side conduction), the current monitoring starts.

Therefore the peak current seen by the current limit detector is almost the same as the peak inductor current.

To set the value of the current limiting resistor (‘RLIM’, between ILIM pin and SW pin), the function of the ILIM

pin must be understood. Refer to Figure 19 to see how the voltage on the ILIM pin changes as current ramps up.

For this analysis note that the worst case has been taken here by using the minimum possible value of the

current sourced (I_ILIM, see Electrical Characteristics table). Also, the maximum value of the ‘hot’ Rds of the lower

FET should be used. For example if the chosen low-side FET is the Si4420DY from Vishay, the typical Rds at

room temperature is 10mΩ (but this is not the value to be used here). The MAX is the relevant number which is

13mΩ. Now applying the thumbrule that at 100°C the Rds goes up typically 1.4 times (for 30V FETs), the Rds to

be used in the actual current limit calculation is 1.4*13mΩ=18.2mΩ. Therefore using 46µA for I_ILIM(see Electrical

Characteristics table) and Rds = 18.2mΩ here will provide the lowest value of current limit (considering

tolerances and temperature for a chosen RLIM resistor). This current limit must obviously be higher than the

actual peak current in the converter under normal operation to ensure that full rated power can be delivered

under all conditions by the converter without ‘inadvertently’ hitting the worst case (lowest value) set current limit.

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GND

I_lowside

I_ILIM= 46 mA

VLIM

CURRENT

VOLTAGE

TO

DETECTOR

v

}

DV

RLIM

}

SW

-v

VILIM = -v + DV (= 0 at CLIM)

v = Rds x I_lowside

DV = RLIM x I_ILIM

Figure 19. Understanding Current Sensing

The detector sets the overcurrent latch as soon the voltage on the ILIM pin crosses below zero. Therefore the

basic design equation for calculating RLIM is:

ΔV = v

(1)

(see Figure 19)

At the point where current limiting occurs (peak inductor current becomes equal to current limit) the resistor for

setting the current limit can be calculated.

But what (peak) current limit value should actually be set? This depends on two factors:

a) There is a natural steady state peak current in the inductor with the converter delivering maximum rated load.

This should be calculated at VIN_MAX(the maximum of the input voltage range):

b) Over and above this steady state value we need to provide an ‘overload margin’. This margin will depend on

the step loads likely to be seen in the application and the response expected.

The equation for calculating the steady state peak current is:

r

I_peak= I_ox (1+ )

2

(2)

where ‘r’ is the current ripple ratio (refer to Application Note AN-1197 for a detailed understanding of how ‘r’

affects all the power components). ‘r’ is given by:

V_o

x (1-D) x 10⁶

r =

I_ox L x f

(3)

where L is in µH, f is in Hz.

Example: Let VIN range from 5.5V to 28V, Vo=5V, Io=3A, L=10µH, f=300kHz. What is the peak current under

normal operation?

Only the highest input voltage must be used for any peak current calculation. At VIN_MAXthe duty cycle is

D=Vo/Vin=5/28=18%. So

5

x (1-0.18) x 10⁶= 0.45

r =

3 x 10 x 300000

(4)

NOTE

In general, as discussed in AN-1197, the optimum value of ‘r’ is between 0.3 to 0.5. Large

inductances (higher than ‘optimum’) may be selected if the output voltage ripple needs to

be decreased but it is not desirable to achieve this by adding more (expensive?) output

specialty caps.

The peak current under normal operation is

r

I_peak= I_ox (1+ )

2

(5)

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0.45

2

I_peak= 3 x (1+

)

= 3.7A

(6)

Conclusions: In this example the peak inductor current under normal operation is 3.7A. Usually it is necessary

only to set the current limit about 20% higher than the peak value. This ‘overload margin’ helps greatly in

handling sudden load changes. A 20% margin would have required the current limit to be set at 3.7*120%=4.44A

(for a steady state peak of 3.7A). Therefore RLIM would need to be

Rds₁₀₀x I_CLIM

RLIM =

46m

(7)

18.2m x 4.44

RLIM =

= 1.76k

46m

(8)

A standard resistor value of 1.78k can be chosen in the example. However, a larger overload margin than the

chosen 20% (say 40%) is recommended for obtaining good dynamic response if the load could suddenly change

from extremely low values (zero to a few mA) right up to maximum load current. In this case, it would require

I_CLIM=3.7*140%=5.2A, requiring RLIM to be 18.2m*5.2/46µ=2.05k (available as a standard value).

Note that excessively high current limits (large RLIM values) will generate severe stresses in the FETs during

abnormal load condition (like a shorted output for example). These peak currents will be even higher if the

inductor saturates sharply. The designer must evaluate the actual application for the expected and actual step

loads so as to select RLIM more optimally. Then it should be decided how much overload margin is really

required, and RLIM selected accordingly. The equations to do this are provided in this section, but the judgement

must remain with the designer, as it depends on the specific application on hand.

Repeating the calculation for a 10µH inductor for a 3.3V/3A rated output, and any low side equivalent FET (with

the same Rds as Si4420DY) we get the following requirement:

For 20% overload margin, select current limit resistor to be 1.69k

For 40% overload margin, select current limit resistor to be 1.96k

Note that if the lower FET Rds is different from the one used in the example above, the current limit resistor

RLIM must be recalculated according the new Rds.

For the evaluation board the selected FET was a dual pack Si4828DY. Its worst case hot Rds is 24.5mΩ. Setting

current limit as 5.5A, the estimated current limit resistor is 5.5 x 24.5 / 46 = 2.93kΩ. A standard value of 2.94kΩ

was chosen for the Bill of Materials.

INDUCTOR and OUTPUT CAPACITOR

The designer is again referred to AN-1197 for the equations required here. In general, ‘r’ is the key parameter

and once that is chosen, the inductance can be calculated. The design table in the referenced Application Note

uses V_Das the drop across the diode in an asynchronous configuration. Also, V_SWis the drop across the Switch

(upper FET). In the case of the LM2647 a reasonable approximation is to set V_D= V_SW= 0 in the design table

available in AN-1197. Then the table can be used easily for selection of the inductor and output capacitor. A step

by step example is also provided for a general buck regulator in the Application Note AN-1207.

Only in the case of the input capacitor, the situation may be different as is explained next.

INPUT CAPACITOR

In a typical single-channel buck regulator, the input capacitor provides most of the pulsed current waveform

demanded by the Switch. However the DC (average) value of the current through a capacitor in steady state

must be zero. Otherwise, the capacitor would start accumulating charge every cycle, and that would clearly not

represent a ‘steady state’ by definition.

Now for the LM2647, there are two ways of calculating and meeting the input capacitance requirement. One way

is to use separate input capacitors for each channel (as in the Evaluation board). The other possibility is to

combine them into a single component. There are advantages and disadvantages to each approach.

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By keeping separate input capacitors the possibility of interaction between the two channels is reduced, and the

layout is a little more forgiving. But two components would require more board space and could also add to the

cost. Though in general, there could also be a situation where the cost of a single component is equal to (or even

exceeds) the combined cost of two separate capacitors. The reason cost can be surely reduced when using one

input capacitor in the LM2647 is because the two channels run 180° out of phase (interleaved switching). It can

be shown that this dramatically reduces the ripple current requirement at the input. See Figure 20 for typical

waveforms to understand how this happens. Remember that ‘frequency’ does not (directly) enter into any

computations of RMS values, so the use of interleaved switching is clearly going to produce a lower RMS value

as can be guessed by eyeballing the waveforms shown in Figure 20.

CH1

I_SW

CH1

I_SW

CH2

I_SW

CH2

I_SW

INPUT

CAP

INPUT

CAP

ONE INPUT CAP

INTERLEAVED

SWITCHING

ONE INPUT CAP

NON INTERLEAVED

SWITCHING

Figure 20. Switch and Input Capacitor Currents

The case of a single input capacitor supplying two channels running out of phase is now discussed in detail and

it shows how to formally calculate the input RMS current capability required. The example represents a very

general case in terms of the output voltages simply to highlight the various possible applications of the LM2647

other than its primary intended application. One of the most important questions to answer here is: what input

voltage really gives the worst possible (highest) input RMS current? This information is required to size the

capacitor correctly.

Example: Consider two channels running at 5V@3A and 3.3V@3A. What is the worst case input capacitor RMS

current if the input varies from 10V to 28V?

Step1: Call the output with the higher voltage as Vo1 and the other as Vo2. Then find the ratio ‘y’ as shown

below

Vo2 3.3

= 0.66

=

y =

5

Vo1

(9)

y is clearly going to be equal to or less than 1 by definition (since Vo2 ≤ Vo1). This step is required for using the

equation presented in the next step.

Step2: The equation for the input current has been derived and it reveals that the worst-case occurs when the

duty cycle of the first channel is

Io1²+ (y x Io2²)

2 x {Io1 + (Io2 x y)}²

D1 =

(10)

where ‘y’ has been defined in Step 1. So

3²+ (0.66 x 3²)

2 x {3 + (3 x 0.66)}²

= 0.3

D1 =

(11)

(12)

Therefore the appropriate input voltage to calculate the worst case RMS input current is

Vo1

D1

5

= 16.7V

V_IN

=

0.3

Step3: Calculate the duty cycle of the other channel when this happens

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Vo2

V_IN

0.2

=

D2 =

(13)

Step4: Calculate input capacitor RMS current by using the known equation

I_IN²= (Io1²• D1) + (Io2²• D2) - [Iol • D1+Io2 • D2]²

(14)

(15)

I_IN²= (3²• 0.3) + (3²• 0.2) - [(3 • 0.3) + (3 • 0.2)]²

Solving

I_IN= 1.5A

(16)

Step5: But what is really the worst case??

It may have simply concluded at this point that "the rating of the input capacitor must be greater or equal than

1.5A, otherwise the life/reliability of the capacitor may be affected severely etc.". And that is true but only under

the single-point load conditions used for the calculation. It will now be seen that the worst case may still have

gone unrecognized. What if maximum load currents are not being drawn simultaneously as was assumed in our

example? It can be shown that the capacitor could actually see higher currents than calculated in Step 4.

Suppose one channel was completely unloaded. So in effect there is only a single output of 5V@3A. The

equation for the RMS current through the input capacitor is then

I_IN= Io D(1-D)

(17)

The function D(1-D) has a maxima at D = 0.5. This would correspond to an input voltage of 5V/0.5 = 10V. And

the input capacitor current at this worst case input voltage would be

I_IN= 3 0.5 (1-0.5)

= 3 x 0.5 = 1.5A

(18)

It is just a coincidence in this application that in both cases (above and at the end of Step 4) we have calculated

the same RMS current rating for the capacitor. In general, Step 4 can certainly yield smaller values than those for

a single channel, and this may mislead us into an improper selection of the input capacitor. It must be

remembered that Step 4 is not necessarily the worst case. We must always take the higher of the two values so

calculated.

Incidentally, the above method for a single channel is also the method to be used to calculate the capacitor rating

when the LM2647 is formally used for single channel operation, or if both channels are being used but separate

input capacitors are being allocated for each channel.

In all cases the input capacitors must be positioned physically close to their respective stages. But if separate

input capacitors are being used for each channel, the input traces to the two inputs must be long and thin so as

to introduce a measure of high frquency decoupling between the now separated stages.

The designer may ask, what is the use of interleaved switching if the result of the interleaved calculation in Step

4 may not even be used in our particular example? Interleaved switching certainly reduces cost because if the

calculation had been carried out for two non-interleaved channels (switching in phase), both delivering maximum

load, the capacitor RMS current would have been much higher.

Note that the equations used in the above sections apply only if the duty cycles of both channels are less than or

equal to 50% (and there is therefore no overlap in the current waveforms). The equations for overlapping

waveforms are out of scope here.

MOSFETs

The selection of the MOSFETS should be done carefully to maximize both efficiency and reliability together.

There is a different set of criteria for selecting the upper FET and lower FET. It will also be seen that using very

fast FETs without deliberate thought, may seem to improve efficiency dramatically on one prototype board but

can impair efficiency on another apparently ‘identical' board, specially at light loads. Therefore, the quest for

improved efficiency must be weighed against the possible penalty for doing this without deeper understanding of

the nuances of synchronous switching buck stages in general. The criteria for selection are briefly:

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a) The upper FET is chosen basically for high switching speed because in a typical synchronous buck regulator

only the upper FET sees the V-I crossover losses (at turn-ON and at turn-OFF). So to maximize efficiency, high

switching speed is certainly needed in this position. This FET position has typically very low conduction losses,

especially in a power supply for mobile applications since the duty cycle is very low. So the Rds is not of much

direct concern here. A possible choice of FET for the upper position on the Evaluation board is the Si4800DY

from Vishay (www.vishay.com). The threshold voltage (MIN value) of a FET in this position can be 0.8V but 1.0V

is preferable. Note however that if the upper FET is chosen so that it switches too fast, it can induce a shoot-

through (called a CdV/dt turn-on of the lower FET) whenever the upper FET turns on hard. Therefore, Q_Gof the

upper FET should not be much less than 8nC.

b) The lower FET sees no V-I crossover loss in principle (under most situations). Also, since it can conduct for

the complete OFF-time, its Rds becomes important, especially at low duty cycles. This FET is therefore chosen

basically for its low Rds, not necessarily speed. A high C_OSSfor this FET position also helps, by reducing the

possibility of CdV/dt turn-on of this FET, by snubbing the rising edge of voltage applied on the lower FET when

the upper FET turns ON. Note that too high a C_OSSvalue will degrade efficiency. An acceptable compromise

figure for C_OSSof the lower FET is 350-800pF. A possible choice of FET for this position is the Si4420DY from

Vishay. The C_OSSof this FET is about 700pF at 24V. The threshold voltage for the lower FET position must also

be 1V or slightly higher. Too high a threshold will prevent the FET from turning ON fully, and too low a value will

increase the likelihood of a CdV/dt turn-on. Also note that one of the factors which can provoke a spurious turn-

on is layout. In particular, the source lead/trace of a given FET must be kept short and the copper area around it

large to reduce inductive spikes during transitions. Gate trace lengths must also be kept short.

Note that the threshold voltage of a FET should have both MIN and MAX limits as per its datasheet. Since it is

important that the FET turn on fully, ensure that the threshold voltage is ensured to be below 3V. Contact the

FET vendor if necessary. If the threshold voltage is too high, foldback might result upon hitiing current limit. This

will result in failure of the output to recover after an overload condition.

EFFICIENCY ESTIMATE

A sample calculation follows based on the low cost FETs used on the Evaluation Board. The device is the

Si4828DY from Vishay.

The extension '_u' stands for the upper FET (half Si4828DY), and '_l' for the lower FET (half Si4828DY). The

general equation is first stated and then the numerical result is quoted (in bold). The case is for V_IN=20V, Vo=5V,

Io=3A. The frequency is set to 300kHz. Note that efficiency estimates are usually based on typical values.

Therefore, in the calculations below the typical value of the gate charge Q_Gis used. For the Si4828DY the typical

values as declared in its datasheet (available at the time of writing this section) are Q_G(upper) = 8nC, Q_G(lower)

= 23nC, Rds(upper) = 24mΩ, Rds(lower) = 14.5mΩ

FET Conduction losses

Vo

Pcond_u = Io²x rds_u x

V_IN

(19)

Pcond_u = 54mW

Vo

V_IN

Pcond_I = Io²x rds_l x (1-

)

(20)

Pcond_l = 98mW

FET Switching Losses

The transition times must first be determined. A simplified equation available in related literature is:

Q_G

~

4.6 x R x

t_r= t_f

P

V_P

(21)

This equation is applied to our case by setting the pulse amplitude Vp to 5V. Suppose the output impedances of

the IC are (in ohms):

Rpon_u = 7

Rpoff_u = 2

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Rpon_l = 7

Rpoff_l = 1

Therefore transition times are

Qg_u

ton_u = 4.6 x Rpon_u x

Vp

(22)

(23)

ton_u = 51.5ns

Qg_u

toff_u = 4.6 x Rpoff_u x

Vp

toff_u = 15ns

Qg_l

ton_l = 4.6 x Rpon_l x

Vp

(24)

(25)

ton_l = 148ns

Qg_l

toff_l = 4.6 x Rpoff_l x

Vp

toff_l = 21ns

The switching loss for any V-I crossover when driving an inductive load is in general

Pcross = 1/2 x V x I x tcross x freq

I (or V)

V (or I)

repetition rate = freq

tcross

Figure 21. Crossover (turn-on or turn-off)

The V-I crossover losses (exist only in upper FET) are:

Pswon_u = 1/2 • VIN • Io • f • ton_u

(26)

(27)

Pswon_u = 464mW

Pswoff_u = 1/2 • VIN • Io • f • toff_u

Pswoff_u = 132mW

There is another loss term associated with charging C_OSSevery cycle, then dumping it into the FET before the

next charge cycle. This applies to both upper and lower FETs.

Pcoss_u = 1/2 • C_{OSS_u}• VIN²• f

Pcoss_l = 1/2 • C_{OSS_l}• VIN²• f

(28)

(29)

From the datasheets of the chosen FETs, 'Coss' are respectively about:

C_{OSS_u}= 250pF

C_{OSS_l}= 500pF

So

Pcoss_u = 15mW

Pcoss_I = 30mW

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Summing up,

Psw_u = 464+132+15=611mW

Psw_I = 30mW

Controller Losses

In addition to the losses in the FETs, there is another loss term associated with the switching, and this is

dissipated in the controller. The LM2647 has to pump in current pulses at each transition to turn-ON or turn-OFF

the FETs. Several simplified or more complicated equations exist for calculat-ing this, but this is most easily

deduced by simply turning to the measured consumption (see Electrical Characteristics table). The current into

the V5 pin is I_{Q_V5}and reflects the driver consumption. This can be as high as 1.5mA (measured at 300kHz). Let

us also include the current into the control sections (VDD pin), which can be as high as 4mA. The total controller

consumption is therefore

P_IC= (I_{Q_V5}+ I_{Q_VDD}) x 5

(30)

P_IC= 28mW

Inductor Losses

The DC resistance (‘DCR') of the chosen inductor is typically is 26mΩ. The DC loss is therefore DCR*Io². The

core losses typically add 10% more to this. Therefore our estimate of total inductor loss is

Pind=1.1 x (DCR x Io²)

(31)

Pind = 257mW

Capacitor Losses

The output capacitor of a typical buck regulator has very low ripple current going through it. So its loss term can

be ignored. The input capacitor however provides the sharp pulses of current for the Switch, and therefore the

RMS current through it can be fairly high. But the dissipation can still be negligible if the ESR is very low. This is

the situation if the input capacitors are monolithic ceramic capacitors as in the Evaluation board (if Tantalum or

Aluminum electrolytic capacitors are used at the input, their dissipation must be accounted for here). The final

efficiency/loss terms are provided in Table 1.

Table 1. Losses and Efficiency

Upper

54

Lower

98

Pcond (mW)

Psw (mW)

611

665

30

PFET (mW)

P_IC(mW)

128

28

257

Pind (mW)

Ptotal (mW)

Pout (=VoxIo) (mW)

1078

15000

93%

P_OUT

Eff =

P_OUT+ P_TOTAL

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Typical efficiency curves for different input voltages are available under Typical Performance Characteristics.

LAYOUT GUIDELINES

For a deeper understanding of Buck converters and the ‘critical traces' please see Application Note AN-1229.

Figure 22 is based on such an understanding of the critical sections and also the pin functions of the LM2647.

Refer to the Typical Application circuit and the LM2647 TSSOP to understand the layout suggestions more

thoroughly. The components shown in Figure 22 are most critical and must be placed close to the device and

connected onto a ground island on the component side. Several vias can then connect to the ground plane at the

locations indicated. The FETs are positioned close to the controller and are also very close to each other to

minimize inductances.

After the critical components are placed, the resistor to the frequency adjust pin (R19) must also be placed close

to the IC connecting to SGND. This will reduce noise pickup and jitter.

The feedback trace can also pick up noise and it must be routed away from sources of noise/EMI, particularly the

FETs and inductors.

Enough copper area must be left around the FETs for thermal dissipation. More details on this are also provided

in AN-1229.

Note that the current limit detector circuit compares the voltage on the ILIM pin with respect to the PGND pin.

Therefore, if the power ground is noisy it can lead to erroneous triggering of the current limit detector. This will

manifest itself as an inability to meet the load requirement despite oversizing the current limit resistor. It can also

lead to failure of the output to recover after encountering an overload condition. Therefore, it is strongly

recommended that a solid ground plane be created as the first internal plane right below the component

side.Several vias should be generously placed to connect the ground nodes of the component layer to this

ground plane.

Figure 22. Critical Component placement (TSSOP)

SETTING OUTPUT VOLTAGE

From the Typical Application circuit on Page 1, it can be seen that R15 and R16 are used to set V_O2whereas

R21 and R22 set V_O1. For either channel, calling the upper resistor (connected to one end of the droop resistor)

R_Uand the lower resistor (connected to ground) R_Lthe following equation is applicable.

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R_U+ R_L

R_L

x V_FB

Vo =

(32)

Therefore from the Bill of Material:

For channel #1 (V_O1= 5V),

R_U= R21 = 43.2k

R_L= R22 = 5.9k

For channel #1 (V_O2= 3.3V),

R_U= R15 = 43.2k

R_L= R16 = 9.53k

So

(43.2 + 5.9) x 0.6

= 4.99V

V_O1

=

5.9

(33)

(34)

(43.2 + 9.53) x 0.6

= 3.32V

V_O2

=

9.53

This is as per the requirement of the primary end-application. Other output voltage values are possible by

adjusting the resistor ratios (but note that there are maximum duty cycle constraints as stated in Electrical

Characteristics table) which will limit the range of output voltages achievable. Note that the upper resistor is

involved in fixing the gain of the error amplifier, and therefore its value has been set to an ‘optimum' value of

43.2k for both channels. This value helps in achieving good step response and ensuring stability. Therefore, in

general, only the lower resistor should be adjusted. However the more experienced designer can judiciously use

the open-loop gain information provided in the next section, to change both upper and lower resistor values if

required.

MODULATOR GAIN/COMPENSATION

The modulator gain is plotted out for various typical values of components in Typical Performance

characteristics. The curves were based on the following information. The plant/modulator gain ‘G' is:

Ro x Vin x (s x C x esr +1)

1

x

G =

A₁x s²+ A₂x s +A₃

V_RAMP

where

•

Ro is the load resistance

s = jω

A1 = LC*(Ro + esr)

A2 = {L + R_PC(Ro + esr) + Ro*esr*C}

A3 = R_o+ R_P

(35)

Here esr is the Equivalent Series Resistance of the output capacitor, Ro is the load resistance, C is the output

capacitance, L is the inductance, and R_Pis the resistance of the power stage (Rds+ DCR etc, typically about

40mΩ). V_RAMPis about 1.6V for the LM2647. The unity gain bandwidth of the error amplifier is taken as 6.5MHz.

Let us assume Type 3 compensation Figure 23.

C1

V_OUT

R2

C2

R3

C3

R1

fb

-

+

COMP

Rbias

V_REF

(0.6V)

Figure 23. Type 3 Compensation

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The design procedure is summarized in Summary of Compensation Design Procedure.

Summary of Compensation Design Procedure

Table 2.

fp1

fz1

fz2

fp2

fp3

Type 3

(C1)

R2, C2

R1sR3, C3

R2, C1sC2

C3, R3

Table 3.

C1

C2

C3

R2

R3

1

fz

1

fp2

- C1

-

2p x R1 fz2

R1 x A

2p x C3 x fp2

2p x C2 x fz1

R1 x A x fp3

Table 4.

fp1

(0)

fz1

fdp

fz2

fdb

fp2

fp3

fesr

fsw/2

A short explanation on Table 3 follows. For example from the table it can be seen that the second zero is created

by the series combination of R1 and R3 resonating with C3. So the frequency of this zero is at 1/2π(R1+R3)*C3.

The solution for calculating the component values follows. For example, C3 is set as

1

fp2

-

C3 =

2p x R1 fz2

(36)

But where should the designer position the poles and zeroes of the error amplifier, taking into account the

modulator gain, so as to achieve good closed loop characteristics? A typical scenario is also provided in Table 4.

For example it suggests that both the first and second zeroes should be positioned at the point where double

pole (fdb) of the output LC filter is. This double pole is known to occur at about.

1

fdp =

2p

LC

(37)

Similarly, the esr zero occurs at fesr which is at

1

fesr =

2p x esr x C

(38)

where C is the output capacitance

Thus all the components can be calculated easily

EVALUATION BOARD DETAILS

The Bill of Materials is now provided for the LM2647 Evaluation board. The schematic is the Typical Application

circuit. See LAYOUT GUIDELINES for more guidance on preferred layout practices and also refer to Application

Note AN-1229. Note that a dual FET pack has been chosen for the Evaluation Board.

The Evaluation board has two outputs V_O1= 5V and V_O2= 3.3V as discussed under SETTING OUTPUT

VOLTAGE section. The rated load on each output is 2A continuous, and 3A peak. A minimum load of 0.1mA

should be maintained on each output in SKIP mode, to ensure regulation.

30

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SNVS210F –JUNE 2003–REVISED APRIL 2013

V

IN

5V

D2

V

IN

Q1

Q3

BOOT2

HDRV2

BOOT1

HDRV1

+

C1

C32

C28

C30

L2

L1

SW2

Vo2

SW1

Vo1

R7

R8

Q4

ILIM2

Q2

ILIM1

C23

C22

C25

C26

+

LDRV2

PGND2

LDRV1

PGND1

SENSE2

COMP2

SENSE1

COMP1

SS1

R6

C5

C6

R9

C14

R15

C17

C7

C4

SS2

FB2

FB1

R14

R20

R18

R17

R21

R23

FPWM

VRON

FPWM

EN

PGOOD

FREQ

C16

R19

V

DD

V5

SGND

R1

C31

R16

R22

C15

C29

Figure 24. Typical Application (Expanded View)

Table 5. Bill of Materials for Figure 24

Designator

C1

Function

Descrtiption

10µF, 25V, X7R

Type

1812

1206

Vendor

TDK

Cin (Ch #2)*

C4

Comp cap (across RC, Ch #2)

Comp cap (series with R, Ch #2)

Comp cap (series with R, Ch #1)

Comp cap (across RC, Ch #1)

Cff (Ch #2)

15pF, 6.3V, X7R

680pF, 6.3V, X7R

15pF, 6.3V, X7R

680pF, 6.3V, X7R

0.1µF, 6.3V/25V, X7R

680pF, 6.3V, X7R

330µF,10V, Ta

Vishay

TDK

C5

C6

C7

C14

C15

C16

C17

C22

C23

C25

C26

C28

C29

C30

C31

C32

R1

Soft-start cap (Ch #2)

Soft-start cap (Ch #1)

Cff (Ch #1)

Cout1 (Ch #2) (optional)

Cout2 (Ch #2)

593 Series

1206

330µF,10V, Ta

Cout1 (Ch #1)

330µF,10V, Ta

Cout2 (Ch #1) (optional)

Cboot (Ch #2)

330µF,10V, Ta

0.1µF, 6.3V, X7R

2.2µF, 25V, X7R

10µF, 25V, X7R

10Ω, 5%

V5 decoupling

1206

Cboot (Ch #1)

1206

VDD decoupling

1206

Cin (Ch #1)*

1812

V5 to VDD series pass

1812

Vishay

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Table 5. Bill of Materials for Figure 24 (continued)

Designator

R6

Function

Comp res (series with C, Ch #2)

RLIM (Ch #2)

Descrtiption

Type

1206

Vendor

Vishay

Sumida

Onsemi

Vishay

TI

57.6k, 1%

2.94k, 1%

57.6k, 1%

12.7k, 1%

43.2k, 1%

9.53k, 1%

12.7k, 1%

22.1k, 1%

12.7k, 1%

43.2k, 1%

5.9k, 1%

R7

1206

R8

RLIM (Ch #1)

1206

R9

Comp res (series with C, Ch #1)

Rff (Ch #2)

1206

R14

R15

R16

R17

R18

R19

R20

R21

R22

R23

L1

1206

Res divider, upper (Ch #2)

Res divider, lower (Ch #2)

Enable pullup

1206

FPWM pullup

1206

Freq Adjust

1206

PGOOD pullup

1206

Res divider, upper (Ch #1)

Res divider, lower (Ch #1)

Rff (Ch #1)

1206

12.7k, 1%

1206

Inductor (Ch #1)

10µH,4.4A

CDRH104R

SOT-23

SO-8

L2

Inductor (Ch #2)

10µH,4.4A

D1

Bootstrap diode (Ch #2)

Bootstrap diode (Ch #1)

Upper FET (Ch #1)

Lower FET (Ch #1)

Upper FET (Ch #2)

Lower FET (Ch #2)

Controller

BAT54LT1

D2

BAT54LT1

Q1

Si4828DY (half)

LM2647

Q2

SO-8

Q3

SO-8

Q4

SO-8

U1

TSSOP

DIP

S1

Dual SPDT switch (see Typical Application)

CKN1276-ND

Grayhill

PCB Layout Diagrams

Figure 25. Top Overlay

32

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SNVS210F –JUNE 2003–REVISED APRIL 2013

Figure 26. Top Layer

Figure 27. Internal Plane 1 (GND)

Figure 28. Internal Plane 2

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Figure 29. Bottom Layer

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SNVS210F –JUNE 2003–REVISED APRIL 2013

REVISION HISTORY

Changes from Revision E (April 2013) to Revision F

Page

•

Changed layout of National Data Sheet to TI format .......................................................................................................... 34

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PACKAGE MATERIALS INFORMATION

www.ti.com

23-Sep-2013

TAPE AND REEL INFORMATION

*All dimensions are nominal

Device

Package Package Pins

Type Drawing

SPQ

Reel

A0

B0

K0

P1

W

Pin1

Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant

(mm) W1 (mm)

LM2647LQ/NOPB

LM2647LQX/NOPB

LM2647MTCX/NOPB

WQFN

TSSOP

NJB

PW

28

1000

4500

2500

178.0

330.0

12.4

16.4

5.3

6.8

5.3

1.3

1.6

8.0

12.0

16.0

Q1

10.2

Pack Materials-Page 1

PACKAGE MATERIALS INFORMATION

www.ti.com

23-Sep-2013

*All dimensions are nominal

Device

Package Type Package Drawing Pins

SPQ

Length (mm) Width (mm) Height (mm)

LM2647LQ/NOPB

LM2647LQX/NOPB

LM2647MTCX/NOPB

WQFN

TSSOP

NJB

PW

28

1000

4500

2500

210.0

367.0

185.0

367.0

35.0

38.0

Pack Materials-Page 2

MECHANICAL DATA

NJB0028A

LQA28A (REV B)

www.ti.com

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TI warrants performance of its components to the specifications applicable at the time of sale, in accordance with the warranty in TI’s terms

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是否无铅：	含铅	是否Rohs认证：	符合
生命周期：	Obsolete	包装说明：	HVQCCN, LCC28,.2SQ,20
Reach Compliance Code：	compliant	ECCN代码：	EAR99
HTS代码：	8542.39.00.01	风险等级：	5.72
其他特性：	OUTPUT VOLTAGE ADJUSTABLE DOWN TO 0.6V	模拟集成电路 - 其他类型：	DUAL SWITCHING CONTROLLER
控制模式：	VOLTAGE-MODE	控制技术：	PULSE WIDTH MODULATION
最大输入电压：	28 V	最小输入电压：	5.5 V
标称输入电压：	15 V	JESD-30 代码：	S-XQCC-N28
JESD-609代码：	e3	长度：	5 mm
湿度敏感等级：	1	功能数量：	1
端子数量：	28	最高工作温度：	125 °C
最低工作温度：	-5 °C	封装主体材料：	UNSPECIFIED
封装代码：	HVQCCN	封装等效代码：	LCC28,.2SQ,20
封装形状：	SQUARE	封装形式：	CHIP CARRIER, HEAT SINK/SLUG, VERY THIN PROFILE
峰值回流温度（摄氏度）：	260	认证状态：	Not Qualified
座面最大高度：	0.8 mm	子类别：	Switching Regulator or Controllers
表面贴装：	YES	切换器配置：	BUCK
最大切换频率：	500 kHz	温度等级：	OTHER
端子面层：	Matte Tin (Sn)	端子形式：	NO LEAD
端子节距：	0.5 mm	端子位置：	QUAD
处于峰值回流温度下的最长时间：	NOT SPECIFIED	宽度：	5 mm
Base Number Matches：	1

型号	制造商	描述	替代类型	文档
LM2647LQ/NOPB	TI	LM2647 Dual Synchronous Buck Regulator Controller	类似代替

型号	制造商	描述	价格	文档
LM2647MTC	NSC	Dual Synchronous Buck Regulator Controller	获取价格
LM2647MTC	TI	DUAL SWITCHING CONTROLLER, 345kHz SWITCHING FREQ-MAX, PDSO28, TSSOP-28	获取价格
LM2647MTC/NOPB	TI	LM2647 Dual Synchronous Buck Regulator Controller	获取价格
LM2647MTCX	NSC	Dual Synchronous Buck Regulator Controller	获取价格
LM2647MTCX	TI	DUAL SWITCHING CONTROLLER, 345kHz SWITCHING FREQ-MAX, PDSO28, TSSOP-28	获取价格
LM2647MTCX/NOPB	TI	LM2647 Dual Synchronous Buck Regulator Controller	获取价格
LM2647_15	TI	LM2647 Dual Synchronous Buck Regulator Controller	获取价格
LM2648	NSC	Two-Phase, Synchronous Step-Down 3-Channel Switching Regulator Controller	获取价格
LM26480	NSC	Externally Programmable Dual High-Current Step-Down DC/DC and Dual Linear Regulators	获取价格
LM26480	TI	Compatible with Advanced Applications Processors and FPGAs, 2 LDOs for Powering Internal Processor Functions and I/Os	获取价格

LM2647LQX/NOPB

LM2647LQX/NOPB 概述

LM2647LQX/NOPB 规格参数

LM2647LQX/NOPB 数据手册

LM2647LQX/NOPB CAD模型

LM2647LQX/NOPB 替代型号

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