LM27313XMF/NOPB [TI]
具有 30V 内部 FET 开关并采用 SOT-23 封装的 1.6MHz 升压转换器 | DBV | 5 | -40 to 125;
| 型号： | LM27313XMF/NOPB |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | 具有 30V 内部 FET 开关并采用 SOT-23 封装的 1.6MHz 升压转换器 | DBV | 5 | -40 to 125 升压转换器 开关 控制器 开关式稳压器 开关式控制器 光电二极管 电源电路 开关式稳压器或控制器 |
| 文件： | 总20页 (文件大小：1013K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LM27313
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SNVS487D –DECEMBER 2006–REVISED APRIL 2013
LM27313/LM27313-Q1
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1
FEATURES
DESCRIPTION
The LM27313 switching regulator is a current-mode
boost converter with a fixed operating frequency of
1.6 MHz.
2
•
LM27313-Q1 is an Automotive Grade Product
that is AEC-Q100 Grade 1 Qualified (-40°C to
+125°C Operating Junction Temperature)
The use of the SOT-23 package, made possible by
the minimal losses of the 800 mA switch, and small
inductors and capacitors result in extremely high
power density. The 30V internal switch makes these
solutions perfect for boosting to voltages of 5V to
28V.
•
•
•
•
•
•
•
•
•
•
30V DMOS FET Switch
1.6 MHz Switching Frequency
Low RDS(ON) DMOS FET
Switch Current up to 800 mA
Wide Input Voltage Range (2.7V–14V)
Low Shutdown Current (<1 µA)
5-Lead SOT-23 Package
This part has a logic-level shutdown pin that can be
used to reduce quiescent current and extend battery
life.
Uses Tiny Capacitors and Inductors
Cycle-by-Cycle Current Limiting
Internally Compensated
Protection is provided through cycle-by-cycle current
limiting and thermal shutdown. Internal compensation
simplifies design and reduces component count.
APPLICATIONS
•
•
•
•
•
White LED Current Source
PDA’s and Palm-Top Computers
Digital Cameras
Portable Phones, Games and Media Players
GPS Devices
Typical Application Circuits
D1
MBR0520
L1/10 mH
5 V
IN
U1
SW
LM27313
FB
V
IN
12V
OUT
R3
51k
R1/117k
SHDN
GND
260 mA
(TYP)
SHDN
GND
C1
2.2 mF
CF
220 pF
C2
4.7 mF
R2
13.3k
D1
MBR0530
L1/10 mH
5 V
IN
U1
SW
V
IN
20V
OUT
130 mA
(TYP)
R3
51k
R1/205k
LM27313
SHDN
GND
FB
SHDN
GND
C1
2.2 mF
CF
120 pF
R2
13.3k
C2
4.7 mF
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
2
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2006–2013, Texas Instruments Incorporated
LM27313
SNVS487D –DECEMBER 2006–REVISED APRIL 2013
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Connection Diagram
Figure 1. 5-Lead SOT-23 Package – Top View
See Package Number DBV
PIN DESCRIPTIONS
Pin
1
Name
SW
Function
Drain of the internal FET switch.
Analog and power ground.
2
GND
FB
3
Feedback point that connects to external resistive divider to set VOUT
Shutdown control input. Connect to VIN if this feature is not used.
Analog and power input.
.
4
SHDN
VIN
5
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings(1)(2)
Storage Temperature Range
Lead Temp. (Soldering, 5 sec.)
Power Dissipation(3)
−65°C to +150°C
300°C
Internally Limited
−0.4V to +6V
−0.4V to +30V
−0.4V to +14.5V
−0.4V to +14.5V
±2 kV
FB Pin Voltage
SW Pin Voltage
Input Supply Voltage
Shutdown Input Voltage
ESD Rating(4)
(Survival)
Human Body Model
(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is to be functional, but does not ensure specific limits. For ensured specifications and conditions see the Electrical
Characteristic table.
(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
(3) The maximum power dissipation which can be safely dissipated for any application is a function of the maximum junction temperature,
TJ(MAX) = 125°C, the junction-to-ambient thermal resistance for the SOT-23 package, θJ-A = 265°C/W, and the ambient temperature, TA.
The maximum allowable power dissipation at any ambient temperature for designs using this device can be calculated using the
formula:
If power dissipation exceeds the maximum specified above, the internal thermal protection circuitry will protect the device by reducing
the output voltage as required to maintain a safe junction temperature.
(4) The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Test method is per JESD22-A114.
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Operating Ratings
VIN
2.7V to 14V
30V
VSW(MAX)
VSHDN
0V to VIN
(1)
Junction Temperature, TJ
-40°C to 125°C
265°C/W
θJ-A (SOT-23-5)
(1) The maximum power dissipation which can be safely dissipated for any application is a function of the maximum junction temperature,
TJ(MAX) = 125°C, the junction-to-ambient thermal resistance for the SOT-23 package, θJ-A = 265°C/W, and the ambient temperature, TA.
The maximum allowable power dissipation at any ambient temperature for designs using this device can be calculated using the
formula:
If power dissipation exceeds the maximum specified above, the internal thermal protection circuitry will protect the device by reducing
the output voltage as required to maintain a safe junction temperature.
Electrical Characteristics
Unless otherwise specified: VIN = 5V, VSHDN = 5V, IL = 0 mA, and TJ = 25°C. Limits in standard typeface are for TJ = 25°C, and
limits in boldface type apply over the full operating temperature range (−40°C ≤ TJ ≤ +125°C). Minimum and Maximum limits
are ensured through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ =
25°C, and are provided for reference purposes only.
Symbol
VIN
Parameter
Input Voltage
Conditions
Min
2.7
Typical
Max
14
Units
V
ISW
Switch Current Limit
See(1)
0.80
1.25
500
A
RDS(ON)
Switch ON Resistance
ISW = 100 mA
Device ON
Device OFF
VSHDN = 0
650
mΩ
1.5
VSHDN(TH) Shutdown Threshold
V
0.50
0
0
ISHDN
Shutdown Pin Bias Current
µA
VSHDN = 5V
2
VFB
IFB
Feedback Pin Reference Voltage VIN = 3V
1.205
1.230
60
1.255
V
Feedback Pin Bias Current
VFB = 1.23V
nA
mA
VSHDN = 5V, Switching
VSHDN = 5V, Not Switching
VSHDN = 0
2.1
3.0
500
1
IQ
Quiescent Current
400
0.024
0.02
1.6
µA
ΔVFB/ΔVIN FB Voltage Line Regulation
2.7V ≤ VIN ≤ 14V
%/V
MHz
%
fSW
DMAX
IL
Switching Frequency
Maximum Duty Cycle
Switch Leakage
1.15
80
1.90
88
Not Switching, VSW = 5V
1
µA
(1) Switch current limit is dependent on duty cycle. Limits shown are for duty cycles ≤ 50%. See Figure 15 in Application Information –
MAXIMUM SWITCH CURRENT section.
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Typical Performance Characteristics
Unless otherwise specified: VIN = 5V, SHDN pin is tied to VIN, TJ = 25°C.
Iq VIN (Active) vs Temperature
Oscillator Frequency vs Temperature
Figure 2.
Figure 3.
Max. Duty Cycle vs Temperature
Feedback Voltage vs Temperature
88.5
88.4
88.3
88.2
88.1
88.0
87.9
87.8
-40 -25
0
25
50
75 100 125
TEMPERATURE (oC)
Figure 4.
Figure 5.
RDS(ON) vs Temperature
Current Limit vs Temperature
Figure 6.
Figure 7.
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Typical Performance Characteristics (continued)
Unless otherwise specified: VIN = 5V, SHDN pin is tied to VIN, TJ = 25°C.
RDS(ON) vs VIN
Efficiency vs Load Current (VOUT = 12V)
Figure 8.
Figure 9.
Efficiency vs Load Current (VOUT = 15V)
Efficiency vs Load Current (VOUT = 20V)
100
100
90
80
70
90
V
= 10V
V
IN
= 10V
IN
80
70
60
50
40
30
20
10
0
V
= 5V
IN
V
= 5V
IN
V
= 3.3V
IN
60
50
40
30
V
= 3.3V
IN
20
10
0
0
100 200
400 500
300
600 700
200
800
1000
0
600
400
LOAD CURRENT (mA)
LOAD CURRENT (mA)
Figure 10.
Figure 11.
Efficiency vs Load Current (VOUT = 25V)
100
90
80
70
60
50
40
30
20
10
0
V
IN
= 10V
V
= 5V
IN
0
50 100 150 200 250 300 350 400
LOAD CURRENT (mA)
Figure 12.
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Block Diagram
Theory of Operation
The LM27313 is a switching converter IC that operates at a fixed frequency of 1.6 MHz using current-mode
control for fast transient response over a wide input voltage range and incorporate pulse-by-pulse current limiting
protection. Because this is current mode control, a 50 mΩ sense resistor in series with the switch FET is used to
provide a voltage (which is proportional to the FET current) to both the input of the pulse width modulation
(PWM) comparator and the current limit amplifier.
At the beginning of each cycle, the S-R latch turns on the FET. As the current through the FET increases, a
voltage (proportional to this current) is summed with the ramp coming from the ramp generator and then fed into
the input of the PWM comparator. When this voltage exceeds the voltage on the other input (coming from the
Gm amplifier), the latch resets and turns the FET off. Since the signal coming from the Gm amplifier is derived
from the feedback (which samples the voltage at the output), the action of the PWM comparator constantly sets
the correct peak current through the FET to keep the output voltage in regulation.
Q1 and Q2 along with R3 - R6 form a bandgap voltage reference used by the IC to hold the output in regulation.
The currents flowing through Q1 and Q2 will be equal, and the feedback loop will adjust the regulated output to
maintain this. Because of this, the regulated output is always maintained at a voltage level equal to the voltage at
the FB node "multiplied up" by the ratio of the output resistive divider.
The current limit comparator feeds directly into the flip-flop, that drives the switch FET. If the FET current reaches
the limit threshold, the FET is turned off and the cycle terminated until the next clock pulse. The current limit
input terminates the pulse regardless of the status of the output of the PWM comparator.
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APPLICATION INFORMATION
SELECTING THE EXTERNAL CAPACITORS
The LM27313 requires ceramic capacitors at the input and output to accommodate the peak switching currents
the part needs to operate. Electrolytic capacitors have resonant frequencies which are below the switching
frequency of the device, and therefore can not provide the currents needed to operate. Electrolytics may be used
in parallel with the ceramics for bulk charge storage which will improve transient response.
When selecting a ceramic capacitor, only X5R and X7R dielectric types should be used. Other types such as
Z5U and Y5F have such severe loss of capacitance due to effects of temperature variation and applied voltage,
they may provide as little as 20% of rated capacitance in many typical applications. Always consult capacitor
manufacturer’s data curves before selecting a capacitor. High-quality ceramic capacitors can be obtained from
Taiyo-Yuden, AVX, and Murata.
SELECTING THE OUTPUT CAPACITOR
A single ceramic capacitor of value 4.7 µF to 10 µF will provide sufficient output capacitance for most
applications. For output voltages below 10V, a 10 µF capacitance is required. If larger amounts of capacitance
are desired for improved line support and transient response, tantalum capacitors can be used in parallel with the
ceramics. Aluminum electrolytics with ultra low ESR such as Sanyo Oscon can be used, but are usually
prohibitively expensive. Typical AI electrolytic capacitors are not suitable for switching frequencies above 500
kHz due to significant ringing and temperature rise due to self-heating from ripple current. An output capacitor
with excessive ESR can also reduce phase margin and cause instability.
SELECTING THE INPUT CAPACITOR
An input capacitor is required to serve as an energy reservoir for the current which must flow into the inductor
each time the switch turns ON. This capacitor must have extremely low ESR and ESL, so ceramic must be used.
We recommend a nominal value of 2.2 µF, but larger values can be used. Since this capacitor reduces the
amount of voltage ripple seen at the input pin, it also reduces the amount of EMI passed back along that line to
other circuitry.
FEED-FORWARD COMPENSATION
Although internally compensated, the feed-forward capacitor Cf is required for stability (see Typical Application
Circuits). Adding this capacitor puts a zero in the loop response of the converter. Without it, the regulator loop
can oscillate. The recommended frequency for the zero fz should be approximately 8 kHz. Cf can be calculated
using the formula:
Cf = 1 / (2 x π x R1 x fz)
(1)
SELECTING DIODES
The external diode used in the typical application should be a Schottky diode. If the switch voltage is less than
15V, a 20V diode such as the MBR0520 is recommended. If the switch voltage is between 15V and 25V, a 30V
diode such as the MBR0530 is recommended. If the switch voltage exceeds 25V, a 40V diode such as the
MBR0540 should be used.
The MBR05xx series of diodes are designed to handle a maximum average current of 500mA. For applications
with load currents to 800mA, a Microsemi UPS5817 can be used.
LAYOUT HINTS
High frequency switching regulators require very careful layout of components in order to get stable operation
and low noise. All components must be as close as possible to the LM27313 device. It is recommended that a 4-
layer PCB be used so that internal ground planes are available.
As an example, a recommended layout of components is shown:
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Figure 13. Recommended PCB Component Layout
Some additional guidelines to be observed:
1. Keep the path between L1, D1, and C2 extremely short. Parasitic trace inductance in series with D1 and C2
will increase noise and ringing.
2. The feedback components R1, R2 and CF must be kept close to the FB pin of the LM27313 to prevent noise
injection on the high impedance FB pin.
3. If internal ground planes are available (recommended) use vias to connect directly to the LM27313 ground at
device pin 2, as well as the negative sides of capacitors C1 and C2.
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the external resistors R1 and R2 (see Typical Application Circuits). A value of
13.3 kΩ is recommended for R2 to establish a divider current of approximately 92 µA. R1 is calculated using the
formula:
R1 = R2 x ( (VOUT / VFB) − 1 )
(2)
DUTY CYCLE
The maximum duty cycle of the switching regulator determines the maximum boost ratio of output-to-input
voltage that the converter can attain in continuous mode of operation. The duty cycle for a given boost
application is defined as:
VOUT + VDIODE - VIN
Duty Cycle =
VOUT + VDIODE - VSW
(3)
This applies for continuous mode operation.
The equation shown for calculating duty cycle incorporates terms for the FET switch voltage and diode forward
voltage. The actual duty cycle measured in operation will also be affected slightly by other power losses in the
circuit such as wire losses in the inductor, switching losses, and capacitor ripple current losses from self-heating.
Therefore, the actual (effective) duty cycle measured may be slightly higher than calculated to compensate for
these power losses. A good approximation for effective duty cycle is :
DC (eff) = (1 - Efficiency x (VIN / VOUT))
(4)
Where the efficiency can be approximated from the curves provided.
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INDUCTANCE VALUE
The first question we are usually asked is: “How small can I make the inductor?” (because they are the largest
sized component and usually the most costly). The answer is not simple and involves trade-offs in performance.
More inductance means less inductor ripple current and less output voltage ripple (for a given size of output
capacitor). More inductance also means more load power can be delivered because the energy stored during
each switching cycle is:
E = L/2 x (lp)2
where
•
“lp” is the peak inductor current.
(5)
An important point to observe is that the LM27313 will limit its switch current based on peak current. This means
that since lp(max) is fixed, increasing L will increase the maximum amount of power available to the load.
Conversely, using too little inductance may limit the amount of load current which can be drawn from the output.
Best performance is usually obtained when the converter is operated in “continuous” mode at the load current
range of interest, typically giving better load regulation and less output ripple. Continuous operation is defined as
not allowing the inductor current to drop to zero during the cycle. It should be noted that all boost converters shift
over to discontinuous operation as the output load is reduced far enough, but a larger inductor stays “continuous”
over a wider load current range.
To better understand these tradeoffs, a typical application circuit (5V to 12V boost with a 10 µH inductor) will be
analyzed.
Since the LM27313 typical switching frequency is 1.6 MHz, the typical period is equal to 1/fSW(TYP), or
approximately 0.625 µs.
We will assume: VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V. The duty cycle is:
Duty Cycle = ((12V + 0.5V - 5V) / (12V + 0.5V - 0.5V)) = 62.5%
(6)
(7)
The typical ON time of the switch is:
(62.5% x 0.625 µs) = 0.390 µs
It should be noted that when the switch is ON, the voltage across the inductor is approximately 4.5V.
Using the equation:
V = L (di/dt)
(8)
We can then calculate the di/dt rate of the inductor which is found to be 0.45 A/µs during the ON time. Using
these facts, we can then show what the inductor current will look like during operation:
Figure 14. 10 µH Inductor Current, 5V–12V Boost
During the 0.390 µs ON time, the inductor current ramps up 0.176A and ramps down an equal amount during the
OFF time. This is defined as the inductor “ripple current”. It can also be seen that if the load current drops to
about 33 mA, the inductor current will begin touching the zero axis which means it will be in discontinuous mode.
A similar analysis can be performed on any boost converter, to make sure the ripple current is reasonable and
continuous operation will be maintained at the typical load current values.
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MAXIMUM SWITCH CURRENT
The maximum FET switch current available before the current limiter cuts in is dependent on duty cycle of the
application. This is illustrated in Figure 15 below which shows typical values of switch current as a function of
effective (actual) duty cycle:
1600
1400
1200
1000
V
= 5V
IN
V
= 3.3V
IN
800
600
400
200
0
V
IN
= 2.7V
0
20
40
60
80
100
DUTY CYCLE (%) = [1 - EFF*(VIN/VOUT))]
Figure 15. Switch Current Limit vs Duty Cycle
CALCULATING LOAD CURRENT
As shown in the figure which depicts inductor current, the load current is related to the average inductor current
by the relation:
ILOAD = IIND(AVG) x (1 - DC)
where
•
"DC" is the duty cycle of the application.
(9)
(10)
(11)
The switch current can be found by:
ISW = IIND(AVG) + ½ (IRIPPLE
)
Inductor ripple current is dependent on inductance, duty cycle, input voltage and frequency:
IRIPPLE = DC x (VIN - VSW) / (fSW x L)
Combining all terms, we can develop an expression which allows the maximum available load current to be
calculated:
ILOAD(max) = (1 - DC) x (ISW(max) - DC (VIN - VSW))
2fL
(12)
The equation shown to calculate maximum load current takes into account the losses in the inductor or turn-OFF
switching losses of the FET and diode. For actual load current in typical applications, we took bench data for
various input and output voltages and displayed the maximum load current available for a typical device in graph
form:
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Figure 16. Max. Load Current vs VIN
DESIGN PARAMETERS VSW AND ISW
The value of the FET "ON" voltage (referred to as VSW in the equations) is dependent on load current. A good
approximation can be obtained by multiplying the "ON Resistance" of the FET times the average inductor
current.
FET on resistance increases at VIN values below 5V, since the internal N-FET has less gate voltage in this input
voltage range (see Typical Performance Characteristics curves). Above VIN = 5V, the FET gate voltage is
internally clamped to 5V.
The maximum peak switch current the device can deliver is dependent on duty cycle. The minimum switch
current value (ISW) is ensured to be at least 800 mA at duty cycles below 50%. For higher duty cycles, see
Typical Performance Characteristics curves.
THERMAL CONSIDERATIONS
At higher duty cycles, the increased ON time of the FET means the maximum output current will be determined
by power dissipation within the LM27313 FET switch. The switch power dissipation from ON-state conduction is
calculated by:
PSW = DC x IIND(AVG)2 x RDS(ON)
(13)
There will be some switching losses as well, so some derating needs to be applied when calculating IC power
dissipation.
MINIMUM INDUCTANCE
In some applications where the maximum load current is relatively small, it may be advantageous to use the
smallest possible inductance value for cost and size savings. The converter will operate in discontinuous mode in
such a case.
The minimum inductance should be selected such that the inductor (switch) current peak on each cycle does not
reach the 800 mA current limit maximum. To understand how to do this, an example will be presented.
In this example, the LM27313 nominal switching frequency is 1.6 MHz, and the minimum switching frequency is
1.15 MHz. This means the maximum cycle period is the reciprocal of the minimum frequency:
TON(max) = 1/1.15M = 0.870 µs
(14)
(15)
(16)
We will assume: VIN = 5V, VOUT = 12V, VSW = 0.2V, and VDIODE = 0.3V. The duty cycle is:
Duty Cycle = ((12V + 0.3V - 5V) / (12V + 0.3V - 0.2V)) = 60.3%
Therefore, the maximum switch ON time is:
(60.3% x 0.870 µs) = 0.524 µs
An inductor should be selected with enough inductance to prevent the switch current from reaching 800 mA in
the 0.524 µs ON time interval (see Figure 17):
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Figure 17. Discontinuous Design, 5V–12V Boost
The voltage across the inductor during ON time is 4.8V. Minimum inductance value is found by:
L = V x (dt/dl)
(17)
(18)
L = 4.8V x (0.524 µs / 0.8 mA) = 3.144 µH
In this case, a 3.3 µH inductor could be used, assuming it provided at least that much inductance up to the 800
mA current value. This same analysis can be used to find the minimum inductance for any boost application.
INDUCTOR SUPPLIERS
Some of the recommended suppliers of inductors for this product include, but are not limited to, Sumida,
Coilcraft, Panasonic, TDK and Murata. When selecting an inductor, make certain that the continuous current
rating is high enough to avoid saturation at peak currents. A suitable core type must be used to minimize core
(switching) losses, and wire power losses must be considered when selecting the current rating.
SHUTDOWN PIN OPERATION
The device is turned off by pulling the shutdown pin low. If this function is not going to be used, the pin should be
tied directly to VIN. If the SHDN function will be needed, a pull-up resistor must be used to VIN (50kΩ to 100 kΩ is
recommended), or the pin must be actively driven high and low. The SHDN pin must not be left unterminated.
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REVISION HISTORY
Changes from Revision C (April 2013) to Revision D
Page
•
Changed layout of National Data Sheet to TI format .......................................................................................................... 12
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PACKAGE OPTION ADDENDUM
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1-Nov-2013
PACKAGING INFORMATION
Orderable Device
Status Package Type Package Pins Package
Eco Plan
Lead/Ball Finish
MSL Peak Temp
Op Temp (°C)
Device Marking
Samples
Drawing
Qty
(1)
(2)
(6)
(3)
(4/5)
LM27313XMF/NOPB
ACTIVE
SOT-23
DBV
5
1000
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
SRPB
LM27313XMFX
NRND
SOT-23
SOT-23
DBV
DBV
5
5
3000
3000
TBD
Call TI
CU SN
Call TI
-40 to 125
-40 to 125
SRPB
SRPB
LM27313XMFX/NOPB
ACTIVE
Green (RoHS
& no Sb/Br)
Level-1-260C-UNLIM
LM27313XQMF/NOPB
LM27313XQMFX/NOPB
ACTIVE
ACTIVE
SOT-23
SOT-23
DBV
DBV
5
5
1000
3000
Green (RoHS
& no Sb/Br)
CU SN
CU SN
Level-1-260C-UNLIM
Level-1-260C-UNLIM
-40 to 125
-40 to 125
SD3B
SD3B
Green (RoHS
& no Sb/Br)
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
1-Nov-2013
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
OTHER QUALIFIED VERSIONS OF LM27313, LM27313-Q1 :
Catalog: LM27313
•
Automotive: LM27313-Q1
•
NOTE: Qualified Version Definitions:
Catalog - TI's standard catalog product
•
Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects
•
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
26-Sep-2013
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
B0
K0
P1
W
Pin1
Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant
(mm) W1 (mm)
LM27313XMFX
SOT-23
DBV
DBV
DBV
5
5
5
3000
1000
3000
178.0
178.0
178.0
8.4
8.4
8.4
3.2
3.2
3.2
3.2
3.2
3.2
1.4
1.4
1.4
4.0
4.0
4.0
8.0
8.0
8.0
Q3
Q3
Q3
LM27313XQMF/NOPB SOT-23
LM27313XQMFX/NOPB SOT-23
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
26-Sep-2013
*All dimensions are nominal
Device
Package Type Package Drawing Pins
SPQ
Length (mm) Width (mm) Height (mm)
LM27313XMFX
SOT-23
SOT-23
SOT-23
DBV
DBV
DBV
5
5
5
3000
1000
3000
210.0
210.0
210.0
185.0
185.0
185.0
35.0
35.0
35.0
LM27313XQMF/NOPB
LM27313XQMFX/NOPB
Pack Materials-Page 2
IMPORTANT NOTICE
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changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest
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TI warrants performance of its components to the specifications applicable at the time of sale, in accordance with the warranty in TI’s terms
and conditions of sale of semiconductor products. Testing and other quality control techniques are used to the extent TI deems necessary
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相关型号：
LM27313XMFX/NOPB
1.6 MHz boost converter with 30V Internal FET switch in SOT-23 5-SOT-23 -40 to 125
TI
LM27313XQMF/NOPB
具有 30V 内部 FET 开关并采用 SOT-23 封装的 1.6MHz 升压转换器，符合 AEC-Q100 标准 | DBV | 5 | -40 to 125
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具有 30V 内部 FET 开关并采用 SOT-23 封装的 1.6MHz 升压转换器，符合 AEC-Q100 标准 | DBV | 5 | -40 to 125
TI
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