LM3405AXMKE/NOPB [TI]

1.6MHz, 1A Constant Current Buck LED Driver with Internal Compensation in Tiny SOT and MSOP PowerPAD Packages;


型号：	LM3405AXMKE/NOPB
厂家：	TEXAS INSTRUMENTS
描述：	1.6MHz, 1A Constant Current Buck LED Driver with Internal Compensation in Tiny SOT and MSOP PowerPAD Packages 开关光电二极管
文件：	总32页 (文件大小：1417K)
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LM3405A

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SNVS508C –OCTOBER 2007–REVISED MAY 2013

LM3405A 1.6MHz, 1A Constant Current Buck LED Driver with Internal Compensation in

Tiny SOT and VSSOP PowerPAD Packages

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FEATURES

DESCRIPTION

The LM3405A is a 1A constant current buck LED

driver designed to provide a simple, high efficiency

solution for driving high power LEDs. With a 0.205V

reference voltage feedback control to minimize power

dissipation, an external resistor sets the current as

needed for driving various types of LEDs. Switching

frequency is internally set to 1.6MHz, allowing small

surface mount inductors and capacitors to be used.

The LM3405A utilizes current-mode control and

internal compensation offering ease of use and

predictable, high performance regulation over a wide

range of operating conditions. With a maximum input

voltage of 22V, it can drive up to 5 High-Brightness

LEDs in series at 1A forward current, with the single

LED forward voltage of approximately 3.7V.

Additional features include user accessible EN/DIM

pin for enabling and PWM dimming of LEDs, thermal

shutdown, cycle-by-cycle current limit and over-

current protection.

•

V_INOperating Range of 3V to 22V

•

Drives up to 5 High-Brightness LEDs in Series

at 1A

•

Thin SOT-6 package and VSSOP-8 PowerPAD

Packages

•

1.6MHz Switching Frequency

EN/DIM Input for Enabling and PWM Dimming

of LEDs

•

300mΩ NMOS Switch

40nA Shutdown Current at V_IN= 5V

Internally Compensated Current-mode Control

Cycle-by-cycle Current Limit

Input Voltage UVLO

Over-current Protection

Thermal Shutdown

APPLICATIONS

•

LED Driver

Constant Current Source

Industrial Lighting

LED Flashlights

LED Lightbulbs

TYPICAL APPLICATION CIRCUIT

Efficiency vs LED Current (V_IN= 12V)

BOOST

OUT

LM3405A

EN/DIM

OFF

GND

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of

Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

All trademarks are the property of their respective owners.

PRODUCTION DATA information is current as of publication date.

Products conform to specifications per the terms of the Texas

Instruments standard warranty. Production processing does not

necessarily include testing of all parameters.

LM3405A

SNVS508C –OCTOBER 2007–REVISED MAY 2013

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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam

during storage or handling to prevent electrostatic damage to the MOS gates.

CONNECTION DIAGRAMS

GND

EN/DIM

GND

VIN

BOOST

GND

EN/DIM

BOOST

Figure 1. 6-Lead SOT Package

See Package Number DDC0006A

Figure 2. 8-Lead VSSOP PowerPAD Package

See Package Number DGN0008A

SOT-6 Pin Descriptions

Pin No.

Name

Application Information

Boost voltage that drives the NMOS output switch. A bootstrap capacitor is connected between the BOOST and

SW pins.

BOOST

GND

Signal and power ground pin. Place the bottom resistor of the feedback network as close as possible to this pin.

Feedback pin. Connect FB to external resistor divider to set output voltage.

Enable control input. Logic high enables operation. Toggling this pin with a periodic logic square wave of varying

EN/DIM duty cycle at different frequencies controls the brightness of LEDs. Do not allow this pin to float or be greater than

V_IN+ 0.3V.

VIN

Input supply voltage. Connect a bypass capacitor locally from this pin to GND.

Switch pin. Connect this pin to the inductor, catch diode, and bootstrap capacitor.

VSSOP-8 PowerPAD Pin Descriptions

Pin No.

Name

Function

Feedback pin. Connect FB to external resistor divider to set output voltage.

2, 7

GND

Signal and power ground pin. Place the bottom resistor of the feedback network as close as possible to this pin. Tie

pins 2, 7 and DAP to one GND plane.

No Connection

BOOST Boost voltage that drives the NMOS output switch. A bootstrap capacitor is connected between the BOOST and SW

pins.

VIN

Switch pin. Connect this pin to the inductor, catch diode, and bootstrap capacitor.

Input supply voltage. Connect a bypass capacitor locally from this pin to GND.

EN/DIM Enable control input. Logic high enables operation. Toggling this pin with a periodic logic square wave of varying

duty cycle at different frequencies controls the brightness of LEDs. Do not allow this pin to float or be greater than

V_IN+ 0.3V.

DAP

Attach to power ground pin

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(1)

ABSOLUTE MAXIMUM RATINGS

If Military/Aerospace specified devices are required, contact the Texas Instruments Sales Office/Distributors for availability

and specifications.

VALUE / UNIT

V_IN

-0.5V to 24V

-0.5V to 30V

-0.5V to 6.0V

-0.5V to 3.0V

-0.5V to (V_IN+ 0.3V)

150°C

SW Voltage

Boost Voltage

Boost to SW Voltage

FB Voltage

EN/DIM Voltage

Junction Temperature

(2)

ESD Susceptibility

2kV

Storage Temperature

-65°C to +150°C

220°C

Soldering Information Infrared/Convection Reflow (15sec)

(1) Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings define the conditions under

which the device is intended to be functional. For specifications and test conditions, see the Electrical Characteristics.

(2) Human body model, 1.5kΩ in series with 100pF.

(1)

RECOMMENDED OPERATING CONDITIONS

VALUE / UNIT

V_IN

3V to 22V

-0.5V to (V_IN+ 0.3V)

2.5V to 5.5V

-40°C to +125°C

SOT 118°C/W

eMSOP8 73°C/W

400mA

EN/DIM voltage

Boost to SW Voltage

Junction Temperature Range

(2)

Thermal Resistance θ_JA

(3)

Thermal Resistance θ_JA

I_LEDSOT package

I_LEDVSSOP PowerPAD package

(1) Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings define the conditions under

which the device is intended to be functional. For specifications and test conditions, see the Electrical Characteristics.

(2) Thermal shutdown will occur if the junction temperature (T_J) exceeds 165°C. The maximum allowable power dissipation (P_D) at any

ambient temperature (T_A) is P_D= (T_J(MAX)– T_A)/θ_JA. This number applies to packages soldered directly onto a 3" x 3" PC board with

2oz. copper on 4 layers in still air. For a 2 layer board using 1 oz. copper in still air, θ_JA= 204°C/W.

(3) Thermal shutdown will occur if the junction temperature (T_J) exceeds 165°C. The maximum allowable power dissipation (P_D) at any

ambient temperature (T_A) is P_D= (T_J(MAX)– T_A)/θ_JA. This number applies to packages soldered directly onto a 1" x 0.75" PC board with

1oz. copper on 4 layers in still air.

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ELECTRICAL CHARACTERISTICS

Unless otherwise specified, V_IN= 12V. Limits in standard type are for T_J= 25°C only; limits in boldface type apply over the

junction temperature (T_J) range of -40°C to +125°C. Minimum and Maximum limits are specified through test, design, or

statistical correlation. Typical values represent the most likely parametric norm, and are provided for reference purposes only.

Parameter

Test Conditions

Min

Typ

0.205

0.01

Max

Units

V_FB

ΔV_FB/(ΔV_INxV_FB

I_FB

Feedback Voltage

0.188

0.220

)

Feedback Voltage Line Regulation V_IN= 3V to 22V

%/V

Feedback Input Bias Current

Under-voltage Lockout

UVLO Hysteresis

Sink/Source

V_INRising

V_INFalling

250

2.74

2.3

2.95

UVLO

1.9

0.44

1.6

f_SW

Switching Frequency

Maximum Duty Cycle

1.2

1.9

MHz

D_MAX

V_FB= 0V

SOT (V_BOOST- V_SW= 3V)

MSOP (V_BOOST- V_SW= 3V)

V_BOOST- V_SW= 3V, V_IN= 3V

Switching, V_FB= 0.195V

V_EN/DIM= 0V

300

360

2.0

600

700

2.8

R_DS(ON)

I_CL

Switch ON Resistance

mΩ

Switch Current Limit

1.2

1.8

µA

Quiescent Current

1.8

I_Q

Quiescent Current (Shutdown)

Enable Threshold Voltage

Shutdown Threshold Voltage

EN/DIM Pin Current

0.3

V_EN/DIMRising

V_{EN/DIM_TH}

V_EN/DIMFalling

0.4

I_EN/DIM

I_SW

Sink/Source

0.01

0.1

µA

Switch Leakage

V_IN= 22V

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TYPICAL PERFORMANCE CHARACTERISTICS

Unless otherwise specified, V_IN= 12V, V_BOOST- V_SW= 5V and T_A= 25°C.

Efficiency vs LED Current (V_IN=5V)

Efficiency vs Input Voltage (I_F= 1A)

Figure 3.

Efficiency vs Input Voltage (I_F= 0.7A)

Figure 4.

Efficiency vs Input Voltage (I_F= 0.35A)

Figure 5.

Figure 6.

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TYPICAL PERFORMANCE CHARACTERISTICS (continued)

Unless otherwise specified, V_IN= 12V, V_BOOST- V_SW= 5V and T_A= 25°C.

V_FBvs Temperature

Oscillator Frequency vs Temperature

Figure 7.

Figure 8.

Current Limit vs Temperature

SOT R_DS(ON)vs Temperature (V_BOOST- V_SW= 3V)

Figure 9.

Figure 10.

Quiescent Current vs Temperature

Startup Response to EN/DIM Signal (V_IN= 15V, I_F= 0.2A)

Figure 11.

Figure 12.

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SNVS508C –OCTOBER 2007–REVISED MAY 2013

BLOCK DIAGRAM

Figure 13. Simplified Block Diagram

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APPLICATION INFORMATION

THEORY OF OPERATION

The LM3405A is a PWM, current-mode controled buck switching regulator designed to provide a simple, high

efficiency solution for driving LEDs with a preset switching frequency of 1.6MHz. This high frequency allows the

LM3405A to operate with small surface mount capacitors and inductors, resulting in LED drivers that need only a

minimum amount of board space. The LM3405A is internally compensated, simple to use, and requires few

external components.

The following description of operation of the LM3405A will refer to the Simplified Block Diagram (Figure 13) and

to the waveforms in Figure 14. The LM3405A supplies a regulated output current by switching the internal NMOS

power switch at constant frequency and variable duty cycle. A switching cycle begins at the falling edge of the

reset pulse generated by the internal oscillator. When this pulse goes low, the output control logic turns on the

internal NMOS power switch. During this on-time, the SW pin voltage (V_SW) swings up to approximately V_IN, and

the inductor current (I_L) increases with a linear slope. I_Lis measured by the current sense amplifier, which

generates an output proportional to the switch current. The sense signal is summed with the regulator’s

corrective ramp and compared to the error amplifier’s output, which is proportional to the difference between the

feedback voltage and V_REF. When the PWM comparator output goes high, the internal power switch turns off until

the next switching cycle begins. During the switch off-time, inductor current discharges through the catch diode

D1, which forces the SW pin to swing below ground by the forward voltage (V_D1) of the catch diode. The

regulator loop adjusts the duty cycle (D) to maintain a constant output current (I_F) through the LED, by forcing FB

pin voltage to be equal to V_REF(0.205V).

D = T /T

ON SW

Voltage

OFF

-V

LPK

Inductor

Current

Figure 14. SW Pin Voltage and Inductor Current Waveforms of LM3405A

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BOOST FUNCTION

Capacitor C3 and diode D2 in Figure 13 are used to generate a voltage V_BOOST. The voltage across C3, V_BOOST

V_SW, is the gate drive voltage to the internal NMOS power switch. To properly drive the internal NMOS switch

during its on-time, V_BOOSTneeds to be at least 2.5V greater than V_SW. A large value of V_BOOST- V_SWis

recommended to achieve better efficiency by minimizing both the internal switch ON resistance (R_DS(ON)), and the

switch rise and fall times. However, V_BOOST- V_SWshould not exceed the maximum operating limit of 5.5V.

When the LM3405A starts up, internal circuitry from V_INsupplies a 20mA current to the BOOST pin, flowing out

of the BOOST pin into C3. This current charges C3 to a voltage sufficient to turn the switch on. The BOOST pin

will continue to source current to C3 until the voltage at the feedback pin is greater than 123mV.

There are various methods to derive V_BOOST

1. From the input voltage (V_IN)

2. From the output voltage (V_OUT

3. From a shunt or series zener diode

4. From an external distributed voltage rail (V_EXT

)

The first method is shown in the Simplified Block Diagram of Figure 13. Capacitor C3 is charged via diode D2 by

V_IN. During a normal switching cycle, when the internal NMOS power switch is off (T_OFF) (refer to Figure 14),

V_BOOSTequals V_INminus the forward voltage of D2 (V_D2), during which the current in the inductor (L1) forward

biases the catch diode D1 (V_D1). Therefore the gate drive voltage stored across C3 is:

V_BOOST- V_SW= V_IN- V_D2+ V_D1

(1)

When the NMOS switch turns on (T_ON), the switch pin rises to:

V_SW= V_IN– (R_DS(ON)x I_L)

(2)

Since the voltage across C3 remains unchanged, V_BOOSTis forced to rise thus reverse biasing D2. The voltage at

V_BOOSTis then:

V_BOOST= 2V_IN– (R_DS(ON)x I_L) – V_D2+ V_D1

(3)

Depending on the quality of the diodes D1 and D2, the gate drive voltage in this method can be slightly less or

larger than the input voltage V_IN. For best performance, ensure that the variation of the input supply does not

cause the gate drive voltage to fall outside the recommended range:

2.5V < V_IN- V_D2+ V_D1< 5.5V

(4)

The second method for deriving the boost voltage is to connect D2 to the output as shown in Figure 15. The gate

drive voltage in this configuration is:

V_BOOST- V_SW= V_OUT– V_D2+ V_D1

(5)

Since the gate drive voltage needs to be in the range of 2.5V to 5.5V, the output voltage V_OUTshould be limited

to a certain range. For the calculation of V_OUT, see OUTPUT VOLTAGE section.

Figure 15. V_BOOSTDerived from V_OUT

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The third method can be used in the applications where both V_INand V_OUTare greater than 5.5V. In these cases,

C3 cannot be charged directly from these voltages; instead C3 can be charged from V_INor V_OUTminus a zener

voltage (V_D3) by placing a zener diode D3 in series with D2 as shown in Figure 16. When using a series zener

diode from the input, the gate drive voltage is V_IN- V_D3- V_D2+ V_D1

Figure 16. V_BOOSTDerived from V_INthrough a Series Zener

An alternate method is to place the zener diode D3 in a shunt configuration as shown in Figure 17. A small

350mW to 500mW, 5.1V zener in a SOT or SOD package can be used for this purpose. A small ceramic

capacitor such as a 6.3V, 0.1µF capacitor (C5) should be placed in parallel with the zener diode. When the

internal NMOS switch turns on, a pulse of current is drawn to charge the internal NMOS gate capacitance. The

0.1µF parallel shunt capacitor ensures that the V_BOOSTvoltage is maintained during this time. Resistor R2 should

be chosen to provide enough RMS current to the zener diode and to the BOOST pin. A recommended choice for

the zener current (I_ZENER) is 1mA. The current I_BOOSTinto the BOOST pin supplies the gate current of the NMOS

power switch. It reaches a maximum of around 3.6mA at the highest gate drive voltage of 5.5V over the

LM3405A operating range.

For the worst case I_BOOST, increase the current by 50%. In that case, the maximum boost current will be:

I_BOOST-MAX= 1.5 x 3.6mA = 5.4mA

(6)

(7)

(8)

R2 will then be given by:

R2 = (V_IN- V_ZENER) / (I_{BOOST_MAX}+ I_ZENER

)

For example, let V_IN= 12V, V_ZENER= 5V, I_ZENER= 1mA, then:

R2 = (12V - 5V) / (5.4mA + 1mA) = 1.09kΩ

Figure 17. V_BOOSTderived from V_INthrough a Shunt Zener

The fourth method can be used in an application which has an external low voltage rail, V_EXT. C3 can be charged

through D2 from V_EXT, independent of V_INand V_OUTvoltage levels. Again for best performance, ensure that the

gate drive voltage, V_EXT- V_D2+ V_D1, falls in the range of 2.5V to 5.5V.

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SETTING THE LED CURRENT

LM3405A is a constant current buck regulator. The LEDs are connected between V_OUTand the FB pin as shown

in the Typical Application Circuit. The FB pin is at 0.205V in regulation and therefore the LED current I_Fis set by

V_FBand resistor R1 from FB to ground by the following equation:

I_F= V_FB/ R1

(9)

I_Fshould not exceed the 1A current capability of LM3405A and therefore R1 minimum must be approximately

0.2Ω. I_Fshould also be kept above 200mA for stable operation, and therefore R1 maximum must be

approximately 1Ω. If average LED currents less than 200mA are desired, the EN/DIM pin can be used for PWM

dimming. See LED PWM DIMMING section.

OUTPUT VOLTAGE

The output voltage is primarily determined by the number of LEDs (n) connected from V_OUTto FB pin and

therefore V_OUTcan be written as :

V_OUT= ((n x V_F) + V_FB

)

(10)

where V_Fis the forward voltage of one LED at the set LED current level (see LED manufacturer datasheet for

forward characteristics curve).

ENABLE MODE / SHUTDOWN MODE

The LM3405A has both enable and shutdown modes that are controlled by the EN/DIM pin. Connecting a

voltage source greater than 1.8V to the EN/DIM pin enables the operation of LM3405A, while reducing this

voltage below 0.4V places the part in a low quiescent current (0.3µA typical) shutdown mode. There is no

internal pull-up on EN/DIM pin, therefore an external signal is required to initiate switching. Do not allow this pin

to float or rise to 0.3V above V_IN. It should be noted that when the EN/DIM pin voltage rises above 1.8V while the

input voltage is greater than UVLO, there is a finite delay before switching starts. During this delay the LM3405A

will go through a power on reset state after which the internal soft-start process commences. The soft-start

process limits the inrush current and brings up the LED current (I_F) in a smooth and controlled fashion. The total

combined duration of the power on reset delay, soft-start delay and the delay to fully establish the LED current is

in the order of 100µs (refer to Figure 22).

The simplest way to enable the operation of LM3405A is to connect the EN/DIM pin to V_INwhich allows self start-

up of LM3405A whenever the input voltage is applied. However, when an input voltage of slow rise time is used

to power the application and if both the input voltage and the output voltage are not fully established before the

soft-start time elapses, the control circuit will command maximum duty cycle operation of the internal power

switch to bring up the output voltage rapidly. When the feedback pin voltage exceeds 0.205V, the duty cycle will

have to reduce from the maximum value accordingly, to maintain regulation. It takes a finite amount of time for

this reduction of duty cycle and this will result in a spike in LED current for a short duration as shown in

Figure 18. In applications where this LED current overshoot is undesirable, EN/DIM pin voltage can be

separately applied and delayed such that V_INis fully established before the EN/DIM pin voltage reaches the

enable threshold. The effect of delaying EN/DIM with respect to V_INon the LED current is shown in Figure 19.

For a fast rising input voltage (200µs for example), there is no need to delay the EN/DIM signal since soft-start

can smoothly bring up the LED current as shown in Figure 20.

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Figure 18. Startup Response to V_INwith 5ms Rise

Time

Figure 19. Startup Response to V_INwith EN/DIM

Delayed

Figure 20. Startup Response to V_INwith 200µs Rise Time

LED PWM DIMMING

The LED brightness can be controlled by applying a periodic pulse signal to the EN/DIM pin and varying its

frequency and/or duty cycle. This so-called PWM dimming method controls the average light output by pulsing

the LED current between the set value and zero. A logic high level at the EN/DIM pin turns on the LED current

whereas a logic low level turns off the LED current. Figure 21 shows a typical LED current waveform in PWM

dimming mode. As explained in the previous section, there is approximately a 100µs delay from the EN/DIM

signal going high to fully establishing the LED current as shown in Figure 22. This 100µs delay sets a maximum

frequency limit for the driving signal that can be applied to the EN/DIM pin for PWM dimming. Figure 23 shows

the average LED current versus duty cycle of PWM dimming signal for various frequencies. The applicable

frequency range to drive LM3405A for PWM dimming is from 100Hz to 5kHz. The dimming ratio reduces

drastically when the applied PWM dimming frequency is greater than 5kHz.

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Figure 21. PWM Dimming of LEDs

Using the EN/DIM Pin

Figure 22. Startup Response to EN/DIM

with I_F= 1A

Figure 23. Average LED Current

Duty Cycle of PWM Dimming Signal at EN/DIM Pin

UNDER-VOLTAGE LOCKOUT

Under-voltage lockout (UVLO) prevents the LM3405A from operating until the input voltage exceeds 2.74V

(typical). The UVLO threshold has approximately 440mV of hysteresis, so the part will operate until V_INdrops

below 2.3V (typical). Hysteresis prevents the part from turning off during power up if V_INis non-monotonic.

CURRENT LIMIT

The LM3405A uses cycle-by-cycle current limit to protect the internal power switch. During each switching cycle,

a current limit comparator detects if the power switch current exceeds 2.0A (typical), and turns off the switch until

the next switching cycle begins.

OVER-CURRENT PROTECTION

The LM3405A has a built in over-current comparator that compares the FB pin voltage to a threshold voltage that

is 60% higher than the internal reference V_REF. Once the FB pin voltage exceeds this threshold level (typically

328mV), the internal NMOS power switch is turned off, which allows the feedback voltage to decrease towards

regulation. This threshold provides an upper limit for the LED current. LED current overshoot is limited to

328mV/R1 by this comparator during transients.

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THERMAL SHUTDOWN

Thermal shutdown limits total power dissipation by turning off the internal power switch when the IC junction

temperature exceeds 165°C. After thermal shutdown occurs, the power switch does not turn on until the junction

temperature drops below approximately 150°C.

DESIGN GUIDE

INDUCTOR (L1)

The Duty Cycle (D) can be approximated quickly using the ratio of output voltage (V_OUT) to input voltage (V_IN):

V_OUT

D =

V_IN

(11)

The catch diode (D1) forward voltage drop and the voltage drop across the internal NMOS must be included to

calculate a more accurate duty cycle. Calculate D by using the following formula:

V_OUT+ V_D1

D =

V_IN+ V_D1- V_SW

(12)

V_SWcan be approximated by:

V_SW= I_Fx R_DS(ON)

(13)

The diode forward drop (V_D1) can range from 0.3V to 0.7V depending on the quality of the diode. The lower V_D1

is, the higher the operating efficiency of the converter.

The inductor value determines the output ripple current (Δi_L, as defined in Figure 14). Lower inductor values

decrease the size of the inductor, but increases the output ripple current. An increase in the inductor value will

decrease the output ripple current. The ratio of ripple current to LED current is optimized when it is set between

0.3 and 0.4 at 1A LED current. This ratio r is defined as:

Di_L

r =

l_F

(14)

One must also ensure that the minimum current limit (1.2A) is not exceeded, so the peak current in the inductor

must be calculated. The peak current (I_LPK) in the inductor is calculated as:

I_LPK= I_F+ Δi_L/2

(15)

When the designed maximum output current is reduced, the ratio r can be increased. At a current of 0.2A, r can

be made as high as 0.7. The ripple ratio can be increased at lighter loads because the net ripple is actually quite

low, and if r remains constant the inductor value can be made quite large. An equation empirically developed for

the maximum ripple ratio at any current below 2A is:

-0.3667

r = 0.387 x I_OUT

(16)

Note that this is just a guideline.

The LM3405A operates at a high frequency allowing the use of ceramic output capacitors without compromising

transient response. Ceramic capacitors allow higher inductor ripple without significantly increasing LED current

ripple. See the output capacitor and feed-forward capacitor sections for more details on LED current ripple.

Now that the ripple current or ripple ratio is determined, the inductance is calculated by:

V_OUT+ V_D1

x (1-D)

L =

I_Fx r x f_SW

(17)

where f_SWis the switching frequency and I_Fis the LED current. When selecting an inductor, make sure that it is

capable of supporting the peak output current without saturating. Inductor saturation will result in a sudden

reduction in inductance and prevent the regulator from operating correctly. Because of the operating frequency of

the LM3405A, ferrite based inductors are preferred to minimize core losses. This presents little restriction since

the variety of ferrite based inductors is huge. Lastly, inductors with lower series resistance (DCR) will provide

better operating efficiency. For recommended inductor selection, refer to Circuit Examples and Recommended

Inductance Range in Table 1.

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Table 1. Recommended Inductance Range

I_F

Inductance Range and Inductor Current Ripple

6.8µH-15µH

1.0A

Inductance

6.8µH

51%

10µH

36%

15µH

24%

(1)

Δi_L/ I_F

10µH-22µH

0.6A

0.2A

Inductance

10µH

58%

15µH

39%

22µH

26%

(1)

Δi_L/ I_F

15µH-27µH

Inductance

15µH

116%

22µH

79%

27µH

65%

(1)

Δi_L/ I_F

(1) Maximum over full range of V_INand V_OUT

INPUT CAPACITOR (C1)

An input capacitor is necessary to ensure that V_INdoes not drop excessively during switching transients. The

primary specifications of the input capacitor are capacitance, voltage rating, RMS current rating, and ESL

(Equivalent Series Inductance). The input voltage rating is specifically stated by the capacitor manufacturer.

Make sure to check any recommended deratings and also verify if there is any significant change in capacitance

at the operating input voltage and the operating temperature. The input capacitor maximum RMS input current

rating (I_RMS-IN) must be greater than:

r²

I_RMS-IN= I_Fx

D x

1 - D +

(18)

It can be shown from the above equation that maximum RMS capacitor current occurs when D = 0.5. Always

calculate the RMS at the point where the duty cycle D, is closest to 0.5. The ESL of an input capacitor is usually

determined by the effective cross sectional area of the current path. A large leaded capacitor will have high ESL

and an 0805 ceramic chip capacitor will have very low ESL. At the operating frequency of the LM3405A, certain

capacitors may have an ESL so large that the resulting inductive impedance (2πfL) will be higher than that

required to provide stable operation. It is strongly recommended to use ceramic capacitors due to their low ESR

and low ESL. A 10µF multilayer ceramic capacitor (MLCC) is a good choice for most applications. In cases

where large capacitance is required, use surface mount capacitors such as Tantalum capacitors and place at

least a 1µF ceramic capacitor close to the V_INpin. For MLCCs it is recommended to use X7R or X5R dielectrics.

Consult capacitor manufacturer datasheet to see how rated capacitance varies over operating conditions.

OUTPUT CAPACITOR (C2)

The output capacitor is selected based upon the desired reduction in LED current ripple. A 1µF ceramic capacitor

results in very low LED current ripple for most applications. Due to the high switching frequency, the 1µF

capacitor alone (without feed-forward capacitor C4) can filter more than 90% of the inductor current ripple for

most applications where the sum of LED dynamic resistance and R1 is larger than 1Ω. Since the internal

compensation is tailored for small output capacitance with very low ESR, it is strongly recommended to use a

ceramic capacitor with capacitance less than 3.3µF.

Given the availability and quality of MLCCs and the expected output voltage of designs using the LM3405A,

there is really no need to review other capacitor technologies. A benefit of ceramic capacitors is their ability to

bypass high frequency noise. A certain amount of switching edge noise will couple through the parasitic

capacitances in the inductor to the output. A ceramic capacitor will bypass this noise. In cases where large

capacitance is required, use Electrolytic or Tantalum capacitors with large ESR, and verify the loop performance

on the bench. Like the input capacitor, multilayer ceramic capacitors are recommended X7R or X5R. Again,

verify actual capacitance at the desired operating voltage and temperature.

Check the RMS current rating of the capacitor. The maximum RMS current rating of the capacitor is:

I_RMS-OUT= I_Fx

(19)

One may select a 1206 size ceramic capacitor for C2 since its current rating is typically higher than 1A, more

than enough for the requirement.

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FEED-FORWARD CAPACITOR (C4)

The feed-forward capacitor (designated as C4) connected in parallel with the LED string is required to provide

multiple benefits to the LED driver design. It greatly improves the large signal transient response and suppresses

LED current overshoot that may otherwise occur during PWM dimming; it also helps to shape the rise and fall

times of the LED current pulse during PWM dimming thus reducing EMI emission; it reduces LED current ripple

by bypassing some of inductor ripple from flowing through the LED. For most applications, a 1µF ceramic

capacitor is sufficient. In fact, the combination of a 1µF feed-forward ceramic capacitor and a 1µF output ceramic

capacitor leads to less than 1% current ripple flowing through the LED. Lower and higher C4 values can be used,

but bench validation is required to ensure the performance meets the application requirement.

Figure 24 shows a typical LED current waveform during PWM dimming without feed-forward capacitor. At the

beginning of each PWM cycle, overshoot can be seen in the LED current. Adding a 1µF feed-forward capacitor

can totally remove the overshoot as shown in Figure 25.

Figure 24. PWM Dimming without Feed-Forward

Capacitor

Figure 25. PWM Dimming with a 1µF Feed-Forward

Capacitor

CATCH DIODE (D1)

The catch diode (D1) conducts during the switch off-time. A Schottky diode is required for its fast switching time

and low forward voltage drop. The catch diode should be chosen such that its current rating is greater than:

I_D1= I_Fx (1-D)

(20)

The reverse breakdown rating of the diode must be at least the maximum input voltage plus appropriate margin.

To improve efficiency, choose a Schottky diode with a low forward voltage drop.

BOOST DIODE (D2)

A standard diode such as the 1N4148 type is recommended. For V_BOOSTcircuits derived from voltages less than

3.3V, a small-signal Schottky diode is recommended for better efficiency. A good choice is the BAT54 small

signal diode.

BOOST CAPACITOR (C3)

A 0.01µF ceramic capacitor with a voltage rating of at least 6.3V is sufficient. The X7R and X5R MLCCs provide

the best performance.

POWER LOSS ESTIMATION

The main power loss in LM3405A includes three basic types of loss in the internal power switch: conduction loss,

switching loss, and gate charge loss. In addition, there is loss associated with the power required for the internal

circuitry of IC.

The conduction loss is calculated as:

(21)

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If the inductor ripple current is fairly small (for example, less than 40%) , the conduction loss can be simplified to:

P_COND= I_F²x R_DS(ON)x D

(22)

The switching loss occurs during the switch on and off transition periods, where voltage and current overlap

resulting in power loss. The simplest means to determine this loss is to empirically measure the rise and fall

times (10% to 90%) of the voltage at the switch pin.

Switching power loss is calculated as follows:

P_SW= 0.5 x V_INx I_Fx f_SWx ( T_RISE+ T_FALL

)

(23)

(24)

(25)

The gate charge loss is associated with the gate charge Q_Grequired to drive the switch:

P_G= f_SWx V_INx Q_G

The power loss required for operation of the internal circuitry:

P_Q= I_Qx V_IN

I_Qis the quiescent operating current, and is typically around 1.8mA for the LM3405A.

The total power loss in the IC is:

P_INTERNAL= P_COND+ P_SW+ P_G+ P_Q

(26)

An example of power losses for a typical application is shown in Table 2:

Table 2. Power Loss Tabulation

Conditions

Power loss

V_IN

V_OUT

I_OUT

V_D1

12V

3.9V

1.0A

0.45V

300mΩ

R_DS(ON)

f_SW

P_COND

108mW

1.6MHz

18ns

T_RISE

T_FALL

I_Q

P_SW

288mW

12ns

1.8mA

1.4nC

P_Q

P_G

22mW

27mW

Q_G

D is calculated to be 0.36

Σ ( P_COND+ P_SW+ P_Q+ P_G) = P_INTERNAL

(27)

(28)

P_INTERNAL= 445mW

PCB Layout Considerations

When planning the layout there are a few things to consider when trying to achieve a clean, regulated output.

The most important consideration when completing the layout is the close coupling of the GND connections of

the input capacitor C1 and the catch diode D1. These ground ends should be close to one another and be

connected to the GND plane with at least two through-holes. Place these components as close to the IC as

possible. The next consideration is the location of the GND connection of the output capacitor C2, which should

be near the GND connections of C1 and D1.

There should be a continuous ground plane on the bottom layer of a two-layer board except under the switching

node island.

The FB pin is a high impedance node and care should be taken to make the FB trace short to avoid noise pickup

that causes inaccurate regulation. The LED current setting resistor R1 should be placed as close as possible to

the IC, with the GND of R1 placed as close as possible to the GND of the IC. The V_OUTtrace to LED anode

should be routed away from the inductor and any other traces that are switching.

High AC currents flow through the V_IN, SW and V_OUTtraces, so they should be as short and wide as possible.

Radiated noise can be decreased by choosing a shielded inductor.

The remaining components should also be placed as close as possible to the IC. See Application Report AN-

1229 (SNVA054) for further considerations and the LM3405A demo board as an example of a four-layer layout.

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LM3405A Circuit Examples

BOOST

OUT

LM3405A

LED1

DC or

PWM

EN/DIM

GND

Figure 26. V_BOOSTderived from V_IN

(V_IN= 5V, I_F= 1A)

Table 3. Bill of Materials for Figure 26

Part ID

Part Value

Part Number

Manufacturer

1A LED Driver

LM3405A

Texas Instruments

Semiconductor

C1, Input Cap

C2, Output Cap

C3, Boost Cap

C4, Feedforward Cap

D1, Catch Diode

D2, Boost Diode

10µF, 6.3V, X5R

C3216X5R0J106M

GRM319R71A105KC01D

0805YC103KAT2A

GRM319R71A105KC01D

MBRM110LT1G

TDK

1µF, 10V, X7R

Murata

0.01µF, 16V, X7R

1µF, 10V, X7R

AVX

Murata

Schottky, 0.37V at 1A, V_R= 10V

Schottky, 0.36V at 15mA

4.7µH, 1.6A

ON Semiconductor

CMDSH-3

Central Semiconductor

SLF6028T-4R7M1R6

WSL2010R2000FEA

LXK2-PW14

TDK

0.2Ω, 0.5W, 1%

Vishay

Lumileds

LED1

1.5A, White LED

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BOOST

OUT

LM3405A

LED1

DC or

PWM

EN/DIM

GND

Figure 27. V_BOOSTderived from V_OUT

(V_IN= 12V, I_F= 1A)

Table 4. Bill of Materials for Figure 27

Part ID

Part Value

Part Number

Manufacturer

1A LED Driver

LM3405A

Texas Instruments

Panasonic

Murata

C1, Input Cap

C2, Output Cap

C3, Boost Cap

C4, Feedforward Cap

D1, Catch Diode

D2, Boost Diode

10µF, 25V, X5R

ECJ-3YB1E106K

GRM319R71A105KC01D

0805YC103KAT2A

GRM319R71A105KC01D

SS13

1µF, 10V, X7R

0.01µF, 16V, X7R

1µF, 10V, X7R

AVX

Murata

Schottky, 0.5V at 1A, V_R= 30V

Schottky, 0.36V at 15mA

4.7µH, 1.6A

Vishay

CMDSH-3

Central Semiconductor

TDK

SLF6028T-4R7M1R6

WSL2010R2000FEA

LXK2-PW14

0.2Ω, 0.5W, 1%

Vishay

LED1

1.5A, White LED

Lumileds

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BOOST

OUT

LM3405A

LED1

DC or

PWM

EN/DIM

GND

Figure 28. V_BOOSTderived from V_INthrough a Shunt Zener Diode (D3) (V_IN= 18V, I_F= 1A)

Table 5. Bill of Materials for Figure 28

Part ID

Part Value

1A LED Driver

Part Number

Manufacturer

LM3405A

Texas Instruments

Semiconductor

Panasonic

Murata

C1, Input Cap

C2, Output Cap

C3, Boost Cap

C4, Feedforward Cap

C5, Shunt Cap

D1, Catch Diode

D2, Boost Diode

D3, Zener Diode

10µF, 25V, X5R

ECJ-3YB1E106K

GRM319R71A105KC01D

0805YC103KAT2A

GRM319R71A105KC01D

GRM219R71C104KA01D

SS13

1µF, 10V, X7R

0.01µF, 16V, X7R

1µF, 10V, X7R

AVX

Murata

0.1µF, 16V, X7R

Schottky, 0.5V at 1A, V_R= 30V

Schottky, 0.36V at 15mA

4.7V, 350mW, SOT-23

6.8µH, 1.5A

Murata

Vishay

CMDSH-3

Central Semiconductor

Fairchild

TDK

BZX84C4V7

SLF6028T-6R8M1R5

WSL2010R2000FEA

CRCW08051K91FKEA

LXK2-PW14

0.2Ω, 0.5W, 1%

Vishay

1.91kΩ, 1%

Vishay

LED1

1.5A, White LED

Lumileds

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BOOST

OUT

LM3405A

LED1

DC or

PWM

EN/DIM

GND

Figure 29. V_BOOSTderived from V_INthrough a Series Zener Diode (D3)

(V_IN= 15V, I_F= 1A)

Table 6. Bill of Materials for Figure 29

Part ID

Part Value

1A LED Driver

Part Number

Manufacturer

Texas Instruments

LM3405A

C1, Input Cap

C2, Output Cap

C3, Boost Cap

C4, Feedforward Cap

D1, Catch Diode

D2, Boost Diode

D3, Zener Diode

10µF, 25V, X5R

ECJ-3YB1E106K

GRM319R71A105KC01D

0805YC103KAT2A

GRM319R71A105KC01D

SS13

Panasonic

Murata

1µF, 10V, X7R

0.01µF, 16V, X7R

1µF, 10V, X7R

AVX

Murata

Schottky, 0.5V at 1A, V_R= 30V

Schottky, 0.36V at 15mA

11V, 350mW, SOT-23

6.8µH, 1.5A

Vishay

CMDSH-3

Central Semiconductor

Fairchild

TDK

BZX84C11

SLF6028T-6R8M1R5

WSL2010R2000FEA

LXK2-PW14

0.2Ω, 0.5W, 1%

Vishay

LED1

1.5A, White LED

Lumileds

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BOOST

OUT

LM3405A

LED1

LED2

DC or

PWM

EN/DIM

GND

Figure 30. V_BOOSTderived from V_OUTthrough a Series Zener Diode (D3)

( V_IN= 18V, I_F= 1A )

Table 7. Bill of Materials for Figure 30

Part ID

Part Value

1A LED Driver

Part Number

Manufacturer

Texas Instruments

LM3405A

C1, Input Cap

C2, Output Cap

C3, Boost Cap

C4, Feedforward Cap

D1, Catch Diode

D2, Boost Diode

D3, Zener Diode

10µF, 25V, X5R

ECJ-3YB1E106K

GRM319R71A105KC01D

0805YC103KAT2A

GRM319R71A105KC01D

SS13

Panasonic

Murata

1µF, 16V, X7R

0.01µF, 16V, X7R

1µF, 16V, X7R

AVX

Murata

Schottky, 0.5V at 1A, V_R= 30V

Schottky, 0.36V at 15mA

3.6V, 350mW, SOT-23

6.8µH, 1.5A

Vishay

CMDSH-3

Central Semiconductor

Fairchild

TDK

BZX84C3V6

SLF6028T-6R8M1R5

WSL2010R2000FEA

LXK2-PW14

0.2Ω, 0.5W, 1%

Vishay

LED1

1.5A, White LED

Lumileds

LED2

1.5A, White LED

LXK2-PW14

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Figure 31. LED MR16 Lamp Application

( V_IN= 12V AC, I_F= 0.75A )

Table 8. Bill of Materials for Figure 31

Part ID

Part Value

1A LED Driver

Part Number

Manufacturer

Texas Instruments

LM3405A

C1, Input Cap

C2, Output Cap

C3, Boost Cap

C5, Input Cap

D1, Catch Diode

D2, Boost Diode

D3, Rectifier Diode

D4, Rectifier Diode

D5, Rectifier Diode

D6, Rectifier Diode

10µF, 25V, X5R

ECJ-3YB1E106K

GRM319R71A105KC01D

0805YC103KAT2A

ECE-A1EN221U

SS13

Panasonic

1µF, 10V, X7R

Murata

0.01µF, 16V, X7R

AVX

220µF, 25V, electrolytic

Schottky, 0.5V at 1A, V_R= 30V

Schottky, 0.36V at 15mA

Schottky, 0.385V at 500mA

6.8µH, 1.5A

Panasonic

Vishay

CMDSH-3

Central Semiconductor

TDK

CMHSH5-2L

SLF6028T-6R8M1R5

ERJ8BQFR27

LXHL-PW09

0.27Ω, 0.33W, 1%

Panasonic

LED1

1A, White LED

Lumileds

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REVISION HISTORY

Changes from Revision B (May 2013) to Revision C

Page

•

Changed layout of National Data Sheet to TI format .......................................................................................................... 23

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PACKAGE OPTION ADDENDUM

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2-May-2013

PACKAGING INFORMATION

Orderable Device

LM3405AXMK/NOPB

LM3405AXMKE/NOPB

LM3405AXMKX/NOPB

LM3405AXMY/NOPB

LM3405AXMYX/NOPB

Status Package Type Package Pins Package

Eco Plan Lead/Ball Finish

MSL Peak Temp

Op Temp (°C)

-40 to 125

Top-Side Markings

Samples

Drawing

Qty

(1)

(2)

(3)

(4)

ACTIVE

SOT

DDC

1000

Green (RoHS

& no Sb/Br)

CU SN

Level-1-260C-UNLIM

SSEB

ACTIVE

DDC

DGN

250

Green (RoHS

& no Sb/Br)

-40 to 125

SSEB

SVSA

3000

1000

3500

Green (RoHS

& no Sb/Br)

-40 to 125

MSOP-

PowerPAD

Green (RoHS

& no Sb/Br)

MSOP-

Green (RoHS

& no Sb/Br)

PowerPAD

⁽¹⁾The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs.

LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.

NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available.

OBSOLETE: TI has discontinued the production of the device.

⁽²⁾Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability

information and additional product content details.

TBD: The Pb-Free/Green conversion plan has not been defined.

Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that

lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.

Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between

the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.

Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight

in homogeneous material)

⁽³⁾MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.

(4)

Multiple Top-Side Markings will be inside parentheses. Only one Top-Side Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a

continuation of the previous line and the two combined represent the entire Top-Side Marking for that device.

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information

provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and

continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.

TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.

Addendum-Page 1

PACKAGE OPTION ADDENDUM

www.ti.com

2-May-2013

In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

Addendum-Page 2

PACKAGE MATERIALS INFORMATION

www.ti.com

11-Oct-2013

TAPE AND REEL INFORMATION

*All dimensions are nominal

Device

Package Package Pins

Type Drawing

SPQ

Reel

Pin1

Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant

(mm) W1 (mm)

LM3405AXMK/NOPB

LM3405AXMKE/NOPB

LM3405AXMKX/NOPB

LM3405AXMY/NOPB

SOT

DDC

DGN

1000

250

178.0

8.4

3.2

5.3

3.2

3.4

1.4

4.0

8.0

3000

1000

8.4

8.0

MSOP-

Power

PAD

12.4

12.0

LM3405AXMYX/NOPB

MSOP-

Power

PAD

DGN

3500

330.0

12.4

5.3

3.4

1.4

8.0

12.0

Pack Materials-Page 1

PACKAGE MATERIALS INFORMATION

www.ti.com

11-Oct-2013

*All dimensions are nominal

Device

Package Type Package Drawing Pins

SPQ

Length (mm) Width (mm) Height (mm)

LM3405AXMK/NOPB

LM3405AXMKE/NOPB

LM3405AXMKX/NOPB

LM3405AXMY/NOPB

SOT

DDC

DGN

1000

250

210.0

367.0

185.0

367.0

35.0

SOT

3000

1000

3500

MSOP-PowerPAD

LM3405AXMYX/NOPB MSOP-PowerPAD

Pack Materials-Page 2

MECHANICAL DATA

DGN0008A

MUY08A (Rev A)

BOTTOM VIEW

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TI has specifically designated certain components as meeting ISO/TS16949 requirements, mainly for automotive use. In any case of use of

non-designated products, TI will not be responsible for any failure to meet ISO/TS16949.

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TI E2E Community

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Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265

LM3405AXMKE/NOPB [TI]

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