LM6211MDC [TI]
IC,OP-AMP,SINGLE,CMOS,DIE;
| 型号： | LM6211MDC |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | IC,OP-AMP,SINGLE,CMOS,DIE 放大器 |
| 文件： | 总20页 (文件大小：1140K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
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June 2006
LM6211
Low Noise, RRO Operational Amplifier with CMOS Input
and 24V Operation
General Description
Features
The LM6211 is a wide bandwidth, low noise op amp with a
wide supply voltage range and a low input bias current. The
LM6211 operates with a single supply voltage of 5V to 24V,
is unity gain stable, has a ground-sensing CMOS input
stage, and offers rail-to-rail output swing.
(Typical 24V supply unless otherwise noted)
n Supply voltage range
n Input referred voltage noise
n Unity gain bandwidth
n 1/f corner frequency
n Slew rate
5V to 24V
5.5 nV/
20 MHz
400 Hz
5.6 V/µs
1.05 mA
5.5 pF
The LM6211 is designed to provide optimal performance in
high voltage, low noise systems. The LM6211 has a unity
gain bandwidth of 20 MHz and an input referred voltage
n Supply current
n Low input capacitance
n Temperature range
n Total harmonic distortion
n Output short circuit current
noise density of 5.5 nV/
at 10 kHz. The LM6211
-40˚C to 125˚C
0.01% 1 kHz, 600Ω
achieves these specifications with a low supply current of
only 1 mA. The LM6211 has a low input bias current of
2.3 pA, an output short circuit current of 25 mA and a slew
rate of 5.6 V/us. The LM6211 also features a low common-
mode input capacitance of 5.5 pF which makes it ideal for
use in wide bandwidth and high gain circuits. The LM6211 is
well suited for low noise applications that require an op amp
with very low input bias currents and a large output voltage
swing, like active loop-filters for wide-band PLLs. A low total
harmonic distortion, 0.01% at 1 kHz with loads as high as
600Ω, also makes the LM6211 ideal for high fidelity audio
and microphone amplifiers.
@
25 mA
Applications
n PLL loop filters
n Low noise active filters
n Strain gauge amplifiers
n Low noise microphone amplifiers
The LM6211 is available in the small SOT package, allowing
the user to implement ultra-small and cost effective board
layouts.
Typical Application
20120303
20120304
© 2006 National Semiconductor Corporation
DS201203
www.national.com
Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Junction Temperature (Note 3)
Soldering Information
+150˚C
Infrared or Convection (20 sec)
Wave Soldering Lead Temp. (10 sec)
235˚C
260˚C
ESD Tolerance (Note 2)
Human Body Model
Machine Model
2000V
Operating Ratings (Note 1)
Temperature Range
Supply Voltage (VS = V+ – V−)
200V
0.3V
−40˚C to +125˚C
5V to 24V
VIN Differential
Supply Voltage (VS = V+ – V−)
Voltage at Input/Output pins
Storage Temperature Range
25V
Package Thermal Resistance (θJA (Note 3))
V+ +0.3V, V− −0.3V
5-Pin SOT23
178˚C/W
−65˚C to +150˚C
5V Electrical Characteristics (Note 4)
Unless otherwise specified, all limits are guaranteed for TA = 25˚C, V+ = 5V, V− = 0V, VCM = VO = V+/2. Boldface limits apply
at the temperature extremes.
Symbol
VOS
Parameter
Conditions
Min
Typ
Max
Units
(Note 6) (Note 5) (Note 6)
Input Offset Voltage
VCM = 0.5V
0.1
2.5
mV
2.8
TC VOS
IB
Input Offset Average Drift
Input Bias Current
VCM = 0.5V (Note 7)
2
µV/C
pA
VCM = 0.5V (Notes 8, 9)
0.5
5
10
nA
IOS
Input Offset Current
Common Mode Rejection
Ratio
VCM = 0.5V
0.1
98
pA
CMRR
0 V ≤ VCM ≤ 3V
0.4 V ≤ VCM ≤ 2.3 V
83
70
85
78
80
0
dB
dB
PSRR
Power Supply Rejection Ratio V+ = 5V to 24V, VCM = 0.5V
98
95
V+ = 4.5V to 25V, VCM = 0.5V
CMVR
AVOL
Input Common-Mode Voltage
Range
CMRR ≥ 65 dB
CMRR ≥ 60 dB
VO = 0.35V to 4.65, RL = 2 kΩ to V+/2
3.3
V
0
2.4
Large Signal Voltage Gain
82
80
85
82
110
110
50
dB
VO = 0.25V to 4.75, RL = 10 kΩ to V+/2
RL = 2 kΩ to V+/2
VO
Output Swing High
Output Swing Low
Output Short Circuit Current
Supply Current
150
165
85
RL = 10 kΩ to V+/2
20
90
mV from
rail
RL = 2 kΩ to V+/2
39
150
170
85
RL = 10 kΩ to V+/2
13
90
IOUT
Sourcing to V+/2
13
10
20
10
16
VID = 100 mV (Note 10)
Sinking to V+/2
mA
mA
30
VID = −100 mV (Note 10)
IS
0.96
1.10
1.25
SR
Slew Rate
AV = +1, 10% to 90% (Note 11)
5.5
17
V/µs
MHz
GBW
en
Gain Bandwidth Product
Input-Referred Voltage Noise
f = 10 kHz
f = 1 kHz
f = 1 kHz
5.5
6.0
0.01
nV/
pA/
in
Input-Referred Current Noise
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2
5V Electrical Characteristics (Note 4) (Continued)
Unless otherwise specified, all limits are guaranteed for TA = 25˚C, V+ = 5V, V− = 0V, VCM = VO = V+/2. Boldface limits apply
at the temperature extremes.
Symbol
Parameter
Conditions
Min
Typ
Max
Units
(Note 6) (Note 5) (Note 6)
0.01
THD
Total Harmonic Distortion
AV = 2, RL = 600Ω to V+/2
%
24V Electrical Characteristics (Note 4)
Unless otherwise specified, all limits are guaranteed for TA = 25˚C, V+ = 24V, V− = 0V, VCM = VO = V+/2. Boldface limits apply
at the temperature extremes.
Symbol
VOS
Parameter
Conditions
Min
Typ
Max
Units
(Note 6) (Note 5) (Note 6)
Input Offset Voltage
VCM = 0.5V
0.25
2.7
mV
3.0
TC VOS
IB
Input Offset Average Drift
Input Bias Current
VCM = 0.5V (Note 7)
2
2
µV/C
pA
VCM = 0.5V (Notes 8, 9)
25
10
nA
IOS
Input Offset Current
Common Mode Rejection
Ratio
VCM = 0.5V
0.1
pA
CMRR
0 ≤ VCM ≤ 21V
0.4 ≤ VCM ≤ 20V
85
70
85
78
80
0
105
dB
dB
V
PSRR
Power Supply Rejection Ratio V+ = 5V to 24V, VCM = 0.5V
98
98
V+ = 4.5V to 25V, VCM = 0.5V
CMVR
AVOL
Input Common-Mode Voltage
Range
CMRR ≥ 65 dB
CMRR ≥ 60 dB
VO = 1.5V to 22.5V, RL = 2 kΩ to V+/2
21.5
0
20.5
Large Signal Voltage Gain
82
77
85
82
120
120
212
48
dB
VO = 1V to 23V, RL = 10 kΩ to V+/2
RL = 2 kΩ to V+/2
VO
Output Swing High
400
520
150
165
350
420
150
170
RL = 10 kΩ to V+/2
mV from
rail
Output Swing Low
RL = 2 kΩ to V+/2
150
38
RL = 10 kΩ to V+/2
IOUT
Output Short Circuit Current
Sourcing to V+/2
20
15
30
20
25
VID = 100 mV (Note 10)
Sinking to V+/2
mA
38
VID = −100 mV (Note 10)
IS
Supply Current
Slew Rate
1.05
5.6
1.25
mA
1.40
SR
AV = +1, VO = 18 VPP
10% to 90% (Note 11)
V/µs
GBW
en
Gain Bandwidth Product
20
5.5
MHz
nV/
Input-Referred Voltage Noise
f = 10 kHz
f = 1 kHz
6.0
in
Input-Referred Current Noise
Total Harmonic Distortion
f = 1 kHz
AV = 2, RL = 2 kΩ to V+/2
0.01
0.01
pA/
%
THD
3
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Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see the Electrical Characteristics Tables.
Note 2: Human Body Model is 1.5 kΩ in series with 100 pF. Machine Model is 0Ω in series with 200 pF.
Note 3: The maximum power dissipation is a function of T
, θ , and T . The maximum allowable power dissipation at any ambient temperature is
JA A
J(MAX)
P
= (T
- T )/θ . All numbers apply for packages soldered directly onto a PC board.
D
J(MAX) A JA
Note 4: Electrical table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very limited self-heating of
the device.
Note 5: Typical values represent the most likely parametric norm at the time of characterization.
Note 6: Limits are 100% production tested at 25˚C. Limits over the operating temperature range are guaranteed through correlations using the Statistical Quality
Control (SQC) method.
Note 7: Offset voltage average drift is determined by dividing the change in V
Note 8: Positive current corresponds to current flowing into the device.
Note 9: Input bias current is guaranteed by design.
at the temperature extremes into the total temperature change.
OS
Note 10: The device is short circuit protected and can source or sink its limit currents continuously. However, care should be taken such that when the output is
driving short circuit currents, the inputs do not see more than 0.3V differential voltage.
Note 11: Slew rate is the average of the rising and falling slew rates.
Connection Diagram
5-Pin SOT23
20120301
Top View
Ordering Information
Package
Part Number
LM6211MF
Package Marking
Transport Media
NSC Drawing
1k Units Tape and Reel
3k Units Tape and Reel
5-Pin SOT-23
AT1A
MF05A
LM6211MFX
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Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, VS = 24V, V+ = VS,
V− = 0 V, VCM = VS/2.
Supply Current vs. Supply Voltage
VOS vs. Supply Voltage
20120319
20120321
20120351
20120318
VOS vs. VCM
VOS vs. VCM
20120320
Input Bias Current vs. VCM
Input Bias Current vs. VCM
20120350
5
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Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, VS = 24V, V+ = VS,
V− = 0 V, VCM = VS/2. (Continued)
Input Bias Current vs. VCM
Input Bias Current vs. VCM
20120352
20120353
Sourcing Current vs. Supply Voltage
Sinking Current vs. Supply Voltage
20120334
20120333
Positive Output Swing vs. Supply Voltage
Negative Output Swing vs. Supply Voltage
20120330
20120332
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Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, VS = 24V, V+ = VS,
V− = 0 V, VCM = VS/2. (Continued)
Positive Output Swing vs. Supply Voltage
Negative Output Swing vs. Supply Voltage
20120331
20120329
Sourcing Current vs. Output Voltage
Sinking Current vs. Output Voltage
20120328
20120327
Sourcing Current vs. Output Voltage
Sinking Current vs. Output Voltage
20120325
20120326
7
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Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, VS = 24V, V+ = VS,
V− = 0 V, VCM = VS/2. (Continued)
Open Loop Gain and Phase with Resistive Load
Open Loop Gain and Phase with Capacitive Load
20120309
20120308
Input Referred Voltage Noise vs. Frequency
THD+N vs. Frequency
20120304
20120317
THD+N vs. Output Amplitude
THD+N vs. Output Amplitude
20120315
20120316
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Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, VS = 24V, V+ = VS,
V− = 0 V, VCM = VS/2. (Continued)
Slew Rate vs. Supply Voltage
Overshoot and Undershoot vs. Capacitive Load
20120313
20120314
Small Signal Transient Response
Large Signal Transient Response
20120322
20120324
Phase Margin vs. Capacitive Load (Stability)
Phase Margin vs. Capacitive Load (Stability)
20120310
20120311
9
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Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, VS = 24V, V+ = VS,
V− = 0 V, VCM = VS/2. (Continued)
Closed Loop Output Impedance vs. Frequency
PSRR vs. Frequency
20120305
20120312
CMRR vs. Frequency
20120306
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10
should be taken to prevent the inputs from seeing more than
0.3V differential voltage, which is the absolute maximum
differential input voltage.
Application Notes
ADVANTAGES OF THE LM6211
Small Size
High Supply Voltage, Low Power Operation
The small footprint of the LM6211 package saves space on
printed circuit boards, and enables the design of smaller and
more compact electronic products. Long traces between the
signal source and the op amp make the signal path suscep-
tible to noise. By using a physically smaller package, the
LM6211 can be placed closer to the signal source, reducing
noise pickup and enhancing signal integrity
The LM6211 has performance guaranteed at supply volt-
ages of 5V and 24V. The LM6211 is guaranteed to be
operational at all supply voltages between 5V and 24V. In
this large range of operation, the LM6211 draws a fairly
constant supply current of 1 mA, while providing a wide
bandwidth of 20 MHz. The wide operating range makes the
LM6211 a versatile choice for a variety of applications rang-
ing from portable instrumentation to industrial control sys-
tems.
STABILITY OF OP AMP CIRCUITS
Stability and Capacitive Loading
Low Input Referred Noise
The LM6211 is designed to be unity gain stable for moderate
capacitive loads, around 100 pF. That is, if connected in a
unity gain buffer configuration, the LM6211 will resist oscil-
lation unless the capacitive load is higher than about 100 pF.
For higher capacitive loads, the phase margin of the op amp
reduces significantly and it tends to oscillate. This is because
an op amp cannot be designed to be stable for high capaci-
tive loads without either sacrificing bandwidth or supplying
higher current. Hence, for driving higher capacitive loads,
the LM6211 needs to be externally compensated.
The LM6211 has very low flatband input referred voltage
noise, 5.5 nV/
. The 1/f corner frequency, also very low,
is about 400 Hz. The CMOS input stage allows for an
extremely low input current (2 pA) and a very low input
referred current noise (0.01 pA/
). This allows the
LM6211 to maintain signal fidelity and makes it ideal for
audio, wireless or sensor based applications.
Low Input Bias Current and High Input Impedance
The LM6211 has a CMOS input stage, which allows it to
have very high input impedance, very small input bias cur-
rents (2 pA) and extremely low input referred current noise
(0.01 pA/
). This level of performance is essential for op
amps used in sensor applications, which deal with extremely
low currents of the order of a few nanoamperes. In this case,
the op amp is being driven by a sensor, which typically has a
source impedance of tens of MΩ. This makes it essential for
the op amp to have a much higher impedance.
Low Input Capacitance
The LM6211 has a comparatively small input capacitance for
a high voltage CMOS design. Low input capacitance is very
beneficial in terms of driving large feedback resistors, re-
quired for higher closed loop gain. Usually, high voltage
CMOS input stages have a large input capacitance, which
when used in a typical gain configuration, interacts with the
feedback resistance to create an extra pole. The extra pole
causes gain-peaking and can compromise the stability of the
op amp. The LM6211 can, however, be used with larger
resistors due to its smaller input capacitance, and hence
provide more gain without compromising stability. This also
makes the LM6211 ideal for wideband transimpedance am-
plifiers, which require a wide bandwidth, low input referred
noise and low input capacitance.
20120337
FIGURE 1. Gain vs. Frequency for an Op Amp
An op amp, ideally, has a dominant pole close to DC, which
causes its gain to decay at the rate of 20 dB/decade with
respect to frequency. If this rate of decay, also known as the
rate of closure (ROC), remains at 20 dB/decade at the unity
gain bandwidth of the op amp, the op amp is stable. If,
however, a large capacitance is added to the output of the op
amp, it combines with the output impedance of the op amp to
create another pole in its frequency response before its unity
gain frequency (Figure 1). This increases the ROC to
40 dB/decade and causes instability.
RRO, Ground Sensing and Current Limiting
The LM6211 has a rail-to-rail output stage, which provides
the maximum possible output dynamic range. This is espe-
cially important for applications requiring a large output
swing, like wideband PLL synthesizers which need an active
loop filter to drive a wide frequency range VCO. The input
common mode range includes the negative supply rail which
allows direct sensing at ground in a single supply operation.
The LM6211 also has a short circuit protection circuit which
limits the output current to about 25 mA sourcing and 38 mA
sinking, and allows the LM6211 to drive short circuit loads
indefinitely. However, while driving short circuit loads care
In such a case a number of techniques can be used to
restore stability to the circuit. The idea behind all these
schemes is to modify the frequency response such that it
can be restored to a ROC of 20 dB/decade, which ensures
stability.
11
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Compensation by External Resistor
Application Notes (Continued)
In some applications it is essential to drive a capacitive load
without sacrificing bandwidth. In such a case, in the loop
compensation is not viable. A simpler scheme for compen-
sation is shown in Figure 3. A resistor, RISO, is placed in
series between the load capacitance and the output. This
introduces a zero in the circuit transfer function, which coun-
teracts the effect of the pole formed by the load capacitance,
and ensures stability.
In the Loop Compensation
Figure 2 illustrates a compensation technique, known as ‘in
the loop’ compensation, that employs an RC feedback circuit
within the feedback loop to stabilize a non-inverting amplifier
configuration. A small series resistance, RS, is used to iso-
late the amplifier output from the load capacitance, CL, and a
small capacitance, CF, is inserted across the feedback re-
sistor to bypass CL at higher frequencies.
20120356
FIGURE 3. Compensation By Isolation Resistor
The value of RISO to be used should be decided depending
on the size of CL and the level of performance desired.
Values ranging from 5Ω to 50Ω are usually sufficient to
ensure stability. A larger value of RISO will result in a system
with lesser ringing and overshoot, but will also limit the
output swing and the short circuit current of the circuit.
20120338
FIGURE 2. In the Loop Compensation
The values for RS and CF are decided by ensuring that the
zero attributed to CF lies at the same frequency as the pole
attributed to CL. This ensures that the effect of the second
pole on the transfer function is compensated for by the
presence of the zero, and that the ROC is maintained at
20 dB/decade. For the circuit shown in Figure 2 the values of
RS and CF are given by Equation (1). Table 1 shows different
values of RS and CF that need to be used for maintaining
stability with different values of CL, as well as the phase
margins to be expected. RF and RIN are assumed to be 10
kΩ, RL is taken as 2 kΩ, while ROUT is taken to be 60Ω.
Stability and Input Capacitance
In certain applications, for example I-V conversion, transim-
pedance photodiode amplification and buffering the output of
current-output DAC, capacitive loading at the input of the op
amp can endanger stability. The capacitance of the source
driving the op amp, the op amp input capacitance and the
parasitic/wiring capacitance contribute to the loading of the
input. This capacitance, CIN, interacts with the feedback
network to introduce a peaking in the closed loop gain of the
circuit, and hence causes instability.
(1)
TABLE 1.
20120349
CL (pF)
250
RS (Ω)
60
CF (pF)
4.5
Phase Margin (˚)
39.8
49.5
53.1
FIGURE 4. Compensating for Input Capacitance
300
60
5.4
This peaking can be eliminated by adding a feedback ca-
pacitance, CF, as shown in Figure 4. This introduces a zero
in the feedback network, and hence a pole in the closed loop
response, and thus maintains stability. An optimal value of
CF is given by Equation (2). A simpler approach is to select
CF = (R1/R2)CIN for a 90˚ phase margin. This approach,
however, limits the bandwidth excessively.
500
60
9
Although this methodology provides circuit stability for any
load capacitance, it does so at the price of bandwidth. The
closed loop bandwidth of the circuit is now limited by RS and
CF.
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Certain performance characteristics are essential for an op
amp if it is to be used in a PLL loop filter. Low input referred
voltage and current noise are essential, as they directly
affect the noise of the filter and hence the phase noise of the
PLL. Low input bias current is also important, as bias current
affects the level of ‘reference spurs’, artifacts in the fre-
quency spectrum of the PLL caused by mismatch or leakage
at the output of the phase detector. A large input and output
swing is beneficial in terms of increasing the flexibility in
biasing the op amp. The op amp can then be biased such
that the output range of the PLL is mapped efficiently onto
the input range of the VCO.
Typical Applications
ACTIVE LOOP FILTER FOR PLLs
A typical phase locked loop, or PLL, functions by creating a
negative feedback loop in terms of the phase of a signal. A
simple PLL consists of three main components: a phase
detector, a loop filter and a voltage controlled oscillator
(VCO). The phase detector compares the phase of the out-
put of the PLL with that of a reference signal, and feeds the
error signal into the loop filter, thus performing negative
feedback. The loop filter performs the important function of
averaging (or low-pass filtering) the error and providing the
VCO with a DC voltage, which allows the VCO to modify its
frequency such that the error is minimized. The performance
of the loop filter affects a number of specifications of the PLL,
like its frequency range, locking time and phase noise.
With a CMOS input, ultra low input bias currents (2 pA) and
low input referred voltage noise (5.5 nV/
), the LM6211
is an ideal op amp for using in a PLL active loop filter. The
LM6211 has a ground sensing input stage, a rail-to-rail out-
put stage, and an operating supply range of 5V - 24V, which
makes it a versatile choice for the design of a wide variety of
active loop filters.
Since a loop filter is a very noise sensitive application, it is
usually suggested that only passive components be used in
its design. Any active devices, like discrete transistors or op
amps, would add significantly to the noise of the circuit and
would hence worsen the in-band phase noise of the PLL. But
newer and faster PLLs, like National’s LMX2430, have a
power supply voltage of less than 3V, which limits the phase-
detector output of the PLL. If a passive loop filter is used with
such circuits, then the DC voltage that can be provided to the
VCO is limited to couple of volts. This limits the range of
frequencies for which the VCO, and hence the PLL, is func-
tional. In certain applications requiring a wider operating
range of frequencies for the PLL, like set-top boxes or base
stations, this level of performance is not adequate and re-
quires active amplification, hence the need for active loop
filters.
Figure 7 shows the LM6211 used with the LMX2430 to
create an RF frequency synthesizer. The LMX2430 detects
the PLL output, compares it with its internal reference clock
and outputs the phase error in terms of current spikes. The
LM6211 is used to create a loop filter which averages the
error and provides a DC voltage to the VCO. The VCO
generates a sine wave at a frequency determined by the DC
voltage at its input. This circuit can provide output signal
frequencies as high as 2 GHz, much higher than a compara-
tive passive loop filter. Compared to a similar passive loop
filter, the LM6211 doesn’t add significantly to the phase noise
of the PLL, except at the edge of the loop bandwidth, as
shown in Figure 6. A peaking of loop gain is expected, since
the loop filter is deliberately designed to have a wide band-
width and a low phase margin so as to minimize locking time.
An active loop filter typically consists of an op amp, which
provides the gain, accompanied by a three or four pole RC
filter. The non-inverting input of the op amp is biased to a
fixed value, usually the mid-supply of the PLL, while a feed-
back network provides the gain as well as one, or two, poles
for low pass filtering. Figure 5 illustrates a typical active loop
filter.
20120303
20120355
FIGURE 5. A Typical Active Loop Filter
FIGURE 6. Effect of LM6211 on Phase Noise of PLL
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Typical Applications (Continued)
20120336
FIGURE 7. LM6211 in the Active Loop Filter for LMX2430
ADC INPUT DRIVER
measured in Effective Number of Bits or ENOB, is only
reduced by 0.3 bits, despite amplifying the input signal by a
gain of 10. Low input bias currents and high input impedance
also help as they prevent the loading of the sensor and allow
the measurement system to function over a large range.
A typical application for a high performance op amp is as an
ADC driver, which delivers the analog signal obtained from
sensors and actuators to ADCs for conversion to the digital
domain and further processing. Important requirements in
this application are a slew rate high enough to drive the ADC
input and low input referred voltage and current noise. If an
op amp is used with an ADC, it is critical that the op amp
noise does not affect the dynamic range of the ADC. The
LM6211, with low input referred voltage and current noise,
provides a great solution for this application. For example,
the LM6211 can be used to drive an ADS121021, a 12-bit
ADC from National. If it provides a gain of 10 to a maximum
input signal amplitude of 100 mV, for a bandwidth as wide as
100 kHz, the average noise seen at the input of the ADC is
only 44.6 µVrms. Hence the dynamic range of the ADC,
Figure 8 shows a circuit for monitoring fluid pressure in a
hydraulic system, in which the LM6211 is used to sense the
error voltage from the pressure sensor. Two LM6211 ampli-
fiers are used to make a difference amplifier which senses
the error signal, amplifies it by a gain of 100, and delivers it
to the ADC input. The ADC converts the error voltage into a
pressure reading to be displayed and drives the DAC, which
changes the voltage driving the resistance bridge sensor.
This is used to control the gain of the pressure measurement
circuit, such that the range of the sensor can be modified to
obtain the best resolution possible.
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Typical Applications (Continued)
20120335
FIGURE 8. Hydraulic Pressure Monitoring System
DAC OUTPUT AMPLIFIER
system performance to improve without any significant deg-
radation of the settling time.
Op amps are often used to improve a DAC’s output driving
capability. High performance op amps are required as I-V
converters at the outputs of high resolution current output
DACs. Since most DACs operate with a single supply of 5V,
a rail-to-rail output swing is essential for this application. A
low offset voltage is also necessary to prevent offset errors in
the waveform generated. Also, the output impedance of
DACs is quite high, more than a few kΩ in some cases, so it
is also advisable for the op amp to have a low input bias
current. An op amp with a high input impedance also pre-
vents the loading of the DAC, and hence, avoids gain errors.
The op amp should also have a slew rate which is fast
enough to not affect the settling time of the DAC output.
The LM6211, with a CMOS input stage, ultra low input bias
current, a wide bandwidth (20 MHz) and a rail-to-rail output
swing for a supply voltage of 24V is an ideal op amp for such
an application. Figure 9 shows a typical circuit for this appli-
cation. The op amp is usually expected to add another time
constant to the system, which worsens the settling time, but
the wide bandwidth of the LM6211 (20 MHz) allows the
20120340
FIGURE 9. DAC Driver Circuit
15
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Typical Applications (Continued)
AUDIO PREAMPLIFIER
With low input referred voltage noise, low supply voltage and
low supply current, and low harmonic distortion, the LM6211
is ideal for audio applications. Its wide unity gain bandwidth
allows it to provide large gain over a wide frequency range
and it can be used to design a preamplifier to drive a load of
as low as 600Ω with less than 0.001% distortion. Two am-
plifier circuits are shown in Figure 10 and Figure 11. Figure
10 is an inverting amplifier, with a 10 kΩ feedback resistor,
R2, and a 1 kΩ input resistor, R1, and hence provides a gain
of −10. Figure 11 is a non-inverting amplifier, using the same
values for R1 and R2, and provides a gain of 11. In either of
these circuits, the coupling capacitor CC1 decides the lower
frequency at which the circuit starts providing gain, while the
feedback capacitor CF decides the frequency at which the
gain starts dropping off. Figure 12 shows the frequency
response of the circuit in Figure 10 with different values of
CF.
20120343
FIGURE 12. Frequency Response of the Non-Inverting
Preamplifier
TRANSIMPEDANCE AMPLIFIER
A transimpedance amplifier converts a small input current
into a voltage. This current is usually generated by a photo-
diode. The transimpedance gain, measured as the ratio of
the output voltage to the input current, is expected to be
large and wide-band. Since the circuit deals with currents in
the range of a few nA, low noise performance is essential.
The LM6211, being a CMOS input op amp, provides a wide
bandwidth and low noise performance while drawing very
low input bias current, and is hence ideal for transimpedance
applications.
A transimpedance amplifier is designed on the basis of the
current source driving the input. A photodiode is a very
common capacitive current source, which requires transim-
pedance gain for transforming its miniscule current into eas-
ily detectable voltages. The photodiode and amplifier’s gain
are selected with respect to the speed and accuracy re-
quired of the circuit. A faster circuit would require a photo-
diode with lesser capacitance and a faster amplifier. A more
sensitive circuit would require a sensitive photodiode and a
high gain. A typical transimpedance amplifier is shown in
Figure 13. The output voltage of the amplifier is given by the
equation VOUT = −IINRF. Since the output swing of the am-
plifier is limited, RF should be selected such that all possible
values of IIN can be detected.
20120341
FIGURE 10. Inverting Audio Amplifier
The LM6211 has a large gain-bandwidth product (20 MHz),
which enables high gains at wide bandwidths. A rail-to-rail
output swing at 24V supply allows detection and amplifica-
tion of a wide range of input currents. A CMOS input stage
with negligible input current noise and low input voltage
noise allows the LM6211 to provide high fidelity amplification
for wide bandwidths. These properties make the LM6211
ideal for systems requiring wide-band transimpedance am-
plification.
20120342
FIGURE 11. Non-Inverting Audio Preamplifier
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16
An essential component for obtaining a maximally flat re-
sponse is the feedback capacitor, CF. The capacitance seen
at the input of the amplifier, CIN, combined with the feedback
resistor, RF, generates a phase lag which causes gain-
peaking and can destabilize the circuit. CIN is usually just the
sum of CD and CCM. The feedback capacitor CF creates a
pole, fP in the noise gain of the circuit, which neutralizes the
zero in the noise gain, fZ, created by the combination of RF
and CIN. If properly positioned, the noise gain pole created
by CF can ensure that the slope of the gain remains at
20 dB/decade till the unity gain frequency of the amplifier is
reached, thus ensuring stability. As shown in Figure 16, fP is
positioned such that it coincides with the point where the
noise gain intersects the op amp’s open loop gain. In this
case, fP is also the overall 3 dB frequency of the transim-
pedance amplifier. The value of CF needed to make it so is
given by Equation (2). A larger value of CF causes excessive
reduction of bandwidth, while a smaller value fails to prevent
gain peaking and maintain stability.
Typical Applications (Continued)
20120344
FIGURE 13. Photodiode Transimpedance Amplifier
(2)
The following parameters are used to design a transimped-
ance amplifier: the amplifier gain-bandwidth product, A0; the
amplifier input capacitance, CCM; the photodiode capaci-
tance, CD; the transimpedance gain required, RF; and the
amplifier output swing. Once a feasible RF is selected using
the amplifier output swing, these numbers can be used to
design an amplifier with the desired transimpedance gain
Calculating CF from Equation (2) can sometimes return un-
reasonably small values ( 1 pF), especially for high speed
<
applications. In these cases, it is often more practical to use
the circuit shown in Figure 15 in order to allow more reason-
able values. In this circuit, the capacitance CF’ is (1+ RB/RA)
times the effective feedback capacitance, CF. A larger ca-
pacitor can now be used in this circuit to obtain a smaller
effective capacitance.
and
a maximally flat frequency response. The input
common-mode capacitance with respect to VCM for the
LM6211 is give in Figure 14.
20120347
FIGURE 15. Modifying CF
20120354
For example, if a CF of 0.5 pF is needed, while only a 5 pF
capacitor is available, RB and RA can be selected such that
RB/RA = 9. This would convert a CF’ of 5 pF into a
FIGURE 14. Input Common-Mode Capacitance vs. VCM
<<
CF of 0.5 pF. This relationship holds as long as RA
RF
17
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Typical Applications (Continued)
20120346
FIGURE 16. Method for CF selection
SENSOR INTERFACES
The low input bias current and low input referred noise of the
LM6211 make it ideal for sensor interfaces. These circuits
are required to sense voltages of the order of a few µV, and
currents amounting to less than a nA, and hence the op amp
needs to have low voltage noise and low input bias current.
Typical applications include infra-red (IR) thermometry, ther-
mocouple amplifiers and pH electrode buffers. Figure 17 is
an example of a typical circuit used for measuring IR radia-
tion intensity, often used for estimating the temperature of an
object from a distance. The IR sensor generates a voltage
proportional to I, which is the intensity of the IR radiation
falling on it. As shown in Figure 17, K is the constant of
20120348
proportionality relating the voltage across the IR sensor (VIN
)
to the radiation intensity, I. The resistances RA and RB are
selected to provide a high gain to amplify this voltage, while
CF is added to filter out the high frequency noise.
FIGURE 17. IR Radiation Sensor
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18
Physical Dimensions inches (millimeters) unless otherwise noted
5-Pin SOT23
NS Package Number MF05A
National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves
the right at any time without notice to change said circuitry and specifications.
For the most current product information visit us at www.national.com.
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device or system whose failure to perform can be reasonably
expected to cause the failure of the life support device or
system, or to affect its safety or effectiveness.
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相关型号：
LM6211MFX/NOPB
IC OP-AMP, 2800 uV OFFSET-MAX, 17 MHz BAND WIDTH, PDSO5, SOT-23, 5 PIN, Operational Amplifier
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