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LMH2100

SNWS020C –NOVEMBER 2007–REVISED OCTOBER 2015

LMH2100 50-MHz to 4-GHz 40-dB Logarithmic Power Detector for CDMA and WCDMA

1 Features

3 Description

The LMH2100 is a 40-dB RF power detector intended

for use in CDMA and WCDMA applications. The

device has an RF frequency range from 50 MHz to

4 GHz. It provides an accurate temperature and

supply compensated output voltage that relates

linearly to the RF input power in dBm. The circuit

operates with a single supply from 2.7 V to 3.3 V.

1

•

Supply Voltage: 2.7 V to 3.3 V

Output Voltage: 0.3 V to 2 V

40-dB Linear in dB Power Detection Range

Shutdown

Multi-Band Operation from 50 MHz to 4 GHz

0.5-dB Accurate Temperature Compensation

External Configurable Output Filter Bandwidth

0.4-mm Pitch DSBGA Package

The LMH2100 has an RF power detection range from

−45 dBm to −5 dBm, and is ideally suited for direct

use in combination with a 30-dB directional coupler.

Additional low-pass filtering of the output signal can

be realized by means of an external resistor and

capacitor. Typical Application: Output RC Low Pass

Filter shows a detector with an additional output low

pass filter. The filter frequency is set with R_Sand C_S.

2 Applications

•

UMTS/CDMA/WCDMA RF Power Control

GSM/GPRS RF Power Control

PA Modules

Typical Application: Feedback (R)C Low Pass Filter

shows a detector with an additional feedback low

pass filter. Resistor R_Pis optional and will lower the

Trans impedance gain (R_TRANS). The filter frequency

IEEE 802.11b, g (WLAN)

is set with C_P//C_TRANSand R_P//R_TRANS

.

The device is active for Enable = High; otherwise it is

in a low power-consumption shutdown mode. To save

power and prevent discharge of an external filter

capacitance, the output (OUT) is high-impedance

during shutdown.

Device Information⁽¹⁾

PART NUMBER

PACKAGE

BODY SIZE (MAX)

LMH2100

DSBGA (6)

1.274 mm × 0.874 mm

(1) For all available packages, see the orderable addendum at

the end of the data sheet.

Typical Application: Output RC Low Pass Filter

Typical Application: Feedback (R)C Low Pass

Filter

COUPLER

ANTENNA

RF

PA

COUPLER

ANTENNA

RF

PA

50 W

VDD

50 W

VDD

R_S

RFIN

EN

OUT

REF

1

6

+

-

2

RFIN

EN

C_S

OUT

ADC

1

LMH2100

2

6

+

-

LMH2100

R_P

C_P

ADC

4

5

3

GND

REF

4

5

3

GND

1

An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,

intellectual property matters and other important disclaimers. PRODUCTION DATA.

LMH2100

SNWS020C –NOVEMBER 2007–REVISED OCTOBER 2015

www.ti.com

Table of Contents

7.3 Feature Description................................................. 23

7.4 Device Functional Modes........................................ 29

Application and Implementation ........................ 30

8.1 Application Information............................................ 30

8.2 Typical Applications ............................................... 33

Power Supply Recommendations...................... 39

1

2

3

4

5

6

Features.................................................................. 1

Applications ........................................................... 1

Description ............................................................. 1

Revision History..................................................... 2

Pin Configuration and Functions......................... 3

Specifications......................................................... 4

6.1 Absolute Maximum Ratings ..................................... 4

6.2 ESD Ratings.............................................................. 4

6.3 Recommended Operating Ratings ........................... 4

6.4 Thermal Information.................................................. 5

6.5 2.7-V DC and AC Electrical Characteristics.............. 5

6.6 Timing Requirements.............................................. 11

6.7 Typical Characteristics............................................ 11

Detailed Description ............................................ 23

7.1 Overview ................................................................. 23

7.2 Functional Block Diagram ....................................... 23

8

9

10 Layout................................................................... 40

10.1 Layout Guidelines ................................................ 40

10.2 Layout Example .................................................... 42

11 Device and Documentation Support ................. 43

11.1 Community Resources.......................................... 43

11.2 Trademarks........................................................... 43

11.3 Electrostatic Discharge Caution............................ 43

11.4 Glossary................................................................ 43

7

12 Mechanical, Packaging, and Orderable

Information ........................................................... 43

4 Revision History

NOTE: Page numbers for previous revisions may differ from page numbers in the current version.

Changes from Revision B (March 2013) to Revision C

Page

•

Added Device Information and Pin Configuration and Functions sections, ESD Ratings table, Feature Description,

Device Functional Modes, Application and Implementation, Power Supply Recommendations, Layout, Device and

Documentation Support, and Mechanical, Packaging, and Orderable Information sections. ................................................ 1

Changes from Revision A (March 2013) to Revision B

Page

•

Changed layout of National Data Sheet to TI format ........................................................................................................... 42

2

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5 Pin Configuration and Functions

YFQ Package

6-Pin DSBGA

Top View

A2

OUT

A1

VDD

B2

REF

B1

RFIN

C2

EN

C1

GND

Pin Functions

I/O

DESCRIPTION

NUMBER

NAME

VDD

A1

A2

B1

Power Supply

Output

Positive supply voltage

OUT

Ground referenced detector output voltage (linear in dB)

RFIN

Analog Input

RF input signal to the detector, internally terminated with 50 Ω.

Reference output, for differential output measurement (without pedestal). Connected

to inverting input of output amplifier.

B2

C1

C2

REF

GND

EN

Reference Output

GND

Power ground

The device is enabled for EN = High, and brought to a low-power shutdown mode for

EN = Low.

Logic Input

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6 Specifications

6.1 Absolute Maximum Ratings

over operating free-air temperature range (unless otherwise noted)⁽¹⁾⁽²⁾

MIN

MAX

3.6

UNIT

V

Supply voltage, V_DDto GND

RF input, input power

RF input, DC voltage

Enable input voltage

10

dBm

mV

V

400

V_SS– 0.4 < V_EN< V_DD+ 0.4

(3)

Junction temperature

150

260

°C

Maximum lead temperature (soldering,10 sec)

Storage temperature, T_stg

°C

−65

150

°C

(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings

only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended

Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. For ensured

specifications and the test conditions, see the 2.7-V DC and AC Electrical Characteristics.

(2) If Military/Aerospace specified devices are required, contact the Texas Instruments Sales Office/ Distributors for availability and

specifications.

(3) The maximum power dissipation is a function of T_J(MAX), R_θJA. The maximum allowable power dissipation at any ambient temperature is

P_D= (T_J(MAX)– T_A)/R_θJA. All numbers apply for packages soldered directly into a PC board.

6.2 ESD Ratings

VALUE

±2000

±200

UNIT

Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001⁽¹⁾

Charged-device model (CDM), per JEDEC specification JESD22-C101⁽²⁾

Machine model

V_(ESD)

Electrostatic discharge

V

(1) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.

(2) JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.

6.3 Recommended Operating Ratings

over operating free-air temperature range (unless otherwise noted)⁽¹⁾

MIN

2.7

–40

50

NOM

MAX

3.3

UNIT

V

Supply voltage

Temperature range

RF frequency range

85

°C

4000

MHz

–45

–58

–5

–18

dBm

dBV

RF input power range⁽²⁾

(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings

only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended

Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. For ensured

specifications and the test conditions, see the 2.7-V DC and AC Electrical Characteristics.

(2) Power in dBV = dBm + 13 when the impedance is 50 Ω.

4

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6.4 Thermal Information

LMH2100

THERMAL METRIC⁽¹⁾

YFQ (DSBGA)

6 PINS

133.7

1.7

UNIT

(2)

R_θJA

R_θJC(top)

R_θJB

ψ_JT

Junction-to-ambient thermal resistance

°C/W

Junction-to-case (top) thermal resistance

Junction-to-board thermal resistance

22.6

Junction-to-top characterization parameter

Junction-to-board characterization parameter

5.7

ψ_JB

22.2

(1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application

report, SPRA953.

(2) The maximum power dissipation is a function of T_J(MAX), R_θJA. The maximum allowable power dissipation at any ambient temperature is

P_D= (T_J(MAX)- T_A)/R_θJA. All numbers apply for packages soldered directly into a PC board.

6.5 2.7-V DC and AC Electrical Characteristics

Unless otherwise specified, all limits are ensured at T_A= 25°C, V_DD= 2.7 V, RF input frequency ƒ = 1855 MHz CW

(Continuous Wave, unmodulated). Maximum and minimum limits apply at the temperature extremes.⁽¹⁾

.

(2)

PARAMETER

SUPPLY INTERFACE

I_DDSupply current

TEST CONDITIONS

MIN

TYP⁽³⁾

MAX

UNIT

Active mode: EN = High, no signal present

at RF_IN

6.3

7.1

7.9

9.2

0.9

1.9

10

mA

Active mode: EN = High, no signal present

at RF_IN

5

T_A= –40°C to +85°C

Shutdown: EN = Low, no signal present at

0.5

RF_IN

Shutdown: EN = Low, no signal present at

RF_IN

.

µA

T_A= –40°C to +85°C

EN = Low: P_IN= 0 dBm⁽⁴⁾

T_A= –40°C to +85°C

LOGIC ENABLE INTERFACE

V_LOW

EN logic low input level (Shutdown T_A= –40°C to +85°C

Mode)

0.6

V

V_HIGH

I_EN

EN logic high input level

Current into EN pin

T_A= –40°C to +85°C

1.1

V

60

nA

RF INPUT INTERFACE

R_INInput resistance

46.7

51.5

56.4

Ω

(1) 2.7-V DC and AC Electrical Characteristics values apply only for factory testing conditions at the temperature indicated. Factory testing

conditions result in very limited self-heating of the device such that T_J= T_A. No specification of parametric performance is indicated in

the electrical tables under conditions of internal self-heating where T_J> T_A.

(2) All limits are ensured by test or statistical analysis.

(3) Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary

over time and will also depend on the application and configuration. The typical values are not tested and are not specified on shipped

production material.

(4) All limits are ensured by design and measurements which are performed on a limited number of samples. Limits represent the mean

±3–sigma values.

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2.7-V DC and AC Electrical Characteristics (continued)

Unless otherwise specified, all limits are ensured at T_A= 25°C, V_DD= 2.7 V, RF input frequency ƒ = 1855 MHz CW

(Continuous Wave, unmodulated). Maximum and minimum limits apply at the temperature extremes.⁽¹⁾

.

(2)

PARAMETER

OUTPUT INTERFACE

TEST CONDITIONS

MIN

TYP⁽³⁾

MAX

UNIT

From positive rail, sourcing,

V_REF= 0 V, I_OUT= 1 mA

15.3

23.9

28.9

22.3

28.3

From positive rail, sourcing,

V_REF= 0 V, I_OUT= 1 mA

T_A= –40°C to +85°C

V_OUT

Output voltage swing

mV

From negative rail, sinking,

V_REF= 2.7 V, I_OUT= 1 mA

13.1

From negative rail, sinking,

V_REF= 2.7 V, I_OUT= 1 mA

T_A= –40°C to +85°C

Sourcing, V_REF= 0 V, V_OUT= 2.6 V

5.8

7.3

8.3

Sourcing, V_REF= 0 V, V_OUT= 2.6 V

T_A= –40°C to +85°C

5.2

6.2

5.4

I_OUT

Output short circuit current

mA

Sinking, V_REF= 2.7 V, V_OUT= 0.1 V

T_A= –40°C to +85°C

BW

Small signal bandwidth

No RF input signal. Measured from REF

input current to V_OUT

416

kHz

R_TRANS

Output amp transimpedance gain

No RF input signal, from I_REFto V_OUT, DC

Positive, V_REFfrom 2.7 V to 0 V

40.7

3.4

43.3

3.9

46.7

kΩ

Positive, V_REFfrom 2.7 V to 0 V

T_A= –40°C to +85°C

3.3

3.8

3.7

SR

Slew rate

V/µs

Negative, V_REFfrom 0 V to 2.7 V

4.4

0.2

Negative, V_REFfrom 0 V to 2.7 V

T_A= –40°C to +85°C

No RF input signal, EN = High, DC

measurement

1.8

4

R_OUT

Output impedance⁽⁵⁾

Ω

No RF input signal, EN = High, DC

measurement

I_OUT,SD

Output leakage current in

shutdown mode

EN = Low, V_OUT= 2 V

T_A= –40°C to +85°C

100

nA

RF DETECTOR TRANSFER

ƒ = 50 MHz, MIN and MAX at T_A= –40°C

to +85°C

1.69

1.67

1.57

1.47

1.38

1.25

1.16

1.77

1.78

1.65

1.55

1.46

1.34

1.82

1.83

1.70

1.60

1.51

1.40

1.30

ƒ = 900 MHz, MIN and MAX at T_A= –40°C

to +85°C

ƒ = 1855 MHz, MIN and MAX at T_A

–40°C to +85°C

=

Maximum output voltage

ƒ = 2500 MHz, MIN and MAX at T_A

–40°C to +85°C

V_OUT,MAX

V

P_IN= −5 dBm⁽⁵⁾

ƒ = 3000 MHz, MIN and MAX at T_A

–40°C to +85°C

ƒ = 3500 MHz, MIN and MAX at T_A

–40°C to +85°C

ƒ = 4000 MHz, MIN and MAX at T_A

–40°C to +85°C

1.25

266

No input signal

207

173

324

365

V_OUT,MINMinimum output voltage (pedestal)

mV

No input signal, T_A= –40°C to +85°C

(5) All limits are ensured by design and measurements which are performed on a limited number of samples. Limits represent the mean

±3–sigma values. The typical value represents the statistical mean value.

6

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2.7-V DC and AC Electrical Characteristics (continued)

Unless otherwise specified, all limits are ensured at T_A= 25°C, V_DD= 2.7 V, RF input frequency ƒ = 1855 MHz CW

(Continuous Wave, unmodulated). Maximum and minimum limits apply at the temperature extremes.⁽¹⁾

.

(2)

PARAMETER

TEST CONDITIONS

MIN

TYP⁽³⁾

MAX

UNIT

ƒ = 50 MHz, MIN and MAX at T_A= –40°C

to +85°C

1.38

1.44

1.49

1.46

1.36

1.27

1.19

1.1

ƒ = 900 MHz, MIN and MAX at T_A= –40°C

to +85°C

1.34

1.27

1.19

1.11

1

1.43

1.32

1.23

1.16

1.05

0.97

ƒ = 1855 MHz, MIN and MAX at T_A

–40°C to +85°C

=

Output voltage range

ƒ = 2500 MHz, MIN and MAX at T_A

–40°C to +85°C

ΔV_OUT

V

P_INfrom −45 dBm to −5 dBm⁽⁵⁾

ƒ = 3000 MHz, MIN and MAX at T_A

–40°C to +85°C

ƒ = 3500 MHz, MIN and MAX at T_A

–40°C to +85°C

ƒ = 4000 MHz, MIN and MAX at T_A

–40°C to +85°C

0.91

1.01

ƒ = 50 MHz

39.6

37.0

40.9

38.2

42.1

39.4

36.5

ƒ = 900 MHz

ƒ = 1855 MHz

ƒ = 2500 MHz

ƒ = 3000 MHz

ƒ = 3500 MHz

f = 4000 MHz

ƒ = 50 MHz

34.5

35.5

K_SLOPE

Logarithmic slope⁽⁵⁾

32.7

33.7

34.6 mV/dB

33.1

31.1

32.1

29.7

30.7

31.6

28.5

29.4

30.3

–50.2

–53.6

–53.2

–52.4

–51.2

–49.1

–47.3

−49.5

−52.7

−52.3

−51.2

−50.1

−47.8

−46.1

1.5

–48.8

ƒ = 900 MHz

–51.8

ƒ = 1855 MHz

ƒ = 2500 MHz

ƒ = 3000 MHz

ƒ = 3500 MHz

ƒ = 4000 MHz

P_IN= −10 dBm at 10 kHz

–51.4

P_INT

Logarithmic intercept⁽⁵⁾

–50.1

–48.9

–46.4

–44.9

dBm

e_n

v_N

Output referred noise⁽⁶⁾

Output referred noise⁽⁵⁾

µV/√Hz

Integrated over frequency band, 1 kHz to

6.5 kHz

100

µV_RMS

Integrated over frequency band, 1 kHz to

6.5 kHz

150

T_A= –40°C to +85°C

P_IN= −10 dBm, ƒ = 1800 MHz

60

PSRR

Power supply rejection ratio⁽⁶⁾

dB

P_IN= −10 dBm, ƒ = 1800 MHz

T_A= –40°C to +85°C

55

(6) This parameter is ensured by design and/or characterization and is not tested in production.

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2.7-V DC and AC Electrical Characteristics (continued)

Unless otherwise specified, all limits are ensured at T_A= 25°C, V_DD= 2.7 V, RF input frequency ƒ = 1855 MHz CW

(Continuous Wave, unmodulated). Maximum and minimum limits apply at the temperature extremes.⁽¹⁾

.

(2)

PARAMETER

TEST CONDITIONS

MIN

TYP⁽³⁾

MAX

UNIT

POWER MEASUREMENT PERFORMANCE

ƒ = 50 MHz

–0.2

0.12

1.2

1.3

0.2

0.3

0.4

0.8

1.1

1.6

1.8

3.3

3.5

4.6

4.9

ƒ = 50 MHz, MIN and MAX at T_A= –40°C

to +85°C

–0.8

–0.4

–1

ƒ = 900 MHz

–0.06

-0.03

0.04

0.13

0.35

0.65

ƒ = 900 MHz, MIN and MAX at T_A= –40°C

to +85°C

ƒ = 1855 MHz

–0.3

–0.7

–0.2

–0.8

–0.1

1

ƒ = 1855 MHz, MIN and MAX at T_A

–40°C to +85°C

=

ƒ = 2500 MHz

Log conformance error⁽⁵⁾

E_LC

dB

ƒ = 2500 MHz, MIN and MAX at T_A

–40°C to +85°C

−40 dBm ≤ P_IN≤ −10 dBm

ƒ = 3000 MHz

ƒ = 3000 MHz, MIN and MAX at T_A

–40°C to +85°C

ƒ = 3500 MHz

-0.036

–1

ƒ = 3500 MHz, MIN and MAX at T_A

–40°C to +85°C

ƒ = 4000 MHz

–0.048

–1

ƒ = 4000 MHz, MIN and MAX at T_A

–40°C to +85°C

ƒ = 50 MHz, MIN and MAX at T_A= –40°C

to +85°C

–0.63

–0.94

0.43

0.30

ƒ = 900 MHz, MIN and MAX at T_A= –40°C

to +85°C

ƒ = 1855 MHz, MIN and MAX at T_A

–40°C to +85°C

=

–0.71

0.33

Variation over temperature⁽⁵⁾

E_VOT

ƒ = 2500 MHz, MIN and MAX at T_A

–40°C to +85°C

–0.88

0.35

dB

−40 dBm ≤ P_IN≤ −10 dBm

ƒ = 3000 MHz, MIN and MAX at T_A

–40°C to +85°C

–1.03

0.37

ƒ = 3500 MHz, MIN and MAX at T_A

–40°C to +85°C

–1.10

0.33

ƒ = 4000 MHz, MIN and MAX at T_A

–40°C to +85°C

–1.12

0.33

ƒ = 50 MHz, MIN and MAX at T_A= –40°C

to +85°C

–0.064

–0.123

–0.050

–0.058

–0.066

–0.082

–0.098

0.066

0.051

0.067

0.074

0.069

0.066

0.072

ƒ = 900 MHz, MIN and MAX at T_A= –40°C

to +85°C

ƒ = 1855 MHz, MIN and MAX at T_A

–40°C to +85°C

=

Measurement Error for a 1-dB

ƒ = 2500 MHz, MIN and MAX at T_A

–40°C to +85°C

E_{1 dB}

Input power step⁽⁵⁾

dB

−40 dBm ≤ P_IN≤ −10 dBm

ƒ = 3000 MHz, MIN and MAX at T_A

–40°C to +85°C

ƒ = 3500 MHz, MIN and MAX at T_A

–40°C to +85°C

ƒ = 4000 MHz, MIN and MAX at T_A

–40°C to +85°C

8

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2.7-V DC and AC Electrical Characteristics (continued)

Unless otherwise specified, all limits are ensured at T_A= 25°C, V_DD= 2.7 V, RF input frequency ƒ = 1855 MHz CW

(Continuous Wave, unmodulated). Maximum and minimum limits apply at the temperature extremes.⁽¹⁾

.

(2)

PARAMETER

TEST CONDITIONS

MIN

TYP⁽³⁾

MAX

UNIT

ƒ = 50 MHz, MIN and MAX at T_A= –40°C

to +85°C

–0.40

0.27

0.22

0.20

0.24

0.29

0.40

0.43

8.6

ƒ = 900 MHz, MIN and MAX at T_A= –40°C

to +85°C

–0.58

–0.29

–0.28

–0.38

–0.60

–0.82

–6.5

–4.7

–5.1

–4.3

–1.5

0.1

ƒ = 1855 MHz, MIN and MAX at T_A

–40°C to +85°C

=

Measurement Error for a 10-dB

ƒ = 2500 MHz, MIN and MAX at T_A

–40°C to +85°C

(5)

E_{10 dB}

Input power step

dB

−40 dBm ≤ P_IN≤ −10 dBm

ƒ = 3000 MHz, MIN and MAX at T_A

–40°C to +85°C

ƒ = 3500 MHz, MIN and MAX at T_A

–40°C to +85°C

ƒ = 4000 MHz, MIN and MAX at T_A

–40°C to +85°C

ƒ = 50 MHz, MIN and MAX at T_A= –40°C

to +85°C

ƒ = 900 MHz, MIN and MAX at T_A= –40°C

to +85°C

14.5

11.0

ƒ = 1855 MHz, MIN and MAX at T_A

–40°C to +85°C

=

Temperature sensitivity

–40°C < TA < 25°C

ƒ = 2500 MHz, MIN and MAX at T_A

–40°C to +85°C

S_T

13.6 mdB/°C

15.8

−40 dBm ≤ P_IN≤ −10 dBm⁽⁵⁾

ƒ = 3000 MHz, MIN and MAX at T_A

–40°C to +85°C

ƒ = 3500 MHz, MIN and MAX at T_A

–40°C to +85°C

16.9

ƒ = 4000 MHz, MIN and MAX at T_A

–40°C to +85°C

0.5

17.3

ƒ = 50 MHz, MIN at T_A= –40°C to +85°C

ƒ = 900 MHz, MIN at T_A= –40°C to +85°C

–10.5

0.5

2.6

ƒ = 1855 MHz, MIN at T_A= –40°C to

+85°C

–11.3

–10.6

–11.2

–12.9

–17.8

3.4

ƒ = 2500 MHz, MIN at T_A= –40°C to

+85°C

Temperature sensitivity

25°C < T_A< 85°C

5.8

S_T

mdB/°C

−40 dBm ≤ P_IN≤ −10 dBm⁽⁵⁾

ƒ = 3000 MHz, MIN at T_A= –40°C to

+85°C

6.1

5.5

ƒ = 3500 MHz, MIN at T_A= –40°C to

+85°C

ƒ = 4000 MHz, MIN at T_A= –40°C to

+85°C

ƒ = 50 MHz, MAX at T_A= –40°C to +85°C

ƒ = 900 MHz, MAX at T_A= –40°C to +85°C

–5.4

0.3

8.6

14.5

ƒ = 1855 MHz, MAX at T_A= –40°C to

+85°C

–3.1

–1.6

0.9

11.0

ƒ = 2500 MHz, MAX at T_A= –40°C to

+85°C

Temperature sensitivity

−40°C < T_A< 25°C⁽⁵⁾

P_IN= −10 dBm

13.6

S_T

mdB/°C

ƒ = 3000 MHz, MAX at T_A= –40°C to

+85°C

15.8

16.9

17.3

ƒ = 3500 MHz, MAX at T_A= –40°C to

+85°C

2.5

ƒ = 4000 MHz, MAX at T_A= –40°C to

+85°C

2.7

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2.7-V DC and AC Electrical Characteristics (continued)

Unless otherwise specified, all limits are ensured at T_A= 25°C, V_DD= 2.7 V, RF input frequency ƒ = 1855 MHz CW

(Continuous Wave, unmodulated). Maximum and minimum limits apply at the temperature extremes.⁽¹⁾

.

(2)

PARAMETER

TEST CONDITIONS

MIN

TYP⁽³⁾

MAX

UNIT

ƒ = 50 MHz, MIN and MAX at T_A= –40°C

to +85°C

–10.5

0.5

2.6

3.3

ƒ = 900 MHz, MIN and MAX at T_A= –40°C

to +85°C

–10.5

–11.3

–10.6

–11.2

–12.9

–17.8

ƒ = 1855 MHz, MIN and MAX at T_A

–40°C to +85°C

=

Temperature sensitivity

25°C < T_A< 85°C⁽⁵⁾

P_IN= −10 dBm

ƒ = 2500 MHz, MIN and MAX at T_A

–40°C to +85°C

S_T

5.4 mdB/°C

ƒ = 3000 MHz, MIN and MAX at T_A

–40°C to +85°C

6.1

4.4

ƒ = 3500 MHz, MIN and MAX at T_A

–40°C to +85°C

ƒ = 4000 MHz, MIN and MAX at T_A

–40°C to +85°C

–1.1

ƒ = 50 MHz, MIN at T_A= –40°C to +85°C

ƒ = 900 MHz, MIN at T_A= –40°C to +85°C

–9.2

–7.4

–8.6

–10.5

ƒ = 1855 MHz, MIN at T_A= –40°C to

+85°C

–8.2

-7.3

–6.5

–5.6

–4.4

–1.9

–7.2

ƒ = 2500 MHz, MIN at T_A= –40°C to

Maximum input power for E_LC= 1 +85°C

P_MAX

P_MIN

DR

dBm

dB

dB⁽⁵⁾

ƒ = 3000 MHz, MIN at T_A= –40°C to

+85°C

–6.3

–6.9

–11.1

ƒ = 3500 MHz, MIN at T_A= –40°C to

+85°C

ƒ = 4000 MHz, MIN at T_A= –40°C to

+85°C

ƒ = 50 MHz, MAX at T_A= –40°C to +85°C

ƒ = 900 MHz, MAX at T_A= –40°C to +85°C

–38.9

–43.1

–38.1

–42.3

ƒ = 1855 MHz, MAX at T_A= –40°C to

+85°C

–42.2

–40.6

–38.7

–35.9

–33.5

–41

-38.9

–37

ƒ = 2500 MHz, MAX at T_A= –40°C to

+85°C

Minimum input power for E_LC= 1

dB⁽⁵⁾

ƒ = 3000 MHz, MAX at T_A= –40°C to

+85°C

ƒ = 3500 MHz, MAX at T_A= –40°C to

+85°C

–34.7

–32

ƒ = 4000 MHz, MAX at T_A= –40°C to

+85°C

ƒ = 50 MHz, MIN at T_A= –40°C to +85°C

ƒ = 900 MHz, MIN at T_A= –40°C to +85°C

29.5

33.3

31.6

35.2

ƒ = 1855 MHz, MIN at T_A= –40°C to

+85°C

34.2

34.1

33.4

28.5

22.7

36.5

36.1

35.5

35.1

26.3

ƒ = 2500 MHz, MIN at T_A= –40°C to

+85°C

Dynamic range for E_LC= 1 dB⁽⁵⁾

ƒ = 3000 MHz, MIN at T_A= –40°C to

+85°C

ƒ = 3500 MHz, MIN at T_A= –40°C to

+85°C

ƒ = 4000 MHz, MIN at T_A= –40°C to

+85°C

10

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6.6 Timing Requirements

MIN

NOM

MAX UNIT

Turnon time, no signal at P_IN, Low-High transition EN, V_OUTto 90%

8.2

9.8

t_ON

µs

12

Turnon time, no signal at P_IN, Low-High transition EN, V_OUTto 90%

T_A= –40°C to +85°C

Rise time⁽¹⁾, P_IN= no signal to 0 dBm, V_OUTfrom 10% to 90%

T_A= –40°C to +85°C

2

t_R

µs

12

Fall time⁽¹⁾, P_IN= no signal to 0 dBm, V_OUTfrom 90% to 10%

T_A= –40°C to +85°C

t_F

µs

12

(1) This parameter is ensured by design and/or characterization and is not tested in production.

6.7 Typical Characteristics

Unless otherwise specified, V_DD= 2.7 V, T_A= 25°C, measured on a limited number of samples.

10

85°C

25°C

-40°C

8

85°C

6

25°C

-40°C

4

2

0

2

0

2.2

2.5

2.8

3.1

3.4

650

700

750

800

(mV)

850

900

SUPPLY VOLTAGE (V)

V

ENABLE

Figure 1. Supply Current vs Supply Voltage

Figure 2. Supply Current vs Enable Voltage

45

-38

-40°C

25°C

-40°C

-42

40

85°C

35

30

25

-46

25°C

85°C

-50

-54

10M

100M

1G

10G

100M

1G

10G

FREQUENCY (Hz)

Figure 3. Log Slope vs Frequency

Figure 4. Log Intercept vs Frequency

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7 V, T_A= 25°C, measured on a limited number of samples.

2.0

1.8

1.6

2.5

2.0

1.8

1.6

2.5

2.0

1.5

2.5

1.5

2.5

2.0

-40°C

25°C

1.8

2.0

1.0

1.8

2.0

1.0

1.4

1.6

1.4

1.6

1.4

1.5

-40°C

85°C

25°C

1.0

0.5

1.0

0.5

1.2

1.0

1.2

1.0

0.5

0.0

1.0

0.8

1.0

0.8

0.6

-0.5

85°C

-1.0

-0.5

0.6

-1.0

-0.5

0.8

0.4

0.2

-1.5

0.4

0.2

-1.5

-2.0

-1.0

-2.5

-2.0

-1.0

0.6

0.0

-2.5

-1.5

85°C

-15

85°C

0.4

-1.5

25°C

-40°C

-45

25°C

0.2

0.0

-2.0

0.2

0.0

-2.0

-40°C

-45

-2.5

5

-2.5

5

-55

-35

-25

-5

-55

-35

-25

-15

-5

RF INPUT POWER (dBm)

Figure 6. Mean Output Voltage and Log Conformance Error

Figure 5. Mean Output Voltage and Log Conformance Error

vs

RF Input Power at 900 MHz

RF Input Power at 50 MHz

2.0

1.8

2.5

2.0

1.8

1.6

2.5

2.0

1.6

2.0

1.5

2.5

1.5

2.5

2.0

1.8

2.0

1.0

1.8

2.0

1.0

1.4

1.6

1.4

1.6

1.4

1.5

25°C

1.0

0.5

1.0

0.5

1.2

1.0

1.2

1.0

0.5

0.0

1.0

0.8

1.0

0.8

0.6

-0.5

-1.0

0.6

-1.0

-0.5

0.8

-0.5

0.4

0.2

-1.5

0.4

0.2

-1.5

85°C

-15

85°C

-2.0

-1.0

-2.5

-2.0

0.6

-1.0

0.6

0.0

-2.5

-1.5

-40°C

25°C

-40°C

85°C

-25

0.4

-1.5

-2.0

25°C

-40°C

-45

85°C

0.2

0.0

-2.0

0.2

0.0

-40°C

-45

-2.5

5

-2.5

5

-55

-35

-5

-55

-35

-25

-15

-5

RF INPUT POWER (dBm)

Figure 7. Mean Output Voltage and Log Conformance Error

Figure 8. Mean Output Voltage and Log Conformance Error

vs

RF Input Power at 1855 MHz

RF Input Power at 2500 MHz

2.0

1.8

1.6

2.5

2.0

1.8

1.6

2.5

2.0

1.5

2.5

1.5

2.5

2.0

1.8

2.0

1.0

1.4

1.8

2.0

1.0

1.4

1.6

1.4

1.5

1.6

1.4

1.5

25°C

1.0

0.5

1.0

0.5

1.2

1.0

0.5

1.2

1.0

0.5

0.0

1.0

0.8

-0.5

0.8

0.6

-0.5

-40°C

0.6

-1.0

-0.5

-1.0

-0.5

0.8

0.4

0.2

-1.5

0.4

0.2

-1.5

85°C

-40°C

-2.0

-1.0

-2.0

0.6

-1.0

0.0

-2.5

-1.5

0.0

-2.5

-1.5

0.4

85°C

-25

85°C

0.2

0.0

0.2

0.0

-2.0

25°C

-35

25°C

-35

-40°C

-45

-40°C

-45

-2.5

5

-2.5

5

-55

-25

-15

-5

-55

-15

-5

RF INPUT POWER (dBm)

Figure 9. Mean Output Voltage and Log Conformance Error

Figure 10. Mean Output Voltage and Log Conformance Error

vs

RF Input Power at 3000 MHz

RF Input Power at 3500 MHz

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7 V, T_A= 25°C, measured on a limited number of samples.

2.0

2.5

1.8

2.0

1.6

2.0

1.5

2.5

1.8

1.4

2.0

1.0

1.6

1.5

25°C

1.4

1.2

1.0

0.5

1.2

0.5

1.0

0.0

0.8

-0.5

0.6

0.8

-1.0

-0.5

0.4

-1.5

85°C

-40°C

0.2

-2.0

0.6

-1.0

0.0

0.4

-2.5

-1.5

85°C

0.2

0.0

-2.0

25°C

-35

-40°C

-45

-2.5

5

-55

-25

-15

-5

RF INPUT POWER (dBm)

Figure 11. Mean Output Voltage and Log Conformance Error

Figure 12. Log Conformance Error (Mean ±3 sigma) vs

RF Input Power at 50 MHz

vs

RF Input Power at 4000 MHz

Figure 13. Log Conformance Error (Mean ±3 sigma) vs

RF Input Power at 900 MHz

Figure 14. Log Conformance Error (Mean ±3 sigma) vs

RF Input Power at 1855 MHz

Figure 16. Log Conformance Error (Mean ±3 sigma) vs

RF Input Power at 3000 MHz

Figure 15. Log Conformance Error (Mean ±3 sigma) vs

RF Input Power at 2500 MHz

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7 V, T_A= 25°C, measured on a limited number of samples.

Figure 18. Log Conformance Error (Mean ±3 sigma) vs

Figure 17. Log Conformance Error (Mean ±3 sigma) vs

RF Input Power at 4000 MHz

RF Input Power at 3500 MHz

1.5

1.0

1.5

1.0

-40°C

0.5

-40°C

0.0

-0.5

-1.0

-0.5

-1.0

85°C

-1.0

-1.5

-1.0

-1.5

-55

-45

-35

-25

-15

-5

5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 20. Mean Temperature Drift Error vs

RF Input Power at 900 MHz

Figure 19. Mean Temperature Drift Error vs

RF Input Power at 50 MHz

1.5

1.0

1.5

1.0

0.5

1.0

0.5

-40°C

85°C

0.0

-0.5

-1.0

-0.5

-1.0

85°C

-40°C

-15

-1.0

-1.5

-1.0

-1.5

-55

-45

-35

-25

-15

-5

5

-55

-45

-35

-25

-5

5

RF INPUT POWER (dBm)

Figure 21. Mean Temperature Drift Error vs

RF Input Power at 1855 MHz

Figure 22. Mean Temperature Drift Error vs

RF Input Power at 2500 MHz

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7 V, T_A= 25°C, measured on a limited number of samples.

1.5

1.0

1.5

1.0

0.5

85°C

0.0

-0.5

-1.0

-0.5

-1.0

-40°C

-1.0

-1.5

-1.0

-1.5

-55

-45

-35

-25

-15

-5

5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 23. Mean Temperature Drift Error vs

RF Input Power at 3000 MHz

Figure 24. Mean Temperature Drift Error vs

RF Input Power at 3500 MHz

1.5

1.0

0.5

85°C

0.0

-0.5

-1.0

-1.5

-40°C

-1.5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 25. Mean Temperature Drift Error vs

RF Input Power at 4000 MHz

Figure 26. Temperature Drift Error (Mean ±3 sigma) vs

RF Input Power at 50 MHz

Figure 28. Temperature Drift Error (Mean ±3 sigma) vs

RF Input Power at 1855 MHz

Figure 27. Temperature Drift Error (Mean ±3 sigma) vs

RF Input Power at 900 MHz

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7 V, T_A= 25°C, measured on a limited number of samples.

Figure 29. Temperature Drift Error (Mean ±3 sigma) vs

RF Input Power at 2500 MHz

Figure 30. Temperature Drift Error (Mean ±3 sigma) vs

RF Input Power at 3000 MHz

Figure 32. Temperature Drift Error (Mean ±3 sigma) vs

RF Input Power at 4000 MHz

Figure 31. Temperature Drift Error (Mean ±3 sigma) vs

RF Input Power at 3500 MHz

0.3

0.2

0.3

0.2

-40°C

0.2

-40°C

0.2

0.1

0.0

-0.1

-0.2

-0.1

-0.2

85°C

-0.2

-0.3

-0.2

-0.3

-55

-45

-35

-25

-15

-5

5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 34. Error for 1 dB Input Power Step vs

RF Input Power at 900 MHz

Figure 33. Error for 1 dB Input Power Step vs

RF Input Power at 50 MHz

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7 V, T_A= 25°C, measured on a limited number of samples.

0.3

0.2

0.3

0.2

-40°C

0.1

85°C

0.0

-0.1

-0.2

-0.1

-0.2

85°C

-35

-40°C

-0.2

-0.3

-0.2

-0.3

-55

-45

-35

-25

-15

-5

5

-55

-45

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 35. Error for 1 dB Input Power Step vs

Figure 36. Error for 1 dB Input Power Step vs

RF Input Power at 2500 MHz

RF Input Power at 1855 MHz

0.3

0.2

0.3

0.2

-40°C

0.2

-40°C

0.1

0.0

-0.1

-0.2

-0.1

-0.2

85°C

-35

85°C

-35

-0.2

-0.3

-0.2

-0.3

-55

-45

-25

-15

-5

5

-55

-45

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 37. Error for 1 dB Input Power Step vs

RF Input Power at 3000 MHz

Figure 38. Error for 1 dB Input Power Step vs

RF Input Power at 3500 MHz

0.3

1.00

0.75

1.00

0.3

0.2

-40°C

0.50

0.2

-40°C

0.75

0.1

0.50

0.25

0.0

0.00

-0.25

-0.50

-0.1

-0.2

-0.75

-0.50

-1.00

85°C

-0.2

-0.3

85°C

-0.75

-0.3

-1.00

-55

-45

-35

-25

-15

-5

5

-60

-50

-40

-30

-20

-10

0

RF INPUT POWER (dBm)

Figure 39. Error for 1 dB Input Power step vs

RF Input Power at 4000 MHz

Figure 40. Error for 10 dB Input Power Step vs

RF Input Power at 50 MHz

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7 V, T_A= 25°C, measured on a limited number of samples.

1.00

0.75

1.00

0.75

1.00

-40°C

0.50

0.75

-40°C

0.50

0.25

0.50

0.25

0.00

-0.25

-0.50

-0.25

-0.50

-0.75

85°C

-0.75

-0.50

-1.00

-0.50

-1.00

85°C

-0.75

-1.00

-0.75

-1.00

-60

-50

-40

-30

-20

-10

0

-60

-50

-40

-30

-20

-10

0

RF INPUT POWER (dBm)

Figure 42. Error for 10 dB Input Power Step vs

RF Input Power at 1855 MHz

Figure 41. Error for 10 dB Input Power Step vs

RF Input Power at 900 MHz

1.00

0.75

1.00

0.75

1.00

0.50

0.75

-40°C

0.50

0.25

0.50

0.25

0.00

-0.25

-0.50

-0.25

-0.50

-0.75

-0.50

-1.00

85°C

-1.00

85°C

-0.75

-1.00

-60

-50

-40

-30

-20

-10

0

-60

-50

-40

-30

-20

-10

0

RF INPUT POWER (dBm)

Figure 44. Error for 10 dB Input Power Step vs

RF Input Power at 3000 MHz

Figure 43. Error for 10 dB Input Power Step vs

RF Input Power at 2500 MHz

1.00

0.75

-40°C

1.00

0.50

0.75

0.50

0.25

0.50

0.25

0.00

-0.25

-0.50

-0.25

-0.50

-0.75

-0.50

-1.00

-0.50

-1.00

85°C

-0.75

-1.00

-60

-50

-40

-30

-20

-10

0

-60

-50

-40

-30

-20

-10

0

RF INPUT POWER (dBm)

Figure 46. Error for 10 dB Input Power step vs

RF Input Power at 4000 MHz

Figure 45. Error for 10 dB Input Power Step vs

RF Input Power at 3500 MHz

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7 V, T_A= 25°C, measured on a limited number of samples.

20

15

20

10

20

10

15

-40°C

10

5

10

5

0

-5

-10

-15

-10

-15

85°C

-45

85°C

-45

-10

-20

-15

-20

-15

-20

-55

-35

-25

-15

-5

5

-55

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 47. Mean Temperature Sensitivity vs

RF Input Power at 50 MHz

Figure 48. Mean Temperature Sensitivity vs

RF Input Power at 900 MHz

20

15

20

10

15

20

-40°C

10

15

-40°C

10

5

10

5

0

-5

-10

-5

-10

85°C

-35

-15

-10

-20

-15

-10

-20

-15

-20

-15

-20

-55

-45

-25

-15

-5

5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 49. Mean Temperature Sensitivity vs

RF Input Power at 1855 MHz

Figure 50. Mean Temperature Sensitivity vs

RF Input Power at 2500 MHz

20

15

-40°C

20

-40°C

10

15

10

5

10

5

0

-5

-10

-5

-10

85°C

-35

85°C

-15

-10

-20

-10

-20

-15

-20

-15

-20

-55

-45

-25

-15

-5

5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 51. Mean Temperature Sensitivity vs

RF Input Power at 3000 MHz

Figure 52. Mean Temperature Sensitivity vs

RF Input Power at 3500 MHz

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7 V, T_A= 25°C, measured on a limited number of samples.

20

15

10

15

-40°C

20

10

-40°C

15

10

5

0

5

0

-5

-10

-5

-10

-15

85°C

-15

-10

-20

85°C

-15

-20

-55

-45

-35

-25

-15

-5

5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 53. Mean Temperature Sensitivity vs

RF Input power at 4000 MHz

Figure 54. Temperature Sensitivity (Mean ±3 sigma) vs

RF Input Power at 50 MHz

20

15

20

15

-40°C

10

5

0

5

0

-5

-10

85°C

-15

-20

-55

-20

-45

-35

-25

-15

-5

5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 55. Temperature Sensitivity (Mean ±3 sigma) vs

RF Input Power at 900 MHz

Figure 56. Temperature Sensitivity (Mean ±3 sigma) vs

RF Input Power at 1855 MHz

20

-40°C

15

10

15

10

5

0

5

0

-5

-10

85°C

-15

-20

-55

-20

-55

-45

-35

-25

-15

-5

5

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 57. Temperature Sensitivity (Mean ±3 sigma) vs

RF Input Power at 2500 MHz

Figure 58. Temperature Sensitivity (Mean ±3 sigma) vs

RF Input Power at 3000 MHz

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7 V, T_A= 25°C, measured on a limited number of samples.

20

-40°C

15

10

5

15

10

5

0

-5

-10

-15

-10

-15

-20

85°C

-20

-55

-45

-35

-25

-15

-5

5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 59. Temperature Sensitivity (Mean ±3 sigma) vs

RF Input Power at 3500 MHz

Figure 60. Temperature Sensitivity (mean ±3 sigma) vs.

RF Input Power at 4000 MHz

2.0

1.8

1.6

2.5

CW

IS-95

WCDMA 64 CH

CW

IS-95

WCDMA 64 CH

1.8

2.0

1.6

2.0

1.5

2.5

1.5

2.5

2.0

1.8

1.4

2.0

1.0

1.4

1.0

2.0

1.8

1.6

1.5

1.6

11.2.4

1.2

1.5

CW

1.4

1.2

1.0

0.5

1.0

0.5

1.2

0.5

0.0

-0.5

CW

1.0

0.0

1.0

0.8

0.6

0.8

-0.5

0.6

0.8

-1.0

-0.5

-1.0

-0.5

0.8

0.4

0.2

0.4

-1.5

0.2

-2.0

-1.0

0.6

0.0

-1.0

0.0

0.4

-2.5

-1.5

-2.5

-1.5

WCDMA 64 ch

IS-95

0.4

IS-95 WCDMA 64 ch

0.2

0.0

0.2

0.0

-2.0

-2.5

5

-2.5

5

-55

-45

-35

-25

-15

-5

-55

-45

-35

-25

-15

-5

RF INPUT POWER (dBm)

Figure 61. Output Voltage and Log Conformance Error vs

RF Input Power for Various Modulation Types at 900 MHz

Figure 62. Output Voltage and Log Conformance Error vs

RF Input Power for Various Modulation Types at 1855 MHz

100

10

9

8

7

6

5

4

3

2

1

0

R

75

50

25

0

X

-25

-50

-75

-100

10M

100M

1G

10G

10

100

1k

10k

100k

1M

FREQUENCY (Hz)

Figure 63. RF Input Impedance vs

Figure 64. Output Noise Spectrum vs Frequency

Frequency (Resistance and Reactance)

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7 V, T_A= 25°C, measured on a limited number of samples.

100k

270

80

GAIN

225

270

70

180

60

225

10k

180

135

50

PHASE

90

40

30

20

10

0

45

1k

0

-45

0

-90

-45

100

-90

10

100

1k

10k

100k

1M

100

1k

10k

100k

1a

10a

FREQUENCY (Hz)

Figure 66. Output Amplifier Gain and Phase vs Frequency

Figure 65. Power Supply Rejection Ratio vs Frequency

60

85°C

25°C

85°C

25°C

60

50

60

50

-40°C

50

40

30

20

10

20

10

0

10

0

0.0

0.5

1.0

1.5

(V)

2.0

2.5

3.0

0.0

0.5

1.0

1.5

(V)

2.0

2.5

3.0

V

OUT

Figure 67. Sourcing Output Current vs Output Voltage

Figure 68. Sinking Output Current vs Output Voltage

2.70

0.08

2.70

2.68

-40°C

0.06

85°C

0.06

25°C

85°C

2.66

-40°C

0.04

2.64

0.02

2.62

2.60

0.00

2.60

0

1

2

3

4

5

0

1

2

3

4

5

SINKING CURRENT (mA)

SOURCING CURRENT (mA)

Figure 70. Output Voltage vs Sinking Current

Figure 69. Output Voltage vs Sourcing Current

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7 Detailed Description

7.1 Overview

The LMH2100 is a versatile logarithmic RF power detector suitable for use in power measurement systems. The

LMH2100 is particularly well suited for CDMA and UMTS applications. It produces a DC voltage that is a

measure for the applied RF power.

The core of the LMH2100 is a progressive compression LOG detector consisting of four gain stages. Each of

these saturating stages has a gain of approximately 10 dB and therefore realizes about 10 dB of the detector

dynamic range. The five diode cells perform the actual detection and convert the RF signal to a DC current. This

DC current is subsequently supplied to the transimpedance amplifier at the output, that converts it into an output

voltage. In addition, the amplifier provides buffering of and applies filtering to the detector output signal. To

prevent discharge of filtering capacitors between OUT and GND in shutdown, a switch is inserted at the amplifier

input that opens in shutdown to realize a high impedance output of the device.

7.2 Functional Block Diagram

REF

B2

VDD

A1

R_TRANS

en

EN C2

en

I / I

-

A2 OUT

+

V_REF

+

en

-

RFIN

B1

C1

10 dB

V-V

10 dB

GN

D

R_IN

7.3 Feature Description

7.3.1 Characteristics of the LMH2100

The LMH2100 is a logarithmic RF power detector with approximately 40-dB dynamic range. This dynamic range

plus its logarithmic behavior make the LMH2100 ideal for various applications such as wireless transmit power

control for CDMA and UMTS applications. The frequency range of the LMH2100 is from 50 MHz to 4 GHz, which

makes it suitable for various applications.

The LMH2100 transfer function is accurately temperature compensated. This makes the measurement accurate

for a wide temperature range. Furthermore, the LMH2100 can easily be connected to a directional coupler

because of its 50-Ω input termination. The output range is adjustable to fit the ADC input range. The detector can

be switched into a power saving shutdown mode for use in pulsed conditions.

7.3.2 Accurate Power Measurement

The power measurement accuracy achieved with a power detector is not only determined by the accuracy of the

detector itself, but also by the way it is integrated into the application. In many applications some form of

calibration is employed to improve the accuracy of the overall system beyond the intrinsic accuracy provided by

the power detector. For example, for LOG-detectors calibration can be used to eliminate part to part spread of

the LOG-slope and LOG-intercept from the overall power measurement system, thereby improving its power

measurement accuracy.

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Feature Description (continued)

This section shows how calibration techniques can be used to improve the accuracy of a power measurement

system beyond the intrinsic accuracy of the power detector itself. The main focus of the section is on power

measurement systems using LOG-detectors, specifically the LMH2100, but the more generic concepts can also

be applied to other power detectors. Other factors influencing the power measurement accuracy, such as the

resolution of the ADC reading the detector output signal will not be considered here since they are not

fundamentally due to the power detector.

7.3.2.1 LOG-Conformance Error

Probably the simplest power measurement system that can be realized is obtained when the LOG-detector

transfer function is modelled as a perfect linear-in-dB relationship between the input power and output voltage:

V_OUT,MOD

=

F_DET,MOD(P_IN) = K_SLOPE(P_INœ P_INTERCEPT

)

(1)

in which K_SLOPErepresents the LOG-slope and P_INTERCEPTthe LOG-intercept. The estimator based on this model

implements the inverse of the model equation, that is:

V_OUT

P_EST= F_EST(V_OUT) =

+ P_INTERCEPT

K_SLOPE

(2)

The resulting power measurement error, the LOG-conformance error, is thus equal to:

V_OUT

K_SLOPE

E_LCE= P_EST- P_IN

=

- (P_IN- P_INTERCEPT)

V_OUT- V_OUT,MOD

=

K_SLOPE

(3)

The most important contributions to the LOG-conformance error are generally:

•

The deviation of the actual detector transfer function from an ideal Logarithm (the transfer function is

nonlinear in dB).

Drift of the detector transfer function over various environmental conditions, most importantly temperature;

K_SLOPEand P_INTERCEPTare usually determined for room temperature only.

Part-to-part spread of the (room temperature) transfer function.

The latter component is conveniently removed by means of calibration, that is, if the LOG slope and LOG-

intercept are determined for each individual detector device (at room temperature). This can be achieved by

measurement of the detector output voltage - at room temperature - for a series of different power levels in the

LOG-linear range of the detector transfer function. The slope and intercept can then be determined by means of

linear regression.

An example of this type of error and its relationship to the detector transfer function is depicted in Figure 71.

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Feature Description (continued)

2.0

1.8

1.6

2.5

2.0

1.5

2.5

2.0

1.8

2.0

1.0

1.4

1.6

1.4

1.5

-40°C

85°C

25°C

1.0

0.5

1.2

1.0

0.5

0.0

1.0

0.8

-0.5

0.6

-1.0

-0.5

0.8

0.4

0.2

-1.5

-2.0

-1.0

0.6

0.0

-2.5

-1.5

85°C

-15

0.4

25°C

-40°C

-45

0.2

0.0

-2.0

-2.5

5

-55

-35

-25

-5

RF INPUT POWER (dBm)

Figure 71. LOG-Conformance Error and LOG-Detector Transfer Function

In the center of the detector's dynamic range, the LOG-conformance error is small, especially at room

temperature; in this region the transfer function closely follows the linear-in-dB relationship while K_SLOPEand

P_INTERCEPTare determined based on room temperature measurements. At the temperature extremes the error in

the center of the range is slightly larger due to the temperature drift of the detector transfer function. The error

rapidly increases toward the top and bottom end of the detector's dynamic range; here the detector saturates and

its transfer function starts to deviate significantly from the ideal LOG-linear model. The detector dynamic range is

usually defined as the power range for which the LOG conformance error is smaller than a specified amount.

Often an error of ±1 dB is used as a criterion.

7.3.2.2 Temperature Drift Error

A more accurate power measurement system can be obtained if the first error contribution, due to the deviation

from the ideal LOG-linear model, is eliminated. This is achieved if the actual measured detector transfer function

at room temperature is used as a model for the detector, instead of the ideal LOG-linear transfer function used in

the previous section.

The formula used for such a detector is:

V_OUT,MOD= F_DET(P_IN,T_O)

where

•

T_Orepresents the temperature during calibration (room temperature).

(4)

The transfer function of the corresponding estimator is thus the inverse of this:

-1

P_EST= F [V_OUT(T),T₀]

DET

(5)

In this expression V_OUT(T) represents the measured detector output voltage at the operating temperature T.

The resulting measurement error is only due to drift of the detector transfer function over temperature, and can

be expressed as:

-1

DET

E_DRIFT(T,T₀) =

P_EST- P_IN= F [V_OUT(T),T₀] - P_IN

= F^-1[V_OUT(T),T₀] - F_D^-_E¹_T[V_OUT(T),T)]

DET

(6)

Unfortunately, the (numeric) inverse of the detector transfer function at different temperatures makes this

expression rather impractical. However, since the drift error is usually small V_OUT(T) is only slightly different from

V_OUT(T_O). This means that we can apply the following approximation:

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Feature Description (continued)

E_DRIFT(T₀,T₀)

E_DRIFT(T,T₀) ö

ï

ïT

-1

DET

-1

DET

+ (T - T₀) {F [V_OUT(T),T₀] - F [V_OUT(T),T]}

(7)

This expression is easily simplified by taking the following considerations into account:

•

The drift error at the calibration temperature E(T_O,T_O) equals zero (by definition).

The estimator transfer F_DET(V_OUT,T_O) is not a function of temperature; the estimator output changes over

temperature only due to the temperature dependence of V_OUT

.

•

The actual detector input power P_INis not temperature dependent (in the context of this expression).

The derivative of the estimator transfer function to V_OUTequals approximately 1/K_SLOPEin the LOG-linear

region of the detector transfer function (the region of interest).

Using this, we arrive at:

-1

DET

ï

ïT

(T œ T )

F

[V_OUT(T),T₀]

E_DRIFT(T,T₀) ö

0

-1

DET

ï V

ïT

(T)

ï

OUT

= (T œ T₀)

F

[V_OUT(T),T₀]

ïV_OUT

V_OUT(T) œ V_OUT(T₀)

ö

K_SLOPE

(8)

This expression is very similar to the expression of the LOG-conformance error determined previously. The only

difference is that instead of the output of the ideal LOG-linear model, the actual detector output voltage at the

calibration temperature is now subtracted from the detector output voltage at the operating temperature.

Figure 72 depicts an example of the drift error.

1.5

1.0

0.5

-40°C

0.0

-0.5

-1.0

85°C

-1.0

-1.5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 72. Temperature Drift Error of the LMH2100 at ƒ = 1855 MHz

In agreement with the definition, the temperature drift error is zero at the calibration temperature. Further, the

main difference with the LOG-conformance error is observed at the top and bottom end of the detection range;

instead of a rapid increase the drift error settles to a small value at high and low input power levels due to the

fact that the detector saturation levels are relatively temperature independent.

In a practical application it may not be possible to use the exact inverse detector transfer function as the

algorithm for the estimator. For example it may require too much memory and/or too much factory calibration

time. However, using the ideal LOG-linear model in combination with a few extra data points at the top and

bottom end of the detection range - where the deviation is largest - can already significantly reduce the power

measurement error.

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Feature Description (continued)

7.3.2.2.1 Temperature Compensation

A further reduction of the power measurement error is possible if the operating temperature is measured in the

application. For this purpose, the detector model used by the estimator should be extended to cover the

temperature dependency of the detector.

Since the detector transfer function is generally a smooth function of temperature (the output voltage changes

gradually over temperature), the temperature is in most cases adequately modeled by a first-order or second-

order polynomial (see Equation 9).

V_OUT,MOD= F_DET(P_IN,T₀)[1 + (T-T₀)TC₁(P_IN)

+ (T-T₀)²TC₂(P_IN) + O(T³)]

(9)

The required temperature dependence of the estimator, to compensate for the detector temperature dependence

can be approximated similarly:

P_EST= F_D^-_E¹_T[V_OUT(T),T₀]{1 + (T-T₀)S₁[V_OUT(T)] +

+ (T-T₀)²S₂[V_OUT(T)] + O(T³)}

ö F_DE^-1_T[V_OUT(T),T₀]{1 + (T-T₀)S₁[V_OUT(T)]}

(10)

The last approximation results from the fact that a first-order temperature compensation is usually sufficiently

accurate. The remainder of this section will therefore concentrate on first-order compensation. For second and

higher-order compensation a similar approach can be followed.

Ideally, the temperature drift could be completely eliminated if the measurement system is calibrated at various

temperatures and input power levels to determine the Temperature Sensitivity S₁. In a practical application,

however that is usually not possible due to the associated high costs. The alternative is to use the average

temperature drift in the estimator, instead of the temperature sensitivity of each device individually. In this way it

becomes possible to eliminate the systematic (reproducible) component of the temperature drift without the need

for calibration at different temperatures during manufacturing. What remains is the random temperature drift,

which differs from device to device. Figure 73 illustrates the idea. The graph at the left schematically represents

the behavior of the drift error versus temperature at a certain input power level for a large number of devices.

ERROR

+3s

T

MEAN

-3s

MEAN

-3s

AFTER

TEMPERATURE

CORRECTION

Figure 73. Elimination of the Systematic Component from the Temperature Drift

The mean drift error represents the reproducible - systematic - part of the error, while the mean ± 3 sigma limits

represent the combined systematic plus random error component. Obviously the drift error must be zero at

calibration temperature T₀. If the systematic component of the drift error is included in the estimator, the total drift

error becomes equal to only the random component, as illustrated in the graph at the right of Figure 73. A

significant reduction of the temperature drift error can be achieved in this way only if:

•

The systematic component is significantly larger than the random error component (otherwise the difference

is negligible).

•

The operating temperature is measured with sufficient accuracy.

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Feature Description (continued)

It is essential for the effectiveness of the temperature compensation to assign the appropriate value to the

temperature sensitivity S₁. Two different approaches can be followed to determine this parameter:

•

Determination of a single value to be used over the entire operating temperature range.

Division of the operating temperature range in segments and use of separate values for each of the

segments.

Also for the first method, the accuracy of the extracted temperature sensitivity increases when the number of

measurement temperatures increases. Linear regression to temperature can then be used to determine the two

parameters of the linear model for the temperature drift error: the first order temperature sensitivity S₁and the

best-fit (room temperature) value for the power estimate at T₀: F_DET[V_OUT(T),T₀]. Note that to achieve an overall -

over all temperatures - minimum error, the room temperature drift error in the model can be non-zero at the

calibration temperature (which is not in agreement with the strict definition).

The second method does not have this drawback but is more complex. In fact, segmentation of the temperature

range is a form of higher-order temperature compensation using only a first-order model for the different

segments: one for temperatures below 25°C, and one for temperatures above 25°C. The mean (or typical)

temperature sensitivity is the value to be used for compensation of the systematic drift error component.

Figure 75 shows the temperature drift error without and with temperature compensation using two segments.

With compensation the systematic component is completely eliminated; the remaining random error component

is centered around zero. Note that the random component is slightly larger at −40°C than at 85°C.

Figure 74. Temperature Drift Error without Temperature

Compensation

Figure 75. Temperature Drift Error without with

Temperature Compensation

In a practical power measurement system, temperature compensation is usually only applied to a small power

range around the maximum power level for two reasons:

•

The various communication standards require the highest accuracy in this range to limit interference.

The temperature sensitivity itself is a function of the power level it becomes impractical to store a large

number of different temperature sensitivity values for different power levels.

The 2.7-V DC and AC Electrical Characteristics in the datasheet specifies the temperature sensitivity for the

aforementioned two segments at an input power level of −10 dBm (near the top-end of the detector dynamic

range). The typical value represents the mean which is to be used for calibration.

7.3.2.2.2 Differential Power Errors

Many third generation communication systems contain a power control loop through the base station and mobile

unit that requests both to frequently update the transmit power level by a small amount (typically 1 dB). For such

applications it is important that the actual change of the transmit power is sufficiently close to the requested

power change.

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Feature Description (continued)

The error metrics in the datasheet that describe the accuracy of the detector for a change in the input power are

E_{1 dB}(for a 1-dB change in the input power) and E_{10 dB}(for a 10-dB step, or ten consecutive steps of 1 dB). Since

it can be assumed that the temperature does not change during the power step the differential error equals the

difference of the drift error at the two involved power levels:

E_1dB(P ,T)=

IN

E_DRIFT(P_IN+1dB,T) - E_DRIFT(P_IN,T)

E_DRIFT(P_IN+10dB,T) - E_DRIFT(P_IN,T)

E_10dB(P ,T)=

IN

(11)

It should be noted that the step error increases significantly when one (or both) power levels in the above

expression are outside the detector dynamic range. For E_{10 dB}this occurs when P_INis less than 10 dB below the

maximum input power of the dynamic range, P_MAX

7.4 Device Functional Modes

7.4.1 Shutdown

.

To save power, the LMH2100 can be brought into a low-power shutdown mode. The device is active for EN =

HIGH (V_EN>1.1 V) and in the low-power shutdown mode for EN = LOW (V_EN< 0.6 V). In this state the output of

the LMH2100 is switched to a high impedance mode. Using the shutdown function, care must be taken not to

exceed the absolute maximum ratings. Forcing a voltage to the enable input that is 400 mV higher than V_DDor

400 mV lower than GND will damage the device and further operations is not ensured. The absolute maximum

ratings can also be exceeded when the enable EN is switched to HIGH (from shutdown to active mode) while the

supply voltage is low (off). This should be prevented at all times. A possible solution to protect the part is to add

a resistor of 100 kΩ in series with the enable input.

7.4.1.1 Output Behavior in Shutdown

In order to save power, the LMH2100 can be used in pulsed mode, such that it is active to perform the power

measurement only during a fraction of the time. During the remaining time the device is in low-power shutdown.

Applications using this approach usually require that the output value is available at all times, also when the

LMH2100 is in shutdown. The settling time in active mode, however, should not become excessively large. This

can be realized by the combination of the LMH2100 and a low pass output filter (see Figure 81).

In active mode, the filter capacitor C_Sis charged to the output voltage of the LMH2100, which in this mode has a

low output impedance to enable fast settling. During shutdown-mode, the capacitor should preserve this voltage.

Discharge of C_Sthrough any current path should therefore be avoided in shutdown. The output impedance of the

LMH2100 becomes high in shutdown, such that the discharge current cannot flow from the capacitor top plate,

through R_S, and the LMH2100 devices's OUT pin to GND. This is realized by the internal shutdown mechanism

of the output amplifier and by the switch depicted in Figure 85. Additionally, it should be ensured that the ADC

input impedance is high as well, to prevent a possible discharge path through the ADC.

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8 Application and Implementation

NOTE

Information in the following applications sections is not part of the TI component

specification, and TI does not warrant its accuracy or completeness. TI’s customers are

responsible for determining suitability of components for their purposes. Customers should

validate and test their design implementation to confirm system functionality.

8.1 Application Information

8.1.1 Functionality and Application of RF Power Detectors

This section describes the functional behavior of RF power detectors and their typical application. Based on a

number of key electrical characteristics of RF power detectors, Functionality of RF Power Detectors discusses

the functionality of RF power detectors in general and of the LMH2100 LOG detector in particular. Subsequently,

Typical Applications describes two important applications of the LMH2100 detector.

8.1.1.1 Functionality of RF Power Detectors

An RF power detector is a device that produces a DC output voltage in response to the RF power level of the

signal applied to its input. A wide variety of power detectors can be distinguished, each having certain properties

that suit a particular application. This section provides an overview of the key characteristics of power detectors,

and discusses the most important types of power detectors. The functional behavior of the LMH2100 is

discussed in detail.

8.1.1.1.1 Key Characteristics of RF Power Detectors

Power detectors are used to accurately measure the power of a signal inside the application. The attainable

accuracy of the measurement is therefore dependent upon the accuracy and predictability of the detector transfer

function from the RF input power to the DC output voltage.

Certain key characteristics determine the accuracy of RF detectors and they are classified accordingly:

•

Temperature Stability

Dynamic Range

Waveform Dependency

Transfer Shape

Generally, the transfer function of RF power detectors is slightly temperature dependent. This temperature drift

reduces the accuracy of the power measurement, because most applications are calibrated at room temperature.

In such systems, the temperature drift significantly contributes to the overall system power measurement error.

The temperature stability of the transfer function differs for the various types of power detectors. Generally,

power detectors that contain only one or few semiconductor devices (diodes, transistors) operating at RF

frequencies attain the best temperature stability.

The dynamic range of a power detector is the input power range for which it creates an accurately reproducible

output signal. What is considered accurate is determined by the applied criterion for the detector accuracy; the

detector dynamic range is thus always associated with certain power measurement accuracy. This accuracy is

usually expressed as the deviation of its transfer function from a certain predefined relationship, such as ”linear in

dB" for LOG detectors and ”square-law" transfer (from input RF voltage to DC output voltage) for Mean-Square

detectors. For LOG-detectors, the dynamic range is often specified as the power range for which its transfer

function follows the ideal linear-in-dB relationship with an error smaller than or equal to ±1 dB. Again, the

attainable dynamic range differs considerably for the various types of power detectors.

According to its definition, the average power is a metric for the average energy content of a signal and is not

directly a function of the shape of the signal in time. In other words, the power contained in a 0-dBm sine wave is

identical to the power contained in a 0-dBm square wave or a 0-dBm WCDMA signal; all these signals have the

same average power. Depending on the internal detection mechanism, though, power detectors may produce a

slightly different output signal in response to the aforementioned waveforms, even though their average power

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Application Information (continued)

level is the same. This is due to the fact that not all power detectors strictly implement the definition formula for

signal power, being the mean of the square of the signal. Most types of detectors perform some mixture of peak

detection and average power detection. A waveform independent detector response is often desired in

applications that exhibit a large variety of waveforms, such that separate calibration for each waveform becomes

impractical.

The shape of the detector transfer function from the RF input power to the DC output voltage determines the

required resolution of the ADC connected to it. The overall power measurement error is the combination of the

error introduced by the detector, and the quantization error contributed by the ADC. The impact of the

quantization error on the overall transfer's accuracy is highly dependent on the detector transfer shape, as shown

in Figure 76 and Figure 77.

2

ÂV

ÂV₁

ÂV₂

0

-60

0

-60

0

ÂP

RF INPUT POWER (dBm)

Figure 76. Convex Detector Transfer Function

Figure 77. Linear Transfer Function

Figure 76 and Figure 77 shows two different representations of the detector transfer function. In both graphs the

input power along the horizontal axis is displayed in dBm, since most applications specify power accuracy

requirements in dBm (or dB). The figure on the left shows a convex detector transfer function, while the transfer

function on the right hand side is linear (in dB). The slope of the detector transfer function — the detector

conversion gain – is of key importance for the impact of the quantization error on the total measurement error. If

the detector transfer function slope is low, a change, ΔP, in the input power results only in a small change of the

detector output voltage, such that the quantization error will be relatively large. On the other hand, if the detector

transfer function slope is high, the output voltage change for the same input power change will be large, such

that the quantization error is small. The transfer function on the left has a very low slope at low input power

levels, resulting in a relatively large quantization error. Therefore, to achieve accurate power measurement in this

region, a high-resolution ADC is required. On the other hand, for high input power levels the quantization error

will be very small due to the steep slope of the curve in this region. For accurate power measurement in this

region, a much lower ADC resolution is sufficient. The curve on the right has a constant slope over the power

range of interest, such that the required ADC resolution for a certain measurement accuracy is constant. For this

reason, the LOG-linear curve on the right will generally lead to the lowest ADC resolution requirements for

certain power measurement accuracy.

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Application Information (continued)

8.1.1.1.2 Types of RF Power Detectors

Three different detector types are distinguished based on the four characteristics previously discussed:

•

Diode Detector

(Root) Mean Square Detector

Logarithmic Detectors

8.1.1.1.2.1 Diode Detector

A diode is one of the simplest types of RF detectors. As depicted in Figure 78, the diode converts the RF input

voltage into a rectified current. This unidirectional current charges the capacitor. The RC time constant of the

resistor and the capacitor determines the amount of filtering applied to the rectified (detected) signal.

D

Z

0

VREF

R

C

S

V

S

OUT

Figure 78. Diode Detector

The advantages and disadvantages can be summarized as follows:

•

The temperature stability of the diode detectors is generally very good, since they contain only one

semiconductor device that operates at RF frequencies.

•

The dynamic range of diode detectors is poor. The conversion gain from the RF input power to the output

voltage quickly drops to very low levels when the input power decreases. Typically a dynamic range of 20 dB

to 25 dB can be realized with this type of detector.

•

The response of diode detectors is waveform dependent. As a consequence of this dependency for example

its output voltage for a 0-dBm WCDMA signal is different than for a 0-dBm unmodulated carrier. This is due to

the fact that the diode measures peak power instead of average power. The relation between peak power and

average power is dependent on the wave shape.

The transfer shape of diode detectors puts high requirements on the resolution of the ADC that reads their

output voltage. Especially at low input power levels a very high ADC resolution is required to achieve

sufficient power measurement accuracy (See Figure 76).

8.1.1.1.2.2 (Root) Mean Square Detector

This type of detector is particularly suited for the power measurements of RF modulated signals that exhibits

large peak to average power ratio variations. This is because its operation is based on direct determination of the

average power and not – like the diode detector – of the peak power.

The advantages and disadvantages can be summarized as follows:

•

The temperature stability of (R)MS detectors is almost as good as the temperature stability of the diode

detector; only a small part of the circuit operates at RF frequencies, while the rest of the circuit operates at

low frequencies.

•

The dynamic range of (R)MS detectors is limited. The lower end of the dynamic range is limited by internal

device offsets.

The response of (R)MS detectors is highly waveform independent. This is a key advantage compared to other

types of detectors in applications that employ signals with high peak-to-average power variations. For

example, the (R)MS detector response to a 0-dBm WCDMA signal and a 0-dBm unmodulated carrier is

essentially equal.

•

The transfer shape of R(MS) detectors has many similarities with the diode detector and is therefore subject

to similar disadvantages with respect to the ADC resolution requirements (see Figure 77).

8.1.1.1.2.3 Logarithmic Detectors

The transfer function of a logarithmic detector has a linear in dB response, which means that the output voltage

changes linearly with the RF power in dBm. This is convenient since most communication standards specify

transmit power levels in dBm as well.

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Application Information (continued)

The advantages and disadvantages can be summarized as follows:

•

The temperature stability of the LOG detector transfer function is generally not as good as the stability of

diode and R(MS) detectors. This is because a significant part of the circuit operates at RF frequencies.

•

The dynamic range of LOG detectors is usually much larger than that of other types of detectors.

Since LOG detectors perform a kind of peak detection their response is wave form dependent, similar to

diode detectors.

•

The transfer shape of LOG detectors puts the lowest possible requirements on the ADC resolution (See

Figure 77).

8.2 Typical Applications

RF power detectors can be used in a wide variety of applications. The first example shows the LMH2100 in a

Figure 79, the second application measures the Figure 88.

8.2.1 Application With Transmit Power Control Loop

The key benefit of a transmit power control loop circuit is that it makes the transmit power insensitive to changes

in the Power Amplifier (PA) gain control function, such as changes due to temperature drift. When a control loop

is used, the transfer function of the PA is eliminated from the overall transfer function. Instead, the overall

transfer function is determined by the power detector. The overall transfer function accuracy depends thus on the

RF detector accuracy. The LMH2100 is especially suited for this application, due to the accurate temperature

stability of its transfer function.

Figure 79 shows a block diagram of a typical transmit power control system. The output power of the PA is

measured by the LMH2100 through a directional coupler. The measured output voltage of the LMH2100 is

filtered and subsequently digitized by the ADC inside the baseband chip. The baseband adjusts the PA output

power level by changing the gain control signal of the RF VGA accordingly. With an input impedance of 50 Ω, the

LMH2100 can be directly connected to a 30 dB directional coupler without the need for an additional external

attenuator. The setup can be adjusted to various PA output ranges by selection of a directional coupler with the

appropriate coupling factor.

COUPLER

B

A

S

E

B

A

N

D

RF

VGA

PA

ANTENNA

50 W

GAIN

R_S

RFIN

ADC

OUT

C_S

LMH2100

EN

LOGIC

GND

Figure 79. Transmit Power Control System

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Typical Applications (continued)

8.2.1.1 Design Requirements

Some of the design requirements for this logarithmic RMS power detector include:

Table 1. Design Parameters

DESIGN PARAMETER

Supply voltage

EXAMPLE VALUE

2.7 V

1855 MHz

0 dBm

–5 dBm

2 V

RF input frequency (unmodulated continuous wave)

Minimum power level

Maximum power level

Maximum output voltage

8.2.1.2 Detailed Design Procedure

8.2.1.2.1 Detector Interfacing

For optimal performance of the LMH2100, it is important that all its pins are connected to the surrounding

circuitry in the appropriate way. This section discusses guidelines and requirements for the electrical connection

of each pin of the LMH2100 to ensure proper operation of the device. Starting from a block diagram, the function

of each pin is elaborated. Subsequently, the details of the electrical interfacing are separately discussed for each

pin. Special attention will be paid to the output filtering options and the differences between single ended and

differential interfacing with an ADC.

8.2.1.2.1.1 Concept of Power Measurements

Power measurement systems generally consists of two clearly distinguishable parts with different functions:

1. A power detector device, that generates a DC output signal (voltage) in response to the power level of the

(RF) signal applied to its input.

2. An “estimator” that converts the measured detector output signal into a (digital) numeric value representing

the power level of the signal at the detector input.

A sketch of this conceptual configuration is depicted in Figure 80.

F_EST

MODEL

P

IN

V

OUT

P

EST

F_DET

PARAMETERS

Figure 80. Generic Concept of a Power Measurement System

The core of the estimator is usually implemented as a software algorithm, receiving a digitized version of the

detector output voltage. Its transfer F_ESTfrom detector output voltage to a numerical output should be equal to

the inverse of the detector transfer F_DETfrom (RF) input power to DC output voltage. If the power measurement

system is ideal, that is, if no errors are introduced into the measurement result by the detector or the estimator,

the measured power P_EST- the output of the estimator - and the actual input power P_INshould be identical. In

that case, the measurement error E, the difference between the two, should be identically zero:

E =

P_EST- P_INô 0

∫ P_EST= F_EST[F_DET(P_IN)] = P_IN

-1

∫ F_EST(V_OUT) = F (V_OUT

)

DET

(12)

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From the expression above it follows that one would design the F_ESTtransfer function to be the inverse of the

F_DETtransfer function.

In practice the power measurement error will not be zero, due to the following effects:

•

The detector transfer function is subject to various kinds of random errors that result in uncertainty in the

detector output voltage; the detector transfer function is not exactly known.

•

The detector transfer function might be too complicated to be implemented in a practical estimator.

The function of the estimator is then to estimate the input power P_IN, that is, to produce an output P_ESTsuch that

the power measurement error is - on average - minimized, based on the following information:

1. Measurement of the not completely accurate detector output voltage V_OUT

2. Knowledge about the detector transfer function F_DET, for example the shape of the transfer function, the

types of errors present (part-to-part spread, temperature drift) etc.

Obviously the total measurement accuracy can be optimized by minimizing the uncertainty in the detector output

signal (select an accurate power detector), and by incorporating as much accurate information about the detector

transfer function into the estimator as possible.

The knowledge about the detector transfer function is condensed into a mathematical model for the detector

transfer function, consisting of:

•

A formula for the detector transfer function.

Values for the parameters in this formula.

The values for the parameters in the model can be obtained in various ways. They can be based on

measurements of the detector transfer function in a precisely controlled environment (parameter extraction). If

the parameter values are separately determined for each individual device, errors like part-to-part spread are

eliminated from the measurement system.

Errors may occur when the operating conditions of the detector (for example, the temperature) become

significantly different from the operating conditions during calibration (for example, room temperature). Examples

of simple estimators for power measurements that result in a number of commonly used metrics for the power

measurement error are discussed in LOG-Conformance Error, the Temperature Drift Error, the Temperature

Compensation and Temperature Drift Error.

8.2.1.2.1.2 RF Input

RF parts typically use a characteristic impedance of 50 Ω. To comply with this standard the LMH2100 has an

input impedance of 50 Ω. Using a characteristic impedance other then 50 Ω will cause a shift of the logarithmic

intercept with respect to the value given in the 2.7-V DC and AC Electrical Characteristics. This intercept shift

can be calculated according to Equation 13.

2 R_SOURCE

R_SOURCE+ 50

≈

«

’

◊

P_INT-SHIFT= 10 LOG

(13)

The intercept will shift to higher power levels for R_SOURCE> 50 Ω, and will shift to lower power levels for R_SOURCE

< 50 Ω.

8.2.1.2.1.3 Output and Reference

The possible filtering techniques that can be applied to reduce ripple in the detector output voltage are discussed

in Filtering. In addition two different topologies to connect the LMH2100 to an ADC are elaborated.

8.2.1.2.1.3.1 Filtering

The output voltage of the LMH2100 is a measure for the applied RF signal on the RF input pin. Usually, the

applied RF signal contains AM modulation that causes low frequency ripple in the detector output voltage. CDMA

signals for instance contain a large amount of amplitude variations. Filtering of the output signal can be used to

eliminate this ripple. The filtering can either be realized by a low pass output filter or a low pass feedback filter.

Those two techniques are depicted in Figure 81 and Figure 82.

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VDD

1

R_S

RFIN

OUT

RFIN

OUT

REF

1

6

+

-

2

6

+

-

C_S

ADC

R_P

C_P

LMH2100

EN

REF

4

5

4

5

3

GND

Figure 82. Low Pass Feedback Filter

Figure 81. Low Pass Output Filter

Depending on the system requirements one of the these filtering techniques can be selected. The low pass

output filter has the advantage that it preserves the output voltage when the LMH2100 is brought into shutdown.

This is elaborated in Output Behavior in Shutdown. In the feedback filter, resistor R_Pdischarges capacitor C_Pin

shutdown and therefore changes the output voltage of the device.

A disadvantage of the low pass output filter is that the series resistor R_Slimits the output drive capability. This

may cause inaccuracies in the voltage read by an ADC when the ADC input impedance is not significantly larger

than R_S. In that case, the current flowing through the ADC input induces an error voltage across filter resistor R_S.

The low pass feedback filter doesn’t have this disadvantage.

Note that adding an external resistor between OUT and REF reduces the transfer gain (LOG-slope and LOG-

intercept) of the device. The internal feedback resistor sets the gain of the transimpedance amplifier.

The filtering of the low pass output filter is realized by resistor R_Sand capacitor C_S. The −3 dB bandwidth of this

filter can then be calculated by: ƒ_{−3 dB}= 1 / 2πR_SC_S. The bandwidth of the low pass feedback filter is determined

by external resistor R_Pin parallel with the internal resistor R_TRANS, and external capacitor C_Pin parallel with

internal capacitor C_TRANS(see Figure 85). The −3 dB bandwidth of the feedback filter can be calculated by ƒ_{−3 dB}

= 1 / 2π (R_P//R_TRANS) (C_P+ C_TRANS). The bandwidth set by the internal resistor and capacitor (when no external

components are connected between OUT and REF) equals ƒ_{−3 dB}= 1 / 2π R_TRANSC_TRANS= 450 kHz.

8.2.1.2.1.4 Interface to the ADC

The LMH2100 can be connected to the ADC with a single-ended or a differential topology. The single ended

topology connects the output of the LMH2100 to the input of the ADC and the reference pin is not connected. In

a differential topology, both the output and the reference pins of the LMH2100 are connected to the ADC. The

topologies are depicted in Figure 83 and Figure 84.

VDD

1

VDD

1

RFIN

EN

RFIN

OUT

2

6

+

-

2

6

+

-

R_P

C_P

ADC

LMH2100

R_P

C_P

LMH2100

REF

EN

REF

4

5

3

4

5

3

GND

Figure 84. Differential Application

Figure 83. Single-Ended

The differential topology has the advantage that it is compensated for temperature drift of the internal reference

voltage. This can be explained by looking at the transimpedance amplifier of the LMH2100 (Figure 85).

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REF

C

R

TRANS

I_DET

-

OUT

+

V

REF

+

-

Figure 85. Output Stage of the LMH2100

It can be seen that the output of the amplifier is set by the detection current I_DETmultiplied by the resistor R_TRANS

plus the reference voltage V_REF

:

V_OUT= I_DETR_TRANS+ V_REF

(14)

I_DETrepresents the detector current that is proportional to the RF input power. The equation shows that

temperature variations in V_REFare also present in the output V_OUT. In case of a single ended topology the output

is the only pin that is connected to the ADC. The ADC voltage for single ended is thus:

V_ADC= I_DETR_TRANS+ V_REF

(15)

A differential topology also connects the reference pin, which is the value of reference voltage V_REF. The ADC

reads V_OUT– V_REF

:

V_ADC= V_OUT– V_REF= I_DETR_TRANS

(16)

Equation 16 does not contain the reference voltage V_REFanymore. Temperature variations in this reference

voltage are therefore not measured by the ADC.

8.2.1.3 Application Curves

2.0

1.6

1.2

0.8

0.4

0.0

RF = - 5 dBm

IN

1855 MHz

1.6

RF = -15 dBm

IN

900 MHz

RF = -25 dBm

IN

1.2

50 MHz

2500 MHz

0.8

RF = -35 dBm

IN

3000 MHz

0.4

RF = -45 dBm

IN

0.4

0.0

4000 MHz

0.0

10M

100M

1G

10G

-60 -50 -40 -30 -20 -10

RF INPUT POWER (dBm)

0

10

FREQUENCY (Hz)

Figure 86. Output Voltage vs RF Input Power

Figure 87. Output Voltage vs Frequency

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8.2.2 Application With Voltage Standing Wave Ratio Measurement

Transmission in RF systems requires matched termination by the proper characteristic impedance at the

transmitter and receiver side of the link. In wireless transmission systems though, matched termination of the

antenna can rarely be achieved. The part of the transmitted power that is reflected at the antenna bounces back

toward the PA and may cause standing waves in the transmission line between the PA and the antenna. These

standing waves can attain unacceptable levels that may damage the PA. A Voltage Standing Wave Ratio

(VSWR) measurement is used to detect such an occasion. It acts as an alarm function to prevent damage to the

transmitter.

VSWR is defined as the ratio of the maximum voltage divided by the minimum voltage at a certain point on the

transmission line:

1+ |G|

VSWR =

1 - |G|

(17)

Where Γ = V_REFLECTED/ V_FORWARDdenotes the reflection coefficient.

This means that to determine the VSWR, both the forward (transmitted) and the reflected power levels have to

be measured. This can be accomplished by using two LMH2100 RF power detectors according to Figure 88. A

directional coupler is used to separate the forward and reflected power waves on the transmission line between

the PA and the antenna. One secondary output of the coupler provides a signal proportional to the forward power

wave, the other secondary output provides a signal proportional to the reflected power wave. The outputs of both

RF detectors that measure these signals are connected to a micro-controller or baseband that calculates the

VSWR from the detector output signals.

COUPLER

ANTENNA

RF

PA

MICRO

CONTROLLER

V

DD

RF

IN

OUT

6

1

2

ADC1

REVERSE

POWER

LMH2100

R

C

P1

REF

EN

4

5

3

GND

RF

IN

OUT

REF

1

2

6

ADC2

TRANSMITTED

POWER

LMH2100

R

P2

C

P2

EN

4

5

3

GND

Figure 88. VSWR Application

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9 Power Supply Recommendations

The LMH2100 is designed to operate from an input voltage supply range between 2.7 V to 3.3 V. This input

voltage must be well regulated. Enable voltage levels lower than 400 below GND could lead to incorrect

operation of the device. Also, the resistance of the input supply rail must be low enough to ensure correct

operation of the device.

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10 Layout

10.1 Layout Guidelines

As with any other RF device, careful attention must be paid to the board layout. If the board layout is not properly

designed, unwanted signals can easily be detected or interference will be picked up. This section gives

guidelines for proper board layout for the LMH2100.

Electrical signals (voltages and currents) need a finite time to travel through a trace or transmission line. RF

voltage levels at the generator side and at the detector side can therefore be different. This is not only true for

the RF strip line, but for all traces on the PCB. Signals at different locations or traces on the PCB will be in a

different phase of the RF frequency cycle. Phase differences in, for example, the voltage across neighboring

lines, may result in crosstalk between lines due to parasitic capacitive or inductive coupling. This crosstalk is

further enhanced by the fact that all traces on the PCB are susceptible to resonance. The resonance frequency

depends on the trace geometry. Traces are particularly sensitive to interference when the length of the trace

corresponds to a quarter of the wavelength of the interfering signal or a multiple thereof.

10.1.1 Supply Lines

Because the PSRR of the LMH2100 is finite, variations of the supply can result in some variation at the output.

This can be caused among others by RF injection from other parts of the circuitry or the on/off switching of the

PA.

10.1.1.1 Positive Supply (V_DD

)

In order to minimize the injection of RF interference into the LMH2100 through the supply lines, the phase

difference between the PCB traces connecting to VDD and GND should be minimized. A suitable way to achieve

this is to short both connections for RF. This can be done by placing a small decoupling capacitor between the

VDD and GND. It should be placed as close as possible to the VDD and GND pins of the LMH2100 as indicated

in Figure 91. Be aware that the resonance frequency of the capacitor itself should be above the highest RF

frequency used in the application, because the capacitor acts as an inductor above its resonance frequency.

Low frequency supply voltage variations due to PA switching might result in a ripple at the output voltage. The

LMH2100 has a PSRR of 60 dB for low frequencies.

10.1.1.2 Ground (GND)

The LMH2100 needs a ground plane free of noise and other disturbing signals. It is important to separate the RF

ground return path from the other grounds. This is due to the fact that the RF input handles large voltage swings.

A power level of 0 dBm will cause a voltage swing larger than 0.6 V_PP, over the internal 50-Ω input resistor. This

will result in a significant RF return current toward the source. It is therefore recommended that the RF ground

return path not be used for other circuits in the design. The RF path should be routed directly back to the source

without loops.

10.1.2 RF Input Interface

The LMH2100 is designed to be used in RF applications, having a characteristic impedance of 50Ω. To achieve

this impedance, the input of the LMH2100 needs to be connected via a 50Ω transmission line. Transmission lines

can be easily created on PCBs using microstrip or (grounded) coplanar waveguide (GCPW) configurations. This

section will discuss both configurations in a general way. For more details about designing microstrip or GCPW

transmission lines, a microwave designer handbook is recommended.

10.1.3 Microstrip Configuration

One way to create a transmission line is to use a microstrip configuration. A cross section of the configuration is

shown in Figure 89, assuming a two-layer PCB.

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Layout Guidelines (continued)

METAL CONDUCTOR

W

FR4 PCB

H

GROUND PLANE

Figure 89. Microstrip Configuration

A conductor (trace) is placed on the topside of a PCB. The bottom side of the PCB has a fully copper ground

plane. The characteristic impedance of the microstrip transmission line is a function of the width W, height H, and

the dielectric constant ε_r.

Characteristics such as height and the dielectric constant of the board have significant impact on transmission

line dimensions. A 50-Ω transmission line may result in impractically wide traces. A typical 1.6-mm thick FR4

board results in a trace width of 2.9 mm, for instance. This is impractical for the LMH2100 because the pad width

of the 6-Bump DSBGA package is 0.24 mm. The transmission line has to be tapered from 2.9 mm to 0.24 mm.

Significant reflections and resonances in the frequency transfer function of the board may occur due to this

tapering.

10.1.4 GCPW Configuration

A transmission line in a (grounded) coplanar waveguide (GCPW) configuration will give more flexibility in terms of

trace width. The GCPW configuration is constructed with a conductor surrounded by ground at a certain

distance, S, on the top side. Figure 90 shows a cross section of this configuration. The bottom side of the PCB is

a ground plane. The ground planes on both sides of the PCB should be firmly connected to each other by

multiple vias. The characteristic impedance of the transmission line is mainly determined by the width W and the

distance S. In order to minimize reflections, the width W of the center trace should match the size of the package

pad. The required value for the characteristic impedance can subsequently be realized by selection of the proper

gap width S.

METAL CONDUCTOR

S

W

H

FR4 PCB

GROUND PLANE

Figure 90. GCPW Configuration

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Layout Guidelines (continued)

10.1.5 Reference (REF)

The Reference pin can be used to compensate for temperature drift of the internal reference voltage as

described in Interface to the ADC. The REF pin is directly connected to the inverting input of the transimpedance

amplifier. Thus, RF signals and other spurious signals couple directly through to the output. Introduction of RF

signals can be prevented by connecting a small capacitor between the REF pin and ground. The capacitor

should be placed close to the REF pin as depicted in Figure 91.

10.1.6 Output (OUT)

The OUT pin is sensitive to crosstalk from the RF input, especially at high power levels. The ESD diode between

OUT and VDD may rectify the crosstalk, but may add an unwanted inaccurate DC component to the output

voltage.

The board layout should minimize crosstalk between the detectors input RFIN and the detectors output. Using an

additional capacitor connected between the output and the positive supply voltage (VDD pin) or GND can

prevent this. For optimal performance this capacitor should be placed as close as possible to the OUT pin of the

LMH2100.

10.2 Layout Example

DECOUPLING

CAPACITOR

GND

VDD

OUT

REF

EN

CROSSTALK FILTER

CAPACITOR

RFIN

TRANSMISSION LINE

GND

Figure 91. Recommended LMH2100 Board Layout

42

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LMH2100

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SNWS020C –NOVEMBER 2007–REVISED OCTOBER 2015

11 Device and Documentation Support

11.1 Community Resources

The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective

contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of

Use.

TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration

among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help

solve problems with fellow engineers.

Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and

contact information for technical support.

11.2 Trademarks

E2E is a trademark of Texas Instruments.

All other trademarks are the property of their respective owners.

11.3 Electrostatic Discharge Caution

This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with

appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.

ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more

susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.

11.4 Glossary

SLYZ022 — TI Glossary.

This glossary lists and explains terms, acronyms, and definitions.

12 Mechanical, Packaging, and Orderable Information

The following pages include mechanical, packaging, and orderable information. This information is the most

current data available for the designated devices. This data is subject to change without notice and revision of

this document. For browser-based versions of this data sheet, refer to the left-hand navigation.

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43

Product Folder Links: LMH2100

MECHANICAL DATA

YFQ0006xxx

D

0.600±0.075

E

TMD06XXX (Rev B)

D: Max = 1.274 mm, Min =1.214 mm

E: Max = 0.874 mm, Min =0.814 mm

4215075/A

12/12

A. All linear dimensions are in millimeters. Dimensioning and tolerancing per ASME Y14.5M-1994.

B. This drawing is subject to change without notice.

NOTES:

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型号：	LMH2100TM/NOPB
厂家：	TEXAS INSTRUMENTS
描述：	用于 CDMA 和 WCDMA 的 50MHz 至 4GHz 40dB 对数功率检测器 \| YFQ \| 6 \| -40 to 85 CD 电信电信集成电路
文件：	总49页 (文件大小：1612K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

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