LMR23615QDRRRQ1 [TI]
SIMPLE SWITCHER® 36V、1.5A 同步降压转换器 | DRR | 12 | -40 to 125;型号: | LMR23615QDRRRQ1 |
厂家: | TEXAS INSTRUMENTS |
描述: | SIMPLE SWITCHER® 36V、1.5A 同步降压转换器 | DRR | 12 | -40 to 125 转换器 |
文件: | 总33页 (文件大小:2369K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
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LMR23615-Q1
ZHCSHR8 –MARCH 2018
LMR23615-Q1 SIMPLE SWITCHER® 36V、1.5A 同步降压转换器
1 特性
3 说明
•
符合汽车应用 标准
LMR23615-Q1 SIMPLE SWITCHER®是一款简便易用
的 36V、1.5A 同步降压稳压器。该器件具有 4V 至
36V 的宽输入范围,适用于各种 适用于 从工业到汽车
各类应用中非稳压电源的电压调节。此器件采用峰值电
流模式控制来实现简单控制环路补偿和逐周期电流限
制。它具有 75µA 的静态电流,因此适用于电池供电型
系统。2µA 的超低关断电流可进一步延长电池使用寿
命。内部环路补偿意味着用户不用承担设计环路补偿组
件的枯燥工作。这样还能够最大限度地减少外部元件
数。
1
•
具有符合 AEC-Q100 标准的下列特性:
–
器件温度 1 级:-40℃ 至 +125℃ 的环境运行温
度范围
–
–
器件 HBM ESD 分类等级 H2
器件组件充电模式 (CDM) ESD 分类等级 C4A
•
•
•
•
•
•
•
•
•
•
•
•
•
4V 至 36V 的输入范围
1.5A 持续输出电流
集成同步整流
最短打开时间:60ns
方便易用的内部补偿
该器件的扩展系列产品能够以引脚到引脚兼容的封装提
供 2.5A (LMR23625-Q1) 和 3A (LMR23630-Q1) 负载
电流选项,从而可以实现简单且最佳的 PCB 布局。精
密使能输入简化了稳压器控制和系统电源排序。保护功
能 特性 包括逐周期电流限制、间断模式短路保护和过
多功率耗散而引起的热关断。
可调开关频率
轻负载下的脉冲频率调制 (PFM) 模式
与外部时钟频率同步
软启动至预偏置负载
支持高占空比运行模式
具有间断模式的输出短路保护
12 引脚 WSON 可湿侧面封装,采用 PowerPAD™
器件信息(1)
结合使用 LMR23615-Q1 和 WEBENCH® 电源设计
器件型号
LMR23615-Q1
封装
封装尺寸(标称值)
WSON (12)
3.00mm x 3.00mm
器创建定制设计
(1) 如需了解所有可用封装,请参阅数据表末尾的可订购产品附
录。
2 应用
•
•
•
汽车信息娱乐系统:仪表组、音响主机、抬头显示
空白
空白
空白
USB 充电
一般电池供电应用
空白
简化原理图
效率与负载间的关系,VIN = 12V
VIN up to 36 V
CIN
100
VIN
90
80
70
60
BOOT
SW
EN/SYNC
AGND
CBOOT
L
VOUT
RFBT
COUT
RFBB
VCC
FB
50
CVCC
VOUT = 5 V
VOUT = 3.3 V
PGND
40
1E-5
0.0001
0.001
0.01
0.1
1
10
IOUT (A)
LMR2
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
English Data Sheet: SNVSAS5
LMR23615-Q1
ZHCSHR8 –MARCH 2018
www.ti.com.cn
目录
7.3 Feature Description................................................. 10
7.4 Device Functional Modes........................................ 16
Application and Implementation ........................ 17
8.1 Application Information............................................ 17
8.2 Typical Applications ................................................ 17
Power Supply Recommendations...................... 23
1
2
3
4
5
6
特性.......................................................................... 1
应用.......................................................................... 1
说明.......................................................................... 1
修订历史记录 ........................................................... 2
Pin Configuration and Functions......................... 3
Specifications......................................................... 4
6.1 Absolute Maximum Ratings ...................................... 4
6.2 ESD Ratings.............................................................. 4
6.3 Recommended Operating Conditions ...................... 4
6.4 Thermal Information.................................................. 5
6.5 Electrical Characteristics........................................... 5
6.6 Timing Characteristics............................................... 6
6.7 Switching Characteristics.......................................... 6
6.8 Typical Characteristics.............................................. 7
Detailed Description .............................................. 9
7.1 Overview ................................................................... 9
7.2 Functional Block Diagram ......................................... 9
8
9
10 Layout................................................................... 23
10.1 Layout Guidelines ................................................. 23
10.2 Layout Example .................................................... 25
11 器件和文档支持 ..................................................... 26
11.1 器件支持................................................................ 26
11.2 接收文档更新通知 ................................................. 26
11.3 社区资源................................................................ 26
11.4 商标....................................................................... 26
11.5 静电放电警告......................................................... 26
11.6 Glossary................................................................ 26
12 机械、封装和可订购信息....................................... 26
7
4 修订历史记录
日期
修订版本
说明
2018 年 3 月
*
初始发行版
2
Copyright © 2018, Texas Instruments Incorporated
LMR23615-Q1
www.ti.com.cn
ZHCSHR8 –MARCH 2018
5 Pin Configuration and Functions
DRR Package
12-Pin WSON With RT
Top View
PGND
NC
1
2
3
4
5
6
12
11
10
9
SW
SW
VIN
BOOT
VCC
FB
PAD
13
VIN
EN/
SYNC
8
7
RT
AGND
Pin Functions
PIN
(1)
I/O
DESCRIPTION
NO.
NAME
Switching output of the regulator. Internally connected to both power MOSFETs. Connect to power
inductor.
1, 2
SW
P
Boot-strap capacitor connection for high-side driver. Connect a high-quality, 100-nF capacitor from
BOOT to SW.
3
4
BOOT
VCC
P
P
Internal bias supply output for bypassing. Connect a 2.2-μF, 16-V or higher capacitance bypass
capacitor from this pin to AGND. Do not connect external loading to this pin. Never short this pin to
ground during operation.
5
6
7
FB
RT
A
A
G
Feedback input to regulator, connect the feedback resistor divider tap to this pin.
Connect a resistor RT from this pin to AGND to program switching frequency. Leave floating for
400-kHz default switching frequency.
AGND
Analog ground pin. Ground reference for internal references and logic. Connect to system ground.
Enable input to regulator. High = On, Low = Off. Can be connected to VIN. Do not float. Adjust the
input undervoltage lockout with two resistors. The internal oscillator can be synchronized to an
external clock by coupling a positive pulse into this pin through a small coupling capacitor. See
EN/SYNC for detail.
8
EN/SYNC
A
9, 10
11
VIN
NC
P
Input supply voltage.
N/A
G
Not for use. Leave this pin floating.
Power ground pin, connected internally to the low side power FET. Connect to system ground,
PAD, AGND, ground pins of CIN and COUT. Path to CIN must be as short as possible.
12
13
PGND
PAD
Low impedance connection to AGND. Connect to PGND on PCB. Major heat dissipation path of
the die. Must be used for heat sinking to ground plane on PCB.
G
(1) A = Analog, P = Power, G = Ground.
Copyright © 2018, Texas Instruments Incorporated
3
LMR23615-Q1
ZHCSHR8 –MARCH 2018
www.ti.com.cn
6 Specifications
6.1 Absolute Maximum Ratings
Over the recommended operating junction temperature range of –40°C to +125°C (unless otherwise noted)(1)
PARAMETER
VIN to PGND
MIN
–0.3
–5.5
–0.3
–0.3
–0.3
–1
MAX
42
UNIT
EN/SYNC to AGND
FB to AGND
42
Input voltages
4.5
V
RT to AGND
4.5
AGND to PGND
SW to PGND
0.3
VIN + 0.3
42
SW to PGND less than 10-ns transients
BOOT to SW
–5
Output voltages
V
–0.3
–0.3
–40
–65
5.5
VCC to AGND
4.5(2)
150
150
Junction temperature, TJ
Storage temperature, Tstg
°C
°C
(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
(2) In shutdown mode, the VCC to AGND maximum value is 5.25 V.
6.2 ESD Ratings
VALUE
±2500
±1000
UNIT
(1)
Human-body model (HBM)
Charged-device model (CDM)
V(ESD)
Electrostatic discharge
V
(1) AEC Q100-002 indicates that HBM stressing shall be in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.3 Recommended Operating Conditions
Over the recommended operating junction temperature range of –40°C to 125°C (unless otherwise noted)
(1)
MIN
4
MAX
36
UNIT
VIN
Input voltage
EN/SYNC
FB
–5
–0.3
1
36
V
1.2
28
Output voltage, VOUT
Output current, IOUT
V
A
0
1.5
125
Operating junction temperature, TJ
–40
°C
(1) Recommended Operating Ratings indicate conditions for which the device is intended to be functional, but do not ensured specific
performance limits. For specified specifications, see Electrical Characteristics.
4
Copyright © 2018, Texas Instruments Incorporated
LMR23615-Q1
www.ti.com.cn
ZHCSHR8 –MARCH 2018
6.4 Thermal Information
LMR23615-Q1
THERMAL METRIC(1)(2)
DDR (WSON)
12 PINS
41.5
UNIT
RθJA
Junction-to-ambient thermal resistance
°C/W
°C/W
°C/W
°C/W
°C/W
°C/W
RθJC(top)
RθJB
Junction-to-case (top) thermal resistance
Junction-to-board thermal resistance
0.3
16.5
ψJT
Junction-to-top characterization parameter
Junction-to-board characterization parameter
Junction-to-case (bottom) thermal resistance
39.1
ψJB
3.4
RθJC(bot)
16.3
(1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
(2) Determine power rating at a specific ambient temperature (TA) with a maximum junction temperature (TJ) of 125°C, which is illustrated in
Recommended Operating Conditions section.
6.5 Electrical Characteristics
Limits apply over the recommended operating junction temperature (TJ) range of –40°C to +125°C, unless otherwise stated.
Minimum and Maximum limits are specified through test, design or statistical correlation. Typical values represent the most
likely parametric norm at TJ = 25°C, and are provided for reference purposes only.
PARAMETER
POWER SUPPLY (VIN PIN)
VIN Operation input voltage
TEST CONDITIONS
MIN
TYP
MAX UNIT
4
3.3
2.9
36
3.9
3.5
4
V
V
Rising threshold
3.7
3.3
2
VIN_UVLO Undervoltage lockout thresholds
Falling threshold
ISHDN
IQ
Shutdown supply current
VEN = 0 V, VIN = 12 V, TJ = –40°C to 125°C
μA
μA
Operating quiescent current (non-
switching)
VIN =12 V, VFB = 1.2 V, TJ = –40°C to 125°C
75
ENABLE (EN/SYNC PIN)
VEN_H
Enable rising threshold voltage
1.4
0.4
1.55
0.4
1.7
V
V
VEN_HYS
VWAKE
Enable hysteresis voltage
Wake-up threshold
V
VIN = 4 V to 36 V, VEN= 2 V
VIN = 4 V to 36 V, VEN= 36 V
10
100
1
nA
μA
IEN
Input leakage current at EN pin
VOLTAGE REFERENCE (FB PIN)
VIN = 4 V to 36 V, TJ = 25°C
VIN = 4 V to 36 V, TJ = –40°C to 125°C
VFB= 1 V
0.985
0.98
1
1
1.015
1.02
VREF
Reference voltage
V
ILKG_FB
Input leakage current at FB pin
10
nA
INTERNAL LDO (VCC PIN)
VCC
Internal LDO output voltage
4.1
3.2
2.8
V
V
Rising threshold
Falling threshold
2.8
2.4
3.6
3.2
VCC undervoltage lockout
thresholds
VCC_UVLO
CURRENT LIMIT
IHS_LIMIT
ILS_LIMIT
IL_ZC
Peak inductor current limit
2.9
1.9
3.9
2.5
4.9
3.2
A
A
A
Valley inductor current limit
Zero cross current limit
–0.04
INTEGRATED MOSFETS
RDS_ON_HS High-side MOSFET ON-resistance
RDS_ON_LS Low-side MOSFET ON-resistance
THERMAL SHUTDOWN
TSHDN Thermal shutdown threshold
THYS Hysteresis
VIN = 12 V, IOUT = 1 A
VIN = 12 V, IOUT = 1 A
160
95
mΩ
mΩ
162
170
15
178
°C
°C
Copyright © 2018, Texas Instruments Incorporated
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LMR23615-Q1
ZHCSHR8 –MARCH 2018
www.ti.com.cn
6.6 Timing Characteristics
Over the recommended operating junction temperature range of –40°C to +125°C (unless otherwise noted)
MIN
NOM
MAX
UNIT
HICCUP MODE
Number of cycles that LS current limit is
tripped to enter hiccup mode
(1)
NOC
64
10
Cycles
ms
TOC
Hiccup retry delay time
SOFT START
Internal soft-start time. The time of internal
reference to increase from 0 V to 1 V
ms
TSS
6
(1) Specified by design.
6.7 Switching Characteristics
Over the recommended operating junction temperature range of –40°C to +125°C (unless otherwise noted)
PARAMETER
TEST CONDITION
MIN
TYP
MAX UNIT
SW (SW PIN)
TON_MIN
Minimum turnon time
Minimum turnoff time
60
90
ns
ns
(1)
TOFF_MIN
100
SYNC (EN/SYNC PIN)
fSW
Oscillator default frequency
RT pin open circuit
340
150
400
200
460 kHz
250 kHz
2425
Minimum adjustable frequency
Maximum adjustable frequency
SYNC frequency range
RT = 198 kΩ with 1% accuracy
RT = 17.8 kΩ with 1% accuracy
fADJ
1750
200
2150
fSYNC
2200 kHz
Amplitude of SYNC clock AC signal
(measured at SYNC pin)
VSYNC
2.8
5.5
V
TSYNC_MIN
Minimum sync clock ON and OFF time
100
ns
(1) Specified by design.
6
Copyright © 2018, Texas Instruments Incorporated
LMR23615-Q1
www.ti.com.cn
ZHCSHR8 –MARCH 2018
6.8 Typical Characteristics
Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 1600 kHz, L = 4.7µH, COUT = 47 µF, TA = 25°C.
100
90
80
70
60
50
40
30
20
10
0
100
90
80
70
60
50
40
30
20
10
0
VIN = 12 V
VIN = 24 V
VIN = 36 V
VIN = 8 V
VIN = 12 V
VIN = 24 V
1E-5
fSW = 1000 kHz
Figure 1. Efficiency vs Load Current
0.0001
0.001
0.01
IOUT (A)
0.1
1
10
1E-5
fSW = 1000 kHz
Figure 2. Efficiency vs Load Current
0.0001
0.001
0.01
IOUT (A)
0.1
1
10
LMR2
LMR2
VOUT = 5 V
VOUT = 3.3 V
100
90
80
70
60
50
40
30
20
10
0
100
90
80
70
60
50
40
30
20
10
0
VIN = 8 V
VIN = 12 V
VIN = 24 V
VIN = 8 V
VIN = 12 V
VIN = 20 V
1E-5
fSW = 2200 kHz
Figure 3. Efficiency vs Load Current
VIN = 12 V
0.0001
0.001
0.01
IOUT (A)
0.1
1
10
1E-5
fSW = 2200 kHz
Figure 4. Efficiency vs Load Current
0.0001
0.001
0.01
IOUT (A)
0.1
1
10
LMR2
LMR2
VOUT = 5 V
VOUT = 3.3 V
5.12
5.11
5.1
5.1
5.09
5.08
5.07
5.06
5.05
5.04
5.03
5.02
5.01
VIN = 24 V
VIN = 36 V
IOUT = 1.5 A
IOUT = 0.2 A
IOUT = 0 A
5.09
5.08
5.07
5.06
5.05
5.04
5.03
5.02
5.01
0
0.2
0.4
0.6
0.8
1
1.2
1.4
1.6
5
10
15
20
25
30
35
40
IOUT (A)
VIN (V)
LMR2
LMR2
fSW = 1000 kHz
VOUT = 5 V
Figure 5. Load Regulation
fSW = 1000 kHz
VOUT = 5 V
Figure 6. Line Regulation
Copyright © 2018, Texas Instruments Incorporated
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LMR23615-Q1
ZHCSHR8 –MARCH 2018
www.ti.com.cn
Typical Characteristics (continued)
Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 1600 kHz, L = 4.7µH, COUT = 47 µF, TA = 25°C.
5.5
5.3
5.1
4.9
4.7
4.5
4.3
4.1
3.9
3.7
3.5
5.5
5.3
5.1
4.9
4.7
4.5
4.3
4.1
3.9
3.7
3.5
IOUT = 0 A
IOUT = 0 A
IOUT = 0.2 A
IOUT = 0.8 A
IOUT = 1.5 A
IOUT = 0.2 A
IOUT = 0.8 A
IOUT = 1.5 A
4
4.5
5
5.5
6
4
4.5
5
5.5
6
VIN (V)
VIN (V)
LMR2
LMR2
fSW = 1000 kHz
VOUT = 5 V
fSW = 2200 kHz
VOUT = 5 V
Figure 7. Dropout Curve
Figure 8. Dropout Curve
80
3.67
3.66
3.65
3.64
3.63
3.62
75
70
65
60
-50
3.61
-50
0
50
100
150
0
50
100
150
Temperature (°C)
Temperature (°C)
D008
D009
VIN = 12 V
VFB = 1.1 V
Figure 9. IQ vs Junction Temperature
Figure 10. VIN UVLO Rising Threshold vs Junction
Temperature
4.5
4
0.425
0.42
LS Limit
HS Limit
3.5
3
0.415
2.5
0.41
2
-50
-50
0
50
100
150
0
50
100
150
Temperature (°C)
Temperature (èC)
D010
LMR2
VIN = 12 V
Figure 11. VIN UVLO Hysteresis vs Junction Temperature
Figure 12. HS and LS Current Limit vs Junction
Temperature
8
Copyright © 2018, Texas Instruments Incorporated
LMR23615-Q1
www.ti.com.cn
ZHCSHR8 –MARCH 2018
7 Detailed Description
7.1 Overview
The LMR23615-Q1 SIMPLE SWITCHER regulator is an easy-to-use synchronous step-down DC/DC converter
operating from 4-V to 36-V supply voltage. The device is capable of delivering up to 1.5-A DC load current with
good thermal performance in a small solution size. An extended family is available in multiple current options
from 1.5 A to 3 A in pin-to-pin compatible packages.
The LMR23615-Q1 employs fixed-frequency peak-current-mode control. The device enters PFM mode at light
load to achieve high efficiency. The device is internally compensated, which reduces design time and requires
few external components. The switching frequency is adjustable from 200 kHz to 2.2 MHz, leave RT pin open for
400-kHz default switching frequency. The LMR23615-Q1 is capable of synchronization to an external clock within
the range of 200 kHz to 2.2 MHz.
Additional features such as precision enable and internal soft start provide a flexible and easy-to-use solution for
a wide range of applications. Protection features include thermal shutdown, VIN and VCC undervoltage lockout
(UVLO), cycle-by-cycle current limit, and hiccup-mode short-circuit protection.
The family requires very few external components and has a pinout designed for simple, optimum PCB layout.
7.2 Functional Block Diagram
EN/SYNC
VCC
SYNC Signal
SYNC
VCC
LDO
VIN
Detector
Enable
Precision
Enable
Internal
SS
CBOOT
HS I Sense
EA
REF
Rc
Cc
TSD
UVLO
PWM CONTROL LOGIC
PFM
Detector
SW
OV/UV
Detector
FB
Slope
Comp
Freq
Foldback
Zero
Cross
HICCUP
Detector
SYNC Signal
LS I
Sense
RT
Oscillator
FB
AGND
PGND
Copyright © 2017, Texas Instruments Incorporated
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LMR23615-Q1
ZHCSHR8 –MARCH 2018
www.ti.com.cn
7.3 Feature Description
7.3.1 Fixed-Frequency Peak-Current-Mode Control
The following operating description of the LMR23615-Q1 refers to the Functional Block Diagram and to the
waveforms in Figure 13. The LMR23615-Q1 is a step-down synchronous buck regulator with integrated high-side
(HS) and low-side (LS) switches (synchronous rectifier). The LMR23615-Q1 supplies a regulated output voltage
by turning on the HS and LS NMOS switches with controlled duty cycle. During high-side switch ON-time, the
SW pin voltage swings up to approximately VIN, and the inductor current iL increase with linear slope (VIN – VOUT
)
/ L. When the HS switch is turned off by the control logic, the LS switch is turned on after an anti-shoot-through
dead time. Inductor current discharges through the LS switch with a slope of –VOUT / L. The control parameter of
a buck converter is defined as duty cycle D = tON / TSW, where tON is the high-side switch ON-time and TSW is the
switching period. The regulator control loop maintains a constant output voltage by adjusting the duty cycle D. In
an ideal buck converter, where losses are ignored, D is proportional to the output voltage and inversely
proportional to the input voltage: D = VOUT / VIN.
VSW
D = tON/ TSW
VIN
tON
tOFF
t
0
-VD
TSW
iL
ILPK
IOUT
DiL
t
0
Figure 13. SW Node and Inductor Current Waveforms in
Continuous Conduction Mode (CCM)
The LMR23615-Q1 employs fixed-frequency peak-current-mode control. A voltage feedback loop is used for
accurate DC voltage regulation by adjusting the peak current command based on voltage offset. The peak
inductor current is sensed from the high-side switch and compared to the peak current threshold to control the
ON-time of the high-side switch. The voltage feedback loop is internally compensated, which allows for fewer
external components, makes it easy to design, and provides stable operation with almost any combination of
output capacitors. The regulator operates with fixed switching frequency at normal load condition. At light load
condition, the LMR23615-Q1 operates in PFM mode to maintain high efficiency.
7.3.2 Adjustable Frequency
The switching frequency can be programmed by the impedance RT from the RT pin to ground. The frequency is
inversely proportional to the RT resistance. The RT pin can be left floating and the LMR23615 will operate at 400
kHz default switching frequency. The RT pin is not designed to be shorted to ground. For a desired frequency,
typical RT resistance can be found by Equation 1. Table 1 gives typical RT values for a given fSW
.
RT(kΩ) = 40200 / fSW(kHz) – 0.6
(1)
10
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ZHCSHR8 –MARCH 2018
Feature Description (continued)
250
200
150
100
50
0
0
500
1000
1500
2000
2500
Switching Frequency (kHz)
C008
Figure 14. RT vs Frequency Curve
Table 1. Typical Frequency Setting RT Resistance
fSW (kHz)
RT (kΩ)
200
200
350
115
500
78.7
53.6
39.2
26.1
19.6
17.8
750
1000
1500
2000
2200
7.3.3 Adjustable Output Voltage
A precision 1-V reference voltage is used to maintain a tightly regulated output voltage over the entire operating
temperature range. The output voltage is set by a resistor divider from output voltage to the FB pin. TI
recommends using 1% tolerance resistors with a low temperature coefficient for the FB divider. Select the low-
side resistor RFBB for the desired divider current and use Equation 2 to calculate high-side RFBT. RFBT in the
range from 10 kΩ to 100 kΩ is recommended for most applications. A lower RFBT value can be used if static
loading is desired to reduce VOUT offset in PFM operation. Lower RFBT reduces efficiency at very light load. Less
static current goes through a larger RFBT and might be more desirable when light load efficiency is critical. But
RFBT larger than 1 MΩ is not recommended because it makes the feedback path more susceptible to noise.
Larger RFBT value requires more carefully designed feedback path on the PCB. The tolerance and temperature
variation of the resistor dividers affect the output voltage regulation.
V
OUT
R
FBT
FBB
FB
R
Figure 15. Output Voltage Setting
VOUT - VREF
RFBT
=
ìRFBB
VREF
(2)
11
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7.3.4 EN/SYNC
The voltage on the EN/SYNC pin controls the ON or OFF operation of LMR23615-Q1. A voltage less than 1 V
(typical) shuts down the device while a voltage higher than 1.6 V (typical) is required to start the regulator. The
EN pin is an input and cannot be left open or floating. The simplest way to enable the operation of the
LMR23615-Q1 is to connect the EN to VIN. This allows self-start-up of the LMR23615-Q1 when VIN is within the
operation range.
Many applications benefit from the employment of an enable divider RENT and RENB (Figure 16) to establish a
precision system UVLO level for the converter. System UVLO can be used for supplies operating from utility
power as well as battery power. It can be used for sequencing, ensuring reliable operation, or supply protection,
such as a battery discharge level. An external logic signal can also be used to drive EN input for system
sequencing and protection.
VIN
RENT
EN/SYNC
RENB
Figure 16. System UVLO by Enable Divider
The EN/SYNC pin also can be used to synchronize the internal oscillator to an external clock. The internal
oscillator can be synchronized by AC coupling a positive edge into the EN/SYNC pin. The AC coupled peak-to-
peak voltage at the EN/SYNC pin must exceed the SYNC amplitude threshold of 2.8 V (typical) to trip the internal
synchronization pulse detector, and the minimum SYNC clock ON- and OFF-times must be longer than 100 ns
(typical). A 3.3-V or a higher amplitude pulse signal coupled through a 1-nF capacitor CSYNC is a good starting
point. Keeping RENT // RENB (RENT parallel with RENB) in the 100-kΩ range is a good choice. RENT is required for
this synchronization circuit, but RENB can be left unmounted if system UVLO is not needed. LMR23615-Q1
switching action can be synchronized to an external clock from 200 kHz to 2.2 MHz. Figure 18 and Figure 19
show the device synchronized to an external system clock.
VIN
RENT
CSYNC
EN/SYNC
RENB
Clock
Source
Figure 17. Synchronize to External Clock
12
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Figure 18. Synchronizing in PWM Mode
Figure 19. Synchronizing in PFM Mode
7.3.5 VCC, UVLO
The LMR23615-Q1 integrates an internal LDO to generate VCC for control circuitry and MOSFET drivers. The
nominal voltage for VCC is 4.1 V. The VCC pin is the output of an LDO and must be properly bypassed. Place
high-quality ceramic capacitor with a value of 2.2 µF to 10 µF, 16 V or higher rated voltage as close as possible
to VCC and grounded to the exposed PAD and ground pins. The VCC output pin must not be loaded or shorted
to ground during operation. Shorting VCC to ground during operation may cause damage to the LMR23615-Q1.
VCC UVLO prevents the LMR23615-Q1 from operating until the VCC voltage exceeds 3.3 V (typical). The VCC
UVLO threshold has 400 mV (typical) of hysteresis to prevent undesired shutdown due to temporary VIN drops.
7.3.6 Minimum ON-time, Minimum OFF-time and Frequency Foldback at Dropout Conditions
Minimum ON-time, TON_MIN, is the smallest duration of time that the HS switch can be on. TON_MIN is typically 60
ns in the LMR23615-Q1. Minimum OFF-time, TOFF_MIN, is the smallest duration that the HS switch can be off.
TOFF_MIN is typically 100 ns in the LMR23615-Q1. In CCM operation, TON_MIN and TOFF_MIN limit the voltage
conversion range given a selected switching frequency.
The minimum duty cycle allowed is:
DMIN = TON_MIN × fSW
(3)
And the maximum duty cycle allowed is:
DMAX = 1 – TOFF_MIN × fSW
(4)
Given fixed TON_MIN and TOFF_MIN, the higher the switching frequency the narrower the range of the allowed duty
cycle. In the LMR23615-Q1, a frequency foldback scheme is employed to extend the maximum duty cycle when
TOFF_MIN is reached. The switching frequency decreases once longer duty cycle is needed under low VIN
conditions. Wide range of frequency foldback allows the LMR23615-Q1 output voltage stay in regulation with a
much lower supply voltage VIN. This leads to a lower effective drop-out voltage.
Given an output voltage, the choice of the switching frequency affects the allowed input voltage range, solution
size and efficiency. The maximum operation supply voltage can be found by:
VOUT
V
=
IN_MAX
f
ì TON_MIN
SW
(5)
At lower supply voltage, the switching frequency decreases once TOFF_MIN is tripped. The minimum VIN without
frequency foldback can be approximated by:
VOUT
V
=
IN_MIN
1- f
ì TOFF _MIN
SW
(6)
Taking considerations of power losses in the system with heavy load operation, VIN_MAX is higher than the result
calculated in Equation 5. With frequency foldback, VIN_MIN is lowered by decreased fSW
.
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2500
2000
1500
1000
500
0.5 A
1.0 A
1.5 A
0
5
5.5
6
6.5
7
7.5
8
Input Voltage (V)
LMR2
Figure 20. Frequency Foldback at Dropout (VOUT = 5 V, fSW = 2100 kHz)
7.3.7 Internal Compensation and CFF
The LMR23615-Q1 is internally compensated as shown in Functional Block Diagram. The internal compensation
is designed such that the loop response is stable over the entire operating frequency and output voltage range.
Depending on the output voltage, the compensation loop phase margin can be low with all ceramic capacitors.
An external feed-forward capacitor CFF is recommended to be placed in parallel with the top resistor divider RFBT
for optimum transient performance.
VOUT
CFF
RFBT
FB
RFBB
Figure 21. Feedforward Capacitor for Loop Compensation
The feed-forward capacitor CFF in parallel with RFBT places an additional zero before the cross over frequency of
the control loop to boost phase margin. The zero frequency can be found by
1
fZ _ CFF
=
2pìCFF ìRFBT
(7)
An additional pole is also introduced with CFF at the frequency of
1
fP _ CFF
=
2pìCFF ìRFBT //RFBB
(8)
The zero fZ_CFF adds phase boost at the crossover frequency and improves transient response. The pole fP-CFF
helps maintaining proper gain margin at frequency beyond the crossover. Table 2 lists the combination of COUT
,
CFF and RFBT for typical applications, designs with similar COUT but RFBT other than recommended value, please
adjust CFF such that (CFF × RFBT) is unchanged and adjust RFBB such that (RFBT / RFBB) is unchanged.
14
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Designs with different combinations of output capacitors need different CFF. Different types of capacitors have
different equivalent series resistance (ESR). Ceramic capacitors have the smallest ESR and need the most CFF.
Electrolytic capacitors have much larger ESR, and the ESR zero frequency would be low enough to boost the
phase up around the crossover frequency. Designs using mostly electrolytic capacitors at the output may not
need any CFF.
1
fZ _ESR
=
2pìC
ìESR
OUT
(9)
The CFF creates a time constant with RFBT that couples in the attenuate output voltage ripple to the FB node. If
the CFF value is too large, it can couple too much ripple to the FB and affect VOUT regulation. Therefore, calculate
CFF base on output capacitors used in the system. At cold temperatures, the value of CFF might change based on
the tolerance of the chosen component. This may reduce its impedance and ease noise coupling on the FB
node. To avoid this, more capacitance can be added to the output or the value of CFF can be reduced.
7.3.8 Bootstrap Voltage (BOOT)
The LMR23615-Q1 provides an integrated bootstrap voltage regulator. A small capacitor between the BOOT and
SW pins provides the gate-drive voltage for the high-side MOSFET. The BOOT capacitor is refreshed when the
high-side MOSFET is off and the low-side switch conducts. The recommended value of the BOOT capacitor is
0.1 μF. For stable performance, TI recommends a ceramic capacitor with an X7R or X5R grade dielectric with a
voltage rating of 16 V or higher over temperature and voltage.
7.3.9 Overcurrent and Short-Circuit Protection
The LMR23615-Q1 is protected from overcurrent conditions by cycle-by-cycle current limit on both the peak and
valley of the inductor current. Hiccup mode is activated if a fault condition persists to prevent overheating.
High-side MOSFET over-current protection is implemented by the nature of the peak-current-mode control. The
HS switch current is sensed when the HS is turned on after a set blanking time. The HS switch current is
compared to the output of the error amplifier (EA) minus slope compensation every switching cycle. See
Functional Block Diagram for more details. The peak current of HS switch is limited by a clamped maximum peak
current threshold IHS_LIMIT, which is constant. Thus, the peak current limit of the high-side switch is not affected
by the slope compensation and remains constant over the full duty cycle range.
The current going through LS MOSFET is also sensed and monitored. When the LS switch turns on, the inductor
current begins to ramp down. The LS switch is not turned OFF at the end of a switching cycle if its current is
above the LS current limit ILS_LIMIT. The LS switch is kept ON so that inductor current keeps ramping down, until
the inductor current ramps below the LS current limit ILS_LIMIT. The LS switch is then turned OFF, and the HS
switch is turned on after a dead time. This is somewhat different than the more typical peak-current limit and
results in Equation 10 for the maximum load current.
V - V
(
)
ì
VOUT
IN
OUT
IOUT _MAX = ILS _LIMIT
+
2ì fSW ìL
V
IN
(10)
If the current of the LS switch is higher than the LS current limit for 64 consecutive cycles, hiccup current-
protection mode is activated. In hiccup mode the regulator is shut down and kept off for 5 ms typically before the
LMR23615-Q1 tries to start again. If overcurrent or short-circuit fault condition still exists, hiccup repeats until the
fault condition is removed. Hiccup mode reduces power dissipation under severe overcurrent conditions,
prevents overheating and potential damage to the device.
7.3.10 Thermal Shutdown
The LMR23615-Q1 provides an internal thermal shutdown to protect the device when the junction temperature
exceeds 170°C (typical). The device is turned off when thermal shutdown activates. Once the die temperature
falls below 155°C (typical), the device reinitiates the power up sequence controlled by the internal soft-start
circuitry.
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7.4 Device Functional Modes
7.4.1 Shutdown Mode
The EN pin provides electrical ON and OFF control for the LMR23615-Q1. When VEN is below 1 V (typical), the
device is in shutdown mode. The LMR23615-Q1 also employs VIN and VCC UVLO protection. If VIN or VCC
voltage is below their respective UVLO level, the regulator is turned off.
7.4.2 Active Mode
The LMR23615-Q1 is in active mode when VEN is above the precision enable threshold, VIN and VCC are above
their respective UVLO level. The simplest way to enable the LMR23615-Q1 is to connect the EN/SYNC pin to
VIN pin. This allows self startup when the input voltage is in the operating range: 4 V to 36 V. See VCC, UVLO
and EN/SYNC for details on setting these operating levels.
In active mode, depending on the load current, the LMR236215-Q1 is in one of three modes:
1. Continuous conduction mode (CCM) with fixed switching frequency when load current is above half of the
peak-to-peak inductor current ripple.
2. Discontinuous conduction mode (DCM) with fixed switching frequency when load current is lower than half of
the peak-to-peak inductor current ripple in CCM operation (only for PFM option).
3. Pulse frequency modulation mode (PFM) when switching frequency is decreased at very light load (only for
PFM option).
7.4.3 CCM Mode
CCM operation is employed in the LMR23615-Q1 when the load current is higher than half of the peak-to-peak
inductor current. In CCM operation, the frequency of operation is fixed, output voltage ripple is at a minimum in
this mode, and the maximum output current of 1.5 A can be supplied by the LMR23615-Q1.
7.4.4 Light Load Operation
When the load current is lower than half of the peak-to-peak inductor current in CCM, the LMR23615-Q1
operates in DCM, also known as diode emulation mode (DEM). In DCM, the LS switch is turned off when the
inductor current drops to IL_ZC (–40 mA typical). Both switching losses and conduction losses are reduced in
DCM, compared to forced PWM operation at light load.
At even lighter current loads, PFM is activated to maintain high efficiency operation. When either the minimum
HS switch ON-time (tON_MIN ) or the minimum peak inductor current IPEAK_MIN (300 mA typical) is reached, the
switching frequency decreases to maintain regulation. In PFM, switching frequency is decreased by the control
loop when load current reduces to maintain output voltage regulation. Switching loss is further reduced in PFM
operation due to less frequent switching actions. The external clock synchronizing is not valid when LMR23615-
Q1 enters into PFM mode.
16
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LMR23615-Q1 is a step-down DC-to-DC regulator. It is typically used to convert a higher DC voltage to a
lower DC voltage with a maximum output current of 1.5 A. The following design procedure can be used to select
components for the LMR23615-Q1. Alternately, the WEBENCH software may be used to generate complete
designs. When generating a design, the WEBENCH software utilizes iterative design procedure and accesses
comprehensive databases of components. See Custom Design With WEBENCH® Tools and ti.com for more
details.
8.2 Typical Applications
The LMR23615-Q1 only requires a few external components to convert from a wide voltage range supply to a
fixed output voltage. Figure 22 shows a basic schematic.
VIN 12 V
CBOOT
0.1 ꢀF
BOOT
SW
VIN
L
VOUT
5 V/1.5 A
CIN
10 ꢀF
4.7 ꢀH
EN/
SYNC
RFBT
88.7 kΩ
PAD
CFF
22 pF
COUT
33 ꢀF
FB
RT
CVCC
2.2 ꢀF
RFBB
22.1 kΩ
VCC
PGND
AGND
RT
24.3 kΩ
Copyright © 2017, Texas Instruments Incorporated
Figure 22. Application Circuit
The external components have to fulfill the needs of the application, but also the stability criteria of the device's
control loop. Table 2 can be used to simplify the output filter component selection.
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Typical Applications (continued)
Table 2. L, COUT, and CFF Typical Values
(1)
fSW (kHz)
VOUT (V)
L (µH)
22
COUT (µF)(2)
CFF (pF)(3)
220
RFBT (kΩ)(4)(5)
3.3
5
200
150
68
51
88.7
243
510
51
33
120
200
12
24
3.3
5
56
See note(3)
See note(3)
100
56
33
10
120
90
15
68
88.7
243
510
51
400
12
24
3.3
5
33
47
See note(3)
(3)
33
22
See note
4.7
5.6
10
68
47
1000
2200
47
22
See note(3)
22
88.7
243
51
12
3.3
5
33
2.2
3.3
33
22
15
88.7
(1) Inductance value is calculated based on VIN = 36 V.
(2) All the COUT values are after derating. Add more when using ceramic capacitors.
(3) High ESR COUT will give enough phase boost and CFF not needed.
(4) RFBT = 0 Ω for VOUT = 1 V. RFBB = 22.1 kΩ for all other VOUT setting.
(5) For designs with RFBT other than recommended value, adjust CFF such that (CFF × RFBT) is unchanged and adjust RFBB such that (RFBT
/ RFBB) is unchanged.
8.2.1 Design Requirements
Detailed design procedure is described based on a design example. For this design example, use the
parameters listed in Table 3 as the input parameters.
Table 3. Design Example Parameters
DESIGN PARAMETER
Input voltage, VIN
EXAMPLE VALUE
12 V typical, range from 8 V to 28 V
Output voltage, VOUT
5 V
1.5 A
Maximum output current IO_MAX
Transient response 0.2 A to 2.5 A
Output voltage ripple
5%
50 mV
400 mV
1600 kHz
Input voltage ripple
Switching frequency fSW
18
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8.2.2 Detailed Design Procedure
8.2.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LMR23615-Q1 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
•
•
•
•
Run electrical simulations to see important waveforms and circuit performance
Run thermal simulations to understand board thermal performance
Export customized schematic and layout into popular CAD formats
Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
8.2.2.2 Output Voltage Setpoint
The output voltage of LMR23615-Q1 is externally adjustable using a resistor divider network. The divider network
is comprised of top feedback resistor RFBT and bottom feedback resistor RFBB. Equation 11 is used to determine
the output voltage:
VOUT - VREF
RFBT
=
ìRFBB
VREF
(11)
Choose the value of RFBB to be 22.1 kΩ. With the desired output voltage set to 5 V and the VREF = 1 V, the RFBB
value can then be calculated using Equation 11. The formula yields to a value 88.7 kΩ.
8.2.2.3 Switching Frequency
The switching frequency can be adjusted by RT resistance from RT pin to ground. Use Equation 1 to calculate
the required value of RT. The device can also be synchronized to an external clock for a desired frequency,
please refer to EN/SYNC for more details.
For 1600 kHz frequency, the calculated RT is 24.5 kΩ, and standard value 24.3 kΩ is selected to set the
frequency approximate to 1600 kHz.
8.2.2.4 Inductor Selection
The most critical parameters for the inductor are the inductance, saturation current and the rated current. The
inductance is based on the desired peak-to-peak ripple current ΔiL. Since the ripple current increases with the
input voltage, the maximum input voltage is always used to calculate the minimum inductance LMIN. Use
Equation 13 to calculate the minimum value of the output inductor. KIND is a coefficient that represents the
amount of inductor ripple current relative to the maximum output current of the device. A reasonable value of
KIND should be 20% to 40%. During an instantaneous short-current or overcurrent operation event, the RMS and
peak inductor current can be high. The inductor current rating must be higher than the current limit of the device.
VOUT ì V
- VOUT
(
)
IN_MAX
DiL =
VIN_MAX ìL ì fSW
(12)
(13)
V
- VOUT
VOUT
IN_MAX
LMIN
=
ì
IOUT ìKIND
VIN_MAX ì fSW
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In general, it is preferable to choose lower inductance in switching power supplies, because it usually
corresponds to faster transient response, smaller DCR, and reduced size for more compact designs. But too low
of an inductance can generate too large of an inductor current ripple such that overcurrent protection at the full
load could be falsely triggered. It also generates more conduction loss and inductor core loss. Larger inductor
current ripple also implies larger output voltage ripple with same output capacitors. With peak current mode
control, it is not recommended to have too small of an inductor current ripple. A larger peak current ripple
improves the comparator signal to noise ratio.
For this design example, choose KIND = 0.4, the minimum inductor value is calculated to be 1.9 µH. Choose the
nearest standard 2.2-μH ferrite inductor with a capability of 3.5-A RMS current, and 6-A saturation current.
8.2.2.5 Output Capacitor Selection
Choose the output capacitor(s), COUT, with care because output capacitance directly affects the steady-state
output-voltage ripple, loop stability, and the voltage over/undershoot during load current transients.
The output ripple is essentially composed of two parts. One is caused by the inductor current ripple going
through the ESR of the output capacitors:
DVOUT_ESR = DiL ìESR = KIND ìIOUT ìESR
(14)
The other is caused by the inductor current ripple charging and discharging the output capacitors:
DiL
KIND ìIOUT
DVOUT _C
=
=
8ì f ìCOUT
8ì f ìCOUT
SW
SW
(15)
The two components in the voltage ripple are not in phase, so the actual peak-to-peak ripple is smaller than the
sum of two peaks.
Output capacitance is usually limited by transient performance specifications if the system requires tight voltage
regulation with presence of large current steps and fast slew rate. When a fast large-load increase uccurs, output
capacitors provide the required charge before the inductor current can slew up to the appropriate level. The
regulator control loop usually needs six or more clock cycles to respond to the output voltage droop. The output
capacitance must be large enough to supply the current difference for six clock cycles to maintain the output
voltage within the specified range. Equation 16 shows the minimum output capacitance needed for specified
output undershoot. When a sudden large load decrease happens, the output capacitors absorb energy stored in
the inductor resulting in an output voltage overshoot. Equation 17 calculates the minimum capacitance required
to keep the voltage overshoot within a specified range.
4ì IOH -IOL
(
)
COUT
>
fSW ì VUS
(16)
2
IOH2 -IOL
COUT
>
2
(VOUT + VOS)2 - VOUT
where
•
•
•
•
•
KIND = Ripple ratio of the inductor ripple current (ΔiL / IOUT
IOL = Low level output current during load transient
IOH = High level output current during load transient
VUS = Target output voltage undershoot
)
VOS = Target output voltage overshoot
(17)
For this design example, the target output ripple is 50 mV. Presuppose ΔVOUT_ESR = ΔVOUT_C = 50 mV, and
chose KIND = 0.4. Equation 14 yields ESR no larger than 83.3 mΩ and Equation 15 yields COUT no smaller than
0.9 μF. For the target over/undershoot range of this design, VUS = VOS = 5% × VOUT = 250 mV. The COUT can be
calculated to be no smaller than 14 μF and 4.1 μF by Equation 16 and Equation 17, respectively. Consider of
derating, one 33-μF, 16 V ceramic capacitor with 5 mΩ ESR is used.
20
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8.2.2.6 Feed-Forward Capacitor
The LMR23615-Q1 is internally compensated. Depending on the VOUT and frequency fSW, if the output capacitor
COUT is dominated by low ESR (ceramic types) capacitors, it could result in low phase margin. To improve the
phase boost an external feed-forward capacitor CFF can be added in parallel with RFBT. Choose CFF so that
phase margin is boosted at the crossover frequency without CFF. A simple estimation for the crossover frequency
(fX) without CFF is shown in Equation 18, assuming COUT has very small ESR, and COUT value is after derating.
8.32
fX
=
VOUT ìCOUT
(18)
Equation 19 for CFF was tested:
1
CFF
=
4pì fX ìRFBT
(19)
For designs with higher ESR, CFF is not needed when COUT has very high ESR; reduce CFF calculated from
Equation 19 with medium ESR. Table 2 can be used as a quick starting point.
For the application in this design example, a 18-pF, 50-V COG capacitor is selected.
8.2.2.7 Input Capacitor Selection
The LMR23615-Q1 device requires high-frequency input decoupling capacitor(s) and a bulk input capacitor,
depending on the application. The typical recommended value for the high frequency decoupling capacitor is 4.7
μF to 10 μF. TI recommends a high-quality ceramic capacitor type X5R or X7R with sufficiency voltage rating is
recommended. To compensate the derating of ceramic capacitors, a voltage rating of twice the maximum input
voltage. Additionally, some bulk capacitance can be required, especially if the LMR23615-Q1 circuit is not
located within approximately 5 cm from the input voltage source. This capacitor is used to provide damping to the
voltage spike due to the lead inductance of the cable or the trace. For this design, two 4.7-μF, 50-V, X7R ceramic
capacitors are used. For high-frequency filtering place a 0.1-µF capacitor as close as possible to the device pins.
8.2.2.8 Bootstrap Capacitor Selection
Every LMR23615-Q1 design requires a bootstrap capacitor (CBOOT). The recommended capacitor is 0.1 μF and
rated 16 V or higher. The bootstrap capacitor is located between the SW pin and the BOOT pin. The bootstrap
capacitor must be a high-quality ceramic type with an X7R or X5R grade dielectric for temperature stability.
8.2.2.9 VCC Capacitor Selection
The VCC pin is the output of an internal LDO for LMR23615-Q1. To insure stability of the device, place a
minimum of 2.2-μF, 16-V, X7R capacitor from VCC pin to ground.
8.2.2.10 Undervoltage Lockout Setpoint
The system UVLO is adjusted using the external voltage divider network of RENT and RENB. The UVLO has two
thresholds, one for power up when the input voltage is rising and one for power down or brownouts when the
input voltage is falling. Use Equation 20 to determine the VIN UVLO level.
RENT + RENB
V
= VENH ì
IN_RISING
RENB
(20)
The EN rising threshold (VENH) for LMR23615-Q1 is set to be 1.55 V (typical). Choose the value of RENB to be
287 kΩ to minimize input current from the supply. If the desired VIN UVLO level is at 6 V, the value of RENT can
be calculated using Equation 21:
V
≈
’
IN_RISING
RENT
=
-1 ìR
∆
∆
÷
ENB
÷
VENH
«
◊
(21)
Equation 21 yields a value of 820 kΩ. The resulting falling UVLO threshold, equals 4.4 V, can be calculated by
Equation 22, where EN hysteresis (VEN_HYS) is 0.4 V (typical).
RENT + RENB
V
= VENH - VEN_HYS
(
ì
)
IN_FALLING
RENB
(22)
21
Copyright © 2018, Texas Instruments Incorporated
LMR23615-Q1
ZHCSHR8 –MARCH 2018
www.ti.com.cn
8.2.3 Application Curves
Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 1600 kHz, L = 4.7 µH, COUT = 47 µF, TA = 25°C.
VOUT = 5 V
IOUT = 1.5 A
fSW = 1600 kHz
VOUT = 5 V
IOUT = 0 mA
fSW = 1600 kHz
Figure 23. CCM Mode
Figure 24. PFM Mode
VIN = 12 V
VOUT = 5 V
IOUT = 1.5 A
VIN = 12 V
VOUT = 5 V
IOUT = 1.5 A
Figure 25. Start-Up by VIN
Figure 26. Start-Up by EN
VIN = 12 V
VOUT = 5 V
VOUT = 5 V
VIN = 8 V to 36 V, 2 V / μs
IOUT = 1.5 A
IOUT = 0.2 A to 1.5 A, 100 mA / μs
Figure 27. Load Transient
Figure 28. Line Transient
22
Copyright © 2018, Texas Instruments Incorporated
LMR23615-Q1
www.ti.com.cn
ZHCSHR8 –MARCH 2018
Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 1600 kHz, L = 4.7 µH, COUT = 47 µF, TA = 25°C.
VOUT = 5 V
IOUT = 1 A to short
VOUT = 5 V
IOUT = short to 1 A
Figure 29. Short Protection
Figure 30. Short Recovery
9 Power Supply Recommendations
The LMR23615-Q1 is designed to operate from an input voltage supply range between 4 V and 36 V. This input
supply must be able to withstand the maximum input current and maintain a stable voltage. The resistance of the
input supply rail must be low enough that an input current transient does not cause a high enough drop at the
LMR23615-Q1 supply voltage that can cause a false UVLO fault triggering and system reset. If the input supply
is located more than a few inches from the LMR23615-Q1, additional bulk capacitance may be required in
addition to the ceramic input capacitors. The amount of bulk capacitance is not critical, but a 47-μF or 100-μF
electrolytic capacitor is a typical choice.
10 Layout
10.1 Layout Guidelines
Layout is a critical portion of good power supply design. The following guidelines will help users design a PCB
with the best power conversion performance, thermal performance, and minimized generation of unwanted EMI.
1. The input bypass capacitor CIN must be placed as close as possible to the VIN and PGND pins. Grounding
for both the input and output capacitors must consist of localized top side planes that connect to the PGND
pin and PAD.
2. Place bypass capacitors for VCC close to the VCC pin and ground the bypass capacitor to device ground.
3. Minimize trace length to the FB pin net. Both feedback resistors, locate RFBT and RFBB close to the FB pin.
Place CFF directly in parallel with RFBT. If VOUT accuracy at the load is important, make sure VOUT sense is
made at the load. Route VOUT sense path away from noisy nodes and preferably through a layer on the other
side of a shielded layer.
4. Use ground plane in one of the middle layers as noise shielding and heat dissipation path.
5. Have a single point ground connection to the plane. Route the ground connections for the feedback and
enable components to the ground plane. This prevents any switched or load currents from flowing in the
analog ground traces. If not properly handled, poor grounding can result in degraded load regulation or
erratic output voltage ripple behavior.
6. Make VIN, VOUT and ground bus connections as wide as possible. This reduces any voltage drops on the
input or output paths of the converter and maximizes efficiency.
7. Provide adequate device heat sinking. Use an array of heat-sinking vias to connect the exposed pad to the
ground plane on the bottom PCB layer. If the PCB has multiple copper layers, these thermal vias can also be
connected to inner layer heat-spreading ground planes. Ensure enough copper area is used for heat-sinking
to keep the junction temperature below 125°C.
Copyright © 2018, Texas Instruments Incorporated
23
LMR23615-Q1
ZHCSHR8 –MARCH 2018
www.ti.com.cn
Layout Guidelines (continued)
10.1.1 Compact Layout for EMI Reduction
Radiated EMI is generated by the high di/dt components in pulsing currents in switching converters. The larger
area covered by the path of a pulsing current, the more EMI is generated. High-frequency ceramic bypass
capacitors at the input side provide primary path for the high di/dt components of the pulsing current. Placing
ceramic bypass capacitor(s) as close as possible to the VIN and PGND pins is the key to EMI reduction.
The SW pin connecting to the inductor must be as short as possible, and just wide enough to carry the load
current without excessive heating. Use short, thick traces or copper pours (shapes) for high-current conduction
path to minimize parasitic resistance. Place the output capacitors close to the VOUT end of the inductor and
closely grounded to PGND pin and exposed PAD.
Place the bypass capacitors on VCC as close as possible to the pin and closely grounded to PGND and the
exposed PAD.
10.1.2 Ground Plane and Thermal Considerations
TI recommends using one of the middle layers as a solid ground plane. Ground plane provides shielding for
sensitive circuits and traces. It also provides a quiet reference potential for the control circuitry. Connect the
AGND and PGND pins to the ground plane using vias right next to the bypass capacitors. PGND pin is
connected to the source of the internal LS switch. They should be connected directly to the grounds of the input
and output capacitors. The PGND net contains noise at switching frequency and may bounce due to load
variations. PGND trace, as well as VIN and SW traces, should be constrained to one side of the ground plane.
The other side of the ground plane contains much less noise and must be used for sensitive routes.
TI recommends providing adequate device heat sinking by utilizing the PAD of the IC as the primary thermal
path. Use a minimum 4 by 2 array of 12 mil thermal vias to connect the PAD to the system ground plane heat
sink. The vias must be evenly distributed under the PAD. Use as much copper as possible, for system ground
plane, on the top and bottom layers for the best heat dissipation. Use a four-layer board with the copper
thickness for the four layers, starting from the top of, 2 oz / 1 oz / 1 oz / 2 oz. Four layer boards with enough
copper thickness provides low current conduction impedance, proper shielding and lower thermal resistance.
The thermal characteristics of the LMR23615-Q1 are specified using the parameter RθJA, which characterize the
junction temperature of silicon to the ambient temperature in a specific system. Although the value of RθJA is
dependent on many variables, it still can be used to approximate the operating junction temperature of the
device. To obtain an estimate of the device junction temperature, one may use Equation 23:
TJ = PD x RθJA + TA
where
•
•
•
•
•
TJ = junction temperature in °C
PD = VIN × IIN × (1 – efficiency) – 1.1 × IOUT2 × DCR in Watt
DCR = Inductor DC parasitic resistance in Ω
RθJA = Junction-to-ambient thermal resistance of the device in °C/W
TA = ambient temperature in °C
(23)
The maximum operating junction temperature of the LMR23615-Q1 is 125°C. RθJA is highly related to PCB size
and layout, as well as environmental factors such as heat sinking and air flow.
10.1.3 Feedback Resistors
To reduce noise sensitivity of the output voltage feedback path, it is important to place the resistor divider and
CFF close to the FB pin, rather than close to the load. The FB pin is the input to the error amplifier, so it is a high
impedance node and very sensitive to noise. Placing the resistor divider and CFF closer to the FB pin reduces the
trace length of FB signal and reduces noise coupling. The output node is a low impedance node, so the trace
from VOUT to the resistor divider can be long if short path is not available.
24
Copyright © 2018, Texas Instruments Incorporated
LMR23615-Q1
www.ti.com.cn
ZHCSHR8 –MARCH 2018
Layout Guidelines (continued)
If voltage accuracy at the load is important, make sure voltage sense is made at the load. Doing so will correct
for voltage drops along the traces and provide the best output accuracy. The voltage sense trace from the load to
the feedback resistor divider should be routed away from the SW node path and the inductor to avoid
contaminating the feedback signal with switch noise, while also minimizing the trace length. This is most
important when high value resistors are used to set the output voltage. It is recommended to route the voltage
sense trace and place the resistor divider on a different layer than the inductor and SW node path, such that
there is a ground plane in between the feedback trace and inductor/SW node polygon. This provides further
shielding for the voltage feedback path from EMI noises.
10.2 Layout Example
Output
Inductor
Output Bypass
Capacitor
PGND
NC
SW
SW
Input Bypass
Capacitor
BOOT
Capacitor
BOOT
VCC
FB
VIN
VIN
VCC
Capacitor
EN/SYNC
AGND
UVLO Adjust
Resistor
RT
RT
Thermal VIA
VIA (Connect to GND Plane)
Output Voltage
Set Resistor
Figure 31. LMR23615-Q1 Layout
版权 © 2018, Texas Instruments Incorporated
25
LMR23615-Q1
ZHCSHR8 –MARCH 2018
www.ti.com.cn
11 器件和文档支持
11.1 器件支持
11.1.1 开发支持
11.1.1.1 使用 WEBENCH® 工具创建定制设计
请单击此处,使用 LMR23615-Q1 器件并借助 WEBENCH® 电源设计器创建定制设计方案。
1. 首先键入输入电压 (VIN)、输出电压 (VOUT) 和输出电流 (IOUT) 要求。
2. 使用优化器拨盘优化关键参数设计,如效率、封装和成本。
3. 将生成的设计与德州仪器 (TI) 的其他解决方案进行比较。
WEBENCH 电源设计器可提供定制原理图以及罗列实时价格和组件供货情况的物料清单。
在多数情况下,可执行以下操作:
•
•
•
•
运行电气仿真,观察重要波形以及电路性能
运行热性能仿真,了解电路板热性能
将定制原理图和布局方案导出至常用 CAD 格式
打印设计方案的 PDF 报告并与同事共享
有关 WEBENCH 工具的详细信息,请访问 www.ti.com.cn/WEBENCH。
11.2 接收文档更新通知
要接收文档更新通知,请导航至 TI.com.cn 上的器件产品文件夹。请单击右上角的提醒我 进行注册,即可每周接收
产品信息更改摘要。有关更改的详细信息,请查看任何已修订文档中包含的修订历史记录。
11.3 社区资源
下列链接提供到 TI 社区资源的连接。链接的内容由各个分销商“按照原样”提供。这些内容并不构成 TI 技术规范,
并且不一定反映 TI 的观点;请参阅 TI 的 《使用条款》。
TI E2E™ 在线社区 TI 的工程师对工程师 (E2E) 社区。此社区的创建目的在于促进工程师之间的协作。在
e2e.ti.com 中,您可以咨询问题、分享知识、拓展思路并与同行工程师一道帮助解决问题。
设计支持
TI 参考设计支持 可帮助您快速查找有帮助的 E2E 论坛、设计支持工具以及技术支持的联系信息。
11.4 商标
PowerPAD, E2E are trademarks of Texas Instruments.
WEBENCH, SIMPLE SWITCHER are registered trademarks of Texas Instruments.
All other trademarks are the property of their respective owners.
11.5 静电放电警告
这些装置包含有限的内置 ESD 保护。 存储或装卸时,应将导线一起截短或将装置放置于导电泡棉中,以防止 MOS 门极遭受静电损
伤。
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 机械、封装和可订购信息
以下页面包含机械、封装和可订购信息。这些信息是指定器件的最新可用数据。数据如有变更,恕不另行通知,也
不会对此文档进行修订。如欲获取此数据表的浏览器版本,请参阅左侧的导航。
26
版权 © 2018, Texas Instruments Incorporated
PACKAGE OPTION ADDENDUM
www.ti.com
23-Jun-2023
PACKAGING INFORMATION
Orderable Device
Status Package Type Package Pins Package
Eco Plan
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
Samples
Drawing
Qty
(1)
(2)
(3)
(4/5)
(6)
LMR23615QDRRRQ1
LMR23615QDRRTQ1
ACTIVE
ACTIVE
WSON
WSON
DRR
DRR
12
12
3000 RoHS & Green
250 RoHS & Green
SN
Level-2-260C-1 YEAR
Level-2-260C-1 YEAR
-40 to 125
-40 to 125
3615Q
3615Q
Samples
Samples
SN
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two
lines if the finish value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
23-Jun-2023
OTHER QUALIFIED VERSIONS OF LMR23615-Q1 :
Catalog : LMR23615
•
NOTE: Qualified Version Definitions:
Catalog - TI's standard catalog product
•
Addendum-Page 2
PACKAGE OUTLINE
DRR0012D
WSON - 0.8 mm max height
SCALE 4.000
PLASTIC SMALL OUTLINE - NO LEAD
3.1
2.9
B
A
PIN 1 INDEX AREA
3.1
2.9
0.1 MIN
(0.05)
S
C
A
L
E
3
0
.
A
SECTION A-A
TYPICAL
0.8
0.7
C
SEATING PLANE
0.08 C
0.05
0.00
EXPOSED
THERMAL PAD
(0.2) TYP
1.7 0.1
6
7
A
A
13
2X
2.5
2.5 0.1
1
12
10X 0.5
0.3
0.2
12X
0.38
0.28
12X
PIN 1 ID
0.1
C A B
C
(OPTIONAL)
0.05
4223146/D 10/2018
NOTES:
1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. The package thermal pad must be soldered to the printed circuit board for thermal and mechanical performance.
www.ti.com
EXAMPLE BOARD LAYOUT
DRR0012D
WSON - 0.8 mm max height
PLASTIC SMALL OUTLINE - NO LEAD
(1.7)
12X (0.53)
SYMM
1
12
12X (0.25)
13
SYMM
(2.5)
10X (0.5)
(1)
(R0.05) TYP
6
7
(0.6)
(2.87)
(
0.2) VIA
TYP
LAND PATTERN EXAMPLE
EXPOSED METAL SHOWN
SCALE:20X
0.07 MIN
ALL AROUND
0.07 MAX
ALL AROUND
EXPOSED METAL
EXPOSED METAL
SOLDER MASK
OPENING
METAL EDGE
SOLDER MASK
OPENING
METAL UNDER
SOLDER MASK
NON SOLDER MASK
DEFINED
SOLDER MASK
DEFINED
(PREFERRED)
SOLDER MASK DETAILS
4223146/D 10/2018
NOTES: (continued)
4. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature
number SLUA271 (www.ti.com/lit/slua271).
5. Vias are optional depending on application, refer to device data sheet. If any vias are implemented, refer to their locations shown
on this view. It is recommended that vias under paste be filled, plugged or tented.
www.ti.com
EXAMPLE STENCIL DESIGN
DRR0012D
WSON - 0.8 mm max height
PLASTIC SMALL OUTLINE - NO LEAD
SYMM
(0.47)
12X (0.53)
1
12
12X (0.25)
METAL
TYP
(0.675)
SYMM
13
10X (0.5)
(1.15)
(R0.05) TYP
6
7
(0.74)
(2.87)
SOLDER PASTE EXAMPLE
BASED ON 0.125 mm THICK STENCIL
EXPOSED PAD
80.1% PRINTED SOLDER COVERAGE BY AREA
SCALE:25X
4223146/D 10/2018
NOTES: (continued)
6. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
www.ti.com
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