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LMV221

SNWS018D –DECEMBER 2006–REVISED JUNE 2016

LMV221 50-MHz to 3.5-GHz 40-dB Logarithmic Power Detector for CDMA and WCDMA

1 Features

3 Description

The LMV221 is a 40-dB RF power detector intended

for use in CDMA and WCDMA applications. The

device has an RF frequency range from 50 MHz to

3.5 GHz. It provides an accurate temperature and

supply-compensated output voltage that relates

linearly to the RF input power in dBm. The circuit

operates with a single supply from 2.7 V to 3.3 V.

1

•

2.7-V to 3.3-V Supply Voltage

40-dB Linear in dB Power Detection Range

0.3-V to 2-V Output Voltage Range

Shutdown

Multi-Band Operation from 50 MHz to 3.5 GHz

0.5-dB Accurate Temperature Compensation

External Configurable Output Filter Bandwidth

The LMV221 has an RF power detection range from

−45 dBm to −5 dBm and is ideally suited for direct

use in combination with a 30-dB directional coupler.

Additional low-pass filtering of the output signal can

be achieved by means of an external resistor and

capacitor. shows a detector with an additional output

low pass filter. The filter frequency is set with R_Sand

C_S.

2.5 mm × 2.2 mm × 0.8 mm 6-pin WSON

Package

2 Applications

•

UMTS/CDMA/WCDMA RF Power Control

GSM/GPRS RF Power Control

PA Modules

shows a detector with an additional feedback low

pass filter. Resistor R_Pis optional and lowers the

trans-impedance gain (R_TRANS). The filter frequency is

IEEE 802.11b, g (WLAN)

set with C_P//C_TRANSand R_P//R_TRANS

.

The device is active for Enable = High, otherwise it is

in a low power consumption shutdown mode. To save

power and prevent discharge of an external filter

capacitance, the output (OUT) is high-impedance

during shutdown.

Device Information⁽¹⁾

PART NUMBER

PACKAGE

BODY SIZE (NOM)

LMV221

WSON (6)

2.50 mm × 2.20 mm

(1) For all available packages, see the orderable addendum at

the end of the data sheet.

space

Typical Application: Output RC Low Pass Filter

Typical Application: Feedback (R)C Low Pass Filter

COUPLER

ANTENNA

RF

ANTENNA

RF

PA

50 W

VDD

R_S

RFIN

OUT

1

RFIN

OUT

1

2

6

+

-

6

+

-

2

C_S

ADC

LMV221

R_P

C_P

ADC

LMV221

REF

EN

REF

4

5

4

5

3

GND

1

An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,

intellectual property matters and other important disclaimers. PRODUCTION DATA.

LMV221

SNWS018D –DECEMBER 2006–REVISED JUNE 2016

www.ti.com

Table of Contents

7.3 Feature Description................................................. 23

7.4 Device Functional Modes........................................ 30

Application and Implementation ........................ 31

8.1 Application Information............................................ 31

8.2 Typical Applications ............................................... 34

Power Supply Recommendations...................... 38

1

2

3

4

5

6

Features.................................................................. 1

Applications ........................................................... 1

Description ............................................................. 1

Revision History..................................................... 2

Pin Configuration and Functions......................... 3

Specifications......................................................... 4

6.1 Absolute Maximum Ratings ...................................... 4

6.2 ESD Ratings.............................................................. 4

6.3 Recommended Operating Conditions....................... 4

6.4 Thermal Information.................................................. 4

6.5 2.7-V DC and AC Electrical Characteristics.............. 5

6.6 Timing Requirements.............................................. 11

6.7 Typical Characteristics............................................ 12

Detailed Description ............................................ 23

7.1 Overview ................................................................. 23

7.2 Functional Block Diagram ....................................... 23

8

9

10 Layout................................................................... 39

10.1 Layout Guidelines ................................................ 39

10.2 Layout Example .................................................... 41

11 Device and Documentation Support ................. 42

11.1 Community Resources.......................................... 42

11.2 Trademarks........................................................... 42

11.3 Electrostatic Discharge Caution............................ 42

11.4 Glossary................................................................ 42

7

12 Mechanical, Packaging, and Orderable

Information ........................................................... 42

4 Revision History

NOTE: Page numbers for previous revisions may differ from page numbers in the current version.

Changes from Revision C (March 2013) to Revision D

Page

•

Added Device Information and Pin Configuration and Functions sections, ESD Ratings table and Thermal

Information table, Feature Description, Device Functional Modes, Application and Implementation, Power Supply

Recommendations, Layout, Device and Documentation Support, and Mechanical, Packaging, and Orderable

Information sections................................................................................................................................................................ 1

Changed R_θJAvalue from 86.6°C/W to 100.4°C/W................................................................................................................. 4

Changes from Revision B (March 2013) to Revision C

Page

•

Changed layout of National Semiconductor data sheet to TI format.................................................................................... 41

2

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5 Pin Configuration and Functions

NGF Package

6-Pin WSON

Top View

VDD

1

OUT

6

5

4

2

3

REF

EN

RFIN

GND

DAP

(GND)

Pin Functions

TYPE

DESCRIPTION

NUMBER

NAME

VDD

1

3

2

Positive supply voltage

Power ground

Power supply

GND

RFIN

Analog input

Logic input

RF input signal to the detector, internally terminated with 50 Ω.

The device is enabled for EN = high, and brought to a low-power shutdown

mode for EN = Low.

4

5

EN

Reference output, for differential output measurement (without pedestal).

Connected to inverting input of output amplifier.

REF

Output

—

6

OUT

GND

Ground referenced detector output voltage (linear in dB)

Ground (must be connected)

DAP

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6 Specifications

6.1 Absolute Maximum Ratings

over operating free-air temperature range (unless otherwise noted)⁽¹⁾

MIN

MAX

UNIT

SUPPLY VOLTAGE

VDD - GND

3.6

V

RF INPUT

Input power

10

dBm

mV

DC voltage

400

Enable input voltage

Junction temperature⁽²⁾

Maximum lead temperature (soldering, 10 seconds)

Storage temperature, T_stg

V_SS– 0.4 V < V_EN< V_DD+ 0.4 V

150

260

°C

–65

150

(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings

only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended

Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

(2) The maximum power dissipation is a function of T_J(MAX), R_θJA. The maximum allowable power dissipation at any ambient temperature is

P_D= (T_J(MAX)– T_A)/R_θJA. All numbers apply for packages soldered directly into a PC board.

6.2 ESD Ratings

VALUE

UNIT

Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001⁽¹⁾

±2000

Charged-device model (CDM), per JEDEC specification JESD22-

C101⁽²⁾

V_(ESD)

Electrostatic discharge

±2000

±200

V

Machine model

(1) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.

(2) JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.

6.3 Recommended Operating Conditions

over operating free-air temperature range (unless otherwise noted)

MIN

2.7

NOM

MAX

3.3

UNIT

V

Supply voltage

Ambient temperature

RF frequency

–40

50

85

°C

3500

–5

MHz

dBm

dBV

–45

–58

RF input power⁽¹⁾

–18

(1) Power in dBV = dBm + 13 when the impedance is 50 Ω.

6.4 Thermal Information

LVM221

THERMAL METRIC⁽¹⁾

NGF (WSON)

UNIT

6 PINS

100.4

120

R_θJA

Junction-to-ambient thermal resistance⁽²⁾

Junction-to-case (top) thermal resistance

Junction-to-board thermal resistance

°C/W

R_θJC(top)

R_θJB

7

ψ_JT

Junction-to-top characterization parameter

Junction-to-board characterization parameter

Junction-to-case (bottom) thermal resistance

69.6

6.9

ψ_JB

R_θJC(bot)

69.9

(1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics, SPRA953.

(2) The maximum power dissipation is a function of T_J(MAX), R_θJA. The maximum allowable power dissipation at any ambient temperature is

P_D= (T_J(MAX)– T_A)/R_θJA. All numbers apply for packages soldered directly into a PC board.

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6.5 2.7-V DC and AC Electrical Characteristics

Unless otherwise specified, all limits are ensured at T_A= 25°C, V_DD= 2.7 V, RF input frequency ƒ = 1855-MHz continuous

wave (CW), modulated.⁽¹⁾

PARAMETER

SUPPLY INTERFACE

TEST CONDITIONS

MIN⁽²⁾

TYP⁽³⁾

MAX⁽²⁾UNIT

Active mode: EN = high, no signal present at RF_IN

6.5

5

7.2

8.5

mA

10

Active mode: EN = high, no signal present at RF_IN

T_A= –40°C to 85°C

Shutdown: EN = low, no signal present at RF_IN

0.5

3

I_DD

Supply current

Shutdown: EN = low, no signal present at RF_IN

T_A= –40°C to 85°C

4

µA

EN = Low: P_IN= 0 dBm⁽⁴⁾

T_A= –40°C to 85°C

10

LOGIC ENABLE INTERFACE

EN logic low input level

V_LOW

T_A= –40°C to 85°C

0.6

V

(shutdown mode)

V_HIGH

I_EN

EN logic high input level

Current into EN pin

T_A= –40°C to 85°C

1.1

40

V

1

µA

RF INPUT INTERFACE

R_INInput resistance

47.1

60

Ω

(1) 2.7-V DC and AC Electrical Characteristics values apply only for factory testing conditions at the temperature indicated. Factory testing

conditions result in very limited self-heating of the device such that T_J= T_A. No specification of parametric performance is indicated in

the electrical tables under conditions of internal self-heating where T_J> T_A.

(2) All limits are ensured by test or statistical analysis.

(3) Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary

over time and also depend on the application and configuration. The typical values are not tested and are not specified on shipped

production material.

(4) All limits are ensured by design and measurements which are performed on a limited number of samples. Limits represent the mean

±3–sigma values. The typical value represents the statistical mean value.

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2.7-V DC and AC Electrical Characteristics (continued)

Unless otherwise specified, all limits are ensured at T_A= 25°C, V_DD= 2.7 V, RF input frequency ƒ = 1855-MHz continuous

wave (CW), modulated.⁽¹⁾

PARAMETER

OUTPUT INTERFACE

TEST CONDITIONS

MIN⁽²⁾

TYP⁽³⁾

MAX⁽²⁾UNIT

From positive rail, sourcing,

V_REF= 0 V, I_OUT= 1 mA

16

40

50

From positive rail, sourcing,

V_REF= 0 V, I_OUT= 1 mA

T_A= –40°C to 85°C

V_OUT

Output voltage swing

mV

40

From negative rail, sinking,

V_REF= 2.7 V, I_OUT= 1 mA

14

From negative rail, sinking,

V_REF= 2.7 V, I_OUT= 1 mA

T_A= –40°C to 85°C

50

Sourcing, V_REF= 0 V, V_OUT= 2.6 V

3

2.7

3

5.4

5.7

Sourcing, V_REF= 0 V, V_OUT= 2.6 V

T_A= –40°C to 85°C

Output short circuit

current

I_OUT

mA

Sinking, V_REF= 2.7 V, V_OUT= 0.1 V

T_A= –40°C to 85°C

2.7

No RF input signal. Measured from REF input

current to V_OUT

BW

Small signal bandwidth

450

kHz

R_TRANS

Output amplifier

transimpedance gain

No RF input signal, from I_REFto V_OUT, DC

35

3

42.7

4.1

55

kΩ

Positive, V_REFfrom 2.7 V to 0 V

T_A= –40°C to 85°C

2.7

3

SR

Slew rate

V/µs

Negative, V_REFfrom 0 V to 2.7 V

4.2

0.6

21

Negative, V_REFfrom 0 V to 2.7 V

T_A= –40°C to 85°C

2.7

No RF input signal, EN = High, DC measurement

5

6

R_OUT

Output impedance⁽⁴⁾

Ω

No RF input signal, EN = High, DC measurement

T_A= –40°C to 85°C

EN = Low, V_OUT= 2 V

300

500

Output leakage current in

shutdown mode

I_OUT,SD

nA

EN = Low, V_OUT= 2 V

T_A= –40°C to 85°C

RF DETECTOR TRANSFER

ƒ = 50 MHz, P_IN= −5 dBm

1.76

1.75

1.61

1.49

1.4

ƒ = 50 MHz, P_IN= −5 dBm

T_A= –40°C to 85°

1.67

1.53

1.42

1.83

1.82

1.68

1.57

ƒ = 900 MHz, P_IN= −5 dBm

T_A= –40°C to 85°C

ƒ = 1855 MHz, P_IN= −5 dBm

T_A= –40°C to 85°C

Maximum output

V_OUT,MAX

V

voltage⁽⁴⁾

ƒ = 2500 MHz, P_IN= −5 dBm

T_A= –40°C to 85°C

ƒ = 3000 MHz, P_IN= −5 dBm

T_A= –40°C to 85°C

1.33

1.21

1.48

1.36

ƒ = 3500 MHz, P_IN= −5 dBm

1.28

ƒ = 3500 MHz, T_A= –40°C to 85°C

6

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2.7-V DC and AC Electrical Characteristics (continued)

Unless otherwise specified, all limits are ensured at T_A= 25°C, V_DD= 2.7 V, RF input frequency ƒ = 1855-MHz continuous

wave (CW), modulated.⁽¹⁾

PARAMETER

TEST CONDITIONS

No input signal

MIN⁽²⁾

TYP⁽³⁾

MAX⁽²⁾UNIT

175

250

350

Minimum output voltage

(pedestal)

No input signal, T_A= –40°C to 85°C

142

388

mV

V_OUT,MIN

Pedestal variation over

temperature

No input signal, relative to 25°C

T_A= –40°C to 85°C

–20

1.37

1.34

1.24

1.14

1.07

0.96

20

1.52

1.47

ƒ = 50 MHz, P_INfrom −45 dBm to −5 dBm

1.44

1.4

ƒ = 50 MHz, P_INfrom −45 dBm to −5 dBm

T_A= –40°C to 85°C

ƒ = 900 MHz, P_INfrom −45 dBm to −5 dBm

T_A= –40°C to 85°C

ƒ = 1855 MHz, P_INfrom −45 dBm to −5 dBm

1.3

ƒ = 1855 MHz, P_INfrom −45 dBm to −5 dBm

T_A= –40°C to +85°C

1.37

V

ΔV_OUT

Output voltage range⁽⁴⁾

ƒ = 2500 MHz, P_INfrom −45 dBm to −5 dBm

1.2

ƒ = 2500 MHz, P_INfrom −45 dBm to −5 dBm

T_A= –40°C to 85°C

1.3

1.2

ƒ = 3000 MHz, P_INfrom −45 dBm to −5 dBm

1.12

1.01

ƒ = 3000 MHz, P_INfrom −45 dBm to −5 dBm

T_A= –40°C to 85°C

ƒ = 3500 MHz, P_INfrom −45 dBm to −5 dBm

T_A= –40°C to 85°C

1.09

ƒ = 50 MHz

39

36.7

34.4

32.6

31

40.5

38.5

35.7

33.8

32.5

31.9

−49.4

−52.8

−51.7

−50

42

40

ƒ = 900 MHz

ƒ = 1855 MHz

37.1

K_SLOPE

Logarithmic slope⁽⁴⁾

mV/dB

35.2

ƒ = 2500 MHz

ƒ = 3000 MHz

34

33.5

ƒ = 3500 MHz

30

ƒ = 50 MHz

–50.4

–54.1

–53.2

–51.8

–51.1

–49.6

–48.3

–51.6

ƒ = 900 MHz

ƒ = 1855 MHz

–50.2

dBm

P_INT

Logarithmic intercept⁽⁴⁾

ƒ = 2500 MHz

–48.3

ƒ = 3000 MHz

−48.9

−46.8

1.5

–46.6

ƒ = 3500 MHz

–44.1

e_n

v_N

Output referred noise⁽⁵⁾

Output referred noise⁽⁴⁾

P_IN= −10 dBm at 10 kHz

Integrated over frequency band, 1 kHz to 6.5 kHz

µV/√Hz

100

µV_RMS

150

Integrated over frequency band, 1 kHz to 6.5 kHz

T_A= –40°C to 85°C

P_IN= −10 dBm, ƒ = 1800 MHz

60

Power supply rejection

ratio⁽⁵⁾

PSRR

dB

P_IN= −10 dBm, ƒ = 1800 MHz

T_A= –40°C to 85°C

55

(5) This parameter is ensured by design and/or characterization and is not tested in production.

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2.7-V DC and AC Electrical Characteristics (continued)

Unless otherwise specified, all limits are ensured at T_A= 25°C, V_DD= 2.7 V, RF input frequency ƒ = 1855-MHz continuous

wave (CW), modulated.⁽¹⁾

PARAMETER

TEST CONDITIONS

MIN⁽²⁾

TYP⁽³⁾

MAX⁽²⁾UNIT

POWER MEASUREMENT PERFORMANCE

ƒ = 50 MHz

−40 dBm ≤ P_IN≤ −10 dBm

–0.6

–1.1

–0.7

–1.24

–0.4

–1.1

–0.43

–1

0.56

1.3

ƒ = 50 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

0.53

0.46

0.48

0.51

0.56

ƒ = 900 MHz

−40 dBm ≤ P_IN≤ −10 dBm

0.37

1.1

ƒ = 900 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

ƒ = 1855 MHz

−40 dBm ≤ P_IN≤ −10 dBm

0.24

ƒ = 1855 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

1.1

dB

E_LC

Log conformance error⁽⁴⁾

ƒ = 2500 MHz

−40 dBm ≤ P_IN≤ −10 dBm

0.56

1.1

ƒ = 2500 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

ƒ = 3000 MHz

−40 dBm ≤ P_IN≤ −10 dBm

–0.87

–1.2

1.34

1.6

ƒ = 3000 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

ƒ = 3500 MHz

−40 dBm ≤ P_IN≤ −10 dBm

–1.73

–2

2.72

2.7

ƒ = 3500 MHz, T_A= –40°C to 85°C

0.84

0.4

ƒ = 50 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

–1.1

1.4

ƒ = 900 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

–1

–1.1

0.38

0.44

0.48

0.5

1.27

ƒ = 1855 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

1.31

dB

Variation over

temperature⁽⁴⁾

E_VOT

ƒ = 2500 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

–1.1

1.15

ƒ = 3000 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

–1.2

0.98

0.85

ƒ = 3500 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

–1.2

0.62

ƒ = 50 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

–0.06

–0.056

–0.069

–0.084

–0.092

–0.1

0.069

0.056

ƒ = 900 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

ƒ = 1855 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

0.069

dB

Measurement error for a

1-dB Input power step⁽⁴⁾

E_{1 dB}

ƒ = 2500 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

0.084

ƒ = 3000 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

0.092

0.1

ƒ = 3500 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

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2.7-V DC and AC Electrical Characteristics (continued)

Unless otherwise specified, all limits are ensured at T_A= 25°C, V_DD= 2.7 V, RF input frequency ƒ = 1855-MHz continuous

wave (CW), modulated.⁽¹⁾

PARAMETER

TEST CONDITIONS

MIN⁽²⁾

TYP⁽³⁾

MAX⁽²⁾UNIT

ƒ = 50 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

–0.65

0.57

ƒ = 900 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

–0.75

–0.88

–0.86

–0.85

–0.76

0.58

ƒ = 1855 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

0.72

dB

Measurement Error for a

10-dB Input power step

E_{10 dB}

(4)

ƒ = 2500 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

0.75

ƒ = 3000 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

0.77

0.74

ƒ = 3500 MHz, T_A= –40°C to 85°C

−40 dBm ≤ P_IN≤ −10 dBm

ƒ = 50 MHz, −40 dBm ≤ P_IN≤ −10 dBm

–7

–6

ƒ = 50 MHz, −40°C < T_A< 25°C

–15

–13.4

–14.1

–13.4

–11.7

–10.5

–12.3

–13.1

–14.7

–15.9

–18

1

(4)

ƒ = 900 MHz,−40 dBm ≤ P_IN≤ −10 dBm

ƒ = 900 MHz, −40°C < T_A< 25°C

−40 dBm ≤ P_IN≤ −10 dBm⁽⁴⁾

1.5

ƒ = 1855 MHz, −40 dBm ≤ P_IN≤ −10 dBm

–5.9

–4.1

–1.8

0.5

ƒ = 1855 MHz, −40°C < T_A< 25°C

−40 dBm ≤ P_IN≤ −10 dBm⁽⁴⁾

2.3

S_T

Temperature sensitivity

mdB/°C

ƒ = 2500 MHz,−40 dBm ≤ P_IN≤ −10 dBm

ƒ = 2500 MHz, −40°C < T_A< 25°C

−40 dBm ≤ P_IN≤ −10 dBm⁽⁴⁾

5.2

8

ƒ = 3000 MHz, −40 dBm ≤ P_IN≤ −10 dBm

ƒ = 3000 MHz, −40°C < T_A< 25°C

−40 dBm ≤ P_IN≤ −10 dBm⁽⁴⁾

ƒ = 3500 MHz, −40 dBm ≤ P_IN≤ −10 dBm

ƒ = 3500 MHz, −40°C < T_A< 25°C

−40 dBm ≤ P_IN≤ −10 dBm⁽⁴⁾

1.2

ƒ = 50 MHz, −40 dBm ≤ P_IN≤ −10 dBm

–6.7

–7.1

–7.6

–8.5

–9.5

ƒ = 50 MHz, 25°C < T_A< 85°C

–1.1

–0.2

−40 dBm ≤ P_IN≤ −10 dBm⁽⁴⁾

ƒ = 900 MHz, −40 dBm ≤ P_IN≤ −10 dBm

ƒ = 900 MHz, 25°C < T_A< 85°C

−40 dBm ≤ P_IN≤ −10 dBm⁽⁴⁾

ƒ =1855 MHz, −40 dBm ≤ P_IN≤ −10 dBm

ƒ =1855 MHz, 25°C < T_A< 85°C

0.42

−40 dBm ≤ P_IN≤ −10 dBm⁽⁴⁾

S_T

Temperature sensitivity

mdB/°C

ƒ = 2500 MHz,−40 dBm ≤ P_IN≤ −10 dBm

ƒ = 2500 MHz, 25°C < T_A< 85°C

0.63

1

−40 dBm ≤ P_IN≤ −10 dBm⁽⁴⁾

ƒ = 3000 MHz, −40 dBm ≤ P_IN≤ −10 dBm

ƒ = 3000 MHz, 25°C < T_A< 85°C

−40 dBm ≤ P_IN≤ −10 dBm⁽⁴⁾

ƒ = 3500 MHz, −40 dBm ≤ P_IN≤ −10 dBm

ƒ = 3500 MHz, 25°C < T_A< 85°C

−40 dBm ≤ P_IN≤ −10 dBm

–21.2

2.5

(4)

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2.7-V DC and AC Electrical Characteristics (continued)

Unless otherwise specified, all limits are ensured at T_A= 25°C, V_DD= 2.7 V, RF input frequency ƒ = 1855-MHz continuous

wave (CW), modulated.⁽¹⁾

PARAMETER

TEST CONDITIONS

ƒ = 50 MHz, P_IN= −10 dBm

MIN⁽²⁾

TYP⁽³⁾

MAX⁽²⁾UNIT

–8.3

ƒ = 50 MHz, –40°C < T_A< 25°C

–15.8

–0.75

P_IN= −10 dBm⁽⁴⁾

ƒ = 900 MHz, P_IN= −10 dBm

–6

–7.4

–6.6

–4.9

–3.4

–8.9

–9.4

–10

ƒ = 900 MHz, –40°C < T_A< 25°C

–14.2

–14.9

–14.5

–13

2.2

P_IN= −10 dBm⁽⁴⁾

ƒ = 1855 MHz, P_IN= −10 dBm

ƒ = 1855 MHz, –40°C < T_A< 25°C

2

P_IN= −10 dBm⁽⁴⁾

S_T

Temperature sensitivity⁽⁴⁾

mdB/°C

ƒ = 2500 MHz, P_IN= −10 dBm

ƒ = 2500 MHz, –40°C < T_A< 25°C

1.3

3.3

P_IN= −10 dBm⁽⁴⁾

ƒ = 3000 MHz, P_IN= −10 dBm

ƒ = 3000 MHz, –40°C < T_A< 25°C

P_IN= −10 dBm⁽⁴⁾

ƒ = 3500 MHz, P_IN= −10 dBm

ƒ = 3500 MHz, –40°C < T_A< 25°C

–12

5.3

P_IN= −10 dBm⁽⁴⁾

ƒ = 50 MHz, P_IN= −10 dBm

ƒ = 50 MHz, 25°C < T_A< 85°C

–12.4

–13.7

–14.6

–15.2

–16.5

–18.1

–5.3

–5

P_IN= −10 dBm⁽⁴⁾

ƒ = 900 MHz, P_IN= −10 dBm

ƒ = 900 MHz, 25°C < T_A< 85°C

P_IN= −10 dBm⁽⁴⁾

ƒ = 1855 MHz, P_IN= −10 dBm

ƒ = 1855 MHz, 25°C < T_A< 85°C

–5.6

P_IN= −10 dBm⁽⁴⁾

S_T

Temperature sensitivity⁽⁴⁾

mdB/°C

ƒ = 2500 MHz, P_IN= −10 dBm

–10.8

–12.2

–13.5

ƒ = 2500 MHz, 25°C < T_A< 85°C

–6.5

–7.9

–9

P_IN= −10 dBm⁽⁴⁾

ƒ = 3000 MHz, P_IN= −10 dBm

ƒ = 3000 MHz, 25°C < T_A< 85°C

P_IN= −10 dBm⁽⁴⁾

ƒ = 3500 MHz, P_IN= −10 dBm

ƒ = 3500 MHz, 25°C < T_A< 85°C

P_IN= −10 dBm⁽⁴⁾

ƒ = 50 MHz

–5.9

–6.1

–5.5

–4.2

–3.7

–2.7

ƒ = 50 MHz, T_A= –40°C to 85°C

ƒ = 900 MHz

–8.85

–9.3

–8.3

–6

ƒ = 900 MHz, MIN at T_A= –40°C to 85°C

ƒ = 1855 MHz

ƒ = 1855 MHz, T_A= –40°C to 85°C

ƒ = 2500 MHz

Maximum input power

for E_LC= 1 dB⁽⁴⁾

P_MAX

dBm

ƒ = 2500 MHz, T_A= –40°C to 85°C

ƒ = 3000 MHz

ƒ = 3000 MHz, T_A= –40°C to 85°C

ƒ = 3500 MHz

–5.4

–7.2

ƒ = 3500 MHz, T_A= –40°C to 85°C

10

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2.7-V DC and AC Electrical Characteristics (continued)

Unless otherwise specified, all limits are ensured at T_A= 25°C, V_DD= 2.7 V, RF input frequency ƒ = 1855-MHz continuous

wave (CW), modulated.⁽¹⁾

PARAMETER

TEST CONDITIONS

MIN⁽²⁾

TYP⁽³⁾

MAX⁽²⁾UNIT

ƒ = 50 MHz

–40.3

ƒ = 50 MHz, T_A= –40°C to 85°C

ƒ = 900 MHz

–38.9

–44.2

–42.9

–40.4

–38.4

–35.3

34.5

ƒ = 900 MHz, MIN at T_A= –40°C to 85°C

ƒ = 1855 MHz

–42.9

ƒ = 1855 MHz, T_A= –40°C to 85°C

ƒ = 2500 MHz

–41.2

dBm

Minimum input power for

E_LC= 1 dB⁽⁴⁾

P_MIN

ƒ = 2500 MHz, T_A= –40°C to 85°C

ƒ = 3000 MHz

–38.6

–35.8

–31.9

ƒ = 3000 MHz, T_A= –40°C to 85°C

ƒ = 3500 MHz

ƒ = 3500 MHz, T_A= –40°C to 85°C

ƒ = 50 MHz

ƒ = 50 MHz, T_A= –40°C to 85°C

ƒ = 900 MHz

31.5

34.4

34

38.1

ƒ = 900 MHz, MIN at T_A= –40°C to 85°C

ƒ = 1855 MHz

37.4

ƒ = 1855 MHz, T_A= –40°C to 85°C

ƒ = 2500 MHz

Dynamic range for E_LC

1 dB⁽⁴⁾

=

DR

dB

36.1

ƒ = 2500 MHz, T_A= –40°C to 85°C

ƒ = 3000 MHz

33.8

32.4

26.2

34.8

ƒ = 3000 MHz, T_A= –40°C to 85°C

ƒ = 3500 MHz

32.7

ƒ = 3500 MHz, T_A= –40°C to 85°C

6.6 Timing Requirements

MIN

NOM

MAX UNIT

Turnon time, no signal at P_IN, low-high transition EN, V_OUTto 90%⁽¹⁾

T_A= –40°C to 85°C

Rise time⁽²⁾, P_IN= no signal to 0 dBm, V_OUTfrom 10% to 90%

T_A= –40°C to 85°C

8

2

10

t_ON

µs

12

t_R

µs

12

Fall time⁽²⁾, P_IN= no signal to 0 dBm, V_OUTfrom 90% to 10%

T_A= –40°C to 85°C

t_F

µs

12

(1) All limits are ensured by design and measurements, which are performed on a limited number of samples. Limits represent the mean

±3-sigma values. The typical value represents the statistical mean value.

(2) This parameter is ensured by design and/or characterization and is not tested in production.

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6.7 Typical Characteristics

Unless otherwise specified, V_DD= 2.7V, T_A= 25°C, measured on a limited number of samples.

10

8

10

8

85°C

25°C

-40°C

85°C

6

25°C

-40°C

4

2

0

2.2

0

650

2.5

2.8

3.1

3.4

700

750

800

(mV)

850

900

SUPPLY VOLTAGE (V)

V

ENABLE

Figure 1. Supply Current vs Supply Voltage

Figure 2. Supply Current vs Enable Voltage

2.0

45

-40°C

45

2.0

1855 MHz

1.6

25°C

85°C

40

1.6

40

900 MHz

1.2

2500 MHz

3000 MHz

3500 MHz

35

0.8

30

0.4

0.0

25

0.0

10M

100M

1G

10G

-60 -50 -40 -30 -20 -10

RF INPUT POWER (dBm)

0

10

FREQUENCY (Hz)

Figure 3. Output Voltage vs RF Input Power

Figure 4. Log Slope vs Frequency

-38

2.0

RF = - 5 dBm

IN

-40°C

1.6

1.2

0.8

0.4

0.0

RF = -15 dBm

IN

-42

RF = -25 dBm

IN

-46

RF = -35 dBm

IN

25°C

-50

85°C

RF = -45 dBm

IN

-50

-54

10M

100M

1G

10G

100M

10G

10M

1G

FREQUENCY (Hz)

Figure 6. Output Voltage vs Frequency

Figure 5. Log Intercept vs Frequency

12

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7V, T_A= 25°C, measured on a limited number of samples.

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

0.0

2.5

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

0.0

2.5

2.0

1.5

-40°C

25°C

85°C

-40°C

25°C

1.0

0.5

0.0

85°C

-0.5

-1.0

-1.5

-2.0

-2.5

-0.5

-1.0

-1.5

-2.0

-2.5

85°C

25°C

85°C

25°C

-40°C

-45

-40°C

-45

-55

-35

-25

-15

-5

5

-55

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

50 MHz

900 MHz

Figure 7. Mean Output Voltage and Log Conformance Error

vs RF Input Power

Figure 8. Mean Output Voltage and Log Conformance Error

vs RF Input Power

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

0.0

2.5

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

0.0

2.5

2.0

1.5

-40°C

25°C

1.0

-40°C

25°C

0.5

85°C

0.0

85°C

-0.5

-1.0

-1.5

-2.0

-2.5

-0.5

-1.0

-1.5

-2.0

-2.5

85°C

25°C

85°C

25°C

-35

-40°C

-45

-40°C

-45

-55

-35

-25

-15

-5

5

-55

-25

-15

-5

5

RF INPUT POWER (dBm)

1855 MHz

2500 MHz

Figure 9. Mean Output Voltage and Log Conformance Error

vs RF Input Power

Figure 10. Mean Output Voltage and Log Conformance Error

vs RF Input Power

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

0.0

2.5

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

0.0

2.5

2.0

1.5

1.0

-40°C

25°C

0.5

0.0

-0.5

-1.0

-1.5

-2.0

-2.5

-0.5

-1.0

-1.5

-2.0

-2.5

85°C

-5

85°C

25°C

85°C

25°C

-35

-40°C

-45

-40°C

-45

-55

-35

-25

-15

-5

5

-55

-25

-15

5

RF INPUT POWER (dBm)

3000 MHz

3500 MHz

Figure 11. Mean Output Voltage and Log Conformance Error

vs RF Input Power

Figure 12. Mean Output Voltage and Log Conformance Error

vs RF Input Power

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7V, T_A= 25°C, measured on a limited number of samples.

50 MHz

900 MHz

Figure 13. Log Conformance Error (Mean ±3 Sigma) vs RF

Input Power

Figure 14. Log Conformance Error (Mean ±3 Sigma) vs RF

Input Powert

1855 MHz

2500 MHz

Figure 15. Log Conformance Error (Mean ±3 Sigma) vs RF

Input Power

Figure 16. Log Conformance Error (Mean ±3 Sigma) vs RF

Input Power

3000 MHz

3500 MHz

Figure 17. Log Conformance Error (Mean ±3 Sigma) vs RF

Input Power

Figure 18. Log Conformance Error (Mean ±3 Sigma) vs RF

Input Power

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7V, T_A= 25°C, measured on a limited number of samples.

1.5

1.0

1.5

1.0

-40°C

1.0

0.5

1.0

0.5

0.0

-0.5

-1.0

-0.5

-1.0

85°C

-35

85°C

-1.5

-1.0

-1.5

-1.0

-1.5

-55

-45

-25

-15

-5

5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

50 MHz

900 MHz

Figure 19. Mean Temperature Drift Error vs Rf Input Power

Figure 20. Mean Temperature Drift Error vs RF Input Power

At 50 MHz

1.5

1.0

1.5

1.0

-40°C

1.0

0.5

0.0

-0.5

-1.0

85°C

-1.5

-1.0

85°C

-1.5

-1.0

-1.5

-55

-45

-35

-25

-15

-5

5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

2500 MHz

1855 MHz

Figure 22. Mean Temperature Drift Error vs RF Input Powert

Figure 21. Mean Temperature Drift Error vs RF Input Power

1.5

1.0

1.5

1.0

0.5

-40°C

0.5

-40°C

0.0

-0.5

-1.0

-0.5

-1.0

85°C

-35

-1.5

-1.0

-1.5

-1.0

85°C

-25

-1.5

-55

-45

-25

-15

-5

5

-55

-45

-35

-15

-5

5

RF INPUT POWER (dBm)

3000 MHz

3500 MHz

Figure 23. Mean Temperature Drift Error vs RF Input Power

Figure 24. Mean Temperature Drift Error vs RF Input Power

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7V, T_A= 25°C, measured on a limited number of samples.

50 MHz

900 MHz

Figure 25. Temperature Drift Error (Mean ±3 Sigma) vs RF

Input Power

Figure 26. Temperature Drift Error (Mean ±3 Sigma) vs RF

Input Power

1855 MHz

2500 MHz

Figure 27. Temperature Drift Error (Mean ±3 Sigma) vs RF

Input Power

Figure 28. Temperature Drift Error (Mean ±3 Sigma) vs RF

Input Power

3000 MHz

3500 MHz

Figure 29. Temperature Drift Error (Mean ±3 Sigma) vs RF

Input Power

Figure 30. Temperature Drift Error (Mean ±3 Sigma) vs RF

Input Power

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7V, T_A= 25°C, measured on a limited number of samples.

0.3

0.2

0.3

0.2

-40°C

25°C

-40°C

0.2

0.1

0.2

0.1

0.0

25°C

85°C

-0.1

-0.2

85°C

-0.1

-0.2

-0.3

-0.2

-0.3

-55

-45

-35

-25

-15

-5

5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

900 MHz

50 MHz

Figure 32. Error For 1-dB Input Power Step vs RF Input

Power

Figure 31. Error For 1-dB Input Power Step vs RF Input

Power

0.3

0.2

0.3

0.2

-40°C

0.2

0.1

85°C

0.1

0.0

25°C

-0.1

-0.2

-0.1

-0.2

85°C

-0.3

-0.2

-0.3

-55

-45

-35

-25

-15

-5

5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

2500 MHz

1855 MHz

Figure 34. Error For 1-dB Input Power Step vs RF Input

Power

Figure 33. Error For 1-dB Input Power Step vs RF Input

Power

0.3

0.2

0.3

0.2

-40°C

0.2

-40°C

25°C

0.2

0.1

0.0

25°C

-0.1

-0.2

-0.1

-0.2

85°C

-0.3

-0.2

-0.3

-0.2

-0.3

-55

-45

-35

-25

-15

-5

5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

3000 MHz

3500 MHz

Figure 35. Error For 1-dB Input Power Step vs RFInput

Power

Figure 36. Error For 1-dB Input Power Step vs RF Input

Power

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7V, T_A= 25°C, measured on a limited number of samples.

1.00

-40°C

0.75

1.00

0.75

1.00

-40°C

0.50

0.75

0.50

0.25

0.50

0.25

25°C

0.00

-0.25

-0.50

-0.25

-0. 0

-0.75

-0.50

-1.00

-0.75

-0.50

-1.00

85°C

-0.75

-1.00

-60

-50

-40

-30

-20

-10

0

-60

-50

-40

-30

-20

-10

0

RF INPUT POWER (dBm)

900 MHz

50 MHz

Figure 38. Error For 10-dB Input Power Step vs RF Input

Power

Figure 37. Error For 10-dB Input Power Step vs RF Input

Power

1.00

25°C

0.75

1.00

-40°C

0.75

1.00

0.50

0.75

0.50

0.25

0.50

0.25

25°C

-40°C

0.00

-0.25

-0. 0

-0.50

-0.75

-0.50

-1.00

-0.75

-0.50

-1.00

85°C

-0.75

-1.00

-60

-50

-40

-30

-20

-10

0

-60

-50

-40

-30

-20

-10

0

RF INPUT POWER (dBm)

1855 MHz

2500 MHz

Figure 39. Error For 10-dB Input Power Step vs RF Input

Power

Figure 40. Error For 10-dB Input Power Step vs RF Input

Power

1.00

-40°C

0.75

-40°C

0.75

1.00

0.50

0.75

0.50

0.25

0.50

0.25

25°C

0.00

-0.25

-0.50

-0.25

-0. 0

85°C

-0.75

-0.50

-0.75

-0.50

-1.00

85°C

-0.75

-1.00

-60

-50

-40

-30

-20

-10

0

-60

-50

-40

-30

-20

-10

0

RF INPUT POWER (dBm)

3000 MHz

3500 MHz

Figure 41. Error For 10-dB Input Power Step vs RF Input

Power

Figure 42. Error For 10-dB Input Power Step vs RF Input

Power

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7V, T_A= 25°C, measured on a limited number of samples.

20

15

20

10

-40°C

20

10

15

10

5

-40°C

0

-5

-30°C

-5

-10

-5

85°C

-15

-10

-20

-15

85°C

-15

-20

-55

-45

-35

-25

-15

-5

5

-55

-45

-35

-25

-5

5

RF INPUT POWER (dBm)

50 MHz

900 MHz

Figure 43. Mean Temperature Sensitivity vs RF Input Power

Figure 44. Mean Temperature Sensitivity vs RF Input Power

20

15

20

-40°C

20

10

15

-40°C

10

5

10

5

0

-5

-30°C

-5

-10

-15

-10

-30°C

85°C

-45

85°C

-15

-10

-20

-15

-20

-15

-20

-55

-45

-35

-25

-15

-5

5

-55

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

1855 MHz

2500 MHz

Figure 45. Mean Temperature Sensitivity vs RF Input Power

Figure 46. Mean Temperature Sensitivity vs RF Input Power

20

15

20

15

20

10

15

-40°C

10

5

-40°C

-30°C

10

5

0

-5

0

-5

85°C

-5

-10

-15

-10

-30°C

-15

85°C

-15

-10

-20

-15

-20

-15

-20

-55

-45

-35

-25

-5

5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

3500 MHz

3000 MHz

Figure 48. Mean Temperature Sensitivity vs RF Input Power

Figure 47. Mean Temperature Sensitivity vs RF Input Power

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7V, T_A= 25°C, measured on a limited number of samples.

50 MHz

900 MHz

Figure 49. Temperature Sensitivity (Mean ±3 Sigma) vs RF

Input Power

Figure 50. Temperature Sensitivity (Mean ±3 Sigma) vs RF

Input Power

1855 MHz

2500 MHz

Figure 51. Temperature Sensitivity (Mean ±3 Sigma) vs RF

Input Power

Figure 52. Temperature Sensitivity (Mean ±3 Sigma) vs RF

Input Power

3000 MHz

3500 MHz

Figure 53. Temperature Sensitivity (Mean ±3 Sigma) vs RF

Input Power

Figure 54. Temperature Sensitivity (Mean ±3 Sigma) vs RF

Input Power

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7V, T_A= 25°C, measured on a limited number of samples.

2.0

1.8

1.6

2.5

2.0

1.5

2.5

2.0

1.5

2.0

1.8

1.6

CW

IS-95

WCDMA 64 CH

CW

IS-95

WCDMA 64 CH

2.0

2.5

2.0

2.5

1.8

2.0

1.4

1.0

1.8

2.0

1.5

1.0

1.4

1.6

1.4

1.5

1.0

1.6

1.4

1.0

1.2

0.5

1.2

1.0

0.5

0.0

CW

1.2

1.0

0.5

0.0

CW

1.0

0.0

1.0

0.8

0.6

-0.5

-1.0

0.8

0.6

-0.5

-1.0

-0.5

0.8

-0.5

0.8

0.4

0.2

-1.5

-2.0

0.4

0.2

-1.5

-2.0

-1.0

0.6

-1.0

0.6

0.0

-2.5

0.0

-2.5

0.4

-1.5

0.4

IS-95

-15

IS-95

WCDMA 64 ch

-35 -25

RF INPUT POWER (dBm)

WCDMA 64 ch

0.2

0.0

-2.0

-2.5

-2.0

-2.5

0.2

0.0

-55

-45

-35

-25

-5

5

-55

-45

-15

-5

5

RF INPUT POWER (dBm)

900 MHz

1855 MHz

Figure 55. Output Voltage and Log Conformance Error vs

RR Input Power for Various Modulation Types

Figure 56. Output Voltage and Log Conformance Error vs

RF Input Power for Various Modulation Types

100

10

9

8

7

6

5

4

3

2

1

0

75

100

R

50

75

50

25

0

-25

-50

-75

X

-50

-100

-75

-100

10M

100M

1G

10G

10

100

1k

10k

100k

1M

FREQUENCY (Hz)

Figure 57. RF Input Impedance vs Frequency (Resistance

and Reactance)

Figure 58. Output Noise Spectrum vs Frequency

270

80

100k

GAIN

225

70

80

60

70

180

10k

60

50

135

50

40

30

PHASE

90

30

45

20

10

1k

20

0

10

0

-45

100

-90

10M

1k

10k

100k

1M

10

100

1k

10k

100k

1M

FREQUENCY (Hz)

Figure 60. Output Amplifier Gain and Phase vs Frequency

Figure 59. Power Supply Rejection Ratio vs Frequency

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Typical Characteristics (continued)

Unless otherwise specified, V_DD= 2.7V, T_A= 25°C, measured on a limited number of samples.

60

50

40

30

20

10

0

60

50

40

30

20

10

0

85°C

25°C

-40°C

0.0

0.5

1.0

1.5

2.0

(V)

2.5

3.0

0.0

0.5

1.0

1.5

2.0

(V)

2.5

3.0

V

OUT

Figure 61. Sourcing Output Current vs Output Voltage

Figure 62. Sinking Output Current vs Output Voltage

2.70

0.08

-40°C

2.68

2.70

-40°C

25°C

85°C

0.06

25°C

85°C

2.66

2.68

0.04

2.66

2.64

0.02

2.62

2.60

0.00

0

1

2

3

4

5

0

1

2

3

4

5

SOURCING CURRENT (mA)

SINKING CURRENT (mA)

Figure 63. Output Voltage vs Sourcing Current

Figure 64. Output Voltage vs Sinking Current

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7 Detailed Description

7.1 Overview

The LMV221 is a versatile logarithmic RF power detector suitable for use in power measurement systems. The

LMV221 is particularly well suited for CDMA and UMTS applications. It produces a DC voltage that is a measure

for the applied RF power.

The core of the LMV221 is a progressive compression LOG detector consisting of four gain stages. Each of

these saturating stages has a gain of approximately 10 dB and therefore achieves about 10 dB of the detector

dynamic range. The five diode cells perform the actual detection and convert the RF signal to a DC current. This

DC current is subsequently supplied to the transimpedance amplifier at the output, which converts it into an

output voltage. In addition, the amplifier provides buffering of and applies filtering to the detector output signal.

To prevent discharge of filtering capacitors between OUT and GND in shutdown, a switch is inserted at the

amplifier input that opens in shutdown to realize a high impedance output of the device.

7.2 Functional Block Diagram

REF

B2

VDD

A1

R_TRANS

en

EN C2

en

I / I

-

A2 OUT

+

V_REF

+

en

-

RFIN

B1

10 dB

V-V

10 dB

R_IN

GND C1

7.3 Feature Description

7.3.1 Characteristics of the LMV221

The LMV221 is a logarithmic RF power detector with approximately 40-dB dynamic range. This dynamic range

plus its logarithmic behavior make the LMV221 ideal for various applications such as wireless transmit power

control for CDMA and UMTS applications. The frequency range of the LMV221 is from 50 MHz to 3.5 GHz,

which makes it suitable for various applications.

The LMV221 transfer function is accurately temperature compensated. This makes the measurement accurate

for a wide temperature range. Furthermore, the LMV221 can easily be connected to a directional coupler

because of its 50-Ω input termination. The output range is adjustable to fit the ADC input range. The detector can

be switched into a power saving shutdown mode for use in pulsed conditions.

7.3.2 Accurate Power Measurement

The power measurement accuracy achieved with a power detector is not only determined by the accuracy of the

detector itself, but also by the way it is integrated into the application. In many applications some form of

calibration is employed to improve the accuracy of the overall system beyond the intrinsic accuracy provided by

the power detector. For example, for LOG-detectors calibration can be used to eliminate part to part spread of

the LOG-slope and LOG-intercept from the overall power measurement system, thereby improving its power

measurement accuracy.

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Feature Description (continued)

Calibration techniques can be used to improve the accuracy of a power measurement system beyond the

intrinsic accuracy of the power detector itself. LOG-Conformance Error and Temperature Drift Error discuss

power measurement systems using LOG-detectors, specifically the LMV221, but the more generic concepts can

also be applied to other power detectors. Other factors influencing the power measurement accuracy, such as

the resolution of the ADC reading the detector output signal, are not considered here because these factors are

not fundamentally due to the power detector.

7.3.2.1 Concept of Power Measurements

Power measurement systems generally consists of two clearly distinguishable parts with different functions:

1. A power detector device, that generates a DC output signal (voltage) in response to the power level of the

(RF) signal applied to its input.

2. An estimator that converts the measured detector output signal into a (digital) numeric value representing the

power level of the signal at the detector input.

This conceptual configuration is shown in Figure 65.

F_EST

MODEL

P

IN

V

OUT

P

EST

F_DET

PARAMETERS

Figure 65. Generic Concept of a Power Measurement System

The core of the estimator is usually implemented as a software algorithm, receiving a digitized version of the

detector output voltage. Its transfer F_ESTfrom detector output voltage to a numerical output must be equal to the

inverse of the detector transfer F_DETfrom (RF) input power to DC output voltage. If the power measurement

system is ideal, that is, if no errors are introduced into the measurement result by the detector or the estimator,

the measured power P_EST(the output of the estimator) and the actual input power P_INmust be identical. In that

case, the measurement error E, the difference between the two, should be identically zero:

E =

P_EST- P_INô 0

∫ P_EST= F_EST[F_DET(P_IN)] = P_IN

-1

∫ F_EST(V_OUT) = F (V_OUT

)

DET

(1)

From Equation 1 it follows that one would design the F_ESTtransfer function to be the inverse of the F_DETtransfer

function.

In practice the power measurement error is not zero, due to the following effects:

•

The detector transfer function is subject to various kinds of random errors that result in uncertainty in the

detector output voltage; the detector transfer function is not exactly known.

•

The detector transfer function might be too complicated to be implemented in a practical estimator.

The function of the estimator is then to estimate the input power P_IN, that is, to produce an output P_ESTsuch that

the power measurement error is, on average, minimized, based on the following information:

1. Measurement of the not-completely-accurate detector output voltage V_OUT

2. Knowledge about the detector transfer function F_DET, for example the shape of the transfer function, the

types of errors present (part-to-part spread, temperature drift), and so forth.

Obviously the total measurement accuracy can be optimized by minimizing the uncertainty in the detector output

signal (select an accurate power detector), and by incorporating as much accurate information about the detector

transfer function into the estimator as possible.

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Feature Description (continued)

The knowledge about the detector transfer function is condensed into a mathematical model for the detector

transfer function, consisting of:

•

A formula for the detector transfer function; and

Values for the parameters in this formula.

The values for the parameters in the model can be obtained in various ways. They can be based on

measurements of the detector transfer function in a precisely controlled environment (parameter extraction). If

the parameter values are separately determined for each individual device, errors like part-to-part spread are

eliminated from the measurement system.

Errors may occur when the operating conditions of the detector (for example, the temperature) become

significantly different from the operating conditions during calibration (for example, room temperature). Examples

of simple estimators for power measurements that result in a number of commonly used metrics for the power

measurement error are discussed in LOG-Conformance Error, Temperature Drift Error, Temperature

Compensation and Temperature Drift Error.

7.3.2.2 LOG-Conformance Error

Probably the simplest power measurement system that can be realized is obtained when the LOG-detector

transfer function is modeled as a perfect linear-in-dB relationship between the input power and output voltage:

V_OUT,MOD

=

F_DET,MOD(P_IN) = K_SLOPE(P_INœ P_INTERCEPT

)

where

•

K_SLOPErepresents the LOG-slope and P_INTERCEPTthe LOG-intercept

(2)

(3)

The estimator based on Equation 2 implements the inverse of the model equation, that is:

V_OUT

P_EST= F_EST(V_OUT) =

+ P_INTERCEPT

K_SLOPE

The resulting power measurement error, the LOG-conformance error, is thus equal to:

V_OUT

K_SLOPE

E_LCE= P_EST- P_IN

=

- (P_IN- P_INTERCEPT)

V_OUT- V_OUT,MOD

=

K_SLOPE

(4)

The most important contributions to the LOG-conformance error are generally:

•

The deviation of the actual detector transfer function from an ideal logarithm (the transfer function is nonlinear

in dB).

Drift of the detector transfer function over various environmental conditions, most importantly temperature;

K_SLOPEand P_INTERCEPTare usually determined for room temperature only.

Part-to-part spread of the (room temperature) transfer function.

The latter component is conveniently removed by means of calibration, that is, if the LOG slope and LOG-

intercept are determined for each individual detector device (at room temperature). This can be achieved by

measurement of the detector output voltage (at room temperature) for a series of different power levels in the

LOG-linear range of the detector transfer function. The slope and intercept can then be determined by means of

linear regression.

An example of this type of error and its relationship to the detector transfer function is shown in Figure 66.

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Feature Description (continued)

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

0.0

2.5

2.0

1.5

-40°C

25°C

1.0

0.5

0.0

85°C

-0.5

-1.0

-1.5

-2.0

-2.5

85°C

25°C

-40°C

-45

-55

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 66. LOG-Conformance Error and LOG-Detector Transfer Function

In the center of the dynamic range of the detector, the LOG-conformance error is small, especially at room

temperature; in this region the transfer function closely follows the linear-in-dB relationship while K_SLOPEand

P_INTERCEPTare determined based on room temperature measurements. At the temperature extremes the error in

the center of the range is slightly larger due to the temperature drift of the detector transfer function. The error

rapidly increases toward the top and bottom end of the detector's dynamic range; here the detector saturates and

its transfer function starts to deviate significantly from the ideal LOG-linear model. The detector dynamic range is

usually defined as the power range for which the LOG conformance error is smaller than a specified amount.

Often an error of ±1 dB is used as a criterion.

7.3.2.3 Temperature Drift Error

A more accurate power measurement system can be obtained if the first error contribution, due to the deviation

from the ideal LOG-linear model, is eliminated. This is achieved if the actual measured detector transfer function

at room temperature is used as a model for the detector, instead of the ideal LOG-linear transfer function used in

the previous section.

The formula used for such a detector is:

V_OUT,MOD= F_DET(P_IN,T_O)

where

•

T_Orepresents the temperature during calibration (room temperature).

(5)

The transfer function of the corresponding estimator is thus the inverse of this:

-1

P_EST= F [V_OUT(T),T₀]

DET

where

•

V_OUT(T) represents the measured detector output voltage at the operating temperature T.

(6)

The resulting measurement error is only due to drift of the detector transfer function over temperature and can be

expressed as in Equation 7:

-1

DET

E_DRIFT(T,T₀) =

P_EST- P_IN= F [V_OUT(T),T₀] - P_IN

= F^-1[V_OUT(T),T₀] - F_D^-_E¹_T[V_OUT(T),T)]

DET

(7)

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Feature Description (continued)

Unfortunately, the (numeric) inverse of the detector transfer function at different temperatures makes this

expression rather impractical. However, because the drift error is usually small V_OUT(T) is only slightly different

from V_OUT(T_O). This means that Equation 8 can be applied:

E_DRIFT(T₀,T₀)

E_DRIFT(T,T₀) ö

ï

ïT

-1

DET

-1

DET

+ (T - T₀) {F [V_OUT(T),T₀] - F [V_OUT(T),T]}

(8)

This expression is easily simplified by taking the following considerations into account:

•

The drift error at the calibration temperature E(T_O,T_O) equals zero (by definition).

The estimator transfer F_DET(V_OUT,T_O) is not a function of temperature; the estimator output changes over

temperature only due to the temperature dependence of V_OUT

.

•

The actual detector input power P_INis not temperature dependent (in the context of this expression).

The derivative of the estimator transfer function to V_OUTequals approximately 1/K_SLOPEin the LOG-linear

region of the detector transfer function (the region of interest).

Taking into account the preceding considerations, the simplified expression would be:

-1

DET

ï

ïT

(T œ T )

F

[V_OUT(T),T₀]

E_DRIFT(T,T₀) ö

0

-1

DET

ï V

ïT

(T)

OUT

ï

= (T œ T₀)

F

[V_OUT(T),T₀]

ïV_OUT

V_OUT(T) œ V_OUT(T₀)

ö

K_SLOPE

(9)

Equation 9 is very similar to Equation 4 determined previously. The only difference is that instead of the output of

the ideal LOG-linear model, the actual detector output voltage at the calibration temperature is now subtracted

from the detector output voltage at the operating temperature.

Figure 67 depicts an example of the drift error.

1.5

1.0

-40°C

1.0

0.5

0.0

-0.5

-1.0

85°C

-1.5

-1.0

-1.5

-55

-45

-35

-25

-15

-5

5

RF INPUT POWER (dBm)

Figure 67. Temperature Drift Error of the LMV221 at ƒ = 1855 MHz

In agreement with the definition, the temperature drift error is zero at the calibration temperature. Further, the

main difference with the LOG-conformance error is observed at the top and bottom end of the detection range;

instead of a rapid increase the drift error settles to a small value at high and low input power levels due to the

fact that the detector saturation levels are relatively temperature independent.

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Feature Description (continued)

In a practical application it may not be possible to use the exact inverse detector transfer function as the

algorithm for the estimator. For example, it may require too much memory or too much factory calibration time.

However, using the ideal LOG-linear model in combination with a few extra data points at the top and bottom end

of the detection range — where the deviation is largest — can already significantly reduce the power

measurement error.

7.3.2.3.1 Temperature Compensation

A further reduction of the power measurement error is possible if the operating temperature is measured in the

application. For this purpose, the detector model used by the estimator should be extended to cover the

temperature dependency of the detector.

Because the detector transfer function is generally a smooth function of temperature (the output voltage changes

gradually over temperature), the temperature is in most cases adequately modeled by a first-order or second-

order polynomial (see Equation 10).

V_OUT,MOD= F_DET(P_IN,T₀)[1 + (T-T₀)TC₁(P_IN)

+ (T-T₀)²TC₂(P_IN) + O(T³)]

(10)

The required temperature dependence of the estimator, to compensate for the detector temperature dependence

can be approximated similarly:

P_EST= F_D^-_E¹_T[V_OUT(T),T₀]{1 + (T-T₀)S₁[V_OUT(T)] +

+ (T-T₀)²S₂[V_OUT(T)] + O(T³)}

ö F_DE^-1_T[V_OUT(T),T₀]{1 + (T-T₀)S₁[V_OUT(T)]}

(11)

The last approximation results from the fact that a first-order temperature compensation is usually sufficiently

accurate. For second and higher-order compensation a similar approach can be followed.

Ideally, the temperature drift could be completely eliminated if the measurement system is calibrated at various

temperatures and input power levels to determine the temperature sensitivity S₁. In a practical application,

however, that is usually not possible due to the associated high costs. The alternative is to use the average

temperature drift in the estimator, instead of the temperature sensitivity of each device individually. In this way it

is possible to eliminate the systematic (reproducible) component of the temperature drift without the need for

calibration at different temperatures during manufacturing. What remains is the random temperature drift, which

differs from device to device. (see Figure 68). The graph at the left of Figure 68 schematically represents the

behavior of the drift error versus temperature at a certain input power level for a large number of devices.

Figure 68. Elimination of the Systematic Component from the Temperature Drift

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Feature Description (continued)

The mean drift error represents the reproducible (systematic) part of the error, while the mean ±3 sigma limits

represent the combined systematic plus random error component. Obviously the drift error must be zero at

calibration temperature T₀. If the systematic component of the drift error is included in the estimator, the total drift

error becomes equal to only the random component, as shown in the graph at the right of Figure 68. A significant

reduction of the temperature drift error can be achieved in this way only if:

•

The systematic component is significantly larger than the random error component (otherwise the difference

is negligible).

•

The operating temperature is measured with sufficient accuracy.

It is essential for the effectiveness of the temperature compensation to assign the appropriate value to the

temperature sensitivity S₁. Two different methods can be followed to determine this parameter:

1. Determination of a single value to be used over the entire operating temperature range.

2. Division of the operating temperature range in segments and use of separate values for each of the

segments.

For the first method, the accuracy of the extracted temperature sensitivity increases when the number of

measurement temperatures increases. Linear regression to temperature can then be used to determine the two

parameters of the linear model for the temperature drift error: the first order temperature sensitivity S₁and the

best-fit (room temperature) value for the power estimate at T₀: F_DET[V_OUT(T),T₀]. Note that to achieve an overall

(over all temperatures) minimum error, the room temperature drift error in the model can be non-zero at the

calibration temperature (which is not in agreement with the strict definition).

The second method does not have this drawback but is more complex. In fact, segmentation of the temperature

range is a form of higher-order temperature compensation using only a first-order model for the different

segments: one for temperatures below 25°C, and one for temperatures above 25°C. The mean (or typical)

temperature sensitivity is the value to be used for compensation of the systematic drift error component.

Figure 69 and Figure 70 show the temperature drift error without and with temperature compensation using two

segments. With compensation the systematic component is completely eliminated; the remaining random error

component is centered around zero. Note that the random component is slightly larger at −40°C than at +85°C.

Figure 69. Temperature Drift Error without Temperature

Figure 70. Temperature Drift Error With Temperature

Compensation

In a practical power measurement system, temperature compensation is usually only applied to a small power

range around the maximum power level for two reasons:

1. The various communication standards require the highest accuracy in this range to limit interference.

2. The temperature sensitivity itself is a function of the power level it becomes impractical to store a large

number of different temperature sensitivity values for different power levels.

The 2.7-V DC and AC Electrical Characteristics specifies the temperature sensitivity for the aforementioned two

segments at an input power level of −10 dBm (near the top-end of the detector dynamic range). The typical value

represents the mean which is to be used for calibration.

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Feature Description (continued)

7.3.2.3.2 Differential Power Errors

Many third-generation communication systems contain a power control loop through the base station and mobile

unit that require frequent updates to the transmit-power level by a small amount (typically 1 dB). For such

applications it is important that the actual change of the transmit power is sufficiently close to the requested

power change.

The error metrics in this data sheet that describe the accuracy of the detector for a change in the input power are

E_{1 dB}(for a 1-dB change in the input power) and E_{10 dB}(for a 10-dB step, or ten consecutive steps of 1 dB).

Because it can be assumed that the temperature does not change during the power step the differential error

equals the difference of the drift error at the two involved power levels:

E_1dB(P ,T)=

IN

E_DRIFT(P_IN+1dB,T) - E_DRIFT(P_IN,T)

E_DRIFT(P_IN+10dB,T) - E_DRIFT(P_IN,T)

E_10dB(P ,T)=

IN

(12)

NOTE

The step error increases significantly when one (or both) power levels in the above

expression are outside the detector dynamic range. For E_{10 dB}this occurs when P_INis less

than 10 dB below the maximum input power of the dynamic range, P_MAX

.

7.4 Device Functional Modes

7.4.1 Shutdown

To save power, the LMV221 can be brought into a low power-shutdown mode. The device is active for EN = high

(V_EN> 1.1 V) and in the low power-shutdown mode for EN = low (V_EN< 0.6 V). In this state the output of the

LMV221 is switched to a high impedance mode. Using the shutdown function, care must be taken not to exceed

the absolute maximum ratings. Forcing a voltage to the enable input that is 400 mV higher than V_DDor 400 mV

lower than GND damages the device, and further operation is not ensured. The absolute maximum ratings can

also be exceeded when the enable EN is switched to high (from shutdown to active mode) while the supply

voltage is low (off). This must be prevented at all times. A possible solution to protect the device is to add a

resistor of 100 kΩ in series with the enable input.

7.4.1.1 Output Behavior in Shutdown

In order to save power, the LMV221 can be used in pulsed mode so that it is active to perform the power

measurement only during a fraction of the time. During the remaining time the device is in low-power shutdown.

Applications using this approach usually require that the output value is available at all times, including when the

LMV221 is in shutdown. The settling time in active mode, however, must not become excessively large. This can

be achieved by the combination of the LMV221 and a low pass output filter (see Figure 75).

In active mode, the filter capacitor C_Sis charged to the output voltage of the LMV221, which in this mode has a

low output impedance to enable fast settling. During shutdown mode, the capacitor should preserve this voltage.

Discharge of C_Sthrough any current path must therefore be avoided in shutdown. The output impedance of the

LMV221 becomes high in shutdown, thus, the discharge current cannot flow from the capacitor top plate, through

R_Sand the device OUT pin to GND. This is detected by the internal shutdown mechanism of the output amplifier

and by the switch depicted in Figure 79. Additionally, the ADC input impedance must be high to prevent a

possible discharge path through the ADC.

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8 Application and Implementation

NOTE

Information in the following applications sections is not part of the TI component

specification, and TI does not warrant its accuracy or completeness. TI’s customers are

responsible for determining suitability of components for their purposes. Customers should

validate and test their design implementation to confirm system functionality.

8.1 Application Information

8.1.1 Functionality and Applications of RF Power Detectors

8.1.1.1 Functionality of RF Power Detectors

An RF power detector is a device that produces a DC output voltage in response to the RF power level of the

signal applied to its input. A wide variety of power detectors can be distinguished, each having certain properties

that suit a particular application. This section provides an overview of the key characteristics of power detectors,

and discusses the most important types of power detectors. The functional behavior of the LMV221 is discussed

in detail.

8.1.1.1.1 Key Characteristics of RF Power Detectors

Power detectors are used to accurately measure the power of a signal inside the application. The attainable

accuracy of the measurement is therefore dependent upon the accuracy and predictability of the detector transfer

function from the RF input power to the DC output voltage.

Certain key characteristics determine the accuracy of RF detectors and they are classified accordingly:

•

Temperature Stability

Dynamic Range

Waveform Dependency

Transfer Shape

Generally, the transfer function of RF power detectors is slightly temperature dependent. This temperature drift

reduces the accuracy of the power measurement, because most applications are calibrated at room temperature.

In such systems, the temperature drift significantly contributes to the overall system power measurement error.

The temperature stability of the transfer function differs for the various types of power detectors. Generally,

power detectors that contain only one or few semiconductor devices (diodes, transistors) operating at RF

frequencies attain the best temperature stability.

The dynamic range of a power detector is the input power range for which it creates an accurately reproducible

output signal. What is considered accurate is determined by the applied criterion for the detector accuracy; the

detector dynamic range is thus always associated with certain power measurement accuracy. This accuracy is

usually expressed as the deviation of its transfer function from a certain predefined relationship, such as linear in

dB for LOG detectors and square-law transfer (from input RF voltage to DC output voltage) for mean-square

detectors. For LOG-detectors, the dynamic range is often specified as the power range for which its transfer

function follows the ideal linear-in-dB relationship with an error smaller than or equal to ±1 dB. Again, the

attainable dynamic range differs considerably for the various types of power detectors.

According to its definition, the average power is a metric for the average energy content of a signal and is not

directly a function of the shape of the signal in time. In other words, the power contained in a 0-dBm sine wave is

identical to the power contained in a 0-dBm square wave or a 0-dBm WCDMA signal; all these signals have the

same average power. Depending on the internal detection mechanism, though, power detectors may produce a

slightly different output signal in response to the aforementioned waveforms, even though their average power

level is the same. This is due to the fact that not all power detectors strictly implement the definition formula for

signal power, being the mean of the square of the signal. Most types of detectors perform some mixture of peak

detection and average power detection. A waveform independent detector response is often desired in

applications that exhibit a large variety of waveforms, such that separate calibration for each waveform becomes

impractical.

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Application Information (continued)

The shape of the detector transfer function from the RF input power to the DC output voltage determines the

required resolution of the ADC connected to it. The overall power measurement error is the combination of the

error introduced by the detector, and the quantization error contributed by the ADC. The impact of the

quantization error on the overall transfer's accuracy is highly dependent on the detector transfer shape, as shown

in Figure 71 and Figure 72.

2

ÂV

ÂV₁

ÂV₂

0

-60

0

-60

0

ÂP

RF INPUT POWER (dBm)

Figure 71. Convex Detector Transfer Function

Figure 72. Linear Transfer Function

Figure 71 and Figure 72 shows two different representations of the detector transfer function. In both Figure 71

and Figure 72 the input power along the horizontal axis is displayed in dBm because most applications specify

power accuracy requirements in dBm (or dB). Figure 71 shows a convex detector transfer function, while the

transfer function on the right hand side is linear (in dB). The slope of the detector transfer function — the detector

conversion gain – is of key importance for the impact of the quantization error on the total measurement error. If

the detector transfer function slope is low, a change, ΔP, in the input power results only in a small change of the

detector output voltage, such that the quantization error is relatively large. On the other hand, if the detector

transfer function slope is high, the output voltage change for the same input power change will be large, such

that the quantization error is small. Figure 72 has a very low slope at low input power levels, resulting in a

relatively large quantization error. Therefore, to achieve accurate power measurement in this region, a high-

resolution ADC is required. On the other hand, for high input power levels the quantization error are very small

due to the steep slope of the curve in this region. For accurate power measurement in this region, a much lower

ADC resolution is sufficient. Figure 71 has a constant slope over the power range of interest, such that the

required ADC resolution for a certain measurement accuracy is constant. For this reason, the LOG-linear curve

in Figure 71 generally leads to the lowest ADC resolution requirements for certain power measurement accuracy.

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Application Information (continued)

8.1.1.1.2 Types of RF Power Detectors

Three different detector types are distinguished based on the four characteristics previously discussed:

•

Diode Detector

(Root) Mean Square (R)MS) Detector

Logarithmic Detectors

8.1.1.1.2.1 Diode Detector

A diode is one of the simplest types of RF detectors. As depicted in Figure 73, the diode converts the RF input

voltage into a rectified current. This unidirectional current charges the capacitor. The RC time constant of the

resistor and the capacitor determines the amount of filtering applied to the rectified (detected) signal.

D

Z

0

VREF

R

S

C

S

V

OUT

Figure 73. Diode Detector

The advantages and disadvantages can be summarized as follows:

•

The temperature stability of the diode detectors is generally very good because they contain only one

semiconductor device that operates at RF frequencies.

•

The dynamic range of diode detectors is poor. The conversion gain from the RF input power to the output

voltage quickly drops to very low levels when the input power decreases. Typically a dynamic range of 20 dB

to 25 dB can be achieved with this type of detector.

•

The response of diode detectors is waveform dependent. As a consequence of this dependency, for example,

its output voltage for a 0-dBm WCDMA signal is different than for a 0-dBm unmodulated carrier. This is due to

the fact that the diode measures peak power instead of average power. The relation between peak power and

average power is dependent on the wave shape.

The transfer shape of diode detectors puts high requirements on the resolution of the ADC that reads their

output voltage. Especially at low input power levels a very high ADC resolution is required to achieve

sufficient power measurement accuracy (See Figure 71).

8.1.1.1.2.2 (Root) Mean Square (R)MS) Detector

This type of detector is particularly suited for the power measurements of RF modulated signals that exhibits

large peak-to-average power ratio variations. This is because its operation is based on direct determination of the

average power and not – like the diode detector – of the peak power.

The advantages and disadvantages can be summarized as follows:

•

The temperature stability of (R)MS detectors is almost as good as the temperature stability of the diode

detector; only a small part of the circuit operates at RF frequencies, while the rest of the circuit operates at

low frequencies.

•

The dynamic range of (R)MS detectors is limited. The lower end of the dynamic range is limited by internal

device offsets.

The response of (R)MS detectors is highly waveform independent. This is a key advantage compared to other

types of detectors in applications that employ signals with high peak-to-average power variations. For

example, the (R)MS detector response to a 0-dBm WCDMA signal and a 0-dBm unmodulated carrier is

essentially equal.

•

The transfer shape of R(MS) detectors has many similarities with the diode detector and is therefore subject

to similar disadvantages with respect to the ADC resolution requirements (see Figure 72).

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Application Information (continued)

8.1.1.1.2.3 Logarithmic Detectors

The transfer function of a logarithmic detector has a linear in dB response, which means that the output voltage

changes linearly with the RF power in dBm. This is convenient because most communication standards specify

transmit power levels in dBm as well.

The advantages and disadvantages can be summarized as follows:

•

The temperature stability of the LOG detector transfer function is generally not as good as the stability of

diode and R(MS) detectors. This is because a significant part of the circuit operates at RF frequencies.

•

The dynamic range of LOG detectors is usually much larger than that of other types of detectors.

Because LOG detectors perform a kind of peak detection their response is wave form dependent, similar to

diode detectors.

•

The transfer shape of LOG detectors puts the lowest possible requirements on the ADC resolution (See

Figure 72).

8.2 Typical Applications

RF power detectors can be used in a wide variety of applications. Figure 74 shows the LMV221 in a transmit

power-control system, and Figure 82 measures the voltage standing wave ratio (VSWR).

8.2.1 Application With Transmit Power Control Loop

The key benefit of a transmit power control loop circuit is that it makes the transmit power insensitive to changes

in the power amplifier (PA) gain control function, such as changes due to temperature drift. When a control loop

is used, the transfer function of the PA is eliminated from the overall transfer function. Instead, the overall

transfer function is determined by the power detector. The overall transfer function accuracy depends thus on the

RF detector accuracy. The LMV221 is especially suited for this application, due to the accurate temperature

stability of its transfer function.

Figure 74 shows a block diagram of a typical transmit power control system. The output power of the PA is

measured by the LMV221 through a directional coupler. The measured output voltage of the LMV221 is filtered

and subsequently digitized by the ADC inside the baseband chip. The baseband adjusts the PA output power

level by changing the gain control signal of the RF VGA accordingly. With an input impedance of 50 Ω, the

LMV221 can be directly connected to a 30-dB directional coupler without the need for an additional external

attenuator. The setup can be adjusted to various PA output ranges by selection of a directional coupler with the

appropriate coupling factor.

COUPLER

B

A

S

E

B

A

N

D

RF

VGA

PA

ANTENNA

50 W

GAIN

R_S

RFIN

ADC

OUT

C_S

LMV221

EN

LOGIC

GND

Figure 74. Transmit Power Control System

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Typical Applications (continued)

8.2.1.1 Design Requirements

Some of the design requirements for this logarithmic RMS power detector include:

Table 1. Design Parameters

DESIGN PARAMETER

Supply voltage

EXAMPLE VALUE

2.7 V

RF input frequency (unmodulated continuous wave)

Minimum input power for E_LC= 1 dB

Maximum input power for E_LC= 1 dB

Maximum output voltage, P_IN= –5 dBm

1855 MHz

–42.9 dBm

–5.5 dBm

1.61 V

8.2.1.2 Detailed Design Procedure

8.2.1.2.1 Detector Interfacing

For optimal performance of the LMV221 device, it is important that all its pins are connected to the surrounding

circuitry in the appropriate way. Starting from the Functional Block Diagram the function of each pin is elaborated

in the following sections. The details of the electrical interfacing are separately discussed for each pin. Output

filtering options and the differences between single ended and differential interfacing with an ADC are also

discussed in detail in the following subsections.

8.2.1.2.1.1 RF Input

RF parts typically use a characteristic impedance of 50 Ω. To comply with this standard the LMV221 has an input

impedance of 50 Ω. Using a characteristic impedance other then 50 Ω causes a shift of the logarithmic intercept

with respect to the value given in the 2.7-V DC and AC Electrical Characteristics. This intercept shift can be

calculated according to Equation 13.

2 R_SOURCE

R_SOURCE+ 50

≈

«

’

◊

P_INT-SHIFT= 10 LOG

(13)

The intercept shifts to higher power levels for R_SOURCE> 50 Ω, and shifts to lower power levels for R_SOURCE

50 Ω.

<

8.2.1.2.1.2 Output and Reference

The possible filtering techniques that can be applied to reduce ripple in the detector output voltage are discussed

in Filtering. In addition, two different topologies to connect the LMV221 to an ADC are elaborated.

8.2.1.2.1.2.1 Filtering

The output voltage of the LMV221 is a measure for the applied RF signal on the RF input pin. Usually, the

applied RF signal contains AM modulation that causes low frequency ripple in the detector output voltage. CDMA

signals, for instance, contain a large amount of amplitude variations. Filtering of the output signal can be used to

eliminate this ripple. The filtering can either be achieved by a low pass output filter or a low pass feedback filter.

Those two techniques are shown in Figure 75 and Figure 76.

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VDD

1

R

S

R

S

OUT

REF

RFIN

OUT

REF

RFIN

1

LMV221

3

6

5

+

2

6

5

+

2

ADC

R_P

C

S

C

S

LMV221

EN

4

-

4

-

3

GND

Figure 75. Low Pass Output Filter

Figure 76. Low Pass Feedback Filter

Depending on the system requirements one of the these filtering techniques can be selected. The low pass

output filter has the advantage that it preserves the output voltage when the LMV221 is brought into shutdown.

This is elaborated in Output Behavior in Shutdown. In the feedback filter, resistor R_Pdischarges capacitor C_Pin

shutdown and therefore changes the output voltage of the device.

A disadvantage of the low pass output filter is that the series resistor R_Slimits the output drive capability. This

may cause inaccuracies in the voltage read by an ADC when the ADC input impedance is not significantly larger

than R_S. In that case, the current flowing through the ADC input induces an error voltage across filter resistor R_S.

The low pass feedback filter does not have this disadvantage.

NOTE

Note that adding an external resistor between OUT and REF reduces the transfer gain

(LOG-slope and LOG-intercept) of the device. The internal feedback resistor sets the gain

of the transimpedance amplifier.

The filtering of the low pass output filter is achieved by resistor R_Sand capacitor C_S. The −3 dB bandwidth of this

filter can then be calculated by: ƒ_{−3 dB}= 1 / 2πR_SC_S. The bandwidth of the low pass feedback filter is determined

by external resistor R_Pin parallel with the internal resistor R_TRANS, and external capacitor C_Pin parallel with

internal capacitor C_TRANS(see Figure 79). The −3 dB bandwidth of the feedback filter can be calculated by ƒ_{−3 dB}

= 1 / 2π (R_P//R_TRANS) (C_P+ C_TRANS). The bandwidth set by the internal resistor and capacitor (when no external

components are connected between OUT and REF) equals ƒ_{−3 dB}= 1 / 2π R_TRANSC_TRANS= 450 kHz.

8.2.1.2.1.3 Interface to the ADC

The LMV221 can be connected to the ADC with a single-ended or a differential topology. The single ended

topology connects the output of the LMV221 to the input of the ADC and the reference pin is not connected. In a

differential topology, both the output and the reference pins of the LMV221 are connected to the ADC. The

topologies are depicted in Figure 77 and Figure 78.

VDD

1

VDD

1

R

S

OUT

REF

RFIN

2

4

6

+

6

5

+

2

ADC

R

P

C

P

R_P

C_P

LMV221

EN

REF

EN

5

-

4

-

3

GND

Figure 77. Single-Ended Application

Figure 78. Differential Application

The differential topology has the advantage that it is compensated for temperature drift of the internal reference

voltage. This can be explained by looking at the transimpedance amplifier of the LMV221 (Figure 79).

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REF

C

R

TRANS

I_DET

-

OUT

+

V

REF

+

-

Figure 79. Output Stage of the LMV221

Equation 14 shows that the output of the amplifier is set by the detection current I_DETmultiplied by the resistor

R_TRANSplus the reference voltage V_REF

:

V_OUT= I_DETR_TRANS+ V_REF

where

•

I_DETrepresents the detector current that is proportional to the RF input power.

(14)

Equation 14 shows that temperature variations in V_REFare also present in the output V_OUT. In case of a single

ended topology the output is the only pin that is connected to the ADC. The ADC voltage for single ended is

thus:

V_ADC= I_DETR_TRANS+ V_REF

(15)

A differential topology also connects the reference pin, which is the value of reference voltage V_REF. The ADC

reads V_OUT– V_REF

:

V_ADC= V_OUT– V_REF= I_DETR_TRANS

(16)

Equation 16 no longer contains the reference voltage V_REFanymore. Temperature variations in this reference

voltage are therefore not measured by the ADC.

8.2.1.3 Application Curves

2.0

1.6

1.2

0.8

0.4

0.0

RF = - 5 dBm

IN

1855 MHz

1.6

RF = -15 dBm

IN

900 MHz

RF = -25 dBm

IN

1.2

50 MHz

2500 MHz

0.8

RF = -35 dBm

IN

3000 MHz

0.4

RF = -45 dBm

IN

0.4

0.0

4000 MHz

0.0

10M

100M

1G

10G

-60 -50 -40 -30 -20 -10

RF INPUT POWER (dBm)

0

10

FREQUENCY (Hz)

Figure 80. Output Voltage vs RF Input Power

Figure 81. Output Voltage vs Frequency

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8.2.2 Application With Voltage Standing Wave Ratio (VSWR) Measurement

Transmission in RF systems requires matched termination by the proper characteristic impedance at the

transmitter and receiver side of the link. In wireless transmission systems, however, matched termination of the

antenna can rarely be achieved. The part of the transmitted power that is reflected at the antenna bounces back

toward the PA and may cause standing waves in the transmission line between the PA and the antenna. These

standing waves can attain unacceptable levels that may damage the PA. A VSWR measurement is used to

detect such an occasion. It acts as an alarm function to prevent damage to the transmitter.

VSWR is defined as the ratio of the maximum voltage divided by the minimum voltage at a certain point on the

transmission line:

1+ |G|

1 - |G|

VSWR =

where

•

Γ = V_REFLECTED/ V_FORWARDdenotes the reflection coefficient.

(17)

This means that to determine the VSWR, both the forward (transmitted) and the reflected power levels must be

measured. This can be accomplished by using two LMV221 RF power detectors according to Figure 82. A

directional coupler is used to separate the forward and reflected power waves on the transmission line between

the PA and the antenna. One secondary output of the coupler provides a signal proportional to the forward power

wave, the other secondary output provides a signal proportional to the reflected power wave. The outputs of both

RF detectors that measure these signals are connected to a microcontroller or baseband that calculates the

VSWR from the detector output signals.

COUPLER

ANTENNA

RF

PA

MICRO

CONTROLLER

V

DD

1

RF

IN

OUT

6

2

4

ADC1

REVERSE

POWER

LMV221

R

P1

C

P1

REF

EN

5

3

GND

RF

IN

OUT

REF

1

2

4

6

ADC2

TRANSMITTED

POWER

LMV221

R

P2

C

P2

EN

5

3

GND

Figure 82. VSWR Application

9 Power Supply Recommendations

The LMV221 is designed to operate from an input voltage supply range from 2.7 V to 3.3 V. This input voltage

must be well regulated. Enable voltage levels lower than 400 mV below GND could lead to incorrect operation of

the device. Also, the resistance of the input supply rail must be low enough to ensure correct operation of the

device.

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10 Layout

10.1 Layout Guidelines

As with any other RF device, careful attention must be paid to the board layout. If the board layout is not properly

designed, unwanted signals can easily be detected or interference picked up.

Electrical signals (voltages and currents) need a finite time to travel through a trace or transmission line. RF

voltage levels at the generator side and at the detector side can therefore be different. This is not only true for

the RF strip line, but for all traces on the PCB. Signals at different locations or traces on the PCB are in a

different phase of the RF frequency cycle. Phase differences in, for example, the voltage across neighboring

lines, may result in crosstalk between lines due to parasitic capacitive, or inductive coupling. This crosstalk is

further enhanced by the fact that all traces on the PCB are susceptible to resonance. The resonance frequency

depends on the trace geometry. Traces are particularly sensitive to interference when the length of the trace

corresponds to a quarter of the wavelength of the interfering signal or a multiple thereof.

10.1.1 Supply Lines

Because the PSRR of the LMV221 is finite, variations of the supply can result in some variation at the output.

This can be caused among others by RF injection from other parts of the circuitry or the on/off switching of the

PA.

10.1.1.1 Positive Supply (VDD)

In order to minimize the injection of RF interference into the LMV221 through the supply lines, the phase

difference between the PCB traces connecting to VDD and GND must be minimized. A suitable way to achieve

this is to short both connections for RF. This can be done by placing a small decoupling capacitor between the

VDD and GND. It must be placed as close to the device VDD and GND pins as possible as shown in Figure 85.

Be aware that the resonance frequency of the capacitor itself must be above the highest RF frequency used in

the application, because the capacitor acts as an inductor above its resonance frequency.

Low frequency-supply voltage variations due to PA switching might result in a ripple at the output voltage. The

LMV221 has a PSRR of 60 dB for low frequencies.

10.1.1.2 Ground (GND)

The LMV221 must have a ground plane free of noise and other disturbing signals. It is important to separate the

RF ground return path from the other grounds. This is due to the fact that the RF input handles large voltage

swings. A power level of 0 dBm causes a voltage swing larger than 0.6 V_PPover the internal 50-Ω input resistor,

resulting in a significant RF return current toward the source. Therefore, TI recommends that the RF ground

return path not be used for other circuits in the design. The RF path must be routed directly back to the source

without loops.

10.1.2 RF Input Interface

The LMV221 is designed to be used in RF applications having a characteristic impedance of 50 Ω. To achieve

this impedance, the input of the LMV221 must be connected via a 50-Ω transmission line. Transmission lines can

be easily created on PCBs using microstrip or (grounded) coplanar waveguide (GCPW) configurations. For more

details about designing microstrip or GCPW transmission lines, TI recommends a microwave designer handbook

is recommended.

10.1.3 Microstrip Configuration

One way to create a transmission line is to use a microstrip configuration. A cross section of the configuration is

shown in Figure 83, assuming a two-layer PCB.

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Layout Guidelines (continued)

METAL CONDUCTOR

W

FR4 PCB

H

GROUND PLANE

Figure 83. Microstrip Configuration

A conductor (trace) is placed on the topside of a PCB. The bottom side of the PCB has a fully copper ground

plane. The characteristic impedance of the microstrip transmission line is a function of the width W, height H, and

the dielectric constant ε_r.

Characteristics such as height and the dielectric constant of the board have significant impact on transmission

line dimensions. A 50-Ω transmission line may result in impractically wide traces. A typical 1.6-mm thick FR4

board results in a trace width of 2.9 mm, for instance. This is impractical for the LMV221 because the pad width

of the 6-pin WSON package is 0.25 mm. The transmission line has to be tapered from 2.9 mm to 0.25 mm.

Significant reflections and resonances in the frequency transfer function of the board may occur due to this

tapering.

10.1.4 GCPW Configuration

A transmission line in a (grounded) coplanar waveguide (GCPW) configuration gives more flexibility in terms of

trace width. The GCPW configuration is constructed with a conductor surrounded by ground at a certain

distance, S, on the top side. Figure 84 shows a cross section of this configuration. The bottom side of the PCB is

a ground plane. The ground planes on both sides of the PCB must be firmly connected to each other by multiple

vias. The characteristic impedance of the transmission line is mainly determined by the width W and the distance

S. In order to minimize reflections, the width W of the center trace must match the size of the package pad. The

required value for the characteristic impedance can subsequently be realized by selection of the proper gap

width S.

METAL CONDUCTOR

S

W

H

FR4 PCB

GROUND PLANE

Figure 84. GCPW Configuration

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Layout Guidelines (continued)

10.1.5 Reference (REF)

The REF pin can be used to compensate for temperature drift of the internal reference voltage as described in

Interface to the ADC. The REF pin is directly connected to the inverting input of the transimpedance amplifier.

Thus, RF signals and other spurious signals couple directly through to the output. Introduction of RF signals can

be prevented by connecting a small capacitor between the REF pin and ground. The capacitor must be placed

close to the REF pin as depicted in Figure 85.

10.1.6 Output (OUT)

The OUT pin is sensitive to crosstalk from the RF input, especially at high power levels. The ESD diode between

OUT and VDD may rectify the crosstalk, but may add an unwanted inaccurate DC component to the output

voltage.

The board layout must minimize crosstalk between the detectors input RFIN and the output of the detector. Using

an additional capacitor connected between the output and the positive supply voltage (VDD pin) or GND can

prevent this. For optimal performance this capacitor must be placed as close as possible to the OUT pin.

10.2 Layout Example

Figure 85. Recommended LMV221 Board Layout

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11 Device and Documentation Support

11.1 Community Resources

The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective

contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of

Use.

TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration

among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help

solve problems with fellow engineers.

Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and

contact information for technical support.

11.2 Trademarks

E2E is a trademark of Texas Instruments.

All other trademarks are the property of their respective owners.

11.3 Electrostatic Discharge Caution

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam

during storage or handling to prevent electrostatic damage to the MOS gates.

11.4 Glossary

SLYZ022 — TI Glossary.

This glossary lists and explains terms, acronyms, and definitions.

12 Mechanical, Packaging, and Orderable Information

The following pages include mechanical, packaging, and orderable information. This information is the most

current data available for the designated devices. This data is subject to change without notice and revision of

this document. For browser-based versions of this data sheet, refer to the left-hand navigation.

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MECHANICAL DATA

NGF0006A

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型号：	LMV221
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LMV221 [TI]

相关型号：

LMV221SD

LMV221SD/NOPB

LMV221SDX

LMV221SDX/NOPB

LMV225

LMV225

LMV225SD

LMV225SD

LMV225SD/NOPB

LMV225SD/NOPB

LMV225SDX

LMV225SDX