OPA657NB/250G4 [TI]
1.6GHz、低噪声、FET 输入运算放大器 | DBV | 5 | -40 to 85;
| 型号： | OPA657NB/250G4 |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | 1.6GHz、低噪声、FET 输入运算放大器 | DBV | 5 | -40 to 85 放大器 光电二极管 运算放大器 放大器电路 |
| 文件： | 总20页 (文件大小：389K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
OPA657
OPA657
SBOS197B – DECEMBER 2001 – REVISED MAY 2004
1.6GHz, Low-Noise, FET-Input
OPERATIONAL AMPLIFIER
FEATURES
DESCRIPTION
The OPA657 combines a high gain bandwidth, low distor-tion,
voltage-feedback op amp with a low voltage noise JFET-input
stage to offer a very high dynamic range amplifier for high
precision ADC (Analog-to-Digital Converter) driving or wideband
transimpedance applications. Photodiode applications will see
improved noise and bandwidth using this decompensated,
high gain bandwidth amplifier.
● HIGH GAIN BANDWIDTH PRODUCT: 1.6GHz
● HIGH BANDWIDTH 275MHz (G = +10)
● LOW INPUT OFFSET VOLTAGE: ±0.25mV
● LOW INPUT BIAS CURRENT: 2pA
● LOW INPUT VOLTAGE NOISE: 4.8nV/ Hz
● HIGH OUTPUT CURRENT: 70mA
Very low level signals can be significantly amplified in a single
OPA657 gain stage with exceptional bandwidth and accuracy.
Having a high 1.6GHz gain bandwidth product will give >
10MHz signal bandwidths up to gains of 160V/V (44dB). The
very low input bias current and capacitance will support this
performance even for relatively high source impedances.
● FAST OVERDRIVE RECOVERY
APPLICATIONS
● WIDEBAND PHOTODIODE AMPLIFIER
● WAFER SCANNING EQUIPMENT
● ADC INPUT AMPLIFIER
Broadband photodetector applications will benefit from the low
voltage noise JFET inputs for the OPA657. The JFET input
contributes virtually no current noise while for broadband
applications, a low voltage noise is also required. The low
4.8nV/ Hz input voltage noise will provide exceptional input
sensitivity for higher bandwidth applications. The example
shown below will give a total equivalent input noise current of
1.8pA/ Hz over a 10MHz bandwidth.
● TEST AND MEASUREMENT FRONT END
● HIGH GAIN PRECISION AMPLIFIER
RELATED OPERATIONAL AMPLIFIER PRODUCTS
200kΩ TRANSIMPEDANCE BANDWIDTH
116
SLEW VOLTAGE
10MHz Bandwidth
VS BW RATE NOISE
DEVICE (V) (MHz) (V/µS) (nV/√HZ) AMPLIFIER DESCRIPTION
106
OPA355 +5 200
OPA655 ±5 400
OPA656 ±5 500
OPA627 ±15 16
THS4601 ±15 180
300
290
170
55
5.80
6
Unity-Gain Stable CMOS
Unity-Gain Stable FET-Input
Unity-Gain Stable FET-Input
Unity-Gain Stable FET-Input
Unity-Gain Stable FET-Input
96
86
76
66
7
4.5
5.4
100
0.1pF
200kΩ
100kHz
1MHz
Frequency
10MHz
50MHz
λ
VO
OPA657
(12pF)
–Vb
Wideband Photodiode Transimpedance Amplifier
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
Copyright © 2001-2004, Texas Instruments Incorporated
www.ti.com
PACKAGE/ORDERING INFORMATION(1)
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
DESIGNATOR(1)
PACKAGE
MARKING
ORDERING
NUMBER(2)
TRANSPORT
MEDIA, QUANTITY
PRODUCT
PACKAGE-LEAD
OPA657U
SO-8 Surface Mount
D
–40°C to +85°C
OPA657U
OPA657U
OPA657U/2K5
OPA657UB
Rails, 100
"
"
"
D
"
"
Tape and Reel, 2500
Rails, 100
OPA657UB
SO-8 Surface Mount
–40°C to +85°C
OPA657UB
"
"
"
"
"
A57
"
OPA657UB/2K5
OPA657N/250
OPA657N/3K
OPA657NB/250
OPA657NB/3K
Tape and Reel, 2500
Tape and Reel, 250
Tape and Reel, 3000
Tape and Reel, 250
Tape and Reel, 3000
OPA657N
SOT23-5
DBV
–40°C to +85°C
"
"
"
"
OPA657NB
SOT23-5
DBV
"
–40°C to +85°C
A57
"
"
"
"
NOTES: (1) For the most current package and ordering infromation, see the Package Option Addendum located at the end of this data sheet.
ABSOLUTE MAXIMUM RATINGS(1)
ELECTROSTATIC
DISCHARGE SENSITIVITY
This integrated circuit can be damaged by ESD. Texas
Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper han-
dling and installation procedures can cause damage.
Supply Voltage ................................................................................. ±6.5V
Internal Power Dissipation........................... See Thermal Characteristics
Differential Input Voltage ..................................................................... ±VS
Input Voltage Range............................................................................ ±VS
Storage Temperature Range .........................................–40°C to +125°C
Lead Temperature ......................................................................... +260°C
Junction Temperature (TJ ) ........................................................... +175°C
ESD Rating (Human Body Model) .................................................. 2000V
(Machine Model) ............................................................ 200V
ESD damage can range from subtle performance degrada-
tion to complete device failure. Precision integrated circuits
may be more susceptible to damage because very small
parametric changes could cause the device not to meet its
published specifications.
NOTE: (1) Stresses above these ratings may cause permanent damage.
Exposure to absolute maximum conditions for extended periods may degrade
device reliability. These are stress ratings only, and functional operation of the
device at these or any other conditions beyond those specified is not implied.
PIN CONFIGURATIONS
Top View
SOT23-5
Top View
SO-8
Output
1
5
4
+VS
–VS
2
3
Noninverting Input
Inverting Input
NC
1
8
NC
Inverting Input
Noninverting Input
–VS
2
3
4
7
6
5
+VS
Output
NC
A57
Pin Orientation/Package Marking
OPA657
2
SBOS197B
www.ti.com
ELECTRICAL CHARACTERISTICS: VS = ±5V
RF = 453Ω, RL = 100Ω, and G = +10, unless otherwise noted. Figure 1 for AC performance.
OPA657U, N (Standard-Grade)
MIN/MAX OVER TEMPERATURE
TYP
0°C to
70°C(2)
–40°C to
+85°C(2)
MIN/ TEST
MAX LEVEL(3)
PARAMETER
CONDITIONS
+25°C
+25°C(1)
UNITS
AC PERFORMANCE (Figure 1)
Small-Signal Bandwidth
G = +7, VO = 200mVp-p
G = +10, VO = 200mVp-p
G = +20, VO = 200mVp-p
G > +40
350
275
90
1600
30
MHz
MHz
MHz
MHz
MHz
dB
MHz
V/µs
ns
Typ
Typ
Typ
Typ
Typ
Typ
Typ
Typ
Typ
Typ
C
C
C
C
C
C
C
C
C
C
C
C
C
C
C
C
C
Gain-Bandwidth Product
Bandwidth for 0.1dB flatness
Peaking at a Gain of +7
Large-Signal Bandwidth
Slew Rate
Rise-and-Fall Time
Settling Time to 0.02%
Harmonic Distortion
2nd-Harmonic
G = +10, 2Vp-p
7
G = +10, 2Vp-p
G = +10, 1V Step
0.2V Step
180
700
1
G = +10, VO = 2V Step
G = +10, f = 5MHz, VO = 2Vp-p
RL = 200Ω
20
ns
–70
–74
–99
–106
4.8
dBc
dBc
dBc
dBc
nV/√Hz
fA/√Hz
Typ
Typ
Typ
Typ
Typ
Typ
RL > 500Ω
RL = 200Ω
RL > 500Ω
f > 100kHz
3rd-Harmonic
Input Voltage Noise
Input Current Noise
f > 100kHz
1.3
DC PERFORMANCE(4)
Open-Loop Voltage Gain (AOL
Input Offset Voltage
Average Offset Voltage Drift
Input Bias Current
)
VCM = 0V, RL = 100Ω
VCM = 0V
70
±0.25
±2
±2
±1
65
64
±2.2
±12
±1800
±900
63
±2.6
±12
±5000
±2500
dB
mV
µV/°C
pA
Min
Max
Max
Max
Max
A
A
A
A
A
±1.8
±12
±20
±10
VCM = 0V
VCM = 0V
VCM = 0V
Input Offset Current
pA
INPUT
Most Positive Input Voltage(5)
+2.5
–4.0
89
+2.0
–3.5
83
+1.9
–3.4
81
+1.8
–3.3
79
V
V
dB
Min
Min
Min
A
A
A
Most Negative Input Voltage(5)
Common-Mode Rejection Ratio (CMRR)
Input Impedance
VCM = ±0.5V
Differential
Common-Mode
1012 || 0.7
1012 || 4.5
Ω || pF
Ω || pF
Typ
Typ
C
C
OUTPUT
Voltage Output Swing
No Load
RL = 100Ω
±3.9
±3.5
+70
–70
±3.7
±3.3
50
V
V
mA
mA
Ω
Typ
Min
Min
Min
Typ
B
A
A
A
C
±3.2
48
–48
±3.1
46
–46
Current Output, Sourcing
Current Output, Sinking
Closed-Loop Output Impedance
–50
G = +10, f = 0.1MHz
0.02
POWER SUPPLY
Specified Operating Voltage
Maximum Operating Voltage Range
Maximum Quiescent Current
Minimum Quiescent Current
Power-Supply Rejection Ratio (+PSRR)
(–PSRR)
±5
V
V
mA
mA
dB
dB
Typ
Max
Max
Min
Min
Min
A
A
A
A
A
A
±6
16
11.7
76
±6
16.2
11.4
74
±6
16.3
11.1
72
14
14
80
80
+VS = 4.50V to 5.50V
–VS = 4.50V to –5.50V
62
60
58
TEMPERATURE RANGE
Specified Operating Range: U,N Package
Thermal Resistance, θJA
U: SO-8
–40 to 85
°C
Typ
Junction-to-Ambient
125
150
°C/W
°C/W
Typ
Typ
N: SOT23-5
NOTES: (1) Junction temperature = ambient for 25°C guaranteed specifications.
(2) Junction temperature = ambient at low temperature limit: junction temperature = ambient +20°C at high temperature limit for over temperature guaranteed
specifications.
(3) Test Levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation.
(C) Typical value only for information.
(4) Current is considered positive out-of-node. VCM is the input common-mode voltage.
(5) Tested < 3dB below minimum specified CMRR at ±CMIR limits.
OPA657
SBOS197B
3
www.ti.com
ELECTRICAL CHARACTERISTICS: VS = ±5V: High Grade DC Specifications(1)
RF = 453Ω, RL = 100Ω, and G = +10, unless otherwise noted.
OPA657UB, NB (High-Grade)
TYP
MIN/MAX OVER TEMPERATURE
0°C to
70°C(3)
–40°C to
+85°C(3)
MIN/ TEST
MAX LEVEL(4)
PARAMETER
CONDITIONS
+25°C
+25°C(2)
UNITS
Input Offset Voltage
Input Offset Voltage Drift
Input Bias Current
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = ±0.5V
±0.1
±2
±1
±0.5
98
82
±0.6
±6
±5
±5
91
±0.85
±6
±450
±450
89
±0.9
±6
±1250
±1250
87
mV
µV/°C
pA
pA
dB
Max
Max
Max
Max
Min
Min
Min
A
A
A
A
A
A
A
Input Offset Current
Common-Mode Rejection Ratio (CMRR)
Power-Supply Rejection Ratio (+PSRR)
(–PSRR)
+VS = 4.5V to 5.5V
–VS = –4.5V to –5.5V
78
68
76
66
74
64
dB
dB
74
NOTES: (1) All other specifications are the same as the standard-grade.
(2) Junction temperature = ambient for 25°C guaranteed specifications.
(3) Junction temperature = ambient at low temperature limit: junction temperature = ambient +20°C at high temperature limit for over temperature
guaranteed specifications.
(4) Test Levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation.
OPA657
4
SBOS197B
www.ti.com
TYPICAL CHARACTERISTICS: VS = ±5V
TA = +25°C, G = +10, RF = 453Ω, RL = 100Ω, unless otherwise noted.
NONINVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
INVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
9
6
9
6
VO = 0.2Vp-p
VO = 0.2Vp-p
RG = 50Ω
G = +7
G = –12
G = –20
G = –50
3
3
0
0
–3
–6
–9
–12
–15
–18
–21
–3
–6
–9
–12
–15
–18
–21
G = +10
G = +20
G = +50
See Figure 1
See Figure 2
0.5
1
10
Frequency (MHz)
100
500
0.5
1
10
Frequency (MHz)
100
500
NONINVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
INVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
26
23
20
17
14
11
8
32
29
26
23
20
17
14
11
8
VO = 0.2Vp-p
VO = 0.2Vp-p
VO = 1Vp-p
VO = 1Vp-p
G = +10
G = –20
RF = 1kΩ
VO = 1Vp-p
VO = 2Vp-p
VO = 5Vp-p
VO = 5Vp-p
5
2
See Figure 1
See Figure 2
–1
–4
5
2
0.5
1
10
Frequency (MHz)
100
500
0.5
1
10
Frequency (MHz)
100
500
NONINVERTING PULSE RESPONSE
G = +10
INVERTING PULSE RESPONSE
0.8
0.6
1.6
0.8
0.6
1.6
G = –20
1.2
1.2
0.4
0.8
0.4
0.8
Large-Signal Right Scale
Small-Signal Left Scale
Large-Signal Right Scale
Small-Signal Left Scale
0.2
0.4
0.2
0.4
0
0
0
0
–0.2
–0.4
–0.6
–0.8
–0.4
–0.8
–1.2
–1.6
–0.2
–0.4
–0.6
–0.8
–0.4
–0.8
–1.2
–1.6
See Figure 1
See Figure 2
Time (10ns/div)
Time (10ns/div)
OPA657
SBOS197B
5
www.ti.com
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, G = +10, RF = 453Ω, RL = 100Ω, unless otherwise noted.
HARMONIC DISTORTION vs OUTPUT VOLTAGE (5MHz)
f = 5MHz
HARMONIC DISTORTION vs LOAD RESISTANCE
–60
–65
–60
–65
2nd Harmonic
VO = 2Vp-p
f = 5MHz
RL = 200Ω
2nd Harmonic
–70
–70
–75
–75
3rd Harmonic
–80
–80
–85
–85
–90
–90
–95
–95
–100
–105
–110
3rd Harmonic
See Figure 1
See Figure 1
–100
–105
100
1k
20
50
0.5
1
5
Resistance (Ω)
Output Voltage Swing (Vp-p)
HARMONIC DISTORTION vs OUTPUT VOLTAGE (1MHz)
f = 1MHz
HARMONIC DISTORTION vs FREQUENCY
–50
–60
–70
–75
VO = 2Vp-p
RL = 200Ω
R
L = 200Ω
2nd Harmonic
–80
–70
–85
2nd Harmonic
–80
–90
–95
–90
3rd Harmonic
–100
–105
–110
3rd Harmonic
See Figure 1
See Figure 1
–100
–110
0.2
1
10
0.5
1
5
Frequency (MHz)
Output Voltage Swing (Vp-p)
HARMONIC DISTORTION vs NONINVERTING GAIN
HARMONIC DISTORTION vs INVERTING GAIN
VO = 2Vp-p
–40
–50
–40
–50
VO = 2Vp-p
f = 5MHz
RL = 200Ω
R
G = 50Ω
f = 5MHz
L = 200Ω
–60
–60
R
2nd Harmonic
2nd Harmonic
3rd Harmonic
–70
–70
–80
–80
3rd Harmonic
–90
–90
–100
–110
–100
–110
See Figure 1, RG Adjusted
See Figure 2, RF Adjusted
5
10
10
50
Gain (V/V)
Gain (V/V)
OPA657
6
SBOS197B
www.ti.com
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, G = +10, RF = 453Ω, RL = 100Ω, unless otherwise noted.
2-TONE, 3RD-ORDER
INPUT CURRENT AND VOLTAGE NOISE DENSITY
INTERMODULATION SPURIOUS
100
10
1
–50
–60
PI
50Ω
20MHz
15MHz
PO
50Ω
OPA657
50Ω
453Ω
–70
50Ω
–80
Input Voltage Noise 4.8nV/√Hz
Input Current Noise 1.3fA/√Hz
10MHz
–90
5MHz
–100
10
1k
10
100
1k
10k
100k
1M
10M
–10
–8
–6
–4
–2
0
2
4
6
8
f (Hz)
Single-Tone Load Power (dBm)
COMMON-MODE REJECTION RATIO AND
POWER-SUPPLY REJECTION RATIO vs FREQUENCY
OPEN-LOOP GAIN AND PHASE
110
100
90
80
70
60
50
40
30
20
90
80
70
60
50
40
30
20
10
0
10
–12
CMRR
–34
20 log(AOL
)
–56
+PSRR
–78
–PSRR
–100
–122
–144
–166
–188
–210
< AOL
–10
10k
100k
1M
10M
100M
100
1k
10k
100k
1M
10M
100M
1G
Frequency (Hz)
Frequency (Hz)
FREQUENCY RESPONSE vs CAPACITIVE LOAD
CL = 10pF
RECOMMENDED RS vs CAPACITIVE LOAD
23
20
17
14
11
8
100
10
1
CL = 22pF
CL = 100pF
VI
RS
VO
1kΩ
OPA657
50Ω
CL
453Ω
5
50Ω
For Maximally Flat Frequency Response
2
1
10
Frequency (MHz)
100
500
100
1k
Capacitive Load (pF)
OPA657
SBOS197B
7
www.ti.com
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, G = +10, RF = 453Ω, RL = 100Ω, unless otherwise noted.
TYPICAL INPUT OFFSET VOLTAGE DRIFT
OVER TEMPERATURE
TYPICAL INPUT BIAS CURRENT
vs COMMON-MODE INPUT VOLTAGE
1.0
0.5
2.0
1.5
1.0
0.5
0
0
–0.5
–1.0
–1.5
–2.0
–0.5
–1.0
–50
–25
0
25
50
75
100
125
–3
–2
–1
0
1
2
3
Ambient Temperature (°C)
Common-Mode Input Voltage (V)
TYPICAL INPUT BIAS CURRENT DRIFT
OVER TEMPERATURE
SUPPLY AND OUTPUT CURRENT vs TEMPERATURE
1000
900
800
700
600
500
400
300
200
100
0
150
125
100
75
18
15
12
9
Supply Current
Left Scale
Right Scale
Sourcing Current
50
6
Left Scale
Sinking Current
25
3
0
0
–50
–25
0
25
50
75
100
125
–50
–25
0
25
50
75
100
125
Ambient Temperature (°C)
Ambient Temperature (°C)
NONINVERTING INPUT OVERDRIVE RECOVERY
INVERTING INPUT OVERDRIVE RECOVERY
Input Voltage
5
4
0.5
5
4
0.25
0.20
0.15
0.10
0.05
0
G = –20
0.4
Input Voltage
Right Scale
Right Scale
3
0.3
3
2
0.2
2
Output Voltage
Left Scale
1
0.1
1
0
0
0
–1
–2
–3
–4
–5
–0.1
–0.2
–0.3
–0.4
–0.5
–1
–2
–3
–4
–5
–0.05
–0.10
–0.15
–0.20
–0.25
Output Voltage
Left Scale
G = +10
See Figure 1
See Figure 2
Time (20ns/div)
Time (20ns/div)
OPA657
8
SBOS197B
www.ti.com
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, G = +10, RF = 453Ω, RL = 100Ω, unless otherwise noted.
CLOSED-LOOP OUTPUT IMPEDANCE
OUTPUT VOLTAGE AND CURRENT LIMITATIONS
1W Internal Power
vs FREQUENCY
5
4
10
1
OPA657
3
ZO
RL = 100Ω
RL = 50Ω
2
453Ω
1
50Ω
0
RL = 25Ω
–1
–2
–3
–4
–5
0.1
0.01
1W Internal Power
–100 –80 –60 –40 –20
0
20
40 60
80 100
0.1
1
10
100
IO (mA)
Frequency (MHz)
COMMON-MODE REJECTION RATIO
vs COMMON-MODE INPUT VOLTAGE
110
90
70
50
–5
–4
–3
–2
–1
0
1
2
3
4
5
Common-Mode Input Voltage (V)
OPA657
SBOS197B
9
www.ti.com
bandwidth for the OPA657. For lower non-inverting gains than
the minimum recommended gain of +7 for the OPA657,
consider the unity gain stable JFET input OPA656.
APPLICATIONS INFORMATION
WIDEBAND, NON-INVERTING OPERATION
The OPA657 provides a unique combination of low input
voltage noise, very high gain bandwidth, and the DC precision
of a trimmed JFET-input stage to give an exceptional high input
impedance, high gain stage amplifier. Its very high Gain Band-
width Product (GBP) can be used to either deliver high signal
bandwidths at high gains, or to extend the achievable bandwidth
or gain in photodiode-transimpedance applications. To achieve
the full performance of the OPA657, careful attention to PC
board layout and component selection is required as discussed
in the following sections of this data sheet.
WIDEBAND, INVERTING GAIN OPERATION
There can be significant benefits to operating the OPA657 as
an inverting amplifier. This is particularly true when a matched
input impedance is required. Figure 2 shows the inverting
gain circuit used as a starting point for the typical character-
istics showing inverting-mode performance.
+5V
+VS
Figure 1 shows the noninverting gain of +10 circuit used as
the basis for most of the Typical Characteristics. Most of the
curves were characterized using signal sources with 50Ω
driving impedance, and with measurement equipment pre-
senting a 50Ω load impedance. In Figure 1, the 50Ω shunt
resistor at the VI terminal matches the source impedance of
the test generator, while the 50Ω series resistor at the VO
terminal provides a matching resistor for the measurement
equipment load. Generally, data sheet voltage swing speci-
fications are at the output pin (VO in Figure 1) while output
power specifications are at the matched 50Ω load. The total
100Ω load at the output combined with the 500Ω total
feedback network load, presents the OPA657 with an effec-
tive output load of 83Ω for the circuit of Figure 1.
+
0.1µF
6.8µF
50Ω Load
VO
50Ω
OPA657
RG
50Ω
RF
1kΩ
50Ω Source
VI
0.1µF
6.8µF
+
–VS
–5V
+5V
+VS
FIGURE 2. Inverting G = –20 Specifications and Test Circuit.
0.1µF
6.8µF
+
Driving this circuit from a 50Ω source, and constraining the
gain resistor (RG) to equal 50Ω will give both a signal
bandwidth and noise advantage. RG in this case is acting as
both the input termination resistor and the gain setting
resistor for the circuit. Although the signal gain for the circuit
of Figure 2 is double that for Figure 1, their noise gains are
equal when the 50Ω source resistor is included. This has the
interesting effect of doubling the equivalent GBP for the
amplifier. This can be seen in comparing the G = +10 and
G = –20 small signal frequency response curves. Both show
about 250MHz bandwidth, but the inverting configuration of
Figure 2 is giving 6dB higher signal gain. If the signal source
is actually the low impedance output of another amplifier, RG
should be increased to the minimum value allowed at the
output of that amplifier and RF adjusted to get the desired
gain. It is critical for stable operation of the OPA657 that this
driving amplifier show a very low output impedance through
frequencies exceeding the expected closed-loop bandwidth
for the OPA657.
50Ω Source
50Ω Load
50Ω
VI
VO
50Ω
OPA657
RF
453Ω
RG
50Ω
6.8µF
0.1µF
+
–VS
–5V
FIGURE 1. Noninverting G = +10 Specifications and Test
Circuit.
Voltage-feedback op amps, unlike current-feedback amplifi-
ers, can use a wide range of resistor values to set their gain.
To retain a controlled frequency response for the noninverting
voltage amplifier of Figure 1, the parallel combination of
RF || RG should always < 150Ω. In the noninverting configura-
tion, the parallel combination of RF || RG will form a pole with
the parasitic input capacitance at the inverting node of the
OPA657 (including layout parasitics). For best performance,
this pole should be at a frequency greater than the closed-loop
Figure 2 also shows the noninverting input tied directly to
ground. Often, a bias current canceling resistor to ground is
included here to null out the DC errors caused by the input
bias currents. This is only useful when the input bias currents
are matched. For a JFET part like the OPA657, the input bias
currents do not match but are so low to begin with (< 5pA)
that DC errors due to input bias currents are negligible.
Hence, no resistor is recommended at the noninverting input
for the inverting signal gain condition.
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WIDEBAND, HIGH SENSITIVITY, TRANSIMPEDANCE
DESIGN
This will give an approximate –3dB bandwidth set by:
f−3dB GPB/2πRFCD) Hz
=
The high GBP and low input voltage and current noise for the
OPA657 make it an ideal wideband-transimpedance ampli-
fier for moderate to high transimpedance gains. Unity-gain
stability in the op amp is not required for application as a
transimpedance amplifier. One transimpedance design ex-
ample is shown on the front page of the data sheet. Designs
that require high bandwidth from a large area detector with
relatively high transimpedance gain will benefit from the low
input voltage noise for the OPA657. This input voltage noise
is peaked up over frequency by the diode source capaci-
tance, and can, in many cases, become the limiting factor to
input sensitivity. The key elements to the design are the
expected diode capacitance (CD) with the reverse bias volt-
age (–VB) applied, the desired transimpedance gain, RF, and
the GBP for the OPA657 (1600MHz). Figure 3 shows a
design from a 50pF source capacitance diode through a
200kΩ transimpedance gain. With these 3 variables set (and
including the parasitic input capacitance for the OPA657
added to CD), the feedback capacitor value (CF) may be set
to control the frequency response.
The example of Figure 3 will give approximately 5MHz flat
bandwidth using the 0.2pF feedback compensation.
If the total output noise is bandlimited to a frequency less
than the feedback pole frequency, a very simple expression
for the equivalent input noise current can be derived as:
2
2
E 2πC F
4kT
RF
EN
RF
(
)
N
D
I
EQ = IN2 +
+
+
3
Where:
iEQ = Equivalent input noise current if the output noise is
bandlimited to F < 1/(2πRFCF).
iN = Input current noise for the op amp inverting input.
eN = Input voltage noise for the op amp.
CD = Diode capacitance.
F = Bandlimiting frequency in Hz (usually a postfilter prior
to further signal processing).
4kT = 1.6E – 21J at T = 290°K
+5V
Evaluating this expression up to the feedback pole frequency
at 3.9MHz for the circuit of Figure 3, gives an equivalent input
noise current of 3.4pA/ Hz. This is much higher than the
1.2fA/ Hz for just the op amp itself. This result is being
dominated by the last term in the equivalent input noise
expression. It is essential in this case to use a low voltage
noise op amp like the OPA657. If lower transimpedance gain,
wider bandwidth solutions are needed, consider the bipolar
input OPA686 or OPA687. These parts offer comparable
gain bandwidth products but much lower input noise voltage
at the expense of higher input current noise.
Supply Decoupling
Not Shown
VO = ID RF
OPA657
RF
200kΩ
λ
ID
CD
CF
0.2pF
50pF
–5V
–VB
LOW GAIN COMPENSATION
FIGURE 3. Wideband, Low Noise, Transimpedance Amplifier.
Where a low gain is desired, and inverting operation is
acceptable, a new external compensation technique may be
used to retain the full slew rate and noise benefits of the
OPA657 while maintaining the increased loop gain and the
associated improvement in distortion offered by the decom-
pensated architecture. This technique shapes the loop gain
for good stability while giving an easily controlled 2nd-order
low-pass frequency response. Considering only the noise
gain for the circuit of Figure 4, the low-frequency noise gain,
(NG1) will be set by the resistor ratios while the high fre-
quency noise gain (NG2) will be set by the capacitor ratios.
The capacitor values set both the transition frequencies and
the high-frequency noise gain. If this noise gain, determined
by NG2 = 1 + CS/CF, is set to a value greater than the
recommended minimum stable gain for the op amp and the
noise gain pole, set by 1/RFCF, is placed correctly, a very well
controlled 2nd-order low-pass frequency response will result.
To achieve a maximally flat 2nd-order Butterworth frequency
response, the feedback pole should be set to:
1/(2πRFCF) = (GPB/(4πRFCD))
Adding the common-mode and differential mode input capaci-
tance (0.7 + 4.5)pF to the 50pF diode source capacitance of
Figure 3, and targeting a 200kΩ transimpedance gain using
the 1600MHz GBP for the OPA657 will require a feedback
pole set to 3.5MHz. This will require a total feedback capaci-
tance of 0.2pF. Typical surface-mount resistors have a para-
sitic capacitance of 0.2pF, therefore, while Figure 3 shows a
0.2pF feedback-compensation capacitor, this will actually be
the parasitic capacitance of the 200kΩ resistor.
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For the values shown in Figure 4, the f–3dB will be approximately
130MHz. This is less than that predicted by simply dividing the
GBP product by NG1. The compensation network controls the
bandwidth to a lower value while providing the full slew rate at
the output and an exceptional distortion performance due to
increased loop gain at frequencies below NG1 • Z0. The
capacitor values shown in Figure 4 are calculated for NG1 = 3
and NG2 = 10.5 with no adjustment for parasitics.
+5V
VO = –2 • VI
OPA657
RG
250Ω
RF
500Ω
VI
CS
27pF
CF
2.9pF
12
9
–5V
6
FIGURE 4. Broadband Low Gain Inverting External Com-
pensation.
3
0
170MHz
–3
To choose the values for both CS and CF, two parameters and
only three equations need to be solved. The first parameter is
the target high-frequency noise gain NG2, which should be
greater than the minimum stable gain for the OPA657. Here,
a target NG2 of 10.5 will be used. The second parameter is the
desired low-frequency signal gain, which also sets the low-
frequency noise gain NG1. To simplify this discussion, we will
target a maximally flat 2nd-order low-pass Butterworth fre-
quency response (Q = 0.707). The signal gain of –2 shown in
Figure 4 will set the low frequency noise gain to
NG1 = 1 + RF/RG (= 3 in this example). Then, using only these
two gains and the GBP for the OPA657 (1600MHz), the key
frequency in the compensation can be determined as:
–6
–9
–12
–15
–18
1
10
100
500
Frequency (MHz)
FIGURE 5. G = –2 Frequency Response with External
Compensation.
Figure 5 shows the measured frequency response for the
circuit of Figure 4. This is showing the expected gain of –2
with exceptional flatness through 70MHz and a –3dB band-
width of 170MHz.
GBP
NG12
NG1
NG2
NG1
NG2
ZO
=
1–
– 1– 2
The real benefit to this compensation is to allow a high slew
rate, exceptional DC precision op amp to provide a low
overshoot, fast settling pulse response. For a 1V output step,
the 700V/µs slew rate of the OPA657 will allow a rise time
limited edge rate (2ns for a 170Mhz bandwidth). While unity-
gain stable op amps may offer comparable bandwidths, their
lower slew rates will extend the settling time for larger steps.
For instance, the OPA656 can also provide a 150MHz gain of
–2 bandwidth implying a 2.3ns transition time. However, the
lower slew rate of this unity gain stable amplifier (290V/us) will
limit a 1V step transition to 3.5ns and delay the settling time as
the slewing transition is recovered. The combination of higher
slew rate and exceptional DC precision for the OPA657 can
yield one of the fastest, most precise, pulse amplifiers using
the circuit of Figure 4.
Physically, this Z0 (10.6MHz for the values shown above) is
set by 1/(2π • RF(CF + CS)) and is the frequency at which the
rising portion of the noise gain would intersect unity gain if
projected back to 0dB gain. The actual zero in the noise gain
occurs at NG1 • Z0 and the pole in the noise gain occurs at
NG2 • Z0. Since GBP is expressed in Hz, multiply Z0 by 2π
and use this to get CF by solving:
1
CF =
(= 2.86pF)
2π • RFZONG2
Finally, since CS and CF set the high-frequency noise gain,
determine CS by [Using NG2 = 10.5]:
CS = (NG2 – 1)CF
(= 27.2pF)
An added benefit to the compensation of Figure 4 is to
increase the loop gain above that achievable at comparable
gains by internally compensated amplifiers. The circuit of
Figure 4 will have lower harmonic distortion through 10Mhz
than the OPA656 operated at a gain of –2.
The resulting closed-loop bandwidth will be approximately
equal to:
(= 130MHz)
f–3dB
ZO GBP
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FREQUENCY RESPONSE CONTROL
OPERATING SUGGESTIONS
Voltage-feedback op amps exhibit decreasing closed-loop
bandwidth as the signal gain is increased. In theory, this
relationship is described by the Gain Bandwidth Product
(GBP) shown in the specifications. Ideally, dividing GBP by
the non-inverting signal gain (also called the Noise Gain, or
NG) will predict the closed-loop bandwidth. In practice, this
only holds true when the phase margin approaches 90°, as
it does in high-gain configurations. At low gains (increased
feedback factors), most high-speed amplifiers will exhibit a
more complex response with lower phase margin. The
OPA657 is compensated to give a maximally flat 2nd-order
Butterworth closed-loop response at a noninverting gain of
+10 (Figure 1). This results in a typical gain of +10 bandwidth
of 275MHz, far exceeding that predicted by dividing the
1600MHz GBP by 10. Increasing the gain will cause the
phase margin to approach 90° and the bandwidth to more
closely approach the predicted value of (GBP/NG). At a gain
of +50 the OPA657 will show the 32MHz bandwidth predicted
using the simple formula and the typical GBP of 1600MHz.
SETTING RESISTOR VALUES TO MINIMIZE NOISE
The OPA657 provides a very low input noise voltage while
requiring a low 14mA of quiescent current. To take full advan-
tage of this low input noise, a careful attention to the other
possible noise contributors is required. Figure 6 shows the op
amp noise analysis model with all the noise terms included. In
this model, all the noise terms are taken to be noise voltage or
current density terms in either nV/ Hz or pA/ Hz
.
ENI
*
EO
OPA657
RS
IBN
ERS
*
RF
√4kTRS
*
Inverting operation offers some interesting opportunities to
increase the available gain-bandwidth product. When the
source impedance is matched by the gain resistor (Figure 2),
the signal gain is –(RF/RG) while the noise gain for bandwidth
purposes is (1 + RF/RG). This cuts the noise gain in half,
increasing the minimum stable gain for inverting operation
under these condition to –12 and the equivalent gain band-
width product to 3.2GHz.
√4kTRF
IBI
RG
4kT
RG
4kT = 1.6E –20J
at 290°K
FIGURE 6. Op Amp Noise Analysis Model.
The total output spot noise voltage can be computed as the
square root of the squared contributing terms to the output
noise voltage. This computation is adding all the contributing
noise powers at the output by superposition, then taking the
square root to get back to a spot noise voltage. Equation 1
shows the general form for this output noise voltage using
the terms shown in Figure 7:
DRIVING CAPACITIVE LOADS
One of the most demanding and yet very common load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an A/D converter — including
additional external capacitance which may be recommended
to improve A/D linearity. A high speed, high open-loop gain
amplifier like the OPA657 can be very susceptible to de-
creased stability and closed-loop response peaking when a
capacitive load is placed directly on the output pin. When the
amplifier’s open loop output resistance is considered, this
capacitive load introduces an additional pole in the signal
path that can decrease the phase margin. Several external
solutions to this problem have been suggested. When the
primary considerations are frequency response flatness, pulse
response fidelity and/or distortion, the simplest and most
effective solution is to isolate the capacitive load from the
feedback loop by inserting a series isolation resistor between
the amplifier output and the capacitive load. This does not
eliminate the pole from the loop response, but rather shifts it
and adds a zero at a higher frequency. The additional zero
acts to cancel the phase lag from the capacitive load pole,
thus increasing the phase margin and improving stability.
(1)
EO
=
ENI2 + IBNRS 2 + 4kTRS NG2 + I R 2 + 4kTRFNG
BI F
(
)
(
)
Dividing this expression by the noise gain (GN = 1 + RF/RG)
will give the equivalent input referred spot noise voltage at
the non-inverting input as shown in Equation 2:
(2)
2
IBIRF
NG
4kTRF
NG
EN = ENI2 + IBNRS 2 + 4kTRS +
+
(
)
Putting high resistor values into Equation 2 can quickly
dominate the total equivalent input referred noise. A source
impedance on the noninverting input of 1.6kΩ will add a
Johnson voltage noise term equal to just that for the amplifier
itself (5nV/ Hz). While the JFET input of the OPA657 is ideal
for high source impedance applications, both the overall
bandwidth and noise may be limited by these higher source
impedances in the non-inverting configuration of Figure 1.
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The Typical Characteristics illustrate Recommended RS vs
Capacitive Load and the resulting frequency response at the
load. In this case, a design target of a maximally flat fre-
quency response was used. Lower values of RS may be used
if some peaking can be tolerated. Also, operating at higher
gains (than the +10 used in the Typical Characteristics) will
require lower values of RS for a minimally peaked frequency
response. Parasitic capacitive loads greater than 2pF can
begin to degrade the performance of the OPA657. Long PC
board traces, unmatched cables, and connections to multiple
devices can easily cause this value to be exceeded. Always
consider this effect carefully, and add the recommended
series resistor as close as possible to the OPA657 output pin
(see Board Layout section).
D.C. ACCURACY AND OFFSET CONTROL
The OPA657 can provide excellent DC accuracy due to its high
open-loop gain, high common-mode rejection, high power-supply
rejection, and its trimmed input offset voltage (and drift) along with
the negligible errors introduced by the low input bias current. For
the best DC precision, a high-grade version (OPA657UB or
OPA657NB) screens the key DC parameters to an even tighter
limit. Both standard- and high-grade versions take advantage of
a new final test technique to 100% test input offset voltage drift
over temperature. This discussion will use the high-grade typical
and min/max electrical characteristics for illustration, however, an
identical analysis applies to the standard-grade version.
The total output DC offset voltage in any configuration and
temperature will be the combination of a number of possible error
terms. In a JFET part like the OPA657, the input bias current
terms are typically quite low but are unmatched. Using bias
current cancellation techniques, more typical in bipolar input
amplifiers, does not improve output DC offset errors. Errors due
to the input bias current will only become dominant at elevated
temperatures. The OPA657 shows the typical 2X increase in
every 10°C common to JFET-input stage amplifiers. Using the
5pA maximum tested value at 25°C, and a 20°C internal self
heating (see thermal analysis), the maximum input bias current
at 85°C ambient will be 5pA • 2(105 – 25)/10 = 1280pA. For
noninverting configurations, this term only begins to be a signifi-
cant term versus the input offset voltage for source impedances
> 750kΩ. This would also be the feedback resistor value for
transimpedance applications (Figure 3) where the output DC
error due to inverting input bias current is on the order of that
contributed by the input offset voltage. In general, except for
these extremely high-impedance values, the output DC errors
due to the input bias current may be neglected.
DISTORTION PERFORMANCE
The OPA657 is capable of delivering a low distortion signal
at high frequencies over a wide range of gains. The distortion
plots in the Typical Characteristics show the typical distortion
under a wide variety of conditions.
Generally, until the fundamental signal reaches very high
frequencies or powers, the 2nd-harmonic will dominate the
distortion with negligible 3rd-harmonic component. Focusing
then on the 2nd-harmonic, increasing the load impedance
improves distortion directly. Remember that the total load
includes the feedback network—in the non-inverting configu-
ration this is sum of RF + RG, while in the inverting configura-
tion this is just RF (Figure 1). Increasing output voltage swing
increases harmonic distortion directly. A 6dB increase in
output swing will generally increase the 2nd-harmonic 12dB
and the 3rd-harmonic 18dB. Increasing the signal gain will also
increase the 2nd-harmonic distortion. Again a 6dB increase in
gain will increase the 2nd- and 3rd-harmonic by about 6dB
even with a constant output power and frequency. And finally,
the distortion increases as the fundamental frequency in-
creases due to the rolloff in the loop gain with frequency.
Conversely, the distortion will improve going to lower frequen-
cies down to the dominant open loop pole at approximately
100kHz. Starting from the –70dBc 2nd-harmonic for a 5MHz,
2Vp-p fundamental into a 200Ω load at G = +10 (from the
Typical Characteristics), the 2nd-harmonic distortion for fre-
quencies lower than 100kHz will be approximately < –90dBc.
After the input offset voltage itself, the most significant term
contributing to output offset voltage is the PSRR for the negative
supply. This term is modeled as an input offset voltage shift due
to changes in the negative power supply voltage (and similarly for
the +PSRR). The high-grade test limit for –PSRR is 68dB. This
translates into 0.40mV/V input offset voltage shift = 10(–68/20). This
low sensitivity to the negative supply voltage would require a 1.5V
change in the negative supply to match the ±0.6mV input offset
voltage error. The +PSRR is tested to a minimum value of 78dB.
This translates into 10(–78/20) = 0.125mV/V sensitivity for the input
offset voltage to positive power-supply changes.
The OPA657 has an extremely low 3rd-order harmonic distor-
tion. This also shows up in the 2-tone 3rd-order intermodulation
spurious (IM3) response curves. The 3rd-order spurious levels
are extremely low (< –80dBc) at low output power levels. The
output stage continues to hold them low even as the fundamen-
tal power reaches higher levels. As the Typical Characteristics
show, the spurious intermodulation powers do not increase as
predicted by a traditional intercept model. As the fundamental
power level increases, the dynamic range does not decrease
significantly. For 2 tones centered at 10MHz, with 4dBm/tone
into a matched 50Ω load (i.e., 1Vp-p for each tone at the load,
which requires 4Vp-p for the overall 2-tone envelope at the
output pin), the Typical Characteristics show a 82dBc difference
between the test tone and the 3rd-order intermodulation spuri-
ous levels. This exceptional performance improves further when
operating at lower frequencies and/or higher load impedances.
As an example, compute the worst-case output DC error for the
transimpedance circuit of Figure 3 at 25°C and then the shift
over the 0°C to 70°C range given the following assumptions.
Negative Power Supply
= –5V ±0.2V with a ±5mV/°C worst-case shift
Positive Power Supply
= +5V ±0.2V with a ±5mV/°C worst-case shift
Initial 25°C Output DC Error Band
= ±0.6mV (OPA657 high-grade input offset voltage limit)
±0.08mV (due to the –PSRR = 0.4mV/V • ±0.2V)
±0.04mV (due to the +PSRR = 0.2mV/V • ±0.2V)
Total = ±0.72mV
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This would be the worst-case error band in volume produc-
tion at 25°C acceptance testing given the conditions stated.
As a worst-case example, compute the maximum TJ using an
OPA657N (SOT23-5 package) in the circuit of Figure 1
operating at the maximum specified ambient temperature of
+85°C and driving a grounded 100Ω load.
Over the temperature range (0°C to 70°C), we can expect the
following worst-case shifting from initial value. A 20°C inter-
nal junction self-heating is assumed here.
PD = 10V • 16.1mA + 52 /(4 • (100Ω || 500Ω)) = 236mW
Maximum TJ = +85°C + (0.24W • 150°C/W) = 121°C.
±0.36mV (OPA656 high-grade input offset drift)
= ±6µV/°C • (70°C + 20°C – 25°C)
All actual applications will be operating at lower internal
power and junction temperature.
±0.11mV (–PSRR of 66dB with 5mV • (70°C – 25°C) supply shift)
±0.04mV (+PSRR of 76dB with 5mV • (70°C – 25°C) supply shift)
Total = ±0.51mV
BOARD LAYOUT
This would be the worst-case shift from an initial offset over
a 0°C to 70°C ambient for the conditions stated. Typical initial
output DC error bands and shifts over temperature will be
much lower than these worst-case estimates.
Achieving optimum performance with a high-frequency am-
plifier like the OPA657 requires careful attention to board
layout parasitics and external component types. Recommen-
dations that will optimize performance include:
In the transimpedance configuration, the CMRR errors can
be neglected since the input common-mode voltage is held
at ground. For noninverting gain configurations (Figure 1),
the CMRR term will need to be considered but will typically
be far lower than the input offset voltage term. With a tested
minimum of 91dB (28uV/V), the added apparent DC error will
be no more than ±0.06mV for a ±2V input swing to the circuit
of Figure 1.
a) Minimize parasitic capacitance to any AC ground for all
of the signal I/O pins. Parasitic capacitance on the output and
inverting input pins can cause instability—on the noninverting
input, it can react with the source impedance to cause
unintentional bandlimiting. To reduce unwanted capacitance,
a window around the signal I/O pins should be opened in all
of the ground and power planes around those pins. Other-
wise, ground and power planes should be unbroken else-
where on the board.
POWER-SUPPLY CONSIDERATIONS
b) Minimize the distance (< 0.25”) from the power-supply
pins to high-frequency 0.1uF decoupling capacitors. At the
device pins, the ground and power plane layout should not
be in close proximity to the signal I/O pins. Avoid narrow
power and ground traces to minimize inductance between
the pins and the decoupling capacitors. The power-supply
connections should always be decoupled with these capaci-
tors. Larger (2.2µF to 6.8µF) decoupling capacitors, effective
at lower frequency, should also be used on the supply pins.
These may be placed somewhat farther from the device and
may be shared among several devices in the same area of
the PC board.
The OPA657 is intended for operation on ±5V supplies.
Single-supply operation is allowed with minimal change from
the stated specifications and performance from a single
supply of +8V to +12V maximum. The limit to lower supply
voltage operation is the useable input voltage range for the
JFET-input stage. Operating from a single supply of +12V
can have numerous advantages. With the negative supply at
ground, the DC errors due to the –PSRR term can be
minimized. Typically, AC performance improves slightly at
+12V operation with minimal increase in supply current.
c) Careful selection and placement of external components
will preserve the high frequency performance of the OPA657.
Resistors should be a very low reactance type. Surface-mount
resistors work best and allow a tighter overall layout. Metal film
and carbon composition axially leaded resistors can also pro-
vide good high-frequency performance. Again, keep their leads
and PC board trace length as short as possible. Never use
wirewound-type resistors in a high-frequency application. Since
the output pin and inverting input pin are the most sensitive to
parasitic capacitance, always position the feedback and series
output resistor, if any, as close as possible to the output pin.
Other network components, such as noninverting input termina-
tion resistors, should also be placed close to the package.
Where double-side component mounting is allowed, place the
feedback resistor directly under the package on the other side
of the board between the output and inverting input pins. Even
with a low parasitic capacitance shunting the external resistors,
excessively high resistor values can create significant time
constants that can degrade performance. Good axial metal film
or surface-mount resistors have approximately 0.2pF in shunt
with the resistor. For resistor values > 1.5kΩ, this parasitic
THERMAL ANALYSIS
The OPA657 will not require heatsinking or airflow in most
applications. Maximum desired junction temperature will set
the maximum allowed internal power dissipation as de-
scribed below. In no case should the maximum junction
temperature be allowed to exceed 175°C.
Operating junction temperature (TJ) is given by TA + PD • θJA.
The total internal power dissipation (PD) is the sum of
quiescent power (PDQ) and additional power dissipated in the
output stage (PDL) to deliver load power. Quiescent power is
simply the specified no-load supply current times the total
supply voltage across the part. PDL will depend on the
required output signal and load but would—for a grounded
resistive load—be at a maximum when the output is fixed at
a voltage equal to 1/2 of either supply voltage (for equal
2
bipolar supplies). Under this condition PDL = VS /(4 • RL)
where RL includes feedback network loading.
Note that it is the power in the output stage and not into the
load that determines internal power dissipation.
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capacitance can add a pole and/or zero below 500MHz that can
effect circuit operation. Keep resistor values as low as possible
consistent with load driving considerations. It has been sug-
gested here that a good starting point for design would be to
keep RF || RG < 150Ω for voltage amplifier applications. Doing
this will automatically keep the resistor noise terms low, and
minimize the effect of their parasitic capacitance.
Transimpedance applications (Figure 3) can use whatever
feedback resistor is required by the application as long as the
feedback-compensation capacitor is set considering all parasitic
capacitance terms on the inverting node.
e) Socketing a high-speed part like the OPA657 is not
recommended. The additional lead length and pin-to-pin ca-
pacitance introduced by the socket can create an extremely
troublesome parasitic network which can make it almost impos-
sible to achieve a smooth, stable frequency response. Best
results are obtained by soldering the OPA657 onto the board.
INPUT AND ESD PROTECTION
The OPA657 is built using a very high-speed complementary
bipolar process. The internal junction breakdown voltages are
relatively low for these very small geometry devices. These
breakdowns are reflected in the Absolute Maximum Ratings
table. All device pins are protected with internal ESD protec-
tion diodes to the power supplies as shown in Figure 7.
d) Connections to other wideband devices on the board
may be made with short direct traces or through onboard
transmission lines. For short connections, consider the trace
and the input to the next device as a lumped capacitive load.
Relatively wide traces (50mils to 100mils) should be used,
preferably with ground and power planes opened up around
them. Estimate the total capacitive load and set RS from the
plot of Recommended RS vs Capacitive Load. Low parasitic
capacitive loads (< 5pF) may not need an RS since the
OPA657 is nominally compensated to operate with a 2pF
parasitic load. Higher parasitic capacitive loads without an RS
are allowed as the signal gain increases (increasing the
unloaded phase margin) If a long trace is required, and the
6dB signal loss intrinsic to a doubly-terminated transmission
line is acceptable, implement a matched-impedance trans-
mission line using microstrip or stripline techniques (consult
an ECL design handbook for microstrip and stripline layout
techniques). A 50Ω environment is normally not necessary
onboard, and in fact a higher impedance environment will
improve distortion as shown in the distortion versus load
plots. With a characteristic board trace impedance defined
based on board material and trace dimensions, a matching
series resistor into the trace from the output of the OPA657
is used as well as a terminating shunt resistor at the input of
the destination device. Remember also that the terminating
impedance will be the parallel combination of the shunt
resistor and the input impedance of the destination device—
this total effective impedance should be set to match the
trace impedance. If the 6dB attenuation of a doubly-termi-
nated transmission line is unacceptable, a long trace can be
series-terminated at the source end only. Treat the trace as
a capacitive load in this case and set the series resistor value
as shown in the plot of RS vs Capacitive Load. This will not
preserve signal integrity as well as a doubly-terminated line.
If the input impedance of the destination device is low, there
will be some signal attenuation due to the voltage divider
formed by the series output into the terminating impedance.
+VCC
External
Pin
Internal
Circuitry
–VCC
FIGURE 7. Internal ESD Protection.
These diodes provide moderate protection to input overdrive
voltages above the supplies as well. The protection diodes
can typically support 30mA continuous current. Where higher
currents are possible (e.g., in systems with ±12V supply parts
driving into the OPA657), current limiting series resistors
should be added into the two inputs. Keep these resistor
values as low as possible since high values degrade both
noise performance and frequency response.
OPA657
16
SBOS197B
www.ti.com
PACKAGE OPTION ADDENDUM
www.ti.com
27-May-2004
PACKAGING INFORMATION
ORDERABLE DEVICE
STATUS(1)
PACKAGE TYPE
PACKAGE DRAWING
PINS
PACKAGE QTY
OPA657N/250
OPA657N/3K
OPA657NB/250
OPA657NB/3K
OPA657U
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
SOP
SOP
SOP
SOP
SOIC
SOIC
SOIC
SOIC
DBV
DBV
DBV
DBV
D
5
5
5
5
8
8
8
8
250
3000
250
3000
100
OPA657U/2K5
OPA657UB
D
2500
100
D
OPA657UB/2K5
D
2500
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
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