SM73303MMX [TI]

5 MHz, Low Noise, RRO, Dual Operational Amplifier with CMOS Input; 5兆赫，低噪声，复制权，与CMOS输入双路运算放大器


型号：	SM73303MMX
厂家：	TEXAS INSTRUMENTS
描述：	5 MHz, Low Noise, RRO, Dual Operational Amplifier with CMOS Input 5兆赫，低噪声，复制权，与CMOS输入双路运算放大器运算放大器
文件：	总18页 (文件大小：457K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

SM73303

SM73303 5 MHz, Low Noise, RRO, Dual Operational Amplifier with CMOS Input

Literature Number: SNOSB94A

July 5, 2011

SM73303

5 MHz, Low Noise, RRO, Dual Operational Amplifier with

CMOS Input

General Description

Features

The SM73303 is a dual operational amplifier with both low

supply voltage and low supply current, making it ideal for

portable applications. The SM73303 CMOS input stage

drives the I_BIAScurrent down to 0.6 pA; this coupled with the

(Typical values, V⁺= 3.3V, T_A= 25°C, unless otherwise spec-

ified)

Input noise voltage

Input bias current

Offset voltage

12.8 nV/

0.6 pA

■

low noise voltage of 12.8 nV/

makes the SM73303 perfect

1.6 mV

80 dB

122 dB

for applications requiring active filters, transimpedance am-

plifiers, and HDD vibration cancellation circuitry.

CMRR

Open loop gain

Rail-to-rail output

GBW

Along with great noise sensitivity, small signal applications

will benefit from the large gain bandwidth of 5 MHz coupled

with the minimal supply current of 1.6 mA and a slew rate of

5.8 V/μs.

The SM73303 provides rail-to-rail output swing into heavy

loads. The input common-mode voltage range includes

ground, which is ideal for ground sensing applications.

5 MHz

5.8 V/µs

1.6 mA

Slew rate

Supply current

Supply voltage range

Operating temperature

8-pin MSOP package

2.7V to 5V

−40°C to 85°C

The SM73303 has a supply voltage spanning 2.7V to 5V and

is offered in an 8-pin MSOP package that functions across the

wide temperature range of −40°C to 85°C. This small package

makes it possible to place the SM73303 next to sensors, thus

reducing external noise pickup.

Applications

Active filters

■

Transimpedance amplifiers

Audio preamp

HDD vibration cancellation circuitry

Typical Application Circuit

30157839

High Gain Band Pass Filter

301578

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Junction Temperature (Note 3)

Mounting Temperature

Infrared or Convection (20 sec)

150°C max

260°C

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Operating Ratings (Note 1)

ESD Tolerance (Note 2)

Human Body Model

Machine Model

Supply Voltage

2.7V to 5V

−40°C to 85°C

2000V

200V

Temperature Range

Supply Voltage (V⁺– V⁻)

Thermal Resistance (θ_JA

)

5.5V

8-Pin MSOP

195°C/W

Storage Temperature Range

−65°C to 150°C

3.3V Electrical Characteristics (Note 4) Unless otherwise specified, all limits are guaranteed for

T_J= 25°C, V⁺= 3.3V, V⁻= 0V. V_CM= V⁺/2. Boldface limits apply at the temperature extremes (Note 5).

Min

(Note 6)

Typ

(Note 7)

Max

(Note 6)

Symbol

V_OS

Parameter

Input Offset Voltage

Condition

V_CM= 1V

Units

1.6

0.6

I_B

Input Bias Current

(Note 8)

115

130

I_OS

Input Offset Current

CMRR

Common Mode Rejection Ratio

0 ≤ V_CM≤ 2.1V

2.7V ≤ V⁺≤ 5V, V_CM= 1V

PSRR

Power Supply Rejection Ratio

CMVR

A_VOL

Common Mode Voltage Range

Open Loop Voltage Gain

−0.2

2.2

For CMRR ≥ 50 dB

Sourcing

R_L= 10 kΩ to V⁺/2,

V_O= 1.65V to 2.9V

122

105

112

Sinking

R_L= 10 kΩ to V⁺/2,

V_O= 0.4V to 1.65V

Sourcing

R_L= 600Ω to V⁺/2,

V_O= 1.65V to 2.8V

Sinking

R_L= 600Ω to V⁺/2,

V_O= 0.5V to 1.65V

R_L= 10 kΩ to V⁺/2

R_L= 600Ω to V⁺/2

R_L= 10 kΩ to V⁺/2

R_L= 600Ω to V⁺/2

Sourcing, V_O= 0V

Sinking, V_O= 3.3V

V_CM= 1V

V_O

Output Swing High

Output Swing Low

Output Current

3.22

3.17

3.29

3.22

0.03

0.07

3.12

3.07

0.12

0.16

0.23

0.27

I_OUT

I_S

Supply Current

1.6

2.0

Slew Rate

(Note 9)

5.8

V/µs

MHz

GBW

Gain Bandwidth

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Min

(Note 6)

Typ

(Note 7)

Max

(Note 6)

Symbol

Parameter

Condition

Units

e_n

i_n

Input-Referred Voltage Noise

Input-Referred Current Noise

f = 1 kHz

12.8

0.01

nV/

pA/

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see the Electrical Characteristics.

Note 2: Human Body Model is 1.5 kΩ in series with 100 pF. Machine Model is 0Ω in series with 100 pF.

Note 3: The maximum power dissipation is a function of T_J(MAX), θ_JAand T_A. The maximum allowable power dissipation at any ambient temperature is

P_D= (T_J(MAX)-T_A)/θ_JA. All numbers apply for packages soldered directly into a PC board.

Note 4: Electrical Table values apply only for factory testing conditions at the temperature indicated. Factor testing conditions result in very limited self-heating

of the device such that T_J= T_A. No guarantee of parametric performance is indicated in the electrical tables under conditions of internal self-heating where T_J>

T_A. Absolute Maximum Ratings indicate junction temperature limits beyond which the device maybe permanently degraded, either mechanically or electrically.

Note 5: Boldface limits apply to temperature range of −40°C to 85°C.

Note 6: All limits are guaranteed by testing or statistical analysis.

Note 7: Typical values represent the most likely parametric norm.

Note 8: Input bias current is guaranteed by design.

Note 9: Number specified is the lower of the positive and negative slew rates.

Connection Diagram

8-Pin MSOP

30157840

Top View

Ordering Information

Package

Part Number

SM73303MM

SM73303MME

SM73303MMX

Package Marking

Transport Media

NSC Drawing

1k Units Tape and Reel

250 Units Tape and Reel

3.5k Units Tape and Reel

8-Pin MSOP

S303

MUA08A

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Simplified Schematic

30157829

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Typical Performance Characteristics Unless otherwise specified, V⁺3.3V, T_J= 25°C.

Supply Current vs. Supply Voltage

Offset Voltage vs. Common Mode

30157806

30157805

Input Bias Current vs. Common Mode

30157827

30157826

Input Bias Current vs. Common Mode

Output Positive Swing vs. Supply Voltage

30157885

30157825

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Output Negative Swing vs. Supply Voltage

Output Positive Swing vs. Supply Voltage

30157802

30157801

Output Negative Swing vs. Supply Voltage

Sinking Current vs. V_OUT

30157884

30157803

Sourcing Current vs. V_OUT

PSRR vs. Frequency

30157804

30157831

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CMRR vs. Frequency

Crosstalk Rejection

30157836

30157837

Inverting Large Signal Pulse Response

Inverting Small Signal Pulse Response

30157835

30157833

Non-Inverting Large Signal Pulse Response

Non-Inverting Small Signal Pulse Response

30157834

30157832

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Open Loop Frequency vs. R_L

Open Loop Frequency Response vs. C_L

Voltage Noise vs. Frequency

Open Loop Frequency Response over Temperature

30157821

30157822

Open Loop Frequency Response vs. C_L

30157823

30157828

30157824

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by preserving phase margin in the overall feedback loop.

Increased capacitive drive is possible by increasing the value

of C_F. This in turn will slow down the pulse response.

Application Information

With the low supply current of only 1.6 mA, the SM73303 of-

fers users the ability to maximize battery life. This makes the

SM73303 ideal for battery powered systems. The SM73303’s

rail-to-rail output swing provides the maximum possible dy-

namic range at the output. This is particularly important when

operating on low supply voltages.

CAPACITIVE LOAD TOLERANCE

The SM73303, when in a unity-gain configuration, can directly

drive large capacitive loads in unity-gain without oscillation.

The unity-gain follower is the most sensitive configuration to

capacitive loading; direct capacitive loading reduces the

phase margin of amplifiers. The combination of the amplifier’s

output impedance and the capacitive load induces phase lag.

This results in either an underdamped pulse response or os-

cillation. To drive a heavier capacitive load, the circuit in

Figure 1 can be used.

30157809

FIGURE 2. Indirectly Driving a Capacitive Load with DC

Accuracy

DIFFERENCE AMPLIFIER

The difference amplifier allows the subtraction of two voltages

or, as a special case, the cancellation of a signal common to

two inputs. It is useful as a computational amplifier in making

a differential to single-ended conversion or in rejecting a com-

mon mode signal.

30157807

FIGURE 1. Indirectly Driving a Capacitive Load using

Resistive Isolation

In Figure 1, the isolation resistor R_ISOand the load capacitor

C_Lform a pole to increase stability by adding more phase

margin to the overall system. The desired performance de-

pends on the value of R_ISO. The bigger the R_ISOresistor value,

the more stable V_OUTwill be.

The circuit in Figure 2 is an improvement to the one in Figure

1 because it provides DC accuracy as well as AC stability. If

there were a load resistor in Figure 1, the output would be

voltage divided by R_ISOand the load resistor. Instead, in Fig-

ure 2, R_Fprovides the DC accuracy by using feed-forward

techniques to connect V_INto R_L. Due to the input bias current

of the SM73303, the designer must be cautious when choos-

ing the value of R_F. C_Fand R_ISOserve to counteract the loss

of phase margin by feeding the high frequency component of

the output signal back to the amplifier’s inverting input, there-

30157810

FIGURE 3. Difference Amplifier

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SINGLE-SUPPLY INVERTING AMPLIFIER

There may be cases where the input signal going into the

amplifier is negative. Because the amplifier is operating in

single supply voltage, a voltage divider using R₃and R₄is

implemented to bias the amplifier so the inverting input signal

is within the input common voltage range of the amplifier. The

capacitor C₁is placed between the inverting input and resistor

R₁to block the DC signal going into the AC signal source,

V_IN. The values of R₁and C₁affect the cutoff frequency, fc =

½π R₁C₁. As a result, the output signal is centered around

mid-supply (if the voltage divider provides V⁺/2 at the non-

inverting input). The output can swing to both rails, maximiz-

ing the signal-to-noise ratio in a low voltage system.

FIGURE 4. Single-supply Inverting Amplifier

INSTRUMENTATION AMPLIFIER

Measurement of very small signals with an amplifier requires

close attention to the input impedance of the amplifier, the

overall signal gain from both inputs to the output, as well as,

the gain from each input to the output. This is because we are

only interested in the difference of the two inputs and the

common signal is considered noise. A classic solution is an

instrumentation amplifier. Instrumentation amplifiers have a

finite, accurate, and stable gain. Also they have extremely

high input impedances and very low output impedances. Fi-

nally they have an extremely high CMRR so that the amplifier

can only respond to the differential signal.

Three-Op-Amp Instrumentation Amplifier

A typical instrumentation amplifier is shown in Figure 5.

30157815

30157842

FIGURE 5. Three-Op-Amp Instrumentation Amplifier

There are two stages in this configuration. The last stage, the

output stage, is a differential amplifier. In an ideal case the

two amplifiers of the first stage, the input stage, would be set

up as buffers to isolate the inputs. However they cannot be

connected as followers due to the mismatch of real amplifiers.

The circuit in Figure 5 utilizes a balancing resistor between

the two amplifiers to compensate for this mismatch. The prod-

uct of the two stages of gain will be the gain of the instrumen-

tation amplifier circuit. Ideally, the CMRR should be infinite.

However the output stage has a small non-zero common

mode gain which results from resistor mismatch.

By Ohm’s Law:

(2)

However:

(3)

In the input stage of the circuit, current is the same across all

resistors. This is due to the high input impedance and low

input bias current of the SM73303. With the node equations

we have:

So we have:

(4)

(1)

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Now looking at the output of the instrumentation amplifier:

Low Pass Filter

The following shows a very simple low pass filter.

(5)

Substituting from Equation 4:

(6)

This shows the gain of the instrumentation amplifier to be:

−K(2a+1)

Typical values for this circuit can be obtained by setting: a =

12 and K = 4. This results in an overall gain of −100.

Three SM73303 amplifiers are used along with 1% resistors

to minimize resistor mismatch. Resistors used to build the

circuit are: R₁= 21.6 kΩ, R₁₁= 1.8 kΩ, R₂= 2.5 kΩ with K =

40 and a = 12. This results in an overall gain of −K(2a+1) =

−1000.

30157853

FIGURE 7. Low Pass Filter

The transfer function can be expressed as follows:

By KCL:

Two-Op-Amp Instrumentation Amplifier

A two-op-amp instrumentation amplifier can also be used to

make a high-input impedance DC differential amplifier Figure

6). As in the three op amp circuit, this instrumentation ampli-

fier requires precise resistor matching for good CMRR. R₄

should be equal to R₁, and R₃should equal R₂.

(7)

(8)

(9)

Simplifying this further results in:

Now, substituting ω=2πf, so that the calculations are in f(Hz)

rather than in ω(rad/s), and setting the DC gain

and

30157813

FIGURE 6. Two-Op-Amp Instrumentation Amplifier

ACTIVE FILTERS

(10)

set:

Active filters are circuits with amplifiers, resistors, and capac-

itors. The use of amplifiers instead of inductors, which are

used in passive filters, enhances the circuit performance

while reducing the size and complexity of the filter. The sim-

plest active filters are designed using an inverting op amp

configuration where at least one reactive element has been

added to the configuration. This means that the op amp will

provide "frequency-dependent" amplification, since reactive

elements are frequency dependent devices.

(11)

Low pass filters are known as lossy integrators because they

only behave as integrators at higher frequencies. The general

form of the bode plot can be predicted just by looking at the

transfer function. When the f/f_Oratio is small, the capacitor is,

in effect, an open circuit and the amplifier behaves at a set

DC gain. Starting at f_O, which is the −3 dB corner, the capac-

itor will have the dominant impedance and hence the circuit

will behave as an integrator and the signal will be attenuated

and eventually cut. The bode plot for this filter is shown in

Figure 8.

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Looking at the transfer function, it is clear that when f/f_Ois

small, the capacitor is open and therefore, no signal is getting

to the amplifier. As the frequency increases the amplifier

starts operating. At f = f_Othe capacitor behaves like a short

circuit and the amplifier will have a constant, high frequency

gain of H_O. Figure 10 shows the transfer function of this high

pass filter.

30157859

FIGURE 8. Low Pass Filter Transfer Function

High Pass Filter

The transfer function of a high pass filter can be derived in

much the same way as the previous example. A typical first

order high pass filter is shown below:

30157864

FIGURE 10. High Pass Filter Transfer Function

Band Pass Filter

Combining a low pass filter and a high pass filter will generate

a band pass filter. Figure 11 offers an example of this type of

circuit.

30157860

FIGURE 9. High Pass Filter

Writing the KCL for this circuit :

(V₁denotes the voltage between C and R₁)

(12)

(13)

30157866

FIGURE 11. Band Pass Filter

In this network the input impedance forms the high pass filter

while the feedback impedance forms the low pass filter. If the

designer chooses the corner frequencies so that f₁< f₂, then

all the frequencies between, f₁≤ f ≤ f₂, will pass through the

filter while frequencies below f₁and above f₂will be cut off.

Solving these two equations to find the transfer function and

using:

The transfer function can be easily calculated using the same

methodology as before and is shown in Figure 12.

(high frequency gain)

Which gives:

and

(15)

(14)

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Where

The transfer function of each filter needs to be calculated. The

derivations will be more trivial if each stage of the filter is

shown on its own.

The three components are:

(16)

30157870

30157868

FIGURE 12. Band Pass Filter Transfer Function

STATE VARIABLE ACTIVE FILTER

State variable active filters are circuits that can simultane-

ously represent high pass, band pass, and low pass filters.

The state variable active filter uses three separate amplifiers

to achieve this task. A typical state variable active filter is

shown in Figure 13. The first amplifier in the circuit is con-

nected as a gain stage. The second and third amplifiers are

connected as integrators, which means they behave as low

pass filters. The feedback path from the output of the third

amplifier to the first amplifier enables this low frequency signal

to be fed back with a finite and fairly low closed loop gain. This

is while the high frequency signal on the input is still gained

up by the open loop gain of the first amplifier. This makes the

first amplifier a high pass filter. The high pass signal is then

fed into a low pass filter. The outcome is a band pass signal,

meaning the second amplifier is a band pass filter. This signal

is then fed into the third amplifiers input and so, the third am-

plifier behaves as a simple low pass filter.

30157871

For A₁the relationship between input and output is:

(17)

This relationship depends on the output of all the filters. The

input-output relationship for A₂can be expressed as:

(18)

And finally this relationship for A₃is as follows:

(19)

Re-arranging these equations, one can find the relationship

between V_Oand V_IN(transfer function of the low pass filter),

V_O1and V_IN(transfer function of the high pass filter), and

V_O2and V_IN(transfer function of the band pass filter) These

relationships are as follows:

30157869

FIGURE 13. State Variable Active Filter

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Low Pass Filter

Designing a band pass filter with a center frequency of 10 kHz

and Quality Factor of 5.5

To do this, first consider the Quality Factor. It is best to pick

convenient values for the capacitors. C₂= C₃= 1000 pF. Also,

choose R₁= R₄= 30 kΩ. Now values of R₅and R₆need to be

calculated. With the chosen values for the capacitors and re-

sistors, Q reduces to:

(20)

High Pass Filter

Band Pass Filter

(24)

R₅= 10R₆

R₆= 1.5 kΩ

R₅= 15 kΩ

(25)

(21)

(22)

Also, for

ωc = 2πf = 62.8 kHz.

10 kHz, the center frequency is

Using the expressions above, the appropriate resistor values

will be R₂= R₃= 16 kΩ.

The DC gain of this circuit is:

The center frequency and Quality Factor for all of these filters

is the same. The values can be calculated in the following

manner:

(26)

(23)

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Physical Dimensions inches (millimeters) unless otherwise noted

8-Pin MSOP

NS Package Number MUA08A

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Following are URLs where you can obtain information on other Texas Instruments products and application solutions:

Products

Audio

Applications

www.ti.com/audio

amplifier.ti.com

dataconverter.ti.com

www.dlp.com

Communications and Telecom www.ti.com/communications

Amplifiers

Data Converters

DLP® Products

DSP

Computers and Peripherals

Consumer Electronics

Energy and Lighting

Industrial

www.ti.com/computers

www.ti.com/consumer-apps

www.ti.com/energy

dsp.ti.com

www.ti.com/industrial

www.ti.com/medical

www.ti.com/security

Clocks and Timers

Interface

www.ti.com/clocks

interface.ti.com

logic.ti.com

Medical

Security

Logic

Space, Avionics and Defense www.ti.com/space-avionics-defense

Transportation and Automotive www.ti.com/automotive

Power Mgmt

Microcontrollers

RFID

power.ti.com

microcontroller.ti.com

www.ti-rfid.com

Video and Imaging

www.ti.com/video

OMAP Mobile Processors www.ti.com/omap

Wireless Connectivity www.ti.com/wirelessconnectivity

TI E2E Community Home Page

e2e.ti.com

Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265

SM73303MMX [TI]

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