THS6012 [TI]
500-mA DUAL DIFFERENTIAL LINE DRIVER; 500 mA的双差分线路驱动器
| 型号： | THS6012 |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | 500-mA DUAL DIFFERENTIAL LINE DRIVER |
| 文件： | 总35页 (文件大小：603K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
Thermally Enchanced SOIC (DWP)
PowerPAD Package
(TOP VIEW)
ADSL Differential Line Driver
400 mA Minimum Output Current Into 25-Ω
Load
1
2
3
4
5
6
7
8
9
10
20
19
18
17
16
15
14
13
12
11
V
1OUT
V
1IN+
1IN–
NC
–
CC
V
2OUT
V
2IN+
2IN–
NC
NC
NC
High Speed
– 140 MHz Bandwidth (–3dB) With 25-Ω
CC–
CC+
Load
CC+
– 315 MHz Bandwidth (–3dB) With 100-Ω
Load
– 1300 V/µs Slew Rate, G = 5
NC
NC
NC
NC
Low Distortion
– –72 dB 3rd Order Harmonic Distortion at
NC
NC
f = 1 MHz, 25-Ω Load, and 20 V
PP
Independent Power Supplies for Low
Crosstalk
Wide Supply Range ±4.5 V to ±16 V
Thermal Shutdown and Short Circuit
Protection
Cross Section View Showing PowerPAD
Improved Replacement for AD815
Evaluation Module Available
MicroStar Junior (GQE) Package
(TOP VIEW)
description
The THS6012 contains two high-speed drivers
capable of providing 400 mA output current (min)
into a 25 Ω load. These drivers can be configured
differentially to drive a 50-Vp-p output signal over
low-impedance lines. The drivers are current
feedback amplifiers, designed for the high slew
rates necessary to support low total harmonic
(SIDE VIEW)
distortion (THD) in xDSL applications. The THS6012 is ideally suited for asymmetrical digital subscriber line
(ADSL) applications at the central office, where it supports the high-peak voltage and current requirements of
this application.
Separate power supply connections for each driver are provided to minimize crosstalk. The THS6012 is
available in the small surface-mount, thermally enhanced 20-pin PowerPAD package.
HIGH-SPEED xDSL LINE DRIVER/RECEIVER FAMILY
DEVICE
THS6002
THS6012
THS6022
THS6032
THS6062
THS7002
DRIVER RECEIVER
DESCRIPTION
•
Dual differential line drivers and receivers
500-mA dual differential line driver
250-mA dual differential line driver
Low-power ADSL central office line driver
Low-noise ADSL receiver
•
•
•
•
•
•
Low-noise programmable gain ADSL receiver
CAUTION: The THS6012 provides ESD protection circuitry. However, permanent damage can still occur if this device is subjected
to high-energy electrostatic discharges. Proper ESD precautions are recommended to avoid any performance degradation or loss
of functionality.
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments Incorporated.
Copyright 2000, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
1
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
AVAILABLE OPTIONS
PACKAGED DEVICE
PowerPAD PLASTIC
SMALL OUTLINE
(DWP)
T
A
MicroStar Junior
(GQE)
EVALUATION
MODULE
†
0°C to 70°C
THS6012CDWP
THS6012IDWP
THS6012CGQE
THS6012IGQE
THS6012EVM
—
–40°C to 85°C
†
The PWP packages are available taped and reeled. Add an R suffix to the device type (i.e.,
THS6012CPWPR)
functional block diagram
Driver 1
3
2
V
+
CC
4
5
1IN+
1IN–
+
1OUT
_
1
V
V
CC–
Driver 2
18
CC+
17
16
+
2IN+
2IN–
19
20
2OUT
_
V
CC–
Terminal Functions
TERMINAL
NAME
1OUT
DWP PACKAGE
TERMINAL NO.
GQE PACKAGE
TERMINAL NO.
2
5
A3
F1
1IN–
1IN+
2OUT
2IN–
2IN+
4
D1
19
A7
16
F9
17
D9
V
V
3, 18
1, 20
B1, B9
A4, A6
NA
CC+
CC–
NC
6, 7, 8 ,9, 10, 11, 12, 13,
14, 15
2
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
pin assignments
MicroStar Junior (GQE) Package
(TOP VIEW)
1
2
3
4
5
6
7
8
9
A
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
B
C
V
V
CC+
CC+
1N+
1IN–
NC
NC
NC
NC
D
E
F
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
2IN+
2IN–
G
H
J
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC
NOTE: Shaded terminals are used for thermal connection to the ground plane.
3
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
†
absolute maximum ratings over operating free-air temperature (unless otherwise noted)
Supply voltage, V
to V
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 V
CC–
CC+
Input voltage, V (driver and receiver) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±V
I
CC
Output current, I (driver) (see Note 1) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 800 mA
O
Differential input voltage, V
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V
ID
Continuous total power dissipation at (or below) T = 25°C (see Note 1) . . . . . . . . . . . . . . . . . . . . . . . . . . 5.8 W
Operating free air temperature, T . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to 85°C
Storage temperature, T
A
A
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to 125°C
stg
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 300°C
†
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
NOTE 1: The THS6012 incorporates a PowerPad on the underside of the chip. This acts as a heatsink and must be connected to a thermal
dissipation plane for proper power dissipation. Failure to do so can result in exceeding the maximum junction temperature, which could
permanently damage the device. See the Thermal Information section of this document for more information about PowerPad
technology.
recommended operating conditions
MIN
±4.5
9
TYP
MAX
±16
32
UNIT
Split supply
Single supply
C suffix
Supply voltage, V
and V
V
CC+
CC–
0
70
Operating free-air temperature, T
°C
A
I suffix
–40
85
4
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
electrical characteristics, V
= ±15 V, R = 25 Ω, R = 1 kΩ, T = 25°C (unless otherwise noted)
CC
L
F
A
dynamic performance
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
V = 200 mV,
G = 1,
R = 25 Ω
L
I
V
V
V
V
V
= ±15 V
= ±5 V
140
CC
CC
CC
CC
CC
R
= 680 Ω,
F
V = 200 mV,
G = 1,
R = 25 Ω
L
I
F
100
120
100
315
265
30
R
= 1 kΩ,
V = 200 mV,
G = 2,
R = 25 Ω
L
I
F
= ±15 V
= ±5 V
R
= 620 Ω,
Small-signal bandwidth (–3 dB)
MHz
V = 200 mV,
G = 2,
R = 820 Ω
F
I
L
R
= 25 Ω,
BW
V = 200 mV,
G = 1,
R = 100 Ω
L
I
F
= ±15 V
= ±15 V
R
= 820 Ω,
V = 200 mV,
G = 2,
R = 100 Ω
L
I
F
V
V
CC
R
= 560 Ω,
= ±5 V,
= 820 Ω
CC
R
F
Bandwidth for 0.1 dB flatness
V = 200 mV,
I
G = 1
MHz
MHz
V
R
= ±15 V,
= 680 Ω
CC
40
F
V
V
V
V
= ±15 V,
= ±5 V,
= ±15 V,
= ±5 V,
V
V
V
V
= 20 V
= 4 V
20
35
CC
CC
CC
CC
O(PP)
O(PP)
O
Full power bandwidth (see Note 3)
= 20 V
,
G = 5
G = 2
G = 2
1300
900
70
(PP)
SR
Slew rate
V/µs
= 5 V
,
(PP)
O
t
s
Settling time to 0.1%
0 V to 10 V Step,
ns
noise/distortion performance
PARAMETER
TEST CONDITIONS
MIN
TYP
–65
–79
MAX
UNIT
V
V
= 20 V
= 2 V
V
= ±15 V,
R
= 680 Ω,
F
O(PP)
CC
G = 2,
f = 1 MHz
R = 680 Ω,
F
O(PP)
THD
Total harmonic distortion
dBc
V
= ±5 V,
CC
G = 2,
V
= 2 V
–76
1.7
O(PP)
f = 1 MHz
V
= ±5 V or ±15 V,
f = 10 kHz,
f = 10 kHz,
CC
G = 2,
V
n
Input voltage noise
nV/√Hz
pA/√Hz
Single-ended
= ±5 V or ±15 V,
Positive (IN+)
Negative (IN–)
11.5
16
V
G = 2
CC
I
n
Input noise current
V
CC
V
CC
V
CC
V
CC
= ±5 V
= ±15 V
= ±5 V
= ±15 V
0.04%
0.05%
0.07°
0.08°
G = 2,
NTSC,
40 IRE Modulation
A
Differential gain error
D
R
= 150 Ω,
L
G = 2,
= 150 Ω,
NTSC,
40 IRE Modulation
φ
D
Differential phase error
Crosstalk
R
L
Driver to driver V = 200 mV,
f = 1 MHz
–62
dB
I
5
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
electrical characteristics, V
(continued)
= ±15 V, R = 25 Ω, R = 1 kΩ, T = 25°C (unless otherwise noted)
CC
L
F
A
dc performance
†
PARAMETER
TEST CONDITIONS
MIN
TYP
1.5
5
MAX
UNIT
V
= ±5 V
CC
Open loop transresistance
MΩ
V
V
V
V
= ±15 V
CC
T
= 25°C
2
5
7
A
V
IO
Input offset voltage
= ±5 V or ±15 V
= ±5 V or ±15 V,
= ±5 V or ±15 V
mV
CC
CC
CC
T
A
= full range
= full range
= 25°C
Input offset voltage drift
T
A
20 µV/°C
T
A
1.5
3
4
Differential input offset voltage
mV
5
T
A
= full range
= 25°C
T
A
9
µA
12
Negative
Positive
T
A
= full range
= 25°C
T
4
10
µA
12
A
I
IB
Input bias current
V
V
= ±5 V or ±15 V
= ±5 V or ±15 V,
CC
T
A
= full range
= 25°C
T
A
1.5
8
µA
11
Differential
T
A
= full range
= full range
Differential input offset voltage drift
T
A
10 µV/°C
CC
input characteristics
†
PARAMETER
TEST CONDITIONS
= ±5 V
MIN
TYP
MAX
UNIT
V
V
±3.6
±3.7
CC
V
ICR
Common-mode input voltage range
V
= ±15 V
±13.4 ±13.5
CC
Common-mode rejection ratio
Differential common-mode rejection ratio
Input resistance
62
70
100
300
1.4
CMRR
V
CC
= ±5 V or ±15 V,
T
A
= full range
dB
R
C
kΩ
I
I
Differential input capacitance
pF
output characteristics
†
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
3
to
–2.8
3.2
to
–3
V
CC
V
CC
V
CC
V
CC
= ±5 V
= ±15 V
= ±5 V
= ±15 V
Single ended
Differential
R
R
= 25 Ω
V
L
L
11.8
to
–11.5
12.5
to
–12.2
V
O
Output voltage swing
6
to
–5.6
6.4
to
–6
= 50 Ω
V
23.6
to
–23 –24.4
25
to
V
V
= ±5 V,
R
R
= 5 Ω
500
CC
L
L
I
I
Output current (see Note 2)
mA
O
= ±15 V,
= 25 Ω
400
500
800
13
CC
Short-circuit output current (see Note 2)
Output resistance
mA
OS
R
Open loop
Ω
O
NOTE 2: A heat sink is required to keep the junction temperature below absolute maximum when an output is heavily loaded or shorted. See
absolute maximum ratings and Thermal Information section.
6
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
electrical characteristics, V
= ±15 V, R = 25 Ω, R = 1 kΩ, T = 25°C (unless otherwise noted)
L F A
CC
power supply
†
PARAMETER
TEST CONDITIONS
Split supply
Single supply
MIN
±4.5
9
TYP
MAX
±16.5
33
UNIT
V
CC
Power supply operating range
V
V
= ±5 V
T
= full range
= 25°C
A
12
CC
CC
A
I
Quiescent current (each driver)
Power supply rejection ratio
T
11.5
–74
–72
13
mA
CC
V
= ±15 V
T
A
= full range
= 25°C
15
T
A
–68
–65
–64
–62
V
CC
= ±5 V
dB
dB
T
A
= full range
= 25°C
PSRR
T
A
V
CC
= ±15 V
T
A
= full range
†
Full range is 0°C to 70°C for the THS6012C and –40°C to 85°C for the THS6012I.
PARAMETER MEASUREMENT INFORMATION
1 kΩ
1 kΩ
1 kΩ
1 kΩ
–
+
–
+
Driver 1
Driver 2
V
O
V
O
V
I
V
I
25 Ω
25 Ω
50 Ω
50 Ω
Figure 1. Input-to-Output Crosstalk Test Circuit
R
R
F
G
15 V
–
V
O
+
V
I
R
25 Ω
L
50 Ω
–15 V
Figure 2. Test Circuit, Gain = 1 + (R /R )
F
G
7
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Supply voltage
3
V
V
Peak-to-peak output voltage
O(PP)
vs Load resistance
vs Free-air temperature
vs Free-air temperature
vs Free-air temperature
vs Frequency
4
5
Input offset voltage
Input bias current
IO
I
IB
6
CMRR Common-mode rejection ratio
Input-to-output crosstalk
7
8
PSRR
Power supply rejection ratio
vs Free-air temperature
vs Frequency
9
Closed-loop output impedance
10
11
12
13, 14
vs Supply voltage
vs Free-air temperature
vs Output step
I
Supply current
CC
SR
Slew rate
V
Input voltage noise
vs Frequency
n
15
I
n
Input current noise
vs Frequency
Normalized frequency response
Output amplitude
vs Frequency
16, 17
18–21
22–25
26, 27
28, 29
30, 31
32, 33
34, 35
32, 33
34, 35
36–38
vs Frequency
Normalized output response
Small and large frequency response
vs Frequency
vs Frequency
Single-ended harmonic distortion
Differential gain
vs Output voltage
DC input offset voltage
Number of 150-Ω loads
DC input offset voltage
Number of 150-Ω loads
Differential phase
Output step response
8
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
TYPICAL CHARACTERISTICS
PEAK-TO-PEAK OUTPUT VOLTAGE
PEAK-TO-PEAK OUTPUT VOLTAGE
vs
vs
SUPPLY VOLTAGE
LOAD RESISTANCE
15
10
15
10
5
V
= ±15 V
= ±5 V
CC
5
0
V
CC
T
R
= 25°C
= 1 kΩ
A
F
0
Gain = 1
–5
–10
–15
V
= ±5 V
–5
–10
–15
CC
T
R
R
= 25°C
= 1 kΩ
= 25 Ω
A
F
L
V
CC
= ±15 V
Gain = 1
5
6
7
8
9
10
11 12 13 14 15
10
100
1000
V
– Supply Voltage – V
CC
R
– Load Resistance – Ω
L
Figure 3
Figure 4
INPUT OFFSET VOLTAGE
vs
FREE-AIR TEMPERATURE
INPUT BIAS CURRENT
vs
FREE-AIR TEMPERATURE
2
1
5
4
3
G = 1
= 1 kΩ
V
I
= ±15 V
CC
IB+
G = 1
R
F
R
= 1 kΩ
F
V
CC
= ±5 V
V
= ±5 V
0
CC
IB+
I
–1
–2
–3
2
1
0
V
CC
= ±15 V
V
I
= ±5 V
V
I
= ±15 V
CC
IB–
CC
IB–
–4
–5
–40
–20
0
20
40
60
80
100
–40
–20
0
20
40
60
80
100
T
A
– Free-Air Temperature – °C
T
A
– Free-Air Temperature – °C
Figure 5
Figure 6
9
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
TYPICAL CHARACTERISTICS
COMMON-MODE REJECTION RATIO
INPUT–TO–OUTPUT CROSSTALK
vs
vs
FREE-AIR TEMPERATURE
FREQUENCY
80
75
70
–20
–30
V
R
R
= ± 15 V
= 1 Ω
= 25 Ω
CC
F
L
Gain = 2
V = 200 mV
I
–40
See Figure 2
V
= ±15 V
CC
Driver 1 = Input
Driver 2 = Output
–50
–60
–70
V
= ±5 V
CC
Driver 1 = Output
Driver 2 = Input
1 kΩ
65
60
1 kΩ
1 kΩ
–
+
V
O
V
I
–80
–90
1 kΩ
–40
–20
0
20
40
60
80
100k
1M
10M
f – Frequency – Hz
Figure 8
100M
500M
T
A
– Free-Air Temperature – °C
Figure 7
POWER SUPPLY REJECTION RATIO
CLOSED-LOOP OUTPUT IMPEDANCE
vs
vs
FREE-AIR TEMPERATURE
FREQUENCY
95
90
85
80
75
100
10
1
G = 1
= 1 kΩ
V
= ±15 V
CC
R = 1 kΩ
F
R
F
Gain = 2
= 25°C
T
A
V
I(PP)
= 1 V
V
CC
= 15 V
V
CC
= 5 V
0.1
V
O
V
CC
= –5 V
1 kΩ
1 kΩ
1 kΩ
–
V
I
+
V
CC
= –15 V
THS6012
1000
0.01
70
65
50 Ω
V
I
Z
o
=
– 1
)
(
V
O
0.001
–40
–20
0
20
40
60
80
100
100k
1M
10M
100M
500M
T
A
– Free-Air Temperature – °C
f – Frequency – Hz
Figure 9
Figure 10
10
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
TYPICAL CHARACTERISTICS
SUPPLY CURRENT
vs
SUPPLY VOLTAGE
SUPPLY CURRENT
vs
FREE-AIR TEMPERATURE
12
11
10
13
12
10
8
V
CC
= ±15 V
V
CC
= ±5 V
9
8
7
6
6
4
T
R
= 25°C
= 1 kΩ
A
F
2
0
Gain = +1
5
5
6
7
8
9
10
11 12 13 14 15
–40
–20
0
20
40
60
80
100
±V
– Supply Voltage – V
CC
T
A
– Free-Air Temperature – °C
Figure 11
Figure 12
SLEW RATE
vs
OUTPUT STEP
SLEW RATE
vs
OUTPUT STEP
1500
1300
1100
1000
900
800
700
600
500
400
300
200
100
V
= ± 15V
CC
V
= ± 5V
CC
Gain = 2
Gain = 5
R
R
= 1 kΩ
= 25 Ω
F
L
R
R
= 1 kΩ
= 25 Ω
F
L
+SR
–SR
+SR
–SR
900
700
500
300
100
0
20
5
10
15
0
5
1
2
3
4
Output Step (Peak–To–Peak) – V
Output Step (Peak–To–Peak) – V
Figure 13
Figure 14
11
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
TYPICAL CHARACTERISTICS
INPUT VOLTAGE AND CURRENT NOISE
vs
FREQUENCY
100
10
1
100
10
1
V
T
A
= ±15 V
= 25°C
CC
I
I
Noise
Noise
n–
n+
V
n
Noise
10
100
1k
10k
100k
f – Frequency – Hz
Figure 15
NORMALIZED FREQUENCY RESPONSE
NORMALIZED FREQUENCY RESPONSE
vs
vs
FREQUENCY
FREQUENCY
2
1
2
R
= 360 Ω
F
R
= 300 Ω
F
1
0
0
–1
–2
–3
–1
–2
R
= 510 Ω
F
–3
–4
–5
R
= 750 Ω
F
R
= 470 Ω
F
R
= 1 kΩ
F
–4
–5
–6
R
= 620 Ω
–6
–7
F
V
V
R
= ±15 V
= 200 mV
= 25 Ω
V
= ±15 V
CC
in
L
CC
V = 200 mV
I
R
–8
= 25 Ω
L
Gain = 2
= 25°C
Gain = 1
= 25°C
–7
–8
–9
R
= 1 kΩ
F
T
T
A
A
–10
100K
100
1M
10M
100M
500M
1M
10M
100M
500M
f – Frequency – Hz
f – Frequency – Hz
Figure 16
Figure 17
12
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
TYPICAL CHARACTERISTICS
OUTPUT AMPLITUDE
vs
OUTPUT AMPLITUDE
vs
FREQUENCY
FREQUENCY
3
2
9
8
R
= 620 Ω
F
R = 510 Ω
F
1
0
7
6
5
–1
R
= 1 kΩ
F
R
= 820 Ω
F
–2
–3
–4
–5
–6
4
3
2
1
R
= 1.5 kΩ
F
R
= 1.2 kΩ
F
V
= ± 5 V
CC
Gain = 1
= 25 Ω
V
= ± 5 V
CC
Gain = 2
R = 25 Ω
L
R
L
V = 200 mV
I
V = 200 mV
I
0
100k
100k
1M
10M
100M
500M
1M
10M
100M
500M
f – Frequency – Hz
f – Frequency – Hz
Figure 18
Figure 19
OUTPUT AMPLITUDE
vs
OUTPUT AMPLITUDE
vs
FREQUENCY
FREQUENCY
70
60
70
60
Gain = 1000
Gain = 1000
50
40
30
20
10
0
50
40
30
20
10
0
Gain = 100
Gain = 100
V
R
R
= ± 5 V
=10 Ω
= 25 Ω
= 2 V
CC
G
L
V
R
R
= ± 5 V
=10 Ω
= 25 Ω
= 2 V
CC
G
L
V
O
V
O
–10
100k
–10
100k
1M
10M
100M
500M
1M
10M
100M
500M
f – Frequency – Hz
f – Frequency – Hz
Figure 20
Figure 21
13
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
TYPICAL CHARACTERISTICS
NORMALIZED OUTPUT RESPONSE
NORMALIZED OUTPUT RESPONSE
vs
vs
FREQUENCY
FREQUENCY
1
1
0
R
= 200 Ω
L
0
–1
–2
–3
–4
–1
–2
–3
–4
R
= 100 Ω
L
R
= 50 Ω
L
R
R
= 25 Ω
L
R
= 25 Ω
L
= 200 Ω
= 100 Ω
–5
–6
–7
L
–5
–6
–7
R
L
R
= 50 Ω
L
V
R
= ±15 V
V
R
= ±15 V
CC
= 1 kΩ
CC
= 1 kΩ
F
F
Gain = 2
V = 200 mV
–8
–9
Gain = 1
V = 200 mV
I
–8
–9
I
100k
1M
10M
100M
500M
100k
1M
10M
100M
500M
f – Frequency – Hz
f – Frequency – Hz
Figure 22
Figure 23
NORMALIZED OUTPUT RESPONSE
NORMALIZED OUTPUT RESPONSE
vs
vs
FREQUENCY
FREQUENCY
3
3
R
= 620 Ω
F
R
= 430 Ω
F
2
1
2
1
R
= 820 Ω
F
0
0
–1
–2
–1
–2
R
= 1 kΩ
F
R
= 620 Ω
F
–3
–4
–5
R
= 1 kΩ
F
–3
–4
–5
–6
V
R
= ±15 V
= 100 Ω
V
= ±15 V
R = 100 Ω
L
CC
L
CC
Gain = 1
V = 200 mV
Gain = 2
V = 200 mV
–6
–7
I
I
100k
1M
10M
100M
500M
100k
1M
10M
f – Frequency – Hz
100M
500M
f – Frequency – Hz
Figure 24
Figure 25
14
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
TYPICAL CHARACTERISTICS
SMALL AND LARGE SIGNAL FREQUENCY RESPONSE
–3
SMALL AND LARGE SIGNAL FREQUENCY RESPONSE
3
V = 500 mV
I
V = 500 mV
I
–6
0
–9
–12
–15
–3
V = 250 mV
I
V = 250 mV
I
–6
–9
V = 125 mV
I
V = 125 mV
I
–18
–21
–24
–27
–30
–12
–15
–18
–21
–24
V = 62.5 mV
I
V = 62.5 mV
I
Gain = 1
Gain = 2
V
R
R
= ± 15 V
= 820 Ω
= 25 Ω
V
R
R
= ± 15 V
= 680 Ω
= 25 Ω
CC
F
L
CC
F
L
100k
1M
10M
100M
500M
100k
1M
10M
100M
500M
f – Frequency – Hz
f – Frequency – Hz
Figure 26
Figure 27
SINGLE–ENDED HARMONIC DISTORTION
SINGLE–ENDED HARMONIC DISTORTION
vs
vs
FREQUENCY
FREQUENCY
–40
–40
V
= ± 15 V
V
= ± 5 V
CC
Gain = 2
CC
Gain = 2
R
R
V
= 680 Ω
= 25 Ω
= 2V
–50
–60
R
R
V
= 680 Ω
= 25 Ω
= 2V
F
L
–50
–60
F
L
O(PP)
O(PP)
–70
–80
–70
–80
3rd Harmonic
2nd Harmonic
2nd Harmonic
3rd Harmonic
–90
–90
–100
–100
100k
1M
10M
100k
1M
10M
f – Frequency – Hz
f – Frequency – Hz
Figure 29
Figure 28
15
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
TYPICAL CHARACTERISTICS
SINGLE–ENDED HARMONIC DISTORTION
SINGLE–ENDED HARMONIC DISTORTION
vs
vs
OUTPUT VOLTAGE
OUTPUT VOLTAGE
–50
–60
–70
–80
–50
–60
–70
–80
V
= ± 5 V
CC
Gain = 2
V
= ± 15 V
CC
Gain = 2
R
R
= 680 Ω
= 25 Ω
F
L
R
R
= 680 Ω
= 25 Ω
F
L
f = 1 MHz
f = 1 MHz
2nd Harmonic
3rd Harmonic
2nd Harmonic
–90
–90
3rd Harmonic
15
–100
–100
5
10
20
1
2
3
4
0
0
V
– Output Voltage – V
V
O(PP)
– Output Voltage – V
O(PP)
Figure 30
Figure 31
DIFFERENTIAL GAIN AND PHASE
vs
DC INPUT OFFSET VOLTAGE
0.05
0.10
V
R
R
= ±15 V
= 150 Ω
= 1 kΩ
CC
L
F
Gain
0.04
0.03
0.02
0.01
0
f = 3.58 MHz
Gain = 2
40 IRE Modulation
0.08
0.06
0.04
Phase
0.02
0
–0.7 –0.5
–0.3 –0.1
0.1
0.3
0.5
0.7
DC Input Offset Voltage – V
Figure 32
16
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
TYPICAL CHARACTERISTICS
DIFFERENTIAL GAIN AND PHASE
vs
DC INPUT OFFSET VOLTAGE
0.05
0.10
0.08
0.06
0.04
V
R
R
= ±5 V
= 150 Ω
= 1 kΩ
CC
L
F
0.04
0.03
0.02
0.01
0
f = 3.58 MHz
Gain = 2
40 IRE Modulation
Gain
Phase
0.02
0
–0.7 –0.5
–0.3 –0.1
0.1
0.3
0.5
0.7
DC Input Offset Voltage – V
Figure 33
DIFFERENTIAL GAIN AND PHASE
vs
NUMBER OF 150-Ω LOADS
0.15
0.25
0.20
0.15
0.10
V
R
= ±15 V
CC
= 1 kΩ
F
Gain = 2
0.12
0.09
0.06
0.03
0
f = 3.58 MHz
40 IRE Modulation
100 IRE Ramp
Phase
Gain
0.05
0
1
2
3
4
5
6
7
8
Number of 150-Ω Loads
Figure 34
17
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
TYPICAL CHARACTERISTICS
DIFFERENTIAL GAIN AND PHASE
vs
NUMBER OF 150-Ω LOADS
0.15
0.25
0.20
0.15
0.10
V
R
= ±5 V
= 1 kΩ
CC
F
Gain = 2
0.12
0.09
0.06
0.03
0
f = 3.58 MHz
40 IRE Modulation
100 IRE Ramp
Gain
0.05
0
Phase
6
1
2
3
4
5
7
8
Number of 150-Ω Loads
Figure 35
400-mV STEP RESPONSE
10-V STEP RESPONSE
400
300
200
100
0
8
6
4
2
0
–100
–200
–2
–4
V
= ±15 V
CC
Gain = 2
V
= ±15 V
CC
Gain = 2
R
R
= 25 Ω
= 1 kΩ
L
F
R
R
= 25 Ω
= 1 kΩ
L
F
t /t = 5 ns
r f
–300
–400
–6
–8
t /t = 300 ps
r f
See Figure 3
See Figure 3
0
50 100 150 200 250 300 350 400 450 500
0
50 100 150 200 250 300 350 400 450 500
t – Time – ns
t – Time – ns
Figure 36
Figure 37
18
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
TYPICAL CHARACTERISTICS
20-V STEP RESPONSE
16
12
8
V
= ±15 V
CC
Gain = 5
R
R
= 25 Ω
= 2 kΩ
L
F
t /t = 5 ns
r f
See Figure 3
4
0
–4
–8
–12
–16
0
50 100 150 200 250 300 350 400 450 500
t – Time – ns
Figure 38
APPLICATION INFORMATION
The THS6012 contains two independent operational amplifiers. These amplifiers are current feedback topology
amplifiers made for high-speed operation. They have been specifically designed to deliver the full power
requirements of ADSL and therefore can deliver output currents of at least 400 mA at full output voltage.
The THS6012 is fabricated using Texas Instruments 30-V complementary bipolar process, HVBiCOM. This
process provides excellent isolation and high slew rates that result in the device’s excellent crosstalk and
extremely low distortion.
independent power supplies
Each amplifier of the THS6012 has its own power supply pins. This was specifically done to solve a problem
that often occurs when multiple devices in the same package share common power pins. This problem is
crosstalk between the individual devices caused by currents flowing in common connections. Whenever the
current required by one device flows through a common connection shared with another device, this current,
inconjunctionwiththeimpedanceinthesharedline, producesanunwantedvoltageonthepowersupply. Proper
power supply decoupling and good device power supply rejection helps to reduce this unwanted signal. What
is left is crosstalk.
However, with independent power supply pins for each device, the effects of crosstalk through common
impedance in the power supplies is more easily managed. This is because it is much easier to achieve low
common impedance on the PCB with copper etch than it is to achieve low impedance within the package with
either bond wires or metal traces on silicon.
19
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
APPLICATION INFORMATION
power supply restrictions
Although the THS6012 is specified for operation from power supplies of ±5 V to ±15 V (or singled-ended power
supply operation from 10 V to 30 V), and each amplifier has its own power supply pins, several precautions must
be taken to assure proper operation.
1. The power supplies for each amplifier must be the same value. For example, if the driver 1 uses ±15 volts,
then the driver 2 must also use ±15 volts. Using ±15 volts for one amplifier and ±5 volts for another amplifier
is not allowed.
2. To save power by powering down one of the amplifiers in the package, the following rules must be followed.
•
The amplifier designated driver 1 must always receive power. This is because the internal startup
circuitry uses the power from the driver 1 device.
•
•
The –V
pins from both drivers must always be at the same potential.
CC
Driver 2 is powered down by simply opening the +V
connection.
CC
The THS6012 incorporates a standard Class A-B output stage. This means that some of the quiescent current
is directed to the load as the load current increases. So under heavy load conditions, accurate power dissipation
calculations are best achieved through actual measurements. For small loads, however, internal power
dissipation for each amplifier in the THS6012 can be approximated by the following formula:
V
O
P
D
2 V
I
V
_ V
D
CC CC
CC
O
R
L
Where:
P
V
= Power dissipation for one amplifier
= Split supply voltage
CC
I
V
R
= Supply current for that particular amplifier
= Output voltage of amplifier
= Load resistance
CC
O
L
To find the total THS6012 power dissipation, we simply sum up both amplifier power dissipation results.
Generally, the worst case power dissipation occurs when the output voltage is one-half the V voltage. One
CC
last note, which is often overlooked: the feedback resistor (R ) is also a load to the output of the amplifier and
F
should be taken into account for low value feedback resistors.
device protection features
The THS6012 has two built-in protection features that protect the device against improper operation. The first
protection mechanism is output current limiting. Should the output become shorted to ground the output current
is automatically limited to the value given in the data sheet. While this protects the output against excessive
current, the device internal power dissipation increases due to the high current and large voltage drop across
the output transistors. Continuous output shorts are not recommended and could damage the device.
Additionally, connection of the amplifier output to one of the supply rails (±V ) can cause failure of the device
CC
and is not recommended.
The second built-in protection feature is thermal shutdown. Should the internal junction temperature rise above
approximately 180 C, the device automatically shuts down. Such a condition could exist with improper heat
sinking or if the output is shorted to ground. When the abnormal condition is fixed, the internal thermal shutdown
circuit automatically turns the device back on.
20
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
APPLICATION INFORMATION
thermal information
The THS6012 is packaged in a thermally-enhanced DWP package, which is a member of the PowerPAD family
of packages. This package is constructed using a downset leadframe upon which the die is mounted
[see Figure 39(a) and Figure 39(b)]. This arrangement results in the lead frame being exposed as a thermal pad
on the underside of the package [see Figure 39(c)]. Because this thermal pad has direct thermal contact with
the die, excellent thermal performance can be achieved by providing a good thermal path away from the thermal
pad.
The PowerPAD package allows for both assembly and thermal management in one manufacturing operation.
During the surface-mount solder operation (when the leads are being soldered), the thermal pad can also be
soldered to a copper area underneath the package. Through the use of thermal paths within this copper area,
heat can be conducted away from the package into either a ground plane or other heat dissipating device. This
is discussed in more detail in the PCB design considerations section of this document.
The PowerPAD package represents a breakthrough in combining the small area and ease of assembly of
surface mount with the, heretofore, awkward mechanical methods of heatsinking.
DIE
Thermal
Pad
Side View (a)
DIE
End View (b)
Bottom View (c)
NOTE A: The thermal pad is electrically isolated from all terminals in the package.
Figure 39. Views of Thermally Enhanced DWP Package
21
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
APPLICATION INFORMATION
recommended feedback and gain resistor values
As with all current feedback amplifiers, the bandwidth of the THS6012 is an inversely proportional function of
the value of the feedback resistor. This can be seen from Figures 17 – 20. The recommended resistors with a
±15 V power supply for the optimum frequency response with a 25-Ω load system are 680-Ω for a gain = 1 and
620-Ω for a gain = 2 or –1. Additionally, using a ±5 V power supply, it is recommended that a 1-kΩ feedback
resistor be used for a gain of 1 and a 820-Ω feedback resistor be used for a gain of 2 or –1. These should be
used as a starting point and once optimum values are found, 1% tolerance resistors should be used to maintain
frequency response characteristics. Because there is a finite amount of output resistance of the operational
amplifier, load resistance can play a major part in frequency response. This is especially true with these drivers,
which tend to drive low-impedance loads. This can be seen in Figure 11, Figure 23, and Figure 24. As the load
resistance increases, the output resistance of the amplifier becomes less dominant at high frequencies. To
compensate for this, the feedback resistor should change. For 100-Ω loads, it is recommended that the
feedback resistor be changed to 820 Ω for a gain of 1 and 560 Ω for a gain of 2 or –1. Although, for most
applications, a feedback resistor value of 1 kΩ is recommended, which is a good compromise between
bandwidth and phase margin that yields a very stable amplifier.
Consistent with current feedback amplifiers, increasing the gain is best accomplished by changing the gain
resistor, not the feedback resistor. This is because the bandwidth of the amplifier is dominated by the feedback
resistor value and internal dominant-pole capacitor. The ability to control the amplifier gain independently of the
bandwidth constitutes a major advantage of current feedback amplifiers over conventional voltage feedback
amplifiers. Therefore, once a frequency response is found suitable to a particular application, adjust the value
of the gain resistor to increase or decrease the overall amplifier gain.
Finally, it is important to realize the effects of the feedback resistance on distortion. Increasing the resistance
decreases the loop gain and increases the distortion. It is also important to know that decreasing load
impedance increases total harmonic distortion (THD). Typically, the third order harmonic distortion increases
more than the second order harmonic distortion.
offset voltage
Theoutputoffsetvoltage,(V )isthesumoftheinputoffsetvoltage(V )andbothinputbiascurrents(I )times
OO
IO
IB
the corresponding gains. The following schematic and formula can be used to calculate the output offset
voltage:
R
F
I
IB–
R
G
+
–
+
V
I
V
O
R
S
I
IB+
R
R
R
R
F
F
V
V
1
I
R
1
I
R
OO
IO
IB
S
IB–
F
G
G
Figure 40. Output Offset Voltage Model
22
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
APPLICATION INFORMATION
noise calculations and noise figure
Noise can cause errors on very small signals. This is especially true for the amplifying small signals. The noise
model for current feedback amplifiers (CFB) is the same as voltage feedback amplifiers (VFB). The only
difference between the two is that the CFB amplifiers generally specify different current noise parameters for
each input while VFB amplifiers usually only specify one noise current parameter. The noise model is shown
in Figure 42. This model includes all of the noise sources as follows:
•
•
•
•
e = Amplifier internal voltage noise (nV/√Hz)
n
IN+ = Noninverting current noise (pA/√Hz)
IN– = Inverting current noise (pA/√Hz)
e
= Thermal voltage noise associated with each resistor (e = 4 kTR )
Rx x
Rx
e
Rs
e
n
R
Noiseless
S
+
_
e
ni
e
no
IN+
IN–
e
Rf
R
F
e
Rg
R
G
Figure 41. Noise Model
The total equivalent input noise density (e ) is calculated by using the following equation:
ni
2
2
2
e
e
IN
R
IN–
R
R
4 kTR
4 kT R
R
n
s
ni
S
F
G
F
G
Where:
–23
k = Boltzmann’s constant = 1.380658 × 10
T = Temperature in degrees Kelvin (273 +°C)
R || R = Parallel resistance of R and R
F
G
F
G
To get the equivalent output noise of the amplifier, just multiply the equivalent input noise density (e ) by the
ni
overall amplifier gain (A ).
V
R
R
F
e
e
A
e
1
(Noninverting Case)
no
ni
ni
V
G
23
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
APPLICATION INFORMATION
noise calculations and noise figure (continued)
As the previous equations show, to keep noise at a minimum, small value resistors should be used. As the
closed-loop gain is increased (by reducing R ), the input noise is reduced considerably because of the parallel
G
resistance term. This leads to the general conclusion that the most dominant noise sources are the source
resistor (R ) and the internal amplifier noise voltage (e ). Because noise is summed in a root-mean-squares
S
n
method, noise sources smaller than 25% of the largest noise source can be effectively ignored. This can greatly
simplify the formula and make noise calculations much easier to calculate.
This brings up another noise measurement usually preferred in RF applications, the noise figure (NF). Noise
figure is a measure of noise degradation caused by the amplifier. The value of the source resistance must be
defined and is typically 50 Ω in RF applications.
2
e
ni
NF
10log
2
e
Rs
Because the dominant noise components are generally the source resistance and the internal amplifier noise
voltage, we can approximate noise figure as:
2
2
e
IN
R
n
S
NF
10log 1
4 kTR
S
Figure 42 shows the noise figure graph for the THS6012.
NOISE FIGURE
vs
SOURCE RESISTANCE
20
T
A
= 25°C
18
16
14
12
10
8
6
4
2
0
10
100
1k
10k
R
– Source Resistance – Ω
s
Figure 42. Noise Figure vs Source Resistance
24
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
APPLICATION INFORMATION
driving a capacitive load
Driving capacitive loads with high performance amplifiers is not a problem as long as certain precautions are
taken. The first is to realize that the THS6012 has been internally compensated to maximize its bandwidth and
slew rate performance. When the amplifier is compensated in this manner, capacitive loading directly on the
output will decrease the device’s phase margin leading to high frequency ringing or oscillations. Therefore, for
capacitive loads of greater than 10 pF, it is recommended that a resistor be placed in series with the output of
the amplifier, as shown in Figure 44. A minimum value of 10 Ω should work well for most applications. For
example, in 75-Ω transmission systems, setting the series resistor value to 75 Ω both isolates any capacitance
loading and provides the proper line impedance matching at the source end.
1 kΩ
1 kΩ
_
Input
10 Ω
Output
LOAD
THS6012
+
C
Figure 43. Driving a Capacitive Load
PCB design considerations
Proper PCB design techniques in two areas are important to assure proper operation of the THS6012. These
areas are high-speed layout techniques and thermal-management techniques. Because the THS6012 is a
high-speed part, the following guidelines are recommended.
Ground plane – It is essential that a ground plane be used on the board to provide all components with a
low inductive ground connection. Although a ground connection directly to a terminal of the THS6012 is not
necessarily required, it is recommended that the thermal pad of the package be tied to ground. This serves
two functions. It provides a low inductive ground to the device substrate to minimize internal crosstalk and
it provides the path for heat removal.
Input stray capacitance – To minimize potential problems with amplifier oscillation, the capacitance at the
inverting input of the amplifiers must be kept to a minimum. To do this, PCB trace runs to the inverting input
must be as short as possible, the ground plane must be removed under any etch runs connected to the
inverting input, and external components should be placed as close as possible to the inverting input. This
isespeciallytrueinthenoninvertingconfiguration. AnexampleofthiscanbeseeninFigure44, whichshows
what happens when 1.8 pF is added to the inverting input terminal in the noninverting configuration. The
bandwidth increases dramatically at the expense of peaking. This is because some of the error current is
flowing through the stray capacitor instead of the inverting node of the amplifier. Although, in the inverting
mode, stray capacitance at the inverting input has little effect. This is because the inverting node is at a
virtual ground and the voltage does not fluctuate nearly as much as in the noninverting configuration.
25
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
APPLICATION INFORMATION
PCB design considerations (continued)
NORMALIZED FREQUENCY RESPONSE
vs
FREQUENCY
3
2
V
= ±15 V
CC
V = 200 mV
I
R
R
= 25 Ω
= 1 kΩ
L
F
1
0
Gain = 1
C = 0 pF
I
(Stray C Only)
–1
–2
C = 1.8 pF
I
1 kΩ
–3
–4
–5
C
in
in
V
out
–
V
+
R
25 Ω
=
L
50 Ω
–6
–7
100
1M
10M
f – Frequency – Hz
100M
500M
Figure 44. Driver Normalized Frequency Response vs Frequency
Proper power supply decoupling – Use a minimum of a 6.8-µF tantalum capacitor in parallel with a 0.1-µF
ceramic capacitor on each supply terminal. It may be possible to share the tantalum among several
amplifiers depending on the application, but a 0.1-µF ceramic capacitor should always be used on the
supply terminal of every amplifier. In addition, the 0.1-µF capacitor should be placed as close as possible
tothesupplyterminal. Asthisdistanceincreases, theinductanceintheconnectingetchmakesthecapacitor
less effective. The designer should strive for distances of less than 0.1 inches between the device power
terminal and the ceramic capacitors.
Because of its power dissipation, proper thermal management of the THS6012 is required. Although there are
many ways to properly heatsink this device, the following steps illustrate one recommended approach for a
multilayer PCB with an internal ground plane.
1. Prepare the PCB with a top side etch pattern as shown in Figure 45. There should be etch for the leads as
well as etch for the thermal pad.
2. Place 18 holes in the area of the thermal pad. These holes should be 13 mils in diameter. They are kept
small so that solder wicking through the holes is not a problem during reflow.
3. It is recommended, but not required, to place six more holes under the package, but outside the thermal
pad area. These holes are 25 mils in diameter. They may be larger because they are not in the area to be
soldered so that wicking is not a problem.
4. Connect all 24 holes, the 18 within the thermal pad area and the 6 outside the pad area, to the internal
ground plane.
26
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
APPLICATION INFORMATION
PCB design considerations (continued)
5. When connecting these holes to the ground plane, do not use the typical web or spoke via connection
methodology. Webconnectionshaveahighthermalresistanceconnectionthatisusefulforslowingtheheat
transfer during soldering operations. This makes the soldering of vias that have plane connections easier.
However, in this application, low thermal resistance is desired for the most efficient heat transfer. Therefore,
the holes under the THS6012 package should make their connection to the internal ground plane with a
complete connection around the entire circumference of the plated through hole.
6. The top-side solder mask should leave exposed the terminals of the package and the thermal pad area with
its five holes. The four larger holes outside the thermal pad area, but still under the package, should be
covered with solder mask.
7. Apply solder paste to the exposed thermal pad area and all of the operational amplifier terminals.
8. With these preparatory steps in place, the THS6012 is simply placed in position and run through the solder
reflow operation as any standard surface-mount component. This results in a part that is properly installed.
Addition 6 vias outside of thermal pad area
but under the package
(Via diameter = 25 mils)
Thermal pad area (0.19 x 0.21) with 18 vias
(Via diameter = 13 mils)
Figure 45. PowerPAD PCB Etch and Via Pattern
The actual thermal performance achieved with the THS6012 in its PowerPAD package depends on the
application. In the previous example, if the size of the internal ground plane is approximately 3 inches × 3 inches,
then the expected thermal coefficient, θ , is about 21.5 C/W. For a given θ , the maximum power dissipation
JA
JA
is shown in Figure 46 and is calculated by the following formula:
T
–T
MAX
A
P
D
JA
Where:
P
= Maximum power dissipation of THS6012 (watts)
= Absolute maximum junction temperature (150°C)
= Free-ambient air temperature (°C)
D
T
MAX
T
A
θ
= θ + θ
JA
JC CA
θ
θ
= Thermal coefficient from junction to case (0.37°C/W)
= Thermal coefficient from case to ambient
JC
CA
27
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
APPLICATION INFORMATION
PCB design considerations (continued)
More complete details of the PowerPAD installation process and thermal management techniques can be found
in the Texas Instruments Technical Brief, PowerPAD Thermally Enhanced Package. This document can be
found at the TI web site (www.ti.com) by searching on the key word PowerPAD. The document can also be
ordered through your local TI sales office. Refer to literature number SLMA002 when ordering.
MAXIMUM POWER DISSIPATION
vs
FREE-AIR TEMPERATURE
9
T
J
= 150°C
PCB Size = 3” x 3”
No Air Flow
8
7
6
θ
= 21.5°C/W
JA
2 oz Trace and
Copper Pad
with Solder
5
4
3
2
θ
= 43.9°C/W
JA
2 oz Trace and Copper Pad
without Solder
1
0
–40
–20
0
20
40
60
80
100
T
A
– Free-Air Temperature – °C
Figure 46. Maximum Power Dissipation vs Free-Air Temperature
28
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
APPLICATION INFORMATION
ADSL
TheTHS6012wasprimarilydesignedasalinedriverandlinereceiverforADSL(asymmetricaldigitalsubscriber
line). The driver output stage has been sized to provide full ADSL power levels of 20 dBm onto the telephone
lines. Although actual driver output peak voltages and currents vary with each particular ADSL application, the
THS6012 is specified for a minimum full output current of 400 mA at its full output voltage of approximately 12
V. ThisperformancemeetsthedemandingneedsofADSLatthecentralofficeendofthetelephoneline. Atypical
ADSL schematic is shown in Figure 47.
15 V
+
THS6012
Driver 1
0.1 µF
6.8 µF
12.5 Ω
+
_
V
I+
1:2
1 kΩ
Telephone Line
100 Ω
1 kΩ
0.1 µF
6.8 µF
+
–15 V
15 V
1 kΩ
15 V
+
2 kΩ
THS6012
Driver 2
0.1 µF
6.8 µF
0.1 µF
12.5 Ω
1 kΩ
+
_
V
I–
–
V
O+
+
THS6062
1 kΩ
–15 V
1 kΩ
1 kΩ
0.1 µF
6.8 µF
+
15 V
–15 V
2 kΩ
1 kΩ
0.1 µF
–
+
V
O–
THS6062
0.01 µF
–15 V
Figure 47. THS6012 ADSL Application
29
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
APPLICATION INFORMATION
ADSL (continued)
The ADSL transmit band consists of 255 separate carrier frequencies each with its own modulation and
amplitude level. With such an implementation, it is imperative that signals put onto the telephone line have as
low a distortion as possible. This is because any distortion either interferes directly with other ADSL carrier
frequencies or it creates intermodulation products that interfere with ADSL carrier frequencies.
The THS6012 has been specifically designed for ultra low distortion by careful circuit implementation and by
taking advantage of the superb characteristics of the complementary bipolar process. Driver single-ended
distortion measurements are shown in Figures 29 – 32. It is commonly known that in the differential driver
configuration, the second order harmonics tend to cancel out. Thus, the dominant total harmonic distortion
(THD)willbeprimarilyduetothethirdorderharmonics. Fortheseteststheloadwas25Ω. Additionally, distortion
should be reduced as the feedback resistance drops. This is because the bandwidth of the amplifier increases,
which allows the amplifier to react faster to any nonlinearities in the closed-loop system.
Another significant point is the fact that distortion decreases as the impedance load increases. This is because
the output resistance of the amplifier becomes less significant as compared to the output load resistance.
general configurations
A common error for the first-time CFB user is to create a unity gain buffer amplifier by shorting the output directly
to the inverting input. A CFB amplifier in this configuration oscillates and is not recommended. The THS6012,
like all CFB amplifiers, must have a feedback resistor for stable operation. Additionally, placing capacitors
directly from the output to the inverting input is not recommended. This is because, at high frequencies, a
capacitor has a very low impedance. This results in an unstable amplifier and should not be considered when
using a current-feedback amplifier. Because of this, integrators and simple low-pass filters, which are easily
implemented on a VFB amplifier, have to be designed slightly differently. If filtering is required, simply place an
RC-filter at the noninverting terminal of the operational-amplifier (see Figure 49).
R
R
F
G
V
R
R
–
O
F
1
1
V
O
V
1
sR1C1
+
I
G
V
I
R1
C1
1
f
–3dB
2 R1C1
Figure 48. Single-Pole Low-Pass Filter
If a multiple pole filter is required, the use of a Sallen-Key filter can work very well with CFB amplifiers. This is
because the filtering elements are not in the negative feedback loop and stability is not compromised. Because
oftheirhighslew-ratesandhighbandwidths, CFBamplifierscancreateveryaccuratesignalsandhelpminimize
distortion. An example is shown in Figure 50.
30
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
APPLICATION INFORMATION
general configurations (continued)
C1
R1 = R2 = R
C1 = C2 = C
Q = Peaking Factor
(Butterworth Q = 0.707)
+
_
V
I
1
2 RC
R1
R2
f
–3dB
C2
R
F
1
R
=
G
R
F
2 –
)
(
R
Q
G
Figure 49. 2-Pole Low-Pass Sallen-Key Filter
There are two simple ways to create an integrator with a CFB amplifier. The first one shown in Figure 51 adds
a resistor in series with the capacitor. This is acceptable because at high frequencies, the resistor is dominant
and the feedback impedance never drops below the resistor value. The second one shown in Figure 52 uses
positive feedback to create the integration. Caution is advised because oscillations can occur because of the
positive feedback.
C1
R
F
R
G
1
S
–
+
V
I
R C1
V
R
R
F
O
F
V
O
V
S
I
G
THS6012
Figure 50. Inverting CFB Integrator
R
R
F
G
For Stable Operation:
R
R
R2
F
≥
R1 || R
–
+
G
A
THS6012
V
O
R
F
1 +
R
V
O
V
I
G
)
(
sR1C1
R1
R2
V
I
C1
R
A
Figure 51. Non-Inverting CFB Integrator
31
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
APPLICATION INFORMATION
general configurations (continued)
Another good use for the THS6012 amplifiers is as very good video distribution amplifiers. One characteristic
of distribution amplifiers is the fact that the differential phase (DP) and the differential gain (DG) are
compromised as the number of lines increases and the closed-loop gain increases. Be sure to use termination
resistors throughout the distribution system to minimize reflections and capacitive loading.
620 Ω
620 Ω
75 Ω Transmission Line
75 Ω
–
+
V
O1
V
I
THS6012
75 Ω
75 Ω
N Lines
75 Ω
V
ON
75 Ω
Figure 52. Video Distribution Amplifier Application
evaluation board
AnevaluationboardisavailablefortheTHS6012(literaturenumberSLOP132). Thisboardhasbeenconfigured
for proper thermal management of the THS6012. The circuitry has been designed for a typical ADSL application
as shown previously in this document. For more detailed information, refer to the THS6012EVM User’s Manual
(literature number SLOU034). To order the evaluation board contact your local TI sales office or distributor.
32
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
MECHANICAL INFORMATION
DWP (R-PDSO-G20)
PowerPAD PLASTIC SMALL-OUTLINE PACKAGE
0.020 (0,51)
0.014 (0,35)
0.010 (0,25)
M
0.050 (1,27)
20
11
Thermal Pad 0.150 (3,81)
(see Note C)
0.170 (4,31) NOM
0.299 (7,59)
0.293 (7,45)
0.430 (10,92)
0.411 (10,44)
0.010 (0,25) NOM
1
10
0.510 (12,95)
0.500 (12,70)
Gage Plane
0.010 (0,25)
+2°–8°
0.050 (1,27)
0.016 (0,40)
Seating Plane
0.004 (0,10)
0.004 (0,10)
0.000 (0,00)
0.096 (2,43) MAX
4073226/B 01/96
NOTES: A. All linear dimensions are in inches (millimeters).
B. This drawing is subject to change without notice.
C. The thermal performance may be enhanced by bonding the thermal pad to an external thermal plane.
PowerPAD is a trademark of Texas Instruments Incorporated.
33
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
MECHANICAL DATA
GQE (S-PLGA-N80)
PLASTIC LAND GRID ARRAY
5,20
4,80
SQ
4,00 TYP
0,50
J
H
G
F
E
D
C
B
A
1
2
3
4
5
6
7
8
9
0,93
0,87
1,00 MAX
Seating Plane
0,08
0,33
0,23
0,05
M
0,08 MAX
4200461/A 10/99
NOTES: A. All linear dimensions are in millimeters.
B. This drawing is subject to change without notice.
C. MicroStar Junior LGA configuration
MicroStar Junior LGA is a trademark of Texas Instruments Incorporated.
34
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
IMPORTANT NOTICE
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Copyright 2000, Texas Instruments Incorporated
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