TPA0312PWPR
更新时间：2024-09-18 18:40:38
品牌：TI
描述：Stereo 2-W Audio Power Amp with 4 Selectable Gain Settings and MUX Control 24-HTSSOP -40 to 85
概述
规格参数
数据手册
替代型号TPA0312PWPR 概述
Stereo 2-W Audio Power Amp with 4 Selectable Gain Settings and MUX Control 24-HTSSOP -40 to 85 音频放大器 音频控制集成电路
TPA0312PWPR 规格参数
| 是否Rohs认证： | 符合 | 生命周期： | Obsolete |
| 零件包装代码： | TSSOP | 包装说明： | GREEN, PLASTIC, HTSSOP-24 |
| 针数： | 24 | Reach Compliance Code： | compliant |
| ECCN代码： | EAR99 | HTS代码： | 8542.39.00.01 |
| Factory Lead Time： | 1 week | 风险等级： | 5.47 |
| Is Samacsys： | N | 商用集成电路类型： | VOLUME CONTROL CIRCUIT |
| 谐波失真： | 1% | JESD-30 代码： | R-PDSO-G24 |
| JESD-609代码： | e4 | 长度： | 7.8 mm |
| 湿度敏感等级： | 2 | 频带数量： | 1 |
| 信道数量： | 2 | 功能数量： | 1 |
| 端子数量： | 24 | 最高工作温度： | 85 °C |
| 最低工作温度： | -40 °C | 封装主体材料： | PLASTIC/EPOXY |
| 封装代码： | HTSSOP | 封装等效代码： | TSSOP24,.25 |
| 封装形状： | RECTANGULAR | 封装形式： | SMALL OUTLINE, HEAT SINK/SLUG, THIN PROFILE, SHRINK PITCH |
| 峰值回流温度（摄氏度）： | 260 | 电源： | 5 V |
| 认证状态： | Not Qualified | 座面最大高度： | 1.2 mm |
| 子类别： | Audio/Video Amplifiers | 最大压摆率： | 10 mA |
| 最大供电电压 (Vsup)： | 5.5 V | 最小供电电压 (Vsup)： | 4.5 V |
| 表面贴装： | YES | 技术： | CMOS |
| 温度等级： | INDUSTRIAL | 端子面层： | Nickel/Palladium/Gold (Ni/Pd/Au) |
| 端子形式： | GULL WING | 端子节距： | 0.65 mm |
| 端子位置： | DUAL | 处于峰值回流温度下的最长时间： | NOT SPECIFIED |
| 宽度： | 4.4 mm | Base Number Matches： | 1 |
TPA0312PWPR 数据手册
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PDF下载TPA0312
www.ti.com
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
2.6-W STEREO AUDIO POWER AMPLIFIER WITH FOUR
SELECTABLE GAIN SETTINGS AND MUX CONTROL
FEATURES
PWP PACKAGE
•
Compatible With PC 99 Desktop Line-Out Into
(TOP VIEW)
10-kΩ Load
1
24
23
22
21
20
19
18
17
16
15
14
13
GND
GAIN0
GAIN1
LOUT+
LLINEIN
LHPIN
GND
•
Internal Gain Control, Which Eliminates
External Gain-Setting Resistors
2
RLINEIN
SHUTDOWN
ROUT+
RHPIN
3
•
•
•
•
•
•
•
•
2.6-W/Ch Output Power Into 3-Ω Load
Input MUX Select Terminal
PC-Beep Input
4
5
6
V
DD
7
PV
PV
DD
DD
Depop Circuitry
8
RIN
LOUT–
LIN
BYPASS
GND
HP/LINE
ROUT–
SE/BTL
PC-BEEP
GND
Stereo Input MUX
9
10
11
12
Fully Differential Input
Low Supply Current and Shutdown Current
Surface-Mount Power Packaging 24-Pin
TSSOP PowerPAD™
DESCRIPTION
The TPA0312 is a stereo audio power amplifier in a 24-pin TSSOP thermally enhanced package capable of
delivering 2.6 W of continuous RMS power per channel into 3-Ω loads. This device minimizes the number of
external components needed, simplifying the design, and freeing up board space for other features. When driving
1 W into 8-Ω speakers, the TPA0312 has less than 0.65% THD+N across its specified frequency range. Included
within this device is integrated depop circuitry that virtually eliminates transients that cause noise in the speakers.
Amplifier gain is internally configured and controlled by way of two terminals (GAIN0 and GAIN1). BTL gain
settings of 6 dB, 10 dB, 15.6 dB, and 21.6 dB (inverting) are provided, whereas SE gain is always configured as
4.1 dB for headphone drive. An internal input MUX allows two sets of stereo inputs to the amplifier. The HP/LINE
terminal allows the user to select which MUX input is active, regardless of whether the amplifier is in SE or BTL
mode. In notebook applications, where internal speakers are driven as BTL and the line outputs (often
headphone drive) are required to be SE, the TPA0312 automatically switches into SE mode when the SE/BTL
input is activated, and this reduces the gain to 4.1 dB.
The TPA0312 consumes only 6 mA of supply current during normal operation. A miserly shutdown mode
reduces the supply current to 150 µA.
The PowerPAD™ package (PWP) delivers a level of thermal performance that was previously achievable only in
TO-220-type packages. Thermal impedances of approximately 35°C/W are readily realized in multilayer PCB
applications. This allows the TPA0312 to operate at full power into 8-Ω loads at an ambient temperature of 85°C.
AVAILABLE OPTIONS
PACKAGED DEVICE
TSSOP(1) (PWP)
TA
–40°C to 85°C
TPA0312PWP
(1) The PWP package is available taped and reeled. To order a taped
and reeled part, add the suffix R to the part number (e.g.,
TPA0312PWPR).
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Copyright © 2000–2004, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
TPA0312
www.ti.com
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
These devices have limited built-in ESD protection. The leads should be shorted together or the device
placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates.
FUNCTIONAL BLOCK DIAGRAM
RHPIN
R
MUX
Volume
Control
RLINEIN
−
+
GAIN0
GAIN1
ROUT+
Volume
Control
RIN
−
+
ROUT−
PC
Beep
PC-BEEP
SE/BTL
PV
DD
MUX
Control
Power
Management
Depop
Circuitry
V
DD
HP/LINE
BYPASS
SHUTDOWN
LHPIN
L
MUX
Volume
Control
GND
LLINEIN
−
+
LOUT+
Volume
Control
LIN
−
+
LOUT−
2
TPA0312
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SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
TERMINAL FUNCTIONS
TERMINAL
I/O
DESCRIPTION
NAME
NO.
11
2
BYPASS
GAIN0
Tap to voltage divider for internal mid-supply bias generator
Bit 0 of gain control
I
I
GAIN1
3
Bit 1 of gain control
1, 12, 13,
24
GND
Ground connection for circuitry. Connected to the thermal pad.
LHPIN
LIN
6
10
5
I
I
Left-channel headphone input, selected when SE/BTL is held high
Common left input for fully differential input. AC ground for single-ended inputs.
Left-channel line input, selected when SE/BTL is held low
LLINEIN
LOUT+
LOUT-
I
4
O
O
Left-channel positive output in BTL mode and positive output in SE mode
Left-channel negative output in BTL mode and high-impedance in SE mode
9
The input for PC Beep mode. PC-BEEP is enabled when a > 1.5-V (peak-to-peak) square wave is input
to PC-BEEP
PC-BEEP
14
I
HP/LINE is the input MUX control input. When the HP/LINE terminal is held high, the headphone inputs
(LHPIN or RHPIN [6, 20]) are active. When the HP/LINE terminal is held low, the line inputs (LLINEIN or
RLINEIN [5, 23]) are active.
HP/LINE
17
I
PVDD
7, 18
20
8
I
I
Power supply for output stage
RHPIN
RIN
Right-channel headphone input, selected when SE/BTL is held high
Common right input for fully differential input. AC ground for single-ended inputs.
Right-channel line input, selected when SE/BTL is held low
I
RLINEIN
ROUT+
ROUT-
SHUTDOWN
SE/BTL
VDD
23
21
16
22
15
19
I
O
O
I
Right-channel positive output in BTL mode and positive output in SE mode
Right-channel negative output in BTL mode and high-impedance in SE mode
Places entire IC in shutdown mode when held low, except PC-BEEP remains active
Hold SE/BTL low for BTL mode and hold high for SE mode.
I
I
Analog VDD input supply. This terminal needs to be isolated from PVDD to achieve highest performance.
Connect to ground. Must be soldered down in all applications to properly secure the device on the PC
board.
Thermal Pad
3
TPA0312
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SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted)(1)
VDD
VI
Supply voltage
6 V
Input voltage
–0.3 V to VDD +0.3 V
Continuous total power dissipation
Operating free-air temperature range
Operating junction temperature range
Storage temperature range
Internally limited (see Dissipation Rating Table)
TA
–40°C to 85°C
–40°C to 150°C
–65°C to 85°C
260°C
TJ
Tstg
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds
(1) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
TA≤ 25°C
DERATING FACTOR
TA = 70°C
TA = 85°C
PWP
2.7 W(1)
21.8 mW/°C
1.7 W
1.4 W
(1) See the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments
Recommended Board for PowerPAD of the before-mentioned document.
RECOMMENDED OPERATING CONDITIONS
MIN
4.5
MAX
UNIT
VDD
VIH
Supply voltage
5.5
V
SE/BTL, HP/LINE, GAIN0, GAIN1
SHUTDOWN
0.8 x VDD
2
High-level input voltage
V
SE/BTL, HP/LINE
GAIN0, GAIN1
0.6 x VDD
0.4 x VDD
0.8
VIL
TA
Low-level input voltage
V
SHUTDOWN
Operating free-air temperature
–40
85
°C
ELECTRICAL CHARACTERISTICS
at specified free-air temperature, VDD = 5 V, TA = 25°C (unless otherwise noted)
PARAMETER
Output offset voltage (measured differentially)
Power supply rejection ratio
TEST CONDITIONS
MIN
TYP
MAX UNIT
|VOO
PSRR
|IIH
|
VI = 0, Av = 6 dB
VDD = 4.5 V to 5.5 V
VDD = 5.5 V, VI = VDD
VDD = 5.5 V, VI = 0 V
BTL mode
25
mV
dB
µA
µA
77
|
High-level input current
1
1
|IIL|
Low-level input current
6
3
10
5
IDD
Supply current
mA
µA
SE mode
IDD(SD)
Supply current, shutdown mode
150
300
4
TPA0312
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SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
OPERATING CHARACTERISTICS
VDD = 5 V, TA = 25°C, RL = 8 Ω , Gain = 6 dB, BTL mode
PARAMETER
TEST CONDITIONS
THD + N= 10%
MIN
TYP
2.6
MAX
UNIT
PO
Output power
RL = 3 Ω
W
THD + N = 1%,
2.05
0.65%
>15
72
THD + N Total harmonic distortion plus noise
PO = 1 W,
f = 20 Hz to 15 kHz
BOM
Maximum output power bandwidth
Supply ripple rejection ratio
Signal-to-noise ratio
THD = 5%
kHz
dB
f = 1 kHz, CB = 0.47 µF
BTL mode
SNR
Vn
105
20
dB
BTL mode
SE mode
CB = 0.47 µF, f = 20 Hz
to 20 kHz
Noise output voltage
Input impedance
µVRMS
18
ZI
See Table 1
TYPICAL CHARACTERISTICS
TABLE OF GRAPHS
FIGURE
1, 4-6, 9-11,
14-16, 18
vs Output power
vs Frequency
THD+N
Vn
Total harmonic distortion plus noise
2, 3, 7, 8, 12,
13, 17, 19
vs Output voltage
vs Bandwidth
vs Frequency
vs Frequency
vs Frequency
vs Frequency
20
21
Output noise voltage
Supply ripple rejection ratio
Crosstalk
22, 23
24, 25
26
Shutdown attenuation
Signal-to-noise ratio
Closed-loop response
Output power
SNR
27
28-30
31, 32
33, 34
35
PO
PD
vs Load resistance
vs Output power
Power dissipation
vs Ambient temperature
5
TPA0312
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SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
TOTAL HARMONIC DISTORTION PLUS NOISE
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
vs
OUTPUT POWER
FREQUENCY
10%
1%
10%
P
R
BTL
= 1.75 W
= 3 Ω
A
= 6 dB
O
V
f = 1 kHz
BTL
L
R
L
= 4 Ω
1%
0.1%
R
L
= 8 Ω
R
L
= 3 Ω
A
= 21.6 dB
A = 6 dB
V
V
0.1%
A
V
= 15.6 dB
1k
0.01%
20
0.01%
0.5 0.75
1
1.25 1.5 1.75
2
2.25 2.5 2.75
3
100
10k 20k
P
O
− Output Power − W
f − Frequency − Hz
Figure 1.
Figure 2.
TOTAL HARMONIC DISTORTION PLUS NOISE
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
vs
FREQUENCY
OUTPUT POWER
10%
1%
10%
1%
R
= 3 Ω
= 6 dB
L
A
V
BTL
f = 15 kHz
f = 1 kHz
P
O
= 1.0 W
P
O
= 0.5 W
0.1%
0.1%
f = 20 Hz
R
L
= 3 Ω
A
BTL
= 6 dB
V
P
O
= 1.75 W
0.01%
20
0.01%
0.01
0.1
1
10
100
1k
10k 20k
P
O
− Output Power − W
f − Frequency − Hz
Figure 3.
Figure 4.
6
TPA0312
www.ti.com
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
TOTAL HARMONIC DISTORTION PLUS NOISE
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
vs
OUTPUT POWER
OUTPUT POWER
10%
1%
10%
1%
f = 15 kHz
f = 1 kHz
f = 15 kHz
f = 1 kHz
f = 20 Hz
f = 20 Hz
0.1%
0.1%
R
A
= 3 Ω
= 21.6 dB
R
A
= 3 Ω
= 15.6 dB
L
L
V
V
BTL
BTL
0.01%
0.01%
0.01
0.1
1
10
0.01
0.1
1
10
P
O
− Output Power − W
P
O
− Output Power − W
Figure 5.
Figure 6.
TOTAL HARMONIC DISTORTION PLUS NOISE
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
vs
FREQUENCY
FREQUENCY
10%
1%
10%
1%
R
A
BTL
= 4 Ω
= 6 dB
P
R
BTL
= 1.75 W
= 3 Ω
L
O
V
L
P
O
= 1.5 W
A
V
= 21.6 dB
A
V
= 6 dB
P
O
= 0.25 W
0.1%
0.1%
P
O
= 1.0 W
A
V
= 15.6 dB
1k
0.01%
20
0.01%
20
100
1k
10k 20k
100
10k 20k
f − Frequency − Hz
f − Frequency − Hz
Figure 7.
Figure 8.
7
TPA0312
www.ti.com
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
TOTAL HARMONIC DISTORTION PLUS NOISE
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
vs
OUTPUT POWER
OUTPUT POWER
10%
1%
10%
R
= 4 Ω
= 6 dB
L
A
V
BTL
f = 15 kHz
f = 15 kHz
f = 1 kHz
1%
f = 1 kHz
f = 20 Hz
0.1%
0.1%
f = 20 Hz
R
L
= 4 Ω
A
V
= 15.6 dB
BTL
0.01%
0.01%
0.01
0.1
1
10
0.01
0.1
1
10
P
O
− Output Power − W
P
O
− Output Power − W
Figure 9.
Figure 10.
TOTAL HARMONIC DISTORTION PLUS NOISE
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
vs
OUTPUT POWER
FREQUENCY
10%
1%
10%
1%
R
= 8 Ω
= 6 dB
L
A
V
f = 15 kHz
BTL
f = 1 kHz
f = 20 Hz
0.1%
0.1%
P
O
= 0.25 W
P
O
= 1.0 W
R
L
= 4 Ω
A
V
= 21.6 dB
BTL
P
O
= 0.5 W
1k
0.01%
20
0.01%
0.01
0.1
1
10
100
10k 20k
P
O
− Output Power − W
f − Frequency − Hz
Figure 11.
Figure 12.
8
TPA0312
www.ti.com
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
TOTAL HARMONIC DISTORTION PLUS NOISE
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
vs
FREQUENCY
OUTPUT POWER
10%
1%
10%
1%
P
R
BTL
= 1 W
= 8 Ω
R
A
BTL
= 8 Ω
= 6 dB
O
L
L
V
f = 15 kHz
f = 1 kHz
A
= 21.6 dB
V
A
V
= 6 dB
0.1%
0.1%
A
V
= 15.6 dB
f = 20 Hz
0.01%
20
0.01%
0.01
0.1
1
10
100
1k
10k 20k
P
O
− Output Power − W
f − Frequency − Hz
Figure 13.
Figure 14.
TOTAL HARMONIC DISTORTION PLUS NOISE
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
vs
OUTPUT POWER
OUTPUT POWER
10%
1%
10%
1%
R
= 8 Ω
= 15.6 dB
L
A
V
BTL
f = 15 kHz
f = 15 kHz
f = 1 kHz
f = 1 kHz
f = 20 Hz
0.1%
0.1%
f = 20 Hz
R
L
= 8 Ω
A
V
= 21.6 dB
BTL
0.01%
0.01%
0.01
0.1
1
10
0.01
0.1
1
10
P
O
− Output Power − W
P
O
− Output Power − W
Figure 15.
Figure 16.
9
TPA0312
www.ti.com
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
TOTAL HARMONIC DISTORTION PLUS NOISE
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
vs
FREQUENCY
OUTPUT POWER
10%
10%
1%
R
A
= 32 Ω
= 4.1 dB
R
A
SE
= 32 Ω
= 4.1 dB
L
L
V
V
SE
1%
f = 15 kHz
P
O
= 25 mW
0.1%
0.1%
f = 1 kHz
f = 20 Hz
P
O
= 50 mW
P
O
= 75 mW
0.01%
0.01%
0.01
0.1
1
20
100
1k
10k 20k
P
O
− Output Power − W
f − Frequency − Hz
Figure 17.
Figure 18.
TOTAL HARMONIC DISTORTION PLUS NOISE
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
vs
FREQUENCY
OUTPUT VOLTAGE
10%
1%
10%
1%
R
= 10 kΩ
= 4.1 dB
R
= 10 kΩ
= 4.1 dB
L
L
A
A
V
V
SE
SE
0.1%
0.1%
f = 20 Hz
f = 15 kHz
V
O
= 1 V
RMS
0.01%
0.01%
f = 1 kHz
0.001%
0.001%
20
100
1k
10k 20k
0.1
1
3
f − Frequency − Hz
V
O
− Output Voltage − V
RMS
Figure 19.
Figure 20.
10
TPA0312
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SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
OUTPUT NOISE VOLTAGE
SUPPLY RIPPLE REJECTION RATIO
vs
vs
BANDWIDTH
FREQUENCY
0
−20
−40
−60
−80
100
V
= 5 V
R
C
= 8 Ω
= 0.47 µF,
= 6 dB
DD
L
90
80
70
60
50
40
30
20
R = 4Ω
L
B
A
V
BTL
A
= 21.6 dB
V
A
V
= 15.6 dB
−100
−120
10
0
A
V
= 6 dB
20
100
1k
10k 20k
10
100
1k
10k
f − Frequency − Hz
BW − Bandwidth − Hz
Figure 21.
Figure 22.
SUPPLY RIPPLE REJECTION RATIO
CROSSTALK
vs
FREQUENCY
vs
FREQUENCY
0
0
−20
−40
−60
−80
P
O
= 1 W
R
C
= 32 Ω
= 0.47 µF,
=4.1 dB
L
R
L
= 8 Ω
B
A = 6 dB
v
BTL
−20
A
V
SE
−40
−60
−80
LEFT TO RIGHT
RIGHT TO LEFT
−100
−120
−100
−120
20
100
1k
10k 20k
20
100
1k
10k 20k
f − Frequency − Hz
f − Frequency − Hz
Figure 23.
Figure 24.
11
TPA0312
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SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
CROSSTALK
vs
FREQUENCY
SHUTDOWN ATTENUATION
vs
FREQUENCY
0
0
−20
−40
−60
−80
V
R
= 1 V
= 10 kΩ
V = 1 V
I RMS
O
RMS
L
A = 4.1 dB
v
SE
−20
−40
−60
−80
R
L
= 10 kΩ, SE
R
L
= 32 Ω, SE
LEFT TO RIGHT
RIGHT TO LEFT
−100
−120
−100
−120
R
L
= 8 Ω, BTL
20
100
1k
10k 20k
20
100
1k
10k 20k
f − Frequency − Hz
f − Frequency − Hz
Figure 25.
Figure 26.
SIGNAL-TO-NOISE RATIO
vs
FREQUENCY
140
130
P
R
BTL
= 1 W
= 8 Ω
O
L
A
V
= 15.6 dB
120
110
A
V
= 6 dB
100
90
A
V
= 21.6 dB
80
70
60
20
100
1k
10k 20k
f − Frequency − Hz
Figure 27.
12
TPA0312
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
180°
www.ti.com
CLOSED-LOOP RESPONSE
10
7.5
Gain
90°
5
2.5
Phase
0°
0
−2.5
−5
R
= 8 Ω
= 6 dB
L
−90°
−180°
A
V
BTL
−7.5
−10
10
100
1k
10k
100k
1M
f − Frequency − Hz
Figure 28.
CLOSED-LOOP RESPONSE
180°
90°
30
25
20
15
Gain
Phase
0°
10
5
R
= 8 Ω
= 15.6 dB
L
−90°
−180°
0
A
V
BTL
−5
−10
10
100
1k
10k
100k
1M
f − Frequency − Hz
Figure 29.
13
TPA0312
www.ti.com
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
CLOSED-LOOP RESPONSE
180°
90°
30
25
Gain
20
15
Phase
0°
10
5
R
= 8 Ω
= 21.6 dB
L
−90°
−180°
0
A
V
BTL
−5
−10
10
100
1k
10k
100k
1M
f − Frequency − Hz
Figure 30.
OUTPUT POWER
vs
LOAD RESISTANCE
OUTPUT POWER
vs
LOAD RESISTANCE
3.5
3
1500
A
BTL
= 6 dB
A = 4.1 dB
V
SE
V
1250
1000
750
2.5
2
10% THD+N
1.5
1
10% THD+N
500
1% THD+N
250
0
0.5
0
1% THD+N
16
0
8
16
24
32
40
48
56
64
0
8
24
32
40
48
56
64
R
L
− Load Resistance − Ω
R
L
− Load Resistance − Ω
Figure 31.
Figure 32.
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TPA0312
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SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
POWER DISSIPATION
vs
OUTPUT POWER
POWER DISSIPATION
vs
OUTPUT POWER
1.8
1.6
0.4
0.35
0.3
3 Ω
1.4
1.2
1
4 Ω
4 Ω
0.25
0.2
0.8
0.6
0.15
0.1
8 Ω
8 Ω
0.4
0.2
f = 1 kHz
BTL
Each Channel
f = 1 kHz
SE
Each Channel
32 Ω
0.05
0
0
0
0.5
1
1.5
2
2.5
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7 0.8
P
O
− Output Power − W
P
O
− Output Power − W
Figure 33.
Figure 34.
POWER DISSIPATION
vs
AMBIENT TEMPERATURE
7
6
Θ
JA1
Θ
JA2
Θ
JA3
Θ
JA4
= 45.9°C/W
= 45.2°C/W
= 31.2°C/W
= 18.6°C/W
Θ
JA4
5
4
Θ
Θ
JA3
3
2
JA1,2
1
0
−40 −20
0
20 40 60 80 100 120 140 160
T
A
− Ambient Temperature − °C
Figure 35.
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TPA0312
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SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
THERMAL INFORMATION
The thermally enhanced PWP package is based on the 24-pin TSSOP, but includes a thermal pad (see Figure
36) to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down
TO-220-type packages have leads formed as gull wings to make them applicable for surface-mount applications.
These packages, however, have only two shortcomings: they do not address the low profile (< 2 mm)
requirements of many of today's advanced systems, and they do not offer a terminal-count high enough to
accommodate increasing integration. On the other hand, traditional low-power, surface-mount packages require
power-dissipation derating that severely limits the usable range of many high-performance analog circuits.
The PowerPAD™ package (thermally enhanced TSSOP) combines fine-pitch, surface-mount technology with
thermal performance comparable to much larger power packages.
The PowerPAD™ package is designed to optimize the heat transfer to the PWB. Because of the small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths
that remove heat from the component. The thermal pad is formed using a patented lead-frame design and
manufacturing technique to provide a direct connection to the heat-generating IC. When this pad is soldered or
otherwise thermally coupled to an external heat dissipator, high power dissipation in the ultrathin, fine-pitch,
surface-mount package can be reliably achieved.
DIE
Side View (a)
Thermal
Pad
DIE
End View (b)
Bottom View (c)
Figure 36. Views of Thermally Enhanced PWP Package
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TPA0312
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SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
APPLICATION INFORMATION
SELECTION OF COMPONENTS
Figure 37 and Figure 38 are schematic diagrams of typical notebook computer application circuits.
C
Right
Head−
phone
Input
IRHP
0.47 µF
RHPIN
20
Signal
R
MUX
Volume
Control
23 RLINEIN
C
IRLINE
Right
Line
0.47 µF
−
ROUT+ 21
Input
Signal
+
8
RIN
Volume
Control
C
RIN
0.47 µF
C
OUTR
PC-BEEP
Input
Signal
330 µF
14
PC-BEEP
−
ROUT− 16
PC-
Beep
V
DD
C
1 kΩ
PCB
+
0.47 µF
100 kΩ
GAIN0
GAIN1
See Note A
PV
DD
18
2, 3
17
V
Gain/
MUX
Control
DD
HP/LINE
C
SR
Depop
Circuitry
0.1 µF
V
DD
15 SE/BTL
19
V
DD
Power
Management
C
BYPASS 11
SHUT−
SR
0.1 µF
C
ILHP
Left
DOWN
22
0.47 µF
Head−
phone
Input
C
BYP
LHPIN
6
5
GND
0.47 µF
To
L
MUX
System
Control
Volume
Control
Signal
1 kΩ
LLINEIN
C
ILLINE
1, 12,
13, 24
0.47 µF
Left
Line
Input
Signal
−
LOUT+
4
C
OUTL
+
330 µF
10
LIN
Volume
Control
C
LIN
0.47 µF
−
LOUT−
9
+
100 kΩ
A. A 0.1-µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower frequency noise
signals, a larger electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier.
Figure 37. Typical TPA0312 Application Circuit Using Single-Ended Inputs and Input MUX
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TPA0312
www.ti.com
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
APPLICATION INFORMATION (continued)
C
IRHP−
0.47 µF
RHPIN
20
IRIN−
C
R
MUX
Volume
Control
Right
Negative
0.47 µF
23 RLINEIN
Differential
Input Signal
−
ROUT+ 21
+
C
IRIN+
0.47 µF
Right
Positive
8
RIN
Volume
Control
Differential
Input Signal
C
OUTR
330 µF
PC-BEEP
Input
14
PC-BEEP
−
PC-
Beep
ROUT− 16
Signal
V
DD
C
PCB
1 kΩ
+
0.47 µF
100 kΩ
See Note A
GAIN0
GAIN1
2, 3
17
PV
V
18
DD
DD
V
SR
Gain/
MUX
Control
DD
HP/LINE
SE/BTL
C
Depop
Circuitry
0.1 µF
15
19
11
22
V
DD
Power
Management
C
SR
BYPASS
0.1 µF
SHUT−
DOWN
C
ILHP
C
BYP
0.47 µF
GND
0.47 µF
To
6
5
LHPIN
System
Control
L
MUX
Volume
Control
1 kΩ
Left
Negative
LLINEIN
1, 12,
13, 24
Differential
Input Signal
4
C
C
ILIN−
OUTL
LOUT+
−
0.47 µF
330 µF
+
Left
Positive
Differential
Input Signal
10
LIN
Volume
Control
C
ILIN
0.47 µF
9
−
LOUT−
+
100 kΩ
A. A 0.1-µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower frequency noise
signals, a larger electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier.
Figure 38. Typical TPA0312 Application Circuit Using Differential Inputs
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TPA0312
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SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
GAIN SETTING VIA GAIN0 AND GAIN1 INPUTS
The gain of the TPA0312 is set by two input terminals, GAIN0 and GAIN1.
Table 1. GAIN SETTINGS
GAIN0
GAIN1
SE/BTL
AV
0
0
1
1
X
0
1
0
1
X
0
0
0
0
1
6 dB
10 dB
15.6 dB
21.6 dB
4.1 dB
The gains listed in Table 1 are realized by changing the taps on the input resistors inside the amplifier. This
causes the input impedance, ZI, to be dependant on the gain setting. The actual gain settings are controlled by
ratios of resistors, so the actual gain distribution from part-to-part is quite good. However, the input impedance
will shift by 30% due to shifts in the actual resistance of the input impedance.
For design purposes, the input network (discussed in the next section) should be designed assuming an input
impedance of 10 kΩ, which is the absolute minimum input impedance of the TPA0312. At the lower gain settings,
the input impedance could increase as high as 115 kΩ.
INPUT RESISTANCE
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the –3-dB or
cutoff frequency also changes by over 6 times. If an additional resistor is connected from the input pin of the
amplifier to ground, as shown in the following figure, the variation of the cutoff frequency is much reduced.
Z
F
C
Z
I
IN
Input
Signal
R
The typical input impedance at each gain setting is given in the table below:
Av
ZI
21.6 dB
15.6 dB
10 dB
6 dB
25 kΩ
45 kΩ
70 kΩ
90 kΩ
The –3-dB frequency can be calculated using Equation 1:
1
ƒ
+
–3 dB
2p CǒR ø RIǓ
(1)
If the filter must be more accurate, the value of the capacitor should be increased while the value of the resistor
to ground should be decreased. In addition, the order of the filter could be increased.
INPUT CAPACITOR, CI
In the typical application, an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and the input impedance of the amplifier, ZI, form a
high-pass filter with the corner frequency determined in Equation 2.
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TPA0312
www.ti.com
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
−3 dB
1
f
+
c(highpass)
2pZ C
I
I
f
c
(2)
The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where ZI is 26 kΩ and the specification calls for a flat bass response down to 65 Hz.
Equation 2 is reconfigured as Equation 3.
1
C
+
I
2pZ f
c
I
(3)
In this example, CI is 94 nF; so, one would likely choose a value in the range of 0.1 nF to 1 µF. A further
consideration for this capacitor is the leakage path from the input source through the input network (CI) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high-gain applications. For this reason, a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at VDD/2, which is likely higher
than the source dc level. Note that it is important to confirm the capacitor polarity in the application.
POWER SUPPLY DECOUPLING, CS
The TPA0312 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to
ensure that the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR)
ceramic capacitor, typically 0.1 µF, placed as close as possible to the device VDD lead, works best. For filtering
lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near the audio
power amplifier is recommended.
MIDRAIL BYPASS CAPACITOR, CBYP
The midrail bypass capacitor, CBYP, is the most critical capacitor and serves several important functions. During
start-up or recovery from shutdown mode, CBYP determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD+N.
Bypass capacitor values of 0.47-µF to 1-µF ceramic or tantalum low-ESR capacitors are recommended for the
best THD and noise performance.
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TPA0312
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SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
OUTPUT COUPLING CAPACITOR, CC
In the typical single-supply SE configuration, an output coupling capacitor (CC) is required to block the dc bias at
the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the output
coupling capacitor and impedance of the load form a high-pass filter governed by Equation 4.
−3 dB
1
fc(high)
+
2pR C
C
L
f
c
(4)
The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher, degrading the bass response. Large values of CC are required to pass low
frequencies into the load. Consider the example where a CC of 330 µF is chosen and loads vary from 3 Ω, 4 Ω,
8 Ω, 32 Ω, 10 kΩ, to 47 kΩ. Table 2 summarizes the frequency response characteristics of each configuration.
Table 2. COMMON LOAD IMPEDANCES VS LOW FREQUENCY OUTPUT CHARACTERISTICS IN SE MODE
RL (Ω)
CC (µF)
330
LOWEST FREQUENCY( Hz)
3
4
161
120
60
330
8
330
32
330
15
10,000
47,000
330
0.05
0.01
330
As Table 2 indicates, most of the bass response is attenuated into a 4-Ω load, an 8-Ω load is adequate,
headphone response is good, and drive into line level inputs (a home stereo, for example) is exceptional.
USING LOW-ESR CAPACITORS
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance, the more the real capacitor behaves like an ideal capacitor.
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TPA0312
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SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
BRIDGE-TIED LOAD VERSUS SINGLE-ENDED MODE
Figure 39 shows a Class-AB audio power amplifier (APA) in a BTL configuration. The TPA0312 BTL amplifier
consists of two Class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration, but initially consider power to the load. The differential drive to the speaker means
that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage
swing on the load as compared to a ground-referenced load. Plugging 2 × VO(PP) into the power equation, where
voltage is squared, yields 4× the output power from the same supply rail and load impedance (see Equation 5).
V
O(PP)
V
+
(rms)
Ǹ
2 2
2
V
(rms)
Power +
R
L
(5)
V
DD
V
O(PP)
2x V
O(PP)
R
L
V
DD
–V
O(PP)
Figure 39. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-Ω speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improve-
ment—which is loudness that can be heard. In addition to increased power, there are frequency response
concerns. Consider the single-supply SE configuration shown in Figure 40. A coupling capacitor is required to
block the dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33 µF to
1000 µF); so, they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of
limiting low-frequency performance of the system. This frequency limiting effect is due to the high-pass filter
network created with the speaker impedance and the coupling capacitance and is calculated with Equation 6.
1
f
+
c
2pR C
L C
(6)
For example, a 68-µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
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TPA0312
www.ti.com
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
V
DD
–3 dB
V
O(PP)
C
C
V
O(PP)
R
L
f
c
Figure 40. Single-Ended Configuration and Frequency Response
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4× the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the Crest Factor and Thermal
Considerations section.
SINGLE-ENDED OPERATION
In SE mode the load is driven from the primary amplifier output for each channel (OUT+, terminals 21 and 4).
The amplifier switches single-ended operation when the SE/BTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier's gain to 4.1 dB.
BTL AMPLIFIER EFFICIENCY
Class-AB amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across
the output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc
voltage drop that varies inversely to output power. The second component is due to the sine-wave nature of the
output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from VDD
.
The internal voltage drop multiplied by the RMS value of the supply current, IDDrms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 41).
I
DD
V
O
I
DD(avg)
V
(LRMS)
Figure 41. Voltage and Current Waveforms for BTL Amplifiers
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are
different between SE and BTL configurations. In an SE application, the current waveform is a half-wave rectified
shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
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TPA0312
www.ti.com
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
P
L
Efficiency of a BTL amplifier +
P
SUP
Where:
2
2
V rms
L
V
V
P
P
P
+
, andV
+
, therefore, P
+
L
LRMS
L
Ǹ
R
2R
2
L
L
p
2V
V
p
V
P
1
p
P
1
p
P
+
+ *
sin(t) dt
[cos(t)]
0
ŕ
P
+ V
I
avg
I
avg +
and
and
p R
DD
SUP
DD DD
R
R
L
L
0
L
Therefore,
2 V
V
DD
P
P
+
SUP
p R
L
P = Power delivered to load
substituting P and P
into Equation 7,
2
L
L
SUP
P
V
= Power drawn from power supply
SUP
V
= RMS voltage on BTL load
P
2 R
L
LRMS
R = Load resistance
L
p V
P
Efficiency of a BTL amplifier +
V = Peak voltage on BTL load
P
DD
+
4 V
2 V
V
I
avg = Average current drawn from the
power supply
DD
DD
p R
P
L
Where:
V
= Power supply voltage
= Efficiency of a BTL amplifier
DD
η
BTL
V
+ Ǹ2 P R
L L
P
(7)
(8)
Therefore,
p Ǹ2 P
R
L
L
h
+
BTL
4 V
DD
Table 3 employs Equation 8 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting in
a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at full
output power is less than in the half-power range. Calculating the efficiency for a specific system is the key to
proper power supply design. For a stereo 1-W audio system with 8-Ω loads and a 5-V supply, the maximum draw
on the power supply is almost 3.25 W.
Table 3. EFFICIENCY VS OUTPUT POWER IN 5-V, 8-Ω, BTL SYSTEMS
OUTPUT POWER
(W)
EFFICIENCY
(%)
PEAK VOLTAGE
(V)
INTERNAL DISSIPATION
(W)
0.25
0.50
1.00
1.25
31.4
44.4
62.8
70.2
2.00
2.83
0.55
0.62
0.59
0.53
4.00
4.47(1)
(1) High peak voltages cause the THD to increase.
24
TPA0312
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SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
A final point to remember about Class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in Equation 8, VDD is in the denominator. This
indicates that as VDD goes down, efficiency goes up.
CREST FACTOR AND THERMAL CONSIDERATIONS
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature, the internal
dissipated power at the average output power level must be used. From the TPA0312 data sheet, one can see
that when the TPA0312 is operating from a 5-V supply into a 3-Ω speaker, 4-W peaks are available. Converting
watts to dB:
P
W
4 W
1 W
P
+ 10Log
+ 10Log
+ 6 dB
dB
P
ref
(9)
Subtracting the headroom restriction to obtain the average listening level without distortion yields:
•
•
•
•
•
6 dB - 15 dB = -9 dB (15-dB crest factor)
6 dB - 12 dB = -6 dB (12-dB crest factor)
6 dB - 9 dB = -3 dB (9-dB crest factor)
6 dB - 6 dB = 0 dB (6-dB crest factor)
6 dB - 3 dB = 3 dB (3-dB crest factor)
Converting dB back into watts:
PdBń10
P
+ 10
P
W
ref
+ 63 mW (18−dB crest factor)
+ 125 mW (15−dB crest factor)
+ 250 mW (9−dB crest factor)
+ 500 mW (6−dB crest factor)
+ 1000 mW (3−dB crest factor)
+ 2000 mW (15−dB crest factor)
(10)
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TPA0312
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SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3-dB crest
factor, against 12-dB and 15-dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for a 5-V, 3-Ω system, the internal dissipation in the TPA0312 and
maximum ambient temperatures is shown in Table 4.
Table 4. TPA0312 POWER RATING, 5-V, 3-Ω, STEREO
PEAK OUTPUT POWER
(W)
POWER DISSIPATION
(W/Channel)
MAXIMUM AMBIENT
TEMPERATURE(1)
AVERAGE OUTPUT POWER
4
4
4
4
4
4
2 W (3 dB)
1.7
1.6
1.4
1.1
0.8
0.6
-3°C
6°C
1000 mW (6 dB)
500 mW (9 dB)
250 mW (12 dB)
125 mW (15 dB)
63 mW (18 dB)
24°C
51°C
78°C
85°C
(1) Package limited to 85°C ambient
Table 5. TPA0312 POWER RATING, 5-V, 8-Ω, STEREO
POWER DISSIPATION
AVERAGE OUTPUT POWER
MAXIMUM AMBIENT
TEMPERATURE(1)
PEAK OUTPUT POWER
(W/Channel)
2.5 W
2.5 W
2.5 W
2.5 W
1250 mW (3-dB crest factor)
1000 mW (4-dB crest factor)
500 mW (7-dB crest factor)
250 mW (10-dB crest factor)
0.55
0.62
0.59
0.53
85°C
85°C
85°C
85°C
(1) Package limited to 85°C ambient
The maximum dissipated power, PDmax, is reached at a much lower output power level for a 3-Ω load than for an
8-Ω load. As a result, this simple formula for calculating PDmax may be used for a 3-Ω application:
2
DD
2V
P
+
Dmax
2
p R
L
(11)
However, in the case of an 8-Ω load, the PDmax occurs at a point well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the PDmax formula for
an 8-Ω load, but do not exceed the maximum ambient temperature of 85°C.
The maximum ambient temperature depends on the heat-sinking ability of the PCB system. The derating factor
for the PWP package is shown in the dissipation rating table (see page 4). Converting this to θJA:
1
1
Θ
+
+
+ 45°CńW
JA
0.022
Derating Factor
(12)
To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are per
channel so the dissipated power needs to be doubled for two-channel operation. Given θJA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0312 is
150°C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
T
Max
T Max
Θ
P
A
J
JA
D
(
)
(
)
+ 150 * 45 0.6 2 + 96°C 15−dB crest factor
(13)
NOTE:
Internal dissipation of 0.6 W is estimated for a 2.6-W system with 15-dB crest factor
per channel. Package limited to 85°C
26
TPA0312
www.ti.com
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
Table 4 and Table 5 show that for some applications no airflow is required to keep junction temperatures in the
specified range. The TPA0312 is designed with thermal protection that turns the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Table 4 and Table 5 were calculated for maximum
listening volume without distortion. When the output level is reduced the numbers in the table change
significantly. Also, using 8-Ω speakers dramatically increases the thermal performance by increasing amplifier
efficiency.
SE/BTL OPERATION
The ability of the TPA0312 to easily switch between BTL and SE modes is one of its most important cost-saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0312, two separate amplifiers drive OUT+ and OUT-. The SE/BTL input (terminal 15) controls
the operation of the follower amplifier that drives LOUT- and ROUT- (terminals 9 and 16). When SE/BTL is held
low, the amplifier is on and the TPA0312 is in the BTL mode. When SE/BTL is held high, the OUT- amplifiers are
in a high-output impedance state, which configures the TPA0312 as an SE driver from LOUT+ and ROUT+
(terminals 4 and 21). IDD is reduced by approximately one-half in SE mode. Control of the SE/BTL input can be
from a logic-level CMOS source or, more typically, from a resistor divider network as shown in Figure 42.
20 RHPIN
R
MUX
Volume
Control
23 RLINEIN
−
ROUT+ 21
+
Volume
Control
8
RIN
C
OUTR
V
DD
330 µF
−
ROUT− 16
SE/BTL 15
+
1 kΩ
100 kΩ
100 kΩ
Figure 42. TPA0312 Resistor Divider Network Circuit
Using a readily available 1/8-in. (3,5-mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed, the 100-kΩ/1-kΩ divider pulls the SE/BTL input low. When a plug is inserted, the 1-kΩ
resistor is disconnected and the SE/BTL input is pulled high. When the input goes high, the OUT- amplifier is
shut down causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (CO) into the headphone jack.
27
TPA0312
www.ti.com
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
INPUT MUX OPERATION
C
IRHP
0.47 µF
Right
Headphone
Input Signal
RHPIN
20
R
MUX
Volume
Control
23 RLINEIN
C
IRLINE
0.47 µF
Right Line
−
ROUT+ 21
Input Signal
+
8
RIN
Volume
Control
C
RIN
0.47 µF
ROUT− 16
−
+
15
2
SE/BTL
HP/LINE
Figure 43. TPA0312 Example Input MUX Circuit
The TPA0312 offers the capability for the designer to use separate headphone inputs (RHPIN, LHPIN) and line
inputs (RLINEIN, LLINEIN). The inputs can be different if the input signal is single-ended. If using a differential
input signal, the inputs must be the same because the inputs share a common RIN, LIN. Although the typical
application in Figure 37 shows the input mux control signal HP/LINE tied to SE/BTL, that configuration is not
required. The input mux can be used to select between two inputs that are used in both SE and BTL modes.
If using the TPA0312 with a single-ended input, the RIN and LIN terminals must be tied through a capacitor to
ground, as shown in Figure 43. RIN and LIN must not be tied to bypass or an offset occurs on the output causing
the device to pop when turning on and off.
Input coupling capacitors can be eliminated when using differential inputs, but are used to obtain maximum
output power. If the input capacitors are eliminated, the dc offset must match the voltage on BYPASS or the
output power is limited.
28
TPA0312
www.ti.com
SLOS335A–DECEMBER 2000–REVISED OCTOBER 2004
PC-BEEP OPERATION
The PC-BEEP input allows a system beep to be sent directly from a computer through the amplifier to the
speakers with few external components. The input is activated automatically. When the PC-BEEP input is active,
both LINEIN and HPIN inputs are deselected, and both the left and right channels are driven in BTL mode with
the signal from PC-BEEP. The gain from the PC-BEEP input to the speakers is fixed at 0.3 V/V and is
independent of the volume setting. When the PC-BEEP input is deselected, the amplifier returns to the previous
operating mode and volume setting. Furthermore, if the amplifier is in shutdown mode, activating PC-BEEP takes
the device out of shutdown, outputs the PC-BEEP signal, then returns the amplifier to shutdown mode.
The preferred input signal is a square wave or pulse train. To be accurately detected, the signal must have a
minimum of 1.5-Vpp amplitude, rise and fall times of less than 0.1 µs and a minimum of eight rising edges. When
the signal is no longer detected, the amplifier returns to its previous operating mode and volume setting.
To ac-couple the PC-BEEP input, choose a coupling-capacitor value to satisfy Equation 14:
1
C
w
PCB
2p ƒ
(100 kW)
PCB
(14)
The PC-BEEP input can also be dc-coupled to avoid using this coupling capacitor. The pin normally rests at
midrail when no signal is present.
SHUTDOWN MODES
The TPA0312 employs a shutdown mode of operation designed to reduce supply current, IDD, to the absolute
minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal should
be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the outputs to
mute and the amplifier to enter a low-current state, IDD = 150 µA. SHUTDOWN should never be left unconnected
because amplifier operation would be unpredictable.
Table 6. HP/LINE, SE/BTL, AND SHUTDOWN FUNCTIONS
INPUTS(1)
SE/BTL
X(2)
AMPLIFIER STATE
HP/LINE
X(2)
SHUTDOWN
Low
INPUT
OUTPUT
Mute
BTL
X(2)
Line
Line
HP
Low
Low
High
Low
High
High
SE
High
High
Low
High
BTL
High
High
HP
SE
(1) Inputs should never be left unconnected.
(2) X = do not care
29
PACKAGE OPTION ADDENDUM
www.ti.com
10-Jun-2014
PACKAGING INFORMATION
Orderable Device
TPA0312PWP
Status Package Type Package Pins Package
Eco Plan
Lead/Ball Finish
MSL Peak Temp
Op Temp (°C)
-40 to 85
Device Marking
Samples
Drawing
Qty
(1)
(2)
(6)
(3)
(4/5)
ACTIVE
HTSSOP
HTSSOP
HTSSOP
PWP
24
24
24
60
Green (RoHS
& no Sb/Br)
CU NIPDAU
CU NIPDAU
CU NIPDAU
Level-2-260C-1 YEAR
Level-2-260C-1 YEAR
Level-2-260C-1 YEAR
TPA0312
TPA0312PWPR
TPA0312PWPRG4
ACTIVE
ACTIVE
PWP
PWP
2000
2000
Green (RoHS
& no Sb/Br)
-40 to 85
TPA0312
TPA0312
Green (RoHS
& no Sb/Br)
-40 to 85
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
10-Jun-2014
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
14-Jul-2012
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
B0
K0
P1
W
Pin1
Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant
(mm) W1 (mm)
TPA0312PWPR
HTSSOP PWP
24
2000
330.0
16.4
6.95
8.3
1.6
8.0
16.0
Q1
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
14-Jul-2012
*All dimensions are nominal
Device
Package Type Package Drawing Pins
HTSSOP PWP 24
SPQ
Length (mm) Width (mm) Height (mm)
367.0 367.0 38.0
TPA0312PWPR
2000
Pack Materials-Page 2
IMPORTANT NOTICE
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TI warrants performance of its components to the specifications applicable at the time of sale, in accordance with the warranty in TI’s terms
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TPA0312PWPR CAD模型
封装焊盘图
TPA0312PWPR 替代型号
| 型号 | 制造商 | 描述 | 替代类型 | 文档 |
| TPA0312PWP | TI | 2.6-W STEREO AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL | 完全替代 |
|
| TPA0312PWPG4 | TI | 2 CHANNEL(S), VOLUME CONTROL CIRCUIT, PDSO24, GREEN, PLASTIC, HTSSOP-24 | 完全替代 |
|
TPA0312PWPR 相关器件
| 型号 | 制造商 | 描述 | 价格 | 文档 |
| TPA0312PWPRG4 | TI | Stereo 2-W Audio Power Amp with 4 Selectable Gain Settings and MUX Control 24-HTSSOP -40 to 85 | 获取价格 |
|
| TPA0312_16 | TI | 2.6-W STEREO AUDIO POWER AMPLIFIER WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL | 获取价格 |
|
| TPA032D01 | TI | 10-W MONO CLASS-D AUDIO POWER AMPLIFIER | 获取价格 |
|
| TPA032D01DCA | TI | 10-W MONO CLASS-D AUDIO POWER AMPLIFIER | 获取价格 |
|
| TPA032D01DCAR | TI | 10-W MONO CLASS-D AUDIO POWER AMPLIFIER | 获取价格 |
|
| TPA032D01DCARG4 | TI | 10-W MONO CLASS-D AUDIO POWER AMPLIFIER | 获取价格 |
|
| TPA032D01_07 | TI | 10-W MONO CLASS-D AUDIO POWER AMPLIFIER | 获取价格 |
|
| TPA032D02 | TI | 10-W STEREO CLASS-D AUDIO POWER AMPLIFIER | 获取价格 |
|
| TPA032D02DCA | TI | 10-W STEREO CLASS-D AUDIO POWER AMPLIFIER | 获取价格 |
|
| TPA032D02DCAG4 | TI | 10-W STEREO CLASS-D AUDIO POWER AMPLIFIER | 获取价格 |
|
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