TPA6120A2RGYR [TI]
High Fidelity Headphone Amplifier;
| 型号： | TPA6120A2RGYR |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | High Fidelity Headphone Amplifier 放大器 商用集成电路 |
| 文件： | 总24页 (文件大小：608K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
HIGH FIDELITY HEADPHONE AMPLIFIER
FEATURES
DESCRIPTION
•
80 mW into 600 Ω From a ±12-V Supply at
0.00014% THD + N
The TPA6120A2 is a high fidelity audio amplifier built
on current-feedback architecture. This high
a
bandwidth, extremely low noise device is ideal for
high performance equipment. The better than 120 dB
of dynamic range exceeds the capabilities of the
human ear, ensuring that nothing audible is lost due
to the amplifier. The solid design and performance of
the TPA6120A2 ensures that music, not the amplifier,
is heard.
•
•
•
•
Current-Feedback Architecture
Greater than 120 dB of Dynamic Range
SNR of 120 dB
Output Voltage Noise of 5 µVrms at
Gain = 2 V/V
•
•
•
•
Power Supply Range: ±5 V to ±15 V
1300 V/µs Slew Rate
Three key features make current-feedback amplifiers
outstanding for audio. The first feature is the high
slew rate that prevents odd order distortion
anomalies. The second feature is current-on-demand
at the output that enables the amplifier to respond
quickly and linearly when necessary without risk of
output distortion. When large amounts of output
power are suddenly needed, the amplifier can re-
spond extremely quickly without raising the noise
floor of the system and degrading the signal-to-noise
ratio. The third feature is the gain-independent fre-
quency response that allows the full bandwidth of the
amplifier to be used over a wide range of gain
settings. The excess loop gain does not deteriorate at
a rate of 20 dB/decade.
Differential Inputs
Independent Power Supplies for Low
Crosstalk
•
Short Circuit and Thermal Protection
APPLICATIONS
•
•
•
•
•
Professional Audio Equipment
Mixing Boards
Headphone Distribution Amplifiers
Headphone Drivers
Microphone Preamplifiers
I/V Gain Stage
Filter and
Stereo Hi−Fi
Headphone Driver
1/2 OPA4134
2.7 nF
C
F
AUDIO DAC
TPA6120A2
R
F
1 kΩ
R
F
−IN A
+IN A
R
I
1 kΩ
OUT A
OUT B
I
L−
OUT
LRCK
BCK
PCM
Audio
Data
1 kΩ
LIN−
LIN+
R
O
LOUT
+IN B
−IN B
R
I
DATA
SCK
10 Ω
Source
I
L+
OUT
1 kΩ
R
F
1 kΩ
1 kΩ
R
F
PCM1792
or
C
2.7 nF
F
1/2 OPA4134
C
DSD1792
2.7 nF
F
ZEROL
ZEROR
R
F
1 kΩ
RIN+
RIN−
1 kΩ
R
F
+IN C
−IN C
R
I
MS
I
R+
OUT
OUT C
OUT D
R
O
ROUT
MDI
MC
Controller
1 kΩ
10 Ω
DYR > 120 dB
for Whole
System!
+IN D
−IN D
R
I
MDO
I
R−
OUT
R
F
1 kΩ
1 kΩ
RST
R
F
1 kΩ
C
F
2.7 nF
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2004, Texas Instruments Incorporated
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted)
(1)
TPA6120A2
Supply voltage, VCC+ to VCC-
Input voltage, VI(2)
33 V
± VCC
Differential input voltage, VID
Minimum load impedance
6 V
8 Ω
Continuous total power dissipation
Operating free–air temperature range, TA
Operating junction temperature range, TJ
Storage temperature range, Tstg
See Dissipation Rating Table
- 40°C to 85°C
- 40°C to 150°C
- 40°C to 125°C
235°C
(3)
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds
(1) Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating
conditions” is not implied. Exposure to absolute–maximum–rated conditions for extended periods may affect device reliability.
(2) When the TPA6120A2 is powered down, the input source voltage must be kept below 600-mV peak.
(3) The TPA6120A2 incorporates an exposed PowerPAD on the underside of the chip. This acts as a heatsink and must be connected to a
thermally dissipating plane for proper power dissipation. Failure to do so may result in exceeding the maximum junction temperature that
could permanently damage the device. See TI Technical Brief SLMA002 for more information about utilizing the PowerPAD thermally
enhanced package.
DISSIPATION RATING TABLE
(1)
θJA
θJC
TA = 25°C
PACKAGE
(°C/W)
(°C/W)
POWER RATING
DWP
44.4
33.8
2.8 W
(1) The PowerPAD must be soldered to a thermal land on the printed-circuit board. See the PowerPAD
Thermally Enhanced Package application note (SLMA002)
AVAILABLE OPTIONS
TA
PACKAGE
PART NUMBER
SYMBOL
-40°C to 85°C
DWP(1)
TPA6120A2DWP
6120A2
(1) The DWP package is available taped and reeled. To order a taped and reeled part, add the suffix R
to the part number (e.g., TPA6120A2DWPR).
RECOMMENDED OPERATING CONDITIONS
MIN
±5
MAX
±15
30
UNIT
Split Supply
Single Supply
Supply voltage, VCC+ and VCC-
V
10
Load impedance
VCC = ±5 V or ±15 V
16
Ω
Operating free–air temperature, TA
-40
85
°C
2
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
ELECTRICAL CHARACTERISTICS
over operating free-air temperature range (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
2
MAX
UNIT
mV
|VIO
|
Input offset voltage (measured differentially)
Power supply rejection ratio
VCC = ±5 V or ±15 V
VCC = 2.5 V to 5.5 V
VCC = ±5 V
5
PSRR
75
dB
±3.6
±3.7
±13.5
11.5
VIC
Common mode input voltage
Supply current (each channel)
V
VCC = ±15 V
±13.4
VCC = ±5 V
13
15
ICC
IO
mA
VCC= ±15 V
Output current (per channel)
Input offset voltage drift
Input resistance
VCC= ±5 V to ±15 V
VCC = ±5 V or ±15 V
700
20
mA
µV/°C
kΩ
ri
300
13
ro
Output resistance
Open Loop
Ω
11.8 to
-11.5
12.5 to
-12.2
VO
Output voltage swing
VCC = ±15 V, RL = 25 Ω
V
3
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
OPERATING CHARACTERISTICS(1)
TA = 25°C, RL = 25 Ω, Gain = 2 V/V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
VCC = ±12 V to ±15 V,
RL = 32 Ω,
VI = 1 VPP
0.00014%
SMTPE ratio = 4:1,
Gain = 2 V/V,
Intermodulation distortion
(SMPTE)
IMD
IM frequency = 60 Hz
High frequency = 7 kHz
VCC = ±12 V to ±15 V,
RL = 64 Ω,
0.000095%
VI = 1 VPP
VCC = ±12 V
VCC = ±15 V
VCC = ±12 V
VCC = ±15 V
PO = 80 mW
PO = 40 mW
PO = 125 mW
PO = 62.5 mW
0.00055%
0.00060%
0.00038%
0.00029%
0.00014%
0.000065%
0.00012%
0.000061%
PO = 100 mW, RL = 32 Ω
f = 1 kHz
PO = 100 mW, RL = 64 Ω
f = 1 kHz
VCC = ±12 V, Gain = 3 V/V
RL = 600 Ω, f = 1 kHz
Total harmonic distortion
plus noise
THD+N
VCC = ±15 V, Gain = 3 V/V
RL = 600 Ω, f = 1 kHz
VO = 15 VPP
RL = 10 kΩ
f = 1 kHz
,
VCC = ±12 V,
Gain = 3 V/V
0.000024%
0.000021%
VO = 15 VPP
RL = 10 kΩ
f = 1 kHz
,
VCC = ±15 V,
Gain = 3 V/V
RL = 32 Ω
f = 10 Hz to 22 kHz
V(RIPPLE) = 1 VPP
VCC= ±12 V
VCC= ±15 V
VCC= ±12 V
VCC= ±15 V
-80
-83
-76
-79
Supply voltage rejection
ratio
kSVR
dB
RL = 64 Ω
f = 10 Hz to 22 kHz
V(RIPPLE) = 1 VPP
Common mode rejection
ratio (differential)
CMRR
SR
VCC = ±5 V or ±15 V
100
dB
VCC = ±15 V, Gain = 5 V/V, VO = 20 VPP
VCC = ±5 V, Gain = 2 V/V, VO = 5 VPP
1300
900
5
Slew rate
V/µs
VCC = ±12 V to ±15 V
RL = 32 Ω to 64 Ω
f = 1 kHz
Gain = 2 V/V
Gain = 100 V/V
Gain = 2 V/V
Gain = 100 V/V
Vn
Output noise voltage
µVrms
50
VCC = ±12 V to ±15 V
RL = 32 Ω to 64 Ω
f = 1 kHz
125
104
SNR
Signal-to-noise ratio
Dynamic range
Crosstalk
dB
VCC = ±12 V
VCC = ±15 V
VCC = ±12 V
VCC = ±15 V
123
125
124
126
RL = 32 Ω, f = 1 kHz
RL = 64 Ω, f = 1 kHz
dB
dB
VCC = ±12 V to ±15 V
RL = 32 Ω to 64 Ω
f = 1 kHz
VI = 1 VRMS
RF = 1 kΩ
-90
(1) For IMD, THD+N, kSVR, and crosstalk, the bandwidth of the measurement instruments was set to 80 kHz.
4
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
DEVICE INFORMATION
Thermally Enhansed SOIC (DWP)
PowerPAD™ Package
Top View
1
2
3
4
5
6
7
8
9
10
20
19
18
17
16
15
14
13
12
11
LVCC−
LOUT
LVCC+
LIN+
LIN−
NC
RVCC−
ROUT
RVCC+
RIN+
RIN−
NC
NC
NC
NC
NC
NC
NC
NC
NC
NC − No internal connection
TERMINAL FUNCTIONS
PIN NAME
PIN NUMBER
I/O
DESCRIPTION
Left channel negative power supply – must be kept at the same potential as
RVCC-.
LVCC-
1
I
LOUT
LVCC+
LIN+
2
O
I
Left channel output
3
Left channel positive power supply
Left channel positive input
Left channel negative input
Not internally connected
4
I
LIN-
5
I
NC
6,7,8,9,10,11,12,13,14,15
-
RIN-
16
17
18
19
I
Right channel negative input
Right channel positive input
Right channel positive power supply
Right channel output
RIN+
RVCC+
ROUT
I
I
O
Right channel negative power supply - must be kept at the same potential as
LVCC-.
RVCC-
20
-
I
Connect to ground. The thermal pad must be soldered down in all
applications to properly secure device on the PCB.
Thermal Pad
-
5
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
1, 2, 3, 4
5
vs Frequency
vs Output voltage
vs Output power
vs Output power
vs Frequency
vs High frequency
vs IM Amplitude
vs Frequency
vs Gain
Total harmonic distortion + noise
6, 7, 8
9
Power dissipation
Supply voltage rejection ratio
10, 11
12
Intermodulation distortion
13
Crosstalk
14
Signal-to-noise ratio
Slew rate
15, 16
17, 18
19, 20
21
vs Output step
Small and large signal frequency response
400-mV step response
10-V step response
20-V step response
22
23
TOTAL HARMONIC DISTORTION + NOISE
TOTAL HARMONIC DISTORTION + NOISE
vs
vs
FREQUENCY
FREQUENCY
0.01
0.01
R
= 10 kW,
L
R
= 600 W,
L
Gain = 3 V/V,
Gain = 3 V/V,
R
F
= 2 kW,
R
F
= 2 kW,
R = 1 kW,
I
R = 1 kW,
I
BW = 80 kHz
BW = 80 kHz
0.001
0.001
V
CC
= +15 V = 15 V
O PP
V
CC
= +12 V = 15 V
O PP
0.0001
V
CC
= +12 V = 12 V
O PP
V
O
= +15 V,
V
= +12 V,
CC
CC
O
P
= 125 mW
P
= 80 mW
V
= +15 V = 23 V
O PP
CC
0.00001
0.0001
10
100
1 k
10 k
50 k
10
100
1 k
10 k
50 k
f − Frequency − Hz
f − Frequency − Hz
Figure 1.
Figure 2.
6
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
TYPICAL CHARACTERISTICS (continued)
TOTAL HARMONIC DISTORTION + NOISE
TOTAL HARMONIC DISTORTION + NOISE
vs
vs
FREQUENCY
FREQUENCY
0.1
1
R
= 64 W,
L
R
= 32 W,
L
Gain = 2 V/V,
Gain = 2 V/V,
R
F
= 1 kW,
R
F
= 1 kW,
R = 1 kW,
I
R = 1 kW,
I
BW = 80 kHz
BW = 80 kHz
0.1
0.01
V
CC
= +15 V, P = 700 mW
O
V
CC
= +15 V, P = 1.5 W
O
0.01
V
CC
= +15 V, P = 1.25 W
O
V
CC
= +15 V, P = 1.35 W
O
V
CC
= +12 V, P = 950 mW
O
V
CC
= +12 V, P = 425 mW
O
0.001
V
CC
= + 12 V, P = 800 mW
O
0.001
V
CC
= +12 V, P = 500 mW
O
0.0001
0.0001
10
100
1 k
10 k
50 k
10
100
1 k
10 k
50 k
f − Frequency − Hz
f − Frequency − Hz
Figure 3.
Figure 4.
TOTAL HARMONIC DISTORTION + NOISE
TOTAL HARMONIC DISTORTION + NOISE
vs
vs
OUTPUT VOLTAGE
OUTPUT POWER
10
10
1
R
= 600 W,
L
R
= 10 kW,
L
Gain = 3 V/V,
f = 1 kHz,
Gain = 3 V/V,
f = 1 kHz,
1
R = 2 kW,
F
R
F
= 2 kW,
R = 1 kW,
I
R = 1 kW,
I
BW = 80 kHz
BW = 80 kHz
0.1
0.1
V
CC
= + 12 V
0.01
0.01
V
CC
= + 15 V
V
CC
= +12 V
0.001
0.001
0.0001
0.0001
V
CC
= +15 V
0.00001
0.00001
3
5
10
15
20
25
30
35
0.01
0.1
0.2
V
O
− Output Voltage − V
PP
P
O
− Output Power − W
Figure 5.
Figure 6.
7
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
TYPICAL CHARACTERISTICS (continued)
TOTAL HARMONIC DISTORTION + NOISE
TOTAL HARMONIC DISTORTION + NOISE
vs
vs
OUTPUT POWER
OUTPUT POWER
10
10
R
= 64 W,
R = 32 W,
L
L
Gain = 2 V/V,
f = 1 kHz,
Gain = 2 V/V,
f = 1 kHz,
R
F
= 1 kW,
R = 1 kW,
F
1
0.1
1
0.1
R = 1 kW,
R = 1 kW,
I
I
BW = 80 kHz
BW = 80 kHz
V
CC
= + 12 V
V
CC
= + 12 V
V
CC
= + 15 V
V
CC
= + 15 V
0.01
0.01
0.001
0.001
0.0001
0.0001
0.01
0.1
− Output Power − W
1
2
0.01
0.1
P − Output Power − W
O
1
2
3 4
P
O
Figure 7.
Figure 8.
POWER DISSIPATION
vs
OUTPUT POWER
SUPPLY VOLTAGE REJECTION RATIO
vs
FREQUENCY
0
2
Mono Operation
V
CC
= +15 V, R = 32 W
L
V
= + 12 V,
CC
V
= 1 V ,
PP
(ripple)
Gain = 2 V/V
BW = 80 kHz
1.8
−10
1.6
1.4
1.2
1
−20
−30
−40
−50
V
CC
= +12 V, R = 32 W
L
Representative of both positive and
negative supplies.
V
CC
R
= +15 V,
L
= 64 W
0.8
−60
−70
−80
64 W
0.6
0.4
V
= +12 V,
L
32 W
CC
R
= 64 W
0.2
0
−90
10
0
0.5
1
1.5
2
2.5
3
3.5
100
1 k
10 k
50 k
f − Frequency − Hz
P
O
− Output Power − W
Figure 9.
Figure 10.
8
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
TYPICAL CHARACTERISTICS (continued)
SUPPLY VOLTAGE REJECTION RATIO
INTERMODULATION DISTORTION
vs
vs
FREQUENCY
HIGH FREQUENCY
−0
0.1
4:1 SMPTE Ratio
V
V
= + 15 V,
(ripple)
CC
V = 1 V ,
I PP
−10
= 1 V
,
PP
Gain = 2 V/V,
Gain = 2 V/V
BW = 80 kHz
IM Frequency = 60 Hz
−20
−30
−40
−50
0.01
Representative of both positive and
negative supplies.
0.001
V
CC
R
= +12 V,
= 32 W
V
CC
R
= +15 V,
= 32 W
L
L
−60
64 W
0.0001
−70
−80
−90
32 W
V
CC
R
= +15 V,
= 64 W
V
CC
R
= +12 V,
L
L
= 64 W
0.00001
2 k
10 k
50 k
10
100
1 k
10 k
50 k
f − High Frequency − Hz
f − Frequency − Hz
Figure 11.
Figure 12.
INTERMODULATION DISTORTION
vs
CROSSTALK
vs
FREQUENCY
IM AMPLITUDE (AT INPUT)
10
1
−60
−70
−80
R
= 1 kW,
F
4:1 SMPTE Ratio
Gain = 2 V/V,
BW = 80 kHz
Gain = 3 V/V,
High Frequency = 7 kHz
IM Frequency = 60 Hz
V
= +15 V,
= 64 W
CC
L
R
V
CC
= +12 V, R = 32 W
L
V
CC
= +12 V,
R
L
= 64 W
0.1
V
CC
= +12 V, R = 64 W
L
−90
V
= +15 V,
= 32 W
CC
L
0.01
R
V
CC
= +15 V, R = 32 W
L
V
= +12 V,
= 32 W
CC
L
R
−100
−110
0.001
0.0001
V
CC
= +15 V, R = 64 W
L
−120
10
0.00001
100
1 k
10 k
50 k
0
1
2
3
4
5
6
7
8
9
10
f − Frequency − Hz
IM Amplitude (At Input) − V
PP
Figure 13.
Figure 14.
9
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
TYPICAL CHARACTERISTICS (continued)
SIGNAL-TO-NOISE RATIO
SIGNAL-TO-NOISE RATIO
vs
vs
GAIN
GAIN
130
130
V
CC
= +12 V
V
CC
= +15 V
THD+N, R = 64 W
I
R = 64 W
I
125
120
115
125
120
115
110
110
105
100
R = 32 W
I
THD+N, R = 32 W
I
105
100
1
10 20 30 40 50 60 70 80 90 100
Gain − V/V
1
10 20 30 40 50 60 70 80 90 100
Gain − V/V
Figure 15.
Figure 16.
SLEW RATE
vs
OUTPUT STEP
SLEW RATE
vs
OUTPUT STEP
1500
1300
1100
900
1000
900
800
700
600
500
400
300
200
100
V
= ± 15 V
V
= ± 5 V
CC
CC
Gain = 5 V/V
R
R
Gain = 2 V/V
R
R
= 1 kΩ
= 25 Ω
= 1 kΩ
= 25 Ω
F
L
F
L
+SR
−SR
+SR
−SR
700
500
300
100
0
20
0
5
5
10
15
1
2
3
4
Output Step (Peak−To−Peak) − V
Output Step (Peak−To−Peak) − V
Figure 17.
Figure 18.
10
TPA6120A2
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SLOS431–MARCH 2004
TYPICAL CHARACTERISTICS (continued)
SMALL AND LARGE SIGNAL
FREQUENCY RESPONSE
SMALL AND LARGE SIGNAL
FREQUENCY RESPONSE
−3
3
0
V = 500 mV
I
V = 500 mV
I
−6
−9
−12
−15
−3
−6
−9
V = 250 mV
I
V = 250 mV
I
V = 125 mV
I
V = 125 mV
I
−18
−21
−24
−27
−30
−12
−15
−18
−21
−24
V = 62.5 mV
I
V = 62.5 mV
I
Gain = 1 V/V
Gain = 2 V/V
V
R
R
= ± 15 V
= 820 Ω
= 25 Ω
V
R
R
= ± 15 V
= 680 Ω
= 25 Ω
CC
CC
F
L
F
L
10 100
1k
10k 100k 1M 10M 100M 500M
f − Frequency − Hz
10 100
1k
10k 100k 1M 10M 100M 500M
f − Frequency − Hz
Figure 19.
Figure 20.
400-mV STEP RESPONSE
10-V STEP RESPONSE
400
300
200
100
0
8
6
4
2
0
−100
−2
−4
V
= ±15 V
CC
V
= ±15 V
CC
Gain = 2 V/V
R
R
Gain = 2 V/V
R
R
= 25 Ω
= 1 kΩ
−200
L
= 25 Ω
= 1 kΩ
L
F
F
t /t = 5 ns
r
f
−300
−400
t /t = 300 ps
−6
−8
r
f
See Figure 3
See Figure 3
0
50 100 150 200 250 300 350 400 450 500
t − Time − ns
0
50 100 150 200 250 300 350 400 450 500
t − Time − ns
Figure 21.
Figure 22.
11
TPA6120A2
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SLOS431–MARCH 2004
TYPICAL CHARACTERISTICS (continued)
20-V STEP RESPONSE
16
12
8
V
= ±15 V
CC
Gain = 5 V/V
R
R
= 25 Ω
= 2 kΩ
L
F
t /t = 5 ns
r
f
See Figure 3
4
0
−4
−8
−12
−16
0
50 100 150 200 250 300 350 400 450 500
t − Time − ns
Figure 23.
12
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
APPLICATION INFORMATION
Current-Feedback Amplifiers
The TPA6120A2 is
a current-feedback amplifier with differential inputs and single-ended outputs.
Current-feedback results in low voltage noise, high open-loop gain throughout a large frequency range, and low
distortion. It can be used in a similar fashion as voltage-feedback amplifiers. The low distortion of the
TPA6120A2 results in a signal-to-noise ratio of 120 dB as well as a dynamic range of 120 dB.
Independent Power Supplies
The TPA6120A2 consists of two independent high-fidelity amplifiers. Each amplifier has its own voltage supply.
This allows the user to leave one of the amplifiers off, saving power, and reducing the heat generated. It also
reduces crosstalk.
Although the power supplies are independent, there are some limitations. When both amplifiers are used, the
same voltage must be applied to each amplifier. For example, if the left channel amplifier is connected to a ±12-V
supply, the right channel amplifier must also be connected to a ±12-V supply. If it is connected to a different
supply voltage, the device may not operate properly and consistently.
When the use of only one amplifier is preferred, it must be the left amplifier. The voltage supply to the left
amplifier is also responsible for internal start-up and bias circuitry of the device. Regardless of whether one or
both amplifiers are used, the VCC- pins of both amplifiers must always be at the same potential.
To power down the right channel amplifier, disconnect the VCC+ pin from the power source.
The two independent power supplies can be tied together on the board to receive their power from the same
source.
Power Supply Decoupling
As with any design, proper power supply decoupling is essential. It prevents noise from entering the device via
the power traces and provides the extra power the device can sometimes require in a rapid fashion. This
prevents the device from being momentarily current starved. Both of these functions serve to reduce distortion,
leaving a clean, uninterrupted signal at the output.
Bulk decoupling capacitors should be used where the main power is brought to the board. Smaller capacitors
should be placed as close as possible to the actual power pins of the device. Because the TPA6120A2 has four
power pins, use four surface mount capacitors. Both types of capacitors should be low ESR.
Resistor Values
R = 1 kW
F
V
−
CC−
R = 1 kW
I
R
O
= 10 W
V
I
+
R
L
R
S
= 50 W
V
CC+
Figure 24. Single-Ended Input With a Noninverting Gain of 2 V/V
In the most basic configuration (see Figure 24), four resistors must be considered, not including the load
impedance. The feedback and input resistors, RF and RI, respectively, determine the closed-loop gain of the
amplifier. RO is a series output resistor designed to protect the amplifier from any capacitance on the output path,
including board and load capacitance. RS is a series input resistor. Because the TPA6120A2 is a
current-feedback amplifier, take care when choosing the feedback resistor.
13
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
APPLICATION INFORMATION (continued)
The value of the feedback resistor should be chosen by using Figure 27 through Figure 32 as guidelines. The
gain can then be set by adjusting the input resistor. The smaller the feedback resistor, the less noise is
introduced into the system. However, smaller values move the dominant pole to higher and higher frequencies,
making the device more susceptible to oscillations. Higher feedback resistor values add more noise to the
system, but pull the dominant pole down to lower frequencies, making the device more stable. Higher impedance
loads tend to make the device more unstable. One way to combat this problem is to increase the value of the
feedback resistor. It is not recommended that the feedback resistor exceed a value of 10 kΩ. The typical value
for the feedback resistor for the TPA6120A2 is 1 kΩ. In some cases, where a high-impedance load is used along
with a relatively large gain and a capacitive load, it may be necessary to increase the value of the feedback
resistor from 1 kΩ to 2 kΩ, thus adding more stability to the system. Another method to deal with oscillations is to
increase the size of RO.
CAUTION:
Do not place a capacitor in the feedback path. Doing so can cause oscillations.
Capacitance at the outputs can cause oscillations. Capacitance from some sources, such as layout, can be
minimized. Other sources, such as those from the load (e.g., the inherent capacitance in a pair of headphones),
cannot be easily minimized. In this case, adjustments to RO and/or RF may be necessary.
The series output resistor should be kept at a minimum of 10 Ω. It is small enough so that the effect on the load
is minimal, but large enough to provide the protection necessary such that the output of the amplifier sees little
capacitance. The value can be increased to provide further isolation, up to 100 Ω.
The series resistor, RS, should be used for two reasons:
1. It prevents the positive input pin from being exposed to capacitance from the line and source.
2. It prevents the source from seeing the input capacitance of the TPA6120A2.
The 50-Ω resistor was chosen because it provides ample protection without interfering in any noticeable way with
the signal. Not shown is another 50-Ω resistor that can be placed on the source side of RS to ground. In that
capacity, it serves as an impedance match to any 50-Ω source.
R
F
= 1 kW
V
CC−
R = 1 kW
I
V
I
R = 10 W
O
−
+
R
L
V
CC+
Figure 25. Single-Ended Input With a Noninverting Gain of -1 V/V
R
F
= 1 kW
V
CC−
R = 1 kW
I
V
V
R = 10 W
O
−
+
I−
I+
R
L
R = 1 kW
I
V
CC+
R = 1 kW
F
Figure 26. Differential Input With a Noninverting Gain of 2 V/V
Figure 26 shows the TPA6120A2 connected with differential inputs. Differential inputs are useful because they
take the greatest advantage of the device's high CMRR. The two feedback resistor values must be kept the
same, as do the input resistor values.
14
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
APPLICATION INFORMATION (continued)
Special note regarding mono operation:
•
If both amplifiers are powered on, but only one channel is to be used, the unused amplifier MUST have a
feedback resistor from the output to the negative input. Additionally, the positive input should be grounded as
close to the pin as possible. Terminate the output as close to the output pin as possible with a 25-Ω load to
ground.
•
These measures should be followed to prevent the unused amplifier from oscillating. If it oscillates, and the
power pins of both amplifiers are tied together, the performance of the amplifier could be seriously degraded.
Checking for Oscillations and Instability
Checking the stability of the amplifier setup is recommended. High frequency oscillations in the megahertz region
can cause undesirable effects in the audio band.
Sometimes, the oscillations can be quite clear. An unexpectedly large draw from the power supply may be an
indication of oscillations. These oscillations can be seen with an oscilloscope. However, if the oscillations are not
obvious, or there is a chance that the system is stable but close to the edge, placing a scope probe with 10 pF of
capacitance can make the oscillations worse, or actually cause them to start.
A network analyzer can be used to determine the inherent stability of a system. An output vs frequency curve
generated by a network analyzer can be a good indicator of stability. At high frequencies, the curve shows
whether a system is oscillating, close to oscillation, or stable. Looking at Figure 27 through Figure 32, several
different phenomena occur. In one scenario, the system is stable because the high frequency rolloff is smooth
and has no peaking. Increasing RF decreases the frequency at which this rolloff occurs (see the Resistor Values
section). Another scenario shows some peaking at high frequency. If the peaking is 2 dB, the amplifier is stable
as there is still 45 degrees of phase margin. As the peaking increases, the phase margin shrinks, the amplifier
and the system, move closer to instability. The same system that has a 2-dB peak has an increased peak when
a capacitor is added to the output. This indicates the system is either on the verge of oscillation or is oscillating,
and corrective action is required.
3
3
R
= 620 Ω
2
1
F
2
1
R
F
= 430 Ω
R
F
= 820 Ω
0
0
−1
−2
−1
−2
R
F
= 1 kΩ
R
= 620 Ω
= 1 kΩ
F
−3
−4
−5
−3
−4
−5
−6
R
F
V
R
= ±15 V
= 100 Ω
V
= ±15 V
R = 100 Ω
L
CC
CC
L
Gain = 1 V/V
V = 200 mV
Gain = 2 V/V
V = 200 mV
−6
−7
I
I
10 100
1k
10k 100k 1M 10M 100M 500M
f − Frequency − Hz
10 100
1k
10k 100k 1M 10M 100M 500M
f − Frequency − Hz
Figure 27. Normalized Output Response vs Frequency
Figure 28. Normalized Output Response vs Frequency
15
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
APPLICATION INFORMATION (continued)
1
1
0
R
= 200 Ω
L
0
−1
−2
−3
−4
−1
−2
−3
−4
R
= 100 Ω
L
R
L
= 25 Ω
R
= 50 Ω
L
R
R
= 200 Ω
= 100 Ω
L
−5
−6
−5
−6
R
L
= 25 Ω
L
V
R
= ±15 V
= 1 kΩ
CC
V
R
= ±15 V
= 1 kΩ
CC
−7
−8
−9
−7
−8
−9
F
F
R
L
= 50 Ω
Gain = 1 V/V
V = 200 mV
Gain = 2 V/V
V = 200 mV
I
I
10 100
1k
10k 100k 1M 10M 100M 500M
f − Frequency − Hz
10 100
1k
10k 100k 1M 10M 100M 500M
f − Frequency − Hz
Figure 29. Normalized Output Response vs Frequency
Figure 30. Normalized Output Response vs Frequency
3
9
2
8
7
R
F
= 510 Ω
R
F
= 620 Ω
1
0
6
5
−1
R
F
= 1 kΩ
R
F
= 820 Ω
F
−2
−3
−4
−5
−6
4
3
2
1
0
R
= 1.5 kΩ
R
F
= 1.2 kΩ
V
= ± 5 V
V
CC
= ± 5 V
CC
Gain = 1 V/V
= 25 Ω
Gain = 2 V/V
R = 25 Ω
L
V = 200 mV
I
R
L
V = 200 mV
I
10 100
1k
10k 100k 1M 10M 100M 500M
f − Frequency − Hz
10 100
1k
10k 100k 1M 10M 100M 500M
f − Frequency − Hz
Figure 31. Output Amplitude vs Frequency
Figure 32. Output Amplitude vs Frequency
PCB Layout
Proper board layout is crucial to getting the maximum performance out of the TPA6120A2.
A ground plane should be used on the board to provide a low inductive ground connection. Having a ground
plane underneath traces adds capacitance, so care must be taken when laying out the ground plane on the
underside of the board (assuming a 2-layer board). The ground plane is necessary on the bottom for thermal
reasons. However, certain areas of the ground plane should be left unfilled. The area underneath the device
where the PowerPAD is soldered down should remain, but there should be no ground plane underneath any of
the input and output pins. This places capacitance directly on those pins and leads to oscillation problems. The
underside ground plane should remain unfilled until it crosses the device side of the input resistors and the
output series resistor. Thermal reliefs should be avoided if possible because of the inductance they introduce.
16
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
APPLICATION INFORMATION (continued)
Despite the removal of the ground plane in critical areas, stray capacitance can still make its way onto the
sensitive outputs and inputs. Place components as close as possible to the pins and reduce trace lengths. See
Figure 33 and Figure 34. It is important for the feedback resistor to be extremely close to the pins, as well as the
series output resistor. The input resistor should also be placed close to the pin. If the amplifier is to be driven in a
noninverting configuration, ground the input close to the device so the current has a short, straight path to the
PowerPAD (gnd).
Too Long
Too Long
R
F
R
I
V
I
−
+
R
O
TPA6120A2
Too Long
Too Long
R
L
Figure 33. Layout That Can Cause Oscillation
Minimized Length of
Feedback Path
Short Trace
Before Resistors
R
F
R
O
V
I
−
+
R
I
R
L
Minimized Length of
the Trace Between
TPA6120A2
Ground as Close to
the Pin as Possible
Output Node and R
O
Figure 34. Layout Designed To Reduce Capacitance On Critical Nodes
Thermal Considerations
Amplifiers can generate quite a bit of heat. Linear amplifiers, as opposed to Class-D amplifiers, are extremely
inefficient, and heat dissipation can be a problem. There is no one to one relationship between output power and
heat dissipation, so the following equations must be used:
P
L
Efficiency of an amplifier +
P
SUP
(1)
Where
2
2
V
V
V
P
LRMS
P
P
+
, and V
+
, therefore, P +
L
per channel
L
LRMS
Ǹ
R
2R
2
L
L
(2)
(3)
P
+ V
I
avg ) V I
CC CC(q)
SUP
CC CC
p
p
V
V
V
2
1
p
P
P
P
2
ŕ
I
+
sin(t) dt + *
[cos(t)]
+
avg
CC
R
pR
p R
L
L
L
0
0
(4)
17
Where
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
APPLICATION INFORMATION (continued)
Ǹ2 P R
+
V
P
L
L
(5)
(6)
Therefore,
V
V
CC
P
P
+
) V
I
CC CC(q)
SUP
p R
L
PL = Power delivered to load (per channel)
PSUP = Power drawn from power supply
VLRMS = RMS voltage on the load
RL = Load resistance
VP = Peak voltage on the load
ICCavg = Average current drawn from the power supply
ICC(q) = Quiescent current (per channel)
VCC = Power supply voltage (total supply voltage = 30 V if running on a ±15-V power supply
η = Efficiency of a SE amplifier
For stereo operation, the efficiency does not change because both PL and PSUP are doubled. This effects the
amount of power dissipated by the package in the form of heat.
A simple formula for calculating the power dissipated, PDISS, is shown in Equation 7:
P
(1
h) P
DISS
SUP
(7)
In stereo operation, PSUP is twice the quantity that is present in mono operation.
The maximum ambient temperature, TA, depends on the heat-sinking ability of the system. θJA for a 20-pin DWP,
whose thermal pad is properly soldered down, is shown in the dissipation rating table.
T
Max
T Max
Θ
P
A
J
JA Diss
(8)
2
Mono Operation
V
CC
= +15 V, R = 32 W
L
1.8
1.6
1.4
1.2
1
V
= +12 V, R = 32 W
L
CC
V
= +15 V,
CC
R
= 64 W
L
0.8
0.6
0.4
V
CC
R
= +12 V,
L
= 64 W
0.2
0
0
0.5
1
1.5
2
2.5
3
3.5
P
O
− Output Power − W
Figure 35. Power Dissipation vs Output Power
18
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
Application Circuit
OPA4134
TPA6120A2
V
V
CC+
V+
V−
CC−
5 V
−5 V
12 V
−12 V
0.1 µF
0.1 µF
10 µF
10 µF
100 µF
10 µF
10 µF
100 µF
C
F
2.7 nF
R
F
1 kW
V−
R
F
1 kW
11
2
−INA
−
OUTA
1
V
3
CC−
+
0.1 mF
R
1 kW
I
4
4
5 V
5
−
V+
LOUT
LIN−
LIN+
4
0.1 µF
2
C
2.7 nF
R
O
10 W
ZEROL
ZEROR
MSEL
LRCK
DATA
BCK
V
2L
1
2
3
F
28
27
26
25
24
23
22
21
20
19
18
17
16
15
10 µF
CC
+
3
R
1 kW
I
0.1 mF
AGND3L
R
F
1 kW
V
CC+
R
F
1 kW
I
I
L−
OUT
V−
11
−INB
5 V
6
−
L+
4
OUT
7
5
PCM
Audio
+
OUTB
AGND2
5
4
Data
Source
V+
V
1
CC
6
10 µF
47 µF
+
+
SCK
V L
COM
7
PCM1792
C
F
2.7 nF
DGND
V
COM
R
8
0.1 µF
47 µF
V
DD
I
REF
9
R
F
1 kW
10 kΩ
MS
AGND1
10
11
V−
R
F
9
11
1 kW
−
MDI
MC
I
I
R−
R+
OUTC
OUT
8
−INC
10
V
CC−
+
0.1 mF
Controller
12
R
I
1 kW
OUT
4
20
0.1 µF
5 V
16
−
V+
AGND3R
2R
MDO
RST
13
14
ROUT
RIN−
C
2.7 nF
F
19
V
CC
RIN+
R
O
10 W
+
18
10 µF
17
0.1 mF
R
I
R
F
1 kW
1 kW
3.3 V
+
V
CC+
R
F
V−
1 kW
13
11
−
14
10 µF
−IND
12
+
OUTD
4
V+
Figure 36. Typical Application Circuit
In many applications, the audio source is digital. It must go through a digital-to-analog converter (DAC) so that
traditional analog amplifiers can drive the speakers or headphones.
Figure 36 shows a complete circuit schematic for such a system. The digital audio is fed into a high performance
DAC. The PCM1792, a Burr-Brown product from TI, is a 24-bit, stereo DAC.
The output of the PCM1792 is current, not voltage, so the OPA4134 is used to convert the current input to a
voltage output. The OPA4134, a Burr-Brown product from TI, is a low-noise, high-speed, high-performance
operational amplifier. CF and RF are used to set the cutoff frequency of the filter. The RC combination in
Figure 36 has a cutoff frequency of 59 kHz. All four amplifiers of the OPA4134 are used so the TPA6120A2 can
be driven differentially.
19
TPA6120A2
www.ti.com
SLOS431–MARCH 2004
The output of the OPA4134 goes into the TPA6120A2. The TPA6120A2 is configured for use with differential
inputs, stereo use, and a gain of 2V/V. Note that the 0.1-uF capacitors are placed at every supply pin of the
TPA6120A2, as well as the 10-Ω series output resistor.
Each output goes to one channel of a pair of stereo headphones, where the listener enjoys crisp, clean, virtually
noise free music with a dynamic range greater than the human ear is capable of detecting.
20
PACKAGE OPTION ADDENDUM
www.ti.com
18-Apr-2006
PACKAGING INFORMATION
Orderable Device
Status (1)
Package Package
Pins Package Eco Plan (2) Lead/Ball Finish MSL Peak Temp (3)
Qty
Type
Drawing
TPA6120A2DWP
ACTIVE
SO
Power
PAD
DWP
20
20
20
25 Green (RoHS & CU NIPDAU Level-2-260C-1 YEAR
no Sb/Br)
TPA6120A2DWPG4
TPA6120A2DWPR
ACTIVE
ACTIVE
SO
Power
PAD
DWP
DWP
25 Green (RoHS & CU NIPDAU Level-2-260C-1 YEAR
no Sb/Br)
SO
Power
PAD
2000 Green (RoHS & CU NIPDAU Level-2-260C-1 YEAR
no Sb/Br)
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and
package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS
compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
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In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI
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Addendum-Page 1
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