TPS40021_技术文档

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

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FEATURES

DESCRIPTION

D

Operating Input Voltage 2.25 V to 5.5 V

The TPS4002x family of dc-to-dc controllers are designed

for non-isolated synchronous buck regulators, providing

enhanced operation and design flexability through user

programmability.

Output Voltage as Low as 0.7 V

1% Internal 0.7 V Reference

Predictive Gate Drive N-Channel MOSFET

Drivers for Higher Efficiency

The TPS4002x utilizes a proprietary Predictive Gate

Drive technology to minimize the diode conduction

losses associated with the high-side and synchronous

rectifier N-channel MOSFET transistions. The integrated

charge pump with boost circuit provides a regulated 5-V

gate drive for both the high side and synchronous rectifier

N-channel MOSFETs. The use of the Predictive Gate

Drive technology and charge pump/boost circuits

combine to provide a highly efficient, smaller and less

expensive converter.

Externally Adjustable Soft-Start and Short

Circuit Current Limit

Programmable Fixed-Frequency

100 KHz-to-1 MHz Voltage-Mode Control

D

Source-Only Current or Source/Sink Current

D

Quick Response Output Transient

Comparators with Power Good Indication

Provide Output Status

Design flexibility is provided through user programmability

of such functions as: operating frequency, short circuit

current detection thresholds, soft-start ramp time, and

external synchronization frequency. The operating

frequency is programmable using a single resistor over a

frequency range of 100 kHz to 1 MHz. Higher operating

frequencies yield smaller component values for a given

converter power level as well as faster loop closure.

D

16-Pin PowerPAD Package

APPLICATIONS

D

Networking Equipment

Telecom Equipment

Base Stations

Servers

DSP Power

VDD

VOUT

TPS40020

VDD 2.25 V − 5.5 V

ILIM/

SYNC

1

2

3

4

5

6

7

8

16

15

BOOT1

HDRV

VOUT

VDD

OSNS

FB

SW 14

BOOT2 13

PVDD 12

LDRV 11

PGND 10

COMP

SS/SD

RT

SGND

9

PWRGD

UDG−02094

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments

semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD and Predictive Gate Drive are trademarks of Texas Instruments.

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Copyright  2004, Texas Instruments Incorporated

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

DESCRIPTION (CONTINUED)

The short circuit current detection is programmable through a single resistor, allowing the short circuit current limit

detection threshold to be easily tailored to accommodate different size (R ) MOSFETs. The short circuit current

DS(on)

function provides pulse-by-pulse current limiting during soft-start and short term transient conditions as well as a fault

counter to handle longer duration short circuit current conditions. If a fault is detected the controller shuts down for

a period of time determined by six (6) consecutive soft-start cycles. The controller automatically retries the output

th

every seventh (7 ) soft-start cycle.

In addition to determining the off time during a fault condition, the soft-start ramp provides a closed loop controlled

ramp of the converter output during startup. Programmability allows the ramp rate to be adjusted for a wide variety

of output L-C component values.

The output voltage transient comparators provide a quick response , first strike, approach to output voltage transients. The

output voltage is sensed through a resistor divider at the OSNS pin. If an overvoltage condition is detected the HDRV gate

drive is shut-off and the LDRV gate drive is turned on until the output is returned to regulation. Similarly, if an output

undervoltage condition is sensed the HDRV gate drive goes to 95% duty cycle to pump the output back up quickly. In either

case, the PowerGood open drain output pulls low to indicate an output voltage out of regulation condition. The PowerGood

output can be daisy-chained to the SS/SD pin or enable pin of other controllers or converters for output voltage sequencing.

The transient comparators can be disabled by simply tying the OSNS pin to VDD.

The TPS4002x can be externally synchronized through the ILIM/SYNC pin up to 1.5× the free-running frequency. This

allows multiple contollers to be synchronized to eliminate EMI concerns due to input beat frequencies between controllers.

INTERNAL BLOCK DIAGRAM

VDD

2

3

VDD

0.719 V

OSNS

VDD

13 BOOT2

12 PVDD

16 BOOT1

15 HDRV

14 SW

SS

CHARGE

PUMP

ACTIVE

PWRGD

9

0.659 V

FB

4

5

0.69 V

+

DRV

PREDICTIVE

GATE

PWM

COMP

DRIVE(tm)

PWM

UVLO

OSC

CLK

LOGIC

PVDD

RT

7

UVLO

IRT

DRV

11 LDRV

10 PGND

FAULT

CLK

VDD

IRT

SS

ACTIVE

CURRENT LIMIT

COMPARATOR

SOFT

START

FAULT

COUNTER

I

−

SS

OC

SS/SD

SGND

6

8

1

ILIM/SYNC

DCHG

+

UVLO

SD

SYNC

0.28 V

1 V

VDD

UVLO

+

DISABLE

V

DD

1.4 V

UDG−02092

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during

storage or handling to prevent electrostatic damage to the MOS gates.

ORDERING INFORMATION

(1)

T

LOAD CURRENT

PACKAGE

PART NUMBER

TPS40020PWP

TPS40021PWP

A

(2)

SOURCE

Plastic HTSSOP (PWP)

−40°C to 85°C

SOURCE/SINK

Plastic HTSSOP (PWP)

(1)

(2)

See page 7 for explanation.

The PWP package is also available taped and reeled. Add an R suffix to the device type (i.e., TPS40020PWPR). See the application section of

the data sheet for PowerPAD drawing and layout information.

ABSOLUTE MAXIMUM RATINGS

over operating free-air temperature range unless otherwise noted

(4)

TPS4002X

UNIT

SS/SD, VDD, PVDD, OSNS

BOOT2, BOOT1

SW

−0.3 to 6

V

SW

+ 6

−3.0 to 10.5

−5

Input voltage range, V

IN

V

SWT (SW transient < 50 ns)

FB, ILIM

−0.3 to 6.0

−0.3 to 6

10

Output voltage range, V

OUT

COMP, PWRGD, RT

PWRGD

Sink current, I

mA

S

Operating virtual junction temperature range, T

−40 to 125

−55 to 150

260

J

Storage temperature, T

stg

°C

Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds

(4)

Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only,

and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is

not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

RECOMMENDED OPERATING CONDITIONS

MIN NOM MAX UNIT

Input voltage, V

IN

2.25

−40

5.5

85

V

Operating junction temperature, T

°C

J

(5)(6)

PWP PACKAGE

(TOP VIEW)

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

ILIM/SYNC

VDD

BOOT1

HDRV

SW

BOOT2

PVDD

LDRV

OSNS

FB

COMP

SS/SD

RT

THERMAL

PAD

PGND

PWRGD

SGND

(5) For more information on the PWP package, refer to TI Technical Brief, Literature No. SLMA002.

(6) PowerPADt heat slug must be connected to SGND (Pin 8), or electrically isolated from all other pins.

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

ELECTRICAL CHARACTERISTICS

T = −40°C to 85°C, T = T

V

= 5.0 V (unless otherwise noted)

J

A, DD

PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

UNIT

INPUT SUPPLY

V

Input voltage range, VDD

PVDD pin voltage

Switching current

Quiescent current

Shutdown current

Minimum on-voltage

Hysteresis

2.25

5.50

5.2

DD

V

= 3.3 V

4.9

3.5

PVDD

DD

500 kHz, No load on HDRV, LDRV

FB = 0.8 V

5.0

2.0

3.0

I

mA

DD

SS/SD = 0 V, Outputs OFF

0.38

2.05

130

1.00

2.15

200

1.95

80

V

UVLO

mV

OSCILLATOR

2.25 V ≤ V

≤ 5.00 V,

R

= 69.8 kΩ

= 34.8 kΩ

425

800

500

950

575

1100

1.07

0.41

DD

T

f

Accuracy

kHz

V

OSC

R

T

DD

V

Ramp voltage

V

−V

0.80

0.24

0.93

0.31

RAMP

PEAK VAL

Ramp valley voltage

VAL

PWM

V

= V , R = 34.8 kΩ,

DD

OSNS

DD

T

85%

90%

94%

95%

= 3.3 V, FB = 0 V

d

Maximum duty cycle

MAX

V

= V , R = 70 kΩ,

DD

OSNS

DD

T

= 5.0 V, FB = 0 V

Minimum duty cycle

0%

MIN

(2)

t

Minimum HDRV on-time

250

ns

ERROR AMPLIFIER

V

Feedback input voltage

Input bias current

−40°C ≤ T ≤ 85°C, 2.25V ≤ V

DD

≤ 5.00V

0.685 0.690

30

0.697

130

V

FB

A

I

nA

BIAS

V

High-level output voltage

I

= 0.5 mA, V

= GND

2.0

2.5

0.08

7

OH

OL

OH

FB

V

Low-level output voltage

= 0.5 mA, V

= GND

= V

DD

0.15

OL

FB

I

High-level output source current

Low-level output sink current

V

3

OH

FB

mA

= V

DD

8

OL

FB

(1)

Gain bandwidth

G

5

10

85

MHz

dB

BW

(1)

Open loop gain

A

OL

55

CURRENT LIMIT

I

Current limit sink current

2.25 V ≤ V

DD

≤ 5.00 V,

R

T

= 69.8 kΩ

165

−20

190

0

215

20

µA

SINK

V

Current limit offset voltage

mV

OS

ON

SS

t

Minimum HDRV on−time in overcurrent

Switch leading-edge blanking pulse time

Soft-start cycles

V

DD

= 3.3 V

200

140

6

300

ns

(1)

cycles

V

Current limit input voltage range

2

VDD

5.4

ILIM

SOFT START

I

Soft-start source current

Outputs = OFF

2.0

3.3

µA

SS

(1)

(2)

Ensured by design. Not production tested.

Operation below the minimum on-time could result in overlap of the HDRV and LDRV outputs.

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

ELECTRICAL CHARACTERISTICS (continued)

T = −40°C to 85°C, T = T

V

= 5.0 V (unless otherwise noted)

J

A, DD

PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

UNIT

SHUTDOWN

V

Shutdown threshold voltage

0.22

0.25

0.26

0.28

0.29

0.32

SD

V

Device enable threshold voltage

EN

OUTPUT DRIVER

V

− V

(SW)

= 3.3 V,

(BOOT1)

R

High-side driver pull-up resistance

High-side driver pull-down resistance

1.0

0.8

2.5

1.5

5.0

3.0

HDHI

I

100mA

SOURCE =

V

− V

(SW)

100mA

(BOOT1)

HDLO

Ω

I

SINK =

R

Low-side driver pull-up resistance

Low-side driver pull-down resistance

Low-side driver rise time

P

= 3.3 V, I

100 mA

SOURCE =

1.0

2.5

0.80

15

5.0

1.50

35

LDHI

VDD

= 3.3 V, I 100 mA

SINK =

0.45

LDLO

VDD

t

LRISE

LFALL

HRISE

HFALL

Low-side driver fall time

10

25

C

= 1 nF

ns

LOAD

High-side driver rise time

15

35

High-side driver fall time

10

25

THERMAL SHUTDOWN

Shutdown temperature

(1)

165

15

T

SD

°C

Hysteresis

CHARGE PUMP

R

VB2

R

B2P

R

PB1

R

VDD to BOOT2

BOOT2 to PVDD

PVDD to BOOT1

V

DD

V

DD

V

DD

= 5.0 V,

I

10 mA

2.8

2.9

6.6

5.6

5.9

10.4

8.4

DS(on)

SOURCE =

= 5.0 V,

10 mA

Ω

8.9

POWER GOOD

V

= 0.8 V,

I

0.5 mA,

OSNS

= 3.3 V

PWRGD =

V

Pull-down voltage

50

6

90

10

140

14

mV

PGD

DD

Output sense high to power good low delay

time

0.7 V≤ V

≤ 0.8 V, I

≤ 0.7 V, I

0.5 mA,

PWRGD =

OSNS

t

ONHPL

V

DD

= 3.3 V

Output sense low to power good low delay

time

0.6 V≤ V

0.5 mA,

PWRGD =

OSNS

= 3.3 V

t

6

10

14

ONLPL

V

DD

µs

V

= 0.7 V, I

= 3.3 V, 0.0 V ≤ V

0.5 mA,

≤ 0.4 V

OSNS

DD

PWRGD =

t

Shutdown high to power good high delay time

Shutdown low to power good low delay time

2

4

6

SDHPH

SS/SD

V

= 0.7 V, I

= 3.3 V, 0.0 V ≤ V

SS/SD

0.5 mA,

≤ 0.4 V

OSNS

DD

PWRGD =

t

0.5

140

1.5

500

3.0

SDLPL

Output sense high to nominal to power good

high delay time

0.7 V≤ V

≤ 0.8 V, I

0.5 mA,

PWRGD =

OSNS

= 3.3 V

t

1000

ONHPH

V

DD

ns

Output sense low to nominal to power good

high delay time

0.6 V≤ V

≤ 0.7 V, I

0.5 mA,

PWRGD =

OSNS

= 3.3 V

t

ONLPH

V

DD

TRANSIENT COMPARATORS

Overvoltage output threshold voltage

23

8

29

15

35

22

V

OV

Hysteresis

Referenced to V

mV

V

FB

Undervoltage output threshold voltage

Hysteresis

−37

8

−31

15

−25

22

V

UV

OSNS minimum disable voltage

0.5

DIS

DD

(1)

Ensured by design. Not production tested.

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

ELECTRICAL CHARACTERISTICS (continued)

T = −40°C to 85°C, T = T

V

= 5.0 V (unless otherwise noted)

J

A, DD

PARAMETER

SYNCHRONIZATION

TEST CONDITIONS

MIN

TYP

MAX

UNIT

V

Synchronization enable low threshold voltage

0.7

50

ENSY

BLNK

MIN

V

Synchronization current limit enable threshold

voltage

V

Referenced to VDD

−0.7

t

Minimum synchronization input pulse width

35

ns

PREDICTIVE DELAY

V

Sense voltage to modulate delay

Maximum delay modulation

Counter delay/bit time

−200

65

mV

ns

SWP

LDRV OFF-to-HDRV ON

HDRV OFF-to-LDRV ON

40

2.5

55

90

6.2

105

6.5

t

LDHD

4.5

80

Maximum delay modulation

Counter delay/bit time

t

HDLD

2.2

5.0

RECTIFIER ZERO CURRENT COMPARATOR

Sense voltage to turn off

rectifier MOSFET

V

TPS40020 LDRV output = OFF

−5

−2.5

150

2

mV

ns

SW

(1)

t

Zero current blanking time

ZBLNK

(1)

Ensured by design. Not production tested.

TERMINAL FUNCTIONS

TERMINAL

I/O

DESCRIPTION

NAME

NO.

This pin provides a bootstrapped supply for the high side FET driver, enabling the gate of the high side FET to be

driven above the input supply rail. Connect a capacitor from this pin to the SW pin.

BOOT1

16

I

This pin provides a secondary bootstrapping necessary for generation of PVDD. Connect a capacitor from this

pin to SW.

BOOT2

13

COMP

FB

5

4

O

I

Output of the error amplifier. Refer to Electrical Characteristics table for loading constraints.

Inverting input of the error amplifier. In normal operation, V

FB

is equal to the internal reference level of 690 mV.

The gate drive output for the high side N-channel MOSFET switch is bootstrapped to near PVDD for good

enhancementof the high-side switch. The HDRV switches from BOOT1 to SW.

HDRV

15

O

The current limit pin is used to set the current limit threshold. A current sink from this pin to GND sets the threshold

voltage for output short circuit current across a resistor connected to VDD. Synchronization is accomplished by

pulling IMAX to less than 1 V for a period greater than the minimum pulse width and then releasing. An open

collector or drain device should be used. These pulses must be of higher frequency than the free running frequency

of the local oscillator.

ILIM/SYNC

1

I

Gate drive output for the low-side synchronous rectifier N-channel MOSFET. LDRV switches from PVDD to

PGND.

LDRV

OSNS

11

3

O

The output sense pin is connected to a resistor divider from VOUT to GND (identical to the main feedback loop)

and is used to sense power good condition and provides reference for the transient comparators.

PGND

10

9

O

−

Power (high-current) ground used by LDRV.

PWRGD

Power good. This is an open-drain output which connects to the supply via an external resistor.

This pin is the regulated output of the charge-pump and provides the supply voltage for the LDRV driver stage.

PVDD also drives the bootstrap circuit which generates the voltage on BOOT1.

PVDD

12

O

RT

7

8

I

External pin for programming the oscillator frequency. Connnected a resistor between this pin and GND.

Signal ground

SGND

−

The soft-start/shutdown pin provides user programmable soft-start timing and shutdown capability for the

controller.

SS/SD

6

I

This pin, used for overcurrent, zero-current, and in the anti-cross conduction sensing is connected to the switched

node on the converter. Output short circuit is detected by sensing the voltage at this pin with respect to VDD while

the high-side switch is on. Zero current is detected by sensing the pin voltage with respect to ground when the

low-side rectifier MOSFET is on.

SW

14

2

I

VDD

Power input for the device. Maximum voltage is 5.5 V. De-coupling of this pin is required.

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

APPLICATION INFORMATION

The TPS4002x series of devices are low-input voltage, synchronous, voltage mode-buck controllers. A typical

application circuit is shown in Figure 1. These controllers are designed to allow construction of

high-performance dc-to-dc converters with input voltages from 2.25 V to 5.5 V, and output voltages as low as

690 mV. Using a top side N-channel MOSFET for the primary buck switch results in lower switch resistance

for a given gate charge.

The device controls the delays from main switch off to rectifier turn on and from rectifier turn off to main switch

turn on in a way that minimizes diode losses (both conduction and recovery) in the synchronous rectifier. The

reduction in these losses is significant and can mean that for a given converter power level, smaller FETs can

be used, or that heat sinking can be reduced or even eliminated.

The TPS40021 is the controller of choice for most general purpose synchronous buck designs, operating in two

quadrant mode (i.e. source or sink current) full time. This choice provides the best performance for output

voltage load transient response over the widest load current range.

The TPS40020 operates in single quadrant mode (source current only) full time, allowing the paralleling of

converters. Single quadrant operation ensures one converter does pull current from a paralleled converter. A

converter using one of these controllers emulates a non-synchronous buck converter at light loads. When

current in the output inductor attempts to reverse, an internal zero-current detection circuit turns OFF the

synchronous rectifier and causes the current flow in the inductor to become discontinuous. At average load

currents greater than the peak amplitude of the inductor ripple current, the converter returns to operation as

a synchronous buck converter to maximize efficiency.

The controller provides for a coarse short circuit current-limit function that provides pulse-by-pulse current

limiting, as well as integrates short circuit current pulses to determine the existence of a persistant fault state

at the converter output. If a fault is detected, the converter shuts down for a period of time (determined by six

soft-start cycles) and then restarts. The current-limit threshold is adjustable with a single resistor connected

from VDD to the ILIM/SYNC pin. This overcurrent function is designed to protect against catastrophic faults

only, and cannot be guaranteed to protect against all overcurrent conditions.

The controller implements a closed-loop soft start function. Startup ramp time is set by a single external

capacitor connected to the SS/SD pin. The SS/SD pin also doubles as a shutdown function.

VOLTAGE REFERENCE

The bandgap cell is designed with a trimmed, curvature corrected (< 1%) 0.69-V output, allowing output

voltages as low as 690 mV to be obtained.

Oscillator

The ramp waveform is a saw-tooth form at the PWM frequency with a peak voltage of 1.25 V, and a valley of

0.3 V. The PWM duty cycle is limited to a maximum of 97%, allowing the bootstrap and charge pump capacitors

to charge during every cycle.

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APPLICATION INFORMATION

Bootstrap/Charge Pump

The TPS4002X series includes a charge pump to boost the drive voltage to the power MOSFET’s to higher

levels when the input supply is low. A capacitor connected from PVDD to PGND is the storage cap for the pump.

A capacitor connected from SW to BOOT2 gets charged every switching cycle while LDRV is high and its charge

is dumped on the PVDD capacitor when HDRV goes high. An internal switch disables the charge pump when

the voltage on PVDD reaches approximately 4.8 V and enables pumping when PVDD falls to approximately

4.6 V. The high-side driver uses the capacitor from SW to BOOT1 as its power supply. When SW is low, this

capacitor charges from the PVDD capacitor. When the SW pin goes high, this capacitor provides above-rail

drive for the high-side N-channel FET.

PVDD, BOOT1 and BOOT2 are pre-charged to the VDD voltage during a shutdown condition. For low-input

voltage converters, utilizing higher gate threshold voltage MOSFETs, it may be necessary to add an Schottky

diode from VDD (anode) to BOOT1 to guarantee sufficient voltage for initial start up. Once switching starts the

charge pump reverses bias on the Schottky diode.

When operating the TPS40020 under no load or extremely light-load conditions the controller will be operating

in discontinuous Mode (DCM); reverse current is prevented from flowing in the synchronous rectifier. In DCM

the on times for both the HDRV and LDRV pulses can become too narrow to provide adequate charging of

PVDD and BOOT1 outputs, causing their voltages to collapse. Insufficient PVDD and BOOT1 voltages prevent

the external MOSFETS from becomming fully enhanced, causing loss of converter output regulation. Schottky

diodes from VIN (anode) to PVDD, and VIN (anode) to BOOT1, as well as a pre-load can be added to maintain

PVDD and BOOT1 at voltage levels sufficient enough to fully enhance the external MOSFETs. The amount of

pre-load typically ranges from 50 mA to 100 mA depending on operating conditions and external MOSFET

selection.

Drivers

The HDRV and LDRV MOSFET drivers are capable of driving gate-to-source voltages up to 5.0 V. Using

appropriate MOSFETs, a 25-A converter can be achieved. The LDRV driver switches between VDD and

ground, while the HDRV driver is referenced to SW and switches between BOOT1 and SW. The maximum

voltage between BOOT1 and SW is 5.0 V when PVDD is in regulation.

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APPLICATION INFORMATION

Figure 1. Typical Application

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APPLICATION INFORMATION

Synchronous Rectification and Predictive Delay

In a normal buck converter, when the main switch turns off, current is flowing to the load in the inductor. This

current cannot be stopped immediately without using infinite voltage. To give this current a path to flow and

maintain voltage levels at a safe level, a rectifier or catch device is used. This device can be either a plain diode,

or it can be a controlled active device if a control signal is available to drive it. The TPS4002X provides a signal

to drive an N−channel MOSFET as a rectifier. This control signal is carefully coordinated with the drive signal

for the main switch so that there is absolute minimum dead time from the time that the rectifier FET turns off

and the main switch turns on, and minimum delay from when the main switch turns off and the rectifier FET

turns on. This TI−patented function, predictive delay, uses information from the current switching cycle to adjust

the delays that are used for the next cycle. Figure 2 shows the switch-node voltage waveform for a

synchronously rectified buck converter. Illustrated are the relative effects of a fixed delay drive scheme

(constant, pre-set delays for the turnoff to turn on intervals), an adaptive delay drive scheme (variable delays

based upon voltages sensed on the current switching cycle) and the predictive delay drive scheme. Note that

the longer the time spent in diode conduction during the rectifier conduction period, the lower the efficiency. Also,

not shown in the figure, is the fact that the predictive delay circuit can actually prevent the body diode from

becoming forward biased at all while at the same time avoiding cross conduction or shoot through. This results

in a significant power savings when the main FET turns on. There is no reverse recovery loss in the body diode

of the rectifier FET.

GND

Channel Conduction

Body Diode Conduction

Fixed Delay

Adaptive Delay

Predictive Delay

UDG−01144

Figure 2. Switch Node Waveforms for Synchronous Buck Converter

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APPLICATION INFORMATION

Output Short Circuit Protection

Output short circuit protection in the TPS4002x is sensed by looking at the voltage across the main FET while

it is on. If the voltage exceeds a pre-set threshold, the current pulse is terminated, and a counter inside the

device is incremented. If this counter fills up, a fault condition is declared and the chip disables switching for

a period of time and then attempts to restart the converter with a full soft-start cycle. The more detailed

explanation follows.

In each switching cycle, a comparator looks at the voltage across the top side FET while it is on. If the voltage

across that FET exceeds a programmable threshold voltage, then the current switching pulse is terminated and

a 3-bit counter (eight counts) is incremented by one count. If during the switching cycle the top side FET voltage

does not exceed a preset threshold, then this counter is decremented by one count. (The counter does not wrap

around from seven to zero or from zero to seven). If the counter reaches a full count of seven, the device

declares that a fault condition exists at the output of the converter. In this state, switching stops and the soft-start

capacitor is discharged. The counter is decremented by one by the soft start cap discharge. When the soft-start

capacitor is fully discharged, the discharge circuit is turned off and the cap is allowed to charge up at the nominal

charging rate, When the soft-start capacitor reaches approximately 1.3 V, it is discharged again and the

overcurrent counter is decremented by one count. The capacitor is charged and discharged, and the counter

decremented until the count reaches zero (a total of six times). When this happens, the outputs are again

enabled as the soft-start capacitor generates a reference ramp for the converter to follow while attempting to

restart. During this soft-start interval (whether or not the controller is attempting to do a fault recovery or starting

for the first time), pulse-by-pulse current limiting is in effect, but overcurrent pulses are not counted to declare

a fault until the soft-start cycle has been completed. It is possible to have a supply try to bring up a short circuit

for the duration of the soft-start period plus seven switching cycles. Power stage designs should take this into

account if it makes a difference thermally. Figure 3 shows the details of the overcurrent operation.

(+)

V

TS

(−)

Overcurrent

Threshold

Voltgage

Internal PWM

V

TS

0 V

External

Main Drive

Normal Cycle

Overcurrent

Cycle

UDG−03029

Figure 3. Switch Node Waveforms for Synchronous Buck Converter

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APPLICATION INFORMATION

Figure 4 shows the behavior of key signals during initial startup, during a fault and a successfully fault recovery.

At time t0, power is applied to the converter. The voltage on the soft-start capacitor (V ) begins to ramp up

CSS

At t1, the soft-start period is over and the converter is regulating its output at the desired voltage level. From

t0 to t1, pulse-by-pulse current limiting was in effect, and from t1 onward, overcurrent pulses are counted for

purposes of determining if a fault exists. At t2, a heavy overload is applied to the converter. This overload is

in excess of the overcurrent threshold, the converter starts limiting current and the output voltage falls to some

level depending on the overload applied. During the period from t2 to t3, the counter is counting overcurrent

pulses and at time t3 reaches a full count of 7. The soft-start capacitor is then discharged, the outputs are

disabled, the counter decremented, and a fault condition is declared.

V

DD

ꢀ 1.3 V

ꢀ 0.6 V

0.6 V

V

CSS

FAULT

I

LOAD

V

OUT

t

t4

t0

t1

t2 t3

t5

t6

t7

t8

t9

t10

Counter

0

6

5

4

3

2

1

0

1

2

3

4

5

6

7

cycles

UDG−03187

Figure 4. Overcurrent/Fault Waveforms

When the soft-start capacitor is fully discharged, it begins charging again at the same rate that it does on startup,

with a nominal 3-µA current source. As the capacitor voltage reaches full charge, it is discharged again and the

counter is decremented by one count. These transitions occur at t3 through t9. At t9, the counter has been

decremented to zero. Now the fault logic is cleared, the outputs are enabled and the converter attempts to

restart with a full soft-start cycle. The converter comes into regulation at t10.

The internal SS signal is a diode drop below V

. When V

reaches one diode drop above ground, (≅0.6 V)

CSS

the output (V

) begins it’s soft-start ramp.

OUT

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APPLICATION INFORMATION

Setting the Short Circuit Current Limit Threshold

Connecting a resistor from VDD to ILIM sets the current limit. A current sink in the chip causes a voltage drop

across the resistor connected to ILIM. This voltage drop is the short circuit current threshold for the part. The

current that the ILIM pin sinks is dependent on the value of the resistor connected to RT and is given by:

0.69 V

I

+ 19.0

ILIM

R

T

(1)

The tolerance of the current sink is too loose to do an accurate current limit. The main purpose is for hard fault

protection of the power switches. Given the tolerance of the ILIM sink current, and the R range for a

DS(on)

MOSFET, it is generally possible to apply a load that thermally damages the converter. This device is intended

for embedded converters where load characteristics are defined and can be controlled. A small capacitor can

be added between ILIM and VDD for filtering. However, capacitors should not be used if the synchronization

function is to be used.

Soft-Start and Shutdown

The soft-start and shutdown functions are common to the SS/SD pin. The voltage at this pin is the controlling

voltage sent to the error amplifier during startup. This reduces the transient current required to charge the output

capacitor at startup, and allows for a smooth startup with no overshoot of the output voltage. A shutdown feature

can be implemented as shown in Figure 5.

3.3 µA

6

SS/SD

C

SS

SHUTDOWN

TPS4002x

Figure 5. Shutdown Implementation

I

SS

C

+

t

(F)

SS

V

FB

(2)

where

D t is the start up time in seconds

SS

Switching Frequency

The switching frequency is programmed by a resistor from RT to SGND. Nominal switching frequency can be

calculated by:

3

37.736 10

(

)

R (kW) +

* 5.09 kW

T

(

)

kHz

f

OSC

(3)

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APPLICATION INFORMATION

Synchronization

The TPS4002x can be synchronized to an external reference frequency higher than the free running oscillator

frequency. The recommended method is to use a diode and a push pull drive signal as shown in Figure 6.

PREFERRED

VDD

ALTERNATE

VDD

Minumize

Output/Stray

Capacitance

on ILIM Node

TPS4002XPWP

1

ILIM/SYNC

VDD

1

2

ILIM/SYNC

VDD

2

50 ns to 100 ns

UDG−03032

50 ns to 100 ns

Figure 6. Synchronization Methods

This design allows synchronization up to the maximum operating frequency of 1 MHz. For best results the

nominal operating frequency of a converter that is to be synchronized should be kept as close as practicable

to the synchronization frequency to avoid excessive noise induced pulse width jitter. A good target is to shoot

for the free run frequency to be 80% of the synchronized frequency. This ensures that the synchronization

source is the frequency determining element in the system and not to adversely affect noise immunity.

Other methods of implementing the synchronization function include using an open collector or open drain

output device directly, or discreet devices to pull the ILIM/SYNC pin down. These do work but performance can

suffer at high frequency because the ILIM/SYNC pin must rise to (V

− 1.0 V) before the next switching cycle

DD

begins. Any time that this requires is directly subtracted from the maximum pulse width available and should

be considered when choosing devices to drive ILIM/SYNC. Consequently, the lowest output capacitance

devices work best.

During a synchronization cycle, the current sink on the ILIM/SYNC pin becomes disabled when ILIM/SYNC is

pulled below 1.0 V. The ILIM/SYNC current sink remains disabled until ILIM/SYNC reaches (V

−1.0 V) This

DD

removes the load on the ILIM/SYNC pin to allow the voltage to slew rapidly depending on the ILIM resistor and

any stray capacitance on the pin. To maximize this slew rate, minimize stray capacitance on this pin.

The duration of the synchronization pulse pulling ILIM/SYNC low shoud be between 50 ns and 100 ns. Longer

durations may limit the maximum obtainable duty cycle.

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APPLICATION INFORMATION

Transient Comparators and Power Good

The TPS4002x makes use of a separate pin, OSNS, to monitor output voltage for these two functions. In normal

operation, OSNS is connected to the output via a resistor divider. It is important to make this divider the same

ratio as the divider for the feedback network so that in normal operation the voltage at OSNS is the same as

the voltage at FB, 0.69 V nominal.

The PWRGD pin is an open drain output that is pulled low when the voltage at OSNS falls outside 0.69 V 4.6%

(approximately). A delay has been purposely built into the PWRGD pin pulling low in response to an out of band

voltage on OSNS, to minimize the need for filtering the signal in the event of a noise glitch causing a brief out

of band OSNS voltage. The PWRGD signal returns to high when the OSNS signal returns to approximately 1%

of nominal (0.69 V 1%).

The transient comparators override the conventional voltage control loop when the output voltage exceeds a

4.6% window. If the output transition is high (i.e. load steps down from 90% load to 10 % load) then the HDRV

gate drive is terminated, 0% duty cycle, the LDRV gate drive is turned on to sink output current until V

returns

OUT

to within 1% of nominal. Conversely when V

drops outside the window (i.e. step load increases from 10%

OUT

load to 90% load) HDRV increases to maximum duty cycle until V

Figure 7.)

returns to within 1% of nominal. (See

OUT

During start-up, the transient comparators control the state of PWRGD as previously described. However, the

operation of the gate drive outputs is not affected. (See Figure 8)

The transient comparators provide an improvement in load transient recovery time if used properly. In some

situations, recovery time may be one half of the time required without transient comparators. Keep in mind that

the transient comparator concept is a double-edged sword. While they provide improved transient recovery

time, they can also lead to instability if incorrectly applied. For proper functionality, design a feedback loop for

the converter that places the closed loop unity gain frequency at least five times higher than the 0 dB frequency

of the output L-C filter. If not, the feedback loop cannot respond to the ring of the L-C on a transient event. The

ring is likely to be large enough to disturb the transient comparators and the result is a power oscillator. Another

helpful action is to ground the feedback loop divider and the OSNS divider at the SGND pin. Make sure both

dividers measure the same physical location on the output bus. These help avoid problems with resistive drops

at higher loads causing problems.

Connecting OSNS to VDD disables the transient comparators. This also disables the PWRGD function.

Alternatively, OSNS and FB can be tied together. This connection allows a proper PWRGD at startup, though

transient performance diminishes.

< 10 µs

4.6%

1%

FB

−1%

− 4.6%

10 µs

PWRGD

µ

500 n s

10

s

SW

98 % Duty Cycle

0 % Duty Cycle

98 % Duty Cycle

0 % Duty Cycle

UDG−03181

Figure 7. Duty Cycle Waveforms

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APPLICATION INFORMATION

V

DD

1.3 V

0.6 V

0.3 V

SS/SD

V

OUT

− 1%

500 ns

V

OUT

1.5 µs

Transient Comparators Enabled

4 µs

PWRGD

Transient Comparators Disabled

UDG−03181

Figure 8. Transient Comparator Waveforms

Layout Considerations

Successful operation of the TPS4002x family of controllers is dependent upon the proper converter layout and

grounding techniques. High current returns for the SR MOSFET’s source, input capacitance, output

capacitance, PVDD capacitance, and input bypass capacitors (if applicable), should be kept on a single ground

plane or wide trace connected to the PGND (pin10) through a short wide trace. Control components connected

to signal ground, as well as the PowerPad thermal pad, should be connected to a single ground plane connected

to SGND (Pin 8) through a short trace. SGND and PGND should be connected at a single point using a narrow

trace.

Proper operation of Predictive Gate Drive technology and I

functions are dependent upon detecting

ZERO

low-voltage thresholds on the SW node. To ensure that the signal at the SW pin accurately represents the

voltage at the main switching node, the connection from SW (pin 14) to the main switching node of the converter

should be kept as short and wide as possible and should ideally be kept on the top level with the power

components. If the SW trace must traverse multiple board layers between the TPS4002x and the main

switching node, multiple vias should be used to minimize the trace impedance.

Gate drive outputs, LDRV and HDRV (pins 11 and 15, respectively) should be kept as short as possible to

minimize inductances in the traces. If the gate drive outputs need to traverse multiple board layers multiple vias

should be used.

Charge pump components, BOOT1, BOOT2, PVDD, and any input bypass capacitors (if required), should be

kept as close as possible to their respective pins. Ceramic bypass capacitors should be used if the input

capacitors are located more than a couple of inches away from the TPS4002X. If a bypass capacitor is not

needed the trace from the input capacitors to VDD (pin2) should be kept as short and wide as possible to

minimize trace impedance. If multiple board layers are traversed multiple vias should be used.

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APPLICATION INFORMATION

Manufacturer’s instructions should be followed for proper layout of the external MOSFETs. Thermal

impedances given in the manufacturer’s datasheets are for a given mounting technique with a specified surface

area under the drain of the MOSFET. PowerPad package information can be found in the APPLICATION

INFORMATION section of this datasheet.

Refer to TPS40021 EVM−001 High Efficiency Synchronous Buck Converter with PWM Controller Evaluation

Module (HPA009) User’s Guide, (Literature No. sluu144A) for a typical board layout.

The PowerPAD package provides low thermal impedance for heat removal from the device. The PowerPAD

derives its name and low thermal impedance from the large bonding pad on the bottom of the device. The circuit

board must have an area of solder-tinned-copper underneath the package. The dimensions of this area

depends on the size of the PowerPAD package. For a 16-pin TSSOP (PWP) package the area is

5 mm x 3.4 mm [3].

X: Minimum PowerPAD = 1.8 mm

Y: Minimum PowerPAD = 1.4 mm

Thermal Pad

6,60 mm

4,50 mm

X

4,30 mm 6,20 mm

1

8

Y

Figure 9. PowerPAD Dimensions

Thermal vias connect this area to internal or external copper planes and should have a drill diameter sufficiently

small so that the via hole is effectively plugged when the barrel of the via is plated with copper. This plug is

needed to prevent wicking the solder away from the interface between the package body and the solder-tinned

area under the device during solder reflow. Drill diameters of 0.33 mm (13 mils) works well when 1-oz copper

is plated at the surface of the board while simultaneously plating the barrel of the via. If the thermal vias are

not plugged when the copper plating is performed, then a solder mask material should be used to cap the vias

with a diameter equal to the via diameter of 0.1 mm minimum. This capping prevents the solder from being

wicked through the thermal vias and potentially creating a solder void under the package. Refer to PowerPAD

[3]

Thermally Enhanced Package for more information on the PowerPAD package.

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TYPICAL CHARACTERISTICS

OSCILLATOR FREQUENCY

vs

JUNCTION TEMPERATURE

MAXIMUM DUTY CYCLE

vs

JUNCTION TEMPERATURE

99

1000

950

900

850

800

750

700

650

600

550

500

450

98

97

R

T

= 35 kΩ

96

95

V

= 3.3 V, R = 69.8 kΩ

T

DD

R

T

= 69.8 kΩ

94

93

V

DD

= 5 V, R = 35 kΩ

T

−50

−25

0

25

50

75

100

125

−50

−25

0

25

50

75

100

125

T

J

− Junction Temperature − °C

T

J

− Junction Temperature − °C

Figure 11

Figure 10

SHUTDOWN SUPPLY CURRENT

vs

REFERENCE VOLTAGE

vs

JUNCTION TEMPERATURE

0.700

0.675

697

695

693

0.650

0.625

0.600

0.575

691

689

687

685

683

0.550

0.525

0.500

−50

−25

0

25

50

75

100

125

−50

−25

0

25

50

75

100

125

T

J

− Junction Temperature − °C

T

J

− Junction Temperature − °C

Figure 12

Figure 13

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TYPICAL CHARACTERISTICS

SOFT-START CURRENT

vs

JUNCTION TEMPERATURE

TIMING RESISTANCE

vs

SWITCHING FREQUENCY

3.50

3.45

3.40

1000

900

800

700

600

V

IN

= 3.9 V

3.35

3.30

3.25

500

400

3.20

3.15

3.10

300

200

3.05

3.00

100

20

40

60

80

100

120

140

160

−50

−25

0

25

50

75

100

125

R

T

− Timing Resistance − kΩ

T

J

− Junction Temperature − °C

Figure 14

Figure 15

SHUTDOWN THRESHOLD VOLTAGE

ILIM OFFSET VOLTAGE

vs

JUNCTION TEMPERATURE

vs

JUNCTION TEMPERATURE

0.30

20

15

10

5

0.29

0.28

V

DD

= 2.0 V

Enable

0.27

0.26

0.25

0

Disable

0.24

0.23

V

= 3.2 V

DD

−5

−10

−15

0.22

0.21

0.20

V

= 4.9 V

DD

−20

−50

−25

0

25

50

75

100

125

−25

0

25

50

75

100

125

T

J

− Junction Temperature − °C

T

J

− Junction Temperature − °C

Figure 16

Figure 17

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TYPICAL CHARACTERISTICS

ILIM SINK CURRENT

vs

JUNCTION TEMPERATURE

200

V

= 3.3 V

= 1.5 V

= 300 kHz

DD

OUT

V

R

= 3 V

DD

= 69.8 kΩ

V

OUT

198

196

T

f

SW

194

192

190

LDRV

SW

188

186

184

182

180

t − Time − 1 µs/div

−50

−25

0

25

50

75

100

125

T

J

− Junction Temperature − °C

Figure 19. TPS40020 Discontinuous Mode (DCM)

Figure 18

V

DD

= 3.3 V

DD

= 1.5 V

f

SW

= 300 kHz

SS/SD

V

OUT

V

OUT

LDRV

SW

t − Time − 1 µs/div

t − Time − 20 ms/div

Figure 21. Output Current Fault Operation

Figure 20. TPS40020 I

Detection − DCM

ZERO

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TYPICAL CHARACTERISTICS

V

= 3.3 V

V

= 3.3 V

DD

SS/SD

= 1.5 V

= 300 kHz

= 1.5 V

= 300 kHz

OUT

f

I

f

I

SW

PVDD

= 5 A

LOAD

SW

V

OUT

PWRGD

V

OUT

t − Time − 25 µs/div

t − Time − 1 ms/div

Figure 22. Start−Up Operation Without

Transient Comparators

Figure 23. PVDD Hysteresis

V

= 3.3 V

= 1.5 V

= 300 kHz

= 5 A

DD

V

= 3.3 V

DD

SD

V

OUT

= 1.5 V

= 300 kHz

OUT

f

SW

f

I

SW

I

LOAD

= 5 A

LOAD

SS/SD

COMP

SW

VOUT

PWRGD

t − Time − 1 ms/div

t − Time − 200 µs/div

Figure 24. Start−Up Operation With

Transient Comparators

Figure 25. COMP Shutdown Operation

21

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

TYPICAL CHARACTERISTICS

V

= 3.3 V

= 1.5 V

= 300 kHz

DD

OUT

V

f

= 3.3 V

DD

ILIM

SW

= 1.5 V

OUT

SYNC

f

SW

= 330 kHz

SS/SD

PWRGD

LDRV

t − Time − 500 ns/div

t − Time − 1 µs/div

Figure 26. PWRGD Shutdown Operation

Figure 27. External Synchronization

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

REFERENCE DESIGN

This design used the TPS40020 PWM controller to facilitate a step-down application from 3.3-V to 1.5 V. (see

Figure 29) Design specifications include:

D Input voltage: 2.5 V ≤ V ≤ 5.0 V

IN

D Nominal output voltage: 3.3 V

D Output voltage V

D Output current I

: 1.5 V

OUT

: 20 A

OUT

D Switching frequency: 300 kHz

DESIGN PROCEDURE

Setting the Frequency

Choosing the optimum switching frequency is complicated. The higher the frequency, the smaller the

inductance and capacitance needed, so the smaller the size, but then the the switching losses are higher, the

efficiency is poorer. For this evaluation module, 300 kHz is chosen for reasonable efficiency and size.

A resistor R4, which is connected from pin 7 to ground, programs the oscillator frequency. The approximate

operating frequency is calculated in equation (3)

3

37.736 10

(

)

R (kW) +

* 5.09 kW

T

(

)

kHz

f

OSC

(4)

Using equation (2), R is calculated to be 120 kΩ and a 118-kΩ resistor is chosen for 300 kHz operation.

T

Inductance Value

The inductance value can be calculated by equation (2).

V

OUT

L

+

1 *

ǒ Ǔ

(min)

f I

V

RIPPLE

IN(max)

(5)

where I

losses.

is the ripple current flowing through the inductor, which affects the output voltage ripple and core

RIPPLE

Based on 24% ripple current and 300 kHz, the inductance value is calculated to 0.71 µH and a 0.75-µH inductor

(part number is CDEP149−0R7) is chosen. The DCR of this inductor is 1.1 mΩ and the loss is 440 mW, which

is approximately 1.5% of output power.

I

RIPPLE

C

+

OUT(min)

8 f V

RIPPLE

(6)

(7)

V

RIPPLE

ESR

+

OUT

I

With 1.2% output voltage ripple, the needed capacitance is at least 109 µF and its ESR should be less than

3.75 mΩ. Three 2-V, 470-µF, POSCAP capacitors from Sanyo are used. The ESR is 10 mΩ each.

The required input capacitance is calculated in equation (5). The calculated value is approximately 390 µF for

a 100-mV input ripple. Three 6.0-V, 330-µF POSCAP capacitors with 10 mΩ ESR are used to handle 10 A of

RMS input current. Additionally, two ceramic capacitors are used to reduce the switching ripple current.

1

V

C

+ I

D

IN(min)

OUT(max)

(max)

f

OSC

IN(ripple)

(8)

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

REFERENCE DESIGN

UDG−03031

Figure 28. Reference Design Schematic

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

REFERENCE DESIGN

Input and Output Capacitors

The output capacitance and its ESR needed are calculated in equations (5) and (6).

Compensation Design

Voltage-mode control is used in this evaluation module, using R2, R7, R8, C14, C15, and C16 to form a Type-III

compensator network. The L-C frequency of the power stage is approximately 4.9-kHz and the ESR-zero is

around 34 kHz. The overall crossover frequency, f

, is chosen at 43-kHz for reasonable transient response

0db

and stability. Two zeros f and f from the compensator are set at 2.4 kHz and 4 kHz. The two poles, f and

Z1

Z2

P1

f

are set at 34 kHz and 115 kHz. The frequency of poles and zeros are defined by the following equations:

P2

1

f

+

Z1

Z2

P1

P2

2p R7 C14

(9)

1

(assuming R2 ơ R8)

2p R2 C16

(10)

(11)

(12)

1

2p R8 C16

1

(assuming C14 ơ C15)

2p R7 C15

The transfer function for the compensator is calculated in equation (10).

(

)

[

(

)]

^{1 ) s}ǒ^{C14 R7 1 ) s C16 R2 ) R3}

Ǔ_{) s R7 C15 1 ) s R8 C16}

A(s) +

C15

C14

ƪ _{1 )}

ƫ

(

)

s R2 C14

(13)

Figure 30 shows the close loop gain and phase. The overall crossover frequency is approximately 30 kHz. The

phase margin is 57°.

OVERALL GAIN AND PHASE

vs

OSCILLATOR FREQUENCY

50

200

Gain

40

30

150

100

20

10

0

50

0

Phase

−10

−20

−30

−40

−50

−100

−50

100

−150

100 k

1 k

10 k

f

− Oscillator Frequency − kHz

OSC

Figure 29.

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

REFERENCE DESIGN

MOSFETs and Diodes

For a 1.5-V output voltage, the lower the R

of the MOSFET, the higher the efficiency. Due to the high

DS(on)

current and high conduction loss, the MOSFET should have very low conduction resistance (R

) and

DS(on)

thermal resistance. Si7858DP is chosen for its low R

(between 3 mΩ and 4 mΩ) and Power-Pak package.

DS(on)

Current Limiting

Resistor R3 sets the short current limit threshold. The R

of the upper MOSFET is used as a current sensor.

DS(on)

The current limit, I

is initialized at 30% above the maximum output current, I

, which is 28 A. Then

OUT(CL)

OUT(max)

R3 can be calculated in equation (11) and yields a value of 1.4 kΩ. An R3 of 1.43 kΩ is selected.

V

0.69 V

118 kW

FB

₊ǒ₁₉

Ǔ_{+ 111.1 mA}

(

)

₊ǒ Ǔ

I

19

LIM

R4

(14)

(15)

K R

I

DS(on)

I

OUT(CL)

1.5 0.004 28 A

(

)

R3 +

+

+ 1.4 kW

I

LIM

where

D R

is the on-resistor of Q1 (4 mΩ)

DS(on)

D Temperature coefficient, K=1.5

D V = 0.69 V

FB

D R4=118 kΩ

Voltage Sense Regulator

R1 and R2 operate as the output voltage divider. The error amplifier reference voltage (V ) is 0.69 V. The

FB

relationship between the output voltage and divider is described in equation (8). Using a 10-kΩ resistor for R2

and 1.5-V output regulation, R1 is calculated as 8.52 kΩ, 8.66 kΩ is selected for R1.

V

OUT

0.69 V

R1

1.5 V

R1 ) 10 kW

FB

+

³

+

³ R1 + 8.52 kW

R1

R1 ) R2

(16)

Transient Comparator

The output voltage transient comparators provide a quick response, first strike, approach to output voltage

transients. The output voltage is sensed through a resistor divider at the OSNS pin, using R5 and R6 shown

in Figure 28. If an overvoltage condition is detected, the HDRV gate drive is shut off and the LDRV gate drive

is turned on until the output is returned to regulation. Similarly, if an output undervoltage condition is sensed,

the HDRV gate drive goes to 95% duty cycle to pump the output back up quickly. The voltage divider should

be exactly the same as resistors R1 and R2 discussed previously. Resistor R5=8.66 kΩ and R6=10 kΩ in this

evaluation module.

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

REFERENCE DESIGN

TEST RESULTS

Efficiency Curves

The tested efficiency at different loads and input voltages are shown in Figure 30. The maximum efficiency is

as high as 93% at 1.5-V output. The efficiency is around 88% when the load current (I

) is 20 A.

LOAD

EFFICIENCY

vs

OUTPUT LOAD CURRENT

0.95

V

IN

= 2.5 V

0.90

0.85

V

IN

= 4.0 V

0.80

0.75

V

IN

= 5.0 V

V

IN

= 3.3 V

0.70

0

5

10

− Load Current − A

15

20

I

LOAD

Figure 30.

Typical Operating Waveforms

Typical operating waveforms are shown in Figure 31 and 32.

V

I

= 3.3 V

V

I

= 3.3 V

IN

= 20 A

LOAD

V

(10 mV/div)

OUTac

V

SW

(2 V/div)

t − Time − 1 µs/div

Figure 32

Figure 31

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

REFERENCE DESIGN

Transient Response and Output Ripple Voltage

The output ripple is about 15 mV

voltage is about 35 mV.

at 20-A output. When the load changes from 4 A to 20 A, the overshooting

P−P

Figures 33 and 34 show the transient waveform with and without the transient comparator. Using the transient

comparator yields a settling time of 10-µs faster than without.

The output ripple is about 15 mV

at 20-A output which is shown in Figure 33. When the load changes from

P−P

0 A to 13 A, the overshoot voltage is approximately 80 mV, and the undershoot is is approximately 60 mV as

shown in Figure 35. When the transient comparator is triggered, the powergood (PWRGD) signal goes low.

V

(50 mV/div)

OUTac

I

(10 A/div)

OUT

t − Time − 200 µs/div

V

IN

= 3.3 V

V

IN

= 3.3 V

WIthout Transient

Comparator

WIth Transient

Comparator

WIth Transient

Comparator

WIthout Transient

Comparator

I

(10 A/div)

OUT

I

(10 A/div)

OUT

t − Time − 10 µs/div

Figure 33. Transient Response Undershoot

Figure 34. Transient Response Overshoot

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SLUS535C − MARCH 2003 − REVISED SEPTEMBER 2004

REFERENCE DESIGN

V

PWRGD

(2.5 V/div)

V

(100 mV/div)

OUTac

I

(10 A/div)

OUT

t − Time − 200 µs/div

Figure 35. Transient Response

29

PACKAGE OPTION ADDENDUM

www.ti.com

8-Mar-2005

PACKAGING INFORMATION

Orderable Device

TPS40020PWP

Status ⁽¹⁾

ACTIVE

Package Package

Pins Package Eco Plan ⁽²⁾Lead/Ball Finish MSL Peak Temp ⁽³⁾

Qty

Type

Drawing

HTSSOP

PWP

16

90 Green (RoHS & CU NIPDAU Level-2-260C-1 YEAR

no Sb/Br)

TPS40020PWPR

TPS40021PWP

HTSSOP

PWP

2000 Green (RoHS & CU NIPDAU Level-2-260C-1 YEAR

no Sb/Br)

90 Green (RoHS & CU NIPDAU Level-2-260C-1 YEAR

no Sb/Br)

TPS40021PWPR

TPS40021PWPRG4

2000 Green (RoHS & CU NIPDAU Level-2-260C-1 YEAR

no Sb/Br)

PREVIEW HTSSOP

2000

None

Call TI

⁽¹⁾The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs.

LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.

NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in

a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available.

OBSOLETE: TI has discontinued the production of the device.

(2)

Eco Plan - May not be currently available - please check http://www.ti.com/productcontent for the latest availability information and additional

product content details.

None: Not yet available Lead (Pb-Free).

Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements

for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered

at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.

Green (RoHS & no Sb/Br): TI defines "Green" to mean "Pb-Free" and in addition, uses package materials that do not contain halogens,

including bromine (Br) or antimony (Sb) above 0.1% of total product weight.

(3)

MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDECindustry standard classifications, and peak solder

temperature.

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is

provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the

accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take

reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on

incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited

information may not be available for release.

In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI

to Customer on an annual basis.

Addendum-Page 1

IMPORTANT NOTICE

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型号：	TPS40021
厂家：	TEXAS INSTRUMENTS
描述：	ENHANCED LOW INPUT VOLTAGE MODE SYNCHRONOUS BUCK CONTROLLER 增强型低输入电压模式同步降压控制器输入元件控制器
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