TPS54140DGQR [TI]
1.5-A, 42V STEP DOWN SWIFT™ DC/DC CONVERTER WITH ECO-MODE™; 1.5 A ， 42V降压SWIFT ™ DC / DC具有Eco-Mode ™转换器
| 型号： | TPS54140DGQR |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | 1.5-A, 42V STEP DOWN SWIFT™ DC/DC CONVERTER WITH ECO-MODE™ |
| 文件： | 总46页 (文件大小：1867K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
TPS54140
www.ti.com .............................................................................................................................................................................................. SLVS889–OCTOBER 2008
1.5-A, 42V STEP DOWN
SWIFT™ DC/DC CONVERTER WITH ECO-MODE™
1
FEATURES
•
•
•
0.8-V Internal Voltage Reference
2
•
3.5V to 42V Input Voltage Range
MSOP10 Package With PowerPAD™
•
•
200-mΩ High-Side MOSFET
Supported by SwitcherPro™ Software Tool
(http://focus.ti.com/docs/toolsw/folders/print/s
witcherpro.html)
High Efficiency at Light Loads with a Pulse
Skipping Eco-Mode™
•
For SWIFT™ Documentation, See the TI
Website at http://www.ti.com/swift
•
•
•
•
•
•
•
116µA Operating Quiescent Current
1.3µA Shutdown Current
300kHz to 2.5MHz Switching Frequency
Synchronizes to External Clock
Adjustable Slow Start/Sequencing
UV and OV Power Good Output
Adjustable UVLO Voltage and Hysteresis
APPLICATIONS
•
12-V and 24-V Industrial and Commercial Low
Power Systems
Aftermarket Auto Accessories: Video, GPS,
Entertainment
•
DESCRIPTION
The TPS54140 device is a 42V, 1.5A, step down regulator with an integrated high side MOSFET. Current mode
control provides simple external compensation and flexible component selection. A low ripple pulse skip mode
reduces the no load, regulated output supply current to 116µA. Using the enable pin, shutdown supply current is
reduced to 1.3µA.
Under voltage lockout is internally set at 2.5V, but can be increased using the enable pin. The output voltage
startup ramp is controlled by the slow start pin that can also be configured for sequencing/tracking. An open
drain power good signal indicates the output is within 93% to 107% of its nominal voltage.
A wide switching frequency range allows efficiency and external component size to be optimized. Frequency fold
back and thermal shutdown protects the part during an overload condition.
The TPS54140 is available in 10 pin thermally enhanced MSOP Power Pad package.
SIMPLIFIED SCHEMATIC
EFFICIENCY
vs
LOAD CURRENT
VIN
PWRGD
90
85
80
75
70
65
60
55
50
TPS54140
EN
BOOT
PH
SS/TR
RT/CLK
COMP
VI = 12 V,
VO = 3.3 V,
fsw = 1200 kHz
VSENSE
GND
0
0.25
0.50 0.75
1
1.25 1.50
1.75
2
Load Current - A
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2
Eco-Mode, PowerPAD, SwitcherPro, SWIFT are trademarks of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2008, Texas Instruments Incorporated
TPS54140
SLVS889–OCTOBER 2008.............................................................................................................................................................................................. www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
ORDERING INFORMATION(1)
TJ
PACKAGE
PART NUMBER(2)
–40°C to 150°C
10 Pin MSOP
TPS54140DGQ
(1) For the most current package and ordering information see the Package Option Addendum at the end of this document, or see the TI
website at www.ti.com.
(2) The DGQ package is also available taped and reeled. Add an R suffix to the device type (i.e., TPS54140DGQR).
ABSOLUTE MAXIMUM RATINGS(1)
Over operating temperature range (unless otherwise noted).
VALUE
–0.3 to 47
–0.3 to 5
55
UNIT
VIN
EN
BOOT
VSENSE
COMP
PWRGD
SS/TR
–0.3 to 3
–0.3 to 3
–0.3 to 6
–0.3 to 3
–0.3 to 3.6
8
Input voltage
V
RT/CLK
PH–BOOT
PH
Output voltage
–0.6 to 47
–2 to 47
±200
V
PH, 10-ns Transient
PAD to GND
EN
Voltage Difference
mV
µA
mA
µA
A
100
BOOT
100
Source current
VSENSE
PH
10
Current Limit
100
RT/CLK
VIN
µA
A
Current Limit
100
COMP
PWRGD
SS/TR
µA
mA
µA
kV
V
Sink current
10
200
Electrostatic Discharge (HBM) QSS 009-105 (JESD22-A114A)
Electrostatic Discharge (CDM) QSS 009-147 (JESD22-C101B.01)
Operating junction temperature
1
500
–40 to 150
–65 to 150
°C
°C
Storage temperature
(1) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
2
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PACKAGE DISSIPATION RATINGS(1)
THERMAL IMPEDANCE
JUNCTION TO AMBIENT
PACKAGE
MSOP
(1) Test board conditions:
57 °C/W
A. 3 inches × 3 inches, 2 layers, thickness: 0.062 inch
B. 2-ounce copper traces located on the top and bottom of the PCB
C. 6 (13 mil diameters) THERMAL VIAS LOCATED UNDER THE DEVICE PACKAGE
ELECTRICAL CHARACTERISTICS
TJ = –40°C to 150°C, VIN = 3.5 to 42V (unless otherwise noted)
PARAMETER
SUPPLY VOLTAGE (VIN PIN)
Operating input voltage
TEST CONDITIONS
MIN
TYP
MAX UNIT
3.5
42
V
V
Internal undervoltage lockout
threshold
No voltage hysteresis, rising and falling
EN = 0 V, 25°C, 3.5 V ≤ VIN ≤ 42 V
VSENSE = 0.83 V, VIN = 12 V, 25°C
2.5
1.3
Shutdown supply current
4
µA
Operating : nonswitching supply
current
116
136
ENABLE AND UVLO (EN PIN)
Enable threshold voltage
No voltage hysteresis, rising and falling, 25°C
Enable threshold +50 mV
0.9
1.25
±3.8
±0.9
±2.9
1.55
V
Input current
µA
µA
Enable threshold ±50 mV
Hysteresis current
VOLTAGE REFERENCE
TJ = 25°C
0.792
0.784
0.8
0.8
0.808
0.816
Voltage reference
HIGH-SIDE MOSFET
On-resistance
V
VIN = 3.5 V, BOOT-PH = 3 V
VIN = 12 V, BOOT-PH = 6 V
300
200
mΩ
410
ERROR AMPLIFIER
Input current
50
97
nA
Error amplifier transconductance (gM) ±2 µA < ICOMP < 2 µA, VCOMP = 1 V
µMhos
±2 µA < ICOMP < 2 µA, VCOMP = 1 V,
VVSENSE = 0.4 V
Error amplifier transconductance (gM)
during slow start
26
µMhos
Error amplifier dc gain
VVSENSE = 0.8 V
10,000
2700
±7
V/V
kHz
µA
Error amplifier bandwidth
Error amplifier source/sink
V(COMP) = 1 V, 100 mV overdrive
COMP to switch current
transconductance
6
A/V
CURRENT LIMIT
Current limit threshold
THERMAL SHUTDOWN
VIN = 12 V, TJ = 25°C
1.8
2.7
A
Thermal shutdown
182
°C
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ELECTRICAL CHARACTERISTICS (continued)
TJ = –40°C to 150°C, VIN = 3.5 to 42V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
TIMING RESISTOR AND EXTERNAL CLOCK (RT/CLK PIN)
Switching Frequency Range using
RT mode
300
450
300
2500
720
kHz
kHz
kHz
fSW
Switching frequency
RT = 200 kΩ
581
Switching Frequency Range using
CLK mode
2200
Minimum CLK pulse width
RT/CLK high threshold
RT/CLK low threshold
40
1.9
0.7
ns
V
2.2
0.5
V
RT/CLK falling edge to PH rising
edge delay
Measured at 500 kHz with RT resistor in series
Measured at 500 kHz
60
ns
PLL lock in time
100
µs
SLOW START AND TRACKING (SS/TR)
Charge current
VSS/TR = 0.4 V
2
45
µA
mV
V
SS/TR-to-VSENSE matching
SS/TR-to-reference crossover
SS/TR discharge current (overload)
SS/TR discharge voltage
VSS/TR = 0.4 V
98% nominal
1.0
112
54
VSENSE = 0 V, V(SS/TR) = 0.4 V
VSENSE = 0 V
µA
mV
POWER GOOD (PWRGD PIN)
VSENSE falling
92%
94%
109%
107%
2%
VSENSE rising
VVSENSE
VSENSE threshold
VSENSE rising
VSENSE falling
Hysteresis
VSENSE falling
Output high leakage
On resistance
VSENSE = VREF, V(PWRGD) = 5.5 V, 25°C
I(PWRGD) = 3 mA, VSENSE < 0.79 V
V(PWRGD) < 0.5 V, II(PWRGD) = 100 µA
10
nA
Ω
50
Minimum VIN for defined output
0.95
1.5
V
4
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DEVICE INFORMATION
PIN CONFIGURATION
MSOP10
(TOP VIEW)
BOOT
VIN
10
9
1
2
3
4
PH
GND
COMP
Thermal
Pad
(11)
8
EN
SS/TR
7
VSENSE
PWRGD
RT/CLK
6
5
PIN FUNCTIONS
PIN
I/O
DESCRIPTION
NAME
NO.
A bootstrap capacitor is required between BOOT and PH. If the voltage on this capacitor is below the
minimum required by the output device, the output is forced to switch off until the capacitor is refreshed.
BOOT
1
O
O
I
Error amplifier output, and input to the output switch current comparator. Connect frequency compensation
components to this pin.
COMP
EN
8
3
Enable pin, internal pull-up current source. Pull below 1.2V to disable. Float to enable. Adjust the input
undervoltage lockout with two resistors.
GND
PH
9
–
I
Ground
10
11
The source of the internal high-side power MOSFET.
POWERPAD
–
GND pin must be electrically connected to the exposed pad on the printed circuit board for proper operation.
An open drain output, asserts low if output voltage is low due to thermal shutdown, dropout, over-voltage or
EN shut down.
PWRGD
6
5
4
O
I
Resistor Timing and External Clock. An internal amplifier holds this pin at a fixed voltage when using an
external resistor to ground to set the switching frequency. If the pin is pulled above the PLL upper threshold,
a mode change occurs and the pin becomes a synchronization input. The internal amplifier is disabled and
the pin is a high impedance clock input to the internal PLL. If clocking edges stop, the internal amplifier is
re-enabled and the mode returns to a resistor set function.
RT/CLK
SS/TR
Slow-start and Tracking. An external capacitor connected to this pin sets the output rise time. Since the
voltage on this pin overrides the internal reference, it can be used for tracking and sequencing.
I
VIN
2
7
I
I
Input supply voltage, 3.5 V to 42 V.
VSENSE
Inverting node of the transconductance ( gm) error amplifier.
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FUNCTIONAL BLOCK DIAGRAM
PWRGD
6
EN
3
VIN
2
Shutdown
UO
Thermal
Shutdown
UVLO
Enable
Comparator
Logic
Shutdown
Shutdown
Logic
OV
Enable
Threshold
Boot
Charge
Voltage
Reference
Minimum
Clamp
Pulse
Boot
UVLO
Current
Sense
ERROR
AMPLIFIER
Skip
PWM
Comparator
VSENSE
SS/TR
7
4
BOOT
1
Logic
And
PWM Latch
Shutdown
Slope
Compensation
PH
10
11
8
COMP
POWERPAD
Frequency
Shift
Maximum
Clamp
Overload
Recovery
GND
9
Oscillator
with PLL
TPS54140 Block Diagram
5
RT/CLK
6
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www.ti.com .............................................................................................................................................................................................. SLVS889–OCTOBER 2008
TYPICAL CHARACTERISTICS
ON RESISTANCE vs JUNCTION TEMPERATURE
VOLTAGE REFERENCE vs JUNCTION TEMPERATURE
0.816
500
375
250
VI = 12 V
VI = 12 V
0.808
0.800
BOOT-PH = 3 V
BOOT-PH = 6 V
0.792
0.784
125
0
-50
-50
-25
0
25
50
75
100
125
150
-25
0
25
50
75
100
125
150
TJ - Junction Temperature - °C
TJ - Junction Temperature - °C
Figure 1.
Figure 2.
SWITCH CURRENT LIMIT vs JUNCTION TEMPERATURE
3.5
SWITCHING FREQUENCY vs JUNCTION TEMPERATURE
610
VI = 12 V,
VI = 12 V
RT = 200 kW
600
3
2.5
2
590
580
570
560
550
-50
-25
0
25
50
75
100
125
150
-50
-25
0
25
50
75
100
125
150
TJ - Junction Temperature - °C
TJ - Junction Temperature - °C
Figure 3.
Figure 4.
SWITCHING FREQUENCY vs RT/CLK RESISTANCE HIGH
FREQUENCY RANGE
2500
SWITCHING FREQUENCY vs RT/CLK RESISTANCE LOW
FREQUENCY RANGE
1000
VI = 12 V,
TJ = 25°C
VI = 12 V,
TJ = 25°C
800
2000
1500
1000
600
400
200
0
500
0
0
25
50
75
100
125
150
175
200
100
200
300
400
500
600
700
800
900
1000
RT/CLK - Resistance - kW
RT/CLK - Resistance - kW
Figure 5.
Figure 6.
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TYPICAL CHARACTERISTICS (continued)
EA TRANSCONDUCTANCE DURING SLOW START vs
JUNCTION TEMPERATURE
EA TRANSCONDUCTANCE vs JUNCTION TEMPERATURE
150
40
VI = 12 V
VI = 12 V
130
30
110
90
20
70
50
10
-50
-50
-25
0
25
50
75
100
125
150
-25
0
25
50
75
100
125
150
TJ - Junction Temperature - °C
TJ - Junction Temperature - °C
Figure 7.
Figure 8.
EN PIN VOLTAGE vs JUNCTION TEMPERATURE
EN PIN CURRENT vs JUNCTION TEMPERATURE
1.40
-3.25
VI = 12 V,
VI = 12 V
VI(EN) = Threshold +50 mV
-3.5
1.30
1.20
1.10
-3.75
-4
-4.25
75
150
-50
-25
0
25
50
75
100
125
150
-25
0
25
50
100
-50
125
TJ - Junction Temperature - °C
TJ - Junction Temperature - °C
Figure 9.
Figure 10.
EN PIN CURRENT vs JUNCTION TEMPERATURE
SS/TR CHARGE CURRENT vs JUNCTION TEMPERATURE
-1
-0.8
VI = 12 V,
VI = 12 V
VI(EN) = Threshold -50 mV
-0.85
-1.5
-0.9
-2
-0.95
-2.5
-1
-50
-3
-50
150
25
125
0
50
75
100
-25
-25
0
25
50
75
100
125
150
TJ - Junction Temperature - °C
TJ - Junction Temperature - °C
Figure 11.
Figure 12.
8
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TYPICAL CHARACTERISTICS (continued)
SS/TR DISCHARGE CURRENT vs JUNCTION
TEMPERATURE
SWITCHING FREQUENCY vs VSENSE
120
115
110
100
80
60
40
20
0
VI = 12 V
VI = 12 V,
TJ = 25°C
105
100
-50
-25
0
25
50
75
100
125
150
0
0.2
0.4
VSENSE - V
0.6
0.8
TJ - Junction Temperature - °C
Figure 13.
Figure 14.
SHUTDOWN SUPPLY CURRENT vs JUNCTION
TEMPERATURE
SHUTDOWN SUPPLY CURRENT vs INPUT VOLTAGE (Vin)
2
2
VI = 12 V
TJ = 25°C
1.5
1.5
1
1
0.5
0.5
0
-50
0
-25
0
25
50
75
100
125
150
0
10
20
30
40
TJ - Junction Temperature - °C
VI - Input Voltage - V
Figure 15.
Figure 16.
VIN SUPPLY CURRENT vs JUNCTION TEMPERATURE
VIN SUPPLY CURRENT vs INPUT VOLTAGE
140
140
TJ = 25oC,
VI = 12 V,
VI(VSENSE) = 0.83 V
VI(VSENSE) = 0.83 V
130
120
130
120
110
100
90
110
100
90
0
20
40
-50
-25
0
25
50
75
100
125
150
VI - Input Voltage - V
TJ - Junction Temperature - °C
Figure 17.
Figure 18.
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TYPICAL CHARACTERISTICS (continued)
PWRGD ON RESISTANCE vs JUNCTION TEMPERATURE
100
PWRGD THRESHOLD vs JUNCTION TEMPERATURE
115
VI = 12 V
VI = 12 V
VSENSE Rising
110
105
80
VSENSE Falling
60
40
100
95
VSENSE Rising
20
0
VSENSE Falling
90
85
-50
-25
0
25
50
75
100
125
150
-50
50
0
25
75
125
-25
100
150
TJ - Junction Temperature - °C
TJ - Junction Temperature - °C
Figure 19.
Figure 20.
BOOT-PH UVLO vs JUNCTION TEMPERATURE
INPUT VOLTAGE (UVLO) vs JUNCTION TEMPERATURE
3
2.5
2.3
2.75
2.50
2
1.8
1.5
2.25
2
-50
-25
0
25
50
75
100
125
150
-50
-25
0
25
50
75
100
125
150
TJ - Junction Temperature - °C
TJ - Junction Temperature - °C
Figure 21.
Figure 22.
SS/TR TO VSENSE OFFSET vs VSENSE
SS/TR TO VSENSE OFFSET vs TEMPERATURE
60
55
50
45
500
400
300
V(SS/TR) = 0.2 V
VI = 12 V
VI = 12 V,
TJ = 25oC
200
100
40
35
30
0
-50
-25
0
25
50
75
100
125
150
0
100
200
300
400
500
600
700
800
TJ - Junction Temperature - °C
VSENSE - mV
Figure 23.
Figure 24.
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OVERVIEW
The TPS54140 device is a 42-V, 1.5-A, step-down (buck) regulator with an integrated high side n-channel
MOSFET. To improve performance during line and load transients the device implements a constant frequency,
current mode control which reduces output capacitance and simplifies external frequency compensation design.
The wide switching frequency of 300kHz to 2500kHz allows for efficiency and size optimization when selecting
the output filter components. The switching frequency is adjusted using a resistor to ground on the RT/CLK pin.
The device has an internal phase lock loop (PLL) on the RT/CLK pin that is used to synchronize the power
switch turn on to a falling edge of an external system clock.
The TPS54140 has a default start up voltage of approximately 2.5V. The EN pin has an internal pull-up current
source that can be used to adjust the input voltage under voltage lockout (UVLO) threshold with two external
resistors. In addition, the pull up current provides a default condition. When the EN pin is floating the device will
operate. The operating current is 116µA when not switching and under no load. When the device is disabled, the
supply current is 1.3µA.
The integrated 200mΩ high side MOSFET allows for high efficiency power supply designs capable of delivering
1.5 amperes of continuous current to a load. The TPS54140 reduces the external component count by
integrating the boot recharge diode. The bias voltage for the integrated high side MOSFET is supplied by a
capacitor on the BOOT to PH pin. The boot capacitor voltage is monitored by an UVLO circuit and will turn the
high side MOSFET off when the boot voltage falls below a preset threshold. The TPS54140 can operate at high
duty cycles because of the boot UVLO. The output voltage can be stepped down to as low as the 0.8V
reference.
The TPS54140 has a power good comparator (PWRGD) which asserts when the regulated output voltage is less
than 92% or greater than 109% of the nominal output voltage. The PWRGD pin is an open drain output which
deasserts when the VSENSE pin voltage is between 94% and 107% of the nominal output voltage allowing the
pin to transition high when a pull-up resistor is used.
The TPS54140 minimizes excessive output overvoltage (OV) transients by taking advantage of the OV power
good comparator. When the OV comparator is activated, the high side MOSFET is turned off and masked from
turning on until the output voltage is lower than 107%.
The SS/TR (slow start/tracking) pin is used to minimize inrush currents or provide power supply sequencing
during power up. A small value capacitor should be coupled to the pin to adjust the slow start time. A resistor
divider can be coupled to the pin for critical power supply sequencing requirements. The SS/TR pin is discharged
before the output powers up. This discharging ensures a repeatable restart after an over-temperature fault,
UVLO fault or a disabled condition.
The TPS54140, also, discharges the slow start capacitor during overload conditions with an overload recovery
circuit. The overload recovery circuit will slow start the output from the fault voltage to the nominal regulation
voltage once a fault condition is removed. A frequency foldback circuit reduces the switching frequency during
startup and overcurrent fault conditions to help control the inductor current.
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DETAILED DESCRIPTION
Fixed Frequency PWM Control
The TPS54140 uses an adjustable fixed frequency, peak current mode control. The output voltage is compared
through external resistors on the VSENSE pin to an internal voltage reference by an error amplifier which drives
the COMP pin. An internal oscillator initiates the turn on of the high side power switch. The error amplifier output
is compared to the high side power switch current. When the power switch current reaches the COMP voltage
level the power switch is turned off. The COMP pin voltage will increase and decrease as the output current
increases and decreases. The device implements a current limit by clamping the COMP pin voltage to a
maximum level. The Eco-Mode™ is implemented with a minimum clamp on the COMP pin.
Slope Compensation Output Current
The TPS54140 adds a compensating ramp to the switch current signal. This slope compensation prevents
sub-harmonic oscillations. The available peak inductor current remains constant over the full duty cycle range.
Pulse Skip Eco-Mode
The TPS54140 enters the pulse skip mode when the voltage on the COMP pin is the minimum clamp value. The
TPS54140 operates in a pulse skip mode at light load currents to improve efficiency. The peak switch current
during the pulse skip mode will be the greater value of 50mA or the peak inductor current that is a function of the
minimum on time, input voltage, output voltage and inductance value. When the load current is low and the
output voltage is within regulation the device will enter a sleep mode and draw only 116µA input quiescent
current. While the device is in sleep mode the output power is delivered by the output capacitor. As the load
current decreases, the time the output capacitor supplies the load current increases and the switching frequency
decreases reducing gate drive and switching losses. As the output voltage drops, the TPS54140 wakes up from
the sleep mode and the power switch turns on to recharge the output capacitor, see Figure 25. The internal PLL
remains operating when in sleep mode. When operating at light load currents in the pulse skip mode the
switching transitions occur synchronously with the external clock signal.
VOUT
(ac)
I
L
PH
Figure 25. Pulse Skip Mode Operation
Bootstrap Voltage (BOOT)
The TPS54140 has an integrated boot regulator and requires a small ceramic capacitor between the BOOT and
PH pin to provide the gate drive voltage for the high side MOSFET. The value of the ceramic capacitor should be
0.1µF. A ceramic capacitor with an X7R or X5R grade dielectric is recommended because of the stable
characteristics over temperature and voltage. To improve drop out, the TPS54140 is designed to operate at
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DETAILED DESCRIPTION (continued)
100% duty cycle as long as the BOOT to PH pin voltage is greater than 2.1V. When the voltage from BOOT to
PH drops below 2.1V, the high side MOSFET is turned off using an UVLO circuit allowing for the low side diode
to conduct which allows refreshing of the BOOT capacitor. Since the supply current sourced from the BOOT
capacitor is low, the high side MOSFET can remain on for more switching cycles than it refreshes, thus, the
effective duty cycle limitation that is attributed to the boot regulator system is high.
Low Dropout Operation
The duty cycle during dropout of the regulator will be mainly determined by the voltage drops across the power
MOSFET, inductor, low side diode and printed circuit board resistance. During operating conditions in which the
input voltage drops, the high side MOSFET can remain on for 100% of the duty cycle to maintain output
regulation or until the BOOT to PH voltage falls below 2.1V.
Once the high side is off, the low side diode will conduct and the BOOT capacitor will be recharged. During this
boot capacitor recharge time, the inductor current will ramp down until the high side MOSFET turns on. The
recharge time is longer than the typical high side off time of previous switching cycles, and thus, the inductor
current ripple is larger resulting in more ripple voltage on the output. The recharge time is a function of the input
voltage, boot capacitor value, and the impedance of the internal boot recharge diode.
Attention needs to be taken in maximum duty cycle applications which experience extended time periods without
a load current. When the voltage across the BOOT capacitors falls below the 2.1V threshold in applications that
have a difference in the input voltage and output voltage that is less than 3V, the high side MOSFET will be
turned off but there is not enough current in the inductor to pull the PH pin down to recharge the boot capacitor.
The regulator will not switch because the boot capacitor is less than 2.1V and the output capacitor will decay until
the difference in the input voltage and output voltage is 2.1V. At this time the boot under voltage lockout is
exceeded and the device will switch until the desired output voltage is reached.
The start and stop voltages are shown in Figure 26 and Figure 27 for 3.3V and 5V applications. The voltages are
plotted versus the load current. The start voltage is defined as the input voltage needed to regulate within 1%.
The stop voltage is defined as the input voltage at which the output drops by 5% or stops switching.
4
5.6
VO = 3.3 V
VO = 5 V
3.8
3.6
3.4
5.4
5.2
Start
Stop
Start
Stop
5
3.2
3
4.8
4.6
0
0.05
0.10
IO - Output Current - A
0.15
0.20
0
0.05
0.10
IO - Output Current - A
0.15
0.20
Figure 26. 3.3V Start/Stop Voltage
Figure 27. 5.0V Start/Stop Voltage
Error Amplifier
The TPS54140 has a transconductance amplifier for the error amplifier. The error amplifier compares the
VSENSE voltage to the lower of the SS/TR pin voltage or the internal 0.8V voltage reference. The
transconductance (gm) of the error amplifier is 97µA/V during normal operation. During the slow start operation,
the transconductance is a fraction of the normal operating gm. When the voltage of the VSENSE pin is below
0.8V and the device is regulating using the SS/TR voltage, the gm is 25µA/V.
The frequency compensation components (capacitor, series resistor and capacitor) are added to the COMP pin
to ground.
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DETAILED DESCRIPTION (continued)
Voltage Reference
The voltage reference system produces a precise ±2% voltage reference over temperature by scaling the output
of a temperature stable bandgap circuit.
Adjusting the Output Voltage
The output voltage is set with a resistor divider from the output node to the VSENSE pin. It is recommended to
use 1% tolerance or better divider resistors. Start with a 10 kΩ for the R2 resistor and use the Equation 1 to
calculate R1. To improve efficiency at very light loads consider using larger value resistors. If the values are too
high the regulator will be more susceptible to noise and voltage errors from the VSENSE input current will be
noticeable
Vout - 0.8V
æ
ö
R1 = R2 ´
ç
÷
0.8 V
è
ø
(1)
Enable and Adjusting Undervoltage Lockout
The TPS54140 is disabled when the VIN pin voltage falls below 2.5 V. If an application requires a higher
undervoltage lockout (UVLO), use the EN pin as shown in Figure 28 to adjust the input voltage UVLO by using
the two external resistors. Though it is not necessary to use the UVLO adjust registers, for operation it is highly
recommended to provide consistent power up behavior. The EN pin has an internal pull-up current source, I1, of
0.9µA that provides the default condition of the TPS54140 operating when the EN pin floats. Once the EN pin
voltage exceeds 1.25V, an additional 2.9µA of hysteresis, Ihys, is added. This additional current facilitates input
voltage hysteresis. Use Equation 2 to set the external hysteresis for the input voltage. Use Equation 3 to set the
input start voltage.
TPS54140
VIN
Ihys
I1
0.9 mA
R1
2.9 mA
+
R2
EN
-
1.25 V
Figure 28. Adjustable Undervoltage Lockout (UVLO)
V
- V
STOP
START
R1=
I
HYS
(2)
(3)
VENA
R2 =
VSTART - VENA
+ I1
R1
Another technique to add input voltage hysteresis is shown in Figure 29. This method may be used, if the
resistance values are high from the previous method and a wider voltage hysteresis is needed. The resistor R3
sources additional hysteresis current into the EN pin.
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DETAILED DESCRIPTION (continued)
TPS54140
VIN
Ihys
R1
R2
I1
0.9 mA
2.9 mA
+
EN
1.25 V
-
VOUT
R3
Figure 29. Adding Additional Hysteresis
VSTART - VSTOP
R1 =
VOUT
+
R3
IHYS
(4)
(5)
VENA
R2 =
VSTART - VENA
VENA
+ I1 -
R1
R3
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DETAILED DESCRIPTION (continued)
Slow Start/Tracking Pin (SS/TR)
The TPS54140 effectively uses the lower voltage of the internal voltage reference or the SS/TR pin voltage as
the power-supply's reference voltage and regulates the output accordingly. A capacitor on the SS/TR pin to
ground implements a slow start time. The TPS54140 has an internal pull-up current source of 2µA that charges
the external slow start capacitor. The calculations for the slow start time (10% to 90%) are shown in Equation 6.
The voltage reference (VREF) is 0.8 V and the slow start current (ISS) is 2µA. The slow start capacitor should
remain lower than 0.47µF and greater than 0.47nF.
Tss(ms) ´ Iss(mA)
Css(nF) =
Vref (V) ´ 0.8
(6)
At power up, the TPS54140 will not start switching until the slow start pin is discharged to less than 40 mV to
ensure a proper power up, see Figure 30.
Also, during normal operation, the TPS54140 will stop switching and the SS/TR must be discharged to 40 mV,
when the VIN UVLO is exceeded, EN pin pulled below 1.25V, or a thermal shutdown event occurs.
The VSENSE voltage will follow the SS/TR pin voltage with a 45mV offset up to 85% of the internal voltage
reference. When the SS/TR voltage is greater than 85% on the internal reference voltage the offset increases as
the effective system reference transitions from the SS/TR voltage to the internal voltage reference (see
Figure 23). The SS/TR voltage will ramp linearly until clamped at 1.7V.
EN
SS/TR
V
SENSE
VOUT
Figure 30. Operation of SS/TR Pin when Starting
Overload Recovery Circuit
The TPS54140 has an overload recovery (OLR) circuit. The OLR circuit will slow start the output from the
overload voltage to the nominal regulation voltage once the fault condition is removed. The OLR circuit will
discharge the SS/TR pin to a voltage slightly greater than the VSENSE pin voltage using an internal pull down of
100µA when the error amplifier is changed to a high voltage from a fault condition. When the fault condition is
removed, the output will slow start from the fault voltage to nominal output voltage.
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DETAILED DESCRIPTION (continued)
Sequencing
Many of the common power supply sequencing methods can be implemented using the SS/TR, EN and PWRGD
pins. The sequential method can be implemented using an open drain output of a power on reset pin of another
device. The sequential method is illustrated in Figure 31 using two TPS54140 devices. The power good is
coupled to the EN pin on the TPS54140 which will enable the second power supply once the primary supply
reaches regulation. If needed, a 1nF ceramic capacitor on the EN pin of the second power supply will provide a
1ms start up delay. Figure 32 shows the results of Figure 31.
TPS54140
PWRGD
EN
EN
EN1
SS/TR
SS /TR
PWRGD1
PWRGD
VOUT1
VOUT2
Figure 31. Schematic for Sequential Start-Up Sequence
Figure 32. Sequential Startup using EN and PWRGD
TPS54140
3
4
6
EN
EN1, EN2
SS/TR
PWRGD
VOUT1
TPS54140
VOUT2
3
4
6
EN
SS/TR
PWRGD
Figure 33. Schematic for Ratiometric Start-Up Sequence
Figure 34. Ratio-Metric Startup using Coupled SS/TR
pins
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DETAILED DESCRIPTION (continued)
Figure 33 shows a method for ratio-metric start up sequence by connecting the SS/TR pins together. The
regulator outputs will ramp up and reach regulation at the same time. When calculating the slow start time the
pull up current source must be doubled in Equation 6. Figure 34 shows the results of Figure 33.
TPS54140
EN
VOUT 1
SS/TR
PWRGD
TPS54140
VOUT 2
EN
R1
SS/TR
R2
PWRGD
R3
R4
Figure 35. Schematic for Ratiometric and Simultaneous Start-Up Sequence
Ratio-metric and simultaneous power supply sequencing can be implemented by connecting the resistor network
of R1 and R2 shown in Figure 35 to the output of the power supply that needs to be tracked or another voltage
reference source. Using Equation 7 and Equation 8, the tracking resistors can be calculated to initiate the Vout2
slightly before, after or at the same time as Vout1. Equation 9 is the voltage difference between Vout1 and Vout2
at the 95% of nominal output regulation.
The deltaV variable is zero volts for simultaneous sequencing. To minimize the effect of the inherent SS/TR to
VSENSE offset (Vssoffset) in the slow start circuit and the offset created by the pullup current source (Iss) and
tracking resistors, the Vssoffset and Iss are included as variables in the equations.
To design a ratio-metric start up in which the Vout2 voltage is slightly greater than the Vout1 voltage when Vout2
reaches regulation, use a negative number in Equation 7 through Equation 9 for deltaV. Equation 9 will result in a
positive number for applications which the Vout2 is slightly lower than Vout1 when Vout2 regulation is achieved.
Since the SS/TR pin must be pulled below 40mV before starting after an EN, UVLO or thermal shutdown fault,
careful selection of the tracking resistors is needed to ensure the device will restart after a fault. Make sure the
calculated R1 value from Equation 7 is greater than the value calculated in Equation 10 to ensure the device can
recover from a fault.
As the SS/TR voltage becomes more than 85% of the nominal reference voltage the Vssoffset becomes larger
as the slow start circuits gradually handoff the regulation reference to the internal voltage reference. The SS/TR
pin voltage needs to be greater than 1.3V for a complete handoff to the internal voltage reference as shown in
Figure 23.
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DETAILED DESCRIPTION (continued)
Vout2 + deltaV
VREF
Vssoffset
R1 =
´
Iss
(7)
VREF ´ R1
Vout2 + deltaV - VREF
R2 =
(8)
(9)
deltaV = Vout1 - Vout2
R1 > 2800 ´ Vout1 - 180 ´ deltaV
(10)
EN
EN
VOUT1
VOUT1
VOUT2
VOUT2
Figure 36. Ratio-metric Startup with Tracking Resistors
Figure 37. Ratiometric Startup with Tracking Resistors
EN
VOUT1
VOUT2
Figure 38. Simultaneous Startup With Tracking Resistor
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DETAILED DESCRIPTION (continued)
Constant Switching Frequency and Timing Resistor (RT/CLK Pin)
The switching frequency of the TPS54140 is adjustable over a wide range from approximately 300kHz to
2500kHz by placing a resistor on the RT/CLK pin. The RT/CLK pin voltage is typically 0.5V and must have a
resistor to ground to set the switching frequency. To determine the timing resistance for a given switching
frequency, use Equation 11 or the curves in Figure 39 or Figure 40. To reduce the solution size one would
typically set the switching frequency as high as possible, but tradeoffs of the supply efficiency, maximum input
voltage and minimum controllable on time should be considered.
The minimum controllable on time is typically 130ns and limits the maximum operating input voltage.
The maximum switching frequency is also limited by the frequency shift circuit. More discussion on the details of
the maximum switching frequency is located below.
206033
RT (kOhm) =
¦sw (kHz)1.0888
(11)
SWITCHING FREQUENCY
SWITCHING FREQUENCY
vs
vs
RT/CLK RESISTANCE HIGH FREQUENCY RANGE
RT/CLK RESISTANCE LOW FREQUENCY RANGE
1000
800
600
400
2500
2000
1500
VI = 12 V,
TJ = 25°C
VI = 12 V,
TJ = 25°C
1000
500
0
200
0
100
200
300
400
500
600
700
800
900
1000
0
25
50
75
100
125
150
175
200
RT/CLK - Clock Resistance - kW
RT/CLK - Clock Resistance - kW
Figure 39. High Range RT
Figure 40. Low Range RT
Overcurrent Protection and Frequency Shift
The TPS54140 implements current mode control which uses the COMP pin voltage to turn off the high side
MOSFET on a cycle by cycle basis. Each cycle the switch current and COMP pin voltage are compared, when
the peak switch current intersects the COMP voltage, the high side switch is turned off. During overcurrent
conditions that pull the output voltage low, the error amplifier will respond by driving the COMP pin high,
increasing the switch current. The error amplifier output is clamped internally, which functions as a switch current
limit.
To increase the maximum operating switching frequency at high input voltages the TPS54140 implements a
frequency shift. The switching frequency is divided by 8, 4, 2, and 1 as the voltage ramps from 0 to 0.8 volts on
VSENSE pin.
The device implements a digital frequency shift to enable synchronizing to an external clock during normal
startup and fault conditions. Since the device can only divide the switching frequency by 8, there is a maximum
input voltage limit in which the device operates and still have frequency shift protection.
During short-circuit events (particularly with high input voltage applications), the control loop has a finite minimum
controllable on time and the output has a very low voltage. During the switch on time, the inductor current ramps
to the peak current limit because of the high input voltage and minimum on time. During the switch off time, the
inductor would normally not have enough off time and output voltage for the inductor to ramp down by the ramp
up amount. The frequency shift effectively increases the off time allowing the current to ramp down.
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DETAILED DESCRIPTION (continued)
Selecting the Switching Frequency
The switching frequency that is selected should be the lower value of the two equations, Equation 12 and
Equation 13. Equation 12 is the maximum switching frequency limitation set by the minimum controllable on time.
Setting the switching frequency above this value will cause the regulator to skip switching pulses.
Equation 13 is the maximum switching frequency limit set by the frequency shift protection. To have adequate
output short circuit protection at high input voltages, the switching frequency should be set to be less than the
fsw(maxshift) frequency. In Equation 13, to calculate the maximum switching frequency one must take into
account that the output voltage decreases from the nominal voltage to 0 volts, the fdiv integer increases from 1 to
8 corresponding to the frequency shift.
In Figure 41, the solid line illustrates a typical safe operating area regarding frequency shift and assumes the
output voltage is zero volts, and the resistance of the inductor is 0.1Ω, FET on resistance of 0.2Ω and the diode
voltage drop is 0.5V. The dashed line is the maximum switching frequency to avoid pulse skipping. Enter these
equations in a spreadsheet or other software or use the SwitcherPro design software to determine the switching
frequency.
æ
ç
ö
÷
IL ´ Rdc + VOUT + Vd
1
fSW max skip
=
´
(
)
ç
÷
tON min
V
- IL ´ R
+ Vd
DS on
IN
(
)
( )
è
ø
(12)
(13)
æ
ö
÷
÷
÷
ç
ç
ç
è
(IL ´Rdc + VOUTSC + Vd)
fdiv
fSW maxshift
=
´
(
)
tON min
(
)
V
IN - 2´ VOUTSC + IL ´ R
+ Rdc + Vd
)
(
(
)
DS on
( )
ø
IL
inductor current
Rdc
inductor resistance
maximum input voltage
output voltage
VIN
VOUT
VOUTSC
Vc
output voltage during short
diode voltage drop
RDS(on)
tON(min)
ƒDIV
switch on resistance
minimum controllable on time
frequency divide equals (1, 2, 4, or 8)
2500
VO = 3.3 V
2000
Shift
1500
Skip
1000
500
0
20
30
10
40
VI - Input Voltage - V
Figure 41. Maximum Switching Frequency vs. Input Voltage
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DETAILED DESCRIPTION (continued)
How to Interface to RT/CLK Pin
The RT/CLK pin can be used to synchronize the regulator to an external system clock. To implement the
synchronization feature connect a square wave to the RT/CLK pin through the circuit network shown in
Figure 42. The square wave amplitude must transition lower than 0.5V and higher than 2.2V on the RT/CLK pin
and have an on time greater than 40 ns and an off time greater than 40 ns. The synchronization frequency range
is 300 kHz to 2200 kHz. The rising edge of the PH will be synchronized to the falling edge of RT/CLK pin signal.
The external synchronization circuit should be designed in such a way that the device will have the default
frequency set resistor connected from the RT/CLK pin to ground should the synchronization signal turn off. It is
recommended to use a frequency set resistor connected as shown in Figure 42 through a 50Ω resistor to
ground. The resistor should set the switching frequency close to the external CLK frequency. It is recommended
to ac couple the synchronization signal through a 10 pF ceramic capacitor to RT/CLK pin and a 4kΩ series
resistor. The series resistor reduces PH jitter in heavy load applications when synchronizing to an external clock
and in applications which transition from synchronizing to RT mode. The first time the CLK is pulled above the
CLK threshold the device switches from the RT resistor frequency to PLL mode. The internal 0.5V voltage source
is removed and the CLK pin becomes high impedance as the PLL starts to lock onto the external signal. Since
there is a PLL on the regulator the switching frequency can be higher or lower than the frequency set with the
external resistor. The device transitions from the resistor mode to the PLL mode and then will increase or
decrease the switching frequency until the PLL locks onto the CLK frequency within 100 microseconds.
When the device transitions from the PLL to resistor mode the switching frequency will slow down from the CLK
frequency to 150 kHz, then reapply the 0.5V voltage and the resistor will then set the switching frequency. The
switching frequency is divided by 8, 4, 2, and 1 as the voltage ramps from 0 to 0.8 volts on VSENSE pin. The
device implements a digital frequency shift to enable synchronizing to an external clock during normal startup
and fault conditions. Figure 43, Figure 44 and Figure 45 show the device synchronized to an external system
clock in continuous conduction mode (ccm) discontinuous conduction (dcm) and pulse skip mode (psm).
TPS54140
10 pF
4 kW
PLL
R
fset
RT/CLK
EXT
Clock
Source
50 W
Figure 42. Synchronizing to a System Clock
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DETAILED DESCRIPTION (continued)
EXT
EXT
VOUT
IL
PH
PH
IL
Figure 43. Plot of Synchronizing in ccm
Figure 44. Plot of Synchronizing in dcm
EXT
IL
PH
Figure 45. Plot of Synchronizing in PSM
Power Good (PWRGD Pin)
The PWRGD pin is an open drain output. Once the VSENSE pin is between 94% and 107% of the internal
voltage reference the PWRGD pin is de-asserted and the pin floats. It is recommended to use a pull-up resistor
between the values of 10 and 100kΩ to a voltage source that is 5.5V or less. The PWRGD is in a defined state
once the VIN input voltage is greater than 1.5V but with reduced current sinking capability. The PWRGD will
achieve full current sinking capability as VIN input voltage approaches 3V.
The PWRGD pin is pulled low when the VSENSE is lower than 92% or greater than 109% of the nominal internal
reference voltage. Also, the PWRGD is pulled low, if the UVLO or thermal shutdown are asserted or the EN pin
pulled low.
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DETAILED DESCRIPTION (continued)
Overvoltage Transient Protection
The TPS54140 incorporates an overvoltage transient protection (OVTP) circuit to minimize voltage overshoot
when recovering from output fault conditions or strong unload transients on power supply designs with low value
output capacitance. For example, when the power supply output is overloaded the error amplifier compares the
actual output voltage to the internal reference voltage. If the VSENSE pin voltage is lower than the internal
reference voltage for a considerable time, the output of the error amplifier will respond by clamping the error
amplifier output to a high voltage. Thus, requesting the maximum output current. Once the condition is removed,
the regulator output rises and the error amplifier output transitions to the steady state duty cycle. In some
applications, the power supply output voltage can respond faster than the error amplifier output can respond, this
actuality leads to the possibility of an output overshoot. The OVTP feature minimizes the output overshoot, when
using a low value output capacitor, by implementing a circuit to compare the VSENSE pin voltage to OVTP
threshold which is 109% of the internal voltage reference. If the VSENSE pin voltage is greater than the OVTP
threshold, the high side MOSFET is disabled preventing current from flowing to the output and minimizing output
overshoot. When the VSENSE voltage drops lower than the OVTP threshold, the high side MOSFET is allowed
to turn on at the next clock cycle.
Thermal Shutdown
The device implements an internal thermal shutdown to protect itself if the junction temperature exceeds 182°C.
The thermal shutdown forces the device to stop switching when the junction temperature exceeds the thermal
trip threshold. Once the die temperature decreases below 182°C, the device reinitiates the power up sequence
by discharging the SS/TR pin.
Small Signal Model for Loop Response
Figure 46 shows an equivalent model for the TPS54140 control loop which can be modeled in a circuit simulation
program to check frequency response and dynamic load response. The error amplifier is a transconductance
amplifier with a gmEA of 97 µA/V. The error amplifier can be modeled using an ideal voltage controlled current
source. The resistor Ro and capacitor Co model the open loop gain and frequency response of the amplifier. The
1mV ac voltage source between the nodes a and b effectively breaks the control loop for the frequency response
measurements. Plotting c/a shows the small signal response of the frequency compensation. Plotting a/b shows
the small signal response of the overall loop. The dynamic loop response can be checked by replacing RL with a
current source with the appropriate load step amplitude and step rate in a time domain analysis. This equivalent
model is only valid for continuous conduction mode designs.
PH
V
O
Power Stage
gm 6 A/V
ps
a
b
R
R1
ESR
R
COMP
L
c
VSENSE
C
OUT
0.8 V
CO
RO
R3
C1
gm
ea
C2
R2
97 mA/V
Figure 46. Small Signal Model for Loop Response
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DETAILED DESCRIPTION (continued)
Simple Small Signal Model for Peak Current Mode Control
Figure 47 describes a simple small signal model that can be used to understand how to design the frequency
compensation. The TPS54140 power stage can be approximated to a voltage-controlled current source (duty
cycle modulator) supplying current to the output capacitor and load resistor. The control to output transfer
function is shown in Equation 14 and consists of a dc gain, one dominant pole, and one ESR zero. The quotient
of the change in switch current and the change in COMP pin voltage (node c in Figure 46) is the power stage
transconductance. The gmPS for the TPS54140 is 6A/V. The low-frequency gain of the power stage frequency
response is the product of the transconductance and the load resistance as shown in Equation 15.
As the load current increases and decreases, the low-frequency gain decreases and increases, respectively. This
variation with the load may seem problematic at first glance, but fortunately the dominant pole moves with the
load current (see Equation 16). The combined effect is highlighted by the dashed line in the right half of
Figure 47. As the load current decreases, the gain increases and the pole frequency lowers, keeping the 0-dB
crossover frequency the same for the varying load conditions which makes it easier to design the frequency
compensation. The type of output capacitor chosen determines whether the ESR zero has a profound effect on
the frequency compensation design. Using high ESR aluminum electrolytic capacitors may reduce the number
frequency compensation components needed to stabilize the overall loop because the phase margin increases
from the ESR zero at the lower frequencies (see Equation 17).
V
O
Adc
VC
R
ESR
fp
R
L
gm
ps
C
OUT
fz
Figure 47. Simple Small Signal Model and Frequency
Response for Peak Current Mode Control
æ
ç
è
ö
÷
ø
s
1+
1+
2p´ fZ
vO
vC
= Adc ´
æ
ç
è
ö
÷
ø
s
2p´ fP
(14)
(15)
Adc = gmps ´ RL
1
f
=
P
C
´R ´ 2p
OUT
L
(16)
(17)
1
f
=
Z
C
´R
´ 2p
OUT
ESR
Small Signal Model for Frequency Compensation
The TPS54140 uses a transconductance amplifier for the error amplifier and readily supports three of the
commonly-used frequency compensation circuits. Compensation circuits Type 2A, Type 2B, and Type 1 are
shown in Figure 48. Type 2 circuits most likely implemented in high bandwidth power-supply designs using low
ESR output capacitors. The Type 1 circuit is used with power-supply designs with high-ESR aluminum
electrolytic or tantalum capacitors.. Equation 18 and Equation 19 show how to relate the frequency response of
the amplifier to the small signal model in Figure 48. The open-loop gain and bandwidth are modeled using the RO
and CO shown in Figure 48. See the application section for a design example using a Type 2A network with a
low ESR output capacitor.
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DETAILED DESCRIPTION (continued)
Equation 18 through Equation 27 are provided as a reference for those who prefer to compensate using the
preferred methods. Those who prefer to use prescribed method use the method outlined in the application
section or use switched information.
V
O
R1
VSENSE
Type 2A
Type 2B
Type 1
gm
ea
COMP
Vref
C2
R3
C1
R3
R2
C2
R
C
O
O
C1
Figure 48. Types of Frequency Compensation
Aol
A0
P1
Z1
P2
A1
BW
Figure 49. Frequency Response of the Type 2A and Type 2B Frequency Compensation
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DETAILED DESCRIPTION (continued)
Aol(V/V)
Ro =
gmea
(18)
gmea
Co =
2p ´ BW (Hz)
(19)
æ
ç
è
ö
÷
ø
s
1+
2p´ fZ1
EA = A0´
æ
ç
è
ö æ
ö
÷
ø
s
s
1+
´ 1+
÷ ç
2p´ fP1
2p´ fP2
ø è
(20)
(21)
(22)
R2
A0 = gmea ´ Ro ´
R1 + R2
R2
R1 + R2
A1 = gmea ´ Ro| | R3 ´
1
P1=
2p´Ro´ C1
(23)
1
Z1=
2p´R3´ C1
(24)
(25)
(26)
(27)
1
P2 =
type 2a
2p ´ R3 | | R ´ (C2 + Co)
1
P2 =
type 2b
2p ´ R3 | | R ´ Co
1
P2 =
type 1
2p ´ R ´ (C2 + CO )
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APPLICATION INFORMATION
Design Guide — Step-By-Step Design Procedure
This example details the design of a high frequency switching regulator design using ceramic output capacitors.
A few parameters must be known in order to start the design process. These parameters are typically determined
at the system level. For this example, we will start with the following known parameters:
Output Voltage
3.3V
Transient Response 0 to 1.5A load step
Maximum Output Current
Input Voltage
ΔVout = 4%
1.5 A
12 V nom. 8V to 18V
< 33 mVpp
7.25 V
Output Voltage Ripple
Start Input Voltage (rising VIN)
Stop Input Voltage (falling VIN)
6.25 V
Selecting the Switching Frequency
The first step is to decide on a switching frequency for the regulator. Typically, the user will want to choose the
highest switching frequency possible since this will produce the smallest solution size. The high switching
frequency allows for lower valued inductors and smaller output capacitors compared to a power supply that
switches at a lower frequency. The switching frequency that can be selected is limited by the minimum on-time of
the internal power switch, the input voltage and the output voltage and the frequency shift limitation.
Equation 12 and Equation 13 must be used to find the maximum switching frequency for the regulator, choose
the lower value of the two equations. Switching frequencies higher than these values will result in pulse skipping
or the lack of overcurrent protection during a short circuit.
The typical minimum on time, tonmin, is 130 ns for the TPS54140. For this example, the output voltage is 3.3 V
and the maximum input voltage is 18 V, which allows for a maximum switch frequency up to 1600 kHz when
including the inductor resistance, on resistance and diode voltage in Equation 12. To ensure overcurrent
runaway is not a concern during short circuits in your design use Equation 13 or the solid curve in Figure 41 to
determine the maximum switching frequency. With an maximum input voltage of 20 V, assuming a diode voltage
of 0.5V, inductor resistance of 100 mΩ, switch resistance of 200 mΩ, an output current of 2.8A, the maximum
switching frequency is approximately 1600kHz.
Choosing the lower of the two values and adding some margin a switching frequency of 1200kHz is used. To
determine the timing resistance for a given switching frequency, use Equation 11 or the curve in Figure 39.
The switching frequency is set by resistor Rt shown in Figure 50.
L1
10 mH
3.3 V at 1.5 A
0.1 mF
C1
COUT
U1
TPS54140DGQ
D1
B220A
+
47 mF/6.3 V
BOOT
VIN
PH
GND
8 - 18 V
C4
2.2 mF 0.1 mF
C2
C3
R3
EN
COMP
VSNS
R1
CF
SS/TR
RT/CLK
RC
332 kW
2.2 mF
31.6 kW
PWRGD
76.8 kW
6.8 pF
CSS
RT
0.01 mF 90.9 kW
R4
CC
R2
61.9 kW
10 kW
2700 pF
Figure 50. High Frequency, 3.3V Output Power Supply Design with Adjusted UVLO.
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Output Inductor Selection (LO)
To calculate the minimum value of the output inductor, use Equation 28.
KIND is a coefficient that represents the amount of inductor ripple current relative to the maximum output current.
The inductor ripple current will be filtered by the output capacitor. Therefore, choosing high inductor ripple
currents will impact the selection of the output capacitor since the output capacitor must have a ripple current
rating equal to or greater than the inductor ripple current. In general, the inductor ripple value is at the discretion
of the designer; however, the following guidelines may be used.
For designs using low ESR output capacitors such as ceramics, a value as high as KIND = 0.3 may be used.
When using higher ESR output capacitors, KIND = 0.2 yields better results. Since the inductor ripple current is
part of the PWM control system, the inductor ripple current should always be greater than 100 mA for
dependable operation. In a wide input voltage regulator, it is best to choose an inductor ripple current on the
larger side. This allows the inductor to still have a measurable ripple current with the input voltage at its
minimum.
For this design example, use KIND = 0.2 and the minimum inductor value is calculated to be 7.6µH. For this
design, a nearest standard value was chosen: 10µH. For the output filter inductor, it is important that the RMS
current and saturation current ratings not be exceeded. The RMS and peak inductor current can be found from
Equation 30 and Equation 31.
For this design, the RMS inductor current is 1.506 A and the peak inductor current is 1.62 A. The chosen
inductor is a MSS6132-103. It has a saturation current rating of 1.64 A and an RMS current rating of 1.9A.
As the equation set demonstrates, lower ripple currents will reduce the output voltage ripple of the regulator but
will require a larger value of inductance. Selecting higher ripple currents will increase the output voltage ripple of
the regulator but allow for a lower inductance value.
The current flowing through the inductor is the inductor ripple current plus the output current. During power up,
faults or transient load conditions, the inductor current can increase above the calculated peak inductor current
level calculated above. In transient conditions, the inductor current can increase up to the switch current limit of
the device. For this reason, the most conservative approach is to specify an inductor with a saturation current
rating equal to or greater than the switch current limit rather than the peak inductor current.
Vinmax - Vout
Vout
Lo min =
´
Io ´ KIND
Vinmax ´ ƒsw
(28)
(29)
I
£ I ´K
IND
RIPPLE
O
2
æ
ç
ç
è
ö
÷
÷
ø
V
´
Vinmax - V
OUT
(
)
1
2
OUT
I
=
I
+
´
(O )
L(rms)
12
Vinmax ´ L ´ f
O SW
(30)
(31)
Iripple
ILpeak = Iout +
2
Output Capacitor
There are three primary considerations for selecting the value of the output capacitor. The output capacitor will
determine the modulator pole, the output voltage ripple, and how the regulators responds to a large change in
load current. The output capacitance needs to be selected based on the more stringent of these three criteria.
The desired response to a large change in the load current is the first criteria. The output capacitor needs to
supply the load with current when the regulator can not. This situation would occur if there are desired hold-up
times for the regulator where the output capacitor must hold the output voltage above a certain level for a
specified amount of time after the input power is removed. The regulator also will temporarily not be able to
supply sufficient output current if there is a large, fast increase in the current needs of the load such as
transitioning from no load to a full load. The regulator usually needs two or more clock cycles for the control loop
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to see the change in load current and output voltage and adjust the duty cycle to react to the change. The output
capacitor must be sized to supply the extra current to the load until the control loop responds to the load change.
The output capacitance must be large enough to supply the difference in current for 2 clock cycles while only
allowing a tolerable amount of droop in the output voltage. Equation 32 shows the minimum output capacitance
necessary to accomplish this.
Where ΔIout is the change in output current, ƒsw is the regulators switching frequency and ΔVout is the
allowable change in the output voltage. For this example, the transient load response is specified as a 4%
change in Vout for a load step from 0A (no load) to 1.5 A (full load). For this example, ΔIout = 1.5-0 = 1.5 A and
ΔVout = 0.04 × 3.3 = 0.132 V. Using these numbers gives a minimum capacitance of 18.9µF. This value does
not take the ESR of the output capacitor into account in the output voltage change. For ceramic capacitors, the
ESR is usually small enough to ignore in this calculation. Aluminum electrolytic and tantalum capacitors have
higher ESR that should be taken into account.
The catch diode of the regulator can not sink current so any stored energy in the inductor will produce an output
voltage overshoot when the load current rapidly decreases, see Figure 51. The output capacitor must also be
sized to absorb energy stored in the inductor when transitioning from a high load current to a lower load current.
The excess energy that gets stored in the output capacitor will increase the voltage on the capacitor. The
capacitor must be sized to maintain the desired output voltage during these transient periods. Equation 33 is
used to calculate the minimum capacitance to keep the output voltage overshoot to a desired value. Where L is
the value of the inductor, IOH is the output current under heavy load, IOL is the output under light load, VF is the
final peak output voltage, and Vi is the initial capacitor voltage. For this example, the worst case load step will be
from 1.5 A to 0 A. The output voltage will increase during this load transition and the stated maximum in our
specification is 4% of the output voltage. This will make Vf = 1.04 × 3.3 = 3.432. Vi is the initial capacitor voltage
which is the nominal output voltage of 3.3 V. Using these numbers in Equation 33 yields a minimum capacitance
of 25.3µF.
Equation 34 calculates the minimum output capacitance needed to meet the output voltage ripple specification.
Where fsw is the switching frequency, Voripple is the maximum allowable output voltage ripple, and Iripple is the
inductor ripple current. Equation 35 yields 0.7µF.
Equation 35 calculates the maximum ESR an output capacitor can have to meet the output voltage ripple
specification. Equation 35 indicates the ESR should be less than 144mΩ.
The most stringent criteria for the output capacitor is 25.3 µF of capacitance to keep the output voltage in
regulation during an unload transient.
Additional capacitance de-ratings for aging, temperature and dc bias should be factored in which will increase
this minimum value. For this example, a 47 µF 6.3V X7R ceramic capacitor with 5 mΩ of ESR will be used.
Capacitors generally have limits to the amount of ripple current they can handle without failing or producing
excess heat. An output capacitor that can support the inductor ripple current must be specified. Some capacitor
data sheets specify the Root Mean Square (RMS) value of the maximum ripple current. Equation 36 can be used
to calculate the RMS ripple current the output capacitor needs to support. For this application, Equation 36 yields
66mA.
2 ´ DIout
Co >
¦sw ´ DVout
(32)
Ioh2 - Iol2
(
)
Co > Lo ´
V¦2 - Vi2
(
)
(33)
(34)
1
1
Co >
´
VORIPPLE
IRIPPLE
8 ´ ¦sw
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V
ORIPPLE
R
<
ESR
I
RIPPLE
(35)
(36)
Vout ´ (Vin max - Vout)
12 ´ Vin max ´ Lo ´ ¦sw
Icorms =
Catch Diode
The TPS54140 requires an external catch diode between the PH pin and GND. The selected diode must have a
reverse voltage rating equal to or greater than Vinmax. The peak current rating of the diode must be greater than
the maximum inductor current. The diode should also have a low forward voltage. Schottky diodes are typically a
good choice for the catch diode due to their low forward voltage. The lower the forward voltage of the diode, the
higher the efficiency of the regulator will be.
Typically, the higher the voltage and current ratings the diode has, the higher the forward voltage will be. Since
the design example has an input voltage up to 18V, a diode with a minimum of 20V reverse voltage will be
selected.
For the example design, the B220A Schottky diode is selected for its lower forward voltage and it comes in a
larger package size which has good thermal characteristics over small devices. The typical forward voltage of the
B220A is 0.50 volts.
The diode must also be selected with an appropriate power rating. The diode conducts the output current during
the off-time of the internal power switch. The off-time of the internal switch is a function of the maximum input
voltage, the output voltage, and the switching frequency. The output current during the off-time is multiplied by
the forward voltage of the diode which equals the conduction losses of the diode. At higher switch frequencies,
the ac losses of the diode need to be taken into account. The ac losses of the diode are due to the charging and
discharging of the junction capacitance and reverse recovery. Equation 37 is used to calculate the total power
dissipation, conduction losses plus ac losses, of the diode.
The B220A has a junction capacitance of 120pF. Using Equation 37, the selected diode will dissipate 0.632
Watts. This power dissipation, depending on mounting techniques, should produce a 16°C temperature rise in
the diode when the input voltage is 18V and the load current is 1.5A.
If the power supply spends a significant amount of time at light load currents or in sleep mode consider using a
diode which has a low leakage current and slightly higher forward voltage drop.
(Vin max - Vout) ´ Iout ´ Vƒd Cj ´ ƒsw ´ Vin2 + Vƒd2
Pd =
+
2
Vin max
(37)
Input Capacitor
The TPS54140 requires a high quality ceramic, type X5R or X7R, input decoupling capacitor of at least 3 µF of
effective capacitance and in some applications a bulk capacitance. The effective capacitance includes any dc
bias effects. The voltage rating of the input capacitor must be greater than the maximum input voltage. The
capacitor must also have a ripple current rating greater than the maximum input current ripple of the TPS54140.
The input ripple current can be calculated using Equation 38.
The value of a ceramic capacitor varies significantly over temperature and the amount of dc bias applied to the
capacitor. The capacitance variations due to temperature can be minimized by selecting a dielectric material that
is stable over temperature. X5R and X7R ceramic dielectrics are usually selected for power regulator capacitors
because they have a high capacitance to volume ratio and are fairly stable over temperature. The output
capacitor must also be selected with the dc bias taken into account. The capacitance value of a capacitor
decreases as the dc bias across a capacitor increases.
For this example design, a ceramic capacitor with at least a 20V voltage rating is required to support the
maximum input voltage. Common standard ceramic capacitor voltage ratings include 4V, 6.3V, 10V, 16V, 25V,
50V or 100V so a 25V capacitor should be selected. For this example, two 2.2µF, 25V capacitors in parallel have
been selected. Table 1 shows a selection of high voltage capacitors. The input capacitance value determines the
input ripple voltage of the regulator. The input voltage ripple can be calculated using Equation 39. Using the
design example values, Ioutmax = 1.5 A, Cin = 4.4µF, ƒsw = 1200 kHz, yields an input voltage ripple of 71 mV
and a rms input ripple current of 0.701A.
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Vin min - Vout
(
)
Vout
Icirms = Iout ´
´
Vin min
Vin min
(38)
(39)
Iout max ´ 0.25
Cin ´ ¦sw
ΔVin =
Table 1. Capacitor Types
VENDOR
VALUE (µF)
EIA Size
VOLTAGE
100 V
50 V
DIALECTRIC
COMMENTS
1.0 to 2.2
1.0 to 4.7
1.0
1210
GRM32 series
Murata
100 V
50 V
1206
2220
2225
1812
1210
1210
1812
GRM31 series
VJ X7R series
1.0 to 2.2
1.0 10 1.8
1.0 to 1.2
1.0 to 3.9
1.0 to 1.8
1.0 to 2.2
1.5 to 6.8
1.0. to 2.2
1.0 to 3.3
1.0 to 4.7
1.0
50 V
100 V
50 V
Vishay
TDK
100 V
100 V
50 V
X7R
C series C4532
C series C3225
100 V
50 V
50 V
100 V
50 V
AVX
X7R dielectric series
1.0 to 4.7
1.0 to 2.2
100 V
Slow Start Capacitor
The slow start capacitor determines the minimum amount of time it will take for the output voltage to reach its
nominal programmed value during power up. This is useful if a load requires a controlled voltage slew rate. This
is also used if the output capacitance is very large and would require large amounts of current to quickly charge
the capacitor to the output voltage level. The large currents necessary to charge the capacitor may make the
TPS54140 reach the current limit or excessive current draw from the input power supply may cause the input
voltage rail to sag. Limiting the output voltage slew rate solves both of these problems.
The slow start time must be long enough to allow the regulator to charge the output capacitor up to the output
voltage without drawing excessive current. Equation 40 can be used to find the minimum slow start time, tss,
necessary to charge the output capacitor, Cout, from 10% to 90% of the output voltage, Vout, with an average
slow start current of Issavg. In the example, to charge the 47µF output capacitor up to 3.3V while only allowing
the average input current to be 0.125A would require a 1 ms slow start time.
Once the slow start time is known, the slow start capacitor value can be calculated using Equation 6. For the
example circuit, the slow start time is not too critical since the output capacitor value is 47µF which does not
require much current to charge to 3.3V. The example circuit has the slow start time set to an arbitrary value of
1ms which requires a 3.3 nF capacitor.
Cout ´ Vout ´ 0.8
Tss >
Issavg
(40)
Bootstrap Capacitor Selection
A 0.1-µF ceramic capacitor must be connected between the BOOT and PH pins for proper operation. It is
recommended to use a ceramic capacitor with X5R or better grade dielectric. The capacitor should have a 10V
or higher voltage rating.
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Under Voltage Lock Out Set Point
The Under Voltage Lock Out (UVLO) can be adjusted using an external voltage divider on the EN pin of the
TPS54140. The UVLO has two thresholds, one for power up when the input voltage is rising and one for power
down or brown outs when the input voltage is falling. For the example design, the supply should turn on and start
switching once the input voltage increases above 7.25V (enabled). After the regulator starts switching, it should
continue to do so until the input voltage falls below 6.25V (UVLO stop).
The programmable UVLO and enable voltages are set using a resistor divider between Vin and ground to the EN
pin. Equation 2 through Equation 3 can be used to calculate the resistance values necessary. For the example
application, a 332kΩ between Vin and EN and a 61.9kΩ between EN and ground are required to produce the
7.25 and 6.25 volt start and stop voltages.
Output Voltage and Feedback Resistors Selection
For the example design, 10.0 kΩ was selected for R2. Using Equation 1, R1 is calculated as 31.25 kΩ. The
nearest standard 1% resistor is 31.6 kΩ. Due to current leakage of the VSENSE pin, the current flowing through
the feedback network should be greater than 1 µA in order to maintain the output voltage accuracy. This
requirement makes the maximum value of R2 equal to 800 kΩ. Choosing higher resistor values will decrease
quiescent current and improve efficiency at low output currents but may introduce noise immunity problems.
Compensation
There are several industry techniques used to compensate DC/DC regulators. The method presented here yields
high phase margins. For most conditions, the regulator will have a phase margin between 60 and 90 degrees.
The method presented here ignores the effects of the slope compensation that is internal to the TPS54140.
Since the slope compensation is ignored, the actual cross over frequency is usually lower than the cross over
frequency used in the calculations.
Use SwitcherPro software for a more accurate design.
The uncompensated regulator will have a dominant pole, typically located between 300 Hz and 3 kHz, due to the
output capacitor and load resistance and a pole due to the error amplifier. One zero exists due to the output
capacitor and the ESR. The zero frequency is higher than either of the two poles.
If left uncompensated, the double pole created by the error amplifier and the modulator would lead to an unstable
regulator. To stabilize the regulator, one pole must be canceled out. One design approach is to locate a
compensating zero at the modulator pole. Then select a cross over frequency that is higher than the modulator
pole. The gain of the error amplifier can be calculated to achieve the desired cross over frequency. The capacitor
used to create the compensation zero along with the output impedance of the error amplifier form a low
frequency pole to provide a minus one slope through the cross over frequency. Then a compensating pole is
added to cancel the zero due to the output capacitors ESR. If the ESR zero resides at a frequency higher than
the switching frequency then it can be ignored.
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To compensate the TPS54140 using this method, first calculate the modulator pole and zero using the following
equations:
Ioutmax
¦p mod =
2 × p × Vout × Cout
(41)
Where Ioutmax is the maximum output current, Cout is the output capacitance and Vout is the nominal output
voltage.
1
¦z mod =
2 ´ p ´ Resr × Cout
(42)
For the example design, the modulator pole is located at 1.5 kHz and the ESR zero is located at 338 kHz.
Next, the designer needs to select a crossover frequency which will determine the bandwidth of the control loop.
The cross over frequency must be located at a frequency at least five times higher than the modulator pole. The
cross over frequency must also be selected so that the available gain of the error amplifier at the cross over
frequency is high enough to allow for proper compensation.
Equation 47 is used to calculate the maximum cross over frequency when the ESR zero is located at a frequency
that is higher than the desired cross over frequency. This will usually be the case for ceramic or low ESR
tantalum capacitors. Aluminum Electrolytic and Tantalum capacitors will typically produce a modulator zero at a
low frequency due to their high ESR.
The example application is using a low ESR ceramic capacitor with 10mΩ of ESR making the zero at 338 kHz.
This value is much higher than typical crossover frequencies so the maximum crossover frequency is calculated
using both Equation 43 and Equation 46.
Using Equation 46 gives a minimum crossover frequency of 7.6 kHz and Equation 43 gives a maximum
crossover frequency of 45.3 kHz.
A crossover frequency of 45 kHz is arbitrarily selected from this range.
F
pmod
F
£ 2100
for ceramic capacitors.
c max
Vout
(43)
(44)
51442
F
£
for Tantalum or Aluminum capacitors.
c max
Vout
Fsw
F
£
for all cases.
³ 5 ´F for all cases.
pmod
c max
5
(45)
(46)
F
c min
Once a cross over frequency, Fc, has been selected, the gain of the modulator at the cross over frequency is
calculated. The gain of the modulator at the cross over frequency is calculated using Equation 47 .
6.6 ´ Rload ´ 2p ´ F ´ C
´ Resr +1
(
)
C
out
Gmod ¦c
=
é
ù
+ Resr +1
2p ´ FC ´ Cout ´ R
(
)
load
ë
û
(47)
For the example problem, the gain of the modulator at the cross over frequency is 0.542. Next, the compensation
components are calculated. A resistor in series with a capacitor is used to create a compensating zero. A
capacitor in parallel to these two components forms the compensating pole. However, calculating the values of
these components varies depending on if the ESR zero is located above or below the cross over frequency. For
ceramic or low ESR tantalum output capacitors, the zero will usually be located above the cross over frequency.
For aluminum electrolytic and tantalum capacitors, the modulator zero is usually located lower in frequency than
the cross over frequency. For cases where the modulator zero is higher than the cross over frequency (ceramic
capacitors).
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Vo
Rc =
G mode ¦c ´ 80 ´ 10-6
(48)
1
Cc =
p × Rc × ¦p mod
(49)
Co × Resr
C¦ =
Rc
(50)
For cases where the modulator zero is less than the cross over frequency (Aluminum or Tantalum capacitors),
the equations are:
Vo ´ Fc
Rc =
G mode ¦c ´ ¦z mod ´ 80 ´ 10-6
(51)
1
Cc =
p × Rc × ¦p mod
(52)
1
C¦ =
2 ´ p ´ Rc ´ ¦z mod
(53)
For the example problem, the ESR zero is located at a higher frequency compared to the cross over frequency
so Equation 50 through Equation 53 are used to calculate the compensation components. For the example
problem, the components are calculated to be: Rc= 76.2kΩ, Cc= 2710pF, and Cf =6.17pF.
The calculated value of the Cf capacitor is not a standard value so a value of 2700pF will be used. 6.8pF is used
for Cc. Rc resistor sets the gain of the error amplifier which determines the cross over frequency. The calculated
Rc resistor is not a standard value, so 76.8kΩ will be used.
APPLICATION CURVES
VIN
VO
VOUT
EN
IO
IL
Figure 51. Load Transmit
Figure 52. Startup With EN
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VOUT
VOUT
IL
PH
VIN
IL
Figure 53. VIN Power Up
Figure 54. Output Ripple CCM
VOUT
VOUT
IL
IL
PH
PH
Figure 55. Output Ripple, DCM
Figure 56. Output Ripple, PSM
VIN
VIN
IL
IL
PH
PH
Figure 57. Input Ripple CCM
Figure 58. Input Ripple DCM
36
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TPS54140
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95
VO = 3.3 V,
VI = 8 V
fsw = 1200 kHz
90
85
80
VIN
VI = 12 V
VI = 16 V
75
70
65
60
IL
PH
55
50
0
0.25
0.50
0.75
1
1.25
1.5
1.75
2
IL - Load Current - A
Figure 59. Input Ripple PSM
Figure 60. Efficiency vs Load Current
60
40
1.015
1.010
1.005
150
100
50
VI = 12 V
Phase
20
0
0
1.000
0.995
Gain
-50
-100
-150
-20
-40
0.990
0.985
1-103
1-104
1-105
1-106
100
0.00
0.25
0.50
0.75
1.00
1.25
1.50
1.75
2.00
f - Frequency - Hz
Load Current - A
Figure 61. Overall Loop Frequency Response
Figure 62. Regulation vs Load Current
1.015
IO = 0.5 A
1.010
1.005
1.000
0.995
0.990
0.985
5
10
15
20
V
- Input Voltage - V
I
Figure 63. Regulation vs Input Voltage
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Power Dissipation Estimate
The following formulas show how to estimate the IC power dissipation under continuous conduction mode (CCM)
operation. These equations should not be used if the device is working in discontinuous conduction mode (DCM).
The power dissipation of the IC includes conduction loss (Pcon), switching loss (Psw), gate drive loss (Pgd) and
supply current (Pq).
Vo
Pcon = Io2 ´ Rdson ´
Vin
(54)
Psw = Vin 2 ´ ¦sw ´ lo ´ 0.25 ´ 10-9
(55)
Pgd = Vin ´ 3 ´ 10-9 ´ ¦sw
(56)
Pq = 116 ´ 10-6 ´ Vin
(57)
Where:
IOUT is the output current (A).
Rdson is the on-resistance of the high-side MOSFET (Ω).
VOUT is the output voltage (V).
VIN is the input voltage (V).
fsw is the switching frequency (Hz).
So
Ptot = Pcon + Psw + Pgd + Pq
(58)
For given TA,
TJ = TA + Rth ´ Ptot
(59)
For given TJMAX = 150°C
TAmax = TJmax - Rth ´ Ptot
(60)
Where:
Ptot is the total device power dissipation (W).
TA is the ambient temperature (°C).
TJ is the junction temperature (°C).
Rth is the thermal resistance of the package (°C/W).
TJMAX is maximum junction temperature (°C).
TAMAX is maximum ambient temperature (°C).
There will be additional power losses in the regulator circuit due to the inductor ac and dc losses, the catch diode
and trace resistance that will impact the overall efficiency of the regulator.
38
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Layout
Layout is a critical portion of good power supply design. There are several signals paths that conduct fast
changing currents or voltages that can interact with stray inductance or parasitic capacitance to generate noise
or degrade the power supplies performance. To help eliminate these problems, the VIN pin should be bypassed
to ground with a low ESR ceramic bypass capacitor with X5R or X7R dielectric. Care should be taken to
minimize the loop area formed by the bypass capacitor connections, the VIN pin, and the anode of the catch
diode. See Figure 64 for a PCB layout example. The GND pin should be tied directly to the power pad under the
IC and the power pad.
The power pad should be connected to any internal PCB ground planes using multiple vias directly under the IC.
The PH pin should be routed to the cathode of the catch diode and to the output inductor. Since the PH
connection is the switching node, the catch diode and output inductor should be located very close to the PH
pins, and the area of the PCB conductor minimized to prevent excessive capacitive coupling. For operation at full
rated load, the top side ground area must provide adequate heat dissipating area. The RT/CLK pin is sensitive to
noise so the RT resistor should be located as close as possible to the IC and routed with minimal lengths of
trace. The additional external components can be placed approximately as shown. It may be possible to obtain
acceptable performance with alternate PCB layouts, however this layout has been shown to produce good
results and is meant as a guideline.
Vout
Output
Capacitor
Output
Inductor
Topside
Ground
Route Boot Capacitor
Catch
Area
Trace on another layer to
provide wide path for
topside ground
Diode
Input
Bypass
Capacitor
BOOT
VIN
PH
GND
Vin
EN
COMP
UVLO
SS/TR
RT/CLK
VSENSE
PWRGD
Compensation
Network
Adjust
Resistor
Divider
Resistors
Slow Start
Capacitor
Frequency
Thermal VIA
Signal VIA
Set Resistor
Figure 64. PCB Layout Example
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PACKAGE OPTION ADDENDUM
www.ti.com
28-Nov-2008
PACKAGING INFORMATION
Orderable Device
Status (1)
Package Package
Pins Package Eco Plan (2) Lead/Ball Finish MSL Peak Temp (3)
Qty
Type
Drawing
TPS54140DGQ
ACTIVE
MSOP-
Power
PAD
DGQ
10
80 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM
no Sb/Br)
TPS54140DGQR
ACTIVE
MSOP-
Power
PAD
DGQ
10
2500 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM
no Sb/Br)
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and
package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS
compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is
provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the
accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take
reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on
incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited
information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI
to Customer on an annual basis.
Addendum-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
1-Dec-2008
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0 (mm)
B0 (mm)
K0 (mm)
P1
W
Pin1
Diameter Width
(mm) W1 (mm)
(mm) (mm) Quadrant
TPS54140DGQR
MSOP-
Power
PAD
DGQ
10
2500
330.0
12.4
5.3
3.3
1.3
8.0
12.0
Q1
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
1-Dec-2008
*All dimensions are nominal
Device
Package Type Package Drawing Pins
MSOP-PowerPAD DGQ 10
SPQ
Length (mm) Width (mm) Height (mm)
370.0 355.0 55.0
TPS54140DGQR
2500
Pack Materials-Page 2
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