TPS61005 [TI]
SINGLE-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD; 与启动，并进入满负载单电池升压转换器
| 型号： | TPS61005 |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | SINGLE-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD |
| 文件： | 总22页 (文件大小：390K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006
SINGLE-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279A – MARCH 2000 – REVISED MAY 2000
Start Up Into a Full Load With Supply
Voltages as Low as 0.9 V Over Full
Temperature Range
Low-EMI Converter (Integrated Antiringing
Switch Across Inductor)
Micro-Size 10-Pin MSOP Package
Minimum 100-mA Output Current From
0.8 V Supply Voltage
Evaluation Modules Available
(TPS6100xEVM–156)
High Power Conversion Efficiency,
up to 90%
Applications Include:
– Single- and Dual-Cell Battery Operated
Products
– MP3-Players and Wireless Headsets
– Pagers and Cordless Phones
– Portable Medical Diagnostic Equipment
– Remote Controls
Power-Save Mode for Improved Efficiency
at Low Output Currents
Device Quiescent Current Less Than 50 µA
Added System Security With Integrated
Low-Battery Comparator
·
description
The TPS6100x devices are boost converters intended for systems that are typically operated from a single- or
dual-cell nickel-cadmium (NiCd), nickel-metal hydride (NiMH), or alkaline battery. The converter output voltage
can be adjusted from 1.5 V to a maximum of 3.3 V and provides a minimum output current of 100 mA. The
converter starts up into a full load with a supply voltage of 0.9 V and stays in operation with supply voltages as
low as 0.8 V.
The converter is based on a fixed-frequency, current-mode pulse-width-modulation (PWM) controller that goes
into power-save mode at low load currents. The current through the switch is limited to a maximum of 1100 mA,
depending on the output voltage. The current sense is integrated to further minimize external component count.
The converter can be disabled to minimize battery drain when the system is put into standby.
A low-EMI mode is implemented to reduce interference and radiated electromagnetic energy that is caused by
the ringing of the inductor when the inductor discharge-current decreases to zero. The device is packaged in
the space saving 10-pin MSOP package.
TPS61006
L1
D1
V
O
= 3.3 V
START UP TIMING INTO 33 Ω LOAD
140
120
33 µH
C
10 µF
i
7
V
OUT
C
o
SW
3
2
6
V
OUT
22 µF
V
BAT
5
R3
100
80
R1
R2
9
LBI
LBO 10
Low Battery
Warning
I
OUT
TPS61006
8
1
NC
EN
FB
3
2
60
40
20
0
R4
10 kΩ
ON
1
0
COMP
OFF
GND
4
C2
33 nF
C1
100 pF
EN
TYPICAL APPLICATION CIRCUIT FOR FIXED
OUTPUT VOLTAGE OPTION
0
2
4
6
8
10 12 14 16 18 20
time – ms
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
Copyright 2000, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
1
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006
SINGLE-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279A – MARCH 2000 – REVISED MAY 2000
AVAILABLE OPTIONS
OUTPUT VOLTAGE
(V)
MARKING DGS
PACKAGE
†
PART NUMBER
T
A
PACKAGE
Adj. from 1.5 V to 3.3 V
TPS61000DGS
TPS61001DGS
TPS61002DGS
TPS61003DGS
TPS61004DGS
TPS61005DGS
TPS61006DGS
ADA
1.5
1.8
2.5
2.8
3.0
3.3
ADB
ADC
–40°C to 85°C
10-Pin MSOP DGS
ADD
ADE
ADF
ADG
†
The DGS package is available taped and reeled. Add R suffix to device type (e.g. TPS61000DGSR) to order quantities of
3000 devices per reel.
Terminal Functions
TERMINAL
I/O
DESCRIPTION
NAME
NO.
Compensation of error amplifier. Connect R-C-C network to set frequency response of control loop. See the
Application section for more details.
COMP
2
1
Chip-enable input. The converter is switched on if EN is set high and is switched off when EN is connected to ground
(shutdown mode).
EN
I
I
Feedback input for adjustable output voltage (TPS61000 only). The output voltage is programmed depending on the
values of resistors R1 and R2. For the fixed output voltage versions (TPS61001, 2, 3, 4, 5, 6), leave the FB pin
unconnected.
FB
3
4
9
GND
LBI
Ground
Low-battery detector input. A low-battery signal is generated at the LBO pin when the voltage on LBI drops below the
I
threshold of 500 mV. Connect LBI to GND or V
pin floating.
if the low-battery detector function is not used. Do not leave this
BAT
Open-drain low-battery detector output. This pin is pulled low if the voltage on LBI drops below the threshold of
LBO
10
O
500 mV. A pull-up resistor should be connected between LBO and V
.
OUT
NC
8
7
6
5
Not connected
SW
I
I
Switch input pin. The node between inductor and anode of the rectifier diode is connected to this pin.
Supply pin
V
V
BAT
O
Output voltage. For the fixed output voltage versions, the integrated resistive divider is connected to this pin.
OUT
DGS PACKAGE
(TOP VIEW)
EN
COMP
FB
LBO
LBI
NC
1
2
3
4
5
10
9
8
GND
7
SW
6
V
V
BAT
OUT
2
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006
SINGLE-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279A – MARCH 2000 – REVISED MAY 2000
functional block diagram
fixed output-voltage option
L1
D1
C
I
C
V
OUT
O
SW
Anti-Ringing
Comparator
and Switch
V
BAT
UVLO
Control Logic
Oscillator
Current Sense
Current Limit
Slope Compensation
EN
LBI/LBO
Comparator
Gate Drive
LBI
V
REF
Comparator
Bandgap
Reference
Error
Amplifier
LBO
GND
COMP
adjustable output-voltage option
L1
D1
C
I
C
O
SW
Anti-Ringing
Comparator
and Switch
V
OUT
V
BAT
UVLO
Control Logic
Oscillator
Current Sense
Current Limit
Slope Compensation
EN
LBI/LBO
Comparator
Gate Drive
LBI
FB
V
REF
Comparator
Bandgap
Reference
Error
Amplifier
LBO
GND
COMP
3
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006
SINGLE-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279A – MARCH 2000 – REVISED MAY 2000
detailed description
controller circuit
The device is based on a current-mode control topology using a constant-frequency pulse-width modulator to
regulate the output voltage. It runs at an oscillator frequency of 500 kHz. The current sense is implemented by
measuring the voltage across the switch. The controller also limits the current through the power switch on a
pulse by pulse basis. Care must be taken that the inductor saturation current is higher than the current limit of
the TPS6100x. This prevents the inductor from going into saturation and therefore protects both device and
inductor. The current limit should not become active during normal operating conditions.
The TPS6100x is designed for high efficiency over a wide output current range. Even at light loads the efficiency
stays high because the controller enters a power-save mode, minimizing switching losses of the converter. In
this mode, the controller only switches if the output voltage trips below a set threshold voltage. It ramps up the
output voltage with one or several pulses, and again goes into the power-save mode once the output voltage
exceeds the threshold voltage. The controller enters the power-save mode when the output current drops to
levels that force the discontinuous current mode. It calculates a minimum duty cycle based on input and output
voltage and uses the calculation for the transition out of the power-save mode into continuous current mode.
The control loop must be externally compensated with an R/C/C network connected to the COMP pin. See the
application section for more details on the design of the compensation network.
device enable
The device is put into operation when EN is set high. During start-up of the converter the input current from the
battery is limited until the voltage on COMP reaches its operating point. The device is put into a shutdown mode
when EN is set to GND. In this mode, the regulator stops switching and all internal control circuitry including
the low-battery comparator is switched off. The output voltage drops to one diode drop below the input voltage
in shutdown.
under-voltage lockout
An under-voltage lockout function prevents the device start-up if the supply voltage on V
is lower than
BAT
approximately 0.7 V. This under-voltage lockout function is implemented in order to prevent the malfunctioning
of the converter. When in operation and the battery is being discharged, the device will automatically enter the
shutdown mode if the voltage on V
drops below approximately 0.7 V.
BAT
IftheENpinishardwiredtoV
andifthevoltageatV
dropstemporarilybelowtheUVLOthresholdvoltage,
BAT
BAT
the device will switch off and will not start up again automatically, even if the supply voltage rises above 0.9 V.
The device will start up again only after a signal change from low to high on EN or if the battery voltage is
completely removed.
4
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006
SINGLE-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279A – MARCH 2000 – REVISED MAY 2000
detailed description (continued)
low Battery detector circuit (LBI and LBO)
The low-battery detector circuit is typically used to supervise the battery voltage and to generate an error flag
when the battery voltage drops below a user-set threshold voltage. The function is active only when the device
is enabled. When the device is disabled, the LBO pin is high impedance. The LBO pin goes active low when
the voltage on the LBI pin decreases below the set threshold voltage of 500 mV ±15 mV, which is equal to the
internal reference voltage. The battery voltage, at which the detection circuit switches, can be programmed with
a resistive divider connected to the LBI pin. The resistive divider scales down the battery voltage to a voltage
level of 500 mV, which is then compared to the LBI threshold voltage. The LBI pin has a build-in hysteresis of
10 mV. Please see the application section for more details about the programming of the LBI threshold.
If the low-battery detection circuit is not used, the LBI pin should be connected to GND (or to V
pin can be left unconnected. Do not let the LBI pin float.
) and the LBO
BAT
low-EMI switch
The device integrates a circuit which removes the ringing that typically appears on the SW-node when the con-
verter enters the discontinuous current mode. In this case, the current through the inductor ramps to zero and
the Schottky diode stops conducting. Due to remaining energy that is stored in parasitic components of diode,
inductor and switch, a ringing on the SW pin is induced. The integrated anti-ringing switch clamps this voltage
internally to V
and therefore dampens this ringing.
BAT
The anti-ringing switch is turned on by a comparator that monitors the voltage between SW and V
. This
OUT
voltage indicates when the diode is reverse biased. The ringing on the SW-node is damped to a large degree,
reducing the electromagnetic interference generated by the switching regulator to a very great extends.
adjustable output voltage
The accuracy of the internal voltage reference, the controller topology, and the accuracy of the external resistor
divider determine the accuracy of the adjustable output voltage version of the TPS61000. The reference voltage
has an accuracy of ±4% over line, load, and temperature. The controller switches between fixed frequency and
pulse-skip mode, depending on load current. This adds an offset to the output voltage that is equivalent to 1%
of V . Using 1% accurate resistors for the feedback divider, a total accuracy of ±6% can be achieved over the
O
complete output current range.
5
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006
SINGLE-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279A – MARCH 2000 – REVISED MAY 2000
†
absolute maximum ratings
Input voltage range, V (V
, V
, COMP, FB, LBO, EN, LBI) . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to 3.6 V
I
BAT OUT
Input voltage, V (SW) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to 7 V
I
Peak current into SW . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1300 mA
Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . See dissipation rating table
Operating free-air temperature range, T . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to 85°C
A
Maximum junction temperature, T . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150°C
J
Storage temperature range, T . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to 150°C
stg
Lead temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260°C
†
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
T
≤ 25 C
DERATING FACTOR
ABOVE T = 25 C
A
T
= 70 C
T = 85 C
A
POWER RATING
A
A
PACKAGE
POWER RATING
POWER RATING
DGS
424 mW
3.4 mW/ C
271 mW
220 mW
recommended operating conditions
MIN NOM
MAX
UNIT
Supply voltage at V
Output current
0.8
100
250
10
V
V
BAT
O
V
V
= 1.2 V
BAT
mA
= 2.4 V
BAT
Inductor
33
µH
µF
µF
°C
Input capacitor
Output capacitor
10
22
Operating junction temperature, T
–40
125
J
6
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006
SINGLE-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279A – MARCH 2000 – REVISED MAY 2000
electricalcharacteristicsoverrecommendedoperatingfree-airtemperaturerange, V
=1.2V,EN
BAT
= V
(unless otherwise noted)
BAT
PARAMETER
TEST CONDITIONS
= 33 Ω
MIN
0.9
TYP
MAX
UNIT
V
V
V
V
Input voltage for start-up
R
R
V
V
V
V
I
L
L
Input voltage for start-up
= 3 kΩ,
T
A
= 25 °C
0.8
I
Input voltage once started
I
O
= 100 mA
= 100 mA
O
0.8
I
Programmable output voltage range TPS61000
I
1.5
3.3
1.55
1.55
1.86
1.86
2.58
2.58
2.58
2.89
2.89
2.89
3.1
O
1.2 V
I
O
I
O
I
O
I
O
I
O
I
O
I
O
I
O
I
O
I
O
I
O
I
O
I
O
I
O
I
O
I
O
= 1 mA
1.44
1.45
1.72
1.74
2.40
2.42
2.42
2.68
2.72
2.72
2.88
2.9
1.5
1.5
1.8
1.8
2.5
2.5
2.5
2.8
2.8
2.8
3.0
3.0
3.0
3.3
3.3
3.3
TPS61001
TPS61002
0.8 V < V < V
I
,
,
= 100 mA
= 1 mA
O
O
1.2 V
0.8 V < V < V
I
= 100 mA
= 1 mA
1.2 V
TPS61003 0.8 V < V < V
,
,
= 100 mA
= 200 mA
= 1 mA
I
O
O
1.6 V < V < V
I
1.2 V
V
O
Output voltage
V
TPS61004 0.8 V < V < V
,
,
= 100 mA
= 200 mA
= 1 mA
I
O
O
1.6 V < V < V
I
1.2 V
TPS61005 0.8 V < V < V
,
,
= 100 mA
= 200 mA
= 1 mA
3.1
I
O
O
1.6 V < V < V
I
2.9
3.1
1.2 V
3.16
3.2
3.4
TPS61006 0.8 V < V < V
,
,
= 100 mA
= 200 mA
3.4
I
O
O
1.6 V < V < V
3.2
3.4
I
V = 0.8 V
100
250
I
I
I
Maximum continuous output current
Switch current limit
mA
A
O
V = 1.8 V
I
TPS61001
TPS61002
TPS61003
TPS61004
TPS61005
TPS61006
TPS61006
0.5
0.65
0.9
0.8 V < V < V
SW
I
O
0.95
1
1.1
V
f
Feedback voltage
468
360
500
500
85%
0.18
515
840
mV
FB
Oscillator frequency
Maximum duty cycle
Switch-on resistance
Line regulation (see Note 1)
kHz
D
MAX
r
V
O
= 3.3 V
0.27
Ω
DS(on)
V = 0.8V to 1.25V,
I
I
= 50 mA
O
0.3
%/V
Load regulation fixed output voltage versions
(see Note 1)
V = 1.2 V;
I
I
= 10 mA to 90 mA
O
0.25%
NOTE 1: Line and load regulation is measured as a percentage deviation from the nominal value (i.e. as percentage deviation from the nominal
output voltage). For line regulation, x %/V stands for ±x% change of the nominal output voltage per 1-V change on the input/supply
voltage. For load regulation, y% stands for ±y% change of the nominal output voltage per the specified current change.
7
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006
SINGLE-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279A – MARCH 2000 – REVISED MAY 2000
electricalcharacteristicsoverrecommendedoperatingfree-airtemperaturerange, V
=1.2V,EN
BAT
= V
(unless otherwise noted) (continued)
BAT
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
V
44
6
Quiescent current drawn from power source
(current into V and into V
I
= 0 mA
= V
BAT
O
I
I
µA
Q
)
V
V
O
= 3.4 V
V
OUT
BAT OUT
EN
I,
Shutdown current from power source
(current into V and into V
V
= 0 V
EN
0.2
5
µA
V
SD
)
BAT OUT
0.2 ×
V
EN low-level input voltage
IL
V
BAT
0.8 ×
V
EN high-level input voltage
EN input current
V
IH
V
BAT
EN = GND or V
BAT
0.1
500
10
1
µA
mV
mV
µA
V
V
IL
LBI low-level input voltage threshold
LBI input hysteresis
V
LBI
voltage decreasing
470
530
I
I
LBI input current
0.01
0.04
0.01
0.01
0.1
0.2
1
V
OL
LBO low-level output voltage
LBO output leakage current
FB input bias current (TPS61000 only)
V
LBI
V
LBI
V
FB
= 0 V, V = 3.3 V,
I
= 50 µA
O
OL
= 3.3 V
= 650 mV, V
µA
µA
LBO
I
= 500 mV
0.1
FB
PARAMETER MEASUREMENT INFORMATION
L1
D1
List of Components:
33 µH
IC1: Only fixed output versions
(unless otherwise noted)
L1: Coilcraft DO3308P–333
D1: Motorola Schottky Diode
MBRM120LT3
C
i
7
C
22 µF
10 µF
o
SW
6
V
OUT
V
BAT
5
R3
R1
R2
C :
Ceramic
I
Low Battery
Warning
9
LBI
LBO 10
C
:
O
Ceramic
TPS6100x
8
1
NC
EN
FB
3
2
R4
10 kΩ
ON
COMP
OFF
GND
4
C1
100 pF
C2
33 nF
Figure 1. Circuit Used For Typical Characteristics Measurements
8
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006
SINGLE-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279A – MARCH 2000 – REVISED MAY 2000
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Output Current
vs Inductor Type
vs Input Voltage
vs Input Voltage
vs Output Current
vs Output Current
vs Input Voltage
vs Input Voltage
vs Load Current
vs Output Voltage
2, 3
4
η
Efficiency
5
I
O
Maximum Output Current
Output Voltage
6
V
V
7
O
O
TPS61000 Output Voltage
No-Load Supply Current
Shutdown Current
8
I
Q
9
I
10
11
12
13
14
15
16
17
SD
V
Minimum Start-Up Input Voltage
Switch current limit
I
I
LIM
Output Voltage Ripple Amplitude
Output Voltage Ripple Amplitude
Load Transient Response
Line Transient Response
Start-Up Timing
EFFICIENCY
vs
OUTPUT CURRENT
EFFICIENCY
vs
OUTPUT CURRENT
100
90
80
70
60
50
40
30
20
10
100
90
V = 2.4 V
I
V = 1.2 V
I
V
= 3.3 V
O
80
70
V
O
= 1.5 V
V
= 2.8 V
V
O
= 3.3 V
O
60
50
40
30
20
10
0
0
1
10
100
1000
1
10
100
1000
I
O
– Output Current – mA
I
O
– Output Current – mA
Figure 2
Figure 3
9
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TPS61000, TPS61001, TPS61002, TPS61003, TPS61004, TPS61005, TPS61006
SINGLE-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279A – MARCH 2000 – REVISED MAY 2000
TYPICAL CHARACTERISTICS
EFFICIENCY
vs
INDUCTOR TYPE
100
V = 1.2 V
I
V
= 3.3 V
= 100 mA
95
90
85
80
75
70
65
60
55
50
O
I
O
Coilcraft
DO1608C
Coilcraft
DS1608C
Coiltronics
UP1B
Coiltronics
UP2B
Sumida
CD43
Sumida
CD54
Inductor Type
Figure 4
EFFICIENCY
MAXIMUM OUTPUT CURRENT
vs
vs
INPUT VOLTAGE
INPUT VOLTAGE
95
90
85
80
75
70
65
60
1
0.90
0.80
0.70
0.60
0.50
0.40
0.30
0.20
0.10
0
I
= 50 mA
O
V
= 3.2 V
O
V
= 2.42 V
O
I
= 100 mA
O
V
= 1.75 V
O
V
= 1.45 V
O
0.80
1.30
1.80
2.30
2.80
3.30
0.8
1
1.2 1.4 1.6 1.8
2
2.2 2.4 2.6 2.8
3
V – Input Voltage – V
I
V – Input Voltage – V
I
Figure 5
Figure 6
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SINGLE-CELL BOOST CONVERTER WITH START-UP INTO FULL LOAD
SLVS279A – MARCH 2000 – REVISED MAY 2000
TYPICAL CHARACTERISTICS
TPS61002/3/6
OUTPUT VOLTAGE
vs
TPS61000
OUTPUT VOLTAGE
vs
OUTPUT CURRENT
OUTPUT CURRENT
3.60
3.40
3.20
3
3.60
3.40
3.20
3
V = 1.2 V
I
3.3 V
V
O
= 3.3 V
2.80
2.60
2.40
2
2.80
V
V
= 2.5 V
O
2.5 V
2.60
2.40
2.20
2.00
2
= 1.8 V
10
O
1.8 V
1.80
1.60
1.80
1.60
1
10
100
1000
0.1
1
100
1000
I
O
– Output Current – mA
I
O
– Output Current – mA
Figure 8
Figure 7
NO-LOAD SUPPLY CURRENT
SHUTDOWN CURRENT
vs
vs
INPUT VOLTAGE
INPUT VOLTAGE
45
40
35
30
25
20
15
10
5
1800
1600
1400
1200
1000
800
600
400
200
0
T
= 85°C
T
= 85°C
= 25°C
A
A
T
A
T
A
= –40°C
T
= 25°C
A
T
= –40°C
A
0
0.80
0.80
1.30
1.80
2.30
2.80
3.30
3.80
1.30
1.80
2.30
2.80
3.30
3.80
V – Input Voltage – V
I
V – Input Voltage – V
I
Figure 9
Figure 10
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TYPICAL CHARACTERISTICS
TPS61000
MINIMUM START-UP INPUT VOLTAGE
SWITCH CURRENT LIMIT
vs
vs
LOAD CURRENT
OUTPUT VOLTAGE
0.90
0.85
0.80
0.75
1.5
V
O
= min 3.2 V
V = 1.2 V
I
1
0.5
0.70
0.65
0.60
0
0
10 20 30 40 50 60 70 80 90 100
1.5 1.7 1.9 2.1 2.3 2.5 2.7 2.9 3.1 3.3 3.5
I
O
– Output Current – mA
V
O
– Output Voltage – V
Figure 11
Figure 12
TPS61006
TPS61006
OUTPUT VOLTAGE RIPPLE AMPLITUDE
OUTPUT VOLTAGE RIPPLE AMPLITUDE
3.36
3.34
3.32
3.30
3.34
3.32
3.30
3.28
3.26
3.24
3.22
V = 1.2 V
I
I
O
= 2 mA
V
OUT
V
SW
2
0
3.20
3.18
0
1
2
3
4
5
0
1
2
3
4
5
time – µs
time – ms
Figure 14
Figure 13
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TYPICAL CHARACTERISTICS
TPS61006
TPS61006
LINE TRANSIENT RESPONSE
LOAD TRANSIENT RESPONSE
3.55
3.45
3.35
3.25
I
R
= 50 mA
V = 1.2 V
O
I
= 33 kΩ
R
= 33 kΩ
C
C
3.4
3.3
3.2
V
OUT
1.2
1
60
40
V
BAT
50 mA
5 mA
0.8
20
0
0
1
2
3
4
5
6
7
8
9
10
0
1
2
3
4
5
6
7
8
9
10
time – ms
time – ms
Figure 15
Figure 16
TPS61006
START-UP TIMING INTO 33 Ω LOAD
140
120
V
OUT
3
2
100
80
I
OUT
60
40
1
0
20
0
EN
0
2
4
6
8
10 12 14 16 18 20
time – ms
Figure 17
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APPLICATION INFORMATION
The TPS6100x boost converter family is intended for systems that are powered by a single-cell NiCd or NiMH
battery with a typical terminal voltage between 0.9 V to 1.6 V. It can also be used in systems that are powered
by two-cell NiCd or NiMH batteries with a typical stack voltage between 1.8 V and 3.2 V. Additionally, single-
or dual-cell, primary and secondary alkaline battery cells can be the power source in systems where the
TPS6100x is used.
programming the TPS61000 adjustable output voltage device
The output voltage of the TPS61000 can be adjusted with an external resistor divider. The typical value of the
voltage on the FB pin is 500 mV in fixed frequency operation and 485 mV in the power-save operation mode.
The maximum allowed value for the output voltage is 3.3 V. The current through the resistive divider should be
about100timesgreaterthanthecurrentintotheFBpin. ThetypicalcurrentintotheFBpinis0.01 µA, thevoltage
across R4 is typically 500 mV. Based on those two values, the recommended value for R4 is in the range of
500 kΩ in order to set the divider current at 1 µA. From that, the value of resistor R3, depending on the needed
output voltage V
, can be calculated using the following equation:
OUT
V
V
O
O
R3
R4
1
500 kΩ
1
(1)
V
500 mV
FB
If, as an example, an output voltage of 2.5 V is needed, a 2 MΩ resistor should be chosen for R3.
D1
L1
33 µH
7
SW
C
10 µF
10 V
C
o
i
22 µF
5
V
OUT
10 V
R5
R3
R4
10
3
Low Battery
Warning
6
9
V
LBO
FB
BAT
R1
LBI
TPS61000
R2
1
8
R
10 kΩ
C
EN
NC
2
COMP
GND
4
C
C
C2
33 nF
C1
100 pF
Figure 18. Typical Application Circuit for Adjustable Output Voltage Option
The output voltage of the adjustable output voltage version changes with the output current. Due to
device-internalgroundshift, whichiscausedbythehighswitchcurrent, theinternalreferencevoltageandhence
the voltage on the FB pin increases with increasing output current. Since the output voltage follows the voltage
on the FB pin, the output voltage rises as well with a rate of 1 mV per 1 mA output current increase. Additionally,
when the converter goes into pulse-skip mode at output currents around 5 mA and lower, the output voltage
drops due to the hysteresis of the controller. This hysteresis is about 15 mV measured on the FB pin.
14
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APPLICATION INFORMATION
programming the low battery comparator threshold voltage
The current through the resistive divider should be about 100 times greater than the current into the LBI pin.
The typical current into the LBI pin is 0.01 µA, the voltage across R2 is equal to the reference voltage that is
generated on chip, which has a value of 500 mV ±15 mV. The recommended value for R2 is therefore in the
range of 500 kΩ. From that, the value of resistor R1 depending on the desired minimum battery voltage V
can be calculated using below equation:
,
BAT
V
V
TRIP
BAT
0.5 V
R1
R2
1
500 kΩ
1
(2)
V
REF
For example, if the low-battery detection circuit should flag an error condition on the LBO output pin at a battery
voltage of 1.0 V, a resistor in the range of 500 kΩ should be chosen for R1.
The output of the low battery comparator is a simple open-drain output that goes active low if the battery voltage
drops below the programmed threshold voltage on LBI. The output requires a pullup resistor with a
recommended value of 1MΩ, and should only be pulled up to the V
floating.
. If not used, the LBO pin can be left
OUT
inductor selection
Theoutputfilterofinductiveswitchingregulatorsisalowpassfilterofsecondorder. Itconsistsofaninductorand
a capacitor, often referred to as storage inductor and output capacitor.
To select an inductor, keep the possible peak inductor current below the current limit threshold of the power
switch in your chosen configuration. For example, the current limit threshold of the TPS61000’s switch is
1100 mA at an output voltage of 3.3 V. The highest peak current through the inductor and the switch depends on
the output load, the input (V
current can be done using the following equation:
) and the output voltage (V
). Estimation of the maximum average inductor
BAT
OUT
V
OUT
I
I
x
(3)
L
OUT
V
x 0.8
BAT
For example, for an output current of 100 mA at 3.3 V, at least 515 mA current will flow through the inductor at a
minimum input voltage of 0.8 V.
The second parameter for choosing the inductor is the desired current ripple in the inductor. Normally it is advis-
able to work with a ripple of less than 20% of the average inductor current. A smaller ripple will reduce the mag-
netic hysteresis losses in the inductor as well as output voltage ripple and EMI. But in the same way, regulation
time at load changes will rise. In addition, a larger inductor will increase the total system costs.
15
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APPLICATION INFORMATION
With those parameters it is possible to calculate the value for the inductor:
V
x V
– V
BAT
OUT
BAT
OUT
Parameter f is the switching frequency and ∆I is the ripple current in the inductor, i.e. 20% x I .
L
(4)
∆I x f x V
L
L
L
In this example, the desired inductor will have the value of 12 µH. With this calculated value and the calculated
currents, it is possible to chose a suitable inductor. Care has to be taken that load transients and losses in the
circuitcanleadtohighercurrentsasestimatedinequation3. Also, thelossesintheinductorcausedbymagnetic
hysteresis losses and copper losses are a major parameter for total circuit efficiency.
The following inductors from different suppliers were tested. All will work with the TPS6100x converter within
their specified parameters:
Table 1. Recommended Inductors
VENDOR
PART NUMBER
Coilcraft
DO1608P Series
DS1608P Series
DO3308 Series
UP1B Series
Coiltronics
UP2B Series
Murata
Sumida
LQH3N Series
CD43 Series
CD54 Series
CDR74B Series
NLC453232T Series
TDK
capacitor selection
The major parameter necessary to define the output capacitor is the maximum allowed output voltage ripple of
the converter. This ripple is determined by two parameters of the capacitor, the capacitance and the ESR. It is
possible to calculate the minimum capacitance needed for the defined ripple, supposing that the ESR is zero.
I
x V
– V
OUT
OUT
BAT
OUT
Parameter f is the switching frequency and ∆V is the maximum allowed ripple.
C
(5)
min
f x ∆V x V
With a chosen ripple voltage of 15 mV, a minimum capacitance of 10 µF is needed. The total ripple will be larger
due to the ESR of the output capacitor. This additional component of the ripple can be calculated using the fol-
lowing equation:
∆V
I
x R
(6)
ESR
OUT
ESR
16
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APPLICATION INFORMATION
An additional ripple of 30 mV is the result of using a tantalum capacitor with a low ESR of 300 mΩ. The total ripple
is the sum of the ripple caused by the capacitance and the ripple caused by the ESR of the capacitor. In this
example, the total ripple will be 45 mV. It is possible to improve the design by enlarging the capacitor or using
smaller capacitors in parallel to reduce the ESR or by using better capacitors with lower ESR, like ceramics. For
example, a 10 µF ceramic capacitor with an ESR of 50 mΩ is used on the evaluation module (EVM). Tradeoffs
have to be made between performance and costs of the converter circuit.
A 10 µF input capacitor is recommended to improve transient behavior of the regulator. A ceramic capacitor or a
tantalum capacitor with a 100 nF ceramic capacitor in parallel placed close to the IC is recommended.
rectifier selection
The rectifier diode has a major impact on the overall converter efficiency. Standard diodes are not suitable for
low-voltageswitchedmodepowersupplies. ASchottkydiodewithlowforwardvoltageandfastreverserecovery
should be used as rectifier to minimize overall losses of the dc-dc converter. The maximum current rating of the
diode must be high enough for the application. The maximum diode current is equal to the maximum current in
the inductor that was calculated in equation 3. The maximum reverse voltage is the output voltage. The chosen
diode should therefore have a reverse voltage rating higher than the output voltage.
Table 2. Recommended Diodes
VENDOR
PART NUMBER
Motorola Surface Mount
MBRM120LT3
MBR0520LT1
1N1517
Motorola Axial Lead
ROHM
RB520S-30
RB160L–40
The typical forward voltage of those diodes is in the range of 0.35 to 0.45 V assuming a peak diode current of
600 mA.
compensation of the control loop
An R/C/C network must be connected to the COMP pin in order to stabilize the control loop of the converter. Both
the pole generated by the inductor L1 and the zero caused by the ESR and capacitance of the output capacitor
must be compensated. The network shown in Figure 19 will satisfy these requirements.
R
10 kΩ
C
COMP
C
C
C2
33 nF
C1
100 pF
Figure 19. Compensation of the Control Loop
17
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APPLICATION INFORMATION
Resistor R and capacitor C depend on the chosen inductance. For a 33 µH inductor, the capacitance of C
C2
C
C2
should be chosen to 33 nF, or in other words, if the inductor is xx µH, the chosen compensation capacitor should
be xx nF, the same number value. The value of the compensation resistor is then chosen based on the require-
ment to have a time constant of 0.3 ms for the R/C network of R and C ; hence for a 33-nF capacitor, a 10-kΩ
C
C2
resistor should be chosen for R .
C
Capacitor C is depending on the ESR and capacitance value of the output capacitor, and on the value chosen
C1
for R . Its value is calculated using following equation:
C
C
x ESR
O
COUT
C
C
(7)
C1
3
R
For a selected output capacitor of 22 µF with an ESR of 0.2 Ω, and R of 33 kΩ, the value of C is in the range of
C
C1
100 pF.
Table 3. Recommended Compensation Components
OUTPUT CAPACITOR
INDUCTOR
R
[kΩ]
C
[pF]
C
C2
[nF]
C
C1
CAPACITANCE
ESR
[µH]
[µF]
[Ω]
33
22
10
10
22
22
22
10
0.2
0.3
0.4
0.1
10
15
33
33
100
100
100
100
33
22
10
10
schematic of TPS6100x evaluation modules (TPS6100xEVM–156)
J1
LP1
R6
TPS6100x
C5
R5
EN
LBO
LBI
NC
C6
COMP
FB
L1
R3
R2
R1
OUT
IN
R4
GND
SW
V
OUT
V
BAT
C2
C1
C3
D1
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SLVS279A – MARCH 2000 – REVISED MAY 2000
APPLICATION INFORMATION
suggested board layout and component placement (21 mm x 21 mm board size)
Figure 20. Top Layer Layout and Component Placement
Figure 21. Bottom Layer Layout and Component Placement
device family products
Other devices in this family are:
PART NUMBER
UCC2941-3/-5/-ADJ
UCC3941-3/-5/-ADJ
UCC29411/2/3
DESCRIPTION
1-V synchronous boost converter with secondary output
1-V low power synchronous boost converter with secondary output
UCC39411/2/3
19
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SLVS279A – MARCH 2000 – REVISED MAY 2000
THERMAL INFORMATION
Implementation of integrated circuits in low-profile and fine-pitch surface-mount packages typically requires
special attention to power dissipation. Many system-dependent issues such as thermal coupling, airflow, added
heat sinks and convection surfaces, and the presence of other heat-generating components affect the power-
dissipation limits of a given component.
Three basic approaches for enhancing thermal performance are listed below:
•
•
•
Improving the power dissipation capability of the PWB design
Improving the thermal coupling of the component to the PWB
Introducing airflow in the system
The maximum junction temperature (T ) of the TPS6100x devices is 125°C. The thermal resistance of the
J
10-pin MSOP package (DSG) is R
= 294°C/W. Specified regulator operation is assured to a maximum
θJA
ambient temperature T of 85 °C. Therefore, the maximum power dissipation is about 130 mW. More power can
be dissipated if the maximum ambient temperature of the application is lower.
A
T
J (MAX )
125°C – 85°C
294°C /W
–A
(8)
P
=
=
= 136 mW
D (MAX )
R
Θ JA
Under normal operating conditions, the sum of all losses generated inside the converter IC is less than 50 mW,
which is well below the maximum allowed power dissipation of 136 mW as calculated in equation 8. Therefore,
power dissipation is given no special attention.
Table 4 shows where the losses inside the converter are generated.
Table 4. Losses Inside the Converter
LOSSES
AMOUNTS
Conduction losses in the switch
36 mW
Switching losses
Gate drive losses
Quiescent current losses
TOTAL
8 mW
2.3 mW
< 1 mW
< 50 mW
20
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SLVS279A – MARCH 2000 – REVISED MAY 2000
MECHANICAL DATA
DGS (S-PDSO-G10)
PLASTIC SMALL-OUTLINE PACKAGE
0,27
0,17
M
0,25
0,50
10
6
0,15 NOM
3,05
2,95
4,98
4,78
Gage Plane
0,25
0°–6°
1
5
0,69
0,41
3,05
2,95
Seating Plane
0,10
0,15
0,05
1,07 MAX
4073272/A 03/98
NOTES: A. All linear dimensions are in millimeters.
B. This drawing is subject to change without notice.
C. Body dimensions do not include mold flash or protrusion.
21
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TI warrants performance of its semiconductor products to the specifications applicable at the time of sale in
accordance with TI’s standard warranty. Testing and other quality control techniques are utilized to the extent
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Customers are responsible for their applications using TI components.
In order to minimize risks associated with the customer’s applications, adequate design and operating
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TI assumes no liability for applications assistance or customer product design. TI does not warrant or represent
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Copyright 2000, Texas Instruments Incorporated
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