TPS61040-Q1_16 [TI]
Low-Power DC-DC Boost Converter in SOT-23 Package;
| 型号： | TPS61040-Q1_16 |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | Low-Power DC-DC Boost Converter in SOT-23 Package |
| 文件： | 总18页 (文件大小：289K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
LOW POWER DC/DC BOOST CONVERTER IN SOT-23 PACKAGE
FEATURES
DESCRIPTION
•
•
•
Qualification in Accordance With AEC-Q100(1)
The TPS61040/41 is
a
high-frequency boost
converter dedicated for small to medium LCD bias
supply and white LED backlight supplies. The device
is ideal to generate output voltages up to 28 V from a
dual cell NiMH/NiCd or a single cell Li-Ion battery.
The part can also be used to generate standard 3.3
V/5 V to 12-V power conversions.
Qualified For Automotive Application
Customer-Specifc Configuration Control Can
Be Supported Along With Major-Change
Approval
•
•
•
1.8-V to 6-V Input Voltage Range
Adjustable Output Voltage Range Up to 28 V
The TPS61040/41 operates with a switching fre-
quency up to 1 MHz. This allows the use of small
external components using ceramic as well as tanta-
lum output capacitors. Together with the tiny SOT23
package, the TPS61040/41 gives a small overall
solution size. The TPS61040 has an internal 400-mA
switch current limit, while the TPS61041 has a
250-mA switch current limit, offering lower output
voltage ripple and allows the use of a smaller form
factor inductor for lower power applications. The low
quiescent current (typically 28 µA) together with an
optimized control scheme, allows device operation at
high efficiencies over the entire load current range.
400-mA (TPS61040) and 250-mA (TPS61041)
Internal Switch Current
•
•
•
•
•
Up to 1-MHz Switching Frequency
28-µA Typical No Load Quiescent Current
1-µA Typical Shutdown Current
Internal Softstart
Available in a Tiny 5-Pin SOT23 Package
(1)Contact Texas Instruments for details. Q100
qualification data available on request.
APPLICATIONS
DBV PACKAGE
(TOP VIEW)
•
•
•
•
•
•
•
LCD Bias Supply
White-LED Supply for LCD Backlights
Digital Still Camera
PDAs, Organizers, and Handheld PCs
Cellular Phones
Internet Audio Player
1
2
3
5
4
V
IN
SW
GND
FB
EN
Standard 3.3 V/5 V to 12 V Conversion
TYPICAL APPLICATION
EFFICIENCY
vs
OUTPUT CURRENT
90
88
86
84
82
80
78
76
74
72
70
L1
10 µH
V
= 18 V
D1
O
V = 5 V
I
V
V
OUT
V
IN
to 28 V
IN
1.8 V to 6.0 V
V = 3.6 V
I
C
FF
R1
1
3
2
5
4
V
SW
IN
C
1 µF
O
V = 2.4 V
I
FB
C
4.7 µF
IN
EN
GND
R2
0.10
1
10
100
I
- Output Current - mA
O
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Copyright © 2005, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
These devices have limited built-in ESD protection. The leads should be shorted together or the device
placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates.
(1)
ORDERING INFORMATION
TJ
SWITCH CURRENT LIMIT
400 mA
SOT23 PACKAGE
TPS61040QDBVRQ1
TPS61041QDBVRQ1
PACKAGE MARKING
PHOQ
-40 to 125°C
250 mA
PHPQ
(1) The DBV package is available in tape & reel. Add R suffix (DBVR) to order quantities of 3000 parts.
FUNCTIONAL BLOCK DIAGRAM
SW
Under Voltage
Lockout
Bias Supply
400 ns Min
VIN
FB
Off Time
Error Comparator
-
S
Power MOSFET
N-Channel
+
RS Latch
Logic
Gate
Driver
V
REF
= 1.233 V
R
Current Limit
R
SENSE
+
_
6 µs Max
On Time
EN
Soft
Start
GND
2
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
Terminal Functions
TERMINAL
I/O
DESCRIPTION
NAME NO.
SW
1
I
Connect the inductor and the Schottky diode to this pin. This is the switch pin and is connected to the drain of the
internal power MOSFET.
GND
FB
2
3
Ground
I
I
I
This is the feedback pin of the device. Connect this pin to the external voltage divider to program the desired output
voltage.
EN
4
5
This is the enable pin of the device. Pulling this pin to ground forces the device into shutdown mode reducing the
supply current to less than 1 µA. This pin should not be left floating and needs to be terminated.
VIN
Supply voltage pin
DETAILED DESCRIPTION
OPERATION
The TPS61040/41 operates with an input voltage range of 1.8 V to 6 V and can generate output voltages up to
28 V. The device operates in a pulse frequency modulation (PFM) scheme with constant peak current control.
This control scheme maintains high efficiency over the entire load current range, and with a switching frequency
up to 1 MHz, the device enables the use of very small external components.
The converter monitors the output voltage, and as soon as the feedback voltage falls below the reference voltage
of typically 1.233 V, the internal switch turns on and the current ramps up. The switch turns off as soon as the
inductor current reaches the internally set peak current of typically 400 mA (TPS61040) or 250 mA (TPS61041).
See the Peak Current Control section for more information. The second criteria that turns off the switch is the
maximum on-time of 6 µs (typical). This is just to limit the maximum on-time of the converter to cover for extreme
conditions. As the switch is turned off, the external Schottky diode is forward biased delivering the current to the
output. The switch remains off for a minimum of 400 ns (typical), or until the feedback voltage drops below the
reference voltage again. Using this PFM peak current control scheme, the converter operates in discontinuous
conduction mode (DCM) where the switching frequency depends on the output current, which results in high
efficiency over the entire load current range. This regulation scheme is inherently stable, allowing a wider
selection range for the inductor and output capacitor.
PEAK CURRENT CONTROL
The internal switch turns on until the inductor current reaches the typical dc current limit (ILIM) of 400 mA
(TPS61040) or 250 mA (TPS61042). Due to the internal propagation delay of typical 100 ns, the actual current
exceeds the dc-current limit threshold by a small amount. The typical peak current limit can be calculated:
Vin
L
I
+ I
)
100 ns
peak(typ)
LIM
Vin
L
Vin
L
I
I
+ 400 mA )
+ 250 mA )
100 ns for the TPS61040
100 ns for the TPS61041
peak(typ)
peak(typ)
(1)
The higher the input voltage and the lower the inductor value, the greater the peak.
By selecting the TPS61040 or TPS61041, it is possible to tailor the design to the specific application current limit
requirements. A lower current limit supports applications requiring lower output power and allows the use of an
inductor with a lower current rating and a smaller form factor. A lower current limit usually has a lower output
voltage ripple as well.
SOFTSTART
All inductive step-up converters exhibit high inrush current during start-up if no special precaution is made. This
can cause voltage drops at the input rail during start up and may result in an unwanted or early system shut
down.
3
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
DETAILED DESCRIPTION (continued)
I
LIM
4
The TPS61040/41 limits this inrush current by increasing the current limit in two steps starting from
for 256
I
LIM
2
cycles to
for the next 256 cycles, and then full current limit (see Figure 14).
ENABLE
Pulling the enable (EN) to ground shuts down the device reducing the shutdown current to 1 µA (typical). Since
there is a conductive path from the input to the output through the inductor and Schottky diode, the output
voltage is equal to the input voltage during shutdown. The enable pin needs to be terminated and should not be
left floating. Using a small external transistor disconnects the input from the output during shutdown as shown in
Figure 18.
UNDERVOLTAGE LOCKOUT
An undervoltage lockout prevents misoperation of the device at input voltages below typical 1.5 V. When the
input voltage is below the undervoltage threshold the main switch is turned off.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature (unless otherwise noted)
(1)
UNIT
(2)
Supply voltages on pin VIN
-0.3 V to 7 V
-0.3 V to VIN + 0.3 V
30 V
(2)
Voltages on pins EN, FB
(2)
Switch voltage on pin SW
Continuous power dissipation
Operating junction temperature
Storage temperature
See Dissipation Rating Table
-40°C to 150°C
-65°C to 150°C
260°C
TJ
TStg
Lead temperature (soldering 10 seconds)
(1) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
(2) All voltage values are with respect to network ground terminal.
DISSIPATION RATING TABLE(1)
T
A ≤ 25°C
DERATING FACTOR
ABOVE TA = 25°C
TA = 70°C
POWER RATING
TA = 85°C
POWER RATING
PACKAGE
POWER RATING
DBV
357 mW
3.5 mW/°C
192 mW
140 mW
(1) The thermal resistance junction to ambient of the 5-pin SOT23 is 250°C/W.
RECOMMENDED OPERATING CONDITIONS
MIN
TYP
MAX UNIT
Vin
VOUT
L
Input voltage range
1.8
6
V
V
Output voltage range
Inductor(1)
Switching frequency(1)
28
2.2
10
µH
MHz
µF
µF
°C
f
1
(1)
Cin
COUT
TJ
Input capacitor
4.7
(1)
Output capacitor
1
Operating junction temperature
-40
125
(1) See the Application Section for further information.
4
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
ELECTRICAL CHARACTERISTICS
VIN = 2.4 V, EN = VIN, TJ = -40°C to 125°C, typical values are at TA = 25°C (unless otherwise noted)
PARAMETER
SUPPLY CURRENT
TEST CONDITIONS
MIN
TYP
MAX UNIT
VIN
Input voltage range
1.8
6
50
1
V
µA
µA
V
IQ
Operating quiescent current
Shutdown current
IOUT = 0 mA, not switching, VFB = 1.3 V
EN = GND
28
0.1
1.5
ISD
VUVLO
ENABLE
VIH
Under-voltage lockout threshold
1.7
EN high level input voltage
EN low level input voltage
EN input leakage current
1.3
V
V
VIL
0.4
1
II
EN = GND or VIN
0.1
µA
POWER SWITCH AND CURRENT LIMIT
Vsw
Maximum switch voltage
Minimum off time
30
550
7.5
V
toff
250
4
400
6
ns
ton
Maximum on time
µs
RDS(ON)
RDS(ON)
MOSFET on-resistance
MOSFET on-resistance
MOSFET leakage current
MOSFET current limit
MOSFET current limit
VIN = 2.4 V; Isw = 200 mA; TPS61040
VIN = 2.4 V; Isw = 200 mA; TPS61041
Vsw = 28 V
600
750
1
1100
1300
10
mΩ
mΩ
µA
mA
mA
ILIM
TPS61040
325
200
400
250
500
325
ILIM
TPS61041
OUTPUT
VOUT
Vref
Adjustable output voltage range(1)
Internal voltage reference
VIN
28
V
V
1.233
IFB
Feedback input bias current
Feedback trip point voltage
VFB = 1.3 V
1
µA
V
VFB
1.8 V ≤ VIN ≤ 6 V
1.2 1.233
0.05
1.27
1.8 V≤ VIN ≤ 6 V; VOUT = 18 V; Iload = 10 mA; Cff
= not connected
(2)
Line regulation
%/V
Load regulation(2)
VIN = 2.4 V; VOUT = 18 V; 0 mA ≤ IOUT ≤ 30 mA
0.15
%/mA
(1) Cannot be production tested. Assured by design.
(2) The line and load regulation depend on the external component selection. See the Application Section for further information.
5
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Load current
1, 2, 3
4
η
Efficiency
vs Input voltage
IQ
Quiescent current
Feedback voltage
Switch current limit
vs Input voltage and temperature
vs Temperature
5
VFB
ISW
6
vs Temperature
7
vs Supply voltage, TPS61041
vs Supply voltage, TPS61040
vs Temperature
8
ICL
Switch current limit
RDSon
9
10
11
12
13
14
RDSon
vs Supply voltage
Line transient response
Load transient response
Start-up behavior
EFFICIENCY
EFFICIENCY
vs
LOAD CURRENT
vs
OUTPUT CURRENT
90
88
86
84
82
80
78
76
74
72
70
90
88
86
84
82
80
78
76
74
72
70
V
O
= 18 V
L = 10 µH
= 18 V
V
O
V = 5 V
I
TPS61040
V = 3.6 V
I
TPS61041
V = 2.4 V
I
0.10
1
10
100
0.10
1
10
100
I
O
- Output Current - mA
I
L
- Load Current - mA
Figure 1.
Figure 2.
6
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
EFFICIENCY
vs
LOAD CURRENT
EFFICIENCY
vs
INPUT VOLTAGE
90
V
90
88
86
84
82
80
78
76
74
= 18 V
L = 10 µH
= 18 V
O
88
V
O
I
O
= 10 mA
86
84
L = 10 µH
I
O
= 5 mA
L = 3.3 µH
82
80
78
76
74
72
70
72
70
1
2
3
4
5
6
0.10
1
10
100
I - Load Current - mA
L
V - Input Voltage - V
I
Figure 3.
Figure 4.
TPS61040
QUIESCENT CURRENT
vs
FEEDBACK VOLTAGE
vs
FREE-AIR TEMPERATIRE
INPUT VOLTAGE
40
1.24
T = 85°C
A
35
30
25
20
15
10
1.238
T = 27°C
A
1.236
1.234
T = -40°C
A
V
CC
= 2.4 V
1.232
1.23
5
0
1.8
2.4
3
3.6
4.2
4.8
5.4
6
-40 -20
0
20
40
60
80 100 120
T - Temperature - °C
A
V - Input Voltage - V
I
Figure 5.
Figure 6.
7
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
TPS61040/41
SWITCH CURRENT LIMIT
vs
TPS61041
CURRENT LIMIT
vs
FREE-AIR TEMPERATURE
SUPPLY VOLTAGE
260
258
256
254
252
430
410
390
370
TPS61040
T
A
= 27°C
350
330
310
290
270
250
248
246
244
TPS61041
250
230
242
240
1.8
2.4
3
3.6
4.2
4.8
5.4
6
-40 -30 -20 -10
0
10 20 30 40 50 60 70 80 90
T
A
- Temperature - °C
V
CC
- Supply Voltage - V
Figure 7.
Figure 8.
TPS61040
TPS61040/41
CURRENT LIMIT
vs
STATIC DRAIN-SOURCE ON-STATE RESISTANCE
vs
SUPPLY VOLTAGE
FREE-AIR TEMPERATURE
420
1200
415
410
405
400
395
390
1000
800
600
400
TPS61041
T
A
= 27°C
TPS61040
200
0
385
380
−40−30 −20 −10 0 10 20 30 40 50 60 70 80 90
1.8
2.4
3
3.6
4.2
4.8
5.4
6
T
A
− Temperature − °C
V
CC
- Supply Voltage - V
Figure 9.
Figure 10.
8
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
TPS61040/41
STATIC DRAIN-SOURCE ON-STATE RESISTANCE
vs
SUPPLY VOLTAGE
1000
V
O
= 18 V
900
800
700
600
500
400
300
200
V
I
2.4 V to 3.4 V
TPS61041
TPS61040
V
O
100 mV/div
100
0
200 µS/div
1.8
2.4
3
3.6
4.2
4.8
5.4
6
V
CC
− Supply Voltage − V
Figure 11.
Figure 12. Line Transient Response
V
O
= 18 V
V
O
= 18 V
V
O
V
O
100 mA/div
5 V/div
EN
1 V/div
V
O
1 mA to 10 mA
I
I
50 mA/div
200 µS/div
Figure 13. Load Transient Rresponse
Figure 14. Start-Up Behavior
9
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
APPLICATION INFORMATION
INDUCTOR SELECTION, MAXIMUM LOAD CURRENT
Since the PFM peak current control scheme is inherently stable, the inductor value does not affect the stability of
the regulator. The selection of the inductor together with the nominal load current, input and output voltage of the
application determines the switching frequency of the converter. Depending on the application, inductor values
between 2.2 µH up to 47 µH are recommended. The maximum inductor value is determined by the maximum on
time of the switch, typically 6 µs. The peak current limit of 400 mA/250 mA (typically) should be reached within
this 6-µs period for proper operation.
The inductor value determines the maximum switching frequency of the converter. Therefore, select the inductor
value that ensures the maximum switching frequency at the converter maximum load current is not exceeded.
The maximum switching frequency is calculated by the following formula:
Vin
(Vout–Vin)
min
fS
+
max
I
L Vout
P
(2)
Where:
IP = Peak current as described in the previous peak current control section
L = Selected inductor value
Vinmin = The highest switching frequency occurs at the minimum input voltage
If the selected inductor value does not exceed the maximum switching frequency of the converter, the next step
is to calculate the switching frequency at the nominal load current using the following formula:
2 I
(Vout–Vin ) Vd)
load
fSǒIloadǓ+
2
I
L
P
(3)
Where:
IP = Peak current as described in the previous peak current control section
L = Selected inductor value
Iload = Nominal load current
Vd = Rectifier diode forward voltage (typically 0.3 V)
A smaller inductor value gives a higher converter switching frequency, but lowers the efficiency.
The inductor value has less effect on the maximum available load current and is only of secondary order. The
best way to calculate the maximum available load current under certain operating conditions is to estimate the
expected converter efficiency at the maximum load current. This number can be taken out of the efficiency
graphs shown in Figure 1, Figure 2, Figure 3, and Figure 4. The maximum load current can then be estimated as
follows:
2
P
I
L
fS
max
I
+ h
load max
Where:
2 (Vout * Vin)
(4)
IP = Peak current as described in the previous peak current control section
L = Selected inductor value
fSmax = Maximum switching frequency as calculated previously
η = Expected converter efficiency. Typically 70% to 85%.
10
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
The maximum load current of the converter is the current at the operation point where the converter starts to
enter the continuous conduction mode. Usually the converter should always operate in discontinuous conduction
mode.
Last, the selected inductor should have a saturation current that meets the maximum peak current of the
converter (as calculated in the peak current control section). Use the maximum value for ILIM for this calculation.
Another important inductor parameter is the dc resistance. The lower the dc resistance, the higher the efficiency
of the converter. See the Table 1 and the Typical Applications section for the inductor selection.
Table 1. Recommended Inductor for Typical LCD Bias Supply (see Figure 15)
DEVICE
INDUCTOR VALUE
10 µH
COMPONENT SUPPLIER
Sumida CR32-100
COMMENTS
High efficiency
10 µH
Sumida CDRH3D16-100
Murata LQH4C100K04
Sumida CDRH3D16-4R7
Murata LQH3C4R7M24
High efficiency
TPS61040
10 µH
High efficiency
4.7 µH
Small solution size
Small solution size
4.7 µH
High efficiency
Small solution size
TPS61041
10 µH
Murata LQH3C100K24
SETTING THE OUTPUT VOLTAGE
The output voltage is calculated as:
R1
R2
+ 1.233 V ǒ1 ) Ǔ
V
out
(5)
For battery powered applications, a high impedance voltage divider should be used with a typical value for R2 of
≤200 kΩ and a maximum value for R1 of 2.2 MΩ. Smaller values might be used to reduce the noise sensitivity of
the feedback pin.
A feed-forward capacitor across the upper feedback resistor R1 is required to provide sufficient overdrive for the
error comparator. Without a feed-forward capacitor, or one whose value is too small, the TPS61040/41 shows
double pulses or a pulse burst instead of single pulses at the switch node (SW), causing higher output voltage
ripple. If this higher output voltage ripple is acceptable, the feed-forward capacitor can be left out.
The lower the switching frequency of the converter, the larger the feed-forward capacitor value required. A good
starting point is to use a 10-pF feed-forward capacitor. As a first estimation, the required value for the
feed-forward capacitor at the operation point can also be calculated using the following formula:
1
C
+
FF
fS
20
2 p
R1
(6)
Where:
R1 = Upper resistor of voltage divider
fS = Switching frequency of the converter at the nominal load current (see the previous section for
calculating the switching frequency)
CFF = Choose a value that comes closest to the result of the calculation
11
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
The larger the feed-forward capacitor the worse the line regulation of the device. Therefore, when concern for
line regulation is paramount, the selected feed-forward capacitor should be as small as possible. See the next
section for more information about line and load regulation.
LINE AND LOAD REGULATION
The line regulation of the TPS61040/41 depends on the voltage ripple on the feedback pin. Usually a 50-mV
peak-to-peak voltage ripple on the feedback pin FB gives good results.
Some applications require a very tight line regulation and can only allow a small change in output voltage over a
certain input voltage range. If no feed-feedforwardforward capacitor CFF is used across the upper resistor of the
voltage feedback divider, the device has the best line regulation. Without the feed-forward capacitor the output
voltage ripple is higher because the TPS61040/41 shows output voltage bursts instead of single pulses on the
switch pin (SW), increasing the output voltage ripple. Increasing the output capacitor value reduces the output
voltage ripple.
If a larger output capacitor value is not an option, a feed-forward capacitor CFF can be used as described in the
previous section. The use of a feed-forward capacitor increases the amount of voltage ripple present on the
feedback pin (FB). The greater the voltage ripple on the feedback pin (≥50 mV), the worse the line regulation.
There are two ways to improve the line regulation further:
1. Use a smaller inductor value to increase the switching frequency which will lower the output voltage ripple,
as well as the voltage ripple on the feedback pin.
2. Add a small capacitor from the feedback pin (FB) to ground to reduce the voltage ripple on the feedback pin
down to 50 mV again. As a starting point, the same capacitor value as selected for the feed-forward
capacitor CFF can be used.
OUTPUT CAPACITOR SELECTION
For best output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have a low
ESR value but tantalum capacitors can be used as well, depending on the application.
Assuming the converter does not show double pulses or pulse bursts on the switch node (SW), the output
voltage ripple can be calculated as:
I
L
I
out
1
P
DV
+
–
) I ESR
ǒ
Ǔ
out
P
C
fS(Iout) Vout ) Vd–Vin
out
(7)
Where:
IP = Peak current as described in the previous Peak Current Control section
L = Selected inductor value
Iout = Nominal load current
fS (Iout) = Switching frequency at the nominal load current as calculated previously
Vd = Rectifier diode forward voltage (typically 0.3 V)
Cout = Selected output capacitor
ESR = Output capacitor ESR value
Refer to Table 2 and typical applications section for choosing the output capacitor.
Table 2. Recommended Input and Output Capacitors
DEVICE
CAPACITOR
4.7 µF/X5R/0805
10 µF/X5R/0805
1.0 µF/X7R/1206
1.0 µF/X5R/1206
4.7 µF/X5R/1210
VOLTAGE RATING
COMPONENT SUPPLIER
Tayo Yuden JMK212BY475MG
Tayo Yuden JMK212BJ106MG
Tayo Yuden TMK316BJ105KL
Tayo Yuden GMK316BJ105KL
Tayo Yuden TMK325BJ475MG
COMMENTS
CIN/COUT
CIN/COUT
COUT
6.3 V
6.3 V
25 V
35 V
25 V
TPS61040/41
COUT
COUT
12
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
INPUT CAPACITOR SELECTION
For good input voltage filtering, low ESR ceramic capacitors are recommended. A 4.7-µF ceramic input capacitor
is sufficient for most of the applications. For better input voltage filtering this value can be increased. See Table 2
and the Typical Applications section for input capacitor recommendations.
DIODE SELECTION
To achieve high efficiency a Schottky diode should be used. The current rating of the diode should meet the
peak current rating of the converter as it is calculated in the section peak current control. Use the maximum
value for ILIM for this calculation. See Table 3 and the Typical Applications section for the selection of the
Schottky diode.
Table 3. Recommended Schottky Diode for Typical LCD Bias Supply (see Figure 15)
DEVICE
REVERSE VOLTAGE
COMPONENT SUPPLIER
ON Semiconductor MBR0530
ON Semiconductor MBR0520
ON Semiconductor MBRM120L
Toshiba CRS02
COMMENTS
30 V
20 V
20 V
30 V
TPS61040/41
High efficiency
LAYOUT CONSIDERATIONS
Typical for all switching power supplies, the layout is an important step in the design; especially at high peak
currents and switching frequencies. If the layout is not carefully done, the regulator might show noise problems
and duty cycle jitter.
The input capacitor should be placed as close as possible to the input pin for good input voltage filtering. The
inductor and diode should be placed as close as possible to the switch pin to minimize the noise coupling into
other circuits. Since the feedback pin and network is a high impedance circuit the feedback network should be
routed away from the inductor. The feedback pin and feedback network should be shielded with a ground plane
or trace to minimize noise coupling into this circuit.
Wide traces should be used for connections in bold as shown in Figure 15. A star ground connection or ground
plane minimizes ground shifts and noise.
D1
L1
V
O
C
FF
R1
V
V
IN
SW
FB
IN
C
O
C
IN
R2
EN
GND
Figure 15. Layout Diagram
13
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
L1
10 µH
D1
V
18 V
OUT
V
IN
1.8 V to 6 V
TPS61040
C
22 pF
FF
R1
2.2 MW
V
IN
SW
FB
C2
1 µF
C1
4.7 µF
L1:
D1:
C1:
C2:
Sumida CR32-100
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden TMK316BJ105KL
EN
GND
R2
160 kW
Figure 16. LCD Bias Supply
L1
10 µH
D1
V
O
18 V
TPS61040
C
FF
R1
2.2 MW
22 pF
V
V
IN
SW
FB
IN
C2
1 µF
1.8 V to 6 V
C1
4.7 µF
DAC or Analog Voltage
0 V = 25 V
1.233 V = 18 V
EN
GND
R2
160 kW
L1:
Sumida CR32-100
D1:
C1:
C2:
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden GMK316BJ105KL
Figure 17. LCD Bias Supply With Adjustable Output Voltage
R3
200 kW
BC857C
L1
10 µH
D1
V
IN
V
OUT
1.8 V to 6 V
18 V / 10 mA
TPS61040
R1
C
FF
2.2 MW
22 pF
V
IN
SW
FB
C2
1 µF
C3
0.1 µF
(Optional)
C1
4.7 µF
R2
160 kW
EN
GND
L1:
D1:
C1:
C2:
Sumida CR32-100
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden TMK316BJ105KL
Figure 18. LCD Bias Supply With Load Disconnect
14
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
D3
V2 = -10 V/15 mA
D2
C4
4.7 µF
C3
1 µF
L1
D1
6.8 µH
V1 = 10 V/15 mA
TPS61040
C
FF
22 pF
R1
1.5 MW
V
SW
FB
IN
V
IN
= 2.7 V to 5 V
C2
1 µF
C1
4.7 µF
L1:
Murata LQH4C6R8M04
D1, D2, D3: Motorola MBR0530
EN
GND
R2
C1:
Tayo Yuden JMK212BY475MG
210 kW
C2, C3, C4: Tayo Yuden EMK316BJ105KF
Figure 19. Positive and Negative Output LCD Bias Supply
L1
6.8 µH
D1
V
O =
12 V/35 mA
TPS61040
C
4.7 pF
FF
R1
1.8 MW
V
IN
3.3 V
V
IN
SW
FB
C2
4.7 µF
C1
10 µF
L1:
Murata LQH4C6R8M04
Motorola MBR0530
Tayo Yuden JMK212BJ106MG
Tayo Yuden EMK316BJ475ML
EN
GND
R2
205 kW
D1:
C1:
C2:
Figure 20. Standard 3.3-V to 12-V Supply
D1
3.3 µH
TPS61040
5 V/45 mA
C
FF
3.3 pF
R1
620 kW
V
1.8 V to 4 V
SW
FB
IN
C2
4.7 µF
C1
4.7 µF
R2
200 kW
EN
GND
L1:
D1:
Murata LQH4C3R3M04
Motorola MBR0530
C1, C2: Tayo Yuden JMK212BY475MG
Figure 21. Dual Battery Cell to 5 V/50-mA Conversion
Efficiency Aprox. Equals 84% at VIN = 2.4 V to VO = 5 V/45 mA
15
TPS61040-Q1
TPS61041-Q1
www.ti.com
SGLS276–JANUARY 2005
L1
D1
10 µH
D2
24 V
V
CC
= 2.7 V to 6 V
V
IN
SW
(Optional)
C1
4.7 µF
FB
L1:
D1:
C1:
C2:
Murata LQH4C100K04
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden TMK316BJ105KL
C2
1 µF
R
82 Ω
EN
PWM
100 Hz to 500 Hz
GND
S
Figure 22. White LED Supply With Adjustable Brightness Control
Using a PWM Signal on the Enable Pin Efficiency Aprox. Equals 86% at VIN = 3 V, ILED = 15 mA
D1
L1
MBRM120L
10 µH
†
C2
D2
24 V
(Optional)
V
CC
= 2.7 V to 6 V
100 nF
V
SW
FB
IN
C1
4.7 µF
R1
120 kW
EN
GND
R
S
110 W
L1:
D1:
C1:
C2:
Murata LQH4C3R3M04
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Standard Ceramic Capacitor
Analog Brightness Control
3.3 V Led Off
R2 160 kW
0 V Iled = 20 mA
A. A smaller output capacitor value for C2 causes a larger LED ripple.
Figure 23. White LED Supply With Adjustable Brightness Control
Using an Analog Signal on the Feedback Pin
16
IMPORTANT NOTICE
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enhancements, improvements, and other changes to its products and services at any time and to discontinue
any product or service without notice. Customers should obtain the latest relevant information before placing
orders and should verify that such information is current and complete. All products are sold subject to TI’s terms
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TI warrants performance of its hardware products to the specifications applicable at the time of sale in
accordance with TI’s standard warranty. Testing and other quality control techniques are used to the extent TI
deems necessary to support this warranty. Except where mandated by government requirements, testing of all
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TI assumes no liability for applications assistance or customer product design. Customers are responsible for
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amplifier.ti.com
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dsp.ti.com
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Digital Control
Military
www.ti.com/broadband
www.ti.com/digitalcontrol
www.ti.com/military
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Logic
interface.ti.com
logic.ti.com
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power.ti.com
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microcontroller.ti.com
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Wireless
www.ti.com/wireless
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Copyright 2005, Texas Instruments Incorporated
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