TPS62615YFDT [TI]
350-mA, 6-MHz HIGH-EFFICIENCY STEP-DOWN CONVERTER IN LOW PROFILE CHIP SCALE PACKAGING (HEIGHT < 0.4mm); 350毫安， 6 MHz的高效率降压转换器低姿态芯片级封装（高度\u003c 0.4mm间距）
| 型号： | TPS62615YFDT |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | 350-mA, 6-MHz HIGH-EFFICIENCY STEP-DOWN CONVERTER IN LOW PROFILE CHIP SCALE PACKAGING (HEIGHT < 0.4mm) |
| 文件： | 总27页 (文件大小：749K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
CSP-6
TPS62619, TPS62612
TPS62615, TPS62616
www.ti.com
SLVS936A –NOVEMBER 2009–REVISED MARCH 2010
350-mA, 6-MHz HIGH-EFFICIENCY STEP-DOWN CONVERTER
IN LOW PROFILE CHIP SCALE PACKAGING (HEIGHT < 0.4mm)
Check for Samples: TPS62619, TPS62612, TPS62615, TPS62616
1
FEATURES
23
•
90% Efficiency at 6MHz Operation
DESCRIPTION
•
31mA Quiescent Current
The TPS6261x device is
a
high-frequency
•
•
•
•
•
•
•
•
•
Wide VIN Range From 2.3V to 5.5V
6MHz Regulated Frequency Operation
Best in Class Load and Line Transient
±2% Total DC Voltage Accuracy
Automatic PFM/PWM Mode Switching
Low Ripple Light-Load PFM Mode
>50dB VIN PSRR (1kHz to 10kHz)
Internal Soft-Start, 100-ms Start-Up Time
synchronous step-down dc-dc converter optimized for
battery-powered portable applications. Intended for
low-power applications, the TPS6261x supports up to
350-mA load current, and allows the use of low cost
chip inductor and capacitors.
With a wide input voltage range of 2.3V to 5.5V, the
device supports applications powered by Li-Ion
batteries with extended voltage range. Different fixed
voltage output versions are available from 1.2V to
2.3V.
Integrated Active Power-Down Sequencing
(Optional)
The TPS6261x operates at a regulated 6-MHz
switching frequency and enters the power-save mode
operation at light load currents to maintain high
efficiency over the entire load current range.
•
•
Three Surface-Mount External Components
Required (One 0603 MLCC Inductor, Two 0402
Ceramic Capacitors)
The PFM mode extends the battery life by reducing
the quiescent current to 31mA (typ) during light load
Complete Sub 0.4-mm Component Profile
Solution
Total Solution Size <10 mm2
operation.
For
low-frequency
noise-sensitive
•
•
applications, the device can be forced into fixed
frequency PWM mode by pulling the MODE pin high.
These features, combined with high PSRR and AC
load regulation performance, make this device
suitable to replace a linear regulator to obtain better
power conversion efficiency.
Available in a 6-Pin NanoFree™ (CSP)
Ultra-Thin Packaging
APPLICATIONS
•
•
•
•
Cell Phones, Smart-Phones
WLAN, GPS and Bluetooth™ Applications
DTV Tuner Applications
DC/DC Micro Modules
l
l
The TPS6261x is available in an 6-pin thin chip-scale
package (CSP, 0.4mm max. height).
100
90
200
180
V
V
= 3.6 V,
= 1.8 V
I
O
80
70
60
50
40
30
20
160
140
12
100
80
Efficiency
PFM/PWM Operation
V
V
OUT
1.8 V @ 350mA
TPS6261x
VIN
BAT
2.9 V .. 5.5 V
L
SW
0.47 mH
Power Loss
PFM/PWM Operation
60
C
I
FB
EN
C
O
40
2.2 mF
4.7 mF
10
0
20
0
MODE
GND
0.1
1
10 100
- Load Current - mA
1000
I
O
Figure 2. Efficiency vs. Load Current
Figure 1. Smallest Solution Size Application
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2
3
NanoFree is a trademark of Texas Instruments.
Bluetooth is a trademark of Bluetooth SIG, Inc.
UNLESS OTHERWISE NOTED this document contains
PRODUCTION DATA information current as of publication date.
Products conform to specifications per the terms of Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2009–2010, Texas Instruments Incorporated
TPS62619, TPS62612
TPS62615, TPS62616
SLVS936A –NOVEMBER 2009–REVISED MARCH 2010
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
ORDERING INFORMATION(1)
PACKAGE
MARKING
CHIP CODE
PART
NUMBER
OUTPUT
VOLTAGE
DEVICE
SPECIFIC FEATURE
(3)
TA
ORDERING(2)
TPS62619
TPS62612
TPS62615
TPS62616
TPS62617(4)
1.8V
1.5V
1.2V
2.15V
1.3V
TPS62619YFD
TPS62612YFD
TPS62615YFD
TPS62616YFD
TPS62617YFD
GD
NA
NC
QD
-40°C to 85°C
(1) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
website at www.ti.com.
(2) The YFD package is available in tape and reel. Add a R suffix (e.g. TPS62619YFDR) to order quantities of 3000 parts. Add a T suffix
(e.g. TPS62619YFDT) to order quantities of 250 parts.
(3) Internal tap points are available to facilitate output voltages in 25mV increments.
(4) Product preview.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted)(1)
UNIT
Voltage at VIN, SW(2)
Voltage at FB(2)
–0.3 V to 7 V
–0.3 V to 3.6 V
–0.3 V to VI + 0.3 V
350 mA
VI
IO
(2)
Voltage at EN, MODE
Peak output current
Power dissipation
Operating temperature range(3)
Maximum operating junction temperature
Storage temperature range
Human body model
Internally limited
–40°C to 85°C
150°C
TA
TJ (max)
Tstg
–65°C to 150°C
2 kV
(4)
ESD rating
Charge device model
1 kV
Machine model
200 V
(1) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
(2) All voltage values are with respect to network ground terminal.
(3) In applications where high power dissipation and/or poor package thermal resistance is present, the maximum ambient temperature may
have to be derated. Maximum ambient temperature (TA(max)) is dependent on the maximum operating junction temperature (TJ(max)), the
maximum power dissipation of the device in the application (PD(max)), and the junction-to-ambient thermal resistance of the part/package
in the application (qJA), as given by the following equation: TA(max)= TJ(max)–(qJA X PD(max)). To achieve optimum performance, it is
recommended to operate the device with a maximum junction temperature of 105°C.
(4) The human body model is a 100-pF capacitor discharged through a 1.5-kΩ resistor into each pin. The machine model is a 200-pF
capacitor discharged directly into each pin.
DISSIPATION RATINGS(1)
POWER RATING
A ≤ 25°C
DERATING FACTOR
ABOVE TA = 25°C
(2)
(2)
PACKAGE
RqJA
RqJB
T
YFD-6
125°C/W
53°C/W
800mW
8mW/°C
(1) Maximum power dissipation is a function of TJ(max), qJA and TA. The maximum allowable power dissipation at any allowable ambient
temperature is PD = [TJ(max)–TA] / qJA
(2) This thermal data is measured with high-K board (4 layers board according to JESD51-7 JEDEC standard).
.
2
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Copyright © 2009–2010, Texas Instruments Incorporated
Product Folder Link(s): TPS62619 TPS62612 TPS62615 TPS62616
TPS62619, TPS62612
TPS62615, TPS62616
www.ti.com
SLVS936A –NOVEMBER 2009–REVISED MARCH 2010
ELECTRICAL CHARACTERISTICS
Minimum and maximum values are at VI = 2.3V to 5.5V, VO = 1.8V, EN = 1.8V, AUTO mode and TA = –40°C to 85°C; Circuit
of Parameter Measurement Information section (unless otherwise noted). Typical values are at VI = 3.6V, VO = 1.8V, EN =
1.8V, AUTO mode and TA = 25°C (unless otherwise noted).
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
SUPPLY CURRENT
VI
IQ
Input voltage range
2.3
5.5
55
V
mA
mA
mA
V
IO = 0mA. Device not switching
31
6.6
Operating quiescent current
IO = 0mA, PWM mode
EN = GND
I(SD)
Shutdown current
0.2
1
UVLO
Undervoltage lockout threshold
2.05
2.1
ENABLE, MODE
VIH
VIL
Ilkg
High-level input voltage
1.0
V
V
Low-level input voltage
Input leakage current
0.4
1
Input connected to GND or VIN
0.01
mA
POWER SWITCH
TPS62612
TPS62615
TPS62617
TPS62619
VI = V(GS) = 3.6V. PWM mode
VI = V(GS) = 2.5V. PWM mode
480
640
mΩ
mΩ
P-channel MOSFET
on resistance
rDS(on)
VI = V(GS) = 3.6V. PWM mode
VI = V(GS) = 2.5V. PWM mode
V(DS) = 5.5V, -40°C ≤ TJ ≤ 85°C
VI = V(GS) = 3.6V. PWM mode
VI = V(GS) = 2.5V. PWM mode
V(DS) = 5.5V, -40°C ≤ TJ ≤ 85°C
270
350
mΩ
mΩ
mA
TPS62616
Ilkg
P-channel leakage current, PMOS
1
140
200
mΩ
mΩ
mA
N-channel MOSFET
TPS6261x
rDS(on)
on resistance
Ilkg
N-channel leakage current, NMOS
1
50
Discharge resistor for power-down
sequence
rDIS
15
Ω
P-MOS current limit
2.3V ≤ VI ≤ 4.8V. Open loop
850
5.4
1100
140
10
1200
mA
°C
Thermal shutdown
Thermal shutdown hysteresis
°C
OSCILLATOR
fSW
Oscillator frequency
TPS6261x
IO = 0mA. PWM mode
6
6.6
MHz
OUTPUT
2.3V ≤ VI ≤ 2.5V, 0mA ≤ IO ≤ 200 mA
2.5V ≤ VI ≤ 2.9V, 0mA ≤ IO ≤ 300 mA
2.9V ≤ VI ≤ 4.8V, 0mA ≤ IO ≤ 350 mA
PFM/PWM operation
0.98×VNOM
0.98×VNOM
0.98×VNOM
VNOM
VNOM
VNOM
1.03×VNOM
1.04×VNOM
1.02×VNOM
V
V
V
Regulated DC
output voltage
2.9V ≤ VI ≤ 5.5V, 0mA ≤ IO ≤ 350 mA
PFM/PWM operation
TPS62612
TPS62615
TPS62617
TPS62619
V(OUT)
2.3V ≤ VI ≤ 2.5V, 0mA ≤ IO ≤ 200 mA
2.5V ≤ VI ≤ 2.9V, 0mA ≤ IO ≤ 300 mA
2.9V ≤ VI ≤ 4.8V, 0mA ≤ IO ≤ 350 mA
PWM operation
Line regulation
Load regulation
VI = VO + 0.5V (min 2.3V) to 5.5V, IO = 200 mA
IO = 0mA to 350 mA
0.13
–0.0002
480
%/V
%/mA
kΩ
Feedback input resistance
TPS62619
IO = 1mA
IO = 1mA
18
mVPP
mVPP
Power-save mode
ripple voltage
ΔVO
TPS62615
TPS62619
TPS62619
22
Power Supply
Rejection Ratio
PSRR
f = 10kHz, IO = 150mA. PWM mode
IO = 0mA, Time from active EN to VO
50
96
dB
Start-up time
ms
Copyright © 2009–2010, Texas Instruments Incorporated
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TPS62619, TPS62612
TPS62615, TPS62616
SLVS936A –NOVEMBER 2009–REVISED MARCH 2010
www.ti.com
PIN ASSIGNMENTS
TPS6261x
CSP-6
(TOP VIEW)
TPS6261x
CSP-6
(BOTTOM VIEW)
VIN
EN
A2
B2
C2
A1
B1
C1
A1
A2
B2
VIN
EN
MODE
SW
MODE
B1
C1
SW
FB
GND
C2 GND
FB
TERMINAL FUNCTIONS
TERMINAL
NO.
I/O
DESCRIPTION
NAME
FB
C1
A2
I
I
Output feedback sense input. Connect FB to the converter’s output.
Power supply input.
VIN
This is the switch pin of the converter and is connected to the drain of the internal Power
MOSFETs.
SW
B1
I/O
This is the enable pin of the device. Connecting this pin to ground forces the device into
shutdown mode. Pulling this pin to VI enables the device. This pin must not be left floating and
must be terminated.
EN
B2
I
This is the mode selection pin of the device. This pin must not be left floating and must be
terminated.
MODE = LOW: The device is operating in regulated frequency pulse width modulation mode
(PWM) at high-load currents and in pulse frequency modulation mode (PFM) at light load
currents.
MODE
GND
A1
C2
I
MODE = HIGH: Low-noise mode enabled, regulated frequency PWM operation forced.
Ground pin.
–
FUNCTIONAL BLOCK DIAGRAM
MODE
EN
VIN
Undervoltage
Lockout
Bias Supply
VIN
Soft-Start
Negative Inductor
Current Detect
Bandgap
V
= 0.8 V
Power Save Mode
Switching Logic
REF
Current Limit
Detect
Thermal
Shutdown
Frequency
Control
R
1
FB
-
Gate Driver
SW
Anti
Shoot-Through
R
V
2
REF
+
GND
4
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Copyright © 2009–2010, Texas Instruments Incorporated
Product Folder Link(s): TPS62619 TPS62612 TPS62615 TPS62616
TPS62619, TPS62612
TPS62615, TPS62616
www.ti.com
SLVS936A –NOVEMBER 2009–REVISED MARCH 2010
PARAMETER MEASUREMENT INFORMATION
TPS6261x
L
SW
VIN
V
O
FB
EN
V
C
I
I
C
O
MODE
GND
List of components:
•
•
•
L = MURATA LQM21PN1R0MC0
CI = MURATA GRM155R60J475M (4.7mF, 6.3V, 0402, X5R)
CO = MURATA GRM155R60J475M (4.7mF, 6.3V, 0402, X5R)
Copyright © 2009–2010, Texas Instruments Incorporated
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Product Folder Link(s): TPS62619 TPS62612 TPS62615 TPS62616
TPS62619, TPS62612
TPS62615, TPS62616
SLVS936A –NOVEMBER 2009–REVISED MARCH 2010
www.ti.com
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Load current
vs Input voltage
3, 4, 5
6
h
Efficiency
Peak-to-peak output ripple voltage
vs Load current
7, 8
Combined line/load transient
response
9, 10
11, 12, 13, 14,
15, 16, 17
Load transient response
AC load transient response
DC output voltage
18
19, 20
21
VO
vs Load current
vs Input voltage
vs Input voltage
vs Input voltage
vs Load current
vs Input voltage
vs Input voltage
PFM/PWM boundaries
Quiescent current
IQ
fs
22
PWM switching frequency
PFM switching frequency
P-channel MOSFET rDS(on)
N-channel MOSFET rDS(on)
PWM operation
23
24
25
rDS(on)
26
27
Power-save mode operation
Mode change response
Start-up
28
29, 30
31
PSRR
Power supply rejection ratio
Spurious output noise (PFM mode)
Spurious output noise (PWM mode)
vs. Frequency
vs. Frequency
vs. Frequency
32
33
34
EFFICIENCY
vs
EFFICIENCY
vs
LOAD CURRENT
LOAD CURRENT
100
90
100
V = 2.7 V
I
V
= 1.8 V
V
= 1.2 V
O
O
V = 2.7 V
I
PFM/PWM Operation
90
PFM/PWM Operation
80
70
60
50
80
70
60
50
V = 3.6 V
I
V = 3.6 V
PFM/PWM Operation
I
PFM/PWM Operation
V
= 4.2 V
I
PFM/PWM Operation
40
30
20
10
V = 3.6 V
40
30
20
10
V
= 4.2 V
I
I
Forced PWM Operation
PFM/PWM Operation
V = 3.6 V
I
Forced PWM Operation
0
0
0.1
1
10
- Load Current - mA
100
1000
0.1
1
10
- Load Current - mA
100
1000
I
I
O
O
Figure 3.
Figure 4.
6
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Copyright © 2009–2010, Texas Instruments Incorporated
Product Folder Link(s): TPS62619 TPS62612 TPS62615 TPS62616
TPS62619, TPS62612
TPS62615, TPS62616
www.ti.com
SLVS936A –NOVEMBER 2009–REVISED MARCH 2010
TYPICAL CHARACTERISTICS (continued)
EFFICIENCY
vs
EFFICIENCY
vs
LOAD CURRENT
INPUT VOLTAGE
91
90
89
100
98
V
= 1.8 V
O
PFM/PWM Operation
L = muRata LQM21PN1R0NGR
L = muRata LQM21PN1R0MC0
96
94
88
87
86
85
84
83
82
81
80
79
78
77
92
90
88
86
84
82
80
78
I
= 100 mA
O
I
= 10 mA
O
L = muRata LQM18PN1R5-B35
76
74
I = 1 mA
O
V = 3.6 V
I
V
= 1.8 V
O
72
70
PFM/PWM Operation
76
1
100
- Load Current - mA
10
1000
2.3
2.7
3.1
3.5
3.9
4.3
4.7
5.1
5.5
I
V - Input Voltage - V
I
O
Figure 5.
Figure 6.
PEAK-TO-PEAK OUTPUT RIPPLE VOLTAGE
PEAK-TO-PEAK OUTPUT RIPPLE VOLTAGE
vs
vs
LOAD CURRENT
LOAD CURRENT
26
24
22
20
18
16
14
12
10
8
20
18
16
14
12
10
8
V
= 1.8 V
O
V
= 1.8 V
O
L = muRata LQM21PN1R5MC0
V = 4.2 V
I
V = 4.8 V
I
V = 3.6 V
I
V = 3.6 V
I
V = 2.9 V
I
V = 2.5 V
I
6
6
4
4
2
2
0
0
0
0
50
100
I
150
200
250
300
350
50
100
I
150
- Load Current - mA
200
250
300
350
- Load Current - mA
O
O
Figure 7.
Figure 8.
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TPS62619, TPS62612
TPS62615, TPS62616
SLVS936A –NOVEMBER 2009–REVISED MARCH 2010
www.ti.com
TYPICAL CHARACTERISTICS (continued)
COMBINED LINE/LOAD TRANSIENT RESPONSE
COMBINED LINE/LOAD TRANSIENT RESPONSE
30 to 150 mA Load Step
3.3 to 3.9 V Line Step
30 to 150 mA Load Step
2.7 to 3.3 V Line Step
V = 3.6 V,
V = 3.6 V,
I
I
V
= 1.8 V
V
= 1.8 V
MODE = Low
MODE = Low
O
O
t - Time - 10 ms/div
t - Time - 10 µs/div
Figure 9.
Figure 10.
LOAD TRANSIENT RESPONSE IN
PFM/PWM OPERATION
LOAD TRANSIENT RESPONSE IN
PFM/PWM OPERATION
30 to 150 mA Load Step
0 to 50 mA Load Step
V = 3.6 V,
V = 3.6 V,
I
I
MODE = Low
V
= 1.8 V
V
= 1.8 V
O
O
MODE = Low
t - Time - 5 µs/div
t - Time - 2 ms/div
Figure 11.
Figure 12.
8
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TPS62615, TPS62616
www.ti.com
SLVS936A –NOVEMBER 2009–REVISED MARCH 2010
TYPICAL CHARACTERISTICS (continued)
LOAD TRANSIENT RESPONSE IN
PFM/PWM OPERATION
LOAD TRANSIENT RESPONSE IN
PFM/PWM OPERATION
30 to 150 mA Load Step
30 to 150 mA Load Step
V = 2.7 V,
I
V = 4.8 V,
I
V
= 1.8 V
MODE = Low
t - Time - 5 µs/div
Figure 13.
O
MODE = Low
V
= 1.8 V
O
t - Time - 5 µs/div
Figure 14.
LOAD TRANSIENT RESPONSE IN
PFM/PWM OPERATION
LOAD TRANSIENT RESPONSE IN
PFM/PWM OPERATION
50 to 350 mA Load Step
50 to 350 mA Load Step
V = 3.6 V,
I
V
V
= 2.9 V,
= 1.8 V
I
V
= 1.8 V
O
O
MODE = Low
MODE = Low
t - Time - 5 µs/div
t - Time - 5 µs/div
Figure 15.
Figure 16.
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TPS62619, TPS62612
TPS62615, TPS62616
SLVS936A –NOVEMBER 2009–REVISED MARCH 2010
www.ti.com
TYPICAL CHARACTERISTICS (continued)
LOAD TRANSIENT RESPONSE
IN PFM/PWM OPERATION
AC LOAD TRANSIENT RESPONSE
V = 3.6 V,
I
V
= 1.8 V
O
50 to 350 mA Load Step
5 to 300 mA Load Sweep
V = 4.8 V,
I
V
= 1.8 V
O
MODE = Low
MODE = Low
t - Time - 10 µs/div
t - Time - 5 µs/div
Figure 17.
Figure 18.
DC OUTPUT VOLTAGE
vs
DC OUTPUT VOLTAGE
vs
LOAD CURRENT
LOAD CURRENT
1.224
1.212
1.2
1.836
1.818
V
= 1.8 V
V
= 1.2 V
O
O
PFM/PWM Operation
V = 3.6 V
V = 4.8 V
I
I
V = 3.6 V
PFM/PWM Operation
I
V = 4.8 V
I
V = 2.5 V
I
1.8
1.782
1.764
V = 3.6 V, PWM Operation
I
V = 3.6 V, PWM Operation
I
V = 2.5 V
I
1.188
1.176
0.1
1
10
- Load Current - mA
100
1000
0.1
1
10
- Load Current - mA
100
1000
I
O
I
O
Figure 19.
Figure 20.
10
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TPS62615, TPS62616
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SLVS936A –NOVEMBER 2009–REVISED MARCH 2010
TYPICAL CHARACTERISTICS (continued)
QUIESCENT CURRENT
vs
PFM/PWM BOUNDARIES
INPUT VOLTAGE
200
180
160
140
50
45
40
35
V
= 1.8 V
Always PWM
O
T
= 85°C
A
T
= 25°C
A
PFM to PWM
Mode Change
120
100
80
30
25
The Switching Mode
Changes at
These Borders
T
= -40°C
20
15
10
A
60
40
PWM to PFM
Mode Change
Always PFM
20
0
5
0
2.5 2.7 2.9 3.1 3.3 3.5 3.7 3.9 4.1 4.3 4.5 4.74.8
2.5 2.8 3.1 3.4 3.7 4.0 4.3 4.6 4.9 5.2 5.5
V - Input Voltage - V
I
V - Input Voltage - V
I
Figure 21.
Figure 22.
PWM SWITCHING FREQUENCY
PFM SWITCHING FREQUENCY
vs
vs
INPUT VOLTAGE
LOAD CURRENT
6.5
6
6.5
6
V
= 1.8 V
I
= 10 mA
O
PFM/PWM Operation
O
5.5
I
= 300 mA
5.5
5
O
5
4.5
4
I
= 200 mA
O
V = 2.5 V
I
I
= 150 mA
O
I
= 100 mA
O
4.5
4
V = 3.6 V
V = 4.8 V
I
I
= 50 mA
I
O
3.5
3
3.5
3
2.5
2
1.5
1
2.5
V
= 1.8V
O
2
0.5
0
1.5
0
20 40
60 80 100 120 140 160 180 200
2.5 2.7 2.9 3.1 3.3 3.5 3.7 3.9 4.1 4.3 4.5 4.7 4.9 5.1 5.3 5.5
V - Input Voltage - V
I
I
- Load Current - mA
O
Figure 23.
Figure 24.
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TYPICAL CHARACTERISTICS (continued)
P-CHANNEL rDS(ON)
vs
N-CHANNEL rDS(ON)
vs
INPUT VOLTAGE
INPUT VOLTAGE
300
275
250
225
200
175
900
850
800
750
700
TPS62619
PWM Mode Operation
TPS62619
PWM Mode Operation
T
= 85°C
A
650
600
550
500
T
= 25°C
A
T = 85°C
A
T
= -40°C
A
T
= 25°C
A
T
A
= -40°C
450
400
350
300
250
200
150
100
150
125
100
75
50
2.5 2.7 2.9 3.1 3.3 3.5 3.7 3.9 4.1 4.3 4.5 4.7 4.9 5.1 5.3 5.5
2.5 2.7 2.9 3.1 3.3 3.5 3.7 3.9 4.1 4.3 4.5 4.7 4.9 5.1 5.3 5.5
V - Input Voltage - V
I
V - Input Voltage - V
I
Figure 25.
Figure 26.
PWM OPERATION
POWER-SAVE MODE OPERATION
V = 3.6 V, V = 1.8 V, I = 150 mA
I
O
O
V = 3.6 V, V = 1.8 V,
I
O
I
= 40 mA
O
MODE = High
MODE = Low
t - Time - 250 ns/div
t - Time - 50 ns/div
Figure 27.
Figure 28.
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TYPICAL CHARACTERISTICS (continued)
MODE CHANGE RESPONSE
MODE CHANGE RESPONSE
V = 3.6 V, V = 1.8 V,
I
O
I
= 40 mA
V = 3.6 V,
O
I
V
I
= 1.8 V,
O
= 40 mA
O
t - Time - 1 µs/div
t - Time - 1 µs/div
Figure 29.
Figure 30.
POWER SUPPLY REJECTION RATIO
vs
START-UP
FREQUENCY
60
V = 3.6 V,
I
55
50
V = 3.6 V,
V
= 1.8 V
I
O
V
I
= 1.8 V,
O
= 0 mA
O
45
40
I
= 150 mA
O
35
30
25
20
15
10
PFM/PWM Operation
MODE = Low
5
0
t - Time - 20 µs/div
1000
0.1
1
10
100
10000
f - Frequency - kHz
Figure 31.
Figure 32.
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TYPICAL CHARACTERISTICS (continued)
SPURIOUS OUTPUT NOISE (PFM MODE)
SPURIOUS OUTPUT NOISE (PWM MODE)
vs
vs
FREQUENCY
FREQUENCY
1 m
500 m
450 m
V = 2.7 V
I
V
= 1.8 V
= 12 Ω
O
V
= 1.8 V
900 m
O
R
V
= 3.6 V
L
I
R
= 100 Ω
L
800 m
700 m
600 m
500 m
400 m
400 m
350 m
V = 4.2 V
I
V
I
= 2.7 V
= 3.6 V
300 m
250 m
V
I
200 m
150 m
100 m
300 m
200 m
100 m
10 n
V
I
= 4.2 V
50 m
5 n
0
Span = 1 MHz
f - Frequency - MHz
10
0
Span = 4 MHz
40
f - Frequency - MHz
Figure 33.
Figure 34.
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DETAILED DESCRIPTION
OPERATION
The TPS6261x is a synchronous step-down converter typically operates at a regulated 6-MHz frequency pulse
width modulation (PWM) at moderate to heavy load currents. At light load currents, the TPS6261x converter
operates in power-save mode with pulse frequency modulation (PFM).
The converter uses a unique frequency locked ring oscillating modulator to achieve best-in-class load and line
response and allows the use of tiny inductors and small ceramic input and output capacitors. At the beginning of
each switching cycle, the P-channel MOSFET switch is turned on and the inductor current ramps up rising the
output voltage until the main comparator trips, then the control logic turns off the switch.
One key advantage of the non-linear architecture is that there is no traditional feed-back loop. The loop response
to change in VO is essentially instantaneous, which explains the transient response. The absence of a traditional,
high-gain compensated linear loop means that the TPS6261x is inherently stable over a range of L and CO.
Although this type of operation normally results in a switching frequency that varies with input voltage and load
current, an internal frequency lock loop (FLL) holds the switching frequency constant over a large range of
operating conditions.
Combined with best in class load and line transient response characteristics, the low quiescent current of the
device (ca. 31mA) allows to maintain high efficiency at light load, while preserving fast transient response for
applications requiring tight output regulation.
Using the YFD package allows for a low profile solution size (0.4mm max height, including external components).
The recommended external components are stated within the application information. The maximum output
current is 350mA when these specific low profile external components are used.
SWITCHING FREQUENCY
The magnitude of the internal ramp, which is generated from the duty cycle, reduces for duty cycles either set of
50%. Thus, there is less overdrive on the main comparator inputs which tends to slow the conversion down. The
intrinsic maximum operating frequency of the converter is about 10MHz to 12MHz, which is controlled to circa.
6MHz by a frequency locked loop.
When high or low duty cycles are encountered, the loop runs out of range and the conversion frequency falls
below 6MHz. The tendency is for the converter to operate more towards a "constant inductor peak current" rather
than a "constant frequency". In addition to this behavior which is observed at high duty cycles, it is also noted at
low duty cycles.
When the converter is required to operate towards the 6MHz nominal at extreme duty cycles, the application can
be assisted by decreasing the ratio of inductance (L) to the output capacitor's equivalent serial inductance (ESL).
This increases the ESL step seen at the main comparator's feed-back input thus decreasing its propagation
delay, hence increasing the switching frequency.
POWER-SAVE MODE
If the load current decreases, the converter will enter Power Save Mode operation automatically. During
power-save mode the converter operates in discontinuous current (DCM) single-pulse PFM mode, which
produces low output ripple compared with other PFM architectures.
When in power-save mode, the converter resumes its operation when the output voltage trips below the nominal
voltage. It ramps up the output voltage with a minimum of one pulse and goes into power-save mode when the
inductor current has returned to a zero steady state. The PFM on-time varies inversely proportional to the input
voltage and proportional to the output voltage giving the regulated switching frequency when in steady-state.
PFM mode is left and PWM operation is entered as the output current can no longer be supported in PFM mode.
As a consequence, the DC output voltage is typically positioned ca. 0.5% above the nominal output voltage and
the transition between PFM and PWM is seamless.
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PFM Mode at Light Load
PFM Ripple
Nominal DC Output Voltage
PWM Mode at Heavy Load
Figure 35. Operation in PFM Mode and Transfer to PWM Mode
MODE SELECTION
The MODE pin allows to select the operating mode of the device. Connecting this pin to GND enables the
automatic PWM and power-save mode operation. The converter operates in regulated frequency PWM mode at
moderate to heavy loads and in the PFM mode during light loads, which maintains high efficiency over a wide
load current range.
Pulling the MODE pin high forces the converter to operate in the PWM mode even at light load currents. The
advantage is that the converter operates with a fixed frequency that allows simple filtering of the switching
frequency for low frequency noise-sensitive applications. In this mode, the efficiency is lower compared to the
power-save mode during light loads.
For additional flexibility, it is possible to switch from power-save mode to forced PWM mode during operation.
This allows efficient power management by adjusting the operation of the converter to the specific system
requirements.
ENABLE
The device starts operation when EN is set high and starts up with the soft start as previously described. For
proper operation, the EN pin must be terminated and must not be left floating.
Pulling the EN pin low forces the device into shutdown, with a shutdown quiescent current of typically 0.1mA. In
this mode, the P and N-channel MOSFETs are turned off, the internal resistor feedback divider is disconnected,
and the entire internal-control circuitry is switched off.
SOFT START
The TPS6261x has an internal soft-start circuit that limits the inrush current during start-up. This limits input
voltage drops when a battery or a high-impedance power source is connected to the input of the converter.
The soft-start system progressively increases the on-time from a minimum pulse-width of 35ns as a function of
the output voltage. This mode of operation continues for c.a. 100ms after enable. Should the output voltage not
have reached its target value by this time, such as in the case of heavy load, the soft-start transitions to a second
mode of operation.
The converter then operates in a current limit mode, specifically the P-MOS current limit is set to half the nominal
limit, and the N-channel MOSFET remains on until the inductor current has reset. After a further 100 ms, the
device ramps up to the full current limit operation if the output voltage has risen above 0.5V (approximately).
Therefore, the start-up time mainly depends on the output capacitor and load current.
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OUTPUT CAPACITOR DISCHARGE
The TPS6261x device can actively discharge the output capacitor when it turns off. The integrated discharge
resistor has a typical resistance of 15 Ω. The required time to discharge the output capacitor at the output node
depends on load current and the output capacitance value.
UNDERVOLTAGE LOCKOUT
The undervoltage lockout circuit prevents the device from misoperation at low input voltages. It prevents the
converter from turning on the switch or rectifier MOSFET under undefined conditions. The TPS6261x device
have a UVLO threshold set to 2.05V (typical). Fully functional operation is permitted down to 2.1V input voltage.
SHORT-CIRCUIT PROTECTION
The TPS6261x integrates a P-channel MOSFET current limit to protect the device against heavy load or short
circuits. When the current in the P-channel MOSFET reaches its current limit, the P-channel MOSFET is turned
off and the N-channel MOSFET is turned on. The regulator continues to limit the current on a cycle-by-cycle
basis.
As soon as the output voltage falls below ca. 0.4V, the converter current limit is reduced to half of the nominal
value. Because the short-circuit protection is enabled during start-up, the device does not deliver more than half
of its nominal current limit until the output voltage exceeds approximately 0.5V. This needs to be considered
when a load acting as a current sink is connected to the output of the converter.
THERMAL SHUTDOWN
As soon as the junction temperature, TJ, exceeds typically 140°C, the device goes into thermal shutdown. In this
mode, the P- and N-channel MOSFETs are turned off. The device continues its operation when the junction
temperature again falls below typically 130°C.
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APPLICATION INFORMATION
INDUCTOR SELECTION
The TPS6261x series of step-down converters have been optimized to operate with an effective inductance
value in the range of 0.3mH to 1.8mH and with output capacitors in the range of 4.7mF to 10mF. The internal
compensation is optimized to operate with an output filter of L = 0.47mH and CO = 4.7mF. Larger or smaller
inductor values can be used to optimize the performance of the device for specific operation conditions. For more
details, see the CHECKING LOOP STABILITY section.
The inductor value affects its peak-to-peak ripple current, the PWM-to-PFM transition point, the output voltage
ripple and the efficiency. The selected inductor has to be rated for its dc resistance and saturation current. The
inductor ripple current (ΔIL) decreases with higher inductance and increases with higher VI or VO.
V
V * V
DI
O
I
O
L
DI +
DI
+ I
)
L
L(MAX)
O(MAX)
2
V
L ƒ
sw
I
with: fSW = switching frequency (6 MHz typical)
L = inductor value
ΔIL = peak-to-peak inductor ripple current
IL(MAX) = maximum inductor current
(1)
In high-frequency converter applications, the efficiency is essentially affected by the inductor AC resistance (i.e.
quality factor) and to a smaller extent by the inductor DCR value. To achieve high efficiency operation, care
should be taken in selecting inductors featuring a quality factor above 25 at the switching frequency. Increasing
the inductor value produces lower RMS currents, but degrades transient response. For a given physical inductor
size, increased inductance usually results in an inductor with lower saturation current.
The total losses of the coil consist of both the losses in the DC resistance (DC)) and the following
frequency-dependent components:
•
•
•
•
The losses in the core material (magnetic hysteresis loss, especially at high switching frequencies)
Additional losses in the conductor from the skin effect (current displacement at high frequencies)
Magnetic field losses of the neighboring windings (proximity effect)
Radiation losses
The following inductor series from different suppliers have been used with the TPS6261x converters.
Table 1. List of Inductors
MANUFACTURER
SERIES
DIMENSIONS (in mm)
2.0 x 1.2 x 0.55 max. height
2.0 x 1.2 x 0.55 max. height
2.0 x 1.2 x 0.55 max. height
2.0 x 1.2 x 0.4 max. height
1.6 x 0.8 x 0.55 max. height
1.6 x 0.8 x 0.4 max. height
LQM21PNR47MC0
LQM21PN1R0MC0
LQM21PN1R5MC0
LQM21P-SAMPLE02
MURATA
LQM18PN1R5-B35-
SAMPLE01
1.6 x 0.8 x 0.33 max. height
1.6 x 0.8 x 0.9 max. height
1.6 x 0.8 x 0.9 max. height
2.0 x 1.2 x 0.55 max. height
BRC1608T1R0
BRC1608T1R5
MLP2012SR82T
TAIYO YUDEN
TDK
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OUTPUT CAPACITOR SELECTION
The advanced fast-response voltage mode control scheme of the TPS6261x allows the use of tiny ceramic
capacitors. Ceramic capacitors with low ESR values have the lowest output voltage ripple and are
recommended. For best performance, the device should be operated with a minimum effective output
capacitance of 1.6mF. The output capacitor requires either an X7R or X5R dielectric. Y5V and Z5U dielectric
capacitors, aside from their wide variation in capacitance over temperature, become resistive at high frequencies.
At nominal load current, the device operates in PWM mode and the overall output voltage ripple is the sum of the
voltage step caused by the output capacitor ESL and the ripple current flowing through the output capacitor
impedance.
At light loads, the output capacitor limits the output ripple voltage and provides holdup during large load
transitions. A 4.7mF capacitor typically provides sufficient bulk capacitance to stabilize the output during large
load transitions. The typical output voltage ripple is 1% of the nominal output voltage VO.
The output voltage ripple during PFM mode operation can be kept very small. The PFM pulse is time controlled,
which allows to modify the charge transferred to the output capacitor by the value of the inductor. The resulting
PFM output voltage ripple and PFM frequency depend in first order on the size of the output capacitor and the
inductor value. The PFM frequency decreases with smaller inductor values and increases with larger once.
Increasing the output capacitor value and the effective inductance will minimize the output ripple voltage.
INPUT CAPACITOR SELECTION
Because of the nature of the buck converter having a pulsating input current, a low ESR input capacitor is
required to prevent large voltage transients that can cause misbehavior of the device or interferences with other
circuits in the system. For most applications, a 2.2-mF capacitor is sufficient. If the application exhibits a noisy or
erratic switching frequency, the remedy will probably be found by experimenting with the value of the input
capacitor.
Take care when using only ceramic input capacitors. When a ceramic capacitor is used at the input and the
power is being supplied through long wires, such as from a wall adapter, a load step at the output can induce
ringing at the VIN pin. This ringing can couple to the output and be mistaken as loop instability or could even
damage the part. Additional "bulk" capacitance (electrolytic or tantalum) should in this circumstance be placed
between CI and the power source lead to reduce ringing than can occur between the inductance of the power
source leads and CI.
CHECKING LOOP STABILITY
The first step of circuit and stability evaluation is to look from a steady-state perspective at the following signals:
•
•
•
Switching node, SW
Inductor current, IL
Output ripple voltage, VO(AC)
These are the basic signals that need to be measured when evaluating a switching converter. When the
switching waveform shows large duty cycle jitter or the output voltage or inductor current shows oscillations, the
regulation loop may be unstable. This is often a result of board layout and/or L-C combination.
As a next step in the evaluation of the regulation loop, the load transient response is tested. The time between
the application of the load transient and the turn on of the P-channel MOSFET, the output capacitor must supply
all of the current required by the load. VO immediately shifts by an amount equal to ΔI(LOAD) x ESR, where ESR
is the effective series resistance of CO. ΔI(LOAD) begins to charge or discharge CO generating a feedback error
signal used by the regulator to return VO to its steady-state value. The results are most easily interpreted when
the device operates in PWM mode.
During this recovery time, VO can be monitored for settling time, overshoot or ringing that helps judge the
converter’s stability. Without any ringing, the loop has usually more than 45° of phase margin.
Because the damping factor of the circuitry is directly related to several resistive parameters (e.g., MOSFET
rDS(on)) that are temperature dependant, the loop stability analysis has to be done over the input voltage range,
load current range, and temperature range.
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LAYOUT CONSIDERATIONS
As for all switching power supplies, the layout is an important step in the design. High-speed operation of the
TPS6261x devices demand careful attention to PCB layout. Care must be taken in board layout to get the
specified performance. If the layout is not carefully done, the regulator could show poor line and/or load
regulation, stability and switching frequency issues as well as EMI problems. It is critical to provide a low
inductance, impedance ground path. Therefore, use wide and short traces for the main current paths.
The input capacitor should be placed as close as possible to the IC pins as well as the inductor and output
capacitor. In order to get an optimum ESL step, the output voltage feedback point (FB) should be taken in the
output capacitor path, approximately 1mm away for it. The feed-back line should be routed away from noisy
components and traces (e.g. SW line).
MODE
VIN
CI
L
ENABLE
CO
GND
VOUT
Figure 36. Suggested Layout (Top)
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SLVS936A –NOVEMBER 2009–REVISED MARCH 2010
APPLICATION CIRCUITS
The following are example circuits.
100Ω
1 mF
V
TPS62616
VIN
IN
2.9 V .. 5.5 V
TPS72018
IN
L
V
= 2.3 V
V
= 1.8 V @ 350 mA
OUT(DC/DC)
OUT(LDO)
BIAS
GND
SW
1 mH
OUT
C
I
FB
EN
C
O
EN
C
O
2.2 mF
4.7 mF
4.7 mF
MODE
GND
L = muRata LQM21PN1R0MC0
CI = muRata GRM155R60J225ME15
CO = muRata GRM155R60J475M
Figure 37. 1.8V Power Rail Featuring Very Low Noise Performance
EFFICIENCY
vs
SPURIOUS OUTPUT NOISE (PFM MODE)
vs.
LOAD CURRENT
FREQUENCY
90
85
80
75
70
65
60
13 m
12 m
11 m
10 m
9 m
V
V
= 2.3 V,
OUT(DC/DC)
V
V
= 3.6 V
IN
OUT(LDO)
= 1.8 V
OUT(LDO)
= 1.8 V
V = 3.6 V
R
= 100 Ω
I
L
V = 3.3 V
I
8 m
7 m
V = 4.2 V
I
6 m
55
50
45
40
35
30
5 m
4 m
3 m
2 m
1 m
L = muRata LQM21PN1R0MC0
1
10
100
1000
0
Span = 1 MHz
f - Frequency - MHz
10
I
- Load Current - mA
O
Figure 38.
Figure 39.
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SPURIOUS OUTPUT NOISE (PWM MODE)
vs.
FREQUENCY
60 m
55 m
50 m
45 m
40 m
V
V
= 3.6 V
IN
OUT(LDO)
= 1.8 V
R
= 12 Ω
L
35 m
30 m
25 m
20 m
15 m
10 m
5 m
0
Span = 4 MHz
f - Frequency - Mhz
40
Figure 40.
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SLVS936A –NOVEMBER 2009–REVISED MARCH 2010
THERMAL INFORMATION
Implementation of integrated circuits in low-profile and fine-pitch surface-mount packages typically requires
special attention to power dissipation. Many system-dependant issues such as thermal coupling, airflow, added
heat sinks, and convection surfaces, and the presence of other heat-generating components, affect the
power-dissipation limits of a given component.
Three basic approaches for enhancing thermal performance are listed below:
•
•
•
Improving the power dissipation capability of the PCB design
Improving the thermal coupling of the component to the PCB
Introducing airflow into the system
The maximum recommended junction temperature (TJ) of the TPS6261x devices is 105°C. The thermal
resistance of the 6-pin CSP package (YFD-6) is RqJA = 125°C/W. Regulator operation is specified to a maximum
steady-state ambient temperature TA of 85°C. Therefore, the maximum power dissipation is about 160 mW.
T
- T
A
105°C - 85°C
125°C/W
J(MAX)
P
=
=
= 160mW
D(MAX)
R
qJA
(2)
PACKAGE SUMMARY
CHIP SCALE PACKAGE
(BOTTOM VIEW)
CHIP SCALE PACKAGE
(TOP VIEW)
A2
B2
C2
A1
B1
C1
YMSCC
LLLL
D
A1
Code:
•
•
•
•
YM — Year Month date Code
S — Assembly site code
CC— Chip code
E
LLLL — Lot trace code
CHIP SCALE PACKAGE DIMENSIONS
The TPS6261x device is available in an 6-bump chip scale package (YFD, NanoFree™). The package
dimensions are given as:
•
•
D = 1.30 ±0.03 mm
E = 0.926 ±0.03 mm
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TPS62619, TPS62612
TPS62615, TPS62616
SLVS936A –NOVEMBER 2009–REVISED MARCH 2010
www.ti.com
REVISION HISTORY
Changes from Original (November 2009) to Revision A
Page
•
Changed device status of TPS62619, TPS62612, and TPS62616 from "Product Preview" to "Production" and added
package marking chip code for TPS62616 in the Ordering Informtion table. ....................................................................... 2
24
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Copyright © 2009–2010, Texas Instruments Incorporated
Product Folder Link(s): TPS62619 TPS62612 TPS62615 TPS62616
PACKAGE OPTION ADDENDUM
www.ti.com
2-Apr-2010
PACKAGING INFORMATION
Orderable Device
TPS62612YFDR
TPS62612YFDT
TPS62615YFDR
TPS62615YFDT
TPS62616YFDR
TPS62616YFDT
TPS62619YFDR
TPS62619YFDT
Status (1)
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
Package Package
Pins Package Eco Plan (2) Lead/Ball Finish MSL Peak Temp (3)
Qty
Type
Drawing
DSBGA
YFD
6
6
6
6
6
6
6
6
3000 Green (RoHS &
no Sb/Br)
SNAGCU
SNAGCU
SNAGCU
SNAGCU
SNAGCU
SNAGCU
SNAGCU
SNAGCU
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
Level-1-260C-UNLIM
DSBGA
DSBGA
DSBGA
DSBGA
DSBGA
DSBGA
DSBGA
YFD
YFD
YFD
YFD
YFD
YFD
YFD
250 Green (RoHS &
no Sb/Br)
3000 Green (RoHS &
no Sb/Br)
250 Green (RoHS &
no Sb/Br)
3000 Green (RoHS &
no Sb/Br)
250 Green (RoHS &
no Sb/Br)
3000 Green (RoHS &
no Sb/Br)
250 Green (RoHS &
no Sb/Br)
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and
package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS
compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
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Addendum-Page 1
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