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UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

UCC2863x 高功率反激式控制器

具有初级侧稳压和峰值功率模式

1 特性

3 说明

1

•

高功率初级侧 CV/CC 稳压

UCC2863x 适用于高功率初级侧稳压反激式变换器。

此器件能够以 CCM 和 DCM 模式运行，适用于具有

宽功率范围的应用。峰值功率模式使得瞬态峰值功率

能够达到标称额定值的 200%，峰值电流只增加

25%，最大限度地增加了变压器利用率。

连续传导模式 (CCM) 和断续传导模式 (DCM) 运行

内置 700V 启动电流源

有源 X 电容器放电（UCC28630 和 UCC28633）

可调节恒定电流 (CC) 模式限制（UCC28630 除

外）

变压器偏置绕组被用来感测输出电压以实现稳压，并用

于低损耗输入电压感测。采用了先进的取样技术，可支

持 CCM 运行，并在 100W 及以上功率范围内实现无

光耦合器设计的出色输出电压稳压性能。

•

高栅极驱动电流，1A 源电流和 2A 灌电流

针对系统待机功率小于 30mW 的低功率模式

业界一流的轻负载 (10%) 效率 >85%

PSR 设计不包含光耦合器 - 可达到 CM 隔离和浪涌

的高要求

高压电流源可实现快速和高效启动。部署了先进的轻负

载模式，可减少控制器和系统在无负载和轻负载状态下

的功率消耗。这些模式支持潜在的系统设计，以满足

30mW 无负载功耗对高达 30W 标称功率和 60W 峰值

功率电源设计的需要。

•

用于开环反馈故障条件下独立间接输出过压的 VDD

OVP

•

针对瞬态过载的峰值功率模式

外部 NTC 的关断引脚接口

保护：过压、过流、过热、过载计时器

(UCC28630)、交流线路 UV、欠压和引脚保护

按照设计，此器件易于使用并整合了多种特性，可实

现多种设计。设计包含大量的保护特性以简化系统设

计。

•

频率抖动以轻松符合 EMI 标准（UCC28632 除

外）

使用 UCC2863x 并借助 WEBENCH^®Power

请参阅表格1《器件比较表》以了解各器件间的具体差

异。

Designer 创建定制设计方案

2 应用

器件信息

器件型号

UCC28630

UCC28631

UCC28632

UCC28633

UCC28634

封装

封装尺寸

•

针对笔记本电脑、游戏机和打印机的交流-直流适配

器

针对工业、打印机、大型家电和 LCD 显示器的开

放式结构开关模式电源 (SMPS)

SOIC (7)

4.90mm x 3.90mm

针对 10W 至 65W 标称功率的高能效交流-直流电

源，（具有高达 200% 的瞬态峰值功率）

简化电路原理图

典型应用测得的稳压

21

V_OUT

V_AC

20.5

20

EMC

Filter

19.5

19

+5% Limit

115V/60 Hz

230V/50 Hz

œ5% Limit

± 1% Typical

18.5

18

1

2

3

4

VSENSE

SD

HV

8

0

10

20

30

40

50

60

70

UCC28630

t^o

CS

VDD

6

5

Output Power (W)

C001

GND

DRV

1

An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,

intellectual property matters and other important disclaimers. PRODUCTION DATA.

English Data Sheet: SLUSBW3

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

www.ti.com.cn

8.4 Device Functional Modes........................................ 52

Applications and Implementation ...................... 53

9.1 Application Information............................................ 53

9.2 Typical Application ................................................. 53

9.3 Dos and Don'ts........................................................ 73

1

2

3

4

5

6

7

特性.......................................................................... 1

应用.......................................................................... 1

说明.......................................................................... 1

修订历史记录 ........................................................... 3

Device Comparison Table..................................... 5

Pin Configuration and Functions......................... 5

Specifications......................................................... 6

7.1 Absolute Maximum Ratings ...................................... 6

7.2 ESD Ratings.............................................................. 6

7.3 Recommended Operating Conditions....................... 6

7.4 Thermal Information.................................................. 7

7.5 Electrical Characteristics........................................... 8

7.6 Typical Characteristics............................................ 10

Detailed Description ............................................ 13

8.1 Overview ................................................................. 13

8.2 Functional Block Diagram ....................................... 14

8.3 Feature Description................................................. 15

9

10 Power Supply Recommendations ..................... 73

11 Layout................................................................... 74

11.1 Layout Guidelines ................................................. 74

11.2 Layout Example .................................................... 75

12 器件和文档支持 ..................................................... 76

12.1 商标....................................................................... 76

12.2 静电放电警告......................................................... 76

12.3 Glossary................................................................ 76

12.4 器件支持................................................................ 76

12.5 文档支持................................................................ 76

13 机械、封装和可订购信息....................................... 77

8

2

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

www.ti.com.cn

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

4 修订历史记录

Changes from Revision C (March 2015) to Revision D

Page

•

已添加 UCC28634 初始发行版。............................................................................................................................................ 1

已删除删除了文本“适用于 65W 标称功率设计”...................................................................................................................... 1

已添加添加了文本“PSR 设计不包含光耦合器 - 可达到 CM 隔离和浪涌的高要求” ................................................................ 1

已添加添加了文本“用于开环反馈故障条件下独立间接输出过压的 VDD OVP” ...................................................................... 1

已添加 Webench 链接。......................................................................................................................................................... 1

已添加添加了文本“请参阅《器件比较表》以了解各器件间的具体差异” ............................................................................... 1

已添加向器件信息表中添加了 UCC28634。.......................................................................................................................... 1

Added UCC28634 to the Device Comparison Table.............................................................................................................. 5

Added UCC28634 to Thermal Information ............................................................................................................................ 7

Added UCC28634 to Electrical Characteristics...................................................................................................................... 8

Changed picture to represent added UCC28634 ................................................................................................................ 10

Added UCC28634 ............................................................................................................................................................... 13

Added UCC28634 ............................................................................................................................................................... 15

Added UCC28634 ............................................................................................................................................................... 16

Changed to correct picture link ............................................................................................................................................ 19

Changed to fix equation typo ............................................................................................................................................... 21

Added UCC28634 ............................................................................................................................................................... 41

Changed to correct typo ...................................................................................................................................................... 41

Changed to correct typo, changed from 4.7 to 47 ............................................................................................................... 41

Added paragraph to clarify the fault protection. .................................................................................................................. 41

Added UCC28634 ............................................................................................................................................................... 42

Added text "For UCC28634, all pin-faults are non-latching." .............................................................................................. 43

Added UCC28634 to the table ............................................................................................................................................ 52

Changed equation to fix typo ............................................................................................................................................... 67

Changes from Revision B (March 2014) to Revision C

Page

•

Changed "No" to "Yes" in Device Comparison Table for Part# UCC28633D, "ACTIVE-X CAPACITOR DISCHARGE"

column .................................................................................................................................................................................... 5

Changed "Handling Ratings" table to "ESD Ratings" table. Moved Storage Temperature and Lead Temperature to

Abs Max Ratings table. .......................................................................................................................................................... 6

Revised Figure 40 ............................................................................................................................................................... 47

Changes from Revision A (January 2014) to Revision B

Page

•

已添加向数据表中添加了 UCC28631、UCC28632 和 UCC28633 器件。............................................................................ 1

已添加添加了 UCC28630 和 UCC28633 的有源 X 电容器放电功能参考。 .......................................................................... 1

已添加添加了可调节恒定电流 (CC) 模式限制项目符号。...................................................................................................... 1

已添加添加了 UCC28630（仅限）的过载计时器参考。........................................................................................................ 1

已添加添加了频率抖动以轻松符合 EMI 标准（UCC28632 除外）。 .................................................................................... 1

已添加向器件信息部分中添加了 UCC28631D、UCC28632D 和 UCC28633D。 ................................................................. 1

Added Device Comparison Table........................................................................................................................................... 5

Added UCC28632 Frequency dither range exception. .......................................................................................................... 8

Added UCC28632 Dither repetition period exception. ........................................................................................................... 8

3

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

www.ti.com.cn

•

Added UCC28633 Wake-up level (rising) exception. ............................................................................................................. 8

Added UCC28633 SD V_WAKE(rise)vs. Temperature exception............................................................................................... 12

Added text to the, "The controller operates in either DCM and CCM..." paragraph. .......................................................... 13

Changed the "Supply the device bias power during latched fault mode" bullet................................................................... 15

Added UCC28630 and UCC28633 only exception to the "AC sense input for X-capacitor discharge detect" bullet. ......... 15

Changed HV Pin Connection diagram. ............................................................................................................................... 15

Added sentence, "In the UCC28631 and the UCC28632, the HV pin can connect to either the AC or DC side of the

bridge.".................................................................................................................................................................................. 16

•

Added VIN(avg) definition. ................................................................................................................................................... 16

Added (UCC28630 and UCC28633 only) to the Active X-Capacitor Discharge section...................................................... 19

Added UCC28633 to the Improved Performance with UCC28630 section.......................................................................... 20

Added UCC28631, UCC28632 and the UCC28633 I_OUT(lim)adjustment note. .................................................................... 37

Added UCC28630 only note to the Primary-Side Overload Timer section. ......................................................................... 38

Added UCC28630 only note added to the Overload Timer Adjustment section. ................................................................ 40

Added CC-Mode I_OUT(lim)Adjustment section. ...................................................................................................................... 41

Added UCC28631, UCC28632 and the UCC28633 to the Fault Sources and Associated Responses table. .................... 42

Added The fault response (latching or auto recovery) depends on the device variant, per Table 4. ................................. 44

Added The fault response (latching or recovery) depends on the device variant, per Table 4. ......................................... 44

Added UCC28633 exception to the External SD Pin Wake Input section. .......................................................................... 45

Added External Wake Input at VSENSE Pin (UCC28633 Only) section.............................................................................. 46

Added UCC28632 exception to the Frequency Dither For EMI section............................................................................... 51

Added External Wake Pulse Calculation at VSENSE Pin (UCC28633 Only) section.......................................................... 66

Changes from Original (January 2014) to Revision A

Page

•

已更改将销售状态从产品预览改为量产数据。....................................................................................................................... 1

4

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

www.ti.com.cn

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

5 Device Comparison Table

Table 1. Device Comparison Table

FEATURES

ACTIVE-X

CAPACITOR

DISCHARGE

ORDER NUMBER

OVERLOAD

TIMER

ADJUSTABLE

CC LIMIT

FREQUENCY

DITHER

SECONDARY-

SIDE WAKE UP

UCC28630D

UCC28631D

UCC28632D

UCC28633D

UCC28634D

Yes

No

Yes

No

Yes

No

SD Pin

Yes

No

SD Pin

Yes

No

Yes

VSENSE Pin

SD Pin

6 Pin Configuration and Functions

7-Pin SOIC

Package D

(Top View)

VSENSE

1

2

3

4

8

HV

SD

CS

6

5

VDD

DRV

GND

PIN Functions

I/O

DESCRIPTION

NAME

CS

NO.

3

I

Current sense input

DRV

GND

HV

5

O

G

P

I

Output drive pin for the external flyback MOSFET

Ground reference connection for all signals

4

8

High-voltage connection to the internal high-voltage start-up current source

SD

2

Latching fault shutdown input pin. May be connected to an external temperature sensor

Bias supply input pin to the device. Decoupled with a 1-µF ceramic bypass capacitor,

connect directly across pins 6-4. Connect an additional hold-up capacitor charged from the

transformer auxiliary bias winding to this pin.

VDD

6

1

P

I

Sense pin for the flyback transformer bias and sense winding for output feedback regulation,

output OVP, and input voltage sense/UV protection

VSENSE

5

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

www.ti.com.cn

7 Specifications

7.1 Absolute Maximum Ratings⁽¹⁾

over operating junction temperature range (unless otherwise noted)

MIN

MAX

700

20

UNIT

Start-up pin voltage

Bias supply voltage

HV

VDD

Current sense input

voltage

CS

–0.3

1.5

V

VSENSE

SD

–0.3

–40

-65

VDD

125

All other input pins

Operating junction temperature range, T_J

Storage temperature, T_stg

125

°C

Lead temperature

260

(1) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. Exposure to absolute-

maximum-rated conditions for extended periods may affect device reliability. These are stress ratings only and functional operation of

the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. All voltages

are with respect to GND. These ratings apply over the junction operating temperature ranges unless otherwise noted.

7.2 ESD Ratings

VALUE

UNIT

Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001⁽²⁾

±2000

V_(ESD)

Electrostatic discharge⁽¹⁾

V

Charged-device model (CDM), per JEDEC specification JESD22-

C101⁽³⁾

±500

(1) Electrostatic discharge (ESD) to measure device sensitivity and immunity to damage caused by assembly line electrostatic discharges

into the device.

(2) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process. Manufacturing with

less than 500-V HBM is possible with the necessary precautions. Pins listed as ±2000 V may actually have higher performance.

(3) JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process. Manufacturing with

less than 250-V CDM is possible with the necessary precautions. Pins listed as ±500 V may actually have higher performance.

7.3 Recommended Operating Conditions

over operating junction temperature range (unless otherwise noted)

MIN

0

NOM

MAX

1.0

UNIT

V

CS input

All other inputs (except HV, CS)

SD pin external capacitance

0

VDD

1

0

nF

R_HV, external resistor on HV pin, see Figure 15

R_P, external pull-up resistor on VSENSE pin, see Figure 21

180

3.8

200

3.9

220

4.0

kΩ

6

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

www.ti.com.cn

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

7.4 Thermal Information

UCC28630

D

UCC28631

D

THERMAL METRIC⁽¹⁾

UNIT

7 PINS

128.5

57.3

7 PINS

128.5

57.3

θ_JA

Junction-to-ambient thermal resistance

Junction-to-case (top) thermal resistance

Junction-to-board thermal resistance

θ_JCtop

θ_JB

83.4

°C/W

ψ_JT

Junction-to-top characterization parameter

Junction-to-board characterization parameter

12.3

ψ_JB

82.1

(1) For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.

UCC28632

D

UCC28633

D

UCC28634

D

THERMAL METRIC⁽¹⁾

UNIT

7 PINS

128.5

57.3

7 PINS

128.5

57.3

7 PINS

128.5

57.3

θ_JA

Junction-to-ambient thermal resistance

Junction-to-case (top) thermal resistance

Junction-to-board thermal resistance

θ_JCtop

θ_JB

83.4

°C/W

ψ_JT

Junction-to-top characterization parameter

Junction-to-board characterization parameter

12.3

ψ_JB

82.1

(1) For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.

7

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

www.ti.com.cn

7.5 Electrical Characteristics

over operating junction temperature range (unless otherwise noted) and VDD = 12 V

PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

UNIT

START-UP CURRENT SOURCE

VDD pin short-circuit charging

current

I_VDD0

I_VDD1

I_LEAK

VDD = 0.2 V, V_HV= 100 V

VDD = 11.9 V, V_HV= 100 V

0.6

1.1

0.9

4.0

0.1

1.2

7.6

0.5

mA

μA

VDD pin final charging current

VDD = 18 V, V_HV= 100 V HV,

current source off, T_A= 25°C

HV current source leakage current

SUPPLY VOLTAGE MONITORING

V_DD(start)

VDD start-up voltage

VDD increasing

13.00

7.3

14.75

8.0

16.50

8.5

V

VDD minimum operating voltage

after start-up

V_DD(stop)

VDD decreasing after start-up

V_DD(hyst)

V_DD(reset)

VDD start – VDD stop level

VDD reset restart level

6.5

5.0

V

3.5

6.5

VDD increasing after start-up,

UCC28630, UCC28631, UCC28632,

UCC28633

16.5

17.5

18.3

V

V_DD(ovp)

VDD over-voltage protection level

VDD increasing after start-up,

UCC28634 only

14.0

6.0

14.85

9.0

15.55

13.0

110

V

(1)

Supply current during normal

operation

V_SENSE= 0.45 V, CS = 0 V See

C_LOAD= 700 pF on DRV

I_DD(run)

mA

μA

Supply current during sleep mode,

between switching pulses

V_SENSE= 8.0 V, V_CS= 1.0 V, light-

load mode at 200 Hz, T_A= 25°C

I_DD(sleep)

90

OSCILLATOR

f_SW(max)

f_SW(min)

D_MAX

Maximum switching frequency

V_SENSE= 0.45 V, V_CS= 0 V

110

120

0.20

70%

600

130

kHz

V_SENSE= 8.0 V, V_CS= 1.0 V, light-

load mode

Minimum switching frequency

Maximum Duty Cycle

Minimum On time

0.18

0.22

V_SENSE= 0.45 V, V_CS= 0 V

V_SENSE= 8.0 V, V_CS= 1.0 V, light-

load mode

t_ON(min)

550

650

ns

f_SW(dith)

t_DITH

Frequency dither range

Dither repetition period

Except UCC28632

± 6.7%

6.0

ms

SHUTDOWN (SD) PIN (EXTERNAL FAULT INPUT)⁽²⁾

(2) (3) (4)

I_PULLUP

Internal pull-up current source

See

,

185

3.2

210

3.5

235

3.8

µA

V

(2) (3)

,

⁽⁴⁾, UCC28630,

UCC28631, UCC28632,UCC28633

V_TRIP(rise)

Fault ok level (rising)

(2) (3)

See

,

⁽⁴⁾, UCC28634 only

2.2

1.7

2.5

2.00

1.5

2.8

2.3

V

(2) (3) (4)

V_TRIP(fall)

Fault trip level (falling)

,

(2) (3) (4)

V_TRIP(hyst)

,

V

(2) (3)

V_WAKE(rise)Wake-up level (rising)

t_WAKEWake delay time

(1) C_LOAD= 700 pF included on DRV pin.

,

⁽⁴⁾Except UCC28633

1.8

2.2

2.6

V

Delay to first DRV pulse

10

µs

(2) The SD pin functions as an NTC input pin (with internal pull-up) during normal operation. The internal pull-up is clamped to 4 V. At start-

up, the external temperature sensor (NTC) must be cool enough that the SD pin pulls up above the V_TRIP(rise)start level. After start-up, if

this pin is pulled below V_TRIP(fall)level, this activates external over-temperature shut-down.

(3) During low power modes (when F_SW< F_SMP(max)), the internal SD pin pull-up is disabled, and the pin functions as a transient wake-up

input. In this case, if the pin is raised above V_WAKE(rise)level, the device wakes from low power sleep mode (rather than waiting for the

scheduled timer-based wake). This is useful for applications that require a response to load transients from zero or near-zero load,

where a wake-up signal can be appropriately coupled to the SD pin from the secondary-side.

(4) A decoupling capacitor on the SD pin should not be required; if used, it must not exceed 1 nF.

8

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

www.ti.com.cn

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

Electrical Characteristics (continued)

over operating junction temperature range (unless otherwise noted) and VDD = 12 V

PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

UNIT

VSENSE Pin (MAGNETIC SENSE)

Required positive voltage at

VSENSE pin during off-time (at

25°C)

Internal output voltage sense

reference level

V_OUT(ref)

7.425

7.500

7.575

V

t_OUT(smp)

V_OUT(ovp)

Vsense sample delay for V_OUT

Measured w.r.t. DRV falling edge

1.7

µs

Internal output voltage sense OVP

level

Measured w.r.t. regulation level,

tracking

120%

CURRENT SENSE (CS) Pin

V_CS(max)

V_CS(min)

V_SLOPE

Peak CS pin voltage level

At maximum modulator demand

At minimum modulator demand

800

172

30

mV

Peak CS pin voltage level

Slope compensation ramp

mV/µs

OVER TEMPERATURE PROTECTION

Thermal protection shutdown

temperature

Default internal setting, latch-off

protection

TEMP_TRIP

125

°C

TEMP_HYSTThermal protection hysteresis

10

GATE DRIVE OUTPUT (DRV)

R_OH

R_OL

High level source resistance

Low level sink resistance

I_OH= 100 mA

I_OL= –100 mA

22

35

Ω

1.2

2.5

9

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

www.ti.com.cn

7.6 Typical Characteristics

1

0.995

0.99

4

3.9

3.8

3.7

3.6

3.5

3.4

3.3

3.2

3.1

3

0.985

0.98

0.975

0.97

0.965

0.96

0.955

0.95

0

50

100

150

0

50

100

150

œ50

Temperature (°C)

C002

C003

Figure 1. I_VDD0Charging Current vs. Temperature

Figure 2. I_VDD1Charging Current vs. Temperature

12

14.9

11.5

11

10.5

10

9.5

9

14.85

14.8

14.75

14.7

8.5

8

14.65

14.6

7.5

7

0

50

100

150

0

50

100

150

œ50

Temperature (°C)

C004

C005

Figure 3. I_DD(run)Current vs. Temperature

Figure 4. V_DD(start)Threshold vs. Temperature

8.2

8.15

8.1

1.036

1.03

1.024

1.018

1.012

1.006

1

8.05

8

7.95

7.9

0.994

0.988

0.982

0.976

0.97

7.85

7.8

7.75

7.7

0

50

100

150

œ50

-50

0

50

100

150

Temperature (°C)

C006

Temperature (^oC)

D001

Figure 5. V_DD(stop)Threshold vs. Temperature

Figure 6. Normalized V_DD(ovp)Threshold vs. Temperature

10

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

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ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

Typical Characteristics (continued)

5

4.95

4.9

7.6

7.58

7.56

7.54

7.52

7.5

4.85

4.8

4.75

4.7

7.48

7.46

7.44

7.42

7.4

4.65

4.6

4.55

4.5

0

50

100

150

0

50

100

150

œ50

Temperature (°C)

C008

C009

Figure 7. V_DD(reset)Threshold vs. Temperature

Figure 8. V_OUT(ref)vs. Temperature

122

121

120

119

118

117

116

115

205

204

203

202

201

200

199

198

197

196

195

0

50

100

150

0

50

100

150

œ50

Temperature (°C)

C010

C011

Figure 9. F_SW(max)vs. Temperature

Figure 10. F_SW(min)vs. Temperature

1.02

1.015

1.01

1.005

1

212

211

210

209

208

207

0.995

0.99

0.985

0.98

0

50

100

150

0

50

100

150

œ50

Temperature (°C)

C012

C013

Figure 11. DRV Programming Current Measure vs.

Temperature

Figure 12. SD Pull-Up vs. Temperature

11

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

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Typical Characteristics (continued)

2.04

2.25

2.23

2.21

2.19

2.17

2.15

2.02

2

1.98

1.96

1.94

1.92

1.9

0

50

100

150

0

50

100

150

œ50

Temperature (°C)

C014

C015

Figure 13. SD V_TRIP(fall)vs. Temperature

Figure 14. SD V_WAKE(rise)vs. Temperature

(except UCC28633)

12

UCC28630, UCC28631

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8 Detailed Description

8.1 Overview

The UCC28630, UCC28631, UCC28633, UCC28633 and UCC28634 family of devices are highly-integrated,

primary-side-regulated (PSR) flyback controllers. The device supports magnetically-sensed output voltage

regulation via the transformer bias winding. This feature eliminates the need for a secondary-side reference,

error amplifier and opto-isolator. The device employs an advanced internal control algorithm that offers accurate

static output voltage regulation against line and load. The fixed-point, magnetic-sampling scheme allows

operation in both continuous conduction mode (CCM) and discontinuous conduction mode (DCM). Additionally,

the device achieves accurate constant-current (CC) control of the output current limit using only primary-side,

current sensing. Uniquely, this CC function operates seamlessly as the operating mode changes between DCM

and CCM operation.

The controller includes an internal, high-voltage (HV) start-up current-source, and employs low-power sleep

modes and switching frequency reduction, to improve light-load efficiency and standby power. The device

typically achieves standby power levels between 0.05% and 0.1% of peak output power.

The controller operates in either DCM and CCM, using a mix of peak current-mode PWM (AM) and switching-

frequency modulation (FM) schemes. The control approach improves performance (efficiency, size and cost) and

can reduce transformer size and cost by allowing operation in CCM with FM during peak overload conditions.

Extensive protection features are incorporated, including output overvoltage protection (OVP), bias rail

overvoltage and undervoltage (OV/UV), active X-capacitor discharge, line undervoltage and brownout protection,

overcurrent overload timer, open- and short-circuit pin protections, peak current adjustment with line and

frequency dither for system EMI reduction. The various devices in the UCC2863x family offer a different mix of

features to suit a wide range of applications and requirements.

13

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

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8.2 Functional Block Diagram

+

V_REFx 120%

OVP Fault

V_REF

t_SMP1

CV Demand

+

Voltage Loop

Compensation

V_O

t_SW

V_OSample

t_SMP2

Min

Demand

FM + AM

Modulator

I_LIMx V_O

8

VSENSE

1

2

3

4

HV

P_LIM

Sleep Timer

Current Loop

Compensation

I_PK(dem)

V_IN

P_INCompute

P_IN=(V_INx I_SW(MID))x(t_ON/t_SW

V_INSample

CC Demand

OCP Fault

)

P_IN

n

Timing and

Trigger

Generation

t_SMP,n

t_ON(min)

t_ON(max)

/

Overload Timer

t_SW

t_ON

I_SW(mid)

t_SW

t_ON

V_O

V_AC(min)

+

Line UV Fault

SD Fault

Line UV Fault

SD Fault

SD

JFET Control

Fault Filtering and

Status Monitor

OVP Fault

OCP Fault

+

VDD OV Fault

VDD UV Fault

Start-Up and

Bias Control

PWM Enable

t_SMP4

V_TRIP(sd)

I_DD(LIMIT)

and

I_HV(MEAS)

Over-Temp Fault

X-Cap Fault

t_SMP3

Isw

Sample

VDD OV Fault

VDD UV Fault

V_VDD(ov)

+

CS

VDD

6

JFET Control

I_HV(meas)

X-Cap Fault

X-Cap Discharge

Detect

V_DD

Internal

Temp

V_VDD(uv)

Sensor

PWM Enable

Q

Over-Temp Fault

+

t_SW

V_TRIP(TEMP)

S

GND

5

DRV

+

t_ON

R

I_PK(dem)

PWM

Comparator

t_ON(max)

t_ON(min)

14

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8.3 Feature Description

The application designer requires some key device internal parameters in order to calculate the required power

stage components and values for a given design specification. Table 7 summarizes the key parameters.

8.3.1 High-Voltage Current Source Start-Up Operation

The controller includes a switched, high-voltage, current source on the HV pin to allow fast start-up, and

eliminates the static power dissipation in a conventional resistive start-up approach. This feature reduces standby

power consumption.

The HV pin has three major functions:

•

Supply the device start-up current

Supply the device bias power during latched fault mode

AC sense input for X-capacitor discharge detect (UCC28630 and UCC28633 only)

The UCC28630 and UCC28633 input supply to the HV start-up pin must be connected to the AC side of the

bridge rectifier as shown in Figure 15, in order to support X-capacitor discharge. More details are given in Active

X-Capacitor Discharge (UCC28630 and UCC28633 only), below. Connection to the AC side of the bridge also

allows faster detection of AC mains removal under latched fault conditions, allowing prompt reset of latched

faults for fast restart.

EMC

Filter

EMC

Filter

V_AC

R_HV

1

2

3

4

VSENSE

SD

HV

8

1

2

3

4

VSENSE

SD

HV

8

R_HV

UCC2863X

CS

VDD

6

5

CS

VDD

6

5

GND

DRV

GND

DRV

(a) AC-side

(b) DC-side

Figure 15. HV Pin Connection: (a) AC-side, (b) DC-side (UCC28631, UCC28632 and UCC28634 only)

15

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

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Feature Description (continued)

In the UCC28631, UCC28632 and UCC28634, the HV pin can connect to either the AC or DC side of the bridge.

The addition of the 200-kΩ external HV resistance (required for X-capacitor discharge sensing) limits the

available charging current for the external bias supply input capacitor. However, for typical values of between 22

µF and 33 µF of input capacitance, start-up bias times of less than 1.5 s are achievable at 90 V_AC. Start-up time

can be estimated using Equation 1.

6

:

;

). °∂ß

¥

_34!24= 2₍₆× #_6$$× ¨ÆF

G

6

F 6_{$$ ≥¥°≤¥_≠°∏}

: ;

:

;

). °∂ß

where

ꢁ × ꢁ

¾

6_{).(°∂ßꢀ}= 6_2-3×

•

N

for AC connection and V_IN(avg)= V_RMSx √2 for DC connection

(1)

For 90 V_AC, if C_VDD= 22 µF and worst case V_{DD(start_max)}= 16.5 V, then t_STARTis 1.002 s.

Figure 16 illustrates the start-up behavior of the controller. The HV current source has built-in short-circuit

protection that limits the initial charge current out of the bias voltage pin until the bias voltage reaches V_DD(sc)

.

This limits the power dissipated in the HV current source in the event of a short circuit on the VDD pin.

Thereafter, the HV current source switches to full available current. The controller remains in a low-power, start-

up mode until the bias voltage reaches V_DD(start), after which the HV current source is turned off and the controller

initiates a start-up sequence.

16

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Feature Description (continued)

The bias voltage decays during the start-up sequence at a rate dependent on the size of the energy storage

capacitor connected to the VDD pin. The VDD storage capacitor must be sized appropriately to ensure adequate

energy storage to supply both the controller bias power and MOSFET drive power during start-up, until the VDD

rail can be supplied through the transformer bias winding. If the bias voltage falls below V_DD(stop)(due to bias

winding fault or an inadequate VDD storage capacitance), the controller stops switching, and transitions into low-

power mode for a time delay of t_RESET(long), or until the bias voltage falls to the V_DD(reset)level, whichever is

shorter. See VDD Capacitor Selection for required VDD capacitor sizing. Once the time delay elapses, the bias

voltage rapidly discharges to the V_DD(reset)level, followed by turn-on of the internal HV current source, and a

normal restart attempt follows.

V_DDcharge current is limited for

V_DD< 1.0 V (Short circuit protection)

V_DD(start)

V_DD(stop)

Rectified bias winding voltage increases with

soft-start, must exceed falling level on bias

capacitor before reaching V_DD(stop)threshold

Controller OFF

Controller ON

HV Current Source OFF

HV Current Source ON

Device Start Up

Normal Operation

Soft Start

Figure 16. Normal Start-Up Sequence,

(assuming V_AC> UV start threshold)

17

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Feature Description (continued)

8.3.2 AC Input UVLO / Brownout Protection

At start-up, once the VDD pin has reached the V_DD(start)level, the internal start-up current source is turned off.

The controller tests the voltage across the bulk capacitor to determine if the level is high enough to allow the

power stage to start, if it has exceeded the rising AC_ONlevel. Because there is no load across the bulk capacitor

at this stage, the bulk voltage can be used as a proxy for the peak of the AC line. In order to measure the bulk

voltage in a low-loss fashion, the controller generates a sequence of three exploratory switching pulses at a

frequency of f_SW(uv), at minimum peak-current demand level V_CS(min)to avoid audible noise, and to deliver

minimum energy to the output of the power stage.

Based on the magnetic sampling information determined via the bias winding during these switching pulses, if

the output voltage is greater than the output overvoltage threshold, the pulsing stops immediately, and the

controller transitions into latched-fault mode. If, however, there is no overvoltage condition detected at the output,

the pulse-set completes. If the sensed line voltage is above the line AC_ONstart threshold, then the controller

starts up normally, and begins to generate the PWM drive pulses that charge and regulate the output voltage.

Alternatively, if the sensed bulk level is below the AC_ONthreshold, then the controller enters low power mode for

the reset period (t_RESET(short)). It then depletes the VDD rail to the V_DD(reset)level. At this point, the start-up

sequence repeats, and the device generates another set of exploratory switching pulses. This sequence repeats

indefinitely until the AC input is increased to a sufficient level that the bulk voltage exceeds the AC_ONlevel.

V_AC(on)threshold

V_BULK

V_ACrectified

V_DD(start)

V_DD(stop)

V_DD

t_RESET(short)

V_DD(reset)

DRV Terminal

Line UV check

exploratory pulses

Line UV check

exploratory pulses

Normal PWM

Apply AC

t_ONUV(max)at f_SW(uv)

Normal PWM soft-

start

Figure 17. AC Input UVLO Detection and Start Up

Once started, the controller regularly monitors the bulk capacitor voltage. Because the ripple on the bulk

capacitor depends on the load level, the device determines the maximum bulk level every 11 ms (approprite for

minimum AC frequency of 47 Hz), so the AC peak can be determined. The controller provides input undervoltage

protection based on the sensed AC peak level. Once the peak drops below the AC_OFFlevel for the delay period

(t_UV(delay)), the PWM switching halts, and the controller enters low-power mode for the reset period (t_RESET(short)).

The device then discharges the bias voltage to the V_DD(reset)level, followed by a restart sequence. The controller

cycles through the AC_ON, monitoring (detailed above) indefinitely until the AC input again rises above the AC_ON

level.

18

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

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ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

Feature Description (continued)

8.3.3 Active X-Capacitor Discharge (UCC28630 and UCC28633 only)

Safety standards such as EN60950 require that any X-capacitors in EMC filters on the AC side of the bridge

rectifier quickly discharge to a safe level when AC is disconnected. This discharge requirement ensures that any

high-voltage level present at the pins of the AC plug does not present an electric shock hazard. The standards

require that the voltage across the X-capacitor decay with a maximum time constant of 1 second. Typically, this

requirement is achieved by including a resistive discharge element in parallel with the X-capacitor. However, this

resistance causes a continuous power dissipation that impacts the standby power performance. The power

dissipation in the discharge resistors depends on the X-capacitor value. Assuming that the discharge resistor

meets the 1-second time-constant requirement, (in other words, the R-C product is 1 second) the dissipation is

described in Equation 2.

0 = 6²× #₈

8

!#

(2)

Thus at 230 V_AC, the discharge resistor causes 5.3-mW dissipation for every 100 nF of X-capacitance – for a

typical 470-nF X-capacitor value, that causes 25 mW to be lost in the discharge resistors.

The safety standard does not mandate that the X-capacitor is fully discharged to zero within one second. It

simply requires the discharge rate to exhibit a 1-s time constant. Figure 18 shows the discharge characteristic

(for a 1-s discharge time constant) versus time, for disconnection at the peak of 90 V_AC, 115 V_AC, 230 V_ACand

264 V_AC. For AC inputs above 115 V_AC, with 1-s discharge time constant, the voltage does not drop below the

Safety-Extra-Low-Voltage (SELV) 60-V level until 1 s or longer. In fact, at 264 V_AC, 1.83 seconds elapse before

reaching 60 V.

400

V_SELV

350

300

250

200

150

100

50

Xcap_90

Xcap_115

Xcap_230

Xcap_264

0

0.2 0.4 0.6 0.8

1

1.2 1.4 1.6 1.8

2

2.2 2.4

Time (ꢀs)

C016

Figure 18. X-Capacitor Discharge with 1-s Time Constant, for Various Voltages

19

UCC28630, UCC28631

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Feature Description (continued)

8.3.3.1 Improved Performance with UCC28630 and UCC28633

In order to reduce standby power and eliminate the standing loss associated with the conventional discharge

resistors, the UCC28630 and the UCC28633 devices incorporate active X-capacitor discharge circuitry. This

circuit periodically monitors the voltage across the X-capacitor to detect any possible DC-condition (which would

indicate that AC mains disconnection has occurred), and then discharges the voltage across the X-capacitor

using the internal HV current source. The X-capacitor discharge function discharges the X-capacitor to the SELV

60-V level in 1 s (as long as the design considerations discussed in this section are followed).

The device internally monitors the current into the HV pin to determine if the voltage across the X-capacitor in the

EMI filter has a sufficient AC ripple component. If insufficient AC content is detected, then a DC condition is

internally flagged. This causes the controller to enter low-power mode for the reset period (t_RESET(short)), followed

by bias voltage discharge to the reset level (V_DD(reset)) , and then the start-up HV current source turns on again to

effectively discharge the X-capacitor by transferring charge to the VDD reservoir capacitor.

Because the device monitors the HV pin to detect a DC condition on the X-capacitor, the system cannot operate

with DC input to the HV pin. Instead, the HV pin must be connected to an AC source only. The device interprets

any DC input on the HV pin as DC across the X-capacitor, indicating an AC-disconnect event. This causes a

repeating cycle of start-up and shutdown. The device requires an external 200-kΩ of resistance on the HV pin, to

limit the current to a level below the saturation point of the internal HV current source. This limit produces a HV

input current that is approximately proportional to AC line, so that the AC content can be sensed.

The size of the X-capacitor that can be discharged depends on the VDD energy storage capacitor. Assuming the

worst case, a maximum X-capacitor disconnect voltage could be at the peak of 264 V_RMS, and assuming that it

should be discharged down to 60-V SELV level, the minimum allowed VDD capacitor can be sized based on the

worst case V_DD(reset)and V_DD(start)levels as described in Equation 3.

6_{!# ∞´}_;F 6

ꢀ7ꢀ F ꢁ0

1ꢀ.0 F ꢁ.5

:

3%,6

#

6$$

R #₈× F

G = #₈× l

p = #₈× (4ꢂ.15)

6_{$$ ≥¥°≤¥ _≠©Æ}_;F 6

:

;

$$ ≤•≥•¥ _≠°∏

(3)

For example, for a 330-nF X-capacitor value, the required VDD capacitor is 15.9 µF, so a 22-µF capacitor

suffices.

: ;

R 330 Æ& × 48.15 = 15.9 J&

#

6$$

(4)

In order to reduce the power consumption from the high voltage AC line, the device pulses current into the HV

pin at a low frequency with very low duty-cycle. The HV current source on-time (t_ON(HV)) , repeats at intervals of

t_SMP(HV). Moreover, the pulsing occurs in bursts, with a time delay between bursts. The sampling occurs in bursts

of 21, at intervals of t_SMP(HV), with a wait time of t_WAIT(HV)between bursts. This reduces the effective average duty-

cycle to a very low value (approximately 0.2%), and minimizes the overhead of X-capacitor sampling current and

device bias consumption overhead to approximately 2 mW of extra standby consumption at high-line 230 V_AC

.

The device enables the X-capacitor monitor in latched fault mode, and in light-load regions where the power level

is below P_LL(%), as a percentage of the nominal rated level. Above the P_LL(%)level, the X-capacitor monitor is

disabled. At this load level the bulk capacitor discharges at a rate that is sufficient to also discharge the X-

capacitor, which appears in parallel with the bulk capacitor once the bulk voltage drops far enough to forward

bias the bridge rectifier diodes. In this case ensure that the bulk capacitance value is not too large for the power

level desired, which in-turn ensures that the bulk capacitor discharge rate is fast enough to discharge the X-

capacitor to meet the 1-s discharge target. This can be calculated in Equation 5.

0_./-× 0

2 l

^,,%p ×¥_{8#!0(§©≥)}

D

#

"5,+

Q

k6_!#(∞´)²F 6_3ꢀ,6²o

(5)

20

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UCC28632, UCC28633, UCC28634

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ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

Feature Description (continued)

Assuming a worst case AC disconnect at the peak at 264 V_RMS(373 V_PK), and a requirement to discharge to

SELV level of 60 V in t_XCAP(dis)of 1 s, for a P_NOMof 65 W at 87% efficiency, this is calculated in Equation 6.

:

;

6ꢀ × 0.12ꢀ

2 × d

h × 1

0.87

#

"5,+

Q

Lsuz J&

2

:

;

373 F 60

(6)

Once the bulk capacitance value is chosen, also ensure that when the bulk capacitor has been discharged down

to the line UV AC_OFFthreshold, that it continues to discharge to an acceptable level during the line UV

persistence delay time (t_UV(delay)) as shown in Equation 7.

≈ P_NOMì P_LL%

’

2ì

ì t

∆

÷

UV(delay )

ꢀ

«

◊

C_BULK

Ç

2ì AC_OFF²- V

2

SELV

(7)

(8)

Again, taking the example above:

:

;

6ꢀ × 0.12ꢀ

2 × l

p × 0.04

0.87

#_"5,+Q

Lswv J&

2

:

;

2 × 6ꢀ F 60

Once the first constraint is satisfied, the second one is also automatically met.

Figure 19. X-Capacitor Discharge Activation, at 230 V_AC, No Load

(red = X-capacitor, blue = bulk-capacitor, both 100 V/div)

21

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Feature Description (continued)

Figure 20. X-Capacitor Decay Rate Without Active Discharge

(time constant dominated by 20-MΩ probe impedance)

(red = X-capacitor, blue = bulk-capacitor, both 100 V/div)

22

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ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

Feature Description (continued)

8.3.4 Magnetic Input and Output Voltage Sensing

A sense winding on the transformer is used to measure the input voltage and output voltage of the power stage.

This winding is typically the converter bias winding. The sense winding should be interfaced to the VSENSE pin

as shown in Figure 21. This interface requires that the voltage across the winding be scaled with a resistor

divider R_A/ R_B, and then offset with a switched, pull-up resistor R_P(in series with a diode) connected to the gate

drive pin DRV.

5

DRV

V_F

R_A

R_B

R_P

N_B

VSENSE

1

Figure 21. VSENSE Pin Interface Arrangement

During the off-time portion of the switching cycle (also referred to as the flyback interval), the resistor divider (R_B/

(R_A+ R_B)) scales the positive voltage swing at the VSENSE pin for output voltage regulation, as shown in

Figure 22. During this interval, since the DRV output is low, the diode in series with R_Pis reverse-biased, and so

R_Pis out-of-circuit.

V_Ox (N_B/N_S)

R_A

VSENSE = V_Ox K1

GND

N_B

V_Ox K1

V_INx (N_B/N_P)

R_B

Figure 22. V_OUTSense Using the Positive Swing on the Sense Winding

23

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UCC28632, UCC28633, UCC28634

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Feature Description (continued)

During the on-time portion of the switching cycle, when the DRV pin goes high (should swing very close to the

value at the VDD pin), the switched pull-up R_Pallows the negative swing on the winding to be level-shifted

positive, and thus also be sensed at the VSENSE pin, as shown in Figure 23. In this way the bias winding may

be used to sense both line input voltage and output voltage.

NOTE

The input voltage sensed by the transformer bias winding is actually the voltage across

the bulk capacitor, not the AC input voltage, because the bulk capacitor voltage appears

across the primary winding when the flyback switch turns on

Uses of the sensed bulk and output voltages:

•

Input AC mains UVLO

Input brownout

Line-dependent peak-current adjustment

Accurate output-current regulation

Output-voltage regulation

Output over-voltage protection (OVP)

5

DRV

V_F

R_P

V_Ox (N_B/N_S)

R_A

R_B

GND

N_B

V_DRVœ V_Fœ V_INx K2

V_INx (N_B/N_P)

V_VSENSE= V_DRVœ V_Fœ V_INx K2

Figure 23. Line Input Sense by Offsetting the Negative Swing on the Sense Winding

In order to protect the VSENSE pin from excessive negative current in the event of a manufacturing fault (such

as an open circuit on R_P), use a series limiting resistor and clamping diode on the VSENSE pin. Combine the

clamping diode and DRV pull-up diode into a single-package common-cathode diode to reduce the component

count of the system. This is illustrated in Figure 24.

R_P

BAV70

R_A

100 ꢀ

1

2

3

4

VSENSE

SD

HV

8

UCC28630

CS

R_B

VDD

6

5

N_B

GND

DRV

Figure 24. VSENSE Pin Protection and Interface to Bias Winding

24

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

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ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

Feature Description (continued)

The device continually adjusts the input voltage sample delay, measuring the sample half-way through the on-

time interval, to ensure the cleanest signal. The device uses same mid-point sample trigger when measuring the

main MOSFET switch current (I_SW). Sampling MOSFET switch current in the middle of the on-time automatically

measures the average current during the on-time, I_{SW(on_avg)}, which is required for the current limit and overload

timer block.

The output voltage sample point is always time relative to the turn-off instant. Internally, the device uses the CS

pin to determine the cycle end, rather than the PWM falling edge on the DRV pin. The device bases this

determination on the instant that the MOSFET switch current drops below the demanded peak current level

(I_PEAK) at the peak current mode comparator. Some delay always occurs from the falling edge on DRV to the

point when the external power MOSFET turns off. This internal timing method ensures a more accurate measure

of I_{SW(on_avg)}, and also ensures that the output voltage sample point is not measured too early, before the leakage

ringing has subsided. The effect of the gate turn-off delay and the adjustment of the output voltage sample point

is illustrated in Figure 25.

I_PK(dem)

Gate turn-off delay

Current Sense

PWM Comparator

PWM drive

FET Gate

V_Osample

delay

Bias Winding

Time

Figure 25. V_OUTSample Adjust for External Gate Delay

25

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UCC28632, UCC28633, UCC28634

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Feature Description (continued)

The sampling of the input voltage and output voltage signals on the bias winding must be synchronized to the on-

time and off-time flyback intervals respectively, because the signals occur during only those intervals in the

switching cycle. Typical waveforms and timing are illustrated in Figure 26.

More conventional knee-point detection schemes, where the sample is measured at the end of the flyback

interval when the secondary-side current has decayed to zero, inherently always operate in discontinuous

conduction mode (DCM). However, the fixed sample-point scheme used on the UCC2863x has the advantages

of being able to operate in regions of fixed frequency, and being able to operate in continuous conduction mode

(CCM). Fixed sample-point schemes conventionally suffer poorer regulation than knee-point schemes, because

there is always current flowing at the sample instant. This current produces a sensing error as a result of the

voltage drop produced across the secondary-side resistance and leakage inductance. This parasitic voltage drop

varies with output voltage, line and load, thus influencing the regulation. The UCC2863x devices uses a novel

internal compensation scheme to adjust for this parasitic voltage drop, and can deliver excellent static line and

load regulation, even when operating heavily in CCM.

DRV

V_INSample

V_INsample

delay

V_Osample

delay

V_Osample

Sense Winding

Primary Current

Secondary Current

Time

Figure 26. V_INand V_OUTSample Trigger Timing

26

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Feature Description (continued)

8.3.5 Fixed-Point Magnetic Sense Sampling Error Sources

To support operation in CCM, and allow operation at fixed frequency over a large percentage of the load range,

the UCC2863x uses fixed-point sampling rather than knee-point detection. When conventionally used, fixed-point

sampling typically suffers from poorer regulation performance. This poor performance results from the voltage

drops across the secondary-side parasitic resistance R_SEC, and the secondary-side leakage inductance from

secondary-side to bias L_{LK(sec_bias)}, as a consequence of the fact that current remains flowing on the secondary-

side when the device measures the output voltage. As shown in Figure 27, the secondary-side pin voltage that

gets reflected to the bias winding is detailed in Equation 9.

6_3%#= 6_/54+ 6_2%#4+ 6_2(≥•£)F6 _{: ;}+ 6_{2# •≥≤}

: ;

, ¨•°´

(9)

Equation 9 can be expanded and rearranged into Equation 10.

,

6_3%#= 6_/54× l1 F ^{,+(≥•£ ¢©°≥ )}p + 6_2%#4+ )_3%#× k2_3%#+ 2_{# •≥≤}_;o F k)_,/!$× 2_{# •≥≤}_;o

:

,

3%#

(10)

VRECT

+

-

+

V_BIAS

C_OUT

VSEC

-

I_LOAD

-

V_O

I_SEC

VR(sec)

+

R_SEC

+

VRC(esr)

R_C(esr)

+

-

VLEAK

L_{LEAK(sec_bias)}

-

Figure 27. Secondary-Side Pin Voltage Contributors with Secondary-Side Current Flow

27

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ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

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Feature Description (continued)

Many elements contribute errors to the sensed secondary-side pin voltage, when measured across the bias

winding:

•

V_L(leak): Negative voltage drop across the sec-bias leakage inductance L_{LK(sec_bias)}; assuming constant

regulated output voltage, this voltage drop is fixed constant offset, because V_OUT/L_SECis constant as long as

the output is in regulation.

•

V_RECT: Positive voltage drop across the output rectifier (assuming use of a conventional diode). This voltage

drop varies with load current and temperature. However, a constant nominal voltage drop can usually be

used, because the increasing forward voltage drop with increasing load current is largely cancelled by the

decrease in forward drop as a result of the temperature rise that results.

•

V_R(sec): This is the drop across the secondary-side winding resistance. This value depends on loading, and

varies in proportion to the primary peak current demand that is set by the modulator.

V_RC(esr): This is the drop across the output capacitor equivalent series resistance (esr). This value depends on

the difference between the secondary-side winding current and the DC load current being drawn.

Typically, the peak secondary-side winding current I_SECis many times larger than the load current, and the

secondary-side winding resistance is typically larger than the output capacitor esr. Thus, the last term in

Equation 10 involving I_LOADcan typically be neglected.

The leakage inductance and secondary-side rectifier terms represent quasi-constant offset terms, so do not

affect regulation to a significant extent. Thus, the quasi-constant offset terms can be accounted for in the

calculation of the required scaling resistors to produce the desired setpoint voltage.

The remaining term that dominates the regulation error in Equation 10 is the drop across the secondary-side

winding resistance and capacitor esr at the sample instant, {I_SECx(R_SEC+ R_C(esr))}. The controller internally

adjusts the control loop reference in proportion to the primary peak current demand in order to null the I_SEC

related error term in the sampled bias winding voltage. Since the peak secondary-side current I_SEC(pk)is the

primary peak current I_PRI(pk)scaled by the transformer turns ratio, the internal control loop reference effectively

varies in approximate proportion to I_SEC, resulting in dramatically improved regulation performance.

This improved regulation performance allows the use of primary-side regulation in a wider range of applications,

and at unprecedented power levels, operating in both CCM and DCM.

28

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Feature Description (continued)

8.3.6 Magnetic Sense Resistor Network Calculations

Because the device uses the VSENSE pin to measure both V_OUTand V_INof the power stage, it is important to

calculate the resistor values correctly. The step-by-step design process is outlined in this section.

8.3.6.1 Step 1

Depending on the power level, choice of transformer size, and required trade-offs between primary MOSFET and

secondary-side rectifier ratings, the transformer turns N_P, N_Sand N_Bwill be chosen first. The controller can

support a wide range of turns ratios.

5

1

DRV

V_F

R_P

R_A

N_B

VSENSE

R_B2

R_B1

Figure 28. Practical Magnetic Sense Setup with Extra Resistor R_B2for Setpoint Fine Adjust

8.3.6.2 Step 2

Once N_P, and N_Bare known, the required value of R_Ain Figure 28 is calculated using Equation 11.

.

2_!= 2₀× l ^"p × +_,).%

.

0

(11)

In this equation, the internal controller gain K_LINEis 49.25 (see Table 7 for key internal controller parameters),

and the internal gains are designed for a fixed value for R_P, (i.e. R_PMUST be 3.9 kΩ).

29

UCC28630, UCC28631

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Feature Description (continued)

8.3.6.3 Step 3

Once N_S, target V_OUT, output rectifier drop V_RECT, and the secondary-side-to-bias leakage inductance L_{LK(sec_bias)}

are known, the required value for R_Bcan be calculated. Referring to Equation 10, L_{LK(sec_bias)}can be

approximated as a percentage of the secondary-side-referred magnetizing inductance L_SEC. (See Magnetic

Sense Resistor Network Selection for details).

2_!

2_"=

.

k6_/54× k1 F %,_{,+ ≥•£ _¢©°≥}_;o ꢀ 6_2ꢁ#4o × @ ^"A

:

3

L

F 1M

6_{/54 ≤•¶}

:

;

(12)

In this case, R_Bmay need to be empirically adjusted to achieve the required exact output set-point, especially if

V_RECTvaries or is not known precisely. For this reason, it is recommended that R_Bshould be implemented on the

system PCB as two parallel resistors R_B1and R_B2as shown in Figure 28, to allow easier fine-tuning of set-point.

For set-point tuning, only R_Bshould be adjusted. R_Ashould never be adjusted, because to do so would affect the

line sense gain and introduce errors into the line voltage measurement.

8.3.6.4 Step 4

Verify that the equivalent Thevenin resistance R_THof the R_A/R_Bcombination falls in the required range of 10 kΩ

to 20 kΩ.

2_!× 2_"

2_!+ 2_"

10 ´3 O 2₄₍< ꢀ0 ´3

2₄₍

=

(13)

(14)

If the Thevenin resistance is outside of that range, then the original choice of turns ratio must be adjusted, and

design steps repeated until a valid value for R_THis determined. This is unlikely to occur in practice, unless an

extreme turns ratio is chosen. If R_THis outside this range, it triggers the VSENSE pin open or short pin-check at

start-up.

30

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Feature Description (continued)

8.3.7 Magnetic Sensing: Power Stage Design Constraints

Because the controller employs fixed-point sampling for output voltage sensing, there are some transformer

design constraints that must be observed. The minimum magnetizing volt-seconds during the on-time interval

occurs at the minimum CS pin voltage, V_CS(min), under light-load conditions. This minimum should be the case at

all line voltages, because the controller compensates for line-dependent peak-current overshoot during turn-off

delay. The choice of transformer turns ratio, transformer inductance (L_PRI), and current sense resistance (R_CS

)

must ensure that the corresponding reset volt-seconds during the flyback interval are sufficient that a valid output

sample is available at the sample point, t_OUT(smp). This constraint is summarized in Equation 15.

6

2_#3

.

1

:

;

#3 ≠©Æ

3

Q

×

: ;

6_/54+ 6_2%#4

,

02)

¥

.

0

:

;

/54 ≥≠∞

where

•

V_RECTis the voltage drop across the output rectifier

(15)

Additionally, the device requires a minimum on-time, t_ON(min), to ensure enough time for the system input voltage

(V_IN) and switch current (I_SW) to be measured. To meet the minimum on-time requirement at maximum line, and

minimum load, the ratio of current sense resistance (R_CS) to transformer inductance (L_PRI) must meet the

constraint shown in Equation 16.

6_{#3 ≠©Æ}

2_#3

1

:

;

Q

×

,

02)

6

¥

:

;

:

/. ≠©Æ

;

). ∞´ _≠°∏

(16)

Equation 15 or Equation 16 sets the limit for the ratio of R_CSto L_PRI, but both need to be verified. See Typical

Application for more details.

31

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Feature Description (continued)

8.3.8 Magnetic Sense Voltage Control Loop

Because the output voltage feedback is inherently a sampled signal obtained from the bias winding, the internal

voltage control loop is most naturally implemented digitally. The internal control loop implements the equivalent

of a PID loop in digital form. Because the output can be sampled only at certain intervals in each switching cycle,

the sample rate is naturally tied to the switching frequency, and the sample rate increases with increasing

frequency. However, the device clamps the sample rate at a normalized maximum rate, f_SMP(max). But because

the device must always synchronize to the next available switching cycle to obtain a new sample of the output

voltage, the effective sample rate varies somewhat around this value.

The digital control loop compensator block diagram is shown in Figure 29. A new sample of output voltage is

supplied to the compensator at the normalized maximum clock rate (f_SMP(max)) , or f_SW, whichever is lower. An

updated output voltage demand signal, y_k, is produced at the same clock rate. This voltage loop demand

represents the required operating point on the modulator curves to keep the output voltage in regulation. The

modulator sets the appropriate switching frequency and peak current demand depending on the load power.

t_SMP

Error e_k

Output y_k

V_O

-

To f_SWand

I_PK(dem)Modulator

Voltage Loop PID

Compensator

1

V_OSample

VSENSE

+

V_REF(adj)

+

K_R(sec)

I_PK(dem)

V_REF

+

Figure 29. Digital Voltage Control Loop Simplified Block Diagram

The control loop PID gain factors are internally fixed values, optimized for flyback power stages in the range

between 20 W and 130 W. The loop is designed to work with magnetizing inductance values in the range

between 200 µH and 1500 µH. Assuming that the output capacitance value is chosen based on required ripple

current rating, then loop stability is not a problem. Adding extra output capacitance does not degrade the loop

performance and the resulting extra output hold-up improves transient response.

The Typical Application section includes gain-phase measurements taken using the 65-W UCC28630EVM-572

evaluation module.

32

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ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

Feature Description (continued)

8.3.9 Peak Current Mode Control

The controller operates in peak current mode. The primary-side switch (MOSFET) current is sensed by a shunt

resistor (R_CS1) connected in series with the source of the FET as shown in Figure 30. The voltage that is

developed across the sense resistor is connected to the CS pin of the controller. The device uses the current

sense signal at the CS pin to terminate the PWM pulse according to the peak current demand of the modulator.

The device automatically applies slope compensation as soon as the duty cycle of the DRV pin pulse exceeds

50%. This compensation provides stable operation up to maximum DRV duty cycle. The device applies this slope

compensation as a downslope on the demand signal at the PWM comparator, so is not measureable at the CS

pin. The device synchronizes the slope compensation signal to the PWM and is active only between 50% and

70% duty cycle, as shown in Figure 31.

Normal operating range for the CS pin is between 0 mV and 800 mV. The R_CS1resistor should be scaled such

that the peak current at maximum peak load and minimum bulk capacitor voltage produces a signal of

approximately 800 mV peak at the CS pin. This resistor value is calculated in conjunction with the calculation of

the required primary magnetizing inductance, as outlined in Notebook Adapter, 19.5 V, 65 W, section.

1

2

3

4

VSENSE

SD

HV

8

UCC28630

CS

VDD

6

5

C_CS

DRV

GND

R_CS2

R_CS1

Figure 30. Primary-Side Current Sensing

33

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Feature Description (continued)

A nominal 100 ns of filtering that is internal to the CS pin helps filter the leading turn-on spike of current.

Depending on PCB layout, an RC filter (R_CS2and C_CS) may be required on the CS pin as shown in Figure 30 to

filter noise and spikes. The capacitor C_CSshould be positioned as close as possible to pins 3 and 4 and tracked

directly to the pins. Series resistor R_CS2should also be located close to pin 3 to minimize noise pick-up. R_CS2

value should not exceed 20 kΩ, because a larger value could be detected as a possible open circuit on the CS

pin during the start-up pin-fault checks. The R-C filter time constant should not be excessive (timing between 100

ns and 200 ns is typical). Otherwise the filter reduces the measured peak current, and allows greater actual peak

current to flow versus the modulator demand level. Such effects force the regulation loop to reduce the switching

frequency to compensate, and at highest line, no load, this can lead to regulation difficulties if the control loop

attempts to drop the frequency so far that it reaches the f_MINlimit.

50%

70%

100 mV_PP

ꢀ

Peak Current Demand

With Slope Compensation

30 mV/ꢀs

PWM Clock at 60 kHz

Figure 31. Peak Current Demand with Slope Compensating Downslope

34

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Feature Description (continued)

8.3.10 I_PEAKAdjust vs. Line

The controller applies a line-dependent reduction in the peak-current demand to correct for the current overshoot

due to the PWM and gate drive propagation delay, with the aim of delivering a constant peak current versus line

at a given power level. This maintains approximately constant switching frequency versus line for a given power

level (until the operation enters into CCM), improves regulation, reduces audio noise, and allows lower standby

power at high line. If not corrected, the current overshoot could become significant at high line, where the

inductor current di/dt is higher. This overshoot would cause a pronounced increase in transferred power per

switching cycle at high line, because power is proportional to I_PK². The effect of the delay on the peak-current

overshoot is illustrated in Figure 32.

V_BULK

Frequency and

Peak Current

Modulator

L_PRI

R_G(off)

I_PK(dem)adjust

vs. V_BULK

V_BULK

DRV

R_G(on)

Gate

Driver

S

R

Q

5

+

CS

LEBs

Filter

3

R_CS1

C_CS

R_CS2

I_PKovershoot

I_PK(dem)

I_PK(adj)

Propagation delay

Low line

High line

Figure 32. Peak-Current Demand Adjustment vs V_BULKto Correct Prop Delay Overshoot

For different power stage designs, the combination of primary magnetizing inductance L_PRI, current sense

resistance R_CSand external MOSFET gate turn-off delay t_OFF(ext), must be verified against Equation 17, to ensure

that the internal peak-current compensation gain range is satisfied. The K_LINE(adj)factor should be within the

range indicated. If the external turn-off delay is too long, then the internal I_PEAKadjustment factor is too low, and

the adjustment at high line is not able to achieve the required level of over-shoot compensation. As noted

previously, this could result in regulation difficulties at no-load, and may cause poor line and load regulation, or

require an increase in output pre-load power.

2_#3

+

= F

× k¥_{02/0 ß°¥•}ꢀ ¥

_;oG P str J °Æ§ < uwrJ

/&& •∏¥

:

;

:

;

:

,).% °§™

,

02)

where

•

where t_PROP(gate)is the internal controller gate-drive turn-off propagation delay, given in Table 7.

(17)

35

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Feature Description (continued)

8.3.11 Primary-Side Constant-Current Limit (CC Mode)

In addition to the peak-current mode PWM function, the device also uses sensed current at the CS pin to

estimate the secondary-side load current. The device samples the CS pin voltage and measures it in the middle

of the on-time, which is effectively the average switch current during the on time, I_{SW(avg_on)}. This measurement

scheme is the case during both DCM and CCM operational modes. The average switch current during the on

time is scaled by the PWM duty cycle to give the I_IN(avg)of the power stage. The power stage input power, P_IN,

can then be estimated as the product of (V_INx I_IN(avg)). The CC mode operation regulates P_INto track (I_OUT(lim)

V_OUT), if P_INincreases to reach P_IN(lim), thereby achieving a regulated constant current as shown in Equation 18.

6_/54× )_{/54 ¨©≠}

x

:

;

0 = 6 × )_{). °∂ß}

=

= 0_{). ¨©≠}

:

;

).

D

(18)

6 × )_{). °∂ß}_;× D

0_{). ¨©≠}_;× D

:

).

:

)

=

= )_{/54 ¨©≠}

:

;

/54

6_/54

(19)

t_SMP1

V_O

P_LIM

V_OSample

I_LIMx V_O

To f_SWand

I_PK(dem)Modulator

+

Output iy_k

Error

ie_k

Current loop

PI compensator

VSENSE

1

V_IN

-

P_INCompute

V_INSample

P=(V_INx I_SW(MID))x(t_ON/t_SW

)

P_IN

t_SMP2

t_ON

t_SW

t_SMP3

I_SW(mid)

I_SW

Sample

3

CS

Figure 33. Digital Current Control Loop Simplified Block Diagram

36

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Feature Description (continued)

Assuming that the power stage efficiency does not change significantly with operating point, by regulating the

input power in inverse proportion to output voltage, this regulates output current. This achieves a brick-wall CC

characteristic, where the output current is regulated as the input voltage changes and as the output voltage rolls

off, regardless of power stage operating mode (CCM or DCM). The CC mode protection eliminates the

characteristic load current tail-out that is typically seen with peak-current mode control as output voltage

collapses and operation goes deeper into CCM mode.

NOTE

As the output voltage decreases in CC mode, the VDD level also decreases. If the

overload is severe, the drop in output voltage causes VDD to drop below the V_DD(stop)UV

level. This drop causes a shutdown for t_RESET(long), as given in Table 7, followed by a

restart attempt.

The constant-current mode output current limit level (I_OUT(lim)) is a function of both the R_CS1resistor and the

transformer turns ratio. The device uses an internal reference and gain for the CC loop, K_CC1and K_CC2, that set

the CC I_OUT(lim)point as a function of the chosen turns ratio, output voltage and current sense resistance as

shown in Equation 20.

1

2_#31

.

+

##1

0

)

=

;

×

:

/54 ¨©≠

.

3

0

+

_##ꢀꢁ6_/54×

.

3

(20)

For the UCC28631, UCC28632 and the UCC28633 devices, the I_OUT(lim)can be adjusted to be a percentage of

the maximum value calculated by equation Equation 20. see CC-Mode I_OUT(lim)Adjustment for more details.

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Feature Description (continued)

8.3.12 Primary-Side Overload Timer (UCC28630 only)

The internal overload timer in the UCC28630 uses the same output load current measurement that is used by

the CC loop. This measurement tracks the power stage thermal stress, and protects the power stage against

output overload. If the output is overloaded for too long such that the power stage would be over-stressed, then

the PWM shuts down, and enters low-power mode for a time period of t_RESET(long); thereafter the device

discharges VDD to the V_DD(reset)level and initiates a hiccup mode restart.

The overload timer operates by taking an estimate of output current, squaring it (assuming the power stage

losses are dominated by resistive I²losses) to produce (K x I²_OUT), where K is a scaling gain factor. The overload

timer is constantly running at every load level, and accumulates at a rate dependent on the difference between

(K x I²_OUT) and the previous level of the timer. If (K x I²_OUT) is greater than the previous timer level, the timer level

continues to increase; if (K x I²_OUT) is less than the previous timer level, then the timer level decreases. At any

steady load, the overload timer level eventually settles at a level proportional to I²_OUT. Because the overload

timer level adjusts at a rate dependent on the difference between (K xI²_OUT) and the previous level, the timer

initially reacts faster to larger differences, but over time settles exponentially at a level proportional to (K x I²_OUT).

As shown in Figure 34, in both the first and second examples, the initial steady load allows the timer to integrate

and settle at a level proportional to the load. The margin to the over-load trip level depends on the historical

loading, lower prior average loading results in greater future over-load capability, and vice versa. The rate at

which the timer reacts to different load steps is set by the chosen time constant (or response rate) per Table 2.

The overload timer can cope with pulsed loads and loads with a complex waveform. Because the rate of

increase and decrease also depends on the load change from the previous load, it also times out faster for

bigger overloads, or allows a smaller overload to run for much longer. The overload timer operates in both

normal CV mode and overload CC mode, or a dynamic mix of both modes.

38

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Feature Description (continued)

P_PEAK

P_TRIP

Load Power

Overload Timer Value

Overload Trip Point

Time

Example 1: Operation at P_RATEDcontinuously; small load

increase after long time œ causes overload timer to trip

P_PEAK

P_TRIP

Load Power

Overload Timer Value

Overload Trip Point

Time

Example 2: Operation at low power continuously; step to peak load

causes fastest overload timer ramp-up rate to trip level

P_PEAK

P_TRIP

Load Power

Overload

Timer Value

Overload Trip Point

Time

Example 3: Operation at low power continuously; repeated short-pulse steps to

peak load œ excessive duty cycle causes eventual overload timer trip

Figure 34. Overload Timer Example Waveforms Under Various Load Scenarios

t_SMP1

V_O

P_LIM(cc)

V_OSample

I_LIMx V_O

Overload

Signal

To Fault

Mgmt Block

2

X²

Block

Overload Timer

Integrator

P_IN

+

VSENSE

1

+

CC/CV Mode

Detect Switch

V_IN

-

P_INCompute

P=(V_INx I_SW(MID)x(t_ON/t_SW

V_INSample

)

P_MEAS(cv)

2

P_TRIP

t_SMP2

t_SW

t_SMP3

t_ON

I_SW(MID)

I_SW

Sample

3

CS

Figure 35. Overload Timer Block Diagram

39

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Feature Description (continued)

8.3.13 Overload Timer Adjustment (UCC28630 only)

The UCC28630 overload timer trip level and time constant are both selectable from a defined list of

combinations. The user can select the overload timer trip level as a percentage of the rated continuous nominal

power, P_NOM(see Figure 41), and the timer response speed. The available choices are detailed in Table 2.

Table 2. Overload Timer Adjustment

R_PROGPROGRAMMING RESISTOR (kΩ)

TIMER CONTINUOUS OPERATION

P_TRIP/P_NOM(%)

TIME CONSTANT AT 200% of P_NOMOR IN

CC MODE (ms)

(E96 series values)

Open, or > 47

20.0

160

135

110

1000

500

12.7

150

9.31

1000

500

7.32

6.04

150

5.11

1000

500

4.42

3.92

150

The desired pull-down resistance on the DRV pin sets the required overload parameters, as shown in Figure 36.

The controller measures the resistance value on the DRV pin at start-up using a low-level test voltage (400 mV

to ensure it is well below the lowest possible power MOSFET gate threshold voltage) and sensing the current

that flows. Thus, based on the resistance R_PROG, the required set of timer parameters can be chosen.

1

2

3

4

VSENSE

SD

HV

8

UCC28630

CS

VDD

6

5

GND

DRV

R_PROG

Figure 36. Overload Timer Setting Adjustment

(with programming pull-down resistor on DRV pin)

To ensure that the sensed current does not sit close to an interval boundary, the resistor values listed in Table 2

(or the closest value possible) should be used. These recommended resistor values position the test current in

the center of each interval.

40

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8.3.14 CC-Mode I_OUT(lim)Adjustment

For the UCC28631, UCC28632, UCC28633 and UCC28634, the pull-down programming resistor on the DRV

pin, as shown in Figure 36, sets the desired CC-Mode limit. The available CC-Mode levels are listed in Table 3,

where the CC limit is given as a percentage of the maximum allowed value from Equation 20.

Table 3. CC-Mode Levels

R_PROG(kΩ)

Open or > 47

20

CC LIMIT

100%

90%

12.7

80%

9.31

75%

7.32

70%

6.04

65%

5.11

60%

4.43

55%

3.92

50%

8.3.15 Fault Protections

The controller has several built-in fault protections. Most faults are subject to internal persistence filtering to avoid

false-tripping due to noise or spurious glitches from external events. When a fault is detected and persists for the

corresponding filter delay time, the device terminates and disables the PWM drive signal. No PWM activity

occurs if the fault (pin faults for example) is detected at start-up . Table 4 lists all fault sources, persistence

delays and the associated response (latching or auto-restart).

In the case of auto-restart (sometimes called hiccup-mode) faults, the device enters low-power mode for a time

period of t_RESET(long)(or t_RESET(short)in the case of AC line UV fault and X-capacitor discharge), then discharges

the VDD pin to the V_DD(reset)level, followed by a restart attempt. The device continues in a repeating shutdown-

delay-restart loop until the fault is removed. Once the fault clears, the controller restarts automatically, there is no

need to remove and re-apply AC input voltage to the system.

Latching faults do not allow any PWM restart attempts until the AC input voltage is removed. In this case the

controller enters low-power mode. During low-power mode, the device regulates the VDD pin between two levels

V_{DD(latch_hi)}and V_{DD(latch_lo)}, as given in Table 7, using the start-up HV current source. This regulation keeps the

controller biased to maintain the latched fault condition as long as AC voltage is present at the input. When the

device loses AC input voltage during latched-fault mode, the controller resets, and restarts when the AC input is

re-applied.

If there is an open-feedback fault due to an open or short on the VSENSE pin or associated external resistor

divider on the aux winding, the output voltage is protected against an over-voltage condition. If the open-

feedback fault occurs before power-up, the fault will be detected by VSENSE pin- fault protection (see next

section 9.3.16), and the controller will not generate any PWM drive signal. This prevents any possible output OV

due to this open-feedback fault condition. If the open-feedback occurs after power-up, when the power stage is

already operating, the open-feedback condition can cause Vout to increase. In this case, the VDD level will also

increase in proportion to Vout (they will track based on the Flyback transformer turns ratio). When the VDD rail

reaches the V_DD(ovp)protection threshold, the PWM will be disabled, and the controller will go to fault mode, as

described above. The V_DD(ovp)protection is used as an indirect back-up OV protection mechanism for the main

output under running open-feedback fault conditions. The level of output OV depends on the ratio of the normal

VDD regulation level to the V_DD(ovp)level. Note that UCC28630/1/2/3 use V_DD(ovp)trip level of 17.5 V nominal,

whereas the UCC28634 uses a lower V_DD(ovp)of 14.85 V nominal, to ensure a lower/tighter level of output OV

under VSENSE open-feedback conditions. As a result, the user must be careful to choose the number of turns in

the transformer aux winding to ensure that the normal VDD regulation is below the V_DD(ovp)protection level, to

avoid false-triggering of the V_DD(ovp)protection.

41

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Table 4. Fault Sources and Associated Responses

RESPONSE

FILTER

DELAY TIME

UCC28631,

UCC28632,

UCC28633

FAULT TYPE

TYPICAL CAUSE

UCC28630

UCC28634

Excessive transformer leakage; system board

fault

(1)

VDD OV

VDD UV

125 μs

Latching

Auto-restart

(1)

Insufficient VDD capacitor; system board fault

125 μs

Auto-restart

Latching

Auto-restart

AC brownout AC voltage removal or extended dip

40 ms

(1)

OverTemp

SD pin low

Internal T_J(max)reached

125 μs

(1)

External NTC over-temperature event

125 μs

Latching

Programmable

Overload timer Excessive load power for too long

Auto-restart

Latching

N/A

(2)

System board fault; system output voltage

Output OV

(1)

125 μs

Auto-restart

back-driven excessively

(3)

VSENSE pin Short or open detected at start-up

No filter

Latching

Auto-restart

(3)

DRV pin

CS pin

Short detected at start-up

No filter

(3)

Short or open detected at start-up

No filter

(3)

Internal fault Internal chip diagnostics fault detected

No filter

(1) The filter delay time is either 125 μs or 2 PWM periods, whichever is longer.

(2) The overload timer delay can be programmed as shown in Table 2.

(3) Because these faults are only identified before PWM commences, noise filtering is not required.

42

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8.3.16 Pin-Fault Detection and Protection

The controller includes protection against most practical pin faults. These faults include open pins, pins shorted

to adjacent pins, pins shorted to GND and pins shorted to VDD. The device performs pin fault checking at start-

up, before the PWM is enabled. Table 5 summarizes the response to pin faults. Most faults cause either a

latched shut-down, or failure to start-up. For UCC28634, all pin-faults are non-latching.

A short-circuit from the HV pin (pin 8) to the VDD pin (pin 6) is unlikely to occur, because pin 7 is not included in

the package. The HV pin and tracking requires additional PCB spacing in any event to meet creepage

requirements. However, if such a fault does occur, the device continues to charge the VDD capacitor through the

HV pin external series resistor, and the power supply starts up and appears to operate normally. But because the

HV and VDD pins are shorted, the internal HV current source cannot switch-out the external HV resistor, so it

always dissipates power. This condition results in a large increase in no-load standby power. A 200-kΩ external

HV resistor, dissipates 66 mW at 115 V_AC, and 265 mW at 230 V_AC. At load levels where the X-capacitor

discharge function is operational, the short to VDD appears to be an AC-disconnect event, and causes the

device to cycle on and off.

Table 5. Pin Faults and Associated Responses

PINS

OPEN

ADJACENT SHORT

GND SHORT

VDD SHORT

NAME

VSENSE

SD

NO.

1

Latched fault

Normal operation

Latched fault

N/A

No start-up

Latched fault

No start-up

2

CS

3

GND

4

Device fails, power supply

damaged

DRV

VDD

no pin

HV

5

6

7

8

Hiccup fault

No start-up

N/A

No start-up

N/A

Latched fault

No start-up

N/A

No start-up

N/A

No start-up

N/A

No start-up

Fault not detected/Hiccup

fault

43

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8.3.17 Over-Temperature Protection

The controller has built-in thermal protection. If the controller junction enters an over-temperature condition, the

controller shuts down. The fault response (latching or auto recovery) depends on the device variant, per Table 4.

There is 10°C hysteresis in the over-temperature trip point, the controller only restarts if the junction temperature

has dropped by at least 10°C below the trip level.

8.3.18 External Fault Input

An external fault input signal may be applied to the controller SD (shutdown) pin. This signal forces the controller

into fault mode. To trigger the fault, the voltage on this pin should be pulled below the fault trip threshold. A

typical application is shown in Figure 37, where this pin is used to shut down the controller in the event of an

over-temperature event as detected by a NTC (negative temperature coefficient) thermistor. The device pulls up

the SD pin internally using a current source. As temperature rises, the external NTC resistance decreases,

reducing the voltage on the pin. When the pin voltage drops to the fault trip threshold, the controller enters fault

mode. The fault response (latching or recovery) depends on the device variant, per Table 4.

VSENSE

HV

4V5

210 mA

UCC28630

1

8

SD

2

Over

Temperature

R_ADJ

VDD

DRV

CS

+

6

5

3

NTC

R1 at 25°C

R2 at T_TRIP

t^o

+

2.0 V

GND

4

Figure 37. Fault Interface to SD Pin

The required trip resistance can be calculated from the internal trip voltage and pull-up current source. Nominally,

this is 9.5 kΩ. Choose the NTC should so that it can achieve this value of resistance at the desired hot-spot trip

temperature. If the NTC resistance is too low at the required trip temperature, connect a standard chip resistor in

series to bring the total resistance up to 9.5 kΩ.

The device internally filters the SD pin with persistence delay as listed in Table 4. An external filter capacitor is

not normally necessary. However, if an application uses an external filter capacitor, the value should be limited to

1 nF maximum. A larger value may impact the useful life of the controller.

44

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8.3.19 External SD Pin Wake Input (except UCC28633)

During low-power modes (when f_SW< f_SMP(max)), the device disables the internal pull-up on the SD pin. This

action allows the pin voltage to fall to GND, and the SD pin then functions as a transient wake-up input. In this

case, if the pin rises above the wake threshold while the device is in low-power sleep mode, the device wakes

and starts PWM pulses immediately. This feature is useful for applications that require a faster response to load

transients from zero or near-zero load, where a wake-up signal can be appropriately coupled to the SD pin from

the secondary side.

Figure 38 describes a typical secondary-side wake circuit and coupling of the wake signal to the controller on the

primary side. This circuit uses a TL103W component which is an integrated reference plus two op-amps in a

convenient SOIC-8 package. Both op-amps are connected to the same internal 2.5-V TL431 type reference, with

a 3-resistor divider chain allowing each op-amp to monitor a different level. The upper op-amp output is low as

long as the device is regulating the output voltage normally. If a sufficiently large load transient occurs while the

primary-side controller is in sleep mode, the output voltage drops below a transient wake level. The upper op-

amp output goes high, driving current through the low-cost wake signal opto-coupler. On the primary side, the

wake opto-coupler pulls up the SD pin above the wake threshold and forces PWM switching as a reaction to the

load transient.

The lower op-amp section monitors the output voltage and its output goes low only when the output voltage is

above a minimum enable threshold for the secondary-side wake-up monitor. This action is necessary so that

under certain conditions, such as a start-up sequence or short-circuit condition (when the output voltage is

already below the transient wake level) that the secondary-side circuit does not continually drive the wake opto-

coupler, which could activate an SD pin fault during pin-fault checking at start-up.

V_OUT

R_PULLUP

HV

VSENSE

LED

8

1

2

3

4

UCC28630

TL103W

SD

Wake-up

signal

+

VDD

DRV

CS

+

2.2 V

6

t^o

2.5 V

GND

5

Figure 38. Typical Secondary-Side Voltage Monitor and Wake-Up Circuit for Interfacing to the SD Pin

45

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8.3.20 External Wake Input at VSENSE Pin (UCC28633 Only)

The UCC28633 device variant supports fast PSR transient response via the VSENSE pin. When the loop

demand drives the modulator frequency below approximately f_SMP(max), the controller enters a low-power sleep

mode for a portion of the switching cycle. The sleep interval varies, depending on the switching frequency

commanded. The sleep interval is longer for lower switching frequency, and longest at f_SW(min). For conventional

PSR controllers, if a load transient occurs during this sleep interval, the controller will not react until the next

timed wake-up, during which the output voltage can drop significantly, depending on the size of the load step and

the amount of output capacitance.

The UCC28633 can respond to fast transient wake signal coupled to the VSENSE pin. If the wake signal

exceeds an internal pin threshold V_SENSE(wake)while the controller is in sleep mode, the sleep interval is

terminated and PWM activity commences within a typical delay time of t_WAKE. This dramatically improves the

response to heavy load transients from zero load, or very light load. If the switching frequency is above f_SMP(max)

,

the controller never enters sleep mode, so wake response on the VSENSE pin never enabled. The

commencement of any sleep interval in the controller is delayed until the resonant ringing on the VENSE pin has

decreased below the V_SENSE(wake)threshold for at least 2 µs. Once the ringing has decreased, the wake response

is enabled, and the sleep interval commences.

V_OUT

EMC

Filter

UCC24650

V_AC

WAKE VDD

5

1

GND

2

UCC28633

VSENSE

1

2

3

4

HV

8

SD

t^o

CS

VDD

DRV

6

5

GND

Figure 39. UCC24650 Secondary-Side Voltage Monitor and Wake-Up Circuit

46

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The wake signal at the VSENSE pin can be generated using a secondary side low power voltage monitor such

as UCC24650, as shown in Figure 39. Further details can be found in the datasheet for UCC24650. This

secondary-side monitor uses the switching activity on the secondary winding to trigger refresh of an internal

sample-and-hold circuit to measure and record the system output voltage at the VDD pin. Thereafter, if the actual

output voltage, sensed at the VDD pin, drops by ΔWAKE% (see UCC24650 detailed datasheet specifications) of

the previously sampled value, the WAKE pin is internally pulled low through a current-limited open-drain switch.

As shown in Figure 39, the main output rectifier diode is positioned at the return side of the secondary winding,

so that the GND-referenced UCC24650 WAKE function can be deployed. In effect, the WAKE pin shorts out the

rectifier diode for a short interval (see UCC24650 detailed datasheet specifications), to draw some current from

the output capacitor through the transformer secondary winding. This sets up a low-level pulse of current that

then rings resonantly in the power circuit magnetizing inductance and parasitic capacitance. The ringing causes

a similar ringing voltage waveform on all transformer windings, including the bias/sense winding, which interfaces

to the VSENSE pin. If the initial pulse of current drawn by the secondary WAKE pin is sufficient, then the ringing

voltage at the VSENSE pin is large enough to exceed the V_SENSE(wake)threshold.

The UCC24650 datasheet Application Information section includes details of how to estimate the amplitude of

the wake-pulse ringing at the WAKE pin. In some cases, especially at higher rated output power, the transformer

magnetizing inductance is lower, while the total switch node capacitance tends to be higher. This reduces the

transformer impedance, and can also result in reduced wake pulse amplitude. In these cases, the UCC24650

WAKE pin output can be augmented with an external PNP circuit Q1, R1 and R2, as shown in Figure 40. In this

case, when the WAKE pin pulls low, Q1 turns on, and draws more current through the secondary winding. A

current limiting resistor R1 is recommended in series with either collector or emitter. Effectively R1 swamps the

UCC24650 internal WAKE pin resistance, R_WAKE. A pull-up resistor R2 from base to emitter is also required, to

ensure that the WAKE pin is adequately pulled up/down during normal switching activity to properly trigger the

internal sample and hold on the VDD pin. The external PNP device Q1 must have at least the same voltage

rating as the main rectifier diode.

V_OUT

EMC

Filter

V_AC

R2

R1

Q1

UCC24650

WAKE VDD

5

1

GND

2

UCC28633

1 VSENSE

SD

HV 8

2

t^o

3 CS

VDD 6

DRV 5

4 GND

Figure 40. Augmented UCC24650 Secondary-Side Voltage Monitor and Wake-Up Circuit

47

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8.3.21 Mode Control and Switching Frequency Modulation

The flyback controller supports applications that require a wide range of operating power levels. This range can

include effectively zero output power in standby conditions, up to a maximum rated continuous power, and then

beyond this, to a mode of peak operating power for a limited time. The modulator operates in multiple modes to

support these power requirements in an efficient way. In some regions, the modulator operates in AM mode at

fixed frequency, where the device adjusts the amplitude of the peak current to regulate the output. In other

regions, the modulator operates in FM mode at fixed peak current, where the device adjusts the switching

frequency to regulate the output. By adjusting only peak current or frequency, (depending on operating region)

the control loop smoothly regulates the power flow of the power stage. The shape of the modulator gain curve

helps counteract the increasing power stage gain as load is decreased.

In the high-power region of the modulator, the device adjusts both peak current and frequency together, to allow

higher power delivery with a modest increase in peak current. In this high-power region, the power stage typically

transitions into continuous-conduction mode (CCM), particularly at low line. The combination of up to 2×

frequency increase and 1.25× peak current increase in CCM allows up to 2× peak power delivery capability for a

given transformer size. Figure 41 provides details regarding the modulator peak current (in mV at the CS pin)

and switching frequency variation vs power demand level. The frequency adjusts from a minimum of 200 Hz up

to a maximum of 120 kHz. The peak-current sense voltage at the CS pin varies from 172 mV to 800 mV. Table 6

summarizes the modulator breakpoints and corresponding percentage power levels.

800

150

640

120

Frequency

(kHz)

V_CS(pk)

(mV)

400

60

30

170

0

0.2

0%

P0

12.5%

P1

30%

P2

45%

P3

70%

P4

100%

P5

Control Loop

Demand Level

Approximate

Power Level

0.025%

P_NOM

3.5%

P_NOM

20%

P_NOM

40%

P_NOM

100%

P_NOM

>=200%

P_NOM

Figure 41. Modulator Modes and Frequency Variations with Power Level

48

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For no load and very light loads (P₀to P₁region) the modulator operates in a pulse frequency modulation (PFM)

mode. In PFM mode, the device maintains a constant peak current in the transformer magnetizing inductance, so

that the energy transferred in each switching cycle is fixed. The magnetic sensing, fixed-point sampling scheme

requires that the device always imposes a minimum peak current. This minimum peak-current demand naturally

results in a minimum transformer magnetizing volt-second product that the device maintains across the input line

voltage range. Ensuring a minimum on-time magnetizing volt-seconds also ensures a balancing volt-second

flyback interval, during which the device guarantees the availability of the output voltage sample. Magnetic

Sensing: Power Stage Design Constraints outlines the transformer design constraints necessary to comply with

the minimum on-time and minimum required volt-seconds.

In the P₀to P₁region, the energy transfer per switching cycle is maximized, which in turn minimizes the switching

frequency and associated switching and drive losses, to improve efficiency. However, due to concerns about

audible noise in this region, the peak current V_CS(min)in this region is limited to 22% of the peak V_CS(max)at the

maximum demand level. This peak-current derating maintains the transformer peak flux density to 22% of the

peak, to minimize transformer-induced audible noise. Assuming a maximum peak flux density of typically 300 mT

at highest peak current, this derating sets the peak flux level at approximately 65 mT in the light-load region.

Empirically, this flux level greatly reduces magnetic audible noise for a variety of power levels and transformer

designs. In this region, the use of sleep modes (where most of the device internal blocks are powered down in

between switching cycles) minimizes the controller power consumption. Minimizing controller power consumption

helps reduce total standby power consumption, and also greatly eases the bias design constraints.

For higher loads above P₁(P₁to P₂region), the device fixes the modulator frequency at a low value above the

audible range, while the peak switch current ramps up from the minimum level, to deliver the increased output

power. Maintaining a fixed low-switching frequency while ramping peak current, minimizes switching losses to

provide good light-load efficiency.

For higher loads above P₂(P₂to P₃region), the device maintains a constant peak-switch current, while the

modulator frequency ramps to its nominal operating value. The normal heavy load (between 40% and 100% of

rated) operating power range lies between P₃and P₄. In this region the device maintains a constant switching

frequency at the nominal value f_SW(nom), and the peak switch current ramps to achieve increased output power.

Fixed-frequency operation at nominal operating power results in consistent EMI and transient load step

performance.

Table 6. Frequency and Peak-Current Modulator Operating Ranges and Breakpoints

MODULATOR

BREAKPOINT

DEMAND LEVEL APPROXIMATE

V_CSPEAK

FREQUENCY f_SW

(kHz)

(%)

POWER

LEVEL % of

P_NOM

(mV)

P_O0

P_O1

P_O2

P_O3

P_O4

P_O5

0

12.5

30

0.025

3.5

172

400

640

800

V_CS(min)

V_CS(nom)

V_CS(bcm)

V_CS(max)

0.200

30

f_SW(min)

f_SW(LL)

20

30

f_SW(LL)

45

40

60

f_SW(nom)

f_SW(max)

70

100

> 200

60

100

120

49

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The peak-power range lies between P₄and P₅. In this region the transformer can operate in CCM depending on

loading and line voltage. By increasing the frequency appropriately, higher average input current can be

processed for the same peak current, so the transformer size does not need to increase substantially for a high-

rated transient peak power. The modulator does, however, also increase the peak current in this region of

operation, requiring a modest increase in transformer size, but this allows a larger transient peak power to be

delivered. The modulator control loop adjusts both the frequency and peak current according to the power

demand so that the increased frequency and peak current meets the load demand.

Figure 42 shows the modulator gain curve, specifically the non-linear modulator gain vs load. At very light loads,

the modulator gain remains low, to help counteract the effect of the higher power stage gain as the load

resistance increases. This low gain helps stabilize the magnetic regulation loop in the light load territory, where

the output voltage sample rate drops with decreasing switching frequency. At heavier loads, the modulator gain

progressively and smoothly increases to help improve transient response. When the switching frequency

increases above the maximum magnetic sense sample rate (f_SMP(max)), the magnetic sense voltage control

sample rate is clamped.

>=200%

P_NOM

100 V

380 V

75 V

100%

P_NOM

Linear Gain

DCM

Operation Up

to Prated

Peak Power Region

œ DCM/CCM

Operation (Line

dependent)

Control Loop

Demand Level

0%

25%

70%

100%

50%

Approximate

Power Level

0.025%

P_NOM

14%

P_NOM

47%

P_NOM

100%

P_NOM

>=200%

P_NOM

Figure 42. Modulator Gain Curves vs Bulk Capacitor Voltage

50

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8.3.22 Frequency Dither For EMI (except UCC28632)

To help ease EMI compliance of the system, the device dithers the switching frequency over time. This dithering

of frequency is active only above the light-load region threshold (P_LL(%)) point on the modulator curve. In the light

load regions, frequency dither is disabled. The frequency dither follows a repeating pattern, in the sequence:

{(f_NOM), (f_NOM+ 6.7%), (f_NOM+ 6.7%), f_NOM), (f_NOM– 6.7%), (f_NOM– 6.7%), (f_NOM), . . . .}

The controller dwells at each frequency for 1 ms. The pattern repeats every 6 ms, as shown graphically in

Figure 43.

NOTE

The device always dithers frequency between 6.7% and –6.7% at every operating point in

the modulator. The dither frequency delta is not an absolute delta, it scales with actual

operating frequency, depending on the exact operating point value.

(f_NOM+ 6.7%)

6 ms

f_NOM

(f_NOM- 6.7%)

Figure 43. Frequency Dither Pattern Details

In order to balance the power flow and reduce and output ripple as a consequence of frequency dithering, the

device automatically adjusts peak-current demand in inverse-proportion to the square-root of the frequency dither

deviation. Thus, since the power flow (in DCM) is given by (½ × L × I²× f_SW), this balances the power flow, and

cancels the output ripple as a consequence of frequency dithering.

51

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8.4 Device Functional Modes

8.4.1 Device Internal Key Parameters

The application designer requires some key device internal parameters in order to calculate the required power

stage components and values for a given design specification . Table 7 summarizes the key parameters.

Table 7. Key Internal Device Parameters

PARAMETER

DESCRIPTION

VALUE

80

UNIT

V_AC

ms

Minimum AC mains input RMS voltage to allow initial start-up, or restart, UCC28630,

UCC28631, UCC28632, UCC28633

AC_ON

Minimum AC mains input RMS voltage to allow initial start-up, or restart, UCC28634

68

Minimum AC mains input RMS voltage below which PWM stops, UCC28630,

UCC28631, UCC28632, UCC28633

65

AC_OFF

Minimum AC mains input RMS voltage below which PWM stops, UCC28634

58

Delay time for which AC mains must remain below AC_OFFlevel to disable PWM, i.e.

brownout delay time

t_UV(delay)

40

Delay time in sleep mode before restart is initiated – applies to AC_UV, X-capacitor

discharge responses

t_RESET(short)

t_RESET(long)

f_SW(uv)

500

1,000

15

ms

kHz

µs

Delay time in sleep mode before restart is initiated – applies to all other auto-restart

faults

Switching frequency used during initial 3-cycle exploratory pulses for AC_ONdetection

at start-up

Maximum on-time used during initial 3-cycle exploratory pulses for AC_ONdetection at

start-up

t_{ON(max_uv)}

2.3

K_LINE

Device internal line sense gain factor

49.25

44.5

69.5

16

K_CC1

Device internal CC mode gain factor

K_CC2

Device internal CC mode offset factor

f_SMP(max)

V_{DD(latch_hi)}

V_{DD(latch_lo)}

t_ON(hv)

Maximum magnetic sense sample rate; in effect when f_SW> f_SMP(max)

Upper VDD regulation level during latched fault mode

Lower VDD regulation level during latched fault mode

HV current source on-time during X-capacitor sampling

HV current source sample repetition rate during X-capacitor sample burst

HV current source wait-time between X-capacitor sampling bursts

Light-load region threshold as % of P_NOM

kHz

V

10

8

V

20

µs

ms

t_SMP(hv)

1

t_WAIT(hv)

P_LL(%)

200

12.5%

1.0

100

3

V_DD(sc)

VDD short-circuit threshold below which charging current is limited

Internal PWM comparator + latch + gate driver aggregate delay

Internal start-up initialization delay

V

ns

ms

V

t_PROP(gate)

t_START(del)

V_SENSE(wake)

VSENSE pin wake threshold for fast transient response (UCC28633 only)

0.8

52

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9 Applications and Implementation

9.1 Application Information

The UCC2863x device is a highly integrated primary-side-regulated (PSR) flyback controller, supporting

magnetically-sensed output voltage regulation via the transformer bias winding. This sensing eliminates the need

for a secondary-side reference, error amplifier and opto-isolator for output voltage regulation. The device delivers

accurate output voltage static load and line regulation, and accurate control of the output constant-current limit.

The fixed-point magnetic sampling scheme allows operation in both continuous conduction mode (CCM) and

discontinuous conduction mode (DCM). The combination of the sampling scheme and high current gate driver

source and sink capability, makes this device ideal for high power flyback converters up to 100 W and beyond.

The modulator adjusts both frequency and peak current in different load regions to maximize efficiency

throughout the operating range. The control approach improves performance (efficiency, size and cost) and can

reduce transformer size and cost by allowing operation in CCM with FM during peak overload conditions. The

modulator supports peak-to-average transient overload power up to 200% of the nominal average rating.

9.2 Typical Application

9.2.1 Notebook Adapter, 19.5 V, 65 W

This design example describes the PWR572 EVM design and outlines the design steps required to design a

constant-voltage, constant-current flyback converter for a 19.5-V/65-W notebook adapter. For all equations and

design steps, refer to Table 7 for definitions and values of key internal device parameters that are relevant for

calculations of external component values.

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Typical Application (continued)

9.2.2 UCC28630 Application Schematic

5

4

3

2

3

° t

~

2

3

2

3

1

4

2

1

Figure 44. Typical Application Circuit for 19.5-V / 65-W Adapter

54

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Typical Application (continued)

9.2.3 Design Requirements

Table 8. Design Requirements

DESIGN PARAMETER

TARGET VALUE

Output voltage

19.5 V

Rated (continuous) output power

Peak (transient) output power

Peak (transient) output power duration

Input AC voltage range

65 W

130 W

2 ms

88 V_RMSto 264 V_RMS

88%

Typical efficiency

Minimum bulk voltage at 88 V_AC/47 Hz and rated (continuous) output power

82 V

9.2.4 Detailed Design Procedure

9.2.4.1 Custom Design With WEBENCH® Tools

Click here to create a custom design using the UCC2863x device with the WEBENCH® Power Designer.

1. Start by entering the input voltage (V_IN), output voltage (V_OUT), and output current (I_OUT) requirements.

2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.

3. Compare the generated design with other possible solutions from Texas Instruments.

The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time

pricing and component availability.

In most cases, these actions are available:

•

Run electrical simulations to see important waveforms and circuit performance

Run thermal simulations to understand board thermal performance

Export customized schematic and layout into popular CAD formats

Print PDF reports for the design, and share the design with colleagues

Get more information about WEBENCH tools at www.ti.com/WEBENCH.

9.2.4.2 Input Bulk Capacitance and Minimum Bulk Voltage

The required bulk capacitance value depends on the target minimum bulk capacitor ripple voltage at minimum

AC input line, minimum line frequency and on the power level of interest. As a way of estimating, use 1.5-μF to

2-μF per Watt of rated, continuous power to achieve approximately 70 V to 80 V minimum at 88 V_RMSinput. This

case indicates a required bulk capacitance of between approximately 100 μF and 130 μF. Alternatively, the

required capacitance may be explicitly calculated for a specific set of requirements using Equation 21.

6_{"5,+ ≠©Æ}

0_/54

D

1

:

;

× Lꢀ.ꢁꢂ × ≥©Æ^-1

F

GM

N

2 × 6

¾

:

;

!# ≠©Æ

#

"5,+

=

k2 × 6_!²_{# ≠©Æ}_;F 6²

o × ¶_{,)ꢃ% ≠©Æ}

: ;

:

;

"5,+ ≠©Æ

(21)

Using the parameters in Table 8, this calculates a required C_BULKof 130 μF.

To help reduce differential mode (DM) emissions for conducted EMC compliance, the bulk capacitance has been

split into two separate capacitors C5 and C7 in Figure 44, with a small DM choke L2 inserted between the

capacitors. The total resulting capacitance of 127 μF is close to the required minimum requirement per

Equation 21, and the design results in a small decrease in the actual bulk capacitor minimum ripple voltage.

Next, verify that the choice of bulk capacitance satisfies the X-capacitor discharge constraints for rate of

discharge by the load when X-capacitor sampling is inactive, per Equation 5. The bulk capacitance should be

less than the value calculated by Equation 22.

55

UCC28630, UCC28631

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ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

www.ti.com.cn

0

2 @ ^./-A× 0_,,(%)× ¥_$ꢀ3

:

;

2 ꢂꢃ × ꢄꢅ12ꢃ × 1

D

#_"5,+Q

=

Lsux J&

ꢆ7ꢆ F ꢂꢄ²

2

k6_{!# ∞´}F 6 ²o

:

;

:

;

3ꢁ,6

(22)

56

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ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

9.2.4.3 Transformer Turn Ratio

Choose the transformer primary-to-secondary-side turns ratio based on the allowed voltage stress for the output

rectifier, or the primary MOSFET. For 19.5-V charger designs, it is valid to choose a turns ratio that allows the

use of a more efficient 100-V Schottky rectifier.

6_{!# ∞´_≠°∏}

.

:

;

0

=

:

;

k6_{2%6 ≤°¥•§}_;× ꢀ$•≤°¥©Æßo F 6_/54+ 6_2%#4

3

:

(23)

For a good Schottky diode with 100-V reverse rating, V_REV(rated), the rectifier forward voltage drop, V_RECT, can be

expected to be in the range of 0.4 V to 0.5 V at 3 A to 5 A, at practical operating temperatures in the region of

100°C. Allowing an 85% derating on the rectifier reverse voltage stress, Equation 23 indicates a required turns

ratio of 5.734 for a maximum AC peak voltage of 373 V (264 V_RMS).

Choose the bias winding turns ratio to set the nominal bias voltage for the device VDD pin. Use an initial

V_BIAS(target)of 12 V.

6_{")!3 ¥°≤ß•¥}_;+ 6

.

:

&

"

=

:

;

6_/54+ 6_2%#4

3

where

•

V_Fis the forward voltage drop of the rectifier on the bias winding.

(24)

For a typical 0.7-V bias-diode drop, this equation calculates to 0.6366.

When the transformer size and type are chosen, the actual turns values can be calculated. Because the turns

need to be rounded to integer values, the actual turns ratios achieved deviates from these targets. Check the

final ratios to ensure that the secondary-side Schottky rectifier stress and the bias winding nominal level are

acceptable. Adjust the specific turns counts to meet the target ratios.

9.2.4.4 Transformer Magnetizing Inductance

Match the power stage design to the modulator curves by ensuring that the boundary conduction mode (BCM –

boundary of operation between DCM and CCM) point coincides with the minimum bulk-capacitor voltage at

minimum line, at rated output power. This choice results in DCM operation at all line voltages for all loads up to

continuous rated load, and minimizes power loss and EMC impacts due to output rectifier reverse recovery

during CCM operation. This design choice allows operation to extend into the CCM region of operation as

required to deliver the transient peak load.

To achieve this design target, the required primary magnetizing inductance, L_PRIis calculated from Equation 25.

In this equation, the value of F_SW(nom)is 60 kHz, taken from the modulator curve region P₃to P₄, in Table 6. The

value of V_BULK(min)is the value that occurs with the actual used bulk capacitance of 127 μF.

1

,

02)

=

ꢀ

0

1

:

1

ꢀ × @ ^2!4%$A × L

ꢁ

M × ¶_{37 ÆØ≠}

:

;

.

0

3

D

6_{"5,+ ≠©Æ}

;

:

;

× 6_/54ꢁ6_2%#4

.

(25)

This calculates a value of 257 μH. Round the value to 260 μH.

57

UCC28630, UCC28631

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ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

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9.2.4.5 Current Sense Resistor R_CS

In addition to choosing L_PRIvalue to map the rated power to the target BCM point at minimum bulk voltage,

choose the R_CSvalue according to Equation 26. This calculation ensures that the resulting peak current, in

conjunction with the chosen value of magnetizing inductance, and the 60-kHz modulator frequency, delivers the

required input power to meet the rated output load power, at minimum bulk voltage ripple.

6_{#3 ¢£≠}

:

;

2_#3

=

0

1

: ;

1

ꢀ × @ ^2!4%$A × L

ꢁ

M

.

0

3

D

6_{"5,+ ≠©Æ}

:

;

× 6_/54ꢁ 6_2%#4

.

where

•

V_CS(bcm)is the modulator peak-current sense level at point P₄(640 mV)

(26)

This equation calculates a value of 207 mΩ. Use the nearest standard E24 value of 200 mΩ.

9.2.4.6 Transformer Constraint Verification

As outlined in Magnetic Sensing: Power Stage Design Constraints, there are constraints on the ratio of R_CS/L_PRI

to ensure the design is consistent with the required volt-seconds for output sampling at minimum load, and with

the controller t_ON(min)at high line. Per Equation 15 and Equation 16, limit the ratio of R_CS/L_PRI

.

6_{#3 ≠©Æ}

2_#3

.

1

17ꢀ ≠6

säy J≥

ꢁ

ꢂꢃ

1

:

;

3

Q

×

=

×

ꢀ {rr

:

;

:

;

,

02)

¥

.

0

6_/54+ 6_2%#4

19ꢄꢅ 6 + ꢆꢄꢃ 6

:

;

/54 ≥≠∞

(27)

(28)

and,

6_{#3 ≠©Æ}

2_#3

1

6

17ꢀ ≠6

ꢁ7ꢁ 6 × ꢂꢃꢄ J≥

:

;

Q

×

=

ꢀ yyr

,

02)

¥

:

;

:

/. ≠©Æ

). ∞´ _≠°∏

In this case, the ratio equates to 769, so both constraints are met.

58

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9.2.4.7 Transformer Selection and Design

After determining the value of current sense resistor R_CS, determine the maximum peak current at maximum

demand point on the modulator. Accommodate for the I_PEAKadjustment for frequency dithering. Use this value

when calculating the margin for core saturation. In this case, I_PK(sat)calculates to 4.13 A.

6_{#3 ≠°∏}× 1ꢀꢁ.7%

¾

:

;

)

=

;

:

0+ ≥°¥

2_#3

(29)

In subsequent calculations of required primary turns etc, the average maximum peak current, I_PK(max), during the

frequency dither period should be used, which calculates to 4.0 A.

6_{#3 ≠°∏}

:

;

)

=

;

:

0+ ≠°∏

2_#3

(30)

Knowing I_PK(max), L_PRIand the turns ratio, the choice of transformer size and core shape and type dictates the

required number of primary, secondary and bias turns, and the size of the air-gap. Various trade-offs, design

preferences, and transformer design targets (size, cost, target losses, etc.) influence the specific choice of

transformer core in any given design.

In the case of the UCC28630EVM-572 (PWR572 EVM), core area-product geometry was used to choose the

minimum core size available to meet the power level. The core geometry factor K_gis a figure-of-merit that

reflects the core power capability, in terms of its physical size, shape and design. It combines the core effective

cross-sectional area, A_e, winding window area, A_w, and the mean length per turn (MLT) of wire around the core.

!

²× !₇

-,4

•

+_'=

(31)

Estimate the required design core geometry, K_G(des), using the required transformer inductance L_PRI, maximum

peak current I_PK(max), allowed maximum core flux density B_maxand a target copper loss budget, P_CU

.

ꢀ

,

02)

^ꢀ× )_{0+ ≥°¥}× )_4/4× O_#5

:

;

+

=

;

:

' §•≥

ꢀ

"

≠°∏

× +₅× 0

#5

where

•

ρ_cuis the resistivity of Copper (approximately 1.7 × 10^-8Ωm at room temperature, 2.2 × 10^-8Ωm at 100°C),

K_uis a winding window utilization factor that accounts for the percentage of the window that is occupied by

Copper

(32)

K_ucan often be as low as 25%, due to the fill factor (gaps between wires), wire insulation (especially for triple-

insulated wire), and the need for insulating tapes and EMC shielding layers. The estimate of the required core

geometry needs an estimate of the aggregate total winding current I_TOT. The analysis models the flyback

transformer primary and secondary windings as a single lumped non-isolated inductor (such as a single winding

buck inductor), only for the purpose of sizing the required core winding window to achieve the target copper loss.

In this case, the secondary-side current amplitude reflects to the primary side so that aggregate total primary

current. I_TOTcan be estimated in Equation 33.

§

^ꢁ%#W

§

W ꢀ )₀₊×

)_4/4= )₀₊

×

3

where

•

d is the primary on-time duty cycle

d_SECis the secondary-side flyback period duty cycle

(33)

At rated power and minimum bulk capacitor voltage, the inductance L_PRIhas been chosen to achieve boundary-

mode conduction, therefore the duty cycle is given in Equation 34.

.

0

.

3

:

;

× 6_/54+ 6_2%#4

§ =

.

0

.

3

:

;

G

F6_{"5,ꢀ ≠©Æ}

+

;

× 6_/54+ 6_2%#4

:

(34)

(35)

and

§_3%#= 1F§

59

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At the boundary conduction point, the primary peak current I_PKis at the level set by the modulator, V_CS(bcm). So

from Equation 33, I_TOTbecomes Equation 36.

Í

Î

Ï

Ð

Ñ

Ò

.

0

:

;

× 6_/54+6_2%#4

6_{#3 ¢£≠}

6_{"5,ꢁ ≠©Æ}

.

3

:

;

:

;

)

=

×

+

4/4

ª

©

ꢀ2

¾

.

0

#3

0

:

;

:

;

G

F6_{"5,ꢁ ≠©Æ}

+

× 6_/54+6_2%#4

G

F6_{"5,ꢁ ≠©Æ}+

;

× 6_/54+6_2%#4

:

;

:

.

3

(36)

Equation 36 calculates I_TOTas 2.6 A. Thus the required design K_G(des), assuming K_Uof 25%, B_maxof 315 mT and

a target of 1-W copper loss, is shown in Equation 37.

txr J(²× 4.13²× 2.6²× 2.2 × 10^-8

+

=

;

:

' §•≥

0.315²× 0.25 × 1.0

(37)

Equation 37 indicates that this design requires a core size and shape with a K_Gof more than 6.9 × 10^-12. A

review of commonly used cores indicated that the RM10/I core set meets this requirement. With A_eof 96.6 mm²,

A_wof 44.2 mm²and mean length per turn (MLT) of 52 mm, K_G(RM10)is 7.9 × 10^–12, giving some margin over the

design target.

(96.6 × 10^ꢀ6)^ꢁ× (44.ꢁ × 10^ꢀ6)

+

=

;

= 7.93ꢁ × 10^ꢀ1ꢁ

:

' 2-10

(5ꢁ × 10^ꢀ3)

(38)

With the chosen core, the actual primary, secondary-side and bias turns can be calculated. The required primary

turns depend on the allowed B_max. For most power ferrites, a value in the region of 315 mT is commonly used.

,

₀₂₎× )_{0+ ≠°∏}

txr J × 4ꢀꢁ

ꢁꢀ315 × 96ꢀ6 J

: ;

.₀=

=

= 34ꢀ18

"

≠°∏

× !_•

(39)

Round N_Pto 34. Now the required secondary-side turns can be calculated, using the previously calculated turns

ratio per Equation 23.

.

5ꢀ7ꢁ4

0

.₃=

= 5ꢀ9ꢁ

(40)

Again, N_Sis rounded to 6. Due to the integer rounding of the turns count, ensure that the actual turns ratio is

within 5% of original target (if outside this range, secondary-side rectifier or primary MOSFET stress may be too

high).

.

0

.

5ꢀ7ꢁ4

3

= 98ꢀ8%

(41)

(42)

From Equation 24, the required bias turns can be calculated using Equation 42.

6_{")!3 ¥°≤ß•¥}_;+ 6

1ꢀ + 0ꢁ7

:

&

._"=

× .₃=

× ꢄ = ꢅꢁ8ꢀ

:

;

6_/54+ 6_2%#4

19ꢁꢂ + 0ꢁꢃꢂ

Again, N_Bis rounded to 4. The effect of integer scaling in the turns is verified by calculating the expected bias

voltage versus target.

.

ꢂ

ꢃ

"

:

;

: ;

F 6 = 19ꢀꢁ + 0ꢀꢂꢁ × F 0ꢀ7 = 1ꢄꢀꢃ 6

&

6_")!3= 6_/54+ 6_2%#4

×

3

(43)

The V_BIAStarget was 12 V, so this is acceptable.

The required core inductance factor, A_L, to achieve the target inductance can be calculated as in Equation 44.

The transformer manufacturer uses this factor to gap the core center leg.

,

txrJ

34^ꢀ

02)

ꢀ

!_,=

=

= ꢀꢀ5 Æ(

.

0

(44)

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Finally, calculate the required air-gap length l_g, based on the required inductance and the core geometry.

.

0

²× J_ꢀ× !_#%.4ꢁ%

¨

≠

¨_ß=

F

,

0ꢁ)

J_≤

where

•

μ₀is the permittivity of free-air

μ_ris the relative permeability of the chosen core ferrite material

A_CENTREis the cross-sectional area of the core center leg

l_mis the core average magnetic path length

(45)

For the RM10/I core in 3C95 material (chosen for low core loss over a wide temperature range), the required air-

gap length is calaulated using Equation 46.

34²× vN × 10^-7× 93.3 J 44.6 ≠

¨_ß=

F

L wsv J≠

txr J

5500

(46)

Typically, the air-gap calculation in Equation 45 underestimates l_g, due to flux fringing in the air-gap. The fringing

causes the affective area of the air-gap A_gto be somewhat larger than the ferrite core center leg A_CENTRE

,

depending on the gap length. This difference requires an increase in the required air-gap length to get the

required inductance, which results in a further increase in fringing. However use Equation 45 to determine an

initial value for l_g, which can then be used to estimate A_g. For round centre legs, the increase in effective area

within the gap can be estimated empirically from Equation 47

ꢀ

¨

0ꢁ51ꢂ

10ꢁ9

ß

!_ß= !_#%.42%× F1+

G

L {uäu J × l1+

p = 10ꢀꢁ31 ≠≠^ꢀ

$

#%.42%

where

•

D_CENTREis the center leg diameter

(47)

(For more information about this subject, download the paper Inductor and Flyback Transformer Design, Lloyd

Dixon, TI Power Supply Design Seminar SLUP127).

Because Equation 45 assumes that A_gequals A_CENTRE, it must be modified using Equation 48.

2

.

0

× J_ꢀ× !_ß

!

¨

G F l ^≠

J_≤

¨_ß=F

×

^ßp

,

0ꢁ)

•

(48)

(49)

Re-iterating the air-gap calculation in Equation 49 .

34²× vN × 10^F7× 102.31 J

44.6 ≠ 102.31 J

G F l p L wxuär J≠

¨_ß= F

×

260 J

5500

96.6 J

Typically, after the second iteration above in Equation 48, the estimated air-gap is very close to the required

value. Further iterations can be made, but should not be necessary.

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9.2.4.8 Slope Compensation Verification

After choosing the current sense resistor, transformer inductance and transformer turns ratio, verify the required

slope compensation against the fixed internal slope compensation. The worst case slope compensation

requirement always occurs at the highest duty cycle operating point (at minimum bulk voltage level).

For stability, the slope compensation should be at least 50% of the difference between the inductor up-slope and

down-slope. [reference Bob Mammano TI Power Supply Design Seminar paper, 2001, SLUP173]. For a flyback

converter, the difference in slopes in CCM is equal the operating duty cycle multiplied by the inductor current

down-slope value. For example, for 50% d_BULK(min)at minimum bulk capacitor voltage, the required slope

compensation ramp is 25% of the inductor current down-slope.

As listed in Table 8, the specified peak-load transient is 130 W for 2 ms. In a worst case, peak transient timing

with respect to the AC phase, the V_BULKminimum level dips to 65 V. This corresponds to a duty cycle of

approximately 63.5% according to Equation 50.

.

0

:

;

× 6_/54ꢀ 6_2%#4

¤

:

;

ꢁꢂ ꢃ × 19ꢄ9ꢅ

.

3

§

=

;

=

= ꢃꢁꢄꢅꢆ

:

"5,+ ≠©Æ

:

¤

:

;;

ꢃꢅ ꢀ ꢁꢂ ꢃ × 19ꢄ9ꢅ

.

0

:

;

G

F6_{"5,+ ≠©Æ}

ꢀ

;

× 6_/54ꢀ 6_2%#4

:

3

(50)

(51)

The required slope compensation ramp is calculated at 63.5% duty cycle.

.

ꢁ

3

:

;

× 6_/ꢀ4ꢂ 6_2ꢃ#4

¤

ꢇꢉ ꢆ × 19ꢅ95

.

r= 0ꢅ5 × 0ꢅꢆꢇ5 × 0ꢅꢈ × F

G = ꢈ7ꢅꢆ ≠6ꢊJ≥

:

,

ꢁ2ꢄ

txr J

This value is within the 30 mV/μs of internal slope compensation provided by the controller.

9.2.4.9 Power MOSFET and Output Rectifier Selection

The initial design target proposed the use of a 100-V Schottky rectifier. The secondary-side reverse voltage

stress can be verified using the final transformer design sh own in Equation 52.

.

ꢀ

ꢁꢂ

3

:

;

=

6_{2%#4 ≤•∂}_;= l × 6_{!# ∞´_≠°∏}_;p + 6_/54+ 6_2%#4

× ꢁ7ꢁ + 19ꢃ9ꢄ = 8ꢄꢃ8 6

:

.

0

(52)

The value derived from Equation 52 is close to the original design target of 85 V.

For 65-W load, the average DC output current is 3.35 A for 19.5-V output. However, to reduce losses, a much

higher current rated diode is typically used, to yield a much lower forward voltage drop V_RECT. As shown in

Figure 44, a 30-A rated diode D7 is used in this case, with a forward drop of approximately 0.45 V at 3.5 A,

100°C.

For the primary-side MOSFET, the peak voltage stress can be estimated using Equation 53.

.

ꢀꢁ

ꢂ

0

:

;

6_{$3 ≠°∏}_;= 6

+

;

!# ∞´ _≠°∏

× 6_/54+ 6_2%#4= ꢀ7ꢀ +

× 19ꢃ9ꢄ = ꢁ8ꢂ 6

:

3

(53)

An allowance of at least 100 V must be added to this figure to account for the leakage inductance spike at turn-

off. This voltage spike depends on the transformer implementation and the amount of leakage inductance, as

well as the specific design of the snubber. A more aggressive snubber may reduce the voltage spike, but at the

expense of higher losses in the snubber. A voltage rating of at least 600 V is recommended for the power

MOSFET to allow for leakage.

The MOSFET rms current at low line, rated load, can be estimated using Equation 54.

.

0

.

3

:

;

× 6_/54+ 6_2%#4

6_{#3 ¢£≠}

ꢂꢃꢄꢅ

:

;

)

=

;

×

=

× ¾ꢂꢃꢇ8 = 1ꢃꢅ1 !

:

02) ≤≠≥

ª

ꢀ2

¾

ꢀ®ꢂꢃꢆ

¾

.

#3

0

:

;

G

F6_{"5,ꢁ ≠©Æ}

+

;

× 6_/54+ 6_2%#4

:

3

(54)

As can be seen in Figure 44, the chosen MOSFET Q1 is a 13-A, 600-V device.

62

UCC28630, UCC28631

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www.ti.com.cn

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

9.2.4.10 Output Capacitor Selection

Select the output capacitor value on the basis of one of the following, depending on which one is the limiting

factor:

•

Required ripple current rating to absorb the high secondary-side peak current

Required esr to achieve a target peak-peak ripple voltage

Required holdup capacitance to achieve target minimum output voltage for a specified load transient from no

load when the device is switching at f_SW(min)

For flyback converters, ripple current rating often dictates the output capacitance value. The required ripple

current rating can be calculated from Equation 55.

ꢀ

ꢂ

3

Ç

È

Ê

Ë

6_{#3 ¢£≠}

6_{"5,+ ≠©Æ}

.

:

;

:

;

0

ꢂ

)

=

;

×

F )_/54

:

#!0 ≤≠≥

2_#3

.

3

ª

.

0

: ;

× 6_/54ꢁ 6_2%#4Gq

:

;

.

3

É

Ì

ã

(55)

At rated 65-W load, I_CAP(rms)= 5.9 A_RMS. Capacitors C11 and C13 in Figure 43 are chosen to meet this ripple

requirement, (each capacitor has a 2.5-A minimum rating at 105°C). Total output capacitance is 1360 μF.

9.2.4.11 Calculation of CC Mode Limit Point

Calculate the expected output constant-current (CC) limit point from Equation 20. As previously noted, K_CC1is

44.5 and K_CC1is 69.5. Thus, I_OUT(lim)in this case is approximately calculated in Equation 56.

1

2_#3

.

+

1

ꢂꢃꢀ

ꢄꢅ

ꢆ

ꢅꢅꢃꢇ

0

##1

)

=

;

×

=

×

= 7ꢃꢂ !

:

/54 ¨©≠

.

3

ꢄꢅ

ꢆ

.

3

0

.

+

ꢁ 6_/54

×

ꢆ9ꢃꢇ ꢁ @19ꢃꢇ ×

A

##ꢀ

(56)

63

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9.2.4.12 VDD Capacitor Selection

Size the VDD capacitor to supply sufficient I_DD(run)current to the device during initial start-up, and also during the

charging phase of the main output capacitors. During the charging phase the bias winding on the transformer

must supply the bias power. When VDD reaches the V_DD(start)threshold, the device consumes I_DD(run)for t_START(del)

before the PWM switching commences. Thereafter, the bias current is the device current plus the MOSFET gate

current. The VDD capacitor must support this higher level of current until the output is sufficiently charged that

the bias winding rail has increased above the V_DD(stop)level.

Calculate the required bias capacitance from the total bias charge associated with the device run current during

the t_START(del)phase, plus the device run current during the output charge phase, plus the primary MOSFET gate

charge current during the output charge phase. The time taken for the output charge phase to reach a sufficient

level to supply the bias can be calculated from the size of the output capacitor, target output regulation voltage,

and the difference between the available CC mode current limit and the maximum load current (assuming that

the output capacitor has to be charged whilst also supplying full rated load current). Assume that the MOSFET is

switched at 60 kHz throughout the charging phase.

Combining these into one equation, the required VDD capacitor can be calculated as shown in Equation 57.

6_{$$ ≥¥Ø∞ _≠°∏}

6_/54× #_/54

:

k)_{$$ ≤µÆ}_;× ¥

_;o + k)_{$$ ≤µÆ}_;+ &_{37 ÆØ≠}_;× 1_{ß ¥Ø¥}_;o × l

p × l

^;p

:

34!24 §•¨

)

_;F )_{/ ≠°∏}

6_{")!3 ÆØ≠}

:

;

:

;

/54 ¨©≠

#

6$$

=

_;g

:

$$ ≥¥Ø∞ _≠°∏

(57)

This can be re-written with the explicit device values substituted:

6_/54× #_/54

8.ꢁ

6_{")!ꢂ JKI}

;

:

;

8≠ × 3≠ +k8≠ + ꢀ0 ´ × 1_{ß PKP}_;o × l

p × l

p

:

)

_;F )_{/ I=T}

:

: ;

:

/54 HEI

#

=

6$$

:

;

ꢃ3.0 F 8.ꢁ

(58)

(59)

For this EVM design, the MOSFET Q_g(tot)is 30 nC, V_BIAS(nom)is 12.6 V. I_O(max)is 3.35 A, so this equates to:

19.5 × suxrJ

7.0 F 3.35

8.5

12.ꢀ

:

;

:

;

8≠ × 3≠ + 8≠ + ꢀ0 ´ × 30Æ × @

A × @

A

#

6$$

=

L sxär J&

:

;

13.0 F 8.5

Choose the next higher standard value, 22 μF.

Verify that the bias capacitance is sufficient to absorb all the X-capacitor energy when it has to be discharged,

per Equation 3. From Figure 44, the value of X-capacitor is 330 nF.

6_{!# ∞´}_;F 6

ꢀ7ꢀ F ꢁ0

1ꢀ F ꢁ.5

:

3%,6

#_6$$R #₈× F

G = ꢀꢀ0 Æ& × l

p = 15.9 J&

:

;

$$ ≤•≥•¥ _≠°∏

(60)

64

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9.2.4.13 Magnetic Sense Resistor Network Selection

The required values for the magnetic sense divider network are calculated from Equation 11 and Equation 12.

For the RM10/I transformer used in the PWR572 EVM, the secondary-side to bias leakage inductance was

measured and found to be approximately 4%. This figure can be reasonably estimated as the ratio of the

inductance value measured across the secondary-side pins with the bias pins shorted together (primary winding

should remain open-circuit), to the inductance value measured across the secondary-side pins with all other

windings open:

,

:

;

=

;

:

,

:

;

3ꢀ# ¢©°≥ _Ø∞•Æ

(61)

R_Ais calculated as shown in Equation 62.

.

4

34

2_!= 2₀× l ^"p × +_,).%= 3ꢀ9 ´3 × l p × 49ꢀꢁ5 = ꢁꢁꢀ597 ´3

0

(62)

The nearest standard E96 value 22.6 kΩ is selected. R_Bmay then be calculated to set V_OUTat 19.5 V.

2_!ꢂꢂꢃꢄ ´3

2_"=

=

= ꢇꢂꢃ10 ´3

.

:

;

: ¤ ;

19ꢃꢅ × 1 F 0ꢃ0ꢆ ꢀ 0ꢃꢆꢅ × ꢆ ꢄ

A

:

l

F 1p

.

3

7ꢃꢅ0

L

F 1M

6_{/54 ≤•¶}

:

;

(63)

The nearest E96 value is 32.4 kΩ, which could be used, but results in some set-point regulation error. As shown

in Figure 44, the setpoint may be fine-tuned by using two parallel resistors for R_B. In this case use values of 39

kΩ and 180 kΩ, to give a net equivalent of 32.05 kΩ, very close to the target value in Equation 63.

Note that the pull-up diode to DRV pin should be a standard switching signal diode such as BAS21 or similar.

The reverse recovery of the diode should be 100 ns or less. A slow-recovery diode clamps the VSENSE pin low

for an initial portion of the flyback interval, and may impair or prevent the ability to take a valid output voltage

sample.

9.2.4.14 Output LED Pre-Load Resistor Calculation

As shown in Figure 44, the output power good LED1 and series resistor R18 form an output pre-load or minimum

load. This pre-load is necessary in order to maintain regulation at no load, or when the power converter output is

disconnected from the load system. Magnetic regulation relies on sensing the output voltage during switching

cycles, so it is necessary to maintain a certain minimum switching frequency f_SW(min)in order to continue sensing

the output voltage. However, generating switching cycles at f_SW(min)transfers energy to the output, which requires

some load on the secondary-side to absorb this energy and prevent the output capacitors from being charged out

of regulation. The minimum energy transferred at f_SW(min)depends on the choice of magnetizing inductance L_PRI

and current sense resistor R_CS

.

ꢁ

6_{#3 ≠©Æ}

ꢀ.1ꢃꢁ

ꢀ.ꢁ

:

p × ꢁꢀꢀ = 19.ꢁꢄ ≠7

:

2_#3

(64)

In order to ensure that the control loop operates at a frequency above the minimum switching frequency, f_SW(min)

(to ensure that the loop has adjustment range up/down as required to maintain regulation), the recommended

minimum pre-load is at least twice the value calculated in Equation 64.

The required value of R18 can then be calculated, assuming a forward voltage drop of 1.8 V for the LED:

6_/54^ꢀF 6_/54× 6_,%$19.ꢁ^ꢀF 19.ꢁ × 1.8

2_,%$

=

= 8.97 ´3

ꢀ × 0_:

ꢀ × 19.ꢀ3≠

;

∞≤•¨Ø°§ _≠©Æ

(65)

Use the next lower E24 value of 8.2 kΩ. For a design without an LED, a pre-load resistor of similar value is still

required across the output voltage.

65

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9.2.5 External Wake Pulse Calculation at VSENSE Pin (UCC28633 Only)

The typical application circuit of Figure 39 may be redrawn as a simplified equivalent circuit as shown in

Figure 45. In this equivalent circuit, the capacitor C_Pis the total parasitic capacitance (MOSFET C_oss, transformer

capacitance, etc), and resistance R_WAKEis the effective internal resistance of the UCC24650 WAKE pin to GND

pin when the internal WAKE pull-down is active (see UCC24650 detailed datasheet specifications).

If all the elements on the primary and secondary of the transformer are referred to the bias winding, this can be

further simplified as in Figure 46.

N_P

V_OUT

L_P

N_S

V_BULK

R_WAKE

R_T

C_P

N_B

V_SENSE

R_B

Figure 45. Simplified Equivalent Circuit of Wake Event with UCC24650

(N_B/N_S) x R_WAKE

R_T

(N_B/N_P)²x L_P

+

(N_B/N_P)²x C_P

(N_B/N_S) x V_OUT

V_SENSE

œ

R_B

Z_LC(bias)

Figure 46. Bias-Referred Simplified Equivalent Circuit of Wake Event with UCC24650

66

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Thus, knowing L_Pand C_P, the power stage impedance Z_LC(bias)(reflected to the bias winding) may be calculated

from Equation 66, and the effective wake resistance can be referred to the bias winding using Equation 67. The

wake pulse amplitude can be calculated from Equation 68. If C_Pis not known, it can be measured by observing

the resonant ring period at the primary drain node, T_RES, and calculating C_Pfrom Equation 69. Worst case values

should be used to estimate the worst case minimum wake pulse amplitude at the VSENSE pin. It should also be

noted that any filter cap on the VSENSE pin (including internal parasitic pin capacitance) adds an RC filter in

conjunction with the Thevenin resistance of the VSENSE divider, R_T, R_B; this delays and further attenuate the

wake pulse amplitude. Additionally, the internal wake comparator requires some over-drive to trip, and exhibits

propagation delay that depends on the amount of overdrive. So some margin should be allowed in the wake

pulse amplitude to ensure that the minimum wake pulse can adequately overdrive the internal wake comparator.

A margin of at least 20% over the threshold V_SENSE(wake)is recommended.

2

.

0_$

0₂

<

= ¨ ²× l

p

.%(>E=O )

%

2

(66)

(67)

(68)

(69)

2

0_$

4_{9#-'(>E=O )}= 4_9#-'× l

p

0

5

<

4_$

0_$

p × l8₁₇₆× (1 F ¿_9#-'%) × p × F

.%(>E=O )

8

= l

G

5'05'_9#-'(LG )

4_#+ 4_$

0

5

<_{.%(>E=O )}+ 4_{9#-'(>E=O )}

2

6

2è

1

%₂= l ^NAOp ×

.

2

If the worst case wake pulse amplitude is too low, then the UCC24650 WAKE output can be augmented with an

external PNP circuit Q1, R1 and R2, as shown in Figure 40. This circuit reduces the effective wake resistance to

ground, so that a larger proportion of the output voltage appears across the transformer secondary pins when the

UCC24650 WAKE activates.

Using the UCC28630EVM-572, (TI Literature Number SLUUAX9) circuit parameters from Figure 44, the nominal

wake pulse amplitude at the VSENSE pin can be estimated. Of course, the rectifying diode D7 in Figure 44

would need to be relocated to return end of the secondary winding (pins 10, 11) to allow UCC24650 to be

deployed.

From observation of the DCM ringing period, the period T_RESwas found to be 1.138 μs. From Equation 69, C_Pis

estimated:

2

6

1

1.138ä

2è

1

260ä

%₂= l ^NAOp × = l

2è

p ×

= 126 L(

.

2

(70)

From Equation 66, the power circuit impedance is:

2

.

%

0_$

0₂

260ä

126L

4

2

<

= ¨ × l p = ¨

× l p = 19.9

34

W

.%(>E=O )

2

(71)

The WAKE pin resistance R_WAKEcan be determined form the UCC24650 datasheet; for now a nominal value of

400 Ω is assumed. Referred to the bias winding (scaled by (N_B/N_S)²), this becomes 178 Ω. Similarly Δ_WAKE%can

be determined from the UCC24650 datasheet; for now, a value of 97% is assumed. From Equation 68, the wake

pulse amplitude can be calculated:

≈

∆

«

’

÷

◊

Z_LC

≈

’ ≈

’

R_B

N_B

N_S

(

bias

)

∆

÷ ∆

÷

V_SENSE

=

ì V_OUT

ì

(

1- D_{W AKE%}

)

ì

WAKE ( pk )

÷ ∆

R_A+ R_B

Z_LC₎+ R_{W AKE}

( (

«

◊ «

◊

bias

)

32.05

4

6

19.9

19.9 +178

≈

∆

«

’ ≈

ì 19.5ì97% ì

’ ≈

’

◊

ì

= 0.743V

÷ ∆

÷

22.6 + 32.05

◊ «

(72)

In this case, the VSENSE wake pulse amplitude would be insufficient to trip the internal wake comparator. If the

power stage had higher L_P, or lower C_P, a larger wake pulse would be produced.

67

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ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

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Alternatively, the effective wake resistance R_WAKEmay be reduced by adding the PNP circuit per Figure 40. This

has been verified using Q1 = FMMTA92 PNP transistor, R1= 100 Ω and R2 = 2.2 kΩ. A wake pulse amplitude of

almost 2 V_PKwas produced at the VSENSE pin, giving generous margin to the internal threshold V_SENSE(wake)

The observed waveforms are shown in Figure 47 for a worst case 0% to 100% (65 W) load transient (where the

PWM is at F_MIN). The PWM is re-activated when V_OUThas dropped by ~3%, rather waiting for the next timed

wake-up (~5 ms later).

.

Figure 48 shows a zoomed waveform of the wake pulsing ringing as measured on the bias winding. It can be

seen that the peak level is approximately 3 V_PK, which would produce a pulse of approximately 1.8 V at the

VSENSE pin (scaled by VSENSE divider resistors R_Tand R_B). As noted in Test and Debug Recommendations,

the VSENSE pin should never be directly probed, doing so affects the regulation setpoint.

Figure 47. Observed Output Voltage (Ch3) and Bias Winding (Ch4)

(showing wake event generated by UCC24650)

Figure 48. Zoom In of Wake-Pulse Ringing

(observed across bias winding (ChB) generated by UCC24650)

68

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9.2.6 Energy Star Average Efficiency and Standby Power

Table 9 summarize the standby power, and Table 10 summarizes the average efficiency performance of the

UCC28630EVM-572, (TI Literature Number SLUUAX9).

Table 9. Standby Power Performance

STANDBY POWER

115 V_AC(mW)

230 V_AC(mW)

57

60

Table 10. Average Efficiency Performance

AVERAGE EFFICIENCY (INCLUDING OUTPUT 76-mΩ CABLE DROP)

LOAD LEVEL (%)

25 (16.25 W)

50 (32.5 W)

75 (48.75 W)

100 (65 W)

115 V_AC(%)

89.44

230 V_AC(%)

89.26

88.98

89.38

88.24

89.10

87.59

88.73

Average

88.6

89.1

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9.2.7 Application Performance Plots

Figure 49. Start-Up from 90-V_AC, 3.35-A CC Load

Figure 50. Start-Up from 230-V_AC, 3.35-A CC Load

Figure 51. Output Rise-Time, 90-V_AC, 3.35-A CC Load

Figure 52. Output Rise-Time, 230-V_AC, 3.35 A CC Load

70

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90

89

88

87

86

85

84

83

82

81

90

85

80

75

70

65

115 VAC

230 VAC

115 VAC

230 VAC

80

0

10

20

30

40

50

60

70

0

1

2

3

4

5

6

7

8

Load Power (W)

C017

C018

Figure 53. Efficiency vs. Load/Line

(cable drop included)

Figure 54. Zoom Light-Load Efficiency vs. Load/Line

(cable drop included)

20

19.9

19.8

19.7

19.6

19.5

19.4

19.3

19.2

19.1

20

90 V/50 Hz

100 V/60 Hz

19.8

120 V/60 Hz

143 V/63 Hz

200 V/47 Hz

230 V/50 Hz

269 V/63 Hz

19.6

19.4

19.2

19

90 V/50 Hz

100 V/60 Hz

120 V/60 Hz

143 V/63 Hz

200 V/47 Hz

230 V/50 Hz

269 V/63 Hz

18.8

18.6

19

0

10

20

30

40

50

60

70

0

10

20

30

40

50

60

70

Output Power (W)

C019

C020

Figure 55. Output Voltage Regulation vs. Line/Load

(without cable drop)

Figure 56. Output Voltage Regulation vs. Line/Load

(with cable drop included)

20

19

18

17

16

15

14

13

12

11

10

90 V/50 Hz

100 V/60 Hz

120 V/60 Hz

143 V/63 Hz

200 V/47 Hz

230 V/50 Hz

269 V/63 Hz

0

2

4

6

8

10

Output Current (A)

C021

Figure 57. CC Mode Regulation vs. Line

Figure 58. Transient Step 5% to 50% Load, 115 V_AC

71

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Figure 59. Transient Step 50% to 100% Load, 115 V_AC

Figure 60. Transient Step 10% to 90% Load, 115 V_AC

35

30

0

50

40

0

Gain

œ30

Phase

25

œ60

30

20

œ90

15

œ120

œ150

œ180

œ210

œ240

œ270

œ300

œ330

œ360

20

œ120

œ150

œ180

œ210

œ240

œ270

œ300

œ330

œ360

10

5

0

œ5

œ10

œ20

œ30

œ40

œ10

œ15

œ20

œ25

10

100

Frequency (Hz)

1000

10

100

Frequency (Hz)

1000

C022

C023

Figure 61. Measured Control Loop Gain/Phase at 300 V_DC

,

Figure 62. Measured Control Loop Gain/Phase at 300 V_DC

,

Full Load 3.35 A

Light Load 0.2 A

72

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ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

9.3 Dos and Don'ts

9.3.1 Test and Debug Recommendations

One important precaution must be noted during test and debug. Do not probe the VSENSE pin with an

oscilloscope probe, meter or differential probe. Doing so adds excessive capacitance to the pin, delaying the pin

rise-time, and causing the regulated system output voltage to increase.

10 Power Supply Recommendations

The VDD power pin for the device requires the placement of low-esr noise-decoupling capacitance as directly as

possible from the VDD pin to the GND pin. Ceramic capacitors with stable dielectric characteristics over

temperature are recommended, such as X7R or better. Depending on the operating temperature range of the

application, X5R may be acceptable, but the drop in capacitance value at high temperature and with applied DC-

bias may not be tolerable. Avoid dielectrics with poor temperature-stability. (such Y5V, Z5U)

The recommended decoupling capacitors are a 1-μF 1206-sized 50-V X7R capacitor, ideally with (but not

essential) a second smaller parallel 100-nF 0603-sized 50-V X7R capacitor. Higher voltage rating parts can also

be used. The use of 25-V rated parts is not recommended, due to the reduction in effective capacitance value

with applied DC bias.

In parallel with the ceramic noise-decoupling capacitor(s), a larger-capacitance energy storage capacitor is also

required, per Equation 58. This energy-storage capacitor does not require low esr, and does not necessarily

need to be located close to the device.

73

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11 Layout

11.1 Layout Guidelines

11.1.1 HV Pin

•

This pin is connected to the rectified AC input, and as such requires appropriate separation to other PCB

traces to meet the application requirements for functional isolation;

•

This pin must have 200 kΩ of external resistance to allow the line voltage to be sensed for the X-capacitor

discharge block. At least two series resistors should be used to reduce the voltage across the pins of each

resistor, with each resistor rated for at least 200 V;

•

The connection to the resistors that feed the HV pin should have separate dedicated rectifying diodes from

the AC input lines, to avoid the DC filtering that the bulk capacitor provides after the main diode bridge; the

lower section of the main diode bridge can be shared by the device and the power stage;

A filtering or noise-decoupling capacitor is not recommended, such a capacitor will degrade the X-capacitor

sampling ability to distinguish AC from DC input.

11.1.2 VDD Pin

•

A 1-µF ceramic decoupling capacitor is recommended, placed as close as possible between the VDD pin and

GND, tracked directly to both pins.

11.1.3 VSENSE Pin

•

The tracking and layout of the VSENSE pin and connecting components is critical to minimizing noise pick-up

and interference in the magnetic sensing block. (See Figure 63 for suggested component placement and

tracking). Reduce the total surface area of traces on the VSENSE net to a minimum.

•

Because the resistance values of R_Aand R_Bare relatively high to minimize power dissipation, the high

impedance makes the VSENSE pin potentially noise-sensitive. To minimize noise pick-up, locate resistors R_A

and R_Bas close as possible to the VSENSE pin, with R_Bin particular placed as directly as possible between

VSENSE and GND pins;

•

Depending on layout, a small noise filter capacitor may be useful on the VSENSE pin, such as C15 shown in

Figure 44. Connect this capacitor as directly as possible between the VSENSE and GND pins. Choose the

value of this capacitor as small as possible, and no greater than 10 pF. A larger value significantly delays the

voltage rise-time at the pin, and affects the regulation set-point;

In case of possible board faults that can pull the VSENSE pin below GND (such as R7 shorted), in order to

protect the pin and limit possible negative current out of the pin, a series resistor R4 (as shown in Figure 44)

and clamping diode from GND are recommended. Maintain the value of R4 between 100 Ω and 500 Ω. A

larger value may affect regulation and line sense accuracy.

For correct line sense operation, the switched pull-up R10 and D4 must be added. The value of R10 must be

3.9 kΩ to match the internal device gain. The switched pull-up diode and the GND clamping diode can be

combined into a dual-diode common-cathode package, such as D4 as shown in Figure 44.

11.1.4 CS Pin

•

A small, external filter capacitor is recommended on the CS pin. Track the filter capacitor as directly as

possible from the CS to GND pin.

•

Referring to Figure 44, a series resistor such as R5 is typically connected between the current sensing

resistor R16 and the CS pin to form an R-C filter. A filter time constant between 100 ns and 200 ns is

recommended. If the filter time constant is made too large, the filtering causes the transformer peak current to

exceed the control loop demand level, which affects regulation and standby power. Place resistor R5 as close

as possible to the CS pin.

•

Reduce the total surface area of traces on the CS net to a minimum.

74

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

www.ti.com.cn

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

Layout Guidelines (continued)

11.1.5 SD Pin

•

Referring to Figure 44, the SD pin is connected to a temperature-sensing NTC RT1 in series with an adjust

resistor R6. The NTC can be tracked to the required hot-spot location, or it can be wired with flying leads to

the required hotspot.

•

Track the RT1 return to GND as directly as possible back to the GND pin of the device. RT1 should not be

connected to a power GND track or plane, in order to minimize error in the trip level.

The device internally filters the SD pin, so an external filter capacitor is not usually required. If the application

design requires an external capacitor, limit the value to 1 nF maximum.

11.1.6 DRV Pin

•

The DRV pin has high internal sink/source current capability. An external gate resistor is recommended. The

value depends on the choice of power MOSFET, efficiency and EMI considerations.

As shown in Figure 44 an anti-parallel path formed by D5 and R13 are placed across the gate resistor R11 to

allow turn-on and turn-off of the MOSFET to be independently adjusted.

A pull-down resistor (such as R15 in this example) on the gate of the external MOSFET is recommended to

prevent the MOSFET gate from floating on if there is an open circuit error in the gate drive path. The value of

R15 also affects the overload timer settings, so carefully choose the value of R15 according to Table 2.

•

Ensure that the noisy gate drive traces are routed away from the sensitive VSENSE pin and CS pin traces.

11.1.7 GND Pin

•

Connect decoupling and noise filter capacitors, as well as sensing resistors directly to the GND pin in a star-

point fashion, ensuring that the current-carrying power tracks (such as the gate drive return) are track

separately to avoid noise and ground-drops that could affect the analogue signal integrity.

11.2 Layout Example

w_!

Ço .ias

ꢀinding

UCC28630

~100 Q

ë{ꢃb{ꢃ

Ië

w_.2

w_.1

G10 pC

{5

ꢂ{

ë55

5wë

G120 pC

100 nC

1 µC

3ꢁ9 lQ

Db5

Ço 5wë

Wumper/

link

Ço ꢂ{

resistor

Figure 63. Recommended PCB Layout for Single-Sided Assembly

75

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

www.ti.com.cn

12 器件和文档支持

12.1 商标

WEBENCH is a registered trademark of Texas Instruments.

All other trademarks are the property of their respective owners.

12.2 静电放电警告

这些装置包含有限的内置 ESD 保护。存储或装卸时，应将导线一起截短或将装置放置于导电泡棉中，以防止 MOS 门极遭受静电损

伤。

12.3 Glossary

SLYZ022 — TI Glossary.

This glossary lists and explains terms, acronyms, and definitions.

12.4 器件支持

12.4.1 开发支持

12.4.1.1 使用 WEBENCH® 工具创建定制设计

请单击此处，借助 WEBENCH® 电源设计器并使用 UCC2863x 器件创建定制设计方案。

1. 首先键入输入电压 (V_IN)、输出电压 (V_OUT) 和输出电流 (I_OUT) 要求。

2. 使用优化器拨盘优化关键参数设计，如效率、封装和成本。

3. 将生成的设计与德州仪器 (TI) 的其他解决方案进行比较。

WEBENCH 电源设计器可提供定制原理图以及罗列实时价格和组件供货情况的物料清单。

在多数情况下，可执行以下操作：

•

运行电气仿真，观察重要波形以及电路性能

运行热性能仿真，了解电路板热性能

将定制原理图和布局方案导出至常用 CAD 格式

打印设计方案的 PDF 报告并与同事共享

有关 WEBENCH 工具的详细信息，请访问 www.ti.com/WEBENCH。

12.5 文档支持

12.5.1 相关文档

《UCC28630EVM-572，65W

SLUSAX9）

标称功率，130W

峰值功率，初级侧稳压适配器模块》

（德州仪器文献编号

12.5.1.1 相关链接

下面的表格列出了快速访问链接。类别包括技术文档、支持与社区资源、工具和软件，以及申请样片或购买产品的

快速链接。

表 11. 相关链接

器件

产品文件夹

请单击此处

样片与购买

请单击此处

技术文档

请单击此处

工具和软件

请单击此处

支持和社区

请单击此处

UCC28630

UCC28631

UCC28632

UCC28633

76

UCC28630, UCC28631

UCC28632, UCC28633, UCC28634

www.ti.com.cn

ZHCSC62D –MARCH 2014–REVISED DECEMBER 2017

13 机械、封装和可订购信息

以下页中包括机械封装、封装和可订购信息。这些信息是针对指定器件可提供的最新数据。数据如有变更，恕不另

行通知和修订此文档。如欲获取此产品说明书的浏览器版本，请参阅左侧的导航。

77

PACKAGE OUTLINE

D0007A

SOIC - 1.75 mm max height

SCALE 2.800

SMALL OUTLINE INTEGRATED CIRCUIT

C

SEATING PLANE

.228-.244 TYP

[5.80-6.19]

.004 [0.1] C

A

PIN 1 ID AREA

8

1

.100

[2.54]

2X

.189-.197

[4.81-5.00]

NOTE 3

.150

[3.81]

4X .050

[1.27]

4

5

7X .012-.020

[0.31-0.51]

B

.150-.157

[3.81-3.98]

NOTE 4

.069 MAX

[1.75]

.010 [0.25]

C A B

.005-.010 TYP

[0.13-0.25]

SEE DETAIL A

.010

[0.25]

.004-.010

[0.11-0.25]

0 - 8

.016-.050

[0.41-1.27]

DETAIL A

TYPICAL

(.041)

[1.04]

4220728/A 01/2018

NOTES:

1. Linear dimensions are in inches [millimeters]. Dimensions in parenthesis are for reference only. Controlling dimensions are in inches.

Dimensioning and tolerancing per ASME Y14.5M.

2. This drawing is subject to change without notice.

3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not

exceed .006 [0.15] per side.

4. This dimension does not include interlead flash.

5. Reference JEDEC registration MS-012, variation AA.

www.ti.com

EXAMPLE BOARD LAYOUT

D0007A

SOIC - 1.75 mm max height

SMALL OUTLINE INTEGRATED CIRCUIT

7X (.061 )

[1.55]

SYMM

SEE

DETAILS

1

8

7X (.024)

[0.6]

(.100 )

[2.54]

SYMM

5

4

4X (.050 )

[1.27]

(.213)

[5.4]

LAND PATTERN EXAMPLE

EXPOSED METAL SHOWN

SCALE:8X

SOLDER MASK

OPENING

SOLDER MASK

OPENING

METAL UNDER

SOLDER MASK

METAL

EXPOSED

METAL

EXPOSED

METAL

.0028 MAX

[0.07]

ALL AROUND

.0028 MIN

[0.07]

ALL AROUND

SOLDER MASK

DEFINED

NON SOLDER MASK

DEFINED

SOLDER MASK DETAILS

4220728/A 01/2018

NOTES: (continued)

6. Publication IPC-7351 may have alternate designs.

7. Solder mask tolerances between and around signal pads can vary based on board fabrication site.

www.ti.com

EXAMPLE STENCIL DESIGN

D0007A

SOIC - 1.75 mm max height

SMALL OUTLINE INTEGRATED CIRCUIT

7X (.061 )

[1.55]

SYMM

1

8

7X (.024)

[0.6]

(.100 )

[2.54]

SYMM

5

4

4X (.050 )

[1.27]

(.213)

[5.4]

SOLDER PASTE EXAMPLE

BASED ON .005 INCH [0.125 MM] THICK STENCIL

SCALE:8X

4220728/A 01/2018

NOTES: (continued)

8. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate

design recommendations.

9. Board assembly site may have different recommendations for stencil design.

www.ti.com

重要声明和免责声明

TI“按原样”提供技术和可靠性数据（包括数据表）、设计资源（包括参考设计）、应用或其他设计建议、网络工具、安全信息和其他资源，

不保证没有瑕疵且不做出任何明示或暗示的担保，包括但不限于对适销性、某特定用途方面的适用性或不侵犯任何第三方知识产权的暗示担

保。

这些资源可供使用 TI 产品进行设计的熟练开发人员使用。您将自行承担以下全部责任：(1) 针对您的应用选择合适的 TI 产品，(2) 设计、验

证并测试您的应用，(3) 确保您的应用满足相应标准以及任何其他功能安全、信息安全、监管或其他要求。

这些资源如有变更，恕不另行通知。TI 授权您仅可将这些资源用于研发本资源所述的 TI 产品的应用。严禁对这些资源进行其他复制或展示。

您无权使用任何其他 TI 知识产权或任何第三方知识产权。您应全额赔偿因在这些资源的使用中对 TI 及其代表造成的任何索赔、损害、成

本、损失和债务，TI 对此概不负责。

TI 提供的产品受 TI 的销售条款或 ti.com 上其他适用条款/TI 产品随附的其他适用条款的约束。TI 提供这些资源并不会扩展或以其他方式更改

TI 针对 TI 产品发布的适用的担保或担保免责声明。

TI 反对并拒绝您可能提出的任何其他或不同的条款。IMPORTANT NOTICE

邮寄地址：Texas Instruments, Post Office Box 655303, Dallas, Texas 75265


型号：	UCC28634D
厂家：	TEXAS INSTRUMENTS
描述：	具有 PSR、峰值功率模式、可调节 CC 限制和频率抖动功能的高功率反激式控制器 \| D \| 7 \| -40 to 125 控制器开关光电二极管
文件：	总86页 (文件大小：3650K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

UCC28634D [TI]

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UCC28700_14

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