FQP16N15 [TRIPATH]
STEREO 200W CLASS-T DIGITAL AUDIO AMPLIFIER DRIVER USING DIGITAL POWER PROCESSING TECHNOLOGY; 200W立体声CLASS -T数字音频放大器驱动器使用数字功率处理技术
| 型号： | FQP16N15 |
| 厂家： | 
TRIPATH TECHNOLOGY INC. |
| 描述： | STEREO 200W CLASS-T DIGITAL AUDIO AMPLIFIER DRIVER USING DIGITAL POWER PROCESSING TECHNOLOGY |
| 文件： | 总34页 (文件大小：664K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Tripath Technology, Inc. - Technical Information
TK2150
STEREO 200W (6Ω) CLASS-T DIGITAL AUDIO AMPLIFIER DRIVER
USING DIGITAL POWER PROCESSINGTM TECHNOLOGY
T e c h n i c a l I n f o r m a t i o n - P r e l i m i n a r y
R e vi s i o n 1 . 0 – D e c e m b e r 2 0 0 2
G E N E R A L D E S C R I P T I O N
T he T K2150 (T C2001/T P2150 chipset) is a two-channel, 200W (6Ω ) per channel Amplifier
Driver that uses T rip ath’s proprietary Digital Power Processing (DPP T M
)
technology.
Class-T amplifiers offer both the audio fidelit y of Class-AB and the power efficiency of
Class-D amplifiers.
Applications
Features
ꢀ
ꢀ
ꢀ
ꢀ
ꢀ
Powered DVD Players
ꢀ
ꢀ
ꢀ
ꢀ
Class-T architecture
Audio/Video Amplifiers & Receivers
Automobile Power Amplifiers
Subwoofer Amplifiers
Pin compatible with Tripath TK2350 Chipset
Proprietary Digital Power Processing technology
“Audiophile” Sound Quality
Pro-audio Amplifiers
ꢀ
ꢀ
0.012% THD+N @ 60W, 8Ω
0.02% IHF-IM @ 30W, 8Ω
ꢀ
ꢀ
High Efficiency
Benefits
ꢀ
ꢀ
93% @ 120W @ 8Ω
91% @ 150W @ 6Ω
ꢀ
ꢀ
ꢀ
Reduced system cost with smaller/less
expensive power supply and heat sink
Signal fidelity equal to high quality
Class-AB amplifiers
Supports wide range of output power levels
ꢀ
Up to 200W/channel (6Ω), single-ended outputs,
@+/- 45V
High dynamic range compatible with
digital media such as CD and DVD
ꢀ
Up to 400W (8Ω), bridged outputs, @+/- 30V
ꢀ
ꢀ
ꢀ
Output over-current protection
Over- and under-voltage protection
Over-temperature protection
Typical Performance for TK2150
THD+N versus Output Power versus Supply Voltage
10
5
RL = 6Ω
Vs = +35V, +40V, +45V
f = 1kHz
2
1
BBM = 40nS
BW = 22hZ - 20kHz(AES17)
NOTE: +45V test uses RFBC=11kΩ
(see Application/ Test Circuit)
0.5
0.2
0.1
%
0.05
0.02
0.01
0.005
0.002
0.001
1
2
5
10
20
50
100
200
W
1
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
Absolute Maximum Ratings TC2001 (Note 1)
SYMBOL
V5
Vlogic
TA
TSTORE
TJMAX
ESDHB
PARAMETER
Value
6
V5+0.3V
-40° to +85°
-55° to 150°
150°
UNITS
V
V
°C
°C
5V Power Supply
Input Logic Level
Operating Free-air Temperature Range
Storage Temperature Range
Maximum Junction Temperature
ESD Susceptibility – Human Body Model (Note 2)
All pins
°C
2000
V
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur.
See the table below for Operating Conditions.
Note 2: Human body model, 100pF discharged through a 1.5KΩ resistor.
Absolute Maximum Ratings TP2150 (Note 3)
SYMBOL
VPP, VNN Supply Voltage
PARAMETER
Value
+/- 65
UNITS
V
VN10
TSTORE
TA
Voltage for FET drive
Storage Temperature Range
Operating Free-air Temperature Range (Note 4)
Junction Temperature
VNN+13
-55º to 150º
-40º to 85º
150º
V
°C
°C
°C
TJ
ESDHB
ESD Susceptibility – Human Body Model (Note 5)
All pins
2000
TBD
V
V
ESDMM
ESD Susceptibility – Machine Model (Note 6)
All pins
Note 3: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur.
See the table below for Operating Conditions.
Note 4: This is a target specification. Characterization is still needed to validate this temperature range.
Note 5: Human body model, 100pF discharged through a 1.5KΩ resistor.
Note 6: Machine model, 220pF – 240pF discharged through all pins.
Operating Conditions TC2001 (Note 7)
SYMBOL
V5
VHI
VLO
TA
PARAMETER
MIN.
4.5
V5-1.0
TYP.
5
MAX. UNITS
Supply Voltage
Logic Input High
Logic Input Low
5.5
V
V
1
V
Operating Temperature Range
°C
-40°
25°
85°
Note 7: Recommended Operating Conditions indicate conditions for which the device is functional.
See Electrical Characteristics for guaranteed specific performance limits.
Operating Conditions TP2150 (Note 8)
SYMBOL
VPP, VNN Supply Voltage
VN10
PARAMETER
MIN.
+/- 15 +/-30 +/- 60
10 12
TYP.
MAX. UNITS
V
V
Voltage for FET drive (Volts above VNN)
9
Note 8: Recommended Operating Conditions indicate conditions for which the device is functional.
See Electrical Characteristics for guaranteed specific performance limits.
2
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
Operating Characteristics TC2001 (Note 9)
SYMBOL
I5
PARAMETER
MIN.
TYP.
50
MAX. UNITS
Supply Current
mA
VIN
VOUTHI
VOUTLO
Input Sensitivity
High Output Voltage
Low Output Voltage
Input DC Bias
0
1.5
V
V
mV
V
V5-0.5
100
2.4
Note 9: Recommended Operating Conditions indicate conditions for which the device is functional.
See Electrical Characteristics for guaranteed specific performance limits.
Thermal Characteristics TC2001
SYMBOL
PARAMETER
Value
UNITS
Junction-to-ambient Thermal Resistance (still air)
C/W
θ
JA
80°
Thermal Characteristics TP2150
SYMBOL
JC
PARAMETER
Junction-to-case Thermal Resistance
Value
TBD
°
UNITS
C/W
θ
Electrical Characteristics TC2001 (Note 10)
TA = 25 °C. See Application/Test Circuit on page 7. Unless otherwise noted, the supply voltage is
VPP=|VNN|=45V.
SYMBOL
Iq
PARAMETER
CONDITIONS
MIN.
TYP.
45
MAX. UNITS
Quiescent Current
V5 = 5V
V5 = 5V
60
mA
(Mute = 0V)
IMUTE
Mute Supply Current
(Mute = 5V)
20
25
mA
VIH
VIL
VOH
VOL
VTOC
High-level input voltage (MUTE)
3.5
4.0
V
V
V
V
V
Low-level input voltage (MUTE)
High-level output voltage (HMUTE) IOH = 3mA
Low-level output voltage (HMUTE) IOL = 3mA
1.0
0.5
TBD
Over Current Sense Voltage
TBD
TBD
1.0
Threshold
IVPPSENSE VPPSENSE Threshold Currents
Over-voltage turn on (muted)
Over-voltage turn off (mute off)
Under-voltage turn off (mute off)
Under-voltage turn on (muted)
Over-voltage turn on (muted)
Over-voltage turn off (mute off)
Under-voltage turn off (mute off)
Under-voltage turn on (muted)
Over-voltage turn on (muted)
Over-voltage turn off (mute off)
Under-voltage turn off (mute off)
Under-voltage turn on (muted)
162
154
79
178
87
µA
µA
µA
µA
V
138
62
72
VVPPSENSE Threshold Voltages with
57.8
55.0
28.2
25.7
174
169
86
63.5
31.1
191
95
49.3
22.1
152
65
V
R
VPP1 = RVPP1 = 357KΩ
V
(Note 11, Note 12)
V
IVNNSENSE VNNSENSE Threshold Currents
µA
µA
µA
µA
V
77
VVNNSENSE Threshold Voltages with
RVNN1 = 324KΩ
Over-voltage turn on (muted)
Over-voltage turn off (mute off)
Under-voltage turn off (mute off)
Under-voltage turn on (muted)
-56.4
-54.8
-27.9
-24.9
-61.9
-30.8
-49.2
-21.1
V
V
RVNN2 = 976KΩ
V
(Note 11, Note 12)
Note 10: Minimum and maximum limits are guaranteed but may not be 100% tested.
3
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
Note 11: These supply voltages are calculated using the IVPPSENSE and IVNNSENSE values shown in the
Electrical Characteristics table. The typical voltage values shown are calculated using a RVPP and RVNN
values without any tolerance variation. The minimum and maximum voltage limits shown include either a
+1% or –1% (+1% for Over-voltage turn on and Under-voltage turn off, -1% for Over-voltage turn off and
Under-voltage turn on) variation of RVPP or RVNN off the nominal 357kohm, 324kohm, and 976kohm
values. These voltage specifications are examples to show both typical and worst case voltage ranges for
the given RVPP and RVNN resistor values. Please refer to the Application Information section for a more
detailed description of how to calculate the over and under voltage trip voltages for a given resistor value.
Note 12: The fact that the over-voltage turn on specifications exceed the absolute maximum of +/-60V for the TK2150
does not imply that the part will work at these elevated supply voltages. It also does not imply that the
TK2150 is tested or guaranteed at these supply voltages. The supply voltages are simply a calculation
based on the process spread of the IVPPSENSE and IVNNSENSE currents (see note 7). The supply
voltage must be maintained below the absolute maximum of +/-60V or permanent damage to the TK2150
may occur.
Electrical Characteristics TP2150 (Note 13)
TA = 25 °C. See Application/Test Circuit on page 7. Unless otherwise noted, the supply voltage is
VPP=|VNN|=45V.
SYMBOL
Iq
PARAMETER
CONDITIONS
MIN.
TYP.
MAX. UNITS
Quiescent Current
VPP = +45V
25
mA
(No load, BBM0=1,BBM1=0,
Mute = 0V)
VNN = -45V (Note 14)
45
mA
IMUTE
Mute Supply Current
(No load, Mute = 5V)
VPP = +45V
VNN = -45V
1
1
mA
mA
Note 13: Minimum and maximum limits are guaranteed but may not be 100% tested.
Note 14: The difference in the VPP and VNN current draw is due to the VN10 regulator sourcing current to the
VNN supply.
Performance Characteristics TK2150 – Single Ended
TA = 25 °C. Unless otherwise noted, the supply voltage is VPP=|VNN|=45V, the input frequency is 1kHz
and the measurement bandwidth is 20kHz. See Application/Test Circuit.
SYMBOL
POUT
PARAMETER
Output Power
(continuous RMS/Channel)
CONDITIONS
MIN.
TYP.
MAX. UNITS
100
135
120
155
W
W
W
W
THD+N = 0.1%, RL = 8Ω
RL = 6Ω
THD+N = 1%,
RL = 8Ω
RL = 6Ω
THD + N Total Harmonic Distortion Plus
Noise
IHF-IM
SNR
CS
0.012
%
P
OUT = 70W/Channel, RL = 8Ω
IHF Intermodulation Distortion
0.02
%
19kHz, 20kHz, 1:1 (IHF), RL = 8Ω
OUT = 30W/Channel
A Weighted, RL = 6Ω,
OUT = 155W/Channel
0dBr = 30W, RL = 8Ω, f = 1kHz
POUT = 150W/Channel, RL = 8Ω
P
Signal-to-Noise Ratio
104.5
dB
P
Channel Separation
Power Efficiency
Amplifier Gain
92
93
13.3
dB
%
V/V
η
AV
P
OUT = 10W/Channel, RL = 6Ω
See Application / Test Circuit
AVERROR
eNOUT
Channel to Channel Gain Error
Output Noise Voltage
0.5
1.0
dB
P
OUT = 10W/Channel, RL = 6Ω
See Application / Test Circuit
A Weighted, no signal, input shorted,
DC offset nulled to zero, RFBC = 11kΩ
No Load, Mute = Logic Low
180
µV
VOFFSET
Output Offset Voltage
-1.0
V
0.1% RFBA, RFBB, RFBC resistors
4
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
TK2150 Block Diagram
LC
Output Left
Filter
Input Left
TC2001
Audio
TP2150
MOSFET
Driver
Output
MOSFETs
Signal
LC
Processor
Input Right
Output Right
Filter
5
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
TC2001 Pinout
28-pin SOIC
(Top View)
28
27
26
1
2
3
4
5
6
7
8
BIASCAP
FBKGND2
DCMP
FBKOUT2
VPWR
FBKGND1
FBKOUT1
HMUTE
Y1
INV2
OAOUT2
BBM0
BBM1
MUTE
INV1
OAOUT1
V5
25
24
23
22
21
20
19
18
17
16
15
9
AGND
10
11
12
13
14
Y1B
Y2B
Y2
NC
VPPSENSE
OVRLDB
VNNSENSE
OCD1
OCD2
REF
TP2150 Pinout
64-pin LQFP
(Top View)
51 50 49 48 47
40 39 38
36 35 34 33
37
46 45 44 43 42 41
52
53
54
55
56
57
58
59
60
61
62
63
64
32
31
30
29
28
27
26
25
24
23
22
21
20
NC
OCS1LN
NC
OCS2LN
OCS2LP
NC
NC
VBOOT2
OCS1LP
NC
NC
VBOOT1
NC
SW -FB
SMPSO
NC
NC
NC
NC
NC
NC
NC
NC
SLEEP
NC
NC
1
2
3
4
5
6
7
8
9
15 16 17 18 19
10 11 12 13 14
6
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
TC2001 Audio Signal Processor Pin Descriptions
Pin
Function
Description
1
BIASCAP
Bandgap reference times two (typically 2.5VDC). Used to set the
common mode voltage for the input op amps. This pin is not capable of
driving external circuitry.
2, 6
3
FBKGND2,
FBKGND1
DCMP
Ground Kelvin feedback (Channels 1 & 2)
Internal mode selection. This pin must be grounded for proper device
operation.
4, 7
FBKOUT2,
FBKOUT1
VPWR
Switching feedback (Channels 1 & 2)
5
8
Test pin. Must be left floating.
HMUTE
Logic output. A logic high indicates both amplifiers are muted, due to the
mute pin state, or a “fault”.
9, 12
10, 11
13
Y1, Y2
Y1B, Y2B
NC
Non-inverted switching modulator outputs.
Inverted switching modulator outputs.
No connect
14
OCD2
Over Current Detect input.
15
REF
Internal bandgap reference voltage; approximately 1.2 VDC.
Over Current Detect input.
16
OCD1
17
VNNSENSE
Negative supply voltage sense input. This pin is used for both over and
under voltage sensing for the VNN supply.
A logic low output indicates the input signal has overloaded the amplifier.
Positive supply voltage sense input. This pin is used for both over and
under voltage sensing for the VPP supply.
Ground.
18
19
OVRLDB
VPPSENSE
20
AGND
V5
21
5 Volt power supply input.
22, 27
23, 28
OAOUT1, OAOUT2 Input stage output pins.
IN1, IN2
Single-ended inputs. Inputs are a “virtual” ground of an inverting opamp
with approximately 2.4VDC bias.
24
MUTE
When set to logic high, both amplifiers are muted and in idle mode.
When low (grounded), both amplifiers are fully operational. If left floating,
the device stays in the mute mode. Ground if not used.
Break-before-make timing control to prevent shoot-through in the output
MOSFETs.
25, 26
BBM1, BBM0
7
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
TP2150 Pin Description
Pin
2,5
6
Function
Description
AGND
V5
Analog ground.
5V power supply input.
7
OCD1
Over-current threshold output (Channel 1)
Soft startup for VN10 controller, this pin should be tied to V5
Over-current threshold output (Channel 2)
Non-inverted switching modulator inputs
Inverted switching modulator inputs
9
10
13,17
14,16
27,57
CSS
OCD2
Y2, Y1
Y2B, Y1B
VBOOT2, VBOOT1
Bootstrapped voltage to supply drive to gate of high-side FET
(Channel 2 & 1)
30,31
33,34
36,48
37,47
39,45
40,44
41,43
OCS2LP, OCS2LN
OCS2HP, OCS2HN
HO2, HO1
Over Current Sense inputs, Channel 2 low-side
Over Current Sense inputs, Channel 2 high-side
High side gate drive output (Channel 2 & 1)
HO2COM, HO1COM
LO2COM, LO1COM
LO2, LO1
Kelvin connection to source of high-side transistor (Channel 2 & 1)
Kelvin connection to source of low-side transistor (Channel 2 & 1)
Low side gate drive output (Channel 2 & 1)
VN10
“Floating” supply input for the FET drive circuitry. This voltage must be stable
and referenced to VNN.
42
50,51
53,54
59
60
62
VNN
OCS1HN, OCS1HP
OCS1LN , OCS1LP
SW-FB
Negative supply voltage.
Over Current Sense inputs, Channel 1 high-side
Over Current Sense inputs, Channel 1 low-side
Feedback for regulating switching power supply output for VN10
Switching power supply output for VN10
SMPSO
SLEEP
This pin is active high. Tie this pin to GND for normal operation. Tie this pin to
+5V to place the part in sleep mode.
1,3,4,8,
11,12,15,
18,19,20,
21,22,23,
24,25,26,
28,31,32,
35,38,46,
49,52,53,
56,58,61,
63,64
NC
Not connected (bonded) internally. Please refer to the Application/Test circuit
for details on the how to connect these pins.
8
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
Application/Test Circuit
9
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
External Components Description (Refer to the Application/Test Circuit)
Components Description
RI
RF
CI
Inverting input resistance to provide AC gain in conjunction with RF. This input is
biased at the BIASCAP voltage (approximately 2.5VDC).
Feedback resistor to set AC gain in conjunction with RI. Please refer to the Amplifier
Gain paragraph, in the Application Information section.
AC input coupling capacitor which, in conjunction with RI, forms a highpass filter at
.
fC = 1 (2πRICI)
RFBA
RFBB
Feedback divider resistor connected to V5. This resistor is normally set at 1kΩ.
Feedback divider resistor connected to AGND. This value of this resistor depends
on the supply voltage setting and helps set the TK2150 gain in conjunction with RI,
RF, RFBA, and RFBC. Please see the Modulator Feedback Design paragraphs in the
Application Information Section.
RFBC
Feedback resistor connected from either the OUT1(OUT2) to FBKOUT1(FBKOUT2)
or speaker ground to FBKGND1(FBKGND2). The value of this resistor depends on
the supply voltage setting and helps set the TK2150 gain in conjunction with RI, RF,
RFBA,, and RFBB. It should be noted that the resistor from OUT1(OUT2) to
P
= VPP2 (2RFBC
FBKOUT1(FBKOUT2) must have a power rating of greater than
.
)
DISS
Please see the Modulator Feedback Design paragraphs in the Application
Information Section.
CFB
Feedback delay capacitor that both lowers the idle switching frequency and filters
very high frequency noise from the feedback signal, which improves amplifier
performance. The value of CFB should be offset between channel 1 and channel 2
so that the idle switching difference is greater than 40kHz. Please refer to the
Application / Test Circuit.
ROFA
ROFB
Potentiometer used to manually trim the DC offset on the output of the TK2350.
Resistor that limits the manual DC offset trim range and allows for more precise
adjustment.
RREF
CA
Bias resistor. Locate close to pin 15 of the TC2001 and ground at pin 20 of the
TC2001.
BIASCAP decoupling capacitor. Should be located close to pin 1 of the TC2001 and
grounded at pin 20 of the TC2001.
DB
Bootstrap diode. This diode charges up the bootstrap capacitors when the output is
low (at VNN) to drive the high side gate circuitry. A fast or ultra fast recovery diode
is recommended for the bootstrap circuitry. In addition, the bootstrap diode must be
able to sustain the entire VPP-VNN voltage. Thus, for most applications, a 150V (or
greater) diode should be used.
CB
High frequency bootstrap capacitor, which filters the high side gate drive supply.
This capacitor must be located as close to VBOOT1 (pin 57 of the TP2150) or
VBOOT2 (pin 27 of the TP2150) for reliable operation. The “negative” side of CB
should be connected directly to the HO1COM (pin 47 of the TP2150) or HO2COM
(pin 37 of the TP2150). Please refer to the Application / Test Circuit.
Bulk bootstrap capacitor that supplements CB during “clipping” events, which result
in a reduction in the average switching frequency.
CBAUX
RB
Bootstrap resistor that limits CBAUX charging current during TK2150 power up
(bootstrap supply charging).
CS
Supply decoupling for the power supply pins. For optimum performance, these
components should be located close to the TC2001 and TP2150 and returned to
their respective ground as shown in the Application/Test Circuit.
Main overvoltage and undervoltage sense resistor for the negative supply (VNN).
Please refer to the Electrical Characteristics Section for the trip points as well as the
hysteresis band. Also, please refer to the Over / Under-voltage Protection section in
the Application Information for a detailed discussion of the internal circuit operation
and external component selection.
RVNN1
RVNN2
Secondary overvoltage and undervoltage sense resistor for the negative supply
(VNN). This resistor accounts for the internal VNNSENSE bias of 1.25V. Nominal
TK2150 – Rev. 1.0/12.02
10
Tripath Technology, Inc. - Technical Information
resistor value should be three times that of RVNN1. Please refer to the Over / Under-
voltage Protection section in the Application Information for a detailed discussion of
the internal circuit operation and external component selection.
RVPP1
Main overvoltage and undervoltage sense resistor for the positive supply (VPP).
Please refer to the Electrical Characteristics Section for the trip points as well as the
hysteresis band. Also, please refer to the Over / Under-voltage Protection section in
the Application Information for a detailed discussion of the internal circuit operation
and external component selection.
RVPP2
Secondary overvoltage and undervoltage sense resistor for the positive supply
(VPP). This resistor accounts for the internal VPPSENSE bias of 2.5V. Nominal
resistor value should be equal to that of RVPP1. Please refer to the Over / Under-
voltage Protection section in the Application Information for a detailed discussion of
the internal circuit operation and external component selection.
RS
Over-current sense resistor. Please refer to the section, Setting the Over-current
Threshold, in the Application Information for a discussion of how to choose the value
of RS to obtain a specific current limit trip point.
ROCR
Over-current “trim” resistor, which, in conjunction with RS, sets the current trip point.
Please refer to the section, Setting the Over-current Threshold, in the Application
Information for a discussion of how to calculate the value of ROCR
.
COCR
Over-current filter capacitor, which filters the overcurrent signal at the OCR pins to
account for the half-wave rectified current sense circuit internal to the TC2001. A
typical value for this component is 220pF. In addition, this component should be
located near pin 14 or pin 16 of the TC2001 as possible.
CHBR
Supply decoupling for the high current Half-bridge supply pins. These components
must be located as close to the output MOSFETs as possible to minimize output
ringing which causes power supply overshoot. By reducing overshoot, these
capacitors maximize both the TP2150 and output MOSFET reliability. These
capacitors should have good high frequency performance including low ESR and
low ESL. In addition, the capacitor rating must be twice the maximum VPP voltage.
Panasonic EB capacitors are ideal for the bulk storage (nominally 33uF) due to their
high ripple current and high frequency design.
RG
DG
Gate resistor, which is used to control the MOSFET rise/ fall times. This resistor
serves to dampen the parasitics at the MOSFET gates, which, in turn, minimizes
ringing and output overshoots. The typical power rating is 1/2 watt.
Gate diode, placed in parallel to the gate resistor. This diode will help discharge the
parasitic capacitance at the MOSFET gates, thus increasing the MOSFET fall time.
This help reduce shoot through current between the top side and bottom side output
MOSFETs. This should be a schottky or ultrafast rectifier. This part may not be
needed depending on the type of output MOSFET used.
CZ
RZ
Zobel capacitor, which in conjunction with RZ, terminates the output filter at high
frequencies. Use a high quality film capacitor capable of sustaining the ripple current
caused by the switching outputs.
Zobel resistor, which in conjunction with CZ, terminates the output filter at high
frequencies. The combination of RZ and CZ minimizes peaking of the output filter
under both no load conditions or with real world loads, including loudspeakers which
usually exhibit a rising impedance with increasing frequency. Depending on the
program material, the power rating of RZ may need to be adjusted. The typical
power rating is 2 watts.
LO
Output inductor, which in conjunction with CO, demodulates (filters) the switching
waveform into an audio signal. Forms a second order filter with a cutoff frequency
of
and a quality factor of
.
LO CO
f
= 1 (2π L
C )
O
Q = RL CO
C
O
CO
Output capacitor, which, in conjunction with LO, demodulates (filters) the switching
waveform into an audio signal. Forms a second order low-pass filter with a cutoff
frequency of
and a quality factor of
. Use
LO CO
f
= 1 (2π L
C )
O
Q = RL CO
C
O
a high quality film capacitor capable of sustaining the ripple current caused by the
switching outputs.
11
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
DD
Drain diode. This diode must be connected from the drain of the high side output
MOSFET to the drain of the low side output MOSFET. This diode absorbs any high
frequency overshoots caused by the output inductor LO during high output current
conditions. In order for this diode to be effective it must be connected directly to the
drains of both the top and bottom side output MOSFET. A fast or ultra fast recovery
diode that can sustain the entire VPP-VNN voltage should be used here. In most
applications a 150V or greater diode must be used.
DS
Source diode. This diode must be connected from the source of the high side
output MOSFET to the source of the low side output MOSFET. This diode absorbs
any high frequency undershoots caused by the output inductor LO during high output
current conditions. In order for this diode to be effective it must be connected
directly to the sources of both the top and bottom sides output MOSFETs. A fast or
ultra fast recovery diode that can sustain the entire VPP-VNN voltage should be
used here. In most applications a 150V or greater diode must be used.
Gate resistor for the output MOSFET for the switchmode power supply. Controls
the rise time, fall time, and reduces ringing for the gate of the output MOSFET for
the switchmode power supply.
RPG
QB
Output MOSFET for the switchmode power supply to generate the VN10. This
output MOSFET must be a P channel device.
DSW
Flywheel diode for the internal VN10 buck converter. This diode also prevents
VN10SW from going more than one diode drop negative with respect to VNN. This
diode should be a Shottky or ultrafast rectifier.
LSW
VN10 generator filter inductor. This inductor should be sized appropriately so that
LSW does not saturate, and VN10 does not overshoot with respect to VNN during
TK2150 turn on.
CSW
VN10 generator filter capacitors. The high frequency capacitor (0.1uF) must be
located close to the VN10 pins (pin 41 and 43 of the TP2150) to maximize device
performance. The bulk capacitor (100uF) should be sized appropriately such that
the VN10 voltage does not overshoot with respect to VNN during TK2150 turn on.
VN10 generator feedback resistor. This resistor sets the nominal VN10 voltage.
With RSWFB equal to 1kΩ, the VN10 voltage generated will typically be 10V above
VNN.
RSWFB
CSWFB
VN10 generator feedback capacitor. This capacitor, in conjunction with RSWFB, filters
the VN10 feedback signal such that the loop is unconditionally stable.
12
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
Typical Performance Characteristics
E fficiency versus Output P ow er
THD +N versus O utput P ower
100
90
10
5
f = 1kHz
B B M = 40nS
V s = + 40V
R L
=
8 Ω
R L
2
1
80
70
60
50
40
30
20
10
0
B W
= 22Hz - 20kHz(A E S 17)
=
6 Ω
0.5
0.2
0.1
%
0.05
R L = 8Ω
0.02
0.01
R L = 6Ω
V s
=
+ 40V
40n S
0.005
BBM
=
BW
T H D +N
= 22H z-20kH z(AES 17)
0.002
0.001
<
10%
0
20
40
60
80
100
120
140
160
1
2
5
10
20
50
100
200
W
W
IntermodulationPerformance
IntermodulationPerformance
+0
+0
RL = 8Ω
RL = 6Ω
19kHz, 20kHz, 1:1
0dBr = 12Vrms
32k FFT
19kHz, 20kHz, 1:1
0dBr = 12Vrms
32k FFT
-20
-20
-40 FS = 96kHz
VS = +40V
-40
-60
F S = 96kHz
VS = +40V
d
B
r
d
B
r
BW = 22Hz - 80kHz
BW = 22Hz - 80kHz
-60
-80
-80
A
A
-100
-120
-100
-120
-140
-140
20
50
100
200
500
1k
2k
5k
10k
30k
20
50
100
200
500
1k
2k
5k
10k
30k
Hz
Hz
N o is e F lo o r
C hanne l S e p aratio n ve rs us F re q ue nc y
-7 0
-7 5
-8 0
+0
V s
=
+4 0 V
P out
P out
=
=
24W
18W
@
@
6 Ω
8 Ω
B B M
=
4 0 n S
-10
3 2 k F F T
F s
B W
0dB r = 12V rm s
=
4 8 k H z
2 2 H z -2 0 k H z (A E S 1 7 )
V s
=
+40V
= 22Hz - 20k H z (A E S 17)
-20
-30
-40
-50
-60
-70
-80
-90
=
B W
-8 5
-9 0
d
B
r
d
B
V
-9 5
-1 0 0
-1 0 5
-1 1 0
-1 1 5
A
R L = 6Ω
R L = 8Ω
-100
-1 2 0
30
50
100
200
500
1k
2k
5k
10k 20k
2 0
5 0
1 0 0
2 0 0
5 0 0
Hz
1k
2k
5k
1 0 k 2 0 k
Hz
13
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
Typical performance Characteristics
THD+N versus Frequency versus Break Before Make
THD+N versus Frequency versus Break Before Make
10
10
RL = 6Ω
RL = 8Ω
5
5
Pout = 25W / Channel
Vs = +40V
Pout = 20W/ Channel
2
1
2
1
Vs = +40V
BW = 22Hz-20kHz(AES17)
BW = 22Hz-20kHz(AES17)
0.5
0.5
0.2
0.1
0.2
0.1
0.05
%
%
%
%
0.05
BBM = 80nS
BBM = 80nS
0.02
0.01
0.005
0.02
0.01
BBM = 40nS
0.005
BBM = 40nS
0.002
0.001
0.002
0.001
20
50
100
200
500
Hz
1k
2k
5k
10k 20k
20
50
100
200
500
Hz
1k
2k
5k
10k 20k
THD+N versus Frequency versus Bandwidth
THD+N versus Frequency versus Bandwidth
10
5
10
5
RL = 8Ω
RL = 6Ω
Pout = 20W/ Channel
Vs = +40V
BBM = 40nS
Pout = 25W/ Channel
Vs = +40V
2
1
2
1
BBM = 40nS
0.5
0.5
0.2
0.1
0.2
0.1
%
0.05
0.05
BW = 30kHz
BW = 30kHz
0.02
0.01
0.02
0.01
BW = 20kHz(AES17)
BW = 20kHz(AES17)
0.005
0.005
0.002
0.001
0.002
0.001
20
50
100
200
500
Hz
1k
2k
5k
10k 20k
20
50
100
200
500
Hz
1k
2k
5k
10k 20k
THD+N versus Output Power versus Supply Voltage
THD+N versus Output Power Versus Supply Voltage
10
5
10
5
RL = 8Ω
RL = 6Ω
Vs = +35V, +40V, +45V
Vs = +35V, +40V, +45V
f = 1kHz
f = 1kHz
2
1
2
BBM = 40nS
BBM = 40nS
1
BW = 22Hz - 20kHz(AES17)
BW = 22hZ - 20kHz(AES17)
NOTE: +45V test uses RFBC=11kΩ
(see Application/ Test Circuit)
0.5
0.5
NOTE: +45V test uses RFBC=11kΩ
(see Application/ Test Circuit)
0.2
0.1
0.2
0.1
%
0.05
0.05
0.02
0.01
0.02
0.01
0.005
0.005
0.002
0.001
0.002
0.001
1
2
5
10
20
50
100
200
1
2
5
10
20
50
100
200
W
W
14
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
Application Information
Figure 1 is a simplified diagram of one channel (Channel 1) of a TK2150 amplifier to assist in
understanding its operation.
TC2001
TP2150
BBM0 26
BBM1
51 OCS1HP
VPP
25
OVER
DB
CB
CS
CURRENT
DETECTION
RS
OAOUT1 22
VN10
RB
OCS1HN
50
57 VBOOT1
RI
RF
CI
V5
QO
RG
RG
INV1 23
ROFB
+
+
-
Y1
9
17
16
48 HO1
47 HO1COM
CBAUX
0.1uF
+
CHBR
AGND
V5
VN10
OUTPUT
FILTER
QO
Processing
&
Modulation
10
Y1B
44 LO1
RL
ROFA
45
LO1COM
COF
2.5V
V5
54 OCS1LP
Offset Trim
Circuit
OVER
CURRENT
DETECTION
RS
CA
5V
53 OCS1LN
BIASCAP
1
VNN
41,43
VN10
VN10
CS
CSW
VNN
VNN
42
6
MUTE 24
REF 15
OCR1
7
VNN
V5
5V
CS
5
AGND
RREF
RVNN1
16
OCR1
COCR
VNNSENSE 17
VPPSENSE 19
V5
VNN
VPP
OVER/
OVER
CURRENT
DETECTION
UNDER
VOLTAGE
DETECTION
ROCR
RVPP1
RFBA
RFBA
RFBC
RVNN2
RVPP1
6
7
FBKOUT1
FBKGND1
V5
V5
RFBC
CFB
V5
21
RFBB
RFBB
5V
CS
8
HMUTE
AGND 20
Analog
Ground
F. BEAD
Power Ground
Figure 1: Simplified TK2150 Amplifier
TK2150 Basic Amplifier Operation
The audio input signal is fed to the processor internal to the TC2001, where a switching pattern is
generated. The average idle (no input) switching frequency is approximately 700kHz. With an
input signal, the pattern is spread spectrum and varies between approximately 200kHz and
1.5MHz depending on input signal level and frequency. Complementary copies of the switching
pattern is output through the Y1 and Y1B pins on the TC2001. These switching patterns are input
to the TP2150 where they are level-shifted by the MOSFET drivers and then output to the gates
(HO1 and LO1) of external power MOSFETs that are connected as a half bridge. The output of
the half bridge is a power-amplified version of the switching pattern that switches between VPP
and VNN. This signal is then low-pass filtered to obtain an amplified reproduction of the audio
input signal.
The TC2001 processor is operated from a 5-volt supply. In the generation of the switching
patterns for the output MOSFETs, the processor inserts a “break-before-make” dead time
between the turn-off of one transistor and the turn-on of the other in order to minimize shoot-
through currents in the external MOSFETs. The dead time can be programmed by setting the
break-before-make control bits, BBM1 and BBM0. Feedback information from the output of the
half-bridge is supplied to the processor via FBKOUT1. Additional feedback information to
account for ground bounce is supplied via FBKGND1.
15
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
The MOSFET drivers in the TP2150 are operated from voltages obtained from VN10 and
LO1COM for the low-side driver, and VBOOT1 and HO1COM for the high-side driver. VN10 must
be a regulated 10V above VNN.
N-Channel MOSFETs are used for both the top and bottom of the half bridge. The gate resistors,
RG, are used to control MOSFET slew rate and thereby minimize voltage overshoots.
Circuit Board Layout
The TK2150 is a power (high current) amplifier that operates at relatively high switching
frequencies. The output of the amplifier switches between VPP and VNN at high speeds while
driving large currents. This high-frequency digital signal is passed through an LC low-pass filter
to recover the amplified audio signal. Since the amplifier must drive the inductive LC output filter
and speaker loads, the amplifier outputs can be pulled above the supply voltage and below
ground by the energy in the output inductance. To avoid subjecting the TK2150 to potentially
damaging voltage stress, it is critical to have a good printed circuit board layout. It is
recommended that Tripath’s layout and application circuit be used for all applications and only be
deviated from after careful analysis of the effects of any changes. Please refer to the TK2150
evaluation board document, RB-TK2150, available on the Tripath website, at www.tripath.com.
The trace that connects the source of the top side output MOSFET to the drain of the bottom side
output MOSFET is very important. This connection should be as wide as possible and as short
as possible. Also a jumper wire of 16 gauge or more can by used in parallel with the trace to
reduce any trace resistance or inductance. Any resistance or inductance on this trace can cause
the switching output to over/undershoot potentially causing damage to both the TP2150 and the
output MOSFETs.
The following components are important to place near either their associated TK2150 or output
MOSFET pins. The recommendations are ranked in order of layout importance, either for proper
device operation or performance considerations.
-
The capacitors, CHBR, provide high frequency bypassing of the amplifier power supplies
and will serve to reduce spikes across the supply rails. Please note that both mosfet
half-bridges must be decoupled separately. In addition, the voltage rating for CHBR
should be at least 150V as this capacitor is exposed to the full supply range, VPP-VNN.
-
CFB removes very high frequency components from the amplifier feedback signals and
lowers the output switching frequency by delaying the feedback signals. In addition,
the value of CFB is different for channel 1 and channel 2 to keep the average switching
frequency difference greater than 40kHz. This minimizes in-band audio noise. Locate
these capacitors as close to their respective TC2001 pin as possible.
-
DD and DS should be placed as close to the drain and source of the output MOSFETs
as possible. DD should be connected directly from the drain of the top side MOSFET to
the drain of the bottom side MOSFET. DS should be connected directly from the source
of the top side MOSFET to the source of the bottom side MOSFET. DD protects the
bottom side output MOSFET from output over/undershoots. DS protects the top side
output MOSFET from output over/undershoots. The over/undershoots are very high
speed transients, if DD and DS are placed too far away from the MOSFETs they will be
ineffective.
-
-
To minimize noise pickup and minimize THD+N, RFBC should be located as close to the
TC2001 as possible. Make sure that the routing of the high voltage feedback lines is
kept far away from the input op amps or significant noise coupling may occur. It is best
to shield the high voltage feedback lines by using a ground plane around these traces
as well as the input section. The feedback and feedback ground traces should be
routed together in parallel.
CB, CSW provides high frequency bypassing for the VN10 and bootstrap supplies. Very
high currents are present on these supplies.
16
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
In general, to enable placement as close to the TK2150, and minimize PCB parasitics, the
capacitors CFB, CB and CSW should be surface mount types, located on the “solder” side of the
board.
Some components are not sensitive to location but are very sensitive to layout and trace routing.
-
-
-
To maximize the damping factor and reduce distortion and noise, the modulator
feedback connections should be routed directly to the pins of the output inductors. LO.
The output filter capacitor, CO, and zobel capacitor, CZ, should be star connected with
the load return. The output ground feedback signal should be taken from this star point.
The modulator feedback resistors, RFBA and RFBB, should all be grounded and attached
to 5V together. These connections will serve to minimize common mode noise via the
differential feedback.
-
The feedback signals that come directly from the output inductors are high voltage and
high frequency in nature. If they are routed close to the input nodes, IN1 and IN2, the
high impedance inverting opamp pins will pick up noise. This coupling will result in
significant background noise, especially when the input is AC coupled to ground, or an
external source such as a CD player or signal generator is connected. Thus, care
should be taken such that the feedback lines are not routed near any of the input
section.
-
To minimize the possibility of any noise pickup, the trace lengths of IN1 and IN2 should
be kept as short as possible. This is most easily accomplished by locating the input
resistors, RI and the input stage feedback resistors, RF as close to the TC2001 as
possible. In addition, the offset trim resistor, ROFB, which connects to either IN1, or IN2,
should be located close to the TC2001 input section.
TK2150 Grounding
Proper grounding techniques are required to maximize TK2150 functionality and performance.
Parametric parameters such as THD+N, Noise Floor and Crosstalk can be adversely affected if
proper grounding techniques are not implemented on the PCB layout. The following discussion
highlights some recommendations about grounding both with respect to the TK2150 as well as
general “audio system” design rules.
The TK2150 is divided into three sections: the input section, which is the TC2001, the MOSFET
driver section, which is the TP2150, and the output (high voltage) section, which is the output
MOSFETs. On the TK2150 evaluation board, the ground is also divided into distinct sections,
Analog Ground (AGND) and Power Ground (PGND). To minimize ground loops and keep the
audio noise floor as low as possible, the two grounds must be only connected at a single point.
Depending on the system design, the single point connection may be in the form of a ferrite bead
or a PCB trace.
The analog ground, must be connected to pin 20 on the TC2001 and pins 2 and 5 on the TP2150.
The ground for the V5 power supply should connect directly to pin 20 of the TC2001. Additionally,
any external input circuitry such as preamps, or active filters, should be referenced to pin 20 on
the TC2001. Special care must be used when connecting the NC pins of the TP2150 in order to
achieve the best noise performance. Pins 1, 3, 4, 8, 11, 12, 18, 19, 20, 22, 23, 24, 25, 61, 62, 63,
64 should be tied to Analog Ground. All of the other NC pins of the TP2150 should be left
floating.
For the power section, Tripath has traditionally used a “star” grounding scheme. Thus, the load
ground returns and the power supply decoupling traces are routed separately back to the power
supply. In addition, any type of shield or chassis connection would be connected directly to the
ground star located at the power supply. These precautions will both minimize audible noise and
enhance the crosstalk performance of the TK2150.
17
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
The TC2001 incorporates a differential feedback system to minimize the effects of ground bounce
and cancel out common mode ground noise. As such, the feedback from the output ground for
each channel needs to be properly sensed. This can be accomplished by connecting the output
ground “sensing” trace directly to the star formed by the output ground return, output capacitor,
CO, and the zobel capacitor, CZ. Refer to the Application / Test Circuit for a schematic
description.
TK2150 Amplifier Gain
The gain of the TK2150 is the product of the input stage gain and the modulator gain for the
TC2001. Please refer to the sections, Input Stage Design, and Modulator Feedback Design, for a
complete explanation of how to determine the external component values.
A
A
VTK2150 = AVINPUTSTAG
E
* AV MODULATOR
R
F
R
FBC * (RFBA + RFBB )
VTK2150 ≈ −
+ 1
R
I
R
FBA * RFBB
For example, using a TC2001 with the following external components,
RI = 20kΩ
RF = 30.1kΩ
R
FBA = 1kΩ
RFBB = 1.1kΩ
RFBC = 10.0kΩ
20k Ω 10.0k Ω * (1.0k Ω + 1.1k Ω)
V
V
A
VTK2150 ≈ −
+ 1 = -13.35
30.1k Ω
1.0k Ω * 1.1k Ω
Input Stage Design
The TC2001 input stage is configured as an inverting amplifier, allowing the system designer
flexibility in setting the input stage gain and frequency response. Figure 2 shows a typical
application where the input stage is a constant gain inverting amplifier. The input stage gain
should be set so that the maximum input signal level will drive the input stage output to 4Vpp.
The gain of the input stage, above the low frequency high pass filter point, is that of a simple
inverting amplifier:
R
R
F
A
VINPUTSTAG E = −
I
18
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
TC2001
OAOUT1
22
V5
CI
RI
RF
INV1 23
+
-
+
INPUT1
AGND
BIASCAP
V5
CI
RI
+
-
28
27
INV2
+
RF
INPUT2
AGND
OAOUT2
Figure 2: TC2001 Input Stage
Input Capacitor Selection
CIN can be calculated once a value for RIN has been determined. CIN and RIN determine the input
low-frequency pole. Typically this pole is set at 10Hz. CIN is calculated according to:
CIN = 1 / (2π x FP x RIN)
where: RIN = Input resistor value in ohms
FP = Input low frequency pole (typically 10Hz)
Modulator Feedback Design
The modulator converts the signal from the input stage to the high-voltage output signal. The
optimum gain of the modulator is determined from the maximum allowable feedback level for the
modulator and maximum supply voltages for the power stage. Depending on the maximum
supply voltage, the feedback ratio will need to be adjusted to maximize performance. The values
of RFBA, RFBB and RFBC (see explanation below) define the gain of the modulator. Once these
values are chosen, based on the maximum supply voltage, the gain of the modulator will be fixed
even with as the supply voltage fluctuates due to current draw.
For the best signal-to-noise ratio and lowest distortion, the maximum modulator feedback voltage
should be approximately 4Vpp. The modulator feedback resistor RFBC should be adjusted so that
the modulator feedback voltage is approximately 4Vpp. This will keep the gain of the modulator
as low as possible and still allow headroom so that the feedback signal does not clip the
modulator feedback stage. Increasing the value of RFBC will increase the modulator gain.
Sometimes increasing the value of RFBC may be necessary to achieve full power for the amplifier
since the input stage for the TC2001 will clip at approximately 4Vpp. This will ensure that the
input stage doesn’t clip before the output stage.
Figure 3 shows how the feedback from the output of the amplifier is returned to the input of the
modulator. The input to the modulator (FBKOUT1/FBKGND1 for channel 1) can be viewed as
inputs to an inverting differential amplifier. RFBA and RFBB bias the feedback signal to
approximately 2.5V and RFBC scales the large OUT1/OUT2 signal to down to 4Vpp.
19
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
1/2 TC2001
V5
RFBA
RFBA
RFBC
FBKOUT1
Processing
&
Modulation
28
27
OUT1
FBKGND1
OUT1 GROUND
RFBC
RFBB
RFBB
AGND
Figure 3: Modulator Feedback
The modulator feedback resistors are:
R
R
FBA = User specified, typically 1KΩ
R
FBA * VPP
FBB
=
(VPP - 4)
R
FBA * VPP
R
A
FBC
=
4
R
FBC * (RFBA + RFBB )
V - MODULATOR
≈
+ 1
R
FBA * RFBB
The above equations assume that VPP=|VNN|.
For example, in a system with VPPMAX=40V and VNNMAX=-40V,
RFBA = 1kΩ, 1%
R
FBB = 1.097kΩ, use 1.1kΩ, 1%
RFBC = 10.0kΩ, use 10.0kΩ, 1%
The resultant modulator gain is:
10.0k Ω * (1.0k Ω + 1.1k Ω)
A
V - MODULATOR
≈
+ 1 = 20.09V/V
1.0k Ω * 1.1k Ω
Mute
When a logic high signal is supplied to MUTE, both amplifier channels are muted (both high- and
low-side transistors are turned off). When a logic level low is supplied to MUTE, both amplifiers
20
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
are fully operational. There is a delay of approximately 200 milliseconds between the de-
assertion of MUTE and the un-muting of the TK2150.
Sleep
When a logic high signal is supplied to SLEEP, the two channels on the TP2150 will be shutdown
and the outputs will be muted. When a logic level low is supplied to SLEEP, both channels are
fully operational.
Turn-on & Turn-off Noise
If turn-on or turn-off noise is present in a TK2150 amplifier, the cause is frequently due to other
circuitry external to the TK2150. While the TK2150 has circuitry to suppress turn-on and turn-off
transients, the combination of the power supply and other audio circuitry with the TK2150 in a
particular application may exhibit audible transients. One solution that will completely eliminate
turn-on and turn-off pops and clicks is to use a relay to connect/disconnect the amplifier from the
speakers with the appropriate timing at power on/off. The relay can also be used to protect the
speakers from a component failure (e.g. shorted output MOSFET), which is a protection
mechanism that some amplifiers have. Circuitry external to the TK2150 would need to be
implemented to detect these failures.
DC Offset
While the DC offset voltages that appear at the speaker terminals of a TK2150 amplifier are
typically small, Tripath recommends that any offsets during operation be nulled out of the
amplifier with a circuit like the one shown connected to IN1 and IN2 in the Test/Application
Circuit.
It should be noted that the DC voltage on the output of a TK2150 amplifier with no load in mute
will not be zero. This offset does not need to be nulled. The output impedance of the amplifier in
mute mode is approximately 10KΩ. This means that the DC voltage drops to essentially zero
when a typical load is connected.
HMUTE
The HMUTE pin on the TC2001 is a 5V logic output that indicates various fault conditions within
the device. These conditions include: over-current, overvoltage and undervoltage. The HMUTE
output is capable of directly driving an LED through a series 2kΩ resistor.
Over-current Protection
The TK2150 has over-current protection circuitry to protect itself and the output transistors from
short-circuit conditions. The TK2150 uses the voltage across a resistor RS (measured via
OCS1HP, OCS1HN, OCS1LP and OCS1LN of the TP2150) that is in series with each output
MOSFET to detect an over-current condition. RS and ROCR are used to set the over-current
threshold. The OCS pins must be Kelvin connected for proper operation. See “Circuit Board
Layout” in Application Information for details.
When the voltage across ROCR becomes greater than VTOC (approximately 1.0V) the TC2001 will
shut off the output stages of its amplifiers. The occurrence of an over-current condition is latched
in the TK2150 and can be cleared by toggling the MUTE input or cycling power.
Setting Over-current Threshold
RS and ROCR determine the value of the over-current threshold, ISC:
ISC = 3580 x (VTOC – IBIAS * ROCR)/(R OCR * RS)
R
OCR = (3580 x VTOC)/(ISC * RS+3580 * IBIAS
)
where:
21
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
RS and ROCR are in Ω
V
TOC = Over-current sense threshold voltage (See Electrical Characteristics Table)
= 1.0V typically
IBIAS = 20uA
For example, to set an ISC of 7A, ROCR = 30.1kΩ and RS will be 10mΩ.
As high-wattage resistors are usually only available in a few low-resistance values (10mΩ, 25mΩ
and 50mΩ), ROCR can be used to adjust for a particular over-current threshold using one of these
values for RS.
It should be noted that the addition of the bulk CHBR capacitor shown in the Application / Test
Diagram will increase the ISC level. Thus, it will be larger than the theoretical value shown above.
Once the designer has settled on a layout and specific CHBR value, the system ISC trip point can
be adjusted by increasing the ROCR value. The ROCR should be increased to a level that allows
expected range of loads to be driven well into clipping without current limiting while still protecting
the output MOSFETs in case of a short circuit condition.
Auto Recovery Circuit for Overcurrent Fault Condition
If an overcurrent fault condition occurs the HMUTE pin (pin 8 of the TC2001) will be latched high
and the amplifier will be muted. The amplifier will remain muted until the MUTE pin (pin 24 of the
TC2001) is toggled high and then low or the power supplies are turned off and then on again.
The circuit shown below in Figure 4 is a circuit that will detect if HMUTE is high and then toggle
the mute pin high and then low, thus resetting the amplifier. The LED, D1 will turn on when
HMUTE is high. The reset time has been set for approximately 2.5 seconds. The duration of the
reset time is controlled by the RC time constant set by R306 and C311. To increase the reset,
time increase the value of C311. To reduce the reset time, reduce the value of C311. Please
note that this circuit is optional and is not included on the RB-TK2150 evaluation boards.
V5
R309
1kΩ , 5%
D1
LED
R311
R306
510kΩ , 5%
R307
10kΩ , 5%
R308
10kΩ , 5%
Q305
2N3906
1kΩ , 5%
R311
1kΩ , 5%
MUTE
Pin 24
C311
10uF, NP
Q302
2N3904
Q303
2N7002
R311
1kΩ , 5%
Jumper
HMUTE
Pin 8
Q304
2N3904
R310
1kΩ , 5%
remove jumper to
enable mute
AGND
Figure 4: Overcurrent Autorecovery Circuit
Over- and Under-Voltage Protection
The TC2001 senses the power rails through external resistor networks connected to VNNSENSE
and VPPSENSE. The over- and under-voltage limits are determined by the values of the resistors
in the networks, as described in the table “Test/Application Circuit Component Values”. If the
supply voltage falls outside the upper and lower limits determined by the resistor networks, the
TC2001 shuts off the output stages of the amplifiers. The removal of the over-voltage or under-
voltage condition returns the TK2150 to normal operation. Please note that trip points specified in
the Electrical Characteristics table are at 25°C and may change over temperature.
The TC2001 has built-in over and under voltage protection for both the VPP and VNN supply
rails. The nominal operating voltage will typically be chosen as the supply “center point.” This
allows the supply voltage to fluctuate, both above and below, the nominal supply voltage.
22
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
VPPSENSE (pin 19) performs the over and undervoltage sensing for the positive supply, VPP.
VNNSENSE (pin 17) performs the same function for the negative rail, VNN. When the current
through RVPPSENSE (or RVNNSENSE) goes below or above the values shown in the Electrical
Characteristics section (caused by changing the power supply voltage), the TK2150 will be
muted. VPPSENSE is internally biased at 2.5V and VNNSENSE is biased at 1.25V.
Once the supply comes back into the supply voltage operating range (as defined by the supply
sense resistors), the TK2150 will automatically be unmuted and will begin to amplify. There is a
hysteresis range on both the VPPSENSE and VNNSENSE pins. If the amplifier is powered up in
the hysteresis band the TK2150 will be muted. Thus, the usable supply range is the difference
between the over-voltage turn-off and under-voltage turn-off for both the VPP and VNN supplies.
It should be noted that there is a timer of approximately 200mS with respect to the over and
under voltage sensing circuit. Thus, the supply voltage must be outside of the user defined
supply range for greater than 200mS for the TK2150 to be muted.
Figure 5 shows the proper connection for the Over / Under voltage sense circuit for both the
VPPSENSE and VNNSENSE pins.
V5
VNN
TC2001
RVNN2
RVNN1
17
19
VNNSENSE
V5
VPP
RVPP1
RVPP1
VPPSENSE
Figure 5: Over / Under voltage sense circuit
The equation for calculating RVPP1 is as follows:
VPP
VPPSENSE
R
VPP1 =
I
Set RVPP2 = RVPP1 .
The equation for calculating RVNNSENSE is as follows:
VNN
R
VNN1 =
I
VNNSENSE
Set
.
R
VNN2 = 3 × RVNN1
IVPPSENSE or IVNNSENSE can be any of the currents shown in the Electrical Characteristics
table for VPPSENSE and VNNSENSE, respectively.
The two resistors, RVPP2 and RVNN2 compensate for the internal bias points. Thus, RVPP1 and
RVNN1 can be used for the direct calculation of the actual VPP and VNN trip voltages without
considering the effect of RVPP2 and RVNN2
.
23
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
Using the resistor values from above, the actual minimum over voltage turn off points will be:
VPP MIN_OV_TUR N_OFF = RVPP1 × IVPPSENSE (MIN_OV_TU RN_OFF)
VNN MIN_OV_TUR N_OFF = −(RVNN1 × IVNNSENSE (MIN_OV_TU RN_OFF)
)
The other three trip points can be calculated using the same formula but inserting the appropriate
IVPPSENSE (or IVNNSENSE) current value. As stated earlier, the usable supply range is the difference
between the minimum overvoltage turn off and maximum under voltage turn-off for both the VPP
and VNN supplies.
VPP RANGE = VPP MIN_OV_TUR N_OFF - VPP MAX_UV_TUR N_OFF
VNN RANGE = VNN MIN_OV_TUR N_OFF - VNN MAX_UV_TUR N_OFF
VN10 Supply and Switch Mode Power Supply Controller
VN10 is an additional supply voltage required by the TP2150. VN10 must be 10 volts more
positive than the nominal VNN. VN10 must track VNN. Generating the VN10 supply requires
some care.
The proper way to generate the voltage for VN10 is to use a 10V-postive supply voltage
referenced to the VNN supply. The TP2150 has an internal switch mode power supply
controller which generates the necessary floating power supply for the MOSFET driver stage
in the TP2150 (nominally 10V with the external components shown in Application / Test
Circuit). The SMPSO pin (pin 60) provides a switching output waveform to drive the gate of a
P channel MOSFET. The source of the P channel MOSFET should be tied to power ground
and the drain of the MOSFET should be tied to the VN10 through a 100uH inductor. The
performance curves shown in this datasheet as well as the efficiency measurements were
done using the internal VN10 generator. Tripath recommends using the internal VN10
generator to power the TP2150. Figure 6 shows how the VN10 generator should be
connected.
TP2150
RSWFB 1kΩ
59 SW-FB
CSWFB
VN10
0.1uF
Switchmode
LSW
QP
Power Supply
VNN
RPG 10Ω
100uH
60
SMPSO
VN10
+
DSW
B1100DICT
CSW
0.1uF
CSW
100uF
VNN
Figure 6: VN10 Generator
In some cases, though, a designer may wish to use an external VN10 generator. The
specification for VN10 quiescent current (200mA typical, 250mA maximum) in the Electrical
Characteristics section states the amount of current needed when an external floating supply is
used. If the internal VN10 generator is not used, Tripath recommends shorting SMPSO(pin 60) to
VNN(pin 42) and SW-FB(pin 59) to VNN(pin 42).
One apparent method to generate the VN10 supply voltage is to use a negative IC regulator to
drop PGND down to 10V (relative to VNN). This method will not work since negative regulators
24
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
only sink current into the regulator output and will not be capable of sourcing the current required
by VN10. Furthermore, problems can arise since VN10 will not track movements in VNN. The
external VN10 supply must be able to source a maximum of 250mA into the VN10 pin. Thus, a
positive supply must be used and must be referenced to the VNN rail. If the external VN10
supply does not track fluctuations in the VNN supply or is not able to source current into the VN10
pin, the TP2150 will not work and can also become permanently damaged.
Figure 7 shows the correct way to power the TP2150:
VPP
V5
VPP
5V
AGND
VN10
PGND
VNN
10V
VNN
F. BEAD
Figure 7: Proper Power Supply Connection
Output Transistor Selection
The key parameters to consider when selecting what MOSFET to use with the TK2150 are drain-
source breakdown voltage (BVdss), gate charge (Qg), and on-resistance (RDS(ON)).
The BVdss rating of the MOSFET needs to be selected to accommodate the voltage swing
between VSPOS and VSNEG as well as any voltage peaks caused by voltage ringing due to
switching transients. With a ‘good’ circuit board layout, a BVdss that is 50% higher than the VPP
and VNN voltage swing is a reasonable starting point. The BVdss rating should be verified by
measuring the actual voltages experienced by the MOSFET in the final circuit.
Ideally a low Qg (total gate charge) and low RDS(ON) are desired for the best amplifier
performance. Unfortunately, these are conflicting requirements since RDS(ON) is inversely
proportional to Qg for a typical MOSFET. The design trade-off is one of cost versus performance.
A lower RDS(ON) means lower I2RDS(ON) losses but the associated higher Qg translates into higher
switching losses (losses = Qg x 10 x 1.2MHz). A lower RDS(ON) also means a larger silicon die
and higher cost. A higher RDS(ON) means lower cost and lower switching losses but higher I2RDSON
losses.
Gate Resistor Selection
The gate resistors, RG, are used to control MOSFET switching rise/fall times and thereby
minimize voltage overshoots. They also dissipate a portion of the power resulting from moving
the gate charge each time the MOSFET is switched. If RG is too small, excessive heat can be
generated in the driver. Large gate resistors lead to slower MOSFET switching, which requires a
larger break-before-make (BBM) delay.
Break-Before-Make (BBM) Timing Control
25
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
The half-bridge power MOSFETs require a deadtime between when one transistor is turned off
and the other is turned on (break-before-make) in order to minimize shoot through currents. The
TC2001 has BBM0 and BBM1 that are logic inputs (connected to logic high or pulled down to
logic low) that control the break-before-make timing of the output transistors according to the
following table.
BBM1
BBM0
Delay
120 ns
80 ns
40 ns
0 ns
0
0
1
1
0
1
0
1
Table 1: BBM Delay
The tradeoff involved in making this setting is that as the delay is reduced, distortion levels
improve but shoot-through and power dissipation increase. All typical curves and performance
information were done with using the 40ns BBM setting. The actual amount of BBM required is
dependent upon other component values and circuit board layout, the value selected should be
verified in the actual application circuit/board. It should also be verified under maximum
temperature and power conditions since shoot-through in the output MOSFETs can increase
under these conditions, possibly requiring a higher BBM setting than at room temperature.
Recommended MOSFETs
The following devices are capable of achieving full performance, both in terms of distortion and
efficiency, for the specified load impedance and voltage range.
Device Information – Recommended MOSFETs
Part Number
IRF520N
Manufacturer
BVDSS (V)
100
ID (A)
9.7
12.8
14
17
23
Qg (nC)
25(max.) 0.20 (max.)
12
15.5
RDS(on) Ω
(
)
PD (W)
48
65
60
70
Package
TO220
TO220
TO220
TO220
TO220
TO220
International Rectifier
Fairchild Semiconductor
ST Microelectronics
International Rectifier
Philips Semiconductor
ST Microelectronics
FQP13N10
STP14NF10
IRF530N
BUK7575-100A
STP24NF10
100
100
100
100
0.142
0.16
37(max.) 0.09 (max.)
25
30
0.064
0.055
99
85
100
26
Note: The devices are listed in ascending current capability not in order of recommendation.
The following information represents qualitative data from system development using the TK2150
and the associated MOSFETs. Recommendations such as maximum supply voltages and gate
resistor values are dependent on the PCB layout and component location. The gate resistor
values were chosen to achieve about 18-80mA of idle current from the VPP supply. This value of
supply current is a good compromise between low power efficiency and high frequency THD+N
performance. As shown in Table 2 below, increasing the gate resistor value will improve high
frequency THD+N performance at the expense of idle current draw. The BBM setting was 40nS
in all cases. It should be understood that different MOSFETs will have different characteristics
and will require some adjustment to the gate resistor to achieve the same idle current.
26
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
10THD+N versus Frequency versus Gate Resistance
10 THD+N versus Frequency versus Gate Resistance
RL = 6Ω
RL = 8Ω
5
5
Pout = 25W /Channel
FET's = FQP13N10
f = 1kHz
Pout = 20W /Channel
FET's = FQP13N10
2
2
1
f = 1kHz
1
BBM = 40nS
BBM = 40nS
VS = +40V
BW = 22Hz - 22kHz
V S = +40V
0.5
0.5
BW = 22Hz - 22kHz
RG = 22 Ω
RG = 33 Ω
RG = 22 Ω
RG = 33 Ω
RG = 46.4 Ω
0.2
0.1
0.2
0.1
%
%
RG = 46.4 Ω
0.05
0.05
0.02
0.01
0.02
0.01
0.005
0.005
0.002
0.001
0.002
0.001
20
50
100
200
500
1k
2k
5k
10k 20k
20
50
100
200
500
Hz
1k
2k
5k
10k 20k
6 ohm and 8 ohm plots of THD+N versus Frequency for variousHzgate resistor values
6 ohms 8 ohms
22 ohms
33 ohms
18mA
20mA
18mA
20mA
80mA
46.4 ohms 80mA
Table 2: Idle current draw for VPP with various gate resistor values
Application Information – Recommended MOSFETs
Part Number Recommended Max
Supply Voltage
Typical Load at
Maximum Supply
8 ohm SE
Recommended
Gate Resistor
22 ohms
Other applications
IRF520N
FQP13N10
+/-45V
+/-45V
Only for 8 ohm SE Loads
6 ohm SE
33 ohms
6 ohm BR at +/25V
8 ohm BR at +/-33V
6 ohm BR at +/-25V
8 ohm BR at +/-33V
6 ohm BR at +/-33V
4 ohm BR at +/-33V
4 ohm BR at +/-35V
STP14NF10
+/-45V
6 ohm SE
33 ohms
IRF530N
BUK7575-100A
STP24NF10
+/-45V
+/-45V
+/-45V
4 ohm SE / 8 ohm BR
4 ohm SE / 6 ohm BR
4 ohm SE / 6 ohm BR
15 ohms
15 ohms
10 ohms
SE stands for Single Ended Outputs and BR stands for Bridged Output
MOSFETs Under Evaluation
The following MOSFETs appear to be suitable for use with the TK2150, and we are waiting for
samples to evaluate. Most of these devices come from the same “family” or generation, as other
recommended MOSFETs. However, experience tells us that we cannot recommend any devices
until we have received samples and fully tested them.
Device Information – MOSFETs Under Evaluation
Part Number
FQP14N15
FQP16N15
FDP2572
Manufacturer
BVDSS (V)
150
ID (A)
14.4
16.4
29
Qg (nC)
18
23
27
18.5
RDS(on) Ω
(
)
PD (W)
104
108
135
95
Package
T0220
TO220
TO247
TO220
Fairchild Semiconductor
Fairchild Semiconductor
Fairchild Semiconductor
Fairchild Semiconductor
.164
0.123
0.045
0.032
150
150
100
FDP3682
32
Note: The devices are listed in ascending current capability not in order of recommendation.
Output Filter Design
One advantage of Tripath amplifiers over PWM solutions is the ability to use higher-cutoff-
frequency filters. This means load-dependent peaking/droop in the 20kHz audio band potentially
caused by the filter can be made negligible. This is especially important for applications where
the user may select a 6-Ohm or 8-Ohm speaker. Furthermore, speakers are not purely resistive
27
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
loads and the impedance they present changes over frequency and from speaker model to
speaker model.
Tripath recommends designing the filter as a 2nd order, 100kHz LC filter. Tripath has obtained
good results with LF = 18uH and CF = 0.15uF.
The core material of the output filter inductor has an effect on the distortion levels produced by a
TK2150 amplifier. Tripath recommends low-mu type-2 iron powder cores because of their low
loss and high linearity (available from Micrometals, www.micrometals.com). The specific core
used on the EB-TK2150 was a T106-2 wound with 44 turns of 22AWG wire.
Tripath also recommends that an RC damper be used after the LC low-pass filter. No-load
operation of a TK2150 amplifier can create significant peaking in the LC filter, which produces
strong resonant currents that can overheat the output MOSFETs and/or other components. The
RC dampens the peaking and prevents problems. Tripath has obtained good results with RZ =
20Ω and CZ = 0.15uF.
Bridging the TK2150
The TK2150 can be bridged by returning the signal from VP1 to the input resistor at INV2. OUT1
will then be a gained version of VP1, and OUT2 will be a gained and inverted version of OAOUT1
(see Figure 8). When the two amplifier outputs are bridged, the apparent load impedance seen
by each output is halved, so the current capability of the output MOSFETs, as well their power
dissipation capability, must be accounted for in the design. In addition, the higher peak currents
caused by driving lower impedance loads will cause additional ringing on the outputs. Thus, the
layout and supply decoupling for low impedance (below 8 ohms) bridged applications must be
extremely good to minimize output ringing and to ensure proper amplifier performance.
TC2001
OAOUT1
22
V5
CI
RI
RF
INV1 23
+
-
+
INPUT1
AGND
BIASCAP
V5
20k
20k
+
-
INV2 28
AGND
OAOUT2
27
Figure 8: Input Stage Setup for Bridging
28
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
The switching outputs, OUT1 and OUT2, are not synchronized, so a common inductor may not
be used with a bridged TK2150. For this same reason, individual zobel networks must be applied
to each output to load each output and lower the Q of each common mode differential LC filter.
Low-frequency Power Supply Pumping
A potentially troublesome phenomenon in single-ended switching amplifiers is power supply
pumping. This phenomenon is caused by current from the output filter inductor flowing into the
power supply output filter capacitors in the opposite direction as a DC load would drain current
from them. Under certain conditions (usually low-frequency input signals), this current can cause
the supply voltage to “pump” (increase in magnitude) and eventually cause over-voltage/under-
voltage shut down. Moreover, since over/under-voltage are not “latched” shutdowns, the effect
would be an amplifier that oscillates between on and off states. If a DC offset on the order of 0.3V
is allowed to develop on the output of the amplifier (see “DC Offset Adjust”), the supplies can be
boosted to the point where the amplifier’s over-voltage protection triggers.
One solution to the pumping issue it to use large power supply capacitors to absorb the pumped
supply current without significant voltage boost. The low-frequency pole used at the input to the
amplifier determines the value of the capacitor required. This works for AC signals only.
A no-cost solution to the pumping problem uses the fact that music has low frequency information
that is correlated in both channels (it is in phase). This information can be used to eliminate
boost by putting the two channels of a TK2150 amplifier out of phase with each other. This works
because each channel is pumping out of phase with the other, and the net effect is a cancellation
of pumping currents in the power supply. The phase of the audio signals needs to be corrected by
connecting one of the speakers in the opposite polarity as the other channel.
Theoretical Efficiency Of A TK2150 Amplifier
The efficiency, η, of an amplifier is:
η = POUT/PIN
The power dissipation of a TK2150 amplifier is primarily determined by the on resistance, RON, of
the output transistors used, and the switching losses of these transistors, PSW. For a TK2150
amplifier, PIN (per channel) is approximated by:
PIN = PDRIVER + PSW + POUT ((RS + RON + RCOIL + RL)/RL)2
where: PDRIVER = Power dissipated in the TP2150 = 1.0W/channel
PSW = 2 x (0.01) x Qg (Qg is the gate charge of MOSFET, in nano-coulombs)
RCOIL = Resistance of the output filter inductor (typically around 50mΩ)
For a 125W RMS per channel, 8Ω load amplifier using FQP13N10 MOSFETs, and an RS of
50mΩ,
PIN = PDRIVER + PSW + POUT ((RS + RON + RCOIL + RL)/RL)2
= .8 + 2 x (0.01) x (12) + 125 x ((0.05 + 0.2414 + 0.05 + 8)/8)2 = 0.8 + 0.24 + 135.9
= 136.94W
In the above calculation the RDS (ON) of 0.065Ω was multiplied by a factor of 1.7 to obtain RON in
order to account for some temperature rise of the MOSFETs. (RDS (ON) typically increases by a
factor of 1.7 for a typical MOSFET as temperature increases from 25ºC to 170ºC.)
So,
η = POUT/PIN = 125/136.94 = 91%
Performance Measurements of a TK2150 Amplifier
Tripath amplifiers operate by modulating the input signal with a high-frequency switching pattern.
This signal is sent through a low-pass filter (external to the TK2150) that demodulates it to
recover an amplified version of the audio input. The frequency of the switching pattern is spread
29
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
spectrum and typically varies between 200kHz and 1.5MHz, which is well above the 20Hz –
22kHz audio band. The pattern itself does not alter or distort the audio input signal but it does
introduce some inaudible noise components.
The measurements of certain performance parameters, particularly those that have anything to
do with noise, like THD+N, are significantly affected by the design of the low-pass filter used on
the output of the TK2150 and also the bandwidth setting of the measurement instrument used.
Unless the filter has a very sharp roll-off just past the audio band or the bandwidth of the
measurement instrument ends there, some of the inaudible noise components introduced by the
Tripath amplifier switching pattern will get integrated into the measurement, degrading it.
Tripath amplifiers do not require large multi-pole filters to achieve excellent performance in
listening tests, usually a more critical factor than performance measurements. Though using a
multi-pole filter may remove high-frequency noise and improve THD+N type measurements
(when they are made with wide-bandwidth measuring equipment), these same filters can
increase distortion due to inductor non-linearity. Multi-pole filters require relatively large
inductors, and inductor non-linearity increases with inductor value.
30
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
TC2001 Package Information
28-pin SOIC
31
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
TP2150 Package Information
64-pin LQFP
32
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
TP2150 Package Information
64-pin LQFP
33
TK2150 – Rev. 1.0/12.02
Tripath Technology, Inc. - Technical Information
PRELIMINARY – This product is still in development. Tripath Technology Inc. reserves the right to
make any changes without further notice to improve reliability, function or design.
This data sheet contains the design specifications for a product in development. Specifications may
change in any manner without notice. Tripath and Digital Power Processing are trademarks of
Tripath Technology Inc. Other trademarks referenced in this document are owned by their respective
companies.
Tripath Technology Inc. reserves the right to make changes without further notice to any products
herein to improve reliability, function or design. Tripath does not assume any liability arising out of the
application or use of any product or circuit described herein; neither does it convey any license under
its patent rights, nor the rights of others.
TRIPATH’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE
SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN CONSENT OF THE
PRESIDENT OF TRIPATH TECHNOLOGY INC.
As used herein:
1.
Life support devices or systems are devices or systems which, (a) are intended for surgical
implant into the body, or (b) support or sustain life, and whose failure to perform, when properly used
in accordance with instructions for use provided in this labeling, can be reasonably expected to result
in significant injury to the user.
2.
A critical component is any component of a life support device or system whose failure to
perform can be reasonably expected to cause the failure of the life support device or system, or to
affect its safety or effectiveness.
Contact Information
TRIPATH TECHNOLOGY, INC
2560 Orchard Parkway, San Jose, CA 95131
408.750.3000 - P
408.750.3001 - F
For more Sales Information, please visit us @ www.tripath.com/cont_s.htm
For more Technical Information, please visit us @ www.tripath.com/data.htm
34
TK2150 – Rev. 1.0/12.02
相关型号：
FQP17N08LJ69Z
Power Field-Effect Transistor, 16.5A I(D), 80V, 0.115ohm, 1-Element, N-Channel, Silicon, Metal-oxide Semiconductor FET, TO-220, 3 PIN
FAIRCHILD
FQP17N40J69Z
Power Field-Effect Transistor, 16A I(D), 400V, 0.27ohm, 1-Element, N-Channel, Silicon, Metal-oxide Semiconductor FET, TO-220, 3 PIN
FAIRCHILD
©2020 ICPDF网 联系我们和版权申明