5962-8980501IA

更新时间:2024-10-08 01:49:41
品牌:ADI
描述:Integrated Circuit True RMS-to-DC Converter

5962-8980501IA 概述

Integrated Circuit True RMS-to-DC Converter 集成电路真RMS至DC转换器 AD转换器

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Integrated Circuit  
True RMS-to-DC Converter  
a
AD536A  
FEATURES  
PIN CONFIGURATIONS AND  
True RMS-to-DC Conversion  
Laser-Trimmed to High Accuracy  
0.2% Max Error (AD536AK)  
0.5% Max Error (AD536AJ)  
Wide Response Capability:  
Computes RMS of AC and DC Signals  
450 kHz Bandwidth: V rms > 100 mV  
2 MHz Bandwidth: V rms > 1 V  
Signal Crest Factor of 7 for 1% Error  
dB Output with 60 dB Range  
Low Power: 1.2 mA Quiescent Current  
Single or Dual Supply Operation  
Monolithic Integrated Circuit  
–55؇C to +125؇C Operation (AD536AS)  
FUNCTIONAL BLOCK DIAGRAMS  
TO-100 (H-10A)  
Package  
TO-116 (D-14) and  
Q-14 Package  
I
OUT  
ABSOLUTE  
14  
13  
12  
11  
10  
9
V
1
2
3
4
5
6
7
+V  
S
IN  
VALUE  
R
BUF IN  
L
25k  
AD536A  
NC  
NC  
25k⍀  
SQUARER  
DIVIDER  
AD536A  
V  
NC  
BUF  
S
COM  
OUT  
CURRENT  
MIRROR  
BUF  
C
NC  
AV  
CURRENT  
MIRROR  
COM  
dB  
SQUARER  
DIVIDER  
+V  
S
BUF  
dB  
R
L
OUT  
25k  
BUF  
IN  
BUF  
8
ABSOLUTE  
VALUE  
I
OUT  
25k⍀  
C
V
AV  
IN  
NC = NO CONNECT  
V  
S
PRODUCT DESCRIPTION  
LCC (E-20A) Package  
The AD536A is a complete monolithic integrated circuit which  
performs true rms-to-dc conversion. It offers performance which  
is comparable or superior to that of hybrid or modular units  
costing much more. The AD536A directly computes the true  
rms value of any complex input waveform containing ac and dc  
components. It has a crest factor compensation scheme which  
allows measurements with 1% error at crest factors up to 7. The  
wide bandwidth of the device extends the measurement capabi-  
lity to 300 kHz with 3 dB error for signal levels above 100 mV.  
NC  
3
NC  
1
NC  
V
2
+V  
IN  
S
20 19  
ABSOLUTE  
VALUE  
4
5
6
7
8
18  
NC  
–V  
S
AD536A  
NC  
17 NC  
SQUARER  
DIVIDER  
16  
NC  
C
AV  
25k⍀  
CURRENT  
MIRROR  
15 NC  
NC  
14  
dB  
COM  
25k⍀  
BUF  
9
10 11 12  
13  
R
An important feature of the AD536A not previously available in  
rms converters is an auxiliary dB output. The logarithm of the  
rms output signal is brought out to a separate pin to allow the  
dB conversion, with a useful dynamic range of 60 dB. Using an  
externally supplied reference current, the 0 dB level can be con-  
veniently set by the user to correspond to any input level from  
0.1 to 2 volts rms.  
BUF BUF  
OUT IN  
NC = NO CONNECT  
I
NC  
OUT  
L
PRODUCT HIGHLIGHTS  
1. The AD536A computes the true root-mean-square level of a  
complex ac (or ac plus dc) input signal and gives an equiva-  
lent dc output level. The true rms value of a waveform is a  
more useful quantity than the average rectified value since it  
relates directly to the power of the signal. The rms value of a  
statistical signal also relates to its standard deviation.  
The AD536A is laser trimmed at the wafer level for input and  
output offset, positive and negative waveform symmetry (dc re-  
versal error), and full-scale accuracy at 7 V rms. As a result, no  
external trims are required to achieve the rated unit accuracy.  
2. The crest factor of a waveform is the ratio of the peak signal  
swing to the rms value. The crest factor compensation  
scheme of the AD536A allows measurement of highly com-  
plex signals with wide dynamic range.  
There is full protection for both inputs and outputs. The input  
circuitry can take overload voltages well beyond the supply lev-  
els. Loss of supply voltage with inputs connected will not cause  
unit failure. The output is short-circuit protected.  
3. The only external component required to perform measure-  
ments to the fully specified accuracy is the capacitor which  
sets the averaging period. The value of this capacitor determines  
the low frequency ac accuracy, ripple level and settling time.  
The AD536A is available in two accuracy grades (J, K) for com-  
mercial temperature range (0°C to +70°C) applications, and one  
grade (S) rated for the –55°C to +125°C extended range. The  
AD536AK offers a maximum total error of 2 mV 0.2% of  
reading, and the AD536AJ and AD536AS have maximum errors  
of 5 mV 0.5% of reading. All three versions are available in  
either a hermetically sealed 14-lead DIP or 10-pin TO-100  
metal can. The AD536AS is also available in a 20-leadless her-  
metically sealed ceramic chip carrier.  
4. The AD536A will operate equally well from split supplies or  
a single supply with total supply levels from 5 to 36 volts.  
The one milliampere quiescent supply current makes the  
device well-suited for a wide variety of remote controllers and  
battery powered instruments.  
5. The AD536A directly replaces the AD536 and provides im-  
proved bandwidth and temperature drift specifications.  
REV. B  
Information furnished by Analog Devices is believed to be accurate and  
reliable. However, no responsibility is assumed by Analog Devices for its  
use, nor for any infringements of patents or other rights of third parties  
which may result from its use. No license is granted by implication or  
otherwise under any patent or patent rights of Analog Devices.  
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.  
Tel: 781/329-4700  
Fax: 781/326-8703  
World Wide Web Site: http://www.analog.com  
© Analog Devices, Inc., 1999  
(@ +25  
؇
C, and ؎15 V dc unless otherwise noted)  
AD536A–SPECIFICATIONS  
Model  
AD536AJ  
Typ  
AD536AK  
Typ  
AD536AS  
Typ  
Min  
Max  
Min  
Max  
Min  
VOUT  
Max  
Units  
TRANSFER FUNCTION  
CONVERSION ACCURACY  
Total Error, Internal Trim1 (Figure 1)  
vs. Temperature, TMIN to +70°C  
+70°C to +125°C  
2
)
2
2
VOUT  
=
avg.(V  
=
avg.(V )  
IN  
VOUT  
=
avg.(V )  
IN  
IN  
؎5 ؎0.5  
0.1 0.01  
؎2 ؎0.2  
0.05 0.005  
؎5 ؎0.5  
؎0.1 ؎0.005  
؎0.3 ؎0.005  
mV % of Reading  
mV % of Reading/°C  
mV % of Reading/°C  
mV % of Reading/V  
% of Reading  
vs. Supply Voltage  
0.1 0.01  
0.2  
0.1 0.01  
0.1  
0.1 0.01  
0.2  
dc Reversal Error  
Total Error, External Trim1 (Figure 2)  
3
0.3  
2
0.1  
3
0.3  
mV % of Reading  
ERROR VS. CREST FACTOR2  
Crest Factor 1 to 2  
Specified Accuracy  
Specified Accuracy  
Specified Accuracy  
Crest Factor = 3  
Crest Factor = 7  
0.1  
1.0  
0.1  
1.0  
0.1  
1.0  
% of Reading  
% of Reading  
FREQUENCY RESPONSE3  
Bandwidth for 1% Additional Error (0.09 dB)  
VIN = 10 mV  
5
45  
120  
5
45  
120  
5
45  
120  
kHz  
kHz  
kHz  
VIN = 100 mV  
VIN = 1 V  
3 dB Bandwidth  
VIN = 10 mV  
VIN = 100 mV  
VIN = 1 V  
90  
450  
2.3  
90  
450  
2.3  
90  
450  
2.3  
kHz  
kHz  
MHz  
AVERAGlNG TlME CONSTANT (Figure 5)  
25  
25  
25  
ms/µF CAV  
INPUT CHARACTERISTICS  
Signal Range, 15 V Supplies  
Continuous rms Level  
0 to 7  
0 to 2  
0 to 7  
0 to 2  
0 to 7  
0 to 2  
V rms  
V peak  
V rms  
V peak  
Peak Transient Input  
20  
7
20  
7
20  
7
Continuous rms Level, 5 V Supplies  
Peak Transient Input, 5 V Supplies  
Maximum Continuous Nondestructive  
Input Level (All Supply Voltages)  
Input Resistance  
25  
20  
2
25  
20  
1
25  
20  
2
V peak  
kΩ  
mV  
13.33  
16.67  
0.8  
13.33  
16.67  
0.5  
13.33  
16.67  
0.8  
Input Offset Voltage  
OUTPUT CHARACTERISTICS  
Offset Voltage, VIN = COM (Figure 1)  
vs. Temperature  
vs. Supply Voltage  
Voltage Swing, 15 V Supplies  
5 V Supply  
1
0.1  
0.1  
+12.5  
2
0.5  
0.1  
0.1  
1
؎2  
؎0.2  
mV  
mV/°C  
mV/V  
V
0.2  
+12.5  
0 to +11  
0 to +2  
0 to +11  
0 to +2  
+12.5  
0 to +11  
0 to +2  
V
dB OUTPUT (Figure 13)  
Error, VlN 7 mV to 7 V rms, 0 dB = 1 V rms  
Scale Factor  
0.4  
3  
؎0.6  
0.2  
3  
؎0.3  
0.5  
3  
؎0.6  
dB  
mV/dB  
Scale Factor TC (Uncompensated, see Fig-  
ure 1 for Temperature Compensation)  
0.033  
+0.33  
20  
0.033  
+0.33  
20  
0.033  
+0.33  
20  
dB/°C  
% of Reading/°C  
I
REF for 0 dB = 1 V rms  
5
1
80  
100  
5
1
80  
100  
5
1
80  
100  
µA  
µA  
IREF Range  
IOUT TERMINAL  
I
OUT Scale Factor  
40  
10  
25  
VS to (+VS  
2.5 V)  
40  
10  
25  
VS to (+VS  
2.5 V)  
40  
10  
25  
VS to (+VS  
2.5 V)  
µA/V rms  
IOUT Scale Factor Tolerance  
Output Resistance  
Voltage Compliance  
20  
30  
20  
30  
20  
30  
%
20  
20  
20  
kΩ  
V
V
BUFFER AMPLIFIER  
Input and Output Voltage Range  
VS to (+VS  
2.5 V)  
VS to (+VS  
2.5 V)  
VS to (+VS  
2.5 V)  
Input Offset Voltage, RS = 25 k  
Input Bias Current  
Input Resistance  
0.5  
؎4  
60  
0.5  
؎4  
60  
0.5  
20  
؎4  
60  
mV  
nA  
20  
20  
108  
108  
108  
Output Current  
(+5 mA,  
(+5 mA,  
(+5 mA,  
130 µA)  
130 µA)  
130 µA)  
Short Circuit Current  
Output Resistance  
Small Signal Bandwidth  
Slew Rate4  
20  
20  
20  
mA  
MHz  
V/µs  
0.5  
0.5  
0.5  
1
5
1
5
1
5
POWER SUPPLY  
Voltage Rated Performance  
Dual Supply  
Single Supply  
15  
15  
15  
V
V
V
3.0  
+5  
18  
+36  
3.0  
+5  
18  
+36  
3.0  
+5  
18  
+36  
Quiescent Current  
Total VS, 5 V to 36 V, TMIN to TMAX  
1.2  
2
1.2  
2
1.2  
2
mA  
TEMPERATURE RANGE  
Rated Performance  
Storage  
0
55  
+70  
+150  
0
55  
+70  
+150  
55  
55  
+125  
+150  
°C  
°C  
NUMBER OF TRANSISTORS  
NOTES  
65  
65  
65  
1Accuracy is specified for 0 V to 7 V rms, dc or 1 kHz sine wave input with the AD536A connected as in the figure referenced.  
2Error vs. crest factor is specified as an additional error for 1 V rms rectangular pulse input, pulsewidth = 200µs.  
3Input voltages are expressed in volts rms, and error is percent of reading.  
4With 2k external pull-down resistor.  
Specifications subject to change without notice.  
Specifications shown in boldface are tested on all production units at final electrical test. Results from those tests are used to calculate outgoing quality levels. All min and max specifications are guaranteed,  
although only those shown in boldface are tested on all production units.  
–2–  
REV. B  
AD536A  
ABSOLUTE MAXIMUM RATINGS1  
STANDARD CONNECTION  
Supply Voltage  
The AD536A is simple to connect for the majority of high accu-  
racy rms measurements, requiring only an external capacitor to  
set the averaging time constant. The standard connection is  
shown in Figure 1. In this configuration, the AD536A will mea-  
sure the rms of the ac and dc level present at the input, but will  
show an error for low frequency inputs as a function of the filter  
capacitor, CAV, as shown in Figure 5. Thus, if a 4 µF capacitor  
is used, the additional average error at 10 Hz will be 0.1%, at  
3 Hz it will be 1%. The accuracy at higher frequencies will be  
according to specification. If it is desired to reject the dc input, a  
capacitor is added in series with the input, as shown in Figure 3,  
the capacitor must be nonpolar. If the AD536A is driven with  
power supplies with a considerable amount of high frequency  
ripple, it is advisable to bypass both supplies to ground with  
0.1 µF ceramic discs as near the device as possible.  
Dual Supply . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 V  
Single Supply . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +36 V  
Internal Power Dissipation2 . . . . . . . . . . . . . . . . . . . . 500 mW  
Maximum Input Voltage . . . . . . . . . . . . . . . . . . . . 25 V Peak  
Buffer Maximum Input Voltage . . . . . . . . . . . . . . . . . . . . . VS  
Maximum Input Voltage . . . . . . . . . . . . . . . . . . . . 25 V Peak  
Storage Temperature Range . . . . . . . . . . . . 55°C to +150°C  
Operating Temperature Range  
AD536AJ/K . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to +70°C  
AD536AS . . . . . . . . . . . . . . . . . . . . . . . . 55°C to +125°C  
Lead Temperature Range  
(Soldering 60 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . +300°C  
ESD Rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1000 V  
NOTES  
1Stresses above those listed under Absolute Maximum Ratings may cause perma-  
nent damage to the device. This is a stress rating only; functional operation of the  
device at these or any other conditions above those indicated in the operational  
section of this specification is not implied. Exposure to absolute maximum rating  
conditions for extended periods may affect device reliability.  
210-Pin Header: θJA = 150°C/W; 20-Leadless LCC: θJA = 95°C/W; 14-Lead Size  
Brazed Ceramic DIP: θJA = 95°C/W.  
C
AV  
ABSOLUTE  
VALUE  
+V  
V
14  
13  
12  
11  
10  
9
1
2
3
4
5
6
7
S
IN  
AD536A  
SQUARER  
DIVIDER  
V  
S
CHIP DIMENSIONS AND PAD LAYOUT  
Dimensions shown in inches and (mm).  
CURRENT  
MIRROR  
V
OUT  
25k  
BUF  
25k⍀  
8
25k⍀  
25k⍀  
AD536A  
V
CURRENT  
MIRROR  
OUT  
BUF  
SQUARER  
DIVIDER  
+V  
S
ABSOLUTE  
VALUE  
V
IN  
C
AV  
V  
S
ORDERING GUIDE  
V
C
+V  
S
IN  
AV  
Temperature  
Range  
Package  
Description  
Package  
Option  
Model  
3
2
1
20 19  
AD536AJD  
AD536AKD  
AD536AJH  
AD536AKH  
AD536AJQ  
AD536AKQ  
AD536ASD  
AD536ASD/883B  
AD536ASE/883B  
AD536ASH  
AD536ASH/883B  
AD536AJCHIPS  
AD536AKH/+  
AD536ASCHIPS  
5962-89805012A  
5962-8980501CA  
5962-8980501IA  
0°C to +70°C  
Side Brazed Ceramic DIP D-14  
Side Brazed Ceramic DIP D-14  
ABSOLUTE  
VALUE  
4
5
6
7
8
18  
17  
16  
15  
14  
V  
0°C to +70°C  
S
AD536A  
0°C to +70°C  
Header  
Header  
Cerdip  
Cerdip  
H-10A  
H-10A  
Q-14  
SQUARER  
DIVIDER  
0°C to +70°C  
0°C to +70°C  
25k⍀  
0°C to +70°C  
Q-14  
CURRENT  
MIRROR  
55°C to +125°C  
55°C to +125°C  
55°C to +125°C  
55°C to +125°C  
55°C to +125°C  
0°C to +70°C  
Side Brazed Ceramic DIP D-14  
Side Brazed Ceramic DIP D-14  
25k⍀  
dB  
BUF  
9
LCC  
E-20A  
H-10A  
H-10A  
10 11 12  
13  
Header  
Header  
Die  
V
OUT  
Figure 1. Standard RMS Connection  
0°C to +70°C  
Header  
Die  
LCC  
H-10A  
E-20A  
55°C to +125°C  
55°C to +125°C  
55°C to +125°C  
55°C to +125°C  
Side Brazed Ceramic DIP D-14  
Header H-10A  
REV. B  
–3–  
AD536A  
The input and output signal ranges are a function of the supply  
voltages; these ranges are shown in Figure 14. The AD536A can  
also be used in an unbuffered voltage output mode by discon-  
necting the input to the buffer. The output then appears unbuf-  
fered across the 25 kresistor. The buffer amplifier can then be  
used for other purposes. Further the AD536A can be used in a  
current output mode by disconnecting the 25 kresistor from  
ground. The output current is available at Pin 8 (Pin 10 on the  
Hpackage) with a nominal scale of 40 µA per volt rms input  
positive out.  
by using a resistive divider between +VS and ground. The values  
of the resistors can be increased in the interest of lowered power  
consumption, since only 5 mA of current flows into Pin 10  
(Pin 2 on the Hpackage). AC input coupling requires only  
capacitor C2 as shown; a dc return is not necessary as it is  
provided internally. C2 is selected for the proper low frequency  
break point with the input resistance of 16.7 k; for a cutoff at  
10 Hz, C2 should be 1 µF. The signal ranges in this connection  
are slightly more restricted than in the dual supply connection.  
The input and output signal ranges are shown in Figure 14. The  
load resistor, RL, is necessary to provide output sink current.  
OPTIONAL EXTERNAL TRIMS FOR HIGH ACCURACY  
If it is desired to improve the accuracy of the AD536A, the  
external trims shown in Figure 2 can be added. R4 is used to  
trim the offset. Note that the offset trim circuit adds 365 in  
series with the internal 25 kresistor. This will cause a 1.5%  
increase in scale factor, which is trimmed out by using R1 as  
shown. Range of scale factor adjustment is 1.5%.  
C2  
The trimming procedure is as follows:  
1. Ground the input signal, VIN, and adjust R4 to give zero  
volts output from Pin 6. Alternatively, R4 can be adjusted to  
give the correct output with the lowest expected value of VIN.  
2. Connect the desired full scale input level to VIN, either dc or  
a calibrated ac signal (1 kHz is the optimum frequency);  
then trim R1, to give the correct output from Pin 6, i.e.,  
1000 V dc input should give 1.000 V dc output. Of course, a  
1.000 V peak-to-peak sine wave should give a 0.707 V dc  
output. The remaining errors, as given in the specifications  
are due to the nonlinearity.  
Figure 3. Single Supply Connection  
The major advantage of external trimming is to optimize device  
performance for a reduced signal range; the AD536A is inter-  
nally trimmed for a 7 V rms full-scale range.  
CHOOSING THE AVERAGING TIME CONSTANT  
The AD536A will compute the rms of both ac and dc signals.  
If the input is a slowly-varying dc signal, the output of the  
AD536A will track the input exactly. At higher frequencies, the  
average output of the AD536A will approach the rms value of  
the input signal. The actual output of the AD536A will differ  
from the ideal output by a dc (or average) error and some  
amount of ripple, as demonstrated in Figure 4.  
Figure 4. Typical Output Waveform for Sinusoidal Input  
Figure 2. Optional External Gain and Output Offset Trims  
The dc error is dependent on the input signal frequency and the  
value of CAV. Figure 5 can be used to determine the minimum  
value of CAV which will yield a given percent dc error above a  
given frequency using the standard rms connection.  
SINGLE SUPPLY CONNECTION  
The applications in Figures l and 2 require the use of approxi-  
mately symmetrical dual supplies. The AD536A can also be  
used with only a single positive supply down to +5 volts, as  
shown in Figure 3. The major limitation of this connection is  
that only ac signals can be measured since the differential input  
stage must be biased off ground for proper operation. This  
biasing is done at Pin 10; thus it is critical that no extraneous  
signals be coupled into this point. Biasing can be accomplished  
The ac component of the output signal is the ripple. There are  
two ways to reduce the ripple. The first method involves using a  
large value of CAV. Since the ripple is inversely proportional to  
C
AV, a tenfold increase in this capacitance will affect a tenfold  
reduction in ripple. When measuring waveforms with high crest  
4–  
REV. B  
AD536A  
The two-pole post-filter uses an active filter stage to provide  
even greater ripple reduction without substantially increasing  
the settling times over a circuit with a one-pole filter. The values  
of CAV, C2, and C3 can then be reduced to allow extremely fast  
settling times for a constant amount of ripple. Caution should  
be exercised in choosing the value of CAV, since the dc error is  
dependent upon this value and is independent of the post filter.  
factors, (such as low duty cycle pulse trains), the averaging time  
constant should be at least ten times the signal period. For  
example, a 100 Hz pulse rate requires a 100 ms time constant,  
which corresponds to a 4 µF capacitor (time constant = 25 ms  
per µF).  
The primary disadvantage in using a large CAV to remove ripple  
is that the settling time for a step change in input level is in-  
creased proportionately. Figure 5 shows that the relationship  
between CAV and 1% settling time is 115 milliseconds for each  
microfarad of CAV. The settling time is twice as great for de-  
creasing signals as for increasing signals (the values in Figure 5  
are for decreasing signals). Settling time also increases for low  
signal levels, as shown in Figure 6.  
For a more detailed explanation of these topics refer to the  
RMS to DC Conversion Application Guide 2nd Edition, available  
from Analog Devices.  
C3  
C2  
C3  
Figure 7. 2-Pole PostFilter  
Figure 5. Error/Settling Time Graph for Use with the Stan-  
dard rms Connection in Figure 1  
Figure 6. Settling Time vs. Input Level  
Figure 8. Performance Features of Various Filter Types  
A better method for reducing output ripple is the use of a  
post-filter.Figure 7 shows a suggested circuit. If a single-pole  
filter is used (C3 removed, RX shorted), and C2 is approximately  
twice the value of CAV, the ripple is reduced as shown in Figure  
8 and settling time is increased. For example, with CAV = 1 µF  
and C2 = 2.2 µF, the ripple for a 60 Hz input is reduced from  
10% of reading to approximately 0.3% of reading. The settling  
time, however, is increased by approximately a factor of 3. The  
values of CAV and C2, can, therefore, be reduced to permit faster  
settling times while still providing substantial ripple reduction.  
AD536A PRINCIPLE OF OPERATION  
The AD536A embodies an implicit solution of the rms equation  
that overcomes the dynamic range as well as other limitations  
inherent in a straightforward computation of rms. The actual  
computation performed by the AD536A follows the equation:  
2
VIN  
V rms = Avg.  
V rms   
REV. B  
5–  
AD536A  
Figure 9 is a simplified schematic of the AD536A; it is subdi-  
vided into four major sections: absolute value circuit (active  
rectifier), squarer/divider, current mirror, and buffer amplifier.  
The input voltage, VIN, which can be ac or dc, is converted to a  
unipolar current I1, by the active rectifier A1, A2. I1 drives one  
input of the squarer/divider, which has the transfer function:  
The current mirror also produces the output current, IOUT,  
which equals 2I4. IOUT can be used directly or converted to a  
voltage with R2 and buffered by A4 to provide a low impedance  
voltage output. The transfer function of the AD536A thus  
results:  
VOUT = 2R2 I rms = VIN rms  
I4 = I12/I3  
The dB output is derived from the emitter of Q3, since the  
voltage at this point is proportional to log VIN. Emitter fol-  
lower, Q5, buffers and level shifts this voltage, so that the dB  
output voltage is zero when the externally supplied emitter  
current (IREF) to Q5 approximates I3.  
The output current, I4, of the squarer/divider drives the current  
mirror through a low-pass filter formed by R1 and the externally  
connected capacitor, CAV. If the R1, CAV time constant is much  
greater than the longest period of the input signal, then I4 is  
effectively averaged. The current mirror returns a current I3,  
which equals Avg. [I4], back to the squarer/divider to complete  
the implicit rms computation. Thus:  
CONNECTIONS FOR dB OPERATION  
A powerful feature added to the AD536A is the logarithmic or  
decibel output. The internal circuit computing dB works accu-  
rately over a 60 dB range. The connections for dB measure-  
ments are shown in Figure 10. The user selects the 0 dB level by  
adjusting R1, for the proper 0 dB reference current (which is set  
to exactly cancel the log output current from the squarer-divider  
at the desired 0 dB point). The external op amp is used to pro-  
vide a more convenient scale and to allow compensation of the  
+0.33%/°C scale factor drift of the dB output pin. The special  
T.C. resistor, R2, is available from Tel Labs in Londonderry,  
N.H. (model Q-81) or from Precision Resistor Inc., Hillside,  
N.J. (model PT146). The averaged temperature coefficients of  
resistors R2 and R3 develop the +3300 ppm needed to reverse  
compensate the dB output. The linear rms output is available at  
Pin 8 on DIP or Pin 10 on header device with an output imped-  
ance of 25 k; thus some applications may require an additional  
buffer amplifier if this output is desired.  
2
I4 = Avg. I1 /I4 = I1 rms  
[
]
dB Calibration:  
1. Set VIN = 1.00 V dc or 1.00 V rms  
2. Adjust R1 for dB out = 0.00 V  
3. Set VIN = +0.1 V dc or 0.10 V rms  
4. Adjust R5 for dB out = 2.00 V  
Any other desired 0 dB reference level can be used by setting  
Figure 9. Simplified Schematic  
VIN and adjusting R1, accordingly. Note that adjusting R5 for  
the proper gain automatically gives the correct temperature  
compensation.  
Figure 10. dB Connection  
6–  
REV. B  
AD536A  
FREQUENCY RESPONSE  
The AD536A utilizes a logarithmic circuit in performing the  
implicit rms computation. As with any log circuit, bandwidth is  
proportional to signal level. The solid lines in the graph below  
represent the frequency response of the AD536A at input levels  
from 10 millivolts to 7 volts rms. The dashed lines indicate the  
upper frequency limits for 1%, 10%, and 3 dB of reading addi-  
tional error. For example, note that a 1 volt rms signal will pro-  
duce less than 1% of reading additional error up to 120 kHz. A  
10 millivolt signal can be measured with 1% of reading addi-  
tional error (100 µV) up to only 5 kHz.  
Figure 12. Error vs. Crest Factor  
Figure 11. High Frequency Response  
AC MEASUREMENT ACCURACY AND CREST FACTOR  
Crest factor is often overlooked in determining the accuracy of  
an ac measurement. Crest factor is defined as the ratio of the  
peak signal amplitude to the rms value of the signal (CF = VP/  
V rms). Most common waveforms, such as sine and triangle  
waves, have relatively low crest factors (<2). Waveforms which  
resemble low duty cycle pulse trains, such as those occurring in  
switching power supplies and SCR circuits, have high crest  
factors. For example, a rectangular pulse train with a 1% duty  
Figure 13. AD536A Error vs. Pulsewidth Rectangular  
Pulse  
cycle has a crest factor of 10 (CF = 1  
η
).  
Figure 12 is a curve of reading error for the AD536A for a 1 volt  
rms input signal with crest factors from 1 to 11. A rectangular  
pulse train (pulsewidth 100 µs) was used for this test since it is  
the worst-case waveform for rms measurement (all the energy is  
contained in the peaks). The duty cycle and peak amplitude  
were varied to produce crest factors from 1 to 11 while main-  
taining a constant 1 volt rms input amplitude.  
Figure 14. AD536A Input and Output Voltage Ranges  
vs. Supply  
REV. B  
7–  
AD536A  
OUTLINE DIMENSIONS  
Dimensions shown in inches and (mm).  
D-14 Package  
TO-116  
H-10A Package  
TO-100  
E-20A Package  
LCC  
8–  
REV. B  

5962-8980501IA CAD模型

  • 引脚图

  • 封装焊盘图

  • 5962-8980501IA 替代型号

    型号 制造商 描述 替代类型 文档
    AD536ASH ADI Integrated Circuit True RMS-to-DC Converter 完全替代
    AD536ASH/883B ADI Integrated Circuit True RMS-to-DC Converter 完全替代
    AD636JHZ ADI Low Level, True RMS-to-DC Converter 类似代替

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