AD7711ARZ-REEL_技术文档

LC²MOS Signal Conditioning ADC

with RTD Excitation Currents

a

AD7711*

FUNCTIONAL BLOCK DIAGRAM

FEATURES

Charge Balancing ADC

24 Bits No Missing Codes

REF

IN (+)

REF

IN (–)

AV

DV

DD

V

DD

REF OUT

BIAS

؎0.0015% Nonlinearity

AV

DD

Two-Channel Programmable Gain Front End

Gains from 1 to 128

One Differential Input

2.5V REFERENCE

4.5␮A

CHARGE-BALANCING A/D

CONVERTER

One Single-Ended Input

AIN1(+)

AIN1(–)

AUTO-ZEROED

DIGITAL

SYNC

Low-Pass Filter with Programmable Filter Cutoffs

Ability to Read/Write Calibration Coefficients

RTD Excitation Current Sources

Bidirectional Microcontroller Serial Interface

Internal/External Reference Option

Single or Dual Supply Operation

Low Power (25 mW typ) with Power-Down Mode

(7 mW typ)

M

U

X

PGA

A = 1 – 128

FILTER

MODULATOR

AIN2

MCLK

IN

CLOCK

GENERATION

200␮A

AV

DD

MCLK

OUT

RTD1

RTD2

SERIAL INTERFACE

200␮A

CONTROL

REGISTER

OUTPUT

REGISTER

APPLICATIONS

RTD Transducers

AD7711

Process Control

Smart Transmitters

Portable Industrial Instruments

AGND DGND

MODE SDATA SCLK

A0

DRDY

V

RFS TFS

SS

and RTD current control can be configured in software using

the bidirectional serial port. The AD7711 contains self-

calibration, system calibration and background calibration

options and also allows the user to read and write the on-chip

calibration registers.

GENERAL DESCRIPTION

The AD7711 is a complete analog front end for low frequency

measurement applications. The device accepts low level signals

directly from a transducer and outputs a serial digital word. It

employs a sigma-delta conversion technique to realize up to

24 bits of no missing codes performance. The input signal is

applied to a proprietary programmable gain front end based

around an analog modulator. The modulator output is pro-

cessed by an on-chip digital filter. The first notch of this digital

filter can be programmed via the on-chip control register allow-

ing adjustment of the filter cutoff and settling time.

CMOS construction ensures low power dissipation, and a soft-

ware programmable power-down mode reduces the standby

power consumption to only 7 mW typical. The part is available

in a 24-lead, 0.3 inch wide, plastic and hermetic dual-in-line

package (DIP) as well as a 24-lead small outline (SOIC)

package.

The part features one differential analog input and one single

ended analog input as well as a differential reference input.

Normally, one of the input channels will be used as the main

channel with the second channel used as an auxiliary input to

periodically measure a second voltage. It can be operated from a

single supply (by tying the V_SSpin to AGND) provided that the

input signals on the analog inputs are more positive than

–30 mV. By taking the V_SSpin negative, the part can convert

signals down to –V_REFon its inputs. The part provides two

current sources that can be used to provide excitation in three-

wire and four-wire RTD configurations. The AD7711 thus

performs all signal conditioning and conversion for a single or

dual channel system.

PRODUCT HIGHLIGHTS

1. The programmable gain front end allows the AD7711 to

accept input signals directly from an RTD transducer,

removing a considerable amount of signal conditioning.

On-chip current sources provide excitation for three-wire and

four-wire RTD configurations.

2. No Missing Codes ensure true, usable, 23-bit dynamic range

coupled with excellent ±0.0015% accuracy. The effects of

temperature drift are eliminated by on-chip self-calibration,

which removes zero-scale and full-scale errors.

3. The AD7711 is ideal for microcontroller or DSP processor

applications with an on-chip control register which allows

control over filter cutoff, input gain, channel selection, signal

polarity, RTD current control and calibration modes.

The AD7711 is ideal for use in smart, microcontroller based

systems. Gain settings, signal polarity, input channel selection

4. The AD7711 allows the user to read and to write the on-chip

calibration registers. This means that the microcontroller has

much greater control over the calibration procedure.

*Protected by U.S. Patent No. 5,134,401.

REV. F

Information furnished by Analog Devices is believed to be accurate and

reliable. However, no responsibility is assumed by Analog Devices for its

use, nor for any infringements of patents or other rights of third parties

which may result from its use. No license is granted by implication or

otherwise under any patent or patent rights of Analog Devices.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.

Tel: 781/329-4700

Fax: 781/326-8703

World Wide Web Site: http://www.analog.com

(AV = +5 V ؎ 5%; DV = +5 V ؎ 5%; V = 0 V or –5 V ؎ 5%; REF IN(+) =

+2.5 V; REF IN(–) = AGND; MCLK IN = 10 MHz unless otherwise stated. All specifications T_MINto T_MAXunless otherwise noted.)

AD7711–SPECIFICATIONS

DD

SS

Parameter

A, S Versions¹

Units

Conditions/Comments

STATIC PERFORMANCE

No Missing Codes

24

22

18

15

Bits min

Guaranteed by Design. For Filter Notches ≤ 60 Hz

For Filter Notch = 100 Hz

For Filter Notch = 250 Hz

For Filter Notch = 500 Hz

For Filter Notch = 1 kHz

12

Output Noise

See Tables I & II

Depends on Filter Cutoffs and Selected Gain

Filter Notches ≤ 60 Hz

Typically ±0.0003%

Excluding Reference

Excluding Reference. For Gains of 1, 2

Excluding Reference. For Gains of 4, 8, 16, 32, 64, 128

Integral Nonlinearity @ +25°C

T_MINto T_MAX

±0.0015

±0.003

See Note 4

1

0.3

See Note 4

0.5

0.25

See Note 4

0.5

0.25

2

% FSR max

Positive Full-Scale Error^{2, 3}

Full-Scale Drift⁵

µV/°C typ

Unipolar Offset Error²

Unipolar Offset Drift⁵

µV/°C typ

For Gains of 1, 2

For Gains of 4, 8, 16, 32, 64, 128

Bipolar Zero Error²

Bipolar Zero Drift⁵

µV/°C typ

ppm/°C typ

% FSR max

µV/°C typ

For Gains of 1, 2

For Gains of 4, 8, 16, 32, 64, 128

Gain Drift

Bipolar Negative Full-Scale Error²@ +25°C

T_MINto T_MAX

±0.003

±0.006

1

Excluding Reference

Typically ±0.0006%

Excluding Reference. For Gains of 1, 2

Excluding Reference. For Gains of 4, 8, 16, 32, 64, 128

Bipolar Negative Full-Scale Drift⁵

0.3

ANALOG INPUTS/REFERENCE INPUTS

Normal-Mode 50 Hz Rejection⁶

Normal-Mode 60 Hz Rejection⁶

DC Input Leakage Current @ +25°C⁶

T_MINto T_MAX

100

10

1

20

dB min

pA max

nA max

pF max

For Filter Notches of 10, 25, 50 Hz, ±0.02 × f_NOTCH

For Filter Notches of 10, 30, 60 Hz, ±0.02 × f_NOTCH

Sampling Capacitance⁶

AIN1/REF IN

Common-Mode Rejection (CMR)

Common-Mode 50 Hz Rejection⁶

Common-Mode 60 Hz Rejection⁶

Common-Mode Voltage Range⁷

Analog Inputs⁸

100

150

V_SSto AV_DD

dB min

V min to V max

At DC

For Filter Notches of 10, 25, 50 Hz, ±0.02 × f_NOTCH

For Filter Notches of 10, 30, 60 Hz, ±0.02 × f_NOTCH

Input Voltage Range⁹

For Normal Operation. Depends on Gain Selected

Unipolar Input Range (B/U Bit of Control Register = 1)

Bipolar Input Range (B/U Bit of Control Register = 0)

10

0 to +V_REF

max

±V_REF

See Table III

2.5

Input Sampling Rate, f_S

AIN2 Offset Error

mV max

Removed by System Calibrations but not by Self-Calibration

AIN2 Offset Drift

1.5

µV/°C typ

Reference Inputs

REF IN(+) – REF IN(–) Voltage¹¹

+2.5 to +5

f_{CLK IN}/256

V min to V max

For Specified Performance. Part Is Functional with

Lower V_REFVoltages

Input Sampling Rate, f_S

REFERENCE OUTPUT

Output Voltage

Initial Tolerance @ +25°C

Drift

Output Noise

Line Regulation (AV_DD

Load Regulation

2.5

±1

20

30

1

1.5

1

V nom

% max

ppm/°C typ

µV typ

mV/V max

mV/mA max

mA max

pk-pk Noise. 0.1 Hz to 10 Hz Bandwidth

Maximum Load Current 1 mA

)

External Current

NOTES

¹Temperature range is as follows: A Version = –40°C to +85°C; S Version = –55°C to +125°C. See also Note 16.

²Applies after calibration at the temperature of interest.

³Positive full-scale error applies to both unipolar and bipolar input ranges.

⁴These errors will be of the order of the output noise of the part as shown in Table I after system calibration. These errors will be 20 µV typical after self-calibration or

background calibration.

⁵Recalibration at any temperature or use of the background calibration mode will remove these drift errors.

⁶These numbers are guaranteed by design and/or characterization.

⁷This common-mode voltage range is allowed, provided that the input voltage on AIN(+) and AIN(–) does not exceed AV _DD+ 30 mV and V_SS– 30 mV.

⁸The analog inputs present a very high impedance dynamic load which varies with clock frequency and input sample rate. The maximum recommended source

resistance depends on the selected gain (see Tables IV and V).

⁹The analog input voltage range on the AIN1(+) input is given here with respect to the voltage on the AIN1(–) input. The input voltage range on the AIN2 input is

with respect to AGND. The absolute voltage on the analog inputs should not go more positive than A V_DD+ 30 mV or go more negative than V_SS– 30 mV.

10

V

= REF IN(+) – REF IN(–).

REF

¹¹The reference input voltage range may be restricted by the input voltage range requirement on the V _BIASinput.

–2–

REV. F

AD7711

Parameter

A, S Versions¹

Units

Conditions/Comments

V_BIASINPUT¹²

Input Voltage Range

AV_DD– 0.85 × V_REF

See V_BIASInput Section

or AV_DD– 3.5

V max

V min

Whichever Is Smaller; +5 V/–5 V or +10 V/0 V

Nominal AV_DD/V_SS

Whichever Is Smaller; +5 V/0 V Nominal AV_DD/V_SS

See V_BIASInput Section

Whichever Is Greater; +5 V/–5 V or +10 V/0 V

Nominal AV_DD/V_SS

or AV_DD– 2.1

V_SS+ 0.85 × V_REF

or V_SS+ 3

or V_SS+ 2.1

65 to 85

V min

dB typ

Whichever Is Greater; +5 V/0 V Nominal AV_DD/V_SS

Increasing with Gain

V_BIASRejection

LOGIC INPUTS

Input Current

±10

µA max

All Inputs except MCLK IN

V_INL, Input Low Voltage

V_INH, Input High Voltage

MCLK IN Only

V_INL, Input Low Voltage

V_INH, Input High Voltage

0.8

2.0

V max

V min

0.8

3.5

V max

V min

LOGIC OUTPUTS

V_OL, Output Low Voltage

V_OH, Output High Voltage

Floating State Leakage Current

Floating State Output Capacitance¹³

0.4

4.0

±10

9

V max

V min

µA max

pF typ

I_SINK= 1.6 mA

I_SOURCE= 100 µA

TRANSDUCER BURNOUT

Current

Initial Tolerance @ +25°C

Drift

4.5

±10

0.1

µA nom

% typ

%/°C typ

RTD EXCITATION CURRENTS (RTD1, RTD2)

Output Current

Initial Tolerance @ +25°C

Drift

Initial Matching @ +25°C

Drift Matching

200

±20

20

±1

3

200

AV_DD– 2

µA nom

% max

ppm/°C typ

% max

ppm/°C typ

nA/V max

V max

Matching Between RTD1 and RTD2 Currents

Matching Between RTD1 and RTD2 Current Drift

AV_DD= +5 V

Line Regulation (AV_DD

Load Regulation

)

Output Compliance

SYSTEM CALIBRATION

Positive Full-Scale Calibration Limit¹⁴

Negative Full-Scale Calibration Limit¹⁴

Offset Calibration Limit¹⁵

(1.05 × V_REF)/GAIN

–(1.05 × V_REF)/GAIN

0.8 × V_REF/GAIN

V max

V min

V max

GAIN Is the Selected PGA Gain (Between 1 and 128)

Input Span¹⁵

(2.1 × V_REF)/GAIN

NOTES

¹²The AD7711 is tested with the following V_BIASvoltages. With AV_DD= +5 V and V_SS= 0 V, V_BIAS= +2.5 V; with AV_DD= +10 V and V_SS= 0 V, V_BIAS= +5 V and

with AV_DD= +5 V and V_SS= –5 V, V_BIAS= 0 V.

¹³Guaranteed by design, not production tested.

¹⁴After calibration, if the analog input exceeds positive full scale, the converter will output all 1s. If the analog input is less than negative full scale, then the device will

output all 0s.

¹⁵These calibration and span limits apply provided the absolute voltage on the analog inputs does not exceed AV _DD+ 30 mV or go more negative than V_SS– 30 mV.

The offset calibration limit applies to both the unipolar zero point and the bipolar zero point.

REV. F

–3–

AD7711–SPECIFICATIONS

Parameter

A, S Versions¹

Units

Conditions/Comments

POWER REQUIREMENTS

Power Supply Voltages

AV_DDVoltage¹⁶

+5 to +10

+5

+10.5

V nom

V max

±5% for Specified Performance

For Specified Performance

DV_DDVoltage¹⁷

AV_DD– V_SSVoltage

Power Supply Currents

AV_DDCurrent

4

4.5

1.5

mA max

DV_DDCurrent

V_SSCurrent

V_SS= –5 V

Power Supply Rejection¹⁸

Positive Supply (AV_DDand DV_DD

Rejection w.r.t. AGND; Assumes V_BIASIs Fixed

)

See Note 19

90

dB typ

Negative Supply (V_SS

Power Dissipation

Normal Mode

)

45

52.5

15

mW max

AV_DD= DV_DD= +5 V, V_SS= 0 V; Typically 25 mW

AV_DD= DV_DD= +5 V, V_SS= –5 V; Typically 30 mW

AV_DD= DV_DD= +5 V, V_SS= 0 V or –5 V; Typically 7 mW

Standby (Power-Down) Dissipation

NOTES

¹⁶The AD7711 is specified with a 10 MHz clock for AV_DDvoltages of +5 V ± 5%. It is specified with an 8 MHz clock for AV_DDvoltages greater than 5.25 V and less

than 10.5 V.

¹⁷The ±5% tolerance on the DV_DDinput is allowed provided that DV_DDdoes not exceed AV_DDby more than 0.3 V.

¹⁸Measured at dc and applies in the selected passband. PSRR at 50 Hz will exceed 120 dB with filter notches of 10 Hz, 25 Hz or 50 Hz. PSRR at 60 Hz will exceed

120 dB with filter notches of 10 Hz, 30 Hz or 60 Hz.

¹⁹PSRR depends on gain: Gain of 1 = 70 dB typ; Gain of 2: 75 dB typ; Gain of 4 = 80 dB typ; Gains of 8 to 128 = 85 dB typ. These numbers can be improved (to

95 dB typ) by deriving the V_BIASvoltage (via Zener diode or reference) from the AV_DDsupply.

Specifications subject to change without notice.

ABSOLUTE MAXIMUM RATINGS*

(T_A= +25°C, unless otherwise noted)

REF OUT to AGND . . . . . . . . . . . . . . . . . . . . –0.3 V to AV_DD

Digital Input Voltage to DGND . . . . . –0.3 V to AV_DD+ 0.3 V

Digital Output Voltage to DGND . . . –0.3 V to DV_DD+ 0.3 V

Operating Temperature Range

Commercial (A Version) . . . . . . . . . . . . . . . . –40°C to +85°C

Extended (S Version) . . . . . . . . . . . . . . . . . –55°C to +125°C

Storage Temperature Range . . . . . . . . . . . . . –65°C to +150°C

Lead Temperature (Soldering, 10 secs) . . . . . . . . . . . . +300°C

Power Dissipation (Any Package) to +75°C . . . . . . . . 450 mW

AV_DDto DV_DD. . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +12 V

AV_DDto V_SS. . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +12 V

AV_DDto AGND . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +12 V

AV_DDto DGND . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +12 V

DV_DDto AGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +6 V

DV_DDto DGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +6 V

V_SSto AGND . . . . . . . . . . . . . . . . . . . . . . . . . +0.3 V to –6 V

V_SSto DGND . . . . . . . . . . . . . . . . . . . . . . . . . +0.3 V to –6 V

Analog Input Voltage to AGND

*Stresses above those listed under Absolute Maximum Ratings may cause perma-

nent damage to the device. This is a stress rating only; functional operation of the

device at these or any other conditions above those listed in the operational

sections of the specification is not implied. Exposure to absolute maximum rating

conditions for extended periods may affect device reliability.

. . . . . . . . . . . . . . . . . . . . . . . . . V_SS– 0.3 V to AV_DD+ 0.3 V

Reference Input Voltage to AGND

. . . . . . . . . . . . . . . . . . . . . . . . . V_SS– 0.3 V to AV_DD+ 0.3 V

ORDERING GUIDE

Temperature Range

Model

Package Option*

AD7711AN

AD7711AR

AD7711AQ

AD7711SQ

–40°C to +85°C

–55°C to +125°C

N-24

R-24

Q-24

EVAL-AD7711EB Evaluation Board

*N = Plastic DIP, Q = Cerdip; R = SOIC.

CAUTION

ESD (electrostatic discharge) sensitive device. The digital control inputs are diode protected;

however, permanent damage may occur on unconnected devices subject to high energy electro-

static fields. Unused devices must be stored in conductive foam or shunts. The protective foam

should be discharged to the destination socket before devices are inserted.

WARNING!

ESD SENSITIVE DEVICE

–4–

REV. F

AD7711

(DV_DD= +5 V ؎ 5%; AV_DD= +5 V or +10 V³؎ 5%; V_SS= 0 V or –5 V ؎ 10%; AGND = DGND =

0 V; f_{CLK IN}= 10 MHz; Input Logic 0 = 0 V, Logic 1 = DV_DD, unless otherwise noted.)

TIMING CHARACTERISTICS^{1, 2}

Limit at T_MIN, T_MAX

(A, S Versions)

Parameter

Units

Conditions/Comments

4, 5

f_{CLK IN}

400

kHz min

Master Clock Frequency: Crystal Oscillator or Externally

Supplied for Specified Performance

10

MHz max

ns min

ns max

ns min

2

t_{CLK IN LO}

0.4 × t_{CLK IN}

50

1000

Master Clock Input Low Time; t_{CLK IN}= 1/f_{CLK IN}

Master Clock Input High Time

Digital Output Rise Time. Typically 20 ns

Digital Output Fall Time. Typically 20 ns

SYNC Pulsewidth

t_{CLK IN HI}

t_r⁶

t_f⁶

t₁

Self-Clocking Mode

t₂

t₃

t₄

0

ns min

ns max

ns min

ns max

ns nom

ns min

ns max

ns min

DRDY to RFS Setup Time

DRDY to RFS Hold Time

A0 to RFS Setup Time

A0 to RFS Hold Time

RFS Low to SCLK Falling Edge

Data Access Time (RFS Low to Data Valid)

SCLK Falling Edge to Data Valid Delay

2 × t_{CLK IN}

0

t₅

t₆₇

t₇₇

t₈

4 × t_{CLK IN}+ 20

t_{CLK IN}/2

t_{CLK IN}/2 + 30

t_{CLK IN}/2

3 × t_{CLK IN}/2

50

t₉

SCLK High Pulsewidth

SCLK Low Pulsewidth

A0 to TFS Setup Time

A0 to TFS Hold Time

TFS to SCLK Falling Edge Delay Time

TFS to SCLK Falling Edge Hold Time

Data Valid to SCLK Setup Time

Data Valid to SCLK Hold Time

t₁₀

t₁₄

t₁₅

t₁₆

t₁₇

t₁₈

t₁₉

0

4 × t_{CLK IN}+ 20

4 × t_{CLK IN}

0

10

REV. F

–5–

AD7711

Limit at T_MIN, T_MAX

(A, S Versions)

Parameter

Units

Conditions/Comments

External Clocking Mode

f_SCLK

t₂₀

t₂₁

t₂₂

f_{CLK IN}/5

0

2 × t_{CLK IN}

0

4 × t_{CLK IN}

10

2 × t_{CLK IN}+ 20

2 × t_{CLK IN}

t_{CLK IN}+ 10

MHz max

ns min

ns max

ns min

ns max

ns min

ns max

ns min

ns max

ns min

ns max

ns min

Serial Clock Input Frequency

DRDY to RFS Setup Time

DRDY to RFS Hold Time

A0 to RFS Setup Time

A0 to RFS Hold Time

Data Access Time (RFS Low to Data Valid)

SCLK Falling Edge to Data Valid Delay

t₂₃

7

t₂₄

7

t₂₅

t₂₆

t₂₇

t₂₈

SCLK High Pulsewidth

SCLK Low Pulsewidth

SCLK Falling Edge to DRDY High

SCLK to Data Valid Hold Time

8

t₂₉

10

t_{CLK IN}+ 10

10

5 × t_{CLK IN}/2 + 50

0

t₃₀

t₃₁

RFS/TFS to SCLK Falling Edge Hold Time

RFS to Data Valid Hold Time

A0 to TFS Setup Time

8

t₃₂

t₃₃

t₃₄

t₃₅

t₃₆

0

A0 to TFS Hold Time

4 × t_{CLK IN}

2 × t_{CLK IN}– SCLK High

30

SCLK Falling Edge to TFS Hold Time

Data Valid to SCLK Setup Time

Data Valid to SCLK Hold Time

NOTES

¹Guaranteed by design, not production tested. All input signals are specified with tr = tf = 5 ns (10% to 90% of 5 V) and timed from a voltage level of 1.6 V.

²See Figures 10 to 13.

³The AD7711 is specified with a 10 MHz clock for AV_DDvoltages of +5 V ± 5%. It is specified with an 8 MHz clock for AV_DDvoltages greater than 5.25 V and less

than 10.5 V.

⁴CLK IN duty cycle range is 45% to 55%. CLK IN must be supplied whenever the AD7711 is not in STANDBY mode. If no clock is present in this case, the device

can draw higher current than specified and possibly become uncalibrated.

⁵The AD7711 is production tested with f_{CLK IN}at 10 MHz (8 MHz for AV_DD> +5.25 V). It is guaranteed by characterization to operate at 400 kHz.

⁶Specified using 10% and 90% points on waveform of interest.

⁷These numbers are measured with the load circuit of Figure 1 and defined as the time required for the output to cross 0.8 V or 2.4 V.

⁸These numbers are derived from the measured time taken by the data output to change 0.5 V when loaded with the circuit of Figure 1. The measured number is then

extrapolated back to remove effects of charging or discharging the 100 pF capacitor. This means that the times quoted in the timing characteristics are the true bus

relinquish times of the part and, as such, are independent of external bus loading capacitances.

Specifications subject to change without notice.

PIN CONFIGURATION

DIP AND SOIC

1.6mA

1

2

24

23

DGND

DV

SCLK

MCLK IN

MCLK OUT

A0

DD

3

22 SDATA

TO OUTPUT

+2.1V

4

21

DRDY

100pF

5

20

RFS

SYNC

MODE

AIN1(+)

AIN1(–)

RTD1

AD7711

TOP VIEW

(Not to Scale)

200␮A

6

19

TFS

7

18

AGND

8

17

AIN2

Figure 1. Load Circuit for Access Time and Bus Relinquish

Time

9

16 REF OUT

10

11

12

15

14

13

RTD2

REF IN(+)

REF IN(–)

V

SS

AV

V

BIAS

DD

–6–

REV. F

AD7711

PIN FUNCTION DESCRIPTION

Pin Mnemonic

Function

1

SCLK

Serial Clock. Logic Input/Output depending on the status of the MODE pin. When MODE is high, the

device is in its self-clocking mode and the SCLK pin provides a serial clock output. This SCLK becomes

active when RFS or TFS goes low and it goes high impedance when either RFS or TFS returns high or when

the device has completed transmission of an output word. When MODE is low, the device is in its external

clocking mode and the SCLK pin acts as an input. This input serial clock can be a continuous clock with all

data transmitted in a continuous train of pulses. Alternatively, it can be a noncontinuous clock with the

information being transmitted to the AD7711 in smaller batches of data.

2

MCLK IN

Master Clock signal for the device. This can be provided in the form of a crystal or external clock. A crystal can

be tied across the MCLK IN and MCLK OUT pins. Alternatively, the MCLK IN pin can be driven with a

CMOS-compatible clock and MCLK OUT left unconnected. The clock input frequency is nominally 10 MHz.

3

4

MCLK OUT

A0

When the master clock for the device is a crystal, the crystal is connected between MCLK IN and MCLK OUT.

Address Input. With this input low, reading and writing to the device is to the control register. With this input

high, access is to either the data register or the calibration registers.

5

6

7

SYNC

Logic Input which allows for synchronization of the digital filters when using a number of AD7711s. It resets

the nodes of the digital filter.

MODE

AIN1(+)

Logic Input. When this pin is high, the device is in its self-clocking mode; with this pin low, the device is in its

external clocking mode.

Analog Input Channel 1. Positive input of the programmable gain differential analog input. The AIN1(+) input

is connected to an output current source which can be used to check that an external transducer has burned out

or gone open circuit. This output current source can be turned on/off via the control register.

8

9

AIN1(–)

RTD1

Analog Input Channel 1. Negative input of the programmable gain differential analog input.

Constant Current Output. A nominal 200 µA constant current is provided at this pin, and this can be used

as the excitation current for RTDs. This current can be turned on or off via the control register.

10

11

RTD2

V_SS

Constant Current Output. A nominal 200 µA constant current is provided at this pin, and this can be used

as the excitation current for RTDs. This current can be turned on or off via the control register. This

second current can be used to eliminate lead resistance errors in three-wire RTD configurations.

Analog Negative Supply, 0 V to –5 V. Tied to AGND for single supply operation. The input voltage on AIN1

or AIN2 should not go > 30 mV negative w.r.t. V_SSfor correct operation of the device.

12

13

AV_DD

V_BIAS

Analog Positive Supply Voltage, +5 V to +10 V.

Input Bias Voltage. This input voltage should be set such that V_BIAS+ 0.85 × V_REF< AV_DDand V_BIAS– 0.85

× V_REF> V_SSwhere V_REFis REF IN(+) – REF IN(–). Ideally, this should be tied halfway between AV_DD

and V_SS. Thus with AV_DD= +5 V and V_SS= 0 V, it can be tied to REF OUT; with AV_DD= +5 V and V_SS

–5 V, it can be tied to AGND, while with AV_DD= +10 V, it can be tied to +5 V.

=

14

15

16

REF IN(–)

REF IN(+)

REF OUT

Reference Input. The REF IN(–) can lie anywhere between AV_DDand V_SSprovided REF IN(+) is greater

than REF IN(–).

Reference Input. The reference input is differential providing that REF IN(+) is greater than REF IN(–).

REF IN(+) can lie anywhere between AV_DDand V_SS.

Reference Output. The internal +2.5 V reference is provided at this pin. This is a single-ended output

which is referred to AGND. It is a buffered output which is capable of providing 1 mA to an external load.

17

18

19

AIN2

AGND

TFS

Analog Input Channel 2. Single-ended programmable gain analog input.

Ground reference point for analog circuitry.

Transmit Frame Synchronization. Active low logic input used to write serial data to the device with serial

data expected after the falling edge of this pulse. In the self-clocking mode, the serial clock becomes active

after TFS goes low. During a write operation to the AD7711, the SDATA line should not return to high

impedance until after TFS returns high.

REV. F

–7–

AD7711

Pin Mnemonic

Function

20

RFS

Receive Frame Synchronization. Active low logic input used to access serial data from the device. In the

self-clocking mode, the SCLK and SDATA lines both become active after RFS goes low. In the external

clocking mode, the SDATA line becomes active after RFS goes low.

21

DRDY

Logic output. A falling edge indicates that a new output word is available for transmission. The DRDY pin

will return high upon completion of transmission of a full output word. DRDY is also used to indicate

when the AD7711 has completed its on-chip calibration sequence.

22 SDATA

Serial Data. Input/Output with serial data being written to either the control register or the calibration

registers and serial data being accessed from the control register, calibration registers or the data register.

During an output data read operation, serial data becomes active after RFS goes low (provided DRDY is

low). During a write operation, valid serial data is expected on the rising edges of SCLK when TFS is low.

The output data coding is natural binary for unipolar inputs and offset binary for bipolar inputs.

23 DV_DD

Digital Supply Voltage, +5 V. DV_DDshould not exceed AV_DDby more than 0.3 V in normal operation.

Ground reference point for digital circuitry.

24 DGND

TERMINOLOGY

POSITIVE FULL-SCALE OVERRANGE

INTEGRAL NONLINEARITY

Positive full-scale overrange is the amount of overhead available

to handle input voltages on AIN1(+) input greater than

AIN1(–) + V_REF/GAIN or on the AIN2 input greater than +

V_REF/GAIN (for example, noise peaks or excess voltages due to

system gain errors in system calibration routines) without intro-

ducing errors due to overloading the analog modulator or to

overflowing the digital filter.

This is the maximum deviation of any code from a straight line

passing through the endpoints of the transfer function. The end-

points of the transfer function are zero-scale (not to be confused

with bipolar zero), a point 0.5 LSB below the first code transi-

tion (000 . . . 000 to 000 . . . 001) and full scale, a point 0.5 LSB

above the last code transition (111 . . . 110 to 111 . . . 111). The

error is expressed as a percentage of full scale.

NEGATIVE FULL-SCALE OVERRANGE

POSITIVE FULL-SCALE ERROR

This is the amount of overhead available to handle voltages on

AIN1(+) below AIN1(–) – V_REF/GAIN or on AIN2 below

–V_REF/GAIN without overloading the analog modulator or over-

flowing the digital filter. Note that the analog input will accept

negative voltage peaks on AIN1(+) even in the unipolar mode

provided that AIN1(+) is greater than AIN1(–) and greater than

V_SS– 30 mV.

Positive full-scale error is the deviation of the last code transi-

tion (111 . . . 110 to 111 . . . 111) from the ideal input full-scale

voltage. For AIN1(+), the ideal full-scale input voltage is

(AIN1(–) + V_REF/GAIN – 3/2 LSBs); for AIN2, the ideal full-

scale input voltage is V_REF/GAIN – 3/2 LSBs. It applies to both

unipolar and bipolar analog input ranges.

UNIPOLAR OFFSET ERROR

OFFSET CALIBRATION RANGE

Unipolar offset error is the deviation of the first code transition

from the ideal voltage. For AIN1(+), the ideal input voltage is

(AIN1(–) + 0.5 LSB); for AIN2, the ideal input is 0.5 LSB

when operating in the unipolar mode.

In the system calibration modes, the AD7711 calibrates its

offset with respect to the analog input. The offset calibration

range specification defines the range of voltages that the

AD7711 can accept and still calibrate offset accurately.

BIPOLAR ZERO ERROR

FULL-SCALE CALIBRATION RANGE

This is the deviation of the midscale transition (0111 . . . 111

to 1000 . . . 000) from the ideal input voltage. For AIN1(+), the

ideal input voltage is (AIN1(–) – 0.5 LSB); for AIN2, the ideal

input is – 0.5 LSB when operating in the bipolar mode.

This is the range of voltages that the AD7711 can accept in the

system calibration mode and still calibrate full-scale correctly.

INPUT SPAN

In system calibration schemes, two voltages applied in sequence

to the AD7711’s analog input define the analog input range.

The input span specification defines the minimum and maxi-

mum input voltages from zero to full-scale that the AD7711

can accept and still calibrate gain accurately.

BIPOLAR NEGATIVE FULL-SCALE ERROR

This is the deviation of the first code transition from the ideal

input voltage. For (AIN1(+), the ideal input voltage is (AIN1(–)

– V_REF/GAIN + 0.5 LSB); for AIN2 the ideal input is – V_REF

GAIN + 0.5 LSB when operating in the bipolar mode.

/

–8–

REV. F

AD7711

CONTROL REGISTER (24 BITS)

A write to the device with the A0 input low writes data to the control register. A read to the device with the A0 input low accesses the

contents of the control register. The control register is 24-bits wide and when writing to the register 24 bits of data must be written

otherwise the data will not be loaded to the control register. In other words, it is not possible to write just the first 12-bits of data into

the control register. If more than 24 clock pulses are provided before TFS returns high, then all clock pulses after the 24th clock

pulse are ignored. Similarly, a read operation from the control register should access 24 bits of data.

MSB

2

MD2

FS11

MD1

FS10

MD0

FS9

G2

G1

G0

CH

PD

WL

FS3

RO

BO

B/U

FS8

FS7

FS6

FS5

FS4

FS2

FS1

FS0

LSB

Operating Mode

MD2

MD1

MD0 Operating Mode

0

Normal Mode. This is the normal mode of operation of the device whereby a read to the device with A0

high accesses data from the data register. This is the default condition of these bits after the internal

power on reset.

0

1

Activate Self-Calibration. This activates self-calibration on the channel selected by CH. This is a one-step

calibration sequence, and when complete, the part returns to normal mode (with MD2, MD1, MD0 of

the control register returning to 0, 0, 0). The DRDY output indicates when this self-calibration is complete.

For this calibration type, the zero-scale calibration is done internally on shorted (zeroed) inputs and the

full-scale calibration is done internally on V_REF

.

0

Activate System Calibration. This activates system calibration on the channel selected by CH. This is a

two-step calibration sequence, with the zero-scale calibration done first on the selected input channel and

DRDY indicating when this zero-scale calibration is complete. The part returns to normal mode at the

end of this first step in the two-step sequence.

0

1

0

1

0

Activate System Calibration. This is the second step of the system calibration sequence with full-scale

calibration being performed on the selected input channel. Once again, DRDY indicates when the full-

scale calibration is complete. When this calibration is complete, the part returns to normal mode.

Activate System Offset Calibration. This activates system offset calibration on the channel selected by

CH. This is a one-step calibration sequence and, when complete, the part returns to normal mode with

DRDY indicating when this system offset calibration is complete. For this calibration type, the zero-scale

calibration is done on the selected input channel and the full-scale calibration is done internally on V_REF

.

1

0

1

Activate Background Calibration. This activates background calibration on the channel selected by CH. If

the background calibration mode is on, then the AD7711 provides continuous self-calibration of the

reference and shorted (zeroed) inputs. This calibration takes place as part of the conversion sequence,

extending the conversion time and reducing the word rate by a factor of six. Its major advantage is that

the user does not have to worry about recalibrating the device when there is a change in the ambient

temperature. In this mode, the shorted (zeroed) inputs and V_REF, as well as the analog input voltage, are

continuously monitored and the calibration registers of the device are automatically updated.

1

0

1

Read/Write Zero-Scale Calibration Coefficients. A read to the device with A0 high accesses the contents

of the zero-scale calibration coefficients of the channel selected by CH. A write to the device with A0 high

writes data to the zero-scale calibration coefficients of the channel selected by CH. The word length for

reading and writing these coefficients is 24 bits, regardless of the status of the WL bit of the control

register. Therefore, when writing to the calibration register 24 bits of data must be written, otherwise the

new data will not be transferred to the calibration register.

Read/Write Full-Scale Calibration Coefficients. A read to the device with A0 high accesses the contents of

the full-scale calibration coefficients of the channel selected by CH. A write to the device with A0 high

writes data to the full-scale calibration coefficients of the channel selected by CH. The word length for

reading and writing these coefficients is 24 bits, regardless of the status of the WL bit of the control

register. Therefore, when writing to the calibration register 24 bits of data must be written, otherwise the

new data will not be transferred to the calibration register.

REV. F

–9–

AD7711

PGA Gain

G2

Gl

G0

Gain

0

1

0

1

0

1

0

1

0

1

0

1

0

1

2

4

8

16

32

64

128

(Default Condition After the Internal Power-On Reset)

Channel Selection

CH

0

1

Channel

AIN1

AIN2

(Default Condition After the Internal Power-On Reset)

(Default Condition After Internal Power-On Reset)

Power-Down

PD

0

Normal Operation

Power-Down

1

Word Length

WL

0

Output Word Length

16-bit

24-bit

1

RTD Excitation Current

IO

0

1

Off

On

Burnout Current

BO

0

1

Off

On

Bipolar/Unipolar Selection (Both Inputs)

B/U

0

1

Bipolar

Unipolar

(Default Condition After Internal Power-On Reset)

Filter Selection (FS11–FS0)

The on-chip digital filter provides a Sinc³(or (Sinx/x)³) filter response. The 12 bits of data programmed into these bits determine

the filter cutoff frequency, the position of the first notch of the filter and the data rate for the part. In association with the gain selec-

tion, it also determines the output noise (and hence the effective resolution) of the device.

The first notch of the filter occurs at a frequency determined by the relationship: filter first notch frequency = (f_{CLK IN}/512)/code

where code is the decimal equivalent of the code in bits FS0 to FS11 and is in the range 19 to 2,000. With the nominal f_{CLK IN}of

10 MHz, this results in a first notch frequency range from 9.76 Hz to 1.028 kHz. To ensure correct operation of the AD7711, the

value of the code loaded to these bits must be within this range. Failure to do this will result in unspecified operation of the device.

Changing the filter notch frequency, as well as the selected gain, impacts resolution. Tables I and II and Figure 2 show the effect of

the filter notch frequency and gain on the effective resolution of the AD7711. The output data rate (or effective conversion time) for

the device is equal to the frequency selected for the first notch of the filter. For example, if the first notch of the filter is selected at

50 Hz, then a new word is available at a 50 Hz rate or every 20 ms. If the first notch is at 1 kHz, a new word is available every 1 ms.

The settling time of the filter to a full-scale step input change is worst case 4 × 1/(output data rate). This settling time is to 100% of

the final value. For example, with the first filter notch at 50 Hz, the settling time of the filter to a full-scale step input change is

80 ms max. If the first notch is at 1 kHz, the settling time of the filter to a full-scale input step is 4 ms max. This settling time can be

reduced to 3 × 1/(output data rate) by synchronizing the step input change to a reset of the digital filter. In other words, if the step

input takes place with SYNC low, the settling time will be 3 × 1/(output data rate). If a change of channels takes place, the settling

time is 3 × 1/(output data rate) regardless of the SYNC input.

The –3 dB frequency is determined by the programmed first notch frequency according to the relationship: filter –3 dB frequency

= 0.262 × first notch frequency.

–10–

REV. F

AD7711

Tables I and II show the output rms noise for some typical notch and –3 dB frequencies. The numbers given are for the bipolar

input ranges with a V_REFof +2.5 V. These numbers are typical and are generated with an analog input voltage of 0 V. The output

noise from the part comes from two sources. The first is the electrical noise in the semiconductor devices used in the implementation

of the modulator (device noise). The second occurs when the analog input signal is converted into the digital domain adding quanti-

zation noise. The device noise is at a low level and is largely independent of frequency. The quantization noise starts at an even

lower level but rises rapidly with increasing frequency to become the dominant noise source. Consequently, lower filter notch set-

tings (below 60 Hz approximately) tend to be device noise dominated while higher notch settings are dominated by quantization

noise. Changing the filter notch and cutoff frequency in the quantization noise dominated region results in a more dramatic im-

provement in noise performance than it does in the device noise dominated region as shown in Table I. Furthermore, quantization

noise is added after the PGA, so effective resolution is independent of gain for the higher filter notch frequencies. Meanwhile, device

noise is added in the PGA and, therefore, effective resolution suffers a little at high gains for lower notch frequencies.

2

At the lower filter notch settings (below 60 Hz), the no missing codes performance of the device is at the 24-bit level. At the higher

settings, more codes will be missed until at 1 kHz notch setting, no missing codes performance is only guaranteed to the 12-bit level.

However, since the effective resolution of the part is 10.5 bits for this filter notch setting, this no missing codes performance should

be more than adequate for all applications.

The effective resolution of the device is defined as the ratio of the output rms noise to the input full scale. This does not remain

constant with increasing gain or with increasing bandwidth. Table II shows the same table as Table I except that the output is now

expressed in terms of effective resolution (the magnitude of the rms noise with respect to 2 × V_REF/GAIN, i.e., the input full scale). It

is possible to do post filtering on the device to improve the output data rate for a given –3 dB frequency and also to further reduce

the output noise (see Digital Filtering section).

Table I. Output Noise vs. Gain and First Notch Frequency

Typical Output RMS Noise (␮V)

First Notch of

Filter and O/P –3 dB

Gain of

1

Gain of

2

Gain of

4

Gain of

8

Gain of

16

Gain of

32

Gain of

64

Gain of

128

Data Rate¹

Frequency

10 Hz²

25 Hz²

30 Hz²

50 Hz²

60 Hz²

100 Hz³

250 Hz³

500 Hz³

1 kHz³

2.62 Hz

6.55 Hz

7.86 Hz

13.1 Hz

15.72 Hz

26.2 Hz

65.5 Hz

131 Hz

262 Hz

1.0

1.8

2.5

4.33

5.28

13

0.78

1.1

1.31

2.06

2.36

6.4

0.48

0.63

0.84

1.2

1.33

3.7

0.33

0.50

0.57

0.64

0.87

1.8

0.25

0.44

0.46

0.54

0.63

1.1

7.5

35

180

0.25

0.41

0.43

0.46

0.62

0.9

4

25

120

0.25

0.38

0.4

0.46

0.6

0.65

2.7

15

0.25

0.38

0.4

0.46

0.56

0.65

1.7

130

75

25

12

0.6 × 10³

3.1 × 10³

0.26 × 10³

1.6 × 10³

140

70

8

40

0.7 × 10³

0.29 × 10³

70

NOTES

¹The default condition (after the internal power-on reset) for the first notch of filter is 60 Hz.

²For these filter notch frequencies, the output rms noise is primarily dominated by device noise and as a result is independent of the value of the reference voltage.

Therefore, increasing the reference voltage will give an increase in the effective resolution of the device (i.e., the ratio of the rms noise to the input full scale is in-

creased since the output rms noise remains constant as the input full scale increases).

³For these filter notch frequencies, the output rms noise is dominated by quantization noise and as a result is proportional to the value of the reference voltage.

Table II. Effective Resolution vs. Gain and First Notch Frequency

Effective Resolution¹(Bits)

First Notch of

Filter and O/P –3 dB

Gain of

1

Gain of

2

Gain of

4

Gain of

8

Gain of

16

Gain of

32

Gain of

64

Gain of

128

Data Rate

Frequency

10 Hz

25 Hz

30 Hz

50 Hz

60 Hz

100 Hz

250 Hz

500 Hz

1 kHz

2.62 Hz

6.55 Hz

7.86 Hz

13.1 Hz

15.72 Hz

26.2 Hz

65.5 Hz

131 Hz

262 Hz

22.5

21.5

21

20

18.5

15

13

21.5

21

20

18.5

15

13

21.5

21

20.5

20

21

20

19.5

18.5

15.5

13

20.5

19.5

19

18

15.5

13

11

19.5

18.5

18

17.5

15.5

12.5

10.5

18.5

17.5

17

15

17.5

16.5

16

20

18.5

15.5

13

16

14.5

12.5

10

12.5

10

10.5

11

NOTE

¹Effective resolution is defined as the magnitude of the output rms noise with respect to the input full scale (i.e., 2 × V_REF/GAIN). The above table applies for

a V_REFof +2.5 V and resolution numbers are rounded to the nearest 0.5 LSB.

REV. F

–11–

AD7711

Figure 2 gives similar information to that outlined in Table I. In this plot, the output rms noise is shown for the full range of available

cutoffs frequencies rather than for some typical cutoff frequencies as in Tables I and II. The numbers given in these plots are typical

values at 25°C.

1000

100

10

10000

1000

100

10

GAIN OF 1

GAIN OF 2

GAIN OF 4

GAIN OF 8

GAIN OF 16

GAIN OF 32

GAIN OF 64

GAIN OF 128

1

0.1

10

0.1

100

1000

10000

10

100

1000

10000

NOTCH FREQUENCY – Hz

Figure 2b. Plot of Output Noise vs. Gain and Notch

Frequency (Gains of 16 to 128)

Figure 2a. Plot of Output Noise vs. Gain and Notch

Frequency (Gains of 1 to 8)

separate supplies for both AV_DDand DV_DD, and in some of

these cases, the analog supply will exceed the +5 V digital sup-

ply (see Power Supplies and Grounding section).

CIRCUIT DESCRIPTION

The AD7711 is a sigma-delta A/D converter with on-chip digital

filtering, intended for the measurement of wide dynamic range,

low frequency signals such as those in RTD applications, indus-

trial control or process control applications. It contains a sigma-

delta (or charge-balancing) ADC, a calibration microcontroller

with on-chip static RAM, a clock oscillator, a digital filter and a

bidirectional serial communications port.

ANALOG

+5V SUPPLY

10␮F

0.1␮F

AV

DV

DD

AIN1(+)

DATA READY

DRDY

DIFFERENTIAL

ANALOG INPUT

The part contains two analog input channels, a programmable

gain differential analog input and a programmable gain single

ended input. The gain range is from 1 to 128 allowing the part

to accept unipolar signals of between 0 mV to +20 mV and 0 V

to +2.5 V or bipolar signals in the range from ±20 mV to ±2.5 V

when the reference input voltage equals +2.5 V. The input

signal to the selected analog input channel is continuously

sampled at a rate determined by the frequency of the master

clock, MCLK IN, and the selected gain (see Table III). A

charge balancing A/D converter (Sigma-Delta Modulator) con-

verts the sampled signal into a digital pulse train whose duty

cycle contains the digital information. The programmable gain

function on the analog input is also incorporated in this sigma-

delta modulator with the input sampling frequency being modi-

fied to give the higher gains. A sinc³digital low-pass filter

processes the output of the sigma-delta modulator and updates

the output register at a rate determined by the first notch fre-

quency of this filter. The output data can be read from the serial

port randomly or periodically at any rate up to the output regis-

ter update rate. The first notch of this digital filter (and hence

its –3 dB frequency) can be programmed via an on-chip control

register. The programmable range for this first notch frequency

is from 9.76 Hz to 1.028 kHz, giving a programmable range for

the –3 dB frequency of 2.58 Hz to 269 Hz.

AIN1(–)

TRANSMIT (WRITE)

RECEIVE (READ)

SERIAL DATA

TFS

RFS

SINGLE-ENDED

ANALOG INPUT

AIN2

AD7711

RTD1

RTD2

SDATA

SCLK

A0

SERIAL CLOCK

ADDRESS INPUT

AGND

ANALOG GROUND

DIGITAL GROUND

V

SS

MODE

DGND

REF OUT

REF IN(+)

+5V

SYNC

MCLK OUT

V

BIAS

REF IN(–)

MCLK IN

Figure 3. Basic Connection Diagram

The AD7711 provides a number of calibration options which

can be programmed via the on-chip control register. A calibra-

tion cycle may be initiated at any time by writing to this control

register. The part can perform self-calibration using the on-chip

calibration microcontroller and SRAM to store calibration pa-

rameters. Other system components may also be included in the

calibration loop to remove offset and gain errors in the input

channel using the system calibration mode. Another option is a

background calibration mode where the part continuously per-

forms self-calibration and updates the calibration coefficients.

Once the part is in this mode, the user does not have to worry

about issuing periodic calibration commands to the device or

asking the device to recalibrate when there is a change in the

ambient temperature or power supply voltage.

The basic connection diagram for the part is shown in Figure 3.

This shows the AD7711 in the external clocking mode with

both the AV_DDand DV_DDpins of the AD7711 being driven

from the analog +5 V supply. Some applications will have

–12–

REV. F

AD7711

The AD7711 gives the user access to the on-chip calibration

registers allowing the microprocessor to read the device’s cali-

bration coefficients and also to write its own calibration coeffi-

cients to the part from prestored values in E²PROM. This gives

the microprocessor much greater control over the AD7711’s

calibration procedure. It also means that the user can verify that

the device has performed its calibration correctly by comparing the

coefficients after calibration with prestored values in E²PROM.

Sigma-delta ADCs are generally described by the order of the

analog low-pass filter. A simple example of a first order sigma-

delta ADC is shown in Figure 5. This contains only a first order

low-pass filter or integrator. It also illustrates the derivation of

the alternative name for these devices: Charge-Balancing ADCs.

DIFFERENTIAL

AMPLIFIER

INTEGRATOR

COMPARATOR

V

IN

2

͐

The AD7711 can be operated in single supply systems provided

that the analog input voltage does not go more negative than

–30 mV. For larger bipolar signals, a V_SSof –5 V is required by

the part. For battery operation, the AD7711 also offers a soft-

ware-programmable standby mode that reduces idle power

consumption to typically 7 mW.

+FS

DAC

–FS

Figure 5. Basic Charge-Balancing ADC

THEORY OF OPERATION

The general block diagram of a sigma-delta ADC is shown in

Figure 4. It contains the following elements:

It consists of a differential amplifier (whose output is the differ-

ence between the analog input and the output of a 1-bit DAC),

an integrator and a comparator. The term charge-balancing,

comes from the fact that this system is a negative feedback loop

that tries to keep the net charge on the integrator capacitor at

zero, by balancing charge injected by the input voltage with

charge injected by the 1-bit DAC. When the analog input is

zero, the only contribution to the integrator output comes from

the 1-bit DAC. For the net charge on the integrator capacitor to

be zero, the DAC output must spend half its time at +FS and

half its time at –FS. Assuming ideal components, the duty cycle

of the comparator will be 50%.

1. A sample-hold amplifier.

2. A differential amplifier or subtracter.

3. An analog low-pass filter.

4. A 1-bit A/D converter (comparator).

5. A 1-bit DAC.

6. A digital low-pass filter.

COMPARATOR

S/H AMP

ANALOG

LOW-PASS

FILTER

+

When a positive analog input is applied, the output of the 1-bit

DAC must spend a larger proportion of the time at +FS, so the

duty cycle of the comparator increases. When a negative input

voltage is applied, the duty cycle decreases.

DIGITAL

FILTER

–

DIGITAL

DATA

DAC

The AD7711 uses a second order sigma-delta modulator and a

digital filter that provides a rolling average of the sampled out-

put. After power-up, or if there is a step change in the input

voltage, there is a settling time that must elapse before valid

data is obtained.

Figure 4. General Sigma-Delta ADC

In operation, the analog signal sample is fed to the subtracter,

along with the output of the 1-bit DAC. The filtered difference

signal is fed to the comparator, whose output samples the differ-

ence signal at a frequency many times that of the analog signal

sampling frequency (oversampling).

Input Sample Rate

The modulator sample frequency for the device remains at

f_{CLK IN}/512 (19.5 kHz @ f_{CLK IN}= 10 MHz) regardless of the

selected gain. However, gains greater than ×1 are achieved by a

combination of multiple input samples per modulator cycle and

a scaling of the ratio of reference capacitor to input capacitor.

As a result of the multiple sampling, the input sample rate of

the device varies with the selected gain (see Table III). The

effective input impedance is 1/C × f_Swhere C is the input sam-

pling capacitance and f_Sis the input sample rate.

Oversampling is fundamental to the operation of sigma-delta

ADCs. Using the quantization noise formula for an ADC:

SNR = (6.02 × number of bits + 1.76) dB,

a 1-bit ADC or comparator yields an SNR of 7.78 dB.

The AD7711 samples the input signal at a frequency of 39 kHz or

greater (see Table III). As a result, the quantization noise is

spread over a much wider frequency than that of the band of

interest. The noise in the band of interest is reduced still further

by analog filtering in the modulator loop, which shapes the

quantization noise spectrum to move most of the noise energy to

frequencies outside the bandwidth of interest. The noise perfor-

mance is thus improved from this 1-bit level to the performance

outlined in Tables I and II and in Figure 2.

Table III. Input Sampling Frequency vs. Gain

Gain

Input Sampling Frequency (f_S)

1

2

4

8

16

32

64

128

f

_{CLK IN}/256 (39 kHz @ f_{CLK IN}= 10 MHz)

2 × f_{CLK IN}/256 (78 kHz @ f_{CLK IN}= 10 MHz)

4 × f_{CLK IN}/256 (156 kHz @ f_{CLK IN}= 10 MHz)

8 × f_{CLK IN}/256 (312 kHz @ f_{CLK IN}= 10 MHz)

The output of the comparator provides the digital input for the

1-bit DAC, so that the system functions as a negative feedback

loop that tries to minimize the difference signal. The digital data

that represents the analog input voltage is contained in the duty

cycle of the pulse train appearing at the output of the compara-

tor. It can be retrieved as a parallel binary data word using a

digital filter.

REV. F

–13–

AD7711

DIGITAL FILTERING

The AD7711’s digital filter behaves like a similar analog filter,

with a few minor differences.

Post Filtering

The on-chip modulator provides samples at a 19.5 kHz output

rate. The on-chip digital filter decimates these samples to pro-

vide data at an output rate which corresponds to the pro-

grammed first notch frequency of the filter. Since the output

data rate exceeds the Nyquist criterion, the output rate for a

given bandwidth will satisfy most application requirements.

However, there may be some applications which require a

higher data rate for a given bandwidth and noise performance.

Applications which need this higher data rate will require some

post filtering following the digital filter of the AD7711.

First, since digital filtering occurs after the A-to-D conversion

process, it can remove noise injected during the conversion

process. Analog filtering cannot do this.

On the other hand, analog filtering can remove noise super-

imposed on the analog signal before it reaches the ADC. Digital

filtering cannot do this and noise peaks riding on signals near

full scale have the potential to saturate the analog modulator

and digital filter, even though the average value of the signal is

within limits. To alleviate this problem, the AD7711 has

overrange headroom built into the sigma-delta modulator and

digital filter which allows overrange excursions of 5% above the

analog input range. If noise signals are larger than this, consid-

eration should be given to analog input filtering, or to reducing

the input channel voltage so that its full scale is half that of the

analog input channel full scale. This will provide an overrange

capability greater than 100% at the expense of reducing the

dynamic range by 1 bit (50%).

For example, if the required bandwidth is 7.86 Hz but the re-

quired update rate is 100 Hz, the data can be taken from the

AD7711 at the 100 Hz rate giving a –3 dB bandwidth of

26.2 Hz. Post filtering can be applied to this to reduce the band-

width and output noise, to the 7.86 Hz bandwidth level, while

maintaining an output rate of 100 Hz.

Post filtering can also be used to reduce the output noise from

the device for bandwidths below 2.62 Hz. At a gain of 128, the

output rms noise is 250 nV. This is essentially device noise or

white noise, and since the input is chopped, the noise has a flat

frequency response. By reducing the bandwidth below 2.62 Hz,

the noise in the resultant passband can be reduced. A reduction

in bandwidth by a factor of two results in a √2 reduction in the

output rms noise. This additional filtering will result in a longer

settling time.

Filter Characteristics

The cutoff frequency of the digital filter is determined by the

value loaded to bits FS0 to FS11 in the control register. At the

maximum clock frequency of 10 MHz, the minimum cutoff

frequency of the filter is 2.58 Hz while the maximum program-

mable cutoff frequency is 269 Hz.

Antialias Considerations

Figure 6 shows the filter frequency response for a cutoff fre-

quency of 2.62 Hz which corresponds to a first filter notch fre-

quency of 10 Hz. This is a (sinx/x)³response (also called sinc³)

that provides >100 dB of 50 Hz and 60 Hz rejection. Program-

ming a different cutoff frequency via FS0–FS11 does not alter

the profile of the filter response; it changes the frequency of the

notches as outlined in the Control Register section.

The digital filter does not provide any rejection at integer mul-

tiples of the modulator sample frequency (n × 19.5 kHz, where

n = 1, 2, 3 . . . ). This means that there are frequency bands,

±f_{3 dB}wide (f_{3 dB}is cutoff frequency selected by FS0 to FS11)

where noise passes unattenuated to the output. However, due to

the AD7711’s high oversampling ratio, these bands occupy only

a small fraction of the spectrum and most broadband noise is

filtered. In any case, because of the high oversampling ratio a

simple, RC, single pole filter is generally sufficient to attenuate

the signals in these bands on the analog input and thus provide

adequate antialiasing filtering.

0

–20

–40

–60

–80

If passive components are placed in front of the AD7711, care

must be taken to ensure that the source impedance is low enough

so as not to introduce gain errors in the system. The dc input

impedance for the AD7711 is over 1 GΩ. The input appears as

a dynamic load which varies with the clock frequency and with

the selected gain (see Figure 7). The input sample rate, as

shown in Table III, determines the time allowed for the analog

input capacitor, C_IN, to be charged. External impedances result

in a longer charge time for this capacitor and this may result

in gain errors being introduced on the analog inputs. Table IV

shows the allowable external resistance/capacitance values such

that no gain error to the 16-bit level is introduced while Table V

shows the allowable external resistance/capacitance values such

that no gain error to the 20-bit level is introduced. Both inputs

of the differential input channel (AIN1) look into similar input

circuitry.

–100

–120

–140

–160

–180

–200

–220

–240

0

10

20

30

40

50

60

FREQUENCY – Hz

Figure 6. Frequency Response of AD7711 Filter

Since the AD7711 contains this on-chip, low-pass filtering,

there is a settling time associated with step function inputs, and

data on the output will be invalid after a step change until the

settling time has elapsed. The settling time depends upon the

notch frequency chosen for the filter. The output data rate

equates to this filter notch frequency and the settling time of the

filter to a full-scale step input is four times the output data pe-

riod. In applications using both input channels, the settling time

of the filter must be allowed to elapse before data from the

second channel is accessed.

–14–

REV. F

AD7711

allowed flow into the transducer and a measurement of the

input voltage on the AIN1 input is taken, it can indicate that the

transducer has burned out or gone open circuit. For normal

operation, this burnout current is turned off by writing a 0 to

the BO bit in the control register.

AD7711

R

INT

7k⍀ TYP

HIGH

IMPEDANCE

>1G⍀

AIN

C

INT

11.5pF TYP

RTD Excitation Current

V

BIAS

The AD7711 also contains two matched 200 µA constant cur-

rent sources which are provided at the RTD1 and RTD2 pins of

the device. These currents can be turned on/off via the control

register. Writing a 1 to the RO bit of the control register enables

these excitation currents.

SWITCHING FREQUENCY DEPENDS

ON f_CLKINAND SELECTED GAIN

2

Figure 7. Analog Input Impedance

Table IV. Typical External Series Resistance Which Will Not

Introduce 16-Bit Gain Error

For four-wire RTD applications, one of these excitation cur-

rents is used to provide the excitation current for the RTD, the

second current source can be left unconnected. For three-wire

RTD configurations, the second on-chip current source can be

used to eliminate errors due to voltage drops across lead resis-

tances. Figures 20 to 22 in the APPLICATIONS section show

some RTD configurations with the AD7711.

External Capacitance (pF)

Gain

0

50

100

500

1000

5000

1

2

4

184 kΩ 45.3 kΩ 27.1 kΩ 7.3 kΩ 4.1 kΩ 1.1 kΩ

88.6 kΩ 22.1 kΩ 13.2 kΩ 3.6 kΩ 2.0 kΩ 560 Ω

41.4 kΩ 10.6 kΩ 6.3 kΩ 1.7 kΩ 970 Ω

270 Ω

120 Ω

The temperature coefficient of the RTD current sources is

typically 20 ppm/°C with a typical matching between the tem-

perature coefficients of both current sources of 3 ppm/°C. For

applications where the absolute value of the temperature coeffi-

cient is too large, the following schemes can be used to remove

the drift error.

8–128 17.6 kΩ 4.8 kΩ 2.9 kΩ 790 Ω 440 Ω

Table V. Typical External Series Resistance Which Will Not

Introduce 20-Bit Gain Error

External Capacitance (pF)

The conversion result from the AD7711 is ratiometric to the

Gain

0

50

145 kΩ 34.5 kΩ 20.4 kΩ 5.2 kΩ 2.8 kΩ 700 Ω

70.5 kΩ 16.9 kΩ 10 kΩ 2.5 kΩ 1.4 kΩ 350 Ω

31.8 kΩ 8.0 kΩ 4.8 kΩ 1.2 kΩ 670 Ω

100

500

1000

5000

V

_REFvoltage. Therefore, if the V_REFvoltage varies with the RTD

1

2

4

temperature coefficient, the temperature drift from the current

source will be removed. For four-wire RTD applications, the

reference voltage can be made ratiometric to RTD current

source by using the second current with a low t.c. resistor to

generate the reference voltage for the part. In this case if a

12.5 kΩ resistor is used, the 200 µA current source generates

+2.5 V across the resistor. This +2.5 V can be applied to the

REF IN(+) input of the AD7711 and with the REF IN(–) input

at ground it will supply a V_REFof 2.5 V for the part. For three-

wire RTD configurations, the reference voltage for the part is

generated by placing a low t.c. resistor (12.5 kΩ for 2.5 V refer-

ence) in series with one of the constant current sources. The

RTD current sources can be driven to within 2 V of AV_DD. The

reference input of the AD7711 is differential so the REF IN(+)

and REF IN(–) of the AD7711 are driven from either side of the

resistor. Both schemes ensure that the reference voltage for the

part tracks the RTD current sources over temperature and,

thereby, removes the temperature drift error.

170 Ω

80 Ω

8–128 13.4 kΩ 3.6 kΩ 2.2 kΩ 550 Ω 300 Ω

The numbers in the above tables assume a full-scale change on

the analog input. In any case, the error introduced due to longer

charging times is a gain error which can be removed using the

system calibration capabilities of the AD7711, provided that the

resultant span is within the span limits of the system calibration

techniques for the AD7711.

ANALOG INPUT FUNCTIONS

Analog Input Ranges

Both analog inputs are programmable gain, input channels

which can handle either unipolar or bipolar input signals. The

AIN1 channel is a differential channel having a common-mode

range from V_SSto AV_DD, provided that the absolute value of the

analog input voltage lies between V_SS–30 mV and AV_DD

+30 mV. The AIN2 input channel is a single-ended input that is

referred to AGND.

Bipolar/Unipolar Inputs

The two analog inputs on the AD7711 can accept either unipo-

lar or bipolar input voltage ranges. Bipolar or unipolar options

are chosen by programming the B/U bit of the control register.

This programs both channels for either unipolar or bipolar

operation. Programming the part for either unipolar or bipolar

operation does not change any of the input signal conditioning;

it simply changes the data output coding. The data coding is

binary for unipolar inputs and offset binary for bipolar inputs.

The dc input leakage current is 10 pA maximum at 25°C

(±1 nA over temperature). This results in a dc offset voltage

developed across the source impedance. However, this dc offset

effect can be compensated for by a combination of the differen-

tial input capability of the part and its system calibration mode.

Burnout Current

The AIN1(+) input of the AD7711 contains a 4.5 µA current

source that can be turned on/off via the control register. This

current source can be used in checking that a transducer has not

burned out or gone open circuit before attempting to take mea-

surements on that channel. If the current is turned on and

The AIN1 input channel is differential and, as a result, the

voltage to which the unipolar and bipolar signals are referenced

is the voltage on the AIN1(–) input. For example, if AIN1(–) is

+1.25 V and the AD7711 is configured for unipolar operation

with a gain of 1 and a V_REFof +2.5 V, the input voltage range

REV. F

–15–

AD7711

on the AIN1(+) input is +1.25 V to +3.75 V. If AIN1(–) is

+1.25 V and the AD7711 is configured for bipolar mode with a

gain of 1 and a V_REFof +2.5 V, the analog input range on the

AIN1(+) input is –1.25 V to +3.75 V. For the AIN2 input, the

input signals are referenced to AGND.

REF OUT

REF IN(+)

REF IN(–)

AD7711

REFERENCE INPUT/OUTPUT

Figure 8. REF OUT/REF IN Connection

V_BIASInput

The V_BIASinput determine at what voltage the internal analog

circuitry is biased. It essentially provides the return path for

analog currents flowing in the modulator and, as such, it should

be driven from a low impedance point to minimize errors.

The AD7711 contains a temperature compensated +2.5 V refer-

ence which has an initial tolerance of ±1%. This reference volt-

age is provided at the REF OUT pin and it can be used as the

reference voltage for the part by connecting the REF OUT pin

to the REF IN(+) pin. This REF OUT pin is a single-ended

output, referenced to AGND, which is capable of providing up

to 1 mA to an external load. In applications where REF OUT is

connected directly to REF IN(+), REF IN(–) should be tied to

AGND to provide the nominal +2.5 V reference for the

AD7711.

For maximum internal headroom, the V_BIASvoltage should be

set halfway between AV_DDand V_SS. The difference between

AV_DDand (V_BIAS+ 0.85 × V_REF) determines the amount of

headroom the circuit has at the upper end, while the difference

between V_SSand (V_BIAS– 0.85 × V_REF) determines the amount

of headroom the circuit has at the lower end. Care should be

taken in choosing a V_BIASvoltage to ensure that it stays within

prescribed limits. For single +5 V operation, the selected V_BIAS

voltage must ensure that V_BIAS± 0.85 × V_REFdoes not exceed

AV_DDor V_SSor that the V_BIASvoltage itself is greater than V_SS

+ 2.1 V and less than AV_DD– 2.1 V. For single +10 V operation

or dual ±5 V operation, the selected V_BIASvoltage must ensure

that V_BIAS× 0.85 × V_REFdoes not exceed AV_DDor V_SSor that

the V_BIASvoltage itself is greater than V_SS+ 3 V or less than

AV_DD– 3 V. For example, with AV_DD= +4.75 V, V_SS= 0 V

and V_REF= +2.5 V, the allowable range for the V_BIASvoltage is

+2.125 V to +2.625 V. With AV_DD= +9.5 V, V_SS= 0 V and

V_REF= +5 V, the range for V_BIASis +4.25 V to +5.25 V. With

AV_DD= +4.75 V, V_SS= –4.75 V and V_REF= +2.5 V, the V_BIAS

range is –2.625 V to +2.625 V.

The reference inputs of the AD7711, REF IN(+) and

REF IN(–), provide a differential reference input capability. The

common-mode range for these differential inputs is from V_SSto

AV_DD. The nominal differential voltage, V_REF(REF IN(+) –

REF IN(–)), is +2.5 V for specified operation, but the reference

voltage can go to +5 V with no degradation in performance

provided that the absolute value of REF IN(+) and REF IN(–)

does not exceed its AV_DDand V_SSlimits and the V_BIASinput

voltage range limits are obeyed. The part is also functional with

V_REFvoltages down to 1 V but with degraded performance as

the output noise will, in terms of LSB size, be larger. REF

IN(+) must always be greater than REF IN(–) for correct opera-

tion of the AD7711.

Both reference inputs provide a high impedance, dynamic load

similar to the analog inputs. The maximum dc input leakage

current is 10 pA (±1 nA over temperature) and source resis-

tance may result in gain errors on the part. The reference inputs

look like the analog input (see Figure 7). In this case, R_INTis

5 kΩ typ and C_INTvaries with gain. The input sample rate is

f_{CLK IN}/256 and does not vary with gain. For gains of 1 to 8 C_INT

is 20 pF; for a gain of 16 it is 10 pF; for a gain of 32 it is 5 pF;

for a gain of 64 it is 2.5 pF; and for a gain of 128 it is 1.25 pF.

The V_BIASvoltage does have an effect on the AV_DDpower sup-

ply rejection performance of the AD7711. If the V_BIASvoltage

tracks the AV_DDsupply, it improves the power supply rejection

from the AV_DDsupply line from 80 dB to 95 dB. Using an

external Zener diode, connected between the AV_DDline and

V_BIAS, as the source for the V_BIASvoltage gives the improvement

in AV_DDpower supply rejection performance.

The digital filter of the AD7711 removes noise from the refer-

ence input just as it does with the analog input, and the same

limitations apply regarding lack of noise rejection at integer

multiples of the sampling frequency. The output noise perfor-

mance outlined in Tables I and II assumes a clean reference. If

the reference noise in the bandwidth of interest is excessive, it

can degrade the performance of the AD7711. Using the on-chip

reference as the reference source for the part (i.e., connecting

REF OUT to REF IN) results in somewhat degraded output

noise performance from the AD7711 for portions of the noise

table that are dominated by the device noise. The on-chip

reference noise effect is eliminated in ratiometric applications

where the reference is used to provide the excitation voltage for

the analog front end. The connection shown in Figure 8 is rec-

ommended when using the on-chip reference. Recommended

reference voltage sources for the AD7711 include the AD580

and AD680 2.5 V references.

USING THE AD7711

SYSTEM DESIGN CONSIDERATIONS

The AD7711 operates differently from successive approxima-

tion ADCs or integrating ADCs. Since it samples the signal

continuously, like a tracking ADC, there is no need for a start

convert command. The output register is updated at a rate

determined by the first notch of the filter and the output can be

read at any time, either synchronously or asynchronously.

Clocking

The AD7711 requires a master clock input, which may be an

external TTL/CMOS compatible clock signal applied to the

MCLK IN pin with the MCLK OUT pin left unconnected.

Alternatively, a crystal of the correct frequency can be con-

nected between MCLK IN and MCLK OUT, in which case the

clock circuit will function as a crystal controlled oscillator. For

lower clock frequencies, a ceramic resonator may be used

instead of the crystal. For these lower frequency oscillators,

external capacitors may be required on either the ceramic reso-

nator or on the crystal.

–16–

REV. F

AD7711

The AD7711 offers self-calibration, system calibration and

background calibration facilities. For calibration to occur on the

selected channel, the on-chip microcontroller must record the

modulator output for two different input conditions. These are

“zero-scale” and “full-scale” points. With these readings, the

microcontroller can calculate the gain slope for the input to

output transfer function of the converter. Internally, the part

works with a resolution of 33 bits to determine its conversion

result of either 16 bits or 24 bits.

The input sampling frequency, the modulator sampling fre-

quency, the –3 dB frequency, output update rate and calibration

time are all directly related to the master clock frequency,

f_{CLK IN.}Reducing the master clock frequency by a factor of two

will halve the above frequencies and update rate and will double

the calibration time.

The current drawn from the DV_DDpower supply is also directly

related to f_{CLK IN}. Reducing f_{CLK IN}by a factor of two will halve

the DV_DDcurrent but will not affect the current drawn from the

AV_DDpower supply.

2

The AD7711 also provides the facility to write to the on-chip

calibration registers and in this manner the span and offset for

the part can be adjusted by the user. The offset calibration regis-

ter contains a value which is subtracted from all conversion

results, while the full-scale calibration register contains a value

which is multiplied by all conversion results. The offset calibra-

tion coefficient is subtracted from the result prior to the multi-

plication by the full-scale coefficient. In the first three modes

outlined here, the DRDY line indicates that calibration is com-

plete by going low. If DRDY is low before (or goes low during)

the calibration command, it may take up to one modulator cycle

before DRDY goes high to indicate that calibration is in

progress. Therefore, DRDY should be ignored for up to one

modulator cycle after the last bit of the calibration command is

written to the control register.

System Synchronization

If multiple AD7711s are operated from a common master clock,

they can be synchronized to update their output registers simul-

taneously. A falling edge on the SYNC input resets the filter and

places the AD7711 into a consistent, known state. A common

signal to the AD7711s’ SYNC inputs will synchronize their

operation. This would normally be done after each AD7711 has

performed its own calibration or has had calibration coefficients

loaded to it.

The SYNC input can also be used to reset the digital filter in

systems where the turn-on time of the digital power supply

(DV_DD) is very long. In such cases, the AD7711 will start oper-

ating internally before the DV_DDline has reached its minimum

operating level, +4.75 V. With a low DV_DDvoltage, the

AD7711’s internal digital filter logic does not operate correctly.

Thus, the AD7711 may have clocked itself into an incorrect

operating condition by the time that DV_DDhas reached its cor-

rect level. The digital filter will be reset upon issue of a calibra-

tion command (whether it is self-calibration, system calibration

or background calibration) to the AD7711. This ensures correct

operation of the AD7711. In systems where the power-on

default conditions of the AD7711 are acceptable, and no cali-

bration is performed after power-on, issuing a SYNC pulse to

the AD7711 will reset the AD7711’s digital filter logic. An R, C

on the SYNC line, with R, C time constant longer than the

DV_DDpower-on time, will perform the SYNC function.

Self-Calibration

In the self-calibration mode with a unipolar input range, the

zero-scale point used in determining the calibration coefficients

is with both inputs shorted (i.e., AIN1(+) = AIN1(–) = V_BIAS

for AIN1 and AIN2 = V_BIASfor AIN2) and the full-scale point is

V_REF. The zero-scale coefficient is determined by converting an

internal shorted inputs node. The full-scale coefficient is deter-

mined from the span between this shorted inputs conversion

and a conversion on an internal V_REFnode. The self-calibration

mode is invoked by writing the appropriate values (0, 0, 1) to

the MD2, MD1 and MD0 bits of the control register. In this

calibration mode, the shorted inputs node is switched into the

modulator first and a conversion is performed; the V_REFnode is

then switched in and another conversion is performed. When

the calibration sequence is complete, the calibration coefficients

updated and the filter resettled to the analog input voltage, the

DRDY output goes low. The self-calibration procedure takes

into account the selected gain on the PGA.

ACCURACY

Sigma-delta ADCs, like VFCs and other integrating ADCs, do

not contain any source of nonmonotonicity and inherently offer

no missing codes performance. The AD7711 achieves excellent

linearity by the use of high quality, on-chip silicon dioxide

capacitors, which have a very low capacitance/voltage coeffi-

cient. The device also achieves low input drift through the use

of chopper stabilized techniques in its input stage. To ensure

excellent performance over time and temperature, the AD7711

uses digital calibration techniques which minimize offset and

gain error.

For bipolar input ranges in the self-calibrating mode, the se-

quence is very similar to that just outlined. In this case, the two

points which the AD7711 calibrates are midscale (bipolar zero)

and positive full scale.

System Calibration

System calibration allows the AD7711 to compensate for

system gain and offset errors as well as its own internal errors.

System calibration performs the same slope factor calculations

as self-calibration but uses voltage values presented by the sys-

tem to the AIN inputs for the zero and full-scale points. System

calibration is a two-step process. The zero-scale point must be

presented to the converter first. It must be applied to the con-

verter before the calibration step is initiated and must remain

stable until the step is complete. System calibration is initiated

by writing the appropriate values (0, 1, 0) to the MD2, MD1

and MD0 bits of the control register. The DRDY output from

the device will signal when the step is complete by going low.

AUTOCALIBRATION

Autocalibration on the AD7711 removes offset and gain errors

from the device. A calibration routine should be initiated on the

device whenever there is a change in the ambient operating

temperature or supply voltage. It should also be initiated if there

is a change in the selected gain, filter notch or bipolar/unipolar

input range. However, if the AD7711 is in its background cali-

bration mode, the above changes are all automatically taken care

of (after the settling time of the filter has been allowed for).

REV. F

–17–

AD7711

After the zero-scale point is calibrated, the full-scale point is

applied and the second step of the calibration process is initiated

by again writing the appropriate values (0, 1, 1) to MD2, MD1

and MD0. Again the full-scale voltage must be set up before the

calibration is initiated and it must remain stable throughout the

calibration step. DRDY goes low at the end of this second step

to indicate that the system calibration is complete. In the unipo-

lar mode, the system calibration is performed between the two

endpoints of the transfer function; in the bipolar mode, it is

performed between midscale and positive full scale.

the self-calibration mode, i.e., shorted inputs and V_REF. The

background calibration mode is invoked by writing 1, 0, 1 to

MD2, MD1, MD0 of the control register. When invoked, the

background calibration mode reduces the output data rate of the

AD7711 by a factor of six while the –3 dB bandwidth remains

unchanged. Its advantage is that the part is continually perform-

ing calibration and automatically updating its calibration coeffi-

cients. As a result, the effects of temperature drift, supply

sensitivity and time drift on zero and full-scale errors are auto-

matically removed. When the background calibration mode is

turned on, the part will remain in this mode until bits MD2,

MD1 and MD0 of the control register are changed. With back-

ground calibration mode on, the first result from the AD7711

will be incorrect as the full-scale calibration will not have been

performed. For a step change on the input, the second output

update will have settled to 100% of the final value.

This two-step system calibration mode offers another feature.

After the sequence has been completed, additional offset or gain

calibrations can be performed by themselves to adjust the zero

reference point or the system gain. This is achieved by perform-

ing the first step of the system calibration sequence (by writing

0, 1, 0 to MD2, MD1, MD0). This will adjust the zero-scale or

offset point but will not change the slope factor from what was

set during a full system calibration sequence.

Table VI summarizes the calibration modes and the calibration

points associated with them. It also gives the duration from

when the calibration is invoked to when valid data is available to

the user.

System calibration can also be used to remove any errors from

an antialiasing filter on the analog input. A simple R, C anti-

aliasing filter on the front end may introduce a gain error on the

analog input voltage but the system calibration can be used to

remove this error.

Span and Offset Limits

Whenever a system calibration mode is used, there are limits on

the amount of offset and span that can be accommodated. The

range of input span in both the unipolar and bipolar modes has

a minimum value of 0.8 × V_REF/GAIN and a maximum value of

2.1 × V_REF/GAIN.

System Offset Calibration

System offset calibration is a variation of both the system cali-

bration and self-calibration. In this case, the zero-scale point

for the system is presented to the AIN input of the converter.

System-offset calibration is initiated by writing 1, 0, 0 to MD2,

MD1, MD0. The system zero-scale coefficient is determined by

converting the voltage applied to the AIN input, while the full-

scale coefficient is determined from the span between this AIN

conversion and a conversion on V_REF. The zero-scale point

should be applied to the AIN input for the duration of the cali-

bration sequence. This is a one-step calibration sequence with

DRDY going low when the sequence is completed. In the uni-

polar mode, the system offset calibration is performed between

the two end points of the transfer function; in the bipolar mode,

it is performed between midscale and positive full scale.

The amount of offset which can be accommodated depends on

whether the unipolar or bipolar mode is being used. This offset

range is limited by the requirement that the positive full-scale

calibration limit is ≤ 1.05 × V_REF/GAIN. Therefore, the offset

range plus the span range cannot exceed 1.05 × V_REF/GAIN. If

the span is at its minimum (0.8 × V_REF/GAIN) the maximum

the offset can be is (0.25 × V_REF/GAIN).

In the bipolar mode, the system offset calibration range is again

restricted by the span range. The span range of the converter in

bipolar mode is equidistant around the voltage used for the

zero-scale point thus the offset range plus half the span range

cannot exceed (1.05 × V_REF/GAIN). If the span is set to 2 × V_REF

/

Background Calibration

GAIN, the offset span cannot move more than ±(0.05 × V_REF

/

The AD7711 also offers a background calibration mode where

the part interleaves its calibration procedure with its normal

conversion sequence. In the background calibration mode, the

same voltages are used as the calibration points as are used in

GAIN) before the endpoints of the transfer function exceed the

input overrange limits ±(1.05 × V_REF/GAIN). If the span range

is set to the minimum ±(0.4 × V_REF/GAIN) the maximum al-

lowable offset range is ±(0.65 × V_REF/GAIN).

Table VI. Calibration Truth Table

Cal Type

MD2, MD1, MD0

Zero-Scale Cal

Shorted Inputs

AIN

–

AIN

Full-Scale Cal

V_REF

–

AIN

V_REF

Sequence

Duration

Self-Cal

System Cal

System Offset Cal

Background Cal

0, 0, 1

0, 1, 0

0, 1, 1

1, 0, 0

1, 0, 1

One Step

Two Step

One Step

9 × 1/Output Rate

4 × 1/Output Rate

9 × 1/Output Rate

6 × 1/Output Rate

Shorted Inputs

–18–

REV. F

AD7711

POWER-UP AND CALIBRATION

The analog and digital supplies to the AD7711 are independent

and separately pinned out to minimize coupling between the

analog and digital sections of the device. The digital filter will

provide rejection of broadband noise on the power supplies,

except at integer multiples of the modulator sampling frequency.

The digital supply (DV_DD) must not exceed the analog positive

supply (AV_DD) by more than 0.3 V in normal operation. If sepa-

rate analog and digital supplies are used, the recommended

decoupling scheme is shown in Figure 9. In systems where

AV_DD= +5 V and DV_DD= +5 V, it is recommended that AV_DD

and DV_DDare driven from the same +5 V supply, although each

supply should be decoupled separately as shown in Figure 9. It

is preferable that the common supply is the system’s analog +5 V

supply.

On power-up, the AD7711 performs an internal reset which sets

the contents of the control register to a known state. However,

to ensure correct calibration for the device a calibration routine

should be performed after power-up.

The power dissipation and temperature drift of the AD7711 are

low and no warm up time is required before the initial calibra-

tion is performed. However, if an external reference is being

used, this reference must have stabilized before calibration is

initiated.

2

Drift Considerations

The AD7711 uses chopper stabilization techniques to minimize

input offset drift. Charge injection in the analog switches and dc

leakage currents at the sampling node are the primary sources of

offset voltage drift in the converter. The dc input leakage cur-

rent is essentially independent of the selected gain. Gain drift

within the converter depends primarily upon the temperature

tracking of the internal capacitors. It is not affected by leakage

currents.

It is also important that power is applied to the AD7711 before

signals at REF IN, AIN or the logic input pins in order to avoid

latch-up. If separate supplies are used for the AD7711 and the

system digital circuitry, then the AD7711 should be powered up

first. If it is not possible to guarantee this, then current limiting

resistors should be placed in series with the logic inputs.

Measurement errors due to offset drift or gain drift can be elimi-

nated at any time by recalibrating the converter or by operating

the part in the background calibration mode. Using the system

calibration mode can also minimize offset and gain errors in the

signal conditioning circuitry. Integral and differential linearity

errors are not significantly affected by temperature changes.

ANALOG

SUPPLY

DIGITAL +5V

SUPPLY

10␮F

0.1␮F

AV

DV

DD

AD7711

POWER SUPPLIES AND GROUNDING

Since the analog inputs and reference input are differential,

most of the voltages in the analog modulator are common-mode

voltages. V_BIASprovides the return path for most of the analog

currents flowing in the analog modulator. As a result, the V_BIAS

input should be driven from a low impedance to minimize errors

due to charging/discharging impedances on this line. When the

internal reference is used as the reference source for the part,

AGND is the ground return for this reference voltage.

Figure 9. Recommended Decoupling Scheme

REV. F

–19–

AD7711

the output data register. It is reset high when the last bit of data

(either 16th bit or 24th bit) is read from the output register. If

data is not read from the output register, the DRDY line will

remain low. The output register will continue to be updated at

the output update rate but DRDY will not indicate this. A read

from the device in this circumstance will access the most recent

word in the output register. If a new data word becomes avail-

able to the output register while data is being read from the

output register, DRDY will not indicate this and the new data

word will be lost to the user. DRDY is not affected by reading

from the control register or the calibration registers.

DIGITAL INTERFACE

The AD7711’s serial communications port provides a flexible

arrangement to allow easy interfacing to industry-standard

microprocessors, microcontrollers and digital signal processors.

A serial read to the AD7711 can access data from the output

register, the control register or from the calibration registers. A

serial write to the AD7711 can write data to the control register

or the calibration registers.

Two different modes of operation are available, optimized for

different types of interface where the AD7711 can act either as

master in the system (it provides the serial clock) or as slave (an

external serial clock can be provided to the AD7711). These

two modes, labelled self-clocking mode and external clocking

mode, are discussed in detail in the following sections.

Data can only be accessed from the output data register when

DRDY is low. If RFS goes low with DRDY high, no data trans-

fer will take place. DRDY does not have any effect on reading

data from the control register or from the calibration registers.

Self-Clocking Mode

Figure 10 shows a timing diagram for reading from the AD7711

in the self-clocking mode. The read operation shows a read from

the AD7711’s output data register. A read from the control

register or calibration registers is similar but in these cases the

DRDY line is not related to the read function. Depending on

the output update rate, it can go low at any stage in the control/

calibration register read cycle without affecting the read and its

status should be ignored. A read operation from either the con-

trol or calibration registers must always read 24 bits of data

from the respective register.

The AD7711 is configured for its self-clocking mode by tying

the MODE pin high. In this mode, the AD7711 provides the

serial clock signal used for the transfer of data to and from the

AD7711. This self-clocking mode can be used with processors

that allow an external device to clock their serial port including

most digital signal processors and microcontrollers such as the

68HC11 and 68HC05. It also allows easy interfacing to serial-

parallel conversion circuits in systems with parallel data commu-

nication, allowing interfacing to 74XX299 universal shift

registers without any additional decoding. In the case of shift

registers, the serial clock line should have a pull-down resistor

instead of the pull-up resistor shown in Figures 10 and 11.

Figure 10 shows a read operation from the AD7711. For the

timing diagram shown, it is assumed that there is a pull-up

resistor on the SCLK output. With DRDY low, the RFS

input is brought low. RFS going low enables the serial clock of

the AD7711 and also places the MSB of the word on the serial

data line. All subsequent data bits are clocked out on a high to

low transition of the serial clock and are valid prior to the fol-

lowing rising edge of this clock. The final active falling edge of

SCLK clocks out the LSB and this LSB is valid prior to the final

active rising edge of SCLK. Coincident with the next falling

edge of SCLK, DRDY is reset high. DRDY going high turns off

the SCLK and the SDATA outputs. This means that the data

hold time for the LSB is slightly shorter than for all other bits.

Read Operation

Data can be read from either the output register, the control

register or the calibration registers. A0 determines whether the

data read accesses data from the control register or from the

output/calibration registers. This A0 signal must remain valid

for the duration of the serial read operation. With A0 high, data

is accessed from either the output register or from the calibra-

tion registers. With A0 low, data is accessed from the control

register.

The function of the DRDY line is dependent only on the output

update rate of the device and the reading of the output data

register. DRDY goes low when a new data word is available in

DRDY (O)

t₃

t₂

A0 (I)

t₅

t₄

RFS (I)

t₉

t₆

SCLK (O)

t₇

t₈

t₁₀

THREE-STATE

LSB

SDATA (O)

MSB

Figure 10. Self-Clocking Mode, Output Data Read Operation

–20–

REV. F

AD7711

Write Operation

Read Operation

Data can be written to either the control register or calibration

registers. In either case, the write operation is not affected by

the DRDY line and the write operation does not have any effect

on the status of DRDY. A write operation to the control register

or the calibration register must always write 24 bits to the re-

spective register.

As with the self-clocking mode, data can be read from either the

output register, the control register or the calibration registers.

A0 determines whether the data read accesses data from the

control register or from the output/calibration registers. This A0

signal must remain valid for the duration of the serial read

operation. With A0 high, data is accessed from either the output

register or from the calibration registers. With A0 low, data is

accessed from the control register.

Figure 11 shows a write operation to the AD7711. A0 deter-

mines whether a write operation transfers data to the control

register or to the calibration registers. This A0 signal must

remain valid for the duration of the serial write operation. The

falling edge of TFS enables the internally generated SCLK

output. The serial data to be loaded to the AD7711 must be

valid on the rising edge of this SCLK signal. Data is clocked

into the AD7711 on the rising edge of the SCLK signal with the

MSB transferred first. On the last active high time of SCLK, the

LSB is loaded to the AD7711. Subsequent to the next falling

edge of SCLK, the SCLK output is turned off. (The timing

diagram of Figure 11 assumes a pull-up resistor on the SCLK

line.)