AD7854BR-REEL [ADI]
IC 1-CH 12-BIT SUCCESSIVE APPROXIMATION ADC, PARALLEL ACCESS, PDSO28, SOIC-28, Analog to Digital Converter;
| 型号： | AD7854BR-REEL |
| 厂家： | 
ADI |
| 描述： | IC 1-CH 12-BIT SUCCESSIVE APPROXIMATION ADC, PARALLEL ACCESS, PDSO28, SOIC-28, Analog to Digital Converter |
| 文件： | 总28页 (文件大小：264K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
3 V to 5 V Single Supply, 200 kSPS
12-Bit Sampling ADCs
a
AD7854/AD7854L*
FUNCTIONAL BLOCK DIAGRAM
FEATURES
AV
DD
AGND
Specified for VDD of 3 V to 5.5 V
Read-Only Operation
AD7854–200 kSPS; AD7854L–100 kSPS
System and Self-Calibration
Low Power
AIN(+)
AIN(–)
AD7854/AD7854L
T/H
DV
DD
2.5V
REFERENCE
Normal Operation
AD7854: 15 mW (VDD = 3 V)
AD7854L: 5.5 mW (VDD = 3 V)
Automatic Power-Down After Conversion (25 W)
AD7854: 1.3 mW 10 kSPS
AD7854L: 650 W 10 kSPS
Flexible Parallel Interface
COMP
REF
IN
/
BUF
REF
OUT
DGND
CLKIN
C
REF1
CHARGE
REDISTRIBUTION
DAC
12-Bit Parallel/8-Bit Parallel (AD7854)
28-Lead DIP, SOIC and SSOP Packages (AD7854)
SAR + ADC
CONTROL
CONVST
C
REF2
CALIBRATION
MEMORY
BUSY
APPLICATIONS
AND CONTROLLER
Battery-Powered Systems (Personal Digital Assistants,
Medical Instruments, Mobile Communications)
Pen Computers
PARALLEL INTERFACE/CONTROL REGISTER
Instrumentation and Control Systems
High Speed Modems
HBEN
CS
WR
DB11–DB0
RD
GENERAL DESCRIPTION
PRODUCT HIGHLIGHTS
1. Operation with either 3 V or 5 V power supplies.
The AD7854/AD7854L is a high speed, low power, 12-bit ADC
that operates from a single 3 V or 5 V power supply, the
AD7854 being optimized for speed and the AD7854L for low
power. The ADC powers up with a set of default conditions at
which time it can be operated as a read-only ADC. The ADC
contains self-calibration and system calibration options to en-
sure accurate operation over time and temperature and has a
number of power-down options for low power applications.
2. Flexible power management options including automatic
power-down after conversion. By using the power manage-
ment options a superior power performance at slower
throughput rates can be achieved:
AD7854: 1 mW typ @ 10 kSPS
AD7854L: 1 mW typ @ 20 kSPS
3. Operates with reference voltages from 1.2 V to AVDD
4. Analog input ranges from 0 V to AVDD
.
The AD7854 is capable of 200 kHz throughput rate while the
AD7854L is capable of 100 kHz throughput rate. The input
track-and-hold acquires a signal in 500 ns and features a pseudo-
differential sampling scheme. The AD7854 and AD7854L input
voltage range is 0 to VREF (unipolar) and –VREF/2 to +VREF/2,
centered at VREF/2 (bipolar). The coding is straight binary in
unipolar mode and twos complement in bipolar mode. Input
signal range is to the supply and the part is capable of convert-
ing full-power signals to 100 kHz.
.
5. Self-calibration and system calibration.
6. Versatile parallel I/O port.
7. Lower power version AD7854L.
CMOS construction ensures low power dissipation of typically
5.4 mW for normal operation and 3.6 µW in power-down mode.
The part is available in 28-lead, 0.6 inch wide dual-in-line pack-
age (DIP), 28-lead small outline (SOIC) and 28-lead small
shrink outline (SSOP) packages.
*Patent pending.
See Page 27 for data sheet index.
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
Fax: 781/326-8703
World Wide Web Site: http://www.analog.com
© Analog Devices, Inc., 2000
AD7854/AD7854L–SPECIFICATIONS1, 2
(AVDD = DVDD = +3.0 V to +5.5 V, REFIN/REFOUT = 2.5 V
External Reference, fCLKIN = 4 MHz (for L Version: 1.8 MHz (0؇C to +70؇C) and 1 MHz (–40؇C to +85؇C)); fSAMPLE = 200 kHz (AD7854), 100 kHz
(AD7854L); TA = TMIN to TMAX, unless otherwise noted.) Specifications in () apply to the AD7854L.
Parameter
A Version1
B Version1
S Version1 Units
Test Conditions/Comments
DYNAMIC PERFORMANCE
Signal to Noise + Distortion Ratio3
(SNR)
70
71
70
dB min
Typically SNR is 72 dB
VIN = 10 kHz Sine Wave, fSAMPLE = 200 kHz
(L Version: fSAMPLE = 100 kHz @ fCLKIN = 2 MHz)
Total Harmonic Distortion (THD)
Peak Harmonic or Spurious Noise
–78
–78
–78
–78
–78
–78
dB max
dB max
VIN = 10 kHz Sine Wave, fSAMPLE = 200 kHz
(L Version: fSAMPLE = 100 kHz @ fCLKIN = 2 MHz)
VIN = 10 kHz Sine Wave, fSAMPLE = 200 kHz
(L Version: fSAMPLE = 100 kHz @ fCLKIN = 2 MHz)
Intermodulation Distortion (IMD)
Second Order Terms
–78
–78
–78
–78
–78
–78
dB typ
dB typ
fa = 9.983 kHz, fb = 10.05 kHz, fSAMPLE = 200 kHz
(L Version: fSAMPLE = 100 kHz @ fCLKIN = 2 MHz)
fa = 9.983 kHz, fb = 10.05 kHz, fSAMPLE = 200 kHz
(L Version: fSAMPLE = 100 kHz @ fCLKIN = 2 MHz)
Third Order Terms
DC ACCURACY
Resolution
Integral Nonlinearity
Differential Nonlinearity
Unipolar Offset Error
12
1
1
3
2
4
2
4
2
4
2
4
12
0.5
1
3
2
4
2
4
2
12
1
1
4
2
4
2
5
2
5
2
5
Bits
LSB max
LSB max
LSB max
LSB typ
LSB max
LSB typ
LSB max
LSB typ
LSB max
LSB typ
LSB max
5 V Reference VDD = 5 V
Guaranteed No Missed Codes to 12 Bits
Unipolar Gain Error
Bipolar Positive Full-Scale Error
Negative Full-Scale Error
Bipolar Zero Error
4
2
4
ANALOG INPUT
Input Voltage Ranges
0 to VREF
0 to VREF
VREF/2
0 to VREF
VREF/2
Volts
Volts
i.e., AIN(+) – AIN(–) = 0 to VREF, AIN(–) can be
biased up but AIN(+) cannot go below AIN(–).
i.e., AIN(+) – AIN(–) = –VREF/2 to +VREF/2, AIN(–)
should be biased to +VREF/2 and AIN(+) can go below
AIN(–) but cannot go below 0 V.
V
REF/2
Leakage Current
Input Capacitance
1
20
1
20
1
20
µA max
pF typ
REFERENCE INPUT/OUTPUT
REFIN Input Voltage Range
Input Impedance
2.3/VDD
150
2.3/VDD
150
2.3/VDD
150
V min/max
kΩ typ
Functional from 1.2 V
REFOUT Output Voltage
REFOUT Tempco
2.3/2.75
20
2.3/2.7
20
2.3/2.7
20
V min/max
ppm/°C typ
LOGIC INPUTS
Input High Voltage, VINH
3
3
3
V min
V min
V max
V max
µA max
pF max
AVDD = DVDD = 4.5 V to 5.5 V
AVDD = DVDD = 3.0 V to 3.6 V
AVDD = DVDD = 4.5 V to 5.5 V
AVDD = DVDD = 3.0 V to 3.6 V
Typically 10 nA, VIN = 0 V or VDD
2.1
0.4
0.6
10
10
2.1
0.4
0.6
10
10
2.1
0.4
0.6
10
10
Input Low Voltage, VINL
Input Current, IIN
4
Input Capacitance, CIN
LOGIC OUTPUTS
Output High Voltage, VOH
ISOURCE = 200 µA
4
2.4
4
2.4
4
2.4
V min
V min
AVDD = DVDD = 4.5 V to 5.5 V
AVDD = DVDD = 3.0 V to 3.6 V
Output Low Voltage, VOL
Floating-State Leakage Current
Floating-State Output Capacitance4
Output Coding
0.4
10
10
0.4
10
10
0.4
10
10
V max
µA max
pF max
ISINK = 0.8 mA
Straight (Natural) Binary
Twos Complement
Unipolar Input Range
Bipolar Input Range
CONVERSION RATE
Conversion Time
Track/Hold Acquisition Time
tCLKIN × 18
4.6 (10)
0.5 (1)
4.6 (9)
0.5 (1)
4.6 (9)
0.5 (1)
µs max
µs min
(L Versions Only, 0°C to +70°C, 1.8 MHz CLKIN)
(L Versions Only, –40°C to +85°C, 1 MHz CLKIN)
–2–
REV. B
AD7854/AD7854L
Parameter
A Version1
B Version1
S Version1 Units
Test Conditions/Comments
POWER REQUIREMENTS
AVDD, DVDD
+3.0/+5.5
5.5 (1.8)
5.5 (1.8)
+3.0/+5.5
5.5 (1.8)
5.5 (1.8)
+3.0/+5.5
6 (1.8)
V min/max
IDD
Normal Mode5
mA max
mA max
AVDD = DVDD = 4.5 V to 5.5 V. Typically 4.5 mA
(1.5 mA);
6 (1.8)
AVDD = DVDD = 3.0 V to 3.6 V. Typically 4.0 mA
(1.5 mA).
Sleep Mode6
With External Clock On
10
400
5
10
400
5
10
400
5
µA typ
µA typ
µA max
Full power-down. Power management bits in control
register set as PMGT1 = 1, PMGT0 = 0.
Partial power-down. Power management bits in
control register set as PMGT1 = 1, PMGT0 = 1.
Typically 1 µA. Full power-down. Power management
bits in control register set as PMGT1 = 1,
PMGT0 = 0.
With External Clock Off
200
200
200
µA typ
Partial power-down. Power management bits in
control register set as PMGT1 = 1, PMGT0 = 1.
Normal Mode Power Dissipation
30 (10)
20 (6.5)
30 (10)
20 (6.5)
30 (10)
20 (6.5)
mW max
mW max
V
V
DD = 5.5 V: Typically 25 mW (8)
DD = 3.6 V: Typically 15 mW (5.4)
Sleep Mode Power Dissipation
With External Clock On
55
36
27.5
18
55
36
27.5
18
55
36
27.5
18
µW typ
µW typ
µW max
µW max
V
V
V
DD = 5.5 V
DD = 3.6 V
DD = 5.5 V: Typically 5.5 µW
With External Clock Off
VDD = 3.6 V: Typically 3.6 µW
SYSTEM CALIBRATION
Offset Calibration Span7
Gain Calibration Span7
+0.05 × VREF/–0.05 × VREF
+0.025 × VREF/–0.025 × VREF
V max/min Allowable Offset Voltage Span for Calibration
V max/min Allowable Full-Scale Voltage Span for Calibration
NOTES
1Temperature ranges as follows: A, B Versions, –40°C to +85°C; S Version, –55°C to +125°C.
2Specifications apply after calibration.
3Not production tested. Guaranteed by characterization at initial product release.
4Sample tested @ +25°C to ensure compliance.
5All digital inputs @ DGND except for CONVST @ DVDD. No load on the digital outputs. Analog inputs @ AGND.
6CLKIN @ DGND when external clock off. All digital inputs @ DGND except for CONVST @ DVDD. No load on the digital outputs. Analog inputs @ AGND.
7The offset and gain calibration spans are defined as the range of offset and gain errors that the AD7854/AD7854L can calibrate. Note also that these are voltage spans
and are not absolute voltages (i.e., the allowable system offset voltage presented at AIN(+) for the system offset error to be adjusted out will be AIN(–) 0.05 × VREF
and the allowable system full-scale voltage applied between AIN(+) and AIN(–) for the system full-scale voltage error to be adjusted out will be VREF 0.025 × VREF
(unipolar mode) and VREF/2 0.025 × VREF (bipolar mode)). This is explained in more detail in the calibration section of the data sheet.
,
Specifications subject to change without notice.
REV. B
–3–
AD7854/AD7854L
(AVDD = DVDD = +3.0 V to +5.5 V; fCLKIN = 4 MHz for AD7854 and 1.8 MHz for AD7854L;
TA = TMIN to TMAX, unless otherwise noted)
TIMING SPECIFICATIONS1
Limit at TMIN, TMAX
(A, B, S Versions)
3 V
Parameter 5 V
Units
Description
2
fCLKIN
500
4
500
4
kHz min
MHz max
MHz max
ns min
ns max
µs max
µs max
ns min
ns min
ns min
ns min
ns min
ns max
ns min
ns max
ns min
ns min
ns max
ns min
ns max
ns min
ns min
ns min
ns min
ns min
ns min
ns min
ns min
ns max
ms typ
Master Clock Frequency
1.8
100
50
4.5
10
15
5
1.8
100
90
4.5
10
15
5
L Version
CONVST Pulsewidth
3
t1
t2
CONVST to BUSY ↑ Propagation Delay
Conversion Time = 18 tCLKIN
L Version 1.8 MHz CLKIN. Conversion Time = 18 tCLKIN
HBEN to RD Setup Time
HBEN to RD Hold Time
CS to RD to Setup Time
CS to RD Hold Time
RD Pulsewidth
Data Access Time After RD
Bus Relinquish Time After RD
tCONVERT
t3
t4
t5
t6
t74
t85
t9
0
0
0
0
55
50
5
40
60
0
5
0
0
55
10
5
1/2 tCLKIN
50
50
40
40
2.5 tCLKIN
70
50
5
40
70
0
5
0
0
70
10
5
1/2 tCLKIN
70
70
60
60
2.5 tCLKIN
t10
t11
t12
t13
t14
t15
t16
t17
Minimum Time Between Reads
HBEN to WR Setup Time
HBEN to WR Hold Time
CS to WR Setup Time
CS to WR Hold Time
WR Pulsewidth
Data Setup Time Before WR
Data Hold Time After WR
New Data Valid Before Falling Edge of BUSY
HBEN High Pulse Duration
4
t18
t19
t20
t21
t22
t23
HBEN Low Pulse Duration
Propagation Delay from HBEN Rising Edge to Data Valid
Propagation Delay from HBEN Falling Edge to Data Valid
CS↑ to BUSY ↑ in Calibration Sequence
Full Self-Calibration Time, Master Clock Dependent (125013
6
tCAL
31.25
31.25
tCLKIN
Internal DAC Plus System Full-Scale Cal Time, Master Clock
Dependent (111124 tCLKIN
System Offset Calibration Time, Master Clock Dependent
(13889 tCLKIN
)
6
tCAL1
27.78
3.47
27.78
3.47
ms typ
ms typ
)
6
tCAL2
)
NOTES
1Sample tested at +25°C to ensure compliance. All input signals are specified with tr = tf = 5 ns (10% to 90% of VDD) and timed from a voltage level of 1.6 V.
2Mark/Space ratio for the master clock input is 40/60 to 60/40.
3The CONVST pulsewidth here only applies for normal operation. When the part is in power-down mode, a different CONVST pulsewidth applies (see Power-Down
section).
4Measured with the load circuit of Figure 1 and defined as the time required for the output to cross 0.8 V or 2.4 V.
5t9 is derived form the measured time taken by the data outputs to change 0.5 V when loaded with the circuit of Figure 1. The measured number is then extrapolated
back to remove the effects of charging or discharging the 50 pF capacitor. This means that the time, t9, quoted in the timing characteristics is the true bus relinquish
time of the part and is independent of the bus loading.
6The typical time specified for the calibration times is for a master clock of 4 MHz. For the L version the calibration times will be longer than those quoted here due to
the 1.8 MHz master clock.
Specifications subject to change without notice.
REV. B
–4–
AD7854/AD7854L
ABSOLUTE MAXIMUM RATINGS1
(TA = +25°C unless otherwise noted)
I
1.6mA
OL
AVDD to AGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V
DVDD to DGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V
AVDD to DVDD . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +0.3 V
Analog Input Voltage to AGND . . . . –0.3 V to AVDD + 0.3 V
Digital Input Voltage to DGND . . . . –0.3 V to DVDD + 0.3 V
Digital Output Voltage to DGND . . . –0.3 V to DVDD + 0.3 V
REFIN/REFOUT to AGND . . . . . . . . . –0.3 V to AVDD + 0.3 V
Input Current to Any Pin Except Supplies2 . . . . . . . . . 10 mA
Operating Temperature Range
TO
OUTPUT
PIN
+2.1V
C
50pF
L
I
200µA
OH
Figure 1. Load Circuit for Digital Output Timing
Specifications
Commercial (A, B Versions) . . . . . . . . . . . –40°C to +85°C
Commercial (S Version) . . . . . . . . . . . . . . –55°C to +125°C
Storage Temperature Range . . . . . . . . . . . –65°C to +150°C
Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . +150°C
Cerdip Package, Power Dissipation . . . . . . . . . . . . . . 450 mW
PIN CONFIGURATION
FOR DIP, SOIC AND SSOP
1
28
27
26
25
24
23
22
21
20
19
18
17
16
15
CONVST
WR
BUSY
CLKIN
DB11
DB10
DB9
θ
JA Thermal Impedance . . . . . . . . . . . . . . . . . . . . . . 75°C/W
2
3
Lead Temperature, (Soldering, 10 secs) . . . . . . . . . +300°C
RD
4
SOIC, SSOP Package, Power Dissipation . . . . . . . . . 450 mW
CS
θ
JA Thermal Impedance . . . 75°C/W (SOIC) 115°C/W (SSOP)
5
REF /REF
IN OUT
AD7854
θ
JC Thermal Impedance . . . 25°C/W (SOIC) 35°C/W (SSOP)
6
AV
DGND
TOP VIEW
DD
(Not to Scale)
Lead Temperature, Soldering
7
AGND
DV
DD
Vapor Phase (60 secs) . . . . . . . . . . . . . . . . . . . . . . +215°C
Infrared (15 secs) . . . . . . . . . . . . . . . . . . . . . . . . . +220°C
8
C
C
DB8
DB7
DB6
DB5
DB4
REF1
9
REF2
10
11
12
13
14
AIN(+)
AIN(–)
HBEN
NOTES
1Stresses above those listed under Absolute Maximum Ratings may cause perma-
nent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those listed in the operational
sections of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
DB0
DB1
DB3
DB2
2Transient currents of up to 100 mA will not cause SCR latchup.
ORDERING GUIDE
Linearity Power
Temperature
Error
(LSB)
Dissipation
(mW)
Package
Option2
Model
Range1
AD7854AQ
AD7854SQ
AD7854AR
AD7854BR
–40°C to +85°C
–55°C to +125°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
1
1
1
1/2
1
1
1
1
15
15
15
15
Q-28
Q-28
R-28
R-28
RS-28
Q-28
R-28
RS-28
AD7854ARS
AD7854LAQ3
AD7854LAR3
AD7854LARS3
EVAL-AD7854CB4
15
5.5
5.5
5.5
EVAL-CONTROL BOARD5
NOTES
1Linearity error refers to the integral linearity error.
2Q = Cerdip; R = SOIC; RS = SSOP.
3L signifies the low power version.
4This can be used as a stand-alone evaluation board or in conjunction with the EVAL-CONTROL BOARD for
evaluation/demonstration purposes.
5This board is a complete unit allowing a PC to control and communicate with all Analog Devices evaluation boards
ending in the CB designator. For more information on Analog Devices products and evaluation boards visit our
World Wide Web home page at http://www.analog.com.
REV. B
–5–
AD7854/AD7854L
PIN FUNCTION DESCRIPTIONS
Pin
Mnemonic
Description
1
CONVST
Convert Start. Logic input. A low to high transition on this input puts the track/hold into its hold
mode and starts conversion. When this input is not used, it should be tied to DVDD
.
2
3
WR
RD
Write Input. Active low logic input. Used in conjunction with CS and HBEN to write to internal registers.
Read Input. Active low logic input. Used in conjunction with CS and HBEN to read from internal
registers.
4
5
CS
Chip Select Input. Active low logic input. The device is selected when this input is active.
REFIN/
REFOUT
Reference Input/Output. This pin is connected to the internal reference through a series resistor and is the
reference source for the analog-to-digital converter. The nominal reference voltage is 2.5 V and this appears
at the pin. This pin can be overdriven by an external reference and can be taken as high as AVDD. When
this pin is tied to AVDD, then the CREF1 pin should also be tied to AVDD
.
6
7
8
AVDD
AGND
CREF1
Analog Positive Supply Voltage, +3.0 V to +5.5 V.
Analog Ground. Ground reference for track/hold, reference and DAC.
Reference Capacitor (0.1 µF multilayer ceramic). This external capacitor is used as a charge source for the
internal DAC. The capacitor should be tied between the pin and AGND.
9
CREF2
Reference Capacitor (0.01 µF ceramic disc). This external capacitor is used in conjunction with the on-chip
reference. The capacitor should be tied between the pin and AGND.
10
11
12
AIN(+)
AIN(–)
HBEN
Analog Input. Positive input of the pseudo-differential analog input. Cannot go below AGND or above
AVDD at any time, and cannot go below AIN(–) when the unipolar input range is selected.
Analog Input. Negative input of the pseudo-differential analog input. Cannot go below AGND or above
AVDD at any time.
High Byte Enable Input. The AD7854 operates in byte mode only but outputs 12 bits of data during a read
cycle with HBEN low. When HBEN is high, then the high byte of data that is written to or read from the
part is on DB0 to DB7. When HBEN is low, then the lowest byte of data being written to the part is on
DB0 to DB7. If reading from the part with HBEN low, then the lowest 12 bits of data appear on pins DB0
to DB11. This allows a single read from the ADC or from the control register in a 16-bit bus system.
However, two reads are needed to access the calibration registers. Also, two writes are necessary to write to
any of the registers.
13–21 DB0–DB8
Data Bits 0 to 8. Three state data I/O pins that are controlled by CS, RD, WR and HBEN. Data output is
straight binary (unipolar mode) or twos complement (bipolar mode).
22
23
DVDD
Digital Supply Voltage, +3.0 V to +5.5 V.
DGND
Digital Ground. Ground reference point for digital circuitry.
24–26 DB9–DB11
Data Bits 9 to 11. Three state data output pins that are controlled by CS, RD and HBEN. Data output is
straight binary (unipolar mode) or twos complement (bipolar mode). These output pins should be tied to
DVDD via 100 kΩ resistors when the AD7854/AD7854L is being interfaced to an 8-bit data bus.
27
28
CLKIN
BUSY
Master Clock Signal for the device (4 MHz for AD7854, 1.8 MHz for AD7854L). Sets the conversion and
calibration times.
Busy Output. The busy output is triggered high by the falling edge of CONVST and remains high until
conversion is completed. BUSY is also used to indicate when the AD7854/AD7854L has completed its on-
chip calibration sequence.
REV. B
–6–
AD7854/AD7854L
TERMINOLOGY
Total Harmonic Distortion
Integral Nonlinearity
Total harmonic distortion (THD) is the ratio of the rms sum of
harmonics to the fundamental. For the AD7854/AD7854L, it is
defined as:
This is the maximum deviation from a straight line passing
through the endpoints of the ADC transfer function. The end-
points of the transfer function are zero scale, a point 1/2 LSB
below the first code transition, and full scale, a point 1/2 LSB
above the last code transition.
2
2
2
2
2
(V2 +V3 +V4 +V5 +V6
)
THD (dB) = 20log
V1
Differential Nonlinearity
where V1 is the rms amplitude of the fundamental and V2, V3,
V4, V5 and V6 are the rms amplitudes of the second through the
sixth harmonics.
This is the difference between the measured and the ideal 1 LSB
change between any two adjacent codes in the ADC.
Unipolar Offset Error
Peak Harmonic or Spurious Noise
This is the deviation of the first code transition (00 . . . 000 to
00 . . . 001) from the ideal AIN(+) voltage (AIN(–) + 1/2 LSB)
when operating in the unipolar mode.
Peak harmonic or spurious noise is defined as the ratio of the
rms value of the next largest component in the ADC output
spectrum (up to fS/2 and excluding dc) to the rms value of the
fundamental. Normally, the value of this specification is deter-
mined by the largest harmonic in the spectrum, but for ADCs
where the harmonics are buried in the noise floor, it will be a
noise peak.
Unipolar Gain Error
This is the deviation of the last code transition (111 . . . 110 to
111 . . . 111) from the ideal, i.e., AIN(–) +VREF/2 – 1.5 LSB,
after the unipolar offset error has been adjusted out.
Bipolar Positive Full-Scale Error
Intermodulation Distortion
This applies to the bipolar modes only and is the deviation of the
last code transition from the ideal AIN(+) voltage. For bipolar
mode, the ideal AIN(+) voltage is (AIN(–) +VREF/2 – 1.5 LSB).
With inputs consisting of sine waves at two frequencies, fa and
fb, any active device with nonlinearities will create distortion
products at sum and difference frequencies of mfa nfb where
m, n = 0, 1, 2, 3, etc. Intermodulation distortion terms are
those for which neither m nor n are equal to zero. For example,
the second order terms include (fa + fb) and (fa – fb), while the
third order terms include (2fa + fb), (2fa – fb), (fa + 2fb) and
(fa – 2fb).
Negative Full-Scale Error
This applies to the bipolar mode only and is the deviation of the
first code transition (10 . . . 000 to 10 . . . 001) from the ideal
AIN(+) voltage (AIN(–) – VREF/2 + 0.5 LSB).
Bipolar Zero Error
This is the deviation of the midscale transition (all 0s to all 1s)
from the ideal AIN(+) voltage (AIN(–) – 1/2 LSB).
Testing is performed using the CCIF standard where two input
frequencies near the top end of the input bandwidth are used. In
this case, the second order terms are usually distanced in fre-
quency from the original sine waves while the third order terms
are usually at a frequency close to the input frequencies. As a
result, the second and third order terms are specified separately.
The calculation of the intermodulation distortion is as per the
THD specification where it is the ratio of the rms sum of the
individual distortion products to the rms amplitude of the sum
of the fundamentals expressed in dBs.
Track/Hold Acquisition Time
The track/hold amplifier returns into track mode and the end of
conversion. Track/Hold acquisition time is the time required for
the output of the track/hold amplifier to reach its final value,
within 1/2 LSB, after the end of conversion.
Signal to (Noise + Distortion) Ratio
This is the measured ratio of signal to (noise + distortion) at the
output of the A/D converter. The signal is the rms amplitude of
the fundamental. Noise is the sum of all nonfundamental sig-
nals up to half the sampling frequency (fS/2), excluding dc. The
ratio is dependent on the number of quantization levels in the
digitization process; the more levels, the smaller the quantiza-
tion noise. The theoretical signal to (noise + distortion) ratio for
an ideal N-bit converter with a sine wave input is given by:
Signal to (Noise + Distortion) = (6.02 N + 1.76) dB
Thus for a 12-bit converter, this is 74 dB.
REV. B
–7–
AD7854/AD7854L
AD7854/AD7854L ON-CHIP REGISTERS
The AD7854/AD7854L powers up with a set of default conditions, and the user need not ever write to the device. In this case the
AD7854/AD7854L will operate as a read-only ADC. The WR pin should be tied to DVDD for operating the AD7854/AD7854L as a
read-only ADC.
Extra features and flexibility such as performing different power-down options, different types of calibrations including system cali-
bration, and software conversion start can be selected by writing to the part.
The AD7854/AD7854L contains a control register, ADC output data register, status register, test register and 10 calibra-
tion registers. The control register is write-only, the ADC output data register and the status register are read-only, and the test and
calibration registers are both read/write registers. The test register is used for testing the part and should not be written to.
Addressing the On-Chip Registers
Writing
To write to the AD7854/AD7854L, a 16-bit word of data must be transferred. This transfer consists of two 8-bit writes. The first
8 bits of data that are written must consist of the 8 LSBs of the 16-bit word and the second 8 bits that are written must consist of the
8 MSBs of the 16-bit word. For each of these 8-bit writes, the data is placed on Pins DB0 to DB7, Pin DB0 being the LSB of each
transfer and Pin DB7 being the MSB of each transfer. The two MSBs of the 16-bit word, ADDR1 and ADDR0, are decoded to
determine which register is addressed, and the 14 LSBs are written to the addressed register. Table I shows the decoding of the
address bits, while Figure 2 shows the overall write register hierarchy.
Table I. Write Register Addressing
ADDR1
ADDR0 Comment
0
0
1
0
1
0
This combination does not address any register.
This combination addresses the TEST REGISTER. The 14 LSBs of data are written to the test register.
This combination addresses the CALIBRATION REGISTER. The 14 least significant data bits are writ-
ten to the selected calibration register.
1
1
This combination addresses the CONTROL REGISTER. The 14 least significant data bits are written to
the control register.
Reading
To read from the various registers the user must first write to Bits 6 and 7 in the Control Register, RDSLT0 and RDSLT1. These
bits are decoded to determine which register is addressed during a read operation. Table II shows the decoding of the read address
bits while Figure 3 shows the overall read register hierarchy. The power-up status of these bits is 00 so that the default read will be
from the ADC output data register. Note: when reading from the calibration registers, the low byte must always be read first.
Once the read selection bits are set in the control register all subsequent read operations that follow are from the selected register
until the read selection bits are changed in the control register.
Table II. Read Register Addressing
RDSLT1 RDSLT0 Comment
0
0
All successive read operations are from the ADC OUTPUT DATA REGISTER. This is the default power-
up setting. There is always four leading zeros when reading from the ADC output data register.
0
1
1
1
0
1
All successive read operations are from the TEST REGISTER.
All successive read operations are from the CALIBRATION REGISTERS.
All successive read operations are from the STATUS REGISTER.
RDSLT1, RDSLT0
DECODE
ADDR1, ADDR0
DECODE
01
TEST
REGISTER
10
11
00
01
TEST
REGISTER
10
11
CALIBRATION
REGISTERS
CONTROL
REGISTER
CALIBRATION
REGISTERS
CONTROL
REGISTER
ADC OUTPUT
DATA REGISTER
GAIN(1)
OFFSET(1)
DAC(8)
GAIN(1)
OFFSET(1)
DAC(8)
GAIN(1)
GAIN(1)
OFFSET(1)
10
GAIN(1)
11
OFFSET(1)
10
GAIN(1)
11
OFFSET(1)
OFFSET(1)
00
01
00
01
CALSLT1, CALSLT0
DECODE
CALSLT1, CALSLT0
DECODE
Figure 3. Read Register Hierarchy/Address Decoding
Figure 2. Write Register Hierarchy/Address Decoding
REV. B
–8–
AD7854/AD7854L
CONTROL REGISTER
The arrangement of the control register is shown below. The control register is a write only register and contains 14 bits of data. The
control register is selected by putting two 1s in ADDR1 and ADDR0. The function of the bits in the control register is described
below. The power-up status of all bits is 0.
MSB
ZERO
ZERO
ZERO
ZERO
PMGT1
PMGT0
RDSLT1
STCAL
RDSLT0
AMODE
CONVST
CALMD
CALSLT1
CALSLT0
LSB
Control Register Bit Function Description
Bit
Mnemonic
Comment
13
12
11
10
ZERO
ZERO
ZERO
ZERO
These four bits must be set to 0 when writing to the control register.
9
8
PMGT1
PMGT0
Power Management Bits. These two bits are used for putting the part into various power-down modes
(See Power-Down section for more details).
7
6
RDSLT1
RDSLT0
Theses two bits determine which register is addressed for the read operations. See Table II.
5
AMODE
Analog Mode Bit. This pin allows two different analog input ranges to be selected. A logic 0 in this bit
position selects range 0 to VREF (i.e., AIN(+) – AIN(–) = 0 to VREF). In this range AIN(+) cannot go
below AIN(–) and AIN(–) cannot go below AGND and data coding is straight binary. A logic 1 in this
bit position selects range –VREF/2 to +VREF/2 (i.e., AIN(+) – AIN(–) = –VREF /2 to +VREF/2). AIN(+)
cannot go below AGND, so for this range, AIN(–) needs to be biased to at least +VREF/2 to allow
AIN(+) to go as low as AIN(–) –VREF/2 V. Data coding is twos complement for this range.
4
3
CONVST
CALMD
Conversion Start Bit. A logic one in this bit position starts a single conversion, and this bit is automati-
cally reset to 0 at the end of conversion. This bit may also used in conjunction with system calibration
(see Calibration section).
Calibration Mode Bit. A 0 here selects self-calibration and a 1 selects a system calibration (see Table III).
2
1
0
CALSLT1
CALSLT0
STCAL
Calibration Selection Bits and Start Calibration Bit. These bits have two functions.
With the STCAL bit set to 1, the CALSLT1 and CALSLT0 bits determine the type of calibration per-
formed by the part (see Table III). The STCAL bit is automatically reset to 0 at the end of calibration.
With the STCAL bit set to 0, the CALSLT1 and CALSLT0 bits are decoded to address the calibration
register for read/write of calibration coefficients (see section on the calibration registers for more details).
Table III. Calibration Selection
CALMD CALSLT1 CALSLT0 Calibration Type
0
0
0
A full internal calibration is initiated. First the internal DAC is calibrated, then the
internal gain error and finally the internal offset error are removed. This is the default setting.
0
0
0
1
0
1
1
0
1
0
1
0
First the internal gain error is removed, then the internal offset error is removed.
The internal offset error only is calibrated out.
The internal gain error only is calibrated out.
A full system calibration is initiated. First the internal DAC is calibrated, followed by the
system gain error calibration, and finally the system offset error calibration.
1
1
1
0
1
1
1
0
1
First the system gain error is calibrated out followed by the system offset error.
The system offset error only is removed.
The system gain error only is removed.
REV. B
–9–
AD7854/AD7854L
STATUS REGISTER
The arrangement of the status register is shown below. The status register is a read-only register and contains 16 bits of data. The
status register is selected by writing to the control register and putting two 1s in RDSLT1 and RDSLT0. The function of the bits in
the status register are described below. The power-up status of all bits is 0.
START
WRITE TO CONTROL REGISTER
SETTING RDSLT0 = RDSLT1 = 1
READ STATUS REGISTER
Figure 4. Flowchart for Reading the Status Register
MSB
ZERO
ZERO
ONE
ZERO
ZERO
BUSY
ZERO
ZERO
PMGT1
PMGT0
STCAL
ONE
AMODE
CALMD
CALSLT1 CALSLT0
LSB
Status Register Bit Function Description
Bit Mnemonic
Comment
15
14
13
12
11
10
ZERO
ZERO
ZERO
ZERO
ZERO
ZERO
These six bits are always 0.
9
8
PMGT1
PMGT0
Power Management Bits. These bits will indicate if the part is in a power-down mode or not. See Table VI
in Power-Down Section for description.
7
6
ONE
ONE
Both these bits are always 1.
5
4
3
AMODE
Analog Mode Bit. When this bit is a 0, the device is set up for the unipolar analog input range. When this
bit is a 1, the device is set up for the bipolar analog input range.
BUSY
Conversion/Calibration Busy Bit. When this bit is 1, this indicates that there is a conversion or calibration
in progress. When this bit is 0, there is no conversion or calibration in progress.
CALMD
Calibration Mode Bit. A 0 in this bit indicates a self-calibration is selected, and a 1 in this bit indicates a
system calibration is selected (see Table III).
2
1
0
CALSLT1
CALSLT0
STCAL
Calibration Selection Bits and Start Calibration Bit. The STCAL bit is read as a 1 if a calibration is in
progress and as a 0 if there is no calibration in progress. The CALSLT1 and CALSLT0 bits indicate
which of the calibration registers are addressed for reading and writing (see section on the Calibration
Registers for more details).
REV. B
–10–
AD7854/AD7854L
CALIBRATION REGISTERS
The AD7854/AD7854L has 10 calibration registers in all, 8 for the DAC, 1 for offset and 1 for gain. Data can be written to or read
from all 10 calibration registers. In self- and system calibration, the part automatically modifies the calibration registers; only if the
user needs to modify the calibration registers should an attempt be made to read from and write to the calibration registers.
Addressing the Calibration Registers
The calibration selection bits in the control register CALSLT1 and CALSLT0 determine which of the calibration registers are
addressed (See Table IV). The addressing applies to both the read and write operations for the calibration registers. The user should
not attempt to read from and write to the calibration registers at the same time.
Table IV. Calibration Register Addressing
CALSLT1
CALSLT0
Comment
0
0
1
1
0
1
0
1
This combination addresses the Gain (1), Offset (1) and DAC Registers (8). Ten registers in total.
This combination addresses the Gain (1) and Offset (1) Registers. Two registers in total.
This combination addresses the Offset Register. One register in total.
This combination addresses the Gain Register. One register in total.
Writing to/Reading from the Calibration Registers
When reading from the calibration registers there are always two
leading zeros for each of the registers.
When writing to the calibration registers a write to the control
register is required to set the CALSLT0 and CALSLT1 bits.
When reading from the calibration registers a write to the con-
trol register is required to set the CALSLT0 and CALSLT1 bits
and also to set the RDSLT1 and RDSLT0 bits to 10 (this
addresses the calibration registers for reading). The calibration
register pointer is reset on writing to the control register setting
the CALSLT1 and CALSLT0 bits, or upon completion of all
the calibration register write/read operations. When reset it
points to the first calibration register in the selected write/read
sequence. The calibration register pointer points to the gain
calibration register upon reset in all but one case, this case
being where the offset calibration register is selected on its own
(CALSLT1 = 1, CALSLT0 = 0). Where more than one cali-
bration register is being accessed, the calibration register pointer
is automatically incremented after each full calibration register
write/read operation. The calibration register address pointer is
incremented after the high byte read or write operation in byte
mode. Therefore when reading from or writing to the calibra-
tion registers, the low byte transfer must be carried out first, i.e.,
HBEN is at logic zero. The order in which the 10 calibration
registers are arranged is shown in Figure 5. Read/Write opera-
tions may be aborted at any time before all the calibration
registers have been accessed, and the next control register write
operation resets the calibration register pointer. The flowchart
in Figure 6 shows the sequence for writing to the calibration
registers. Figure 7 shows the sequence for reading from the cali-
bration registers.
START
WRITE TO CONTROL REGISTER SETTING STCAL = 0
AND CALSLT1, CALSLT0 = 00, 01, 10, 11
CAL REGISTER POINTER IS
AUTOMATICALLY RESET
WRITE TO CAL REGISTER
(ADDR1 = 1, ADDR0 = 0)
CAL REGISTER POINTER IS
AUTOMATICALLY INCREMENTED
LAST
REGISTER
NO
WRITE
OPERATION
OR
ABORT
?
YES
FINISHED
Figure 6. Flowchart for Writing to the Calibration Registers
CALIBRATION REGISTERS
(1)
(2)
(3)
GAIN REGISTER
OFFSET REGISTER
CAL REGISTER
ADDRESS POINTER
DAC 1ST MSB REGISTER
DAC 8TH MSB REGISTER (10)
CALIBRATION REGISTER ADDRESS POINTER POSITION IS
DETERMINED BY THE NUMBER OF CALIBRATION REGISTERS
ADDRESSED AND THE NUMBER OF READ/WRITE OPERATIONS.
Figure 5. Calibration Register Arrangement
REV. B
–11–
AD7854/AD7854L
has a weighting of 1.25%, and so on down to the LSB which has
a weighting of 0.0006%. This gives a resolution of 0.0006% of
VREF approximately. The resolution can also be expressed as
(0.05 × VREF)/213 volts. This equals 0.015 mV, with a 2.5 V
reference. The maximum offset that can be compensated for is
5% of the reference voltage, which equates to 125 mV with a
2.5 V reference and 250 mV with a 5 V
START
WRITE TO CONTROL REGISTER SETTING STCAL = 0, RDSLT1 = 1,
RDSLT0 = 0, AND CALSLT1, CALSLT0 = 00, 01, 10, 11
CAL REGISTER POINTER IS
AUTOMATICALLY RESET
reference.
Q. If a +20 mV offset is present in the analog input signal and the
reference voltage is 2.5 V, what code needs to be written to the
offset register to compensate for the offset ?
READ CAL REGISTER
CAL REGISTER POINTER IS
AUTOMATICALLY INCREMENTED
A. 2.5 V reference implies that the resolution in the offset reg-
ister is 5% × 2.5 V/213 = 0.015 mV. +20 mV/0.015 mV =
1310.72; rounding to the nearest number gives 1311. In
binary terms this is 00 0101 0001 1111, therefore increase
the offset register by 00 0101 0001 1111.
LAST
REGISTER
NO
READ
OPERATION
OR
ABORT
?
This method of compensating for offset in the analog input sig-
nal allows for fine tuning the offset compensation. If the offset
on the analog input signal is known, there is no need to apply
the offset voltage to the analog input pins and do a system cali-
bration. The offset compensation can take place in software.
YES
FINISHED
Figure 7. Flowchart for Reading from the Calibration
Registers
Adjusting the Gain Calibration Register
The gain calibration register contains 16 bits. The two MSBs
are zero and the 14 LSBs contain gain data. As in the offset cali-
bration register the data bits in the gain calibration register are
binary weighted, with the MSB having a weighting of 2.5% of
the reference voltage. The gain register value is effectively multi-
plied by the analog input to scale the conversion result over the
full range. Increasing the gain register compensates for a
smaller analog input range and decreasing the gain register com-
pensates for a larger input range. The maximum analog input
range that the gain register can compensate for is 1.025 times
the reference voltage, and the minimum input range is 0.975
times the reference voltage.
Adjusting the Offset Calibration Register
The offset calibration register contains 16 bits. The two MSBs
are zero and the 14 LSBs contain offset data. By changing the
contents of the offset register, different amounts of offset on the
analog input signal can be compensated for. Decreasing the
number in the offset calibration register compensates for nega-
tive offset on the analog input signal, and increasing the number
in the offset calibration register compensates for positive offset
on the analog input signal. The default value of the offset cali-
bration register is 0010 0000 0000 0000 approximately. This is
not the exact value, but the value in the offset register should be
close to this value. Each of the 14 data bits in the offset register
is binary weighted; the MSB has a weighting of 5% of the refer-
ence voltage, the MSB-1 has a weighting of 2.5%, the MSB-2
REV. B
–12–
AD7854/AD7854L
CIRCUIT INFORMATION
When using the software conversion start for maximum
throughput, the user must ensure the control register write
operation extends beyond the falling edge of BUSY. The falling
edge of BUSY resets the CONVST bit to 0 and allows it to be
reprogrammed to 1 to start the next conversion.
The AD7854/AD7854L is a fast, 12-bit single supply A/D con-
verter. The part requires an external 4 MHz/1.8 MHz master
clock (CLKIN), two CREF capacitors, a CONVST signal to start
conversion and power supply decoupling capacitors. The part
provides the user with track/hold, on-chip reference, calibration
features, A/D converter and parallel interface logic functions on
a single chip. The A/D converter section of the AD7854/
AD7854L consists of a conventional successive-approximation
converter based around a capacitor DAC. The AD7854/
AD7854L accepts an analog input range of 0 to +VREF. VREF
can be tied to VDD. The reference input to the part connected
via a 150 kΩ resistor to the internal 2.5 V reference and to the
on-chip buffer.
TYPICAL CONNECTION DIAGRAM
Figure 8 shows a typical connection diagram for the AD7854/
AD7854L. The AGND and the DGND pins are connected
together at the device for good noise suppression. The first
CONVST applied after power-up starts a self-calibration
sequence. This is explained in the calibration section of the data
sheet. Applying the RD and CS signals causes the conversion
result to be output on the 12 data pins. Note that after power is
applied to AVDD and DVDD, and the CONVST signal is applied,
the part requires (70 ms + 1/sample rate) for the internal refer-
ence to settle and for the self-calibration to be completed.
A major advantage of the AD7854/AD7854L is that a conver-
sion can be initiated in software as well as applying a signal to
the CONVST pin. The part is available in a 28-Lead SSOP
package, and this offers the user considerable space saving advan-
tages over alternative solutions. The AD7854L version typically
consumes only 5.5 mW making it ideal for battery-powered
applications.
4MHz/1.8MHz
OSCILLATOR
ANALOG
SUPPLY
+3V TO +5V
10F
0.1F
0.1F
CONVERSION
START SIGNAL
CONVERTER DETAILS
The master clock for the part is applied to the CLKIN pin.
Conversion is initiated on the AD7854/AD7854L by pulsing the
CONVST input or by writing to the control register and setting
the CONVST bit to 1. On the rising edge of CONVST (or at the
end of the control register write operation), the on-chip track/
hold goes from track to hold mode. The falling edge of the CLKIN
signal which follows the rising edge of CONVST initates the
conversion, provided the rising edge of CONVST (or WR when
converting via the control register) occurs typically at least 10 ns
before this CLKIN edge. The conversion takes 16.5 CLKIN
periods from this CLKIN falling edge. If the 10 ns setup time is
not met, the conversion takes 17.5 CLKIN periods.
AV
DD
DV
DD
CLKIN
CONVST
HBEN
0V TO 2.5V
INPUT
AIN(+)
AIN(–)
C
C
REF1
0.1
F
AD7854/
AD7854L
CS
RD
REF2
WR
0.01F
C/P
BUSY
AGND
DGND
DB0
DB11
REF /REF
IN
OUT
The time required by the AD7854/AD7854L to acquire a signal
depends upon the source resistance connected to the AIN(+)
input. Please refer to the Acquisition Time section for more
details.
0.1nF EXTERNAL REFERENCE
0.1 F ON-CHIP REFERENCE
OPTIONAL
EXTERNAL
REFERENCE
AD780/
REF192
When a conversion is completed, the BUSY output goes low,
and the result of the conversion can be read by accessing the
data through the data bus. To obtain optimum performance
from the part, read or write operations should not occur during
the conversion or less than 200 ns prior to the next CONVST
rising edge. Reading/writing during conversion typically de-
grades the Signal to (Noise + Distortion) by less than 0.5 dBs.
The AD7854 can operate at throughput rates of over 200 kSPS
(up to 100 kSPS for the AD7854L).
Figure 8. Typical Circuit
For applications where power consumption is a major concern,
the power-down options can be programmed by writing to the
part. See Power-Down section for more detail on low power
applications.
With the AD7854L, 100 kSPS throughput can be obtained as
follows: the CLKIN and CONVST signals are arranged to give
a conversion time of 16.5 CLKIN periods as described above
and 1.5 CLKIN periods are allowed for the acquisition time.
With a 1.8 MHz clock, this gives a full cycle time of 10 µs,
which equates to a throughput rate of 100 kSPS.
REV. B
–13–
AD7854/AD7854L
–72
–76
–80
ANALOG INPUT
THD VS. FREQUENCY FOR DIFFERENT
SOURCE IMPEDANCES
The equivalent analog input circuit is shown in Figure 9. Dur-
ing the acquisition interval the switches are both in the track
position and the AIN(+) charges the 20 pF capacitor through
the 125 Ω resistance. On the rising edge of CONVST switches
SW1 and SW2 go into the hold position retaining charge on the
20 pF capacitor as a sample of the signal on AIN(+). The
AIN(–) is connected to the 20 pF capacitor, and this unbalances
the voltage at Node A at the input of the comparator. The
capacitor DAC adjusts during the remainder of the conversion
cycle to restore the voltage at Node A to the correct value. This
action transfers a charge, representing the analog input signal, to
the capacitor DAC which in turn forms a digital representation
of the analog input signal. The voltage on the AIN(–) pin directly
influences the charge transferred to the capacitor DAC at the
hold instant. If this voltage changes during the conversion period,
the DAC representation of the analog input voltage is altered.
Therefore it is most important that the voltage on the AIN(–)
pin remains constant during the conversion period. Further-
more, it is recommended that the AIN(–) pin is always connected
to AGND or to a fixed dc voltage.
R
= 1k⍀
IN
–84
–88
–92
R
= 50⍀, 10nF
IN
AS IN FIGURE 13
0
20
40
60
80 100
INPUT FREQUENCY – kHz
Figure 10. THD vs. Analog Input Frequency
The maximum source impedance depends on the amount of
total harmonic distortion (THD) that can be tolerated. The
THD increases as the source impedance increases. Figure 10
shows a graph of the total harmonic distortion vs. analog input
signal frequency for different source impedances. With the
setup as in Figure 11, the THD is at the –90 dB level. With a
source impedance of 1 kΩ and no capacitor on the AIN(+) pin,
the THD increases with frequency.
125⍀ TRACK
AIN(+)
CAPACITOR
125⍀
DAC
SW1
AIN(–)
20pF
In a single supply application (both 3 V and 5 V), the V+ and
V– of the op amp can be taken directly from the supplies to the
AD7854/AD7854L which eliminates the need for extra external
power supplies. When operating with rail-to-rail inputs and out-
puts at frequencies greater than 10 kHz, care must be taken in
selecting the particular op amp for the application. In particular,
for single supply applications the input amplifiers should be
connected in a gain of –1 arrangement to get the optimum per-
formance. Figure 11 shows the arrangement for a single supply
application with a 50 Ω and 10 nF low-pass filter (cutoff fre-
quency 320 kHz) on the AIN(+) pin. Note that the 10 nF is a
capacitor with good linearity to ensure good ac performance.
Recommended single supply op amps are the AD820 and the
AD820-3V.
HOLD
NODE A
SW2
COMPARATOR
HOLD
TRACK
AGND
Figure 9. Analog Input Equivalent Circuit
Acquisition Time
The track-and-hold amplifier enters its tracking mode on the
falling edge of the BUSY signal. The time required for the
track-and-hold amplifier to acquire an input signal depends on
how quickly the 20 pF input capacitance is charged. There is a
minimum acquisition time of 400 ns. For large source imped-
ances, >2 kΩ, the acquisition time is calculated using the formula:
t
ACQ = 9 × (RIN + 125 Ω) × 20 pF
+3V TO +5V
0.1F
10F
where RIN is the source impedance of the input signal, and
125 Ω, 20 pF is the input R, C.
10k⍀
10k⍀
10k⍀
V
V+
IN
50⍀
(–V
/2 TO +V
/2)
/2
DC/AC Applications
TO AIN(+) OF
AD7854/AD7854L
REF
REF
IC1
V
10nF
(NPO)
For dc applications, high source impedances are acceptable,
provided there is enough acquisition time between conversions
to charge the 20 pF capacitor. For example with RIN = 5 kΩ,
the required acquisition time is 922 ns.
REF
AD820
V–
AD820-3V
10k⍀
Figure 11. Analog Input Buffering
For ac applications, removing high frequency components from
the analog input signal is recommended by use of an RC low-
pass filter on the AIN(+) pin, as shown in Figure 11. In applica-
tions where harmonic distortion and signal to noise ratio are
critical, the analog input should be driven from a low impedance
source. Large source impedances significantly affect the ac per-
formance of the ADC. They may require the use of an input
buffer amplifier. The choice of the amplifier is a function of the
particular application.
REV. B
–14–
AD7854/AD7854L
Transfer Functions
Input Ranges
For the unipolar range the designed code transitions occur
midway between successive integer LSB values (i.e., 1/2 LSB,
3/2 LSBs, 5/2 LSBs . . . FS – 3/2 LSBs). The output coding is
straight binary for the unipolar range with 1 LSB = FS/4096 =
3.3 V/4096 = 0.8 mV when VREF = 3.3 V. The ideal input/
output transfer characteristic for the unipolar range is shown in
Figure 14.
The analog input range for the AD7854/AD7854L is 0 V to
V
REF in both the unipolar and bipolar ranges.
The only difference between the unipolar range and the bipolar
range is that in the bipolar range the AIN(–) should be biased
up to at least +VREF/2 and the output coding is twos comple-
ment (see Table V and Figures 14 and 15).
Table V. Analog Input Connections
OUTPUT
CODE
Analog Input
Range
Input Connections
Connection
Diagram
111...111
111...110
111...101
111...100
AIN(+)
AIN(–)
1
0 V to VREF
VIN
VIN
AGND
VREF/2
Figure 12
Figure 13
VREF/22
NOTES
1Output code format is straight binary.
2Range is VREF/2 biased about VREF/2. Output code format is twos complement.
FS
4096
1LSB =
000...011
000...010
000...001
000...000
Note that the AIN(–) pin on the AD7854/AD7854L can be
biased up above AGND in the unipolar mode, or above VREF/2
in bipolar mode if required. The advantage of biasing the lower
end of the analog input range away from AGND is that the analog
input does not have to swing all the way down to AGND. Thus,
in single supply applications the input amplifier does not have to
swing all the way down to AGND. The upper end of the analog
input range is shifted up by the same amount. Care must be
taken so that the bias applied does not shift the upper end of the
analog input above the AVDD supply. In the case where the ref-
erence is the supply, AVDD, the AIN(–) should be tied to AGND
in unipolar mode or to AVDD/2 in bipolar mode.
0V 1LSB
+FS –1LSB
= (AIN(+) – AIN(–)), INPUT VOLTAGE
V
IN
Figure 14. AD7854/AD7854L Unipolar Transfer
Characteristic
Figure 13 shows the AD7854/AD7854L’s VREF/2 bipolar ana-
log input configuration. AIN(+) cannot go below 0 V, so for the
full bipolar range, AIN(–) should be biased to at least +VREF/2.
Once again the designed code transitions occur midway between
successive integer LSB values. The output coding is twos
complement with 1 LSB = 4096 = 3.3 V/4096 = 0.8 mV. The
ideal input/output transfer characteristic is shown in Figure 15.
TRACK AND HOLD
AMPLIFIER
AIN(+)
DB0
V
= 0 TO V
REF
STRAIGHT
BINARY
FORMAT
IN
. . .
AIN(–)
OUTPUT
CODE
DB11
011...111
011...110
AD7854/AD7854L
(V
/2) – 1LSB
REF
Figure 12. 0 to VREF Unipolar Input Configuration
000...001
000...000
111...111
0V
+ FS – 1LSB
(V
/2) + 1 LSB
FS = V
REF
TRACK AND HOLD
V
REF
AIN(+)
AMPLIFIER
000...010
DB0
2’S
V
= 0 TO V
IN
REF
FS
4096
1LSB =
. . .
000...001
000...000
COMPLEMENT
FORMAT
AIN(–)
DB11
V
/2
REF
AD7854/AD7854L
V
/2
REF
V
= (AIN(+) – AIN(–)), INPUT VOLTAGE
IN
Figure 15. AD7854/AD7854L Bipolar Transfer Characteristic
Figure 13. VREF/2 about VREF/2 Bipolar Input Configuration
REV. B
–15–
AD7854/AD7854L
REFERENCE SECTION
AD7854/AD7854L PERFORMANCE CURVES
Figure 18 shows a typical FFT plot for the AD7854 at 200 kHz
sample rate and 10 kHz input frequency.
For specified performance, it is recommended that when using
an external reference, this reference should be between 2.3 V
and the analog supply AVDD. The connections for the reference
pins are shown below. If the internal reference is being used,
the REFIN/REFOUT pin should be decoupled with a 100 nF
capacitor to AGND very close to the REFIN/REFOUT pin. These
connections are shown in Figure 16.
0
AV = DV = 3.3V
DD
DD
–20
–40
–60
–80
F
F
= 200kHz
= 10kHz
SAMPLE
IN
SNR = 72.04dB
THD = –88.43dB
If the internal reference is required for use external to the ADC,
it should be buffered at the REFIN/REFOUT pin and a 100 nF
capacitor should be connected from this pin to AGND. The typical
noise performance for the internal reference, with 5 V supplies is
150 nV/√Hz @ 1 kHz and dc noise is 100 µV p-p.
ANALOG
SUPPLY
+3V TO +5V
–100
–120
10
F
0.1F
0.1F
0
20
40
60
80
100
FREQUENCY – kHz
AV
DV
DD
DD
C
C
REF1
REF2
Figure 18. FFT Plot
0.1
0.01
0.1
F
F
F
Figure 19 shows the SNR versus frequency for different supplies
and different external references.
AD7854/
AD7854L
74
REF /REF
IN
OUT
AV = DV WITH 2.5V REFERENCE
UNLESS STATED OTHERWISE
DD
DD
73
72
71
70
69
Figure 16. Relevant Connections Using Internal Reference
5.0V SUPPLIES, WITH 5V REFERENCE
The REFIN/REFOUT pin may be overdriven by connecting it to
an external reference. This is possible due to the series resis-
tance from the REFIN/REFOUT pin to the internal reference.
5.0V SUPPLIES
5.0V SUPPLIES, L VERSION
3.3V SUPPLIES
This external reference can be in the range 2.3 V to AVDD
.
When using AVDD as the reference source, the 10 nF capacitor
from the REFIN/REFOUT pin to AGND should be as close as
possible to the REFIN/REFOUT pin, and also the CREF1 pin
should be connected to AVDD to keep this pin at the same volt-
age as the reference. The connections for this arrangement are
shown in Figure 17. When using AVDD it may be necessary to
add a resistor in series with the AVDD supply. This has the effect
of filtering the noise associated with the AVDD supply.
0
20
40
60
80
100
INPUT FREQUENCY – kHz
Figure 19. SNR vs. Frequency
Figure 20 shows the power supply rejection ratio versus fre-
quency for the part. The power supply rejection ratio is defined
as the ratio of the power in ADC output at frequency f to the
power of a full-scale sine wave:
Note that when using an external reference, the voltage present
at the REFIN/REFOUT pin is determined by the external refer-
ence source resistance and the series resistance of 150 kΩ from
the REFIN/REFOUT pin to the internal 2.5 V reference. Thus, a
low source impedance external reference is recommended.
PSRR (dB) = 10 log (Pf/Pfs)
Pf = Power at frequency f in ADC output, Pfs = power of a full-
scale sine wave. Here a 100 mV peak-to-peak sine wave is
coupled onto the AVDD supply while the digital supply is left
unaltered. Both the 3.3 V and 5.0 V supply performances are
shown.
ANALOG
SUPPLY
+3V TO +5V
10F
0.1
F
0.1F
AV
DV
DD
DD
C
C
REF1
REF2
0.1
F
F
AD7854/
AD7854L
0.01
REF /REF
IN
OUT
0.01F
Figure 17. Relevant Connections, AVDD as the Reference
REV. B
–16–
AD7854/AD7854L
POWER-UP TIMES
Using an External Reference
When the AD7854/AD7854L are powered up, the parts are
powered up from one of two conditions. First, when the power
supplies are initially powered up and, secondly, when the parts
are powered up from a software power-down (see last section).
–78
–80
–82
–84
–86
–88
–90
AV = DV = 3.3V/5.0V,
DD DD
100mV pk-pk SINE WAVE ON AV
DD
3.3V
When AVDD and DVDD are powered up, the AD7854/AD7854L
enters a mode whereby the CONVST signal initiates a timeout
followed by a self-calibration. The total time taken for this time-
out and calibration is approximately 70 ms—see Calibration on
Power-Up in the calibration section of this data sheet. The power-
up calibration mode can be disabled if the user writes to the control
register before a CONVST signal is applied. If the timeout and
self-calibration are disabled, then the user must take into account
the time required by the AD7854/AD7854L to power up before
a self-calibration is carried out. This power-up time is the time
taken for the AD7854/AD7854L to power up when power is
first applied (300 µs typ) or the time it takes the external refer-
ence to settle to the 12-bit level—whichever is the longer.
5.0V
0
20
40
60
80
100
INPUT FREQUENCY – kHz
Figure 20. PSRR vs. Frequency
POWER-DOWN OPTIONS
The AD7854/AD7854L provides flexible power management to
allow the user to achieve the best power performance for a given
throughput rate. The power management options are selected
by programming the power management bits, PMGT1 and
PMGT0, in the control register. Table VI summarizes the power-
down options that are available and how they can be selected by
programming the power management bits in the control register.
The AD7854/AD7854L powers up from a full software power-
down in 5 µs typ. This limits the throughput which the part is
capable of to 100 kSPS for the AD7854 and 60 kSPS for the
AD7854L when powering down between conversions. Figure 21
shows how a full power-down between conversions is implemented
using the CONVST pin. The user first selects the power-down
between conversions option by setting the power management
bits, PMGT1 and PMGT0, to 0 and 1 respectively in the control
register (see last section). In this mode the AD7854/AD7854L
automatically enters a full power-down at the end of a conver-
sion, i.e., when BUSY goes low. The falling edge of the next
CONVST pulse causes the part to power up. Assuming the
external reference is left powered up, the AD7854/AD7854L
should be ready for normal operation 5 µs after this falling edge.
The rising edge of CONVST initiates a conversion so the
CONVST pulse should be at least 5 µs wide. The part auto-
matically powers down on completion of the conversion. Where
the software convert start is used, the part may be powered up in
software before a conversion is initiated.
The AD7854/AD7854L can be fully or partially powered down.
When fully powered down, all the on-chip circuitry is powered
down and IDD is 10 µA typ. If a partial power-down is selected,
then all the on-chip circuitry except the reference is powered
down and IDD is 400 µA typ with the external clock running.
Additional power savings may be made if the external clock is off.
The choice of full or partial power-down does not give any
significant improvement in the throughput rate which can be
achieved with a power-down between conversions. This is dis-
cussed in the next section—Power-Up Times. But a partial
power-down does allow the on-chip reference to be used exter-
nally even though the rest of the AD7854/AD7854L circuitry is
powered down. It also allows the AD7854/AD7854L to be pow-
ered up faster after a long power-down period when using the
on-chip reference (See Power-Up Times section—Using the
Internal (On-Chip) Reference).
START CONVERSION ON RISING EDGE
POWER-UP ON FALLING EDGE
4.6s
5s
As can be seen from Table VI, the AD7854/AD7854L can be
programmed for normal operation, a full power-down at the end
of a conversion, a partial power-down at the end of a conversion
and finally a full power-down whether converting or not. The
full and partial power-down at the end of a conversion can be
used to achieve a superior power performance at slower through-
put rates, in the order of 50 kSPS (see Power vs. Throughput Rate
section of this data sheet).
CONVST
tCONVERT
BUSY
FULL
POWER-DOWN
POWER-UP
TIME
NORMAL
OPERATION
POWER-UP
TIME
Figure 21. Using the CONVST Pin to Power Up the AD7854
for a Conversion
Table VI. Power Management Options
PMGT1 PMGT0
Bit
Bit
Comment
0
0
1
1
0
1
0
1
Normal Operation
Full Power-Down After a Conversion
Full Power-Down
Partial Power-Down After a Conversion
REV. B
–17–
AD7854/AD7854L
Using The Internal (On-Chip) Reference
POWER VS. THROUGHPUT RATE
As in the case of an external reference the AD7854/AD7854L can
power up from one of two conditions, power-up after the sup-
plies are connected or power-up from a software power-down.
The main advantage of a full power-down after a conversion is
that it significantly reduces the power consumption of the part
at lower throughput rates. When using this mode of operation,
the AD7854/AD7854L is only powered up for the duration of
the conversion. If the power-up time of the AD7854/AD7854L
is taken to be 5 µs and it is assumed that the current during
power-up is 4.5 mA/1.5 mA typ, then power consumption as a
function of throughput can easily be calculated. The AD7854
has a conversion time of 4.6 µs with a 4 MHz external clock,
and the AD7854L has a conversion time of 9 µs with a 1.8 MHz
clock. This means the AD7854/AD7854L consumes 4.5 mA/
1.5 mA typ for 9.6 µs/14 µs in every conversion cycle if the parts
are powered down at the end of a conversion. The four graphs,
Figures 24, 25, 26 and 27, show the power consumption of the
AD7854 and AD7854L for VDD = 3 V as a function of through-
put. Table VII lists the power consumption for various throughput
rates.
When using the on-chip reference and powering up when AVDD
and DVDD are first connected, it is recommended that the power-
up calibration mode be disabled as explained above. When using
the on-chip reference, the power-up time is effectively the time
it takes to charge up the external capacitor on the REFIN
REFOUT pin. This time is given by the equation:
/
t
UP = 9 × R × C
where R ≈ 150K and C = external capacitor.
The recommended value of the external capacitor is 100 nF;
this gives a power-up time of approximately 135 ms before a
calibration is initiated and normal operation should commence.
When CREF is fully charged, the power-up time from a software
power-down reduces to 5 µs. This is because an internal switch
opens to provide a high impedance discharge path for the refer-
ence capacitor during power-down—see Figure 22. An added
advantage of the low charge leakage from the reference capacitor
during power-down is that even though the reference is being
powered down between conversions, the reference capacitor
holds the reference voltage to within 0.5 LSBs with throughput
rates of 100 samples/second and over with a full power-down
between conversions. A high input impedance op amp like the
AD707 should be used to buffer this reference capacitor if it is
being used externally. Note, if the AD7854/AD7854L is left in
its powered-down state for more than 100 ms, the charge on
CREF will start to leak away and the power-up time will increase.
If this longer power-up time is a problem, the user can use a
partial power-down for the last conversion so the reference
remains powered up.
Table VII. Power Consumption vs. Throughput
Power
AD7854
Power
AD7854L
Throughput Rate
1 kSPS
130 µW
1.3 mW
2.6 mW
6.48 mW
65 µW
10 kSPS
20 kSPS
50 kSPS
650 µW
1.25 mW
3.2 mW
4MHz/1.8MHz
OSCILLATOR
ANALOG
SUPPLY
+3V TO +5V
10F
0.1F
0.1F
CONVERSION
START SIGNAL
DV
AV
DD
DD
CLKIN
0V TO 2.5V
INPUT
AIN(+)
SWITCH OPENS
DURING POWER-DOWN
AD7854/
AD7854L
CONVST
AIN(–)
REF
IN/OUT
ON-CHIP
HBEN
REFERENCE
C
REF1
REF2
EXTERNAL
CAPACITOR
0.1F
AD7854/
AD7854L
CS
RD
WR
TO OTHER CIRCUITRY
C
BUF
0.01F
C/P
BUSY
AGND
DGND
Figure 22. On-Chip Reference During Power-Down
DB0
DB11
REF /REF
IN
OUT
0.1nF EXTERNAL REFERENCE
0.1F ON-CHIP REFERENCE
OPTIONAL
EXTERNAL
REFERENCE
AD780/
REF192
FULL POWER-DOWN
AFTER A CONVERSION
PMGT1 = 0
PMGT0 = 1
Figure 23. Typical Low Power Circuit
REV. B
–18–
AD7854/AD7854L
10
AD7854 FULL POWER-DOWN
= 3V CLKIN = 4MHz
AD7854 FULL POWER-DOWN
= 3V CLKIN = 4MHz
V
DD
V
DD
ON-CHIP REFERENCE
1
ON-CHIP REFERENCE
1
0.1
0.1
0.01
0.01
2
4
6
8
10
0
0
10
20
30
40
50
THROUGHPUT RATE – kSPS
THROUGHPUT RATE – kSPS
Figure 26. Power vs. Throughput AD7854
Figure 24. Power vs. Throughput AD7854
10
AD7854L FULL POWER-DOWN
AD7854L FULL POWER-DOWN
V = 3V CLKIN = 1.8MHz
DD
ON-CHIP REFERENCE
V
= 3V CLKIN = 1.8MHz
DD
1
ON-CHIP REFERENCE
1
0.1
0.1
0.01
0.01
0
10
20
30
40
50
4
8
12
16
20
0
THROUGHPUT RATE – kSPS
THROUGHPUT RATE – kSPS
Figure 27. Power vs. Throughput AD7854L
Figure 25. Power vs. Throughput AD7854L
REV. B
–19–
AD7854/AD7854L
CALIBRATION SECTION
Calibration Overview
AV = DV
DD DD
CONVERSION IS INITIATED
ON THIS EDGE
POWER ON
The automatic calibration that is performed on power-up
ensures that the calibration options covered in this section are
not required in a significant number of applications. A calibration
does not have to be initiated unless the operating conditions
change (CLKIN frequency, analog input mode, reference volt-
age, temperature, and supply voltages). The AD7854/AD7854L
has a number of calibration features that may be required in
some applications, and there are a number of advantages in per-
forming these different types of calibration. First, the internal
errors in the ADC can be reduced significantly to give superior
dc performance; and second, system offset and gain errors can
be removed. This allows the user to remove reference errors
(whether it be internal or external reference) and to make use of
the full dynamic range of the AD7854/AD7854L by adjusting
the analog input range of the part for a specific system.
CONVST
BUSY
32/72 ms
TIMEOUT PERIOD
32/72 ms
AUTOMATIC
CALIBRATION
DURATION
Figure 28. Timing Arrangement for Autocalibration on
Power-On
The CONVST signal is gated with the BUSY internally so that
as soon as the timeout is initiated by the first CONVST pulse all
subsequent CONVST pulses are ignored until the BUSY signal
goes low, 32/72 ms later. The CONVST pulse that follows after
the BUSY signal goes low initiates an automatic full self-
calibration. This takes a further 32/72 ms. After calibration,
the part is accurate to the 12-bit level and the specifications
quoted on the data sheet apply, and all subsequent CONVST
pulses initiate conversions. There is no need to perform another
calibration unless the operating conditions change or unless a
system calibration is required.
There are two main calibration modes on the AD7854/AD7854L,
self-calibration and system calibration. There are various op-
tions in both self-calibration and system calibration as outlined
previously in Table III. All the calibration functions are initi-
ated by writing to the control register and setting the STCAL
bit to 1.
The duration of each of the different types of calibration is given
in Table IX for the AD7854 with a 4 MHz master clock. These
calibration times are master clock dependent. Therefore the
calibration times for the AD7854L (CLKIN = 1.8 MHz) are
larger than those quoted in Table VIII.
This autocalibration at power-on is disabled if the user writes to
the control register before the autocalibration is initiated. If the
control register write operation occurs during the first 32/72 ms
timeout period, then the BUSY signal stays high for the 32/72 ms
and the CONVST pulse that follows the BUSY going low does
not initiate an automatic full self-calibration. It initiates a con-
version and all subsequent CONVST pulses initiate conversions
as well. If the control register write operation occurs when the
automatic full self-calibration is in progress, then the calibration
is not be aborted; the BUSY signal remains high until the auto-
matic full self-calibration is complete.
Table VIII. Calibration Times (AD7854 with 4 MHz CLKIN)
Type of Self-Calibration or System Calibration
Time
Full
Gain + Offset
Offset
31.25 ms
6.94 ms
3.47 ms
3.47 ms
Self-Calibration Description
Gain
There are four different calibration options within the self-
calibration mode. There is a full self-calibration where the
DAC, internal offset, and internal gain errors are removed.
There is the (Gain + Offset) self-calibration which removes the
internal gain error and then the internal offset errors. The inter-
nal DAC is not calibrated here. Finally, there are the self-offset
and self-gain calibrations which remove the internal offset errors
and the internal gain errors respectively.
Automatic Calibration on Power-On
The automatic calibration on power-on is initiated by the first
CONVST pulse after the AVDD and DVDD power on. From the
CONVST pulse the part internally sets a 32/72 ms (4 MHz/
1.8 MHz CLKIN) timeout. This time is large enough to ensure
that the internal reference has settled before the calibration is
performed. However, if an external reference is being used, this
reference must have stabilized before the automatic calibration
is initiated. This first CONVST pulse also triggers the BUSY
signal high, and once the 32/72 ms has elapsed, the BUSY signal
goes low. At this point the next CONVST pulse that is applied
initiates the automatic full self-calibration. This CONVST pulse
again triggers the BUSY signal high, and after 32/72 ms (4 MHz/
1.8 MHz CLKIN), the calibration is completed and the BUSY
signal goes low. This timing arrangement is shown in Figure 28.
The times in Figure 28 assume a 4 MHz/1.8 MHz CLKIN
signal.
The internal capacitor DAC is calibrated by trimming each of
the capacitors in the DAC. It is the ratio of these capacitors to
each other that is critical, and so the calibration algorithm
ensures that this ratio is at a specific value by the end of the
calibration routine. For the offset and gain there are two
separate capacitors, one of which is trimmed during offset
calibration and one of which is trimmed during gain calibration.
In bipolar mode the midscale error is adjusted by an offset cali-
bration and the positive full-scale error is adjusted by the gain
calibration. In unipolar mode the zero-scale error is adjusted by
the offset calibration and the positive full-scale error is adjusted
by the gain calibration.
REV. B
–20–
AD7854/AD7854L
Self-Calibration Timing
MAX SYSTEM FULL SCALE
IS 2.5ꢀ FROM V
Figure 29 shows the timing for a software full self-calibration.
Here the BUSY line stays high for the full length of the self-
calibration. A self-calibration is initiated by writing to the con-
trol register and setting the STCAL bit to 1. The BUSY line
goes high at the end of the write to the control register, and
BUSY goes low when the full self-calibration is complete after a
time tCAL as show in Figure 29.
REF
V
+ SYS OFFSET
REF
V
– 1LSB
V
– 1LSB
REF
REF
SYSTEM OFFSET
ANALOG
INPUT
ANALOG
INPUT
RANGE
RANGE
CALIBRATION
SYS OFFSET
AGND
SYS OFFSET
AGND
t23
MAX SYSTEM OFFSET
IS 5ꢀ OF V
MAX SYSTEM OFFSET
IS 5ꢀ OF V
CS
REF
REF
DATA LATCHED INTO
CONTROL REGISTER
Figure 30. System Offset Calibration
WR
Figure 31 shows a system gain calibration (assuming a system
full scale greater than the reference voltage) where the analog
input range has been increased after the system gain calibration
is completed. A system full-scale voltage less than the reference
voltage may also be accounted for a by a system gain calibration.
Hi-Z
Hi-Z
DATA
VALID
DATA
BUSY
tCAL
MAX SYSTEM FULL SCALE
MAX SYSTEM FULL SCALE
IS 2.5ꢀ FROM V
IS 2.5ꢀ FROM V
REF
REF
Figure 29. Timing Diagram for Full Self-Calibration
SYS FULL S.
SYS FULL S.
For the self-(gain + offset), self-offset and self-gain calibrations,
the BUSY line is triggered high at the end of the write to the
control register and stays high for the full duration of the self-
calibration. The length of time for which BUSY is high depends
on the type of self-calibration that is initiated. Typical values are
given in Table VIII. The timing diagram for the other self-
calibration options is similar to that outlined in Figure 29.
V
– 1LSB
V
– 1LSB
REF
REF
SYSTEM OFFSET
ANALOG
INPUT
ANALOG
INPUT
RANGE
RANGE
CALIBRATION
AGND
AGND
Figure 31. System Gain Calibration
System Calibration Description
Finally in Figure 32 both the system offset error and gain error
are removed by the system offset followed by a system gain cali-
bration. First the analog input range is shifted upwards by the
positive system offset and then the analog input range is
adjusted at the top end to account for the system full scale.
System calibration allows the user to remove system errors
external to the AD7854/AD7854L, as well as remove the errors
of the AD7854/AD7854L itself. The maximum calibration
range for the system offset errors is 5% of VREF, and for the
system gain errors it is 2.5% of VREF. If the system offset or
system gain errors are outside these ranges, the system calibration
algorithm reduces the errors as much as the trim range allows.
MAX SYSTEM FULL SCALE
MAX SYSTEM FULL SCALE
IS 2.5ꢀ FROM V
IS 2.5ꢀ FROM V
REF
REF
Figures 30 through 32 illustrate why a specific type of system
calibration might be used. Figure 30 shows a system offset cali-
bration (assuming a positive offset) where the analog input
range has been shifted upwards by the system offset after the
system offset calibration is completed. A negative offset may
also be removed by a system offset calibration.
V
+ SYS OFFSET
SYS F.S.
REF
SYS F.S.
REF
V
– 1LSB
REF
V
– 1LSB
SYSTEM OFFSET
CALIBRATION
FOLLOWED BY
ANALOG
INPUT
RANGE
ANALOG
INPUT
RANGE
SYSTEM GAIN
CALIBRATION
SYS OFFSET
AGND
SYS OFFSET
AGND
MAX SYSTEM OFFSET
IS 5ꢀ OF V
MAX SYSTEM OFFSET
IS 5ꢀ OF V
REF
REF
Figure 32. System (Gain + Offset) Calibration
REV. B
–21–
AD7854/AD7854L
System Gain and Offset Interaction
The timing for a system (gain + offset) calibration is very similar
to that of Figure 33, the only difference being that the time tCAL1 is
replaced by a shorter time of the order of tCAL2 as the internal
DAC is not calibrated. The BUSY signal signifies when the gain
calibration is finished and when the part is ready for the offset
calibration.
The architecture of the AD7854/AD7854L leads to an interac-
tion between the system offset and gain errors when a system
calibration is performed. Therefore it is recommended to perform
the cycle of a system offset calibration followed by a system gain
calibration twice. When a system offset calibration is performed,
the system offset error is reduced to zero. If this is followed by a
system gain calibration, then the system gain error is now zero,
but the system offset error is no longer zero. A second sequence
of system offset error calibration followed by a system gain cali-
bration is necessary to reduce system offset error to below the
12-bit level. The advantage of doing separate system offset and
system gain calibrations is that the user has more control over
when the analog inputs need to be at the required levels, and the
CONVST signal does not have to be used.
DATA LATCHED INTO
CONTROL REGISTER
t23
CS
WR
DATA
BUSY
CONVST BIT SET
TO 1 IN CONTROL
REGISTER
Hi-Z
Hi-Z
Hi-Z
DATA
VALID
DATA
VALID
Alternatively, a system (gain + offset) calibration can be per-
formed. At the end of one system (gain + offset) calibration, the
system offset error is zero, while the system gain error is reduced
from its initial value. Three system (gain + offset) calibrations
are required to reduce the system gain error to below the 12-bit
error level. There is never any need to perform more than three
system (gain + offset) calibrations.
tCAL1
tCAL2
t23
tSETUP
AIN
V
V
OFFSET
SYSTEM FULL SCALE
Figure 33. Timing Diagram for Full System Calibration
The timing diagram for a system offset or system gain calibra-
tion is shown in Figure 34. Here again a write to the control
register initiates the calibration sequence. At the end of the con-
trol register write operation the BUSY line goes high and it stays
high until the calibration sequence is finished. The analog input
should be set at the correct level for a minimum setup time
(tSETUP) of 100 ns before the CS rising edge and stay at the cor-
rect level until the BUSY signal goes low.
In bipolar mode the midscale error is adjusted for an offset cali-
bration and the positive full-scale error is adjusted for the gain
calibration; in unipolar mode the zero-scale error is adjusted for
an offset calibration and the positive full-scale error is adjusted
for a gain calibration.
System Calibration Timing
The timing diagram in Figure 33 is for a software full system
calibration. It may be easier in some applications to perform
separate gain and offset calibrations so that the CONVST bit in
the control register does not have to be programmed in the
middle of the system calibration sequence. Once the write to the
control register setting the bits for a full system calibration is
completed, calibration of the internal DAC is initiated and the
BUSY line goes high. The full-scale system voltage should be
applied to the analog input pins, AIN(+) and AIN(–) at the start
of calibration. The BUSY line goes low once the DAC and
system gain calibration are complete. Next the system offset
voltage should be applied across the AIN(+) and AIN(–) pins
for a minimum setup time (tSETUP) of 100 ns before the rising
edge of CS. This second write to the control register sets the
CONVST bit to 1 and at the end of this write operation the
BUSY signal is triggered high (note that a CONVST pulse can
be applied instead of this second write to the control register).
The BUSY signal is low after a time tCAL2 when the system offset
calibration section is complete. The full system calibration is now
complete.
t23
CS
DATA LATCHED INTO
CONTROL REGISTER
WR
Hi-Z
Hi-Z
DATA
VALID
DATA
BUSY
tCAL2
tSETUP
AIN
V
OR V
OFFSET
SYSTEM FULL SCALE
Figure 34. Timing Diagram for System Gain or System
Offset Calibration
REV. B
–22–
AD7854/AD7854L
PARALLEL INTERFACE
Reading
The timing diagram for a read cycle is shown in Figure 35. The
CONVST and BUSY signals are not shown here as the read
cycle may occur while a conversion is in progress or after the
conversion is complete.
to DB11. Bringing HBEN high causes the 8 MSBs of the 16-bit
word to be output on pins DB0 to DB7. Note that with this
arrangement the data lines are always active.
t1 = 100ns MIN, t20 = 70ns MIN,
t19
= t20 = 70ns MIN, t21 = t22 = 60ns MAX
t1
The HBEN signal is low for the first read and high for the sec-
ond read. This ensures that it is the lower 12 bits of the 16-bit
word are output in the first read and the 8 MSBs of the 16-bit
word are output in the second read. If required, the HBEN
signal may be high for the first read and low for the second
read to ensure that the high byte is output in the first read
and the lower byte in the second read. The CS and RD sig-
nals are gated together internally and level triggered active
low. Both CS and RD may be tied together as the timing speci-
fication for t5 and t6 are both 0 ns min. The data is output a time t8
after both CS and RD go low. The RD rising edge should be
used to latch the data by the user and after a time t9 the data
lines will go into their high impedance state.
CONVST
CONVERSION IS INITIATED ON THIS EDGE
tCONVERT
t2
BUSY
HBEN
t20
t19
t21
t18
t22
NEW DATA
VALID
(DB0–DB11)
NEW DATA
VALID
(DB8–DB11)
NEW DATA
VALID
(DB0–DB11)
NEW DATA
VALID
(DB8–DB11)
OLD DATA VALID
DATA
ON PINS DB0 TO DB11
ON PINS DB0 TO DB7
Figure 36. Read Cycle Timing Diagram with CS and RD
Tied Low
In Figure 35, the first read outputs the 12 LSBs of the 16-bit
word on pins DB0 to DB11 (DB0 being the LSB of the 12-bit
read). The second read outputs the 8 MSBs of the 16-bit word
on pins DB0 to DB7 (DB0 being the LSB of the 8-bit read). If the
system has a 12-bit or a 16-bit data bus, only one read operation
is necessary to obtain the 12-bit conversion result (12 bits are
output in the first read). A second read operation is not required.
Writing
The timing diagram for a write cycle is shown in Figure 37. The
CONVST and BUSY signals are not shown here as the write
cycle may occur while a conversion is in progress or after the
conversion is complete.
To write a 16-bit word to the AD7854/AD7854L, two 8-bit
writes are required. The HBEN signal must be low for the first
write and high for the second write. This ensures that it is the
lower 8 bits of the 16-bit word are latched in the first write and
the 8 MSBs of the 16-bit word are latched in the second write.
For both write operations the 8 bits of data should be present on
pins DB0 to DB7 (DB0 being the LSB of the 8-bit write). Any
data on pins DB8 to DB11 is ignored when writing to the device.
The CS and WR signals are gated together internally. Both CS
and WR may be tied together as the timing specification for t13
and t14 are both 0 ns min. The data is latched on the rising edge
of WR. The data needs to be set up a time t16 before the WR
rising edge and held for a time t17 after the WR rising edge.
If the system has an 8-bit data bus then two reads are needed.
Pins DB0 to DB7 should be connected the 8-bit data bus. Pins
DB8 to DB11 should be tied to DGND or DVDD via 10 kΩ
resistors. With this arrangement, HBEN is pulled low for the
first read and the 8 LSBs of the 16-bit word are output on pins
DB0 to DB7 (data on pins DB8 to DB11 will be ignored).
HBEN is pulled high for the second read and now the 8 MSBs
of the 16-bit word are output on pins DB0 to DB7.
t3 = 15ns MIN, t4 = 5ns MIN, t5
= t6 = 0ns MIN,
t8 = 50ns MAX, t9 = 5/40ns MIN/MAX, t10 = 70ns MIN
HBEN
t11 = 0ns MIN, t12 = 5ns MIN, t13
= t14 = 0ns MIN,
t4
t3
t4
t15 = 70ns MIN, t16 = 10ns MIN, t17 = 5ns MIN
t3
CS
HBEN
CS
t10
t5
t6
t9
t11
t11
t12
t10
t12
t7
RD
t13
t8
t14
DATA
VALID
DATA
VALID
DATA
t15
t16
WR
t17
Figure 35. Read Cycle Timing Diagram Using CS and RD
DATA
VALID
DATA
VALID
DATA
In the case where the AD7854/AD7854L is operated as a read-
only ADC, the WR pin can be tied permanently high. The read
operation need only consist of one read if the system has a 12-
bit or a 16-bit data bus.
Figure 37. Write Cycle Timing Diagram
Resetting the Parallel Interface
If random data has been inadvertently written to the test regis-
ter, it is necessary to write the 16-bit word 0100 0000 0000
0010 (in two 8-bit bytes) to restore the test register to its
default value.
When both the CS and RD signals are tied permanently low a
different timing arrangement results, as shown in Figure 36.
Here the data is output a time t20 before the falling edge of the
BUSY signal. This allows the falling edge of BUSY to be used
for latching the data. Again if HBEN is low during the conver-
sion the 12 LSBs of the 16-bit word will be output on pins DB0
REV. B
–23–
AD7854/AD7854L
MICROPROCESSOR INTERFACING
AD7854/AD7854L to TMS32020, TMS320C25 and TMS320C5x
A parallel interface between the AD7854/AD7854L and the
TMS32020, TMS320C25 and TMS320C5x family of DSPs are
shown in Figure 39. The memory mapped addresses chosen for
the AD7854/AD7854L should be chosen to fall in the I/O
memory space of the DSPs.
The parallel port on the AD7854/AD7854L allows the device to
be interfaced to microprocessors or DSP processors as a memory
mapped or I/O mapped device. The CS and RD inputs are
common to all memory peripheral interfacing. Typical inter-
faces to different processors are shown in Figures 38 to 41.
In all the interfaces shown, an external timer controls the
CONVST input of the AD7854/AD7854L and the BUSY out-
put interrupts the host DSP. Also, the HBEN pin is connected
to address line A0 (XA0 in the case of the TMS320C30). This
maps the AD7854/AD7854L to two locations in the processor
memory space, ADCaddr and ADCaddr+1. Thus when writing
to the ADC, first the 8 LSBs of the 16-bit are written to address
location ADCaddr and then the 8 MSBs to location ADCaddr+1.
All the interfaces use a 12-bit data bus, so only one read is needed
from location ADCaddr to access the ADC output data register
or the status register. To read from the other registers, the
8 MSBs must be read from location ADCaddr+1. Interfacing
to 8-bit bus systems is similar, except that two reads are
required to obtain data from all the registers.
The parallel interface on the AD7854/AD7854L is fast enough
to interface to the TMS32020 with no extra wait states. In the
TMS320C25 interface, data accesses may be slowed sufficiently
when reading from and writing to the part to require the inser-
tion of one wait state. In such a case, this wait state can be
generated using the single OR gate to combine the CS and
MSC signals to drive the READY line of the TMS320C25, as
shown in Figure 39. Extra wait states are necessary when using
the TMS320C5x at their fastest clock speeds. Wait states can
be programmed via the IOWSR and CWSR registers (please see
TMS320C5x User Guide for details).
Data is read from the ADC using the following instruction:
IN D,ADCaddr
where D is the memory location where the data is to be stored
and ADCaddr is the I/O address of the AD7854/AD7854L.
AD7854/AD7854L to ADSP-21xx
Figure 38 shows the AD7854/AD7854L interfaced to the
ADSP-21xx series of DSPs as a memory mapped device. A
single wait state may be necessary to interface the AD7854/
AD7854L to the ADSP-21xx depending on the clock speed of
the DSP. This wait state can be programmed via the data
memory waitstate control register of the ADSP-21xx (please see
ADSP-2100 Family Users Manual for details). The following
instruction reads data from the AD7854/AD7854L:
Data is written to the ADC using the following two instructions:
OUT D8LSB, ADCaddr
OUT D8MSB, ADCaddr+1
where D8LSB is the memory location where the 8 LSBs of data
are stored, D8MSB is the location where the 8 MSBs of data are
stored and ADCaddr and ADCaddr+1 are the I/O memory
spaces that the AD7854/AD7854L is mapped into.
AX0 = DM(ADCaddr)
Data can be written to the AD7854/AD7854L using the
instructions:
A15–A1
ADDRESS BUS
TMS32020/
TMS320C25/
TMS320C50*
DM (ADCaddr) = AY0
ADDR
EN
CS
IS
DM (ADCaddr+1) = AY1
DECODE
AD7854/
where ADCaddr is the address of the AD7854/AD7854L in
ADSP-21xx data memory, AX0 contains the data read from the
ADC, and AY0 contains the 8 LSBs and AY1 the 8 MSBs of
data written to the AD7854/AD7854L.
READY
MSC
A0
AD7854L*
TMS320C25
ONLY
HBEN
STRB
R/W
WR
RD
A13–A1
ADSP-21xx*
ADDRESS BUS
INTx
BUSY
DB11–DB0
DATA BUS
D23–D0
ADDR
EN
DMS
CS
DECODE
AD7854/
AD7854L*
*ADDITIONAL PINS OMITTED FOR CLARITY
A0
HBEN
Figure 39. AD7854/AD7854L to TMS32020/C25/C5x
Parallel Interface
WR
WR
RD
RD
IRQ2
BUSY
DB11–DB0
D23–D8
DATA BUS
*ADDITIONAL PINS OMITTED FOR CLARITY
Figure 38. AD7854/AD7854L to ADSP-21xx Parallel Interface
REV. B
–24–
AD7854/AD7854L
AD7854/AD7854L to TMS320C30
AD7854/AD7854L to DSP5600x
Figure 40 shows a parallel interface between the AD7854/
AD7854L and the TMS320C3x family of DSPs. The
AD7854/AD7854L is interfaced to the Expansion Bus of the
TMS320C3x. Two wait states are required in this interface.
These can be programmed using the WTCNT bits of the
Expansion Bus Control register (see TMS320C3x Users Guide
for details). Data from the AD7854/AD7854L can be read
using the following instruction:
Figure 41 shows a parallel interface between the AD7854/
AD7854L and the DSP5600x series of DSPs. The AD7854/
AD7854L should be mapped into the top 64 locations of Y data
memory. If extra wait states are needed in this interface, they
can be programmed using the Port A bus control register (please
see DSP5600x User’s Manual for details). Data can be read
from the DSP5600x using the following instruction:
MOVE Y:ADCaddr, X0
LDI *ARn,Rx
Data can be written to the AD7854/AD7854L using the follow-
ing two instructions:
Data can be loaded into the AD7854/AD7854L using the
instructions:
MOVE X0, Y:ADCaddr
STI Ry,*ARn++
STI Rz,*ARn--
MOVE X1, Y:ADCaddr+1
Where ADCaddr is the address in the DSP5600x address space
where ARn is an auxiliary register containing the lower 16 bits
of the address of the AD7854/AD7854L in the TMS320C3x
memory space, Rx is the register into which the ADC data is
loaded during a load operation, Ry contains the 8 LSBs of
data and Rz contains the 8 MSBs of data to be written to the
AD7854/AD7854L.
to which the AD7854/AD7854L has been mapped.
A15–A1
ADDRESS BUS
DSP56000/
DSP56002*
X/Y
ADDR
DECODE
CS
DS
AD7854/
AD7854L*
A0
WR
RD
HBEN
XA12–XA1
EXPANSION ADDRESS BUS
WR
RD
TMS320C30*
ADDR
DECODE
CS
IRQ
BUSY
AD7854/
AD7854L*
DB11–DB0
DATA BUS
D23–D0
XA0
HBEN
IOSTRB
XR/W
WR
*ADDITIONAL PINS OMITTED FOR CLARITY
RD
BUSY
INTx
Figure 41. AD7854/AD7854L to DSP5600x Parallel Interface
DB11–DB0
EXPANSION DATA BUS
XD23–XD0
*ADDITIONAL PINS OMITTED FOR CLARITY
Figure 40. AD7854/AD7854L to TMS320C30 Parallel
Interface
REV. B
–25–
AD7854/AD7854L
Good decoupling is also important. All analog supplies should
be decoupled with a 10 µF tantalum capacitor in parallel with
0.1 µF disc ceramic capacitor to AGND. All digital supplies
should have a 0.1 µF disc ceramic capacitor to DGND. To
achieve the best performance from these decoupling compo-
nents, they must be placed as close as possible to the device,
ideally right up against the device. In systems where a common
supply voltage is used to drive both the AVDD and DVDD of the
AD7854/AD7854L, it is recommended that the system’s AVDD
supply is used. In this case an optional 10 Ω resistor between
the AVDD pin and DVDD pin can help to filter noise from digital
circuitry. This supply should have the recommended analog
supply decoupling capacitors between the AVDD pin of the
AD7854/AD7854L and AGND and the recommended digital
supply decoupling capacitor between the DVDD pin of the
AD7854/AD7854L and DGND.
APPLICATION HINTS
Grounding and Layout
The analog and digital supplies of the AD7854/AD7854L are
independent and separately pinned out to minimize coupling
between the analog and digital sections of the device. The part
has very good immunity to noise on the power supplies as can
be seen by the PSRR versus frequency graph. However, care
should still be taken with regard to grounding and layout.
The printed circuit board on which the AD7854/AD7854L is
mounted should be designed such that the analog and digital
sections are separated and confined to certain areas of the
board. This facilitates the use of ground planes that can be
easily separated. A minimum etch technique is generally best
for ground planes as it gives the best shielding. Digital and
analog ground planes should only be joined in one place. If
the AD7854/AD7854L is the only device requiring an AGND
to DGND connection, then the ground planes should be
connected at the AGND and DGND pins of the AD7854/
AD7854L. If the AD7854/AD7854L is in a system where
multiple devices require AGND to DGND connections, the
connection should still be made at one point only, a star ground
point which should be established as close as possible to the
AD7854/AD7854L.
Evaluating the AD7854/AD7854L Performance
The recommended layout for the AD7854/AD7854L is outlined
in the evaluation board for the AD7854/AD7854L. The evalua-
tion board package includes a fully assembled and tested
evaluation board, documentation, and software for controlling
the board from the PC via the EVAL-CONTROL BOARD.
The EVAL-CONTROL BOARD can be used in conjunction
with the AD7854/AD7854L Evaluation board, as well as many
other Analog Devices evaluation boards ending in the CB desig-
nator, to demonstrate/evaluate the ac and dc performance of the
AD7854/AD7854L.
Avoid running digital lines under the device as these couple
noise onto the die. The analog ground plane should be allowed
to run under the AD7854/AD7854L to avoid noise coupling.
The power supply lines to the AD7854/AD7854L should use as
large a trace as possible to provide low impedance paths and
reduce the effects of glitches on the power supply line. Fast
switching signals like clocks and the data inputs should be
shielded with digital ground to avoid radiating noise to other
sections of the board and clock signals should never be run near
the analog inputs. Avoid crossover of digital and analog signals.
Traces on opposite sides of the board should run at right angles
to each other. This reduces the effects of feedthrough through
the board. A microstrip technique is by far the best but is not
always possible with a double-sided board. In this technique,
the component side of the board is dedicated to ground planes
while signals are placed on the solder side.
The software allows the user to perform ac (fast Fourier trans-
form) and dc (histogram of codes) tests on the AD7854/
AD7854L. It also gives full access to all the AD7854/AD7854L
on-chip registers allowing for various calibration and power-
down options to be programmed.
AD785x Family
All parts are 12 bits, 200 kSPS, 3.0 V to 5.5 V.
AD7853 – Single Channel Serial
AD7854 – Single Channel Parallel
AD7858 – Eight Channel Serial
AD7859 – Eight Channel Parallel
REV. B
–26–
AD7854/AD7854L
PAGE INDEX
Topic
Page
Topic
Page
FEATURES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
GENERAL DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . . 1
PRODUCT HIGHLIGHTS . . . . . . . . . . . . . . . . . . . . . . . . . 1
SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
TIMING SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . 4
ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . 5
ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
PINOUTS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
TERMINOLOGY . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
PIN FUNCTION DESCRIPTION . . . . . . . . . . . . . . . . . . . 7
AD7854/AD7854L ON-CHIP REGISTERS . . . . . . . . . . . . 8
Addressing the On-Chip Registers . . . . . . . . . . . . . . . . . . . 8
Writing/Reading . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
CONTROL REGISTER . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
STATUS REGISTER . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
CALIBRATION REGISTERS . . . . . . . . . . . . . . . . . . . . . . 11
Addressing the Calibration Registers . . . . . . . . . . . . . . . . 11
Writing to/Reading from the Calibration Registers . . . . . . 11
Adjusting the Offset Calibration Register . . . . . . . . . . . . . 12
Adjusting the Gain Calibration Register . . . . . . . . . . . . . . 12
CIRCUIT INFORMATION . . . . . . . . . . . . . . . . . . . . . . . . 13
CONVERTER DETAILS . . . . . . . . . . . . . . . . . . . . . . . . . . 13
TYPICAL CONNECTION DIAGRAM . . . . . . . . . . . . . . 13
ANALOG INPUT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Acquisition Time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
DC/AC Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Input Ranges . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Transfer Functions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
REFERENCE SECTION . . . . . . . . . . . . . . . . . . . . . . . . . . 16
AD7854/AD7854L PERFORMANCE CURVES . . . . . . . . 16
POWER-DOWN OPTIONS . . . . . . . . . . . . . . . . . . . . . . . . 17
POWER-UP TIMES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Using an External Reference . . . . . . . . . . . . . . . . . . . . . . 17
Using the Internal (On-Chip) Reference . . . . . . . . . . . . . 18
POWER VS. THROUGHPUT RATE . . . . . . . . . . . . . . . . 18
CALIBRATION SECTION . . . . . . . . . . . . . . . . . . . . . . . . 20
Calibration Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Automatic Calibration on Power-On . . . . . . . . . . . . . . . . 20
Self-Calibration Description . . . . . . . . . . . . . . . . . . . . . . . 20
Self-Calibration Timing . . . . . . . . . . . . . . . . . . . . . . . . . . 21
System Calibration Description . . . . . . . . . . . . . . . . . . . . 21
System Gain and Offset Interaction . . . . . . . . . . . . . . . . . 22
System Calibration Timing . . . . . . . . . . . . . . . . . . . . . . . 22
PARALLEL INTERFACE . . . . . . . . . . . . . . . . . . . . . . . . . 23
Reading . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Writing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Resetting the Parallel Interface . . . . . . . . . . . . . . . . . . . . . 23
MICROPROCESSOR INTERFACING . . . . . . . . . . . . . . . 24
AD7854/AD7854L to ADSP-21xx . . . . . . . . . . . . . . . . . . 24
AD7854/AD7854L to TMS32020, TMS320C25 and
TMS320C5x . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
AD7854/AD7854L to TMS320C30 . . . . . . . . . . . . . . . . 25
AD7854/AD7854L to DSP5600x . . . . . . . . . . . . . . . . . . 25
APPLICATIONS HINTS . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Grounding and Layout . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Evaluating the AD7854/AD7854L Performance . . . . . . . 26
INDEX . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . 28
TABLE INDEX
#
I
Title
Page
Write Register Addressing . . . . . . . . . . . . . . . . . . . . . . . 8
Read Register Addressing . . . . . . . . . . . . . . . . . . . . . . . 8
II
III Calibration Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
IV Calibrating Register Addressing . . . . . . . . . . . . . . . . . 11
V
Analog Input Connections . . . . . . . . . . . . . . . . . . . . . 15
VI Power Management Options . . . . . . . . . . . . . . . . . . . . 17
VII Power Consumption vs. Throughput . . . . . . . . . . . . . 18
VIII Calibration Times (AD7854 with 4 MHz CLKIN) . . . 20
REV. B
–27–
AD7854/AD7854L
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
28-Lead Cerdip
(Q-28)
0.005 (0.13) MIN
28
0.100 (2.54) MAX
15
0.610 (15.49)
0.500 (12.70)
1
14
0.620 (15.75)
0.590 (14.99)
PIN 1
0.015
(0.38)
MIN
0.150
(3.81)
MIN
1.490 (37.85) MAX
0.225
(5.72)
MAX
0.018 (0.46)
0.008 (0.20)
0.200 (5.08)
0.125 (3.18)
0.026 (0.66) 0.110 (2.79)
0.014 (0.36) 0.090 (2.29)
0.070 (1.78)
0.030 (0.76)
15°
0°
SEATING
PLANE
28-Lead Small Outline Package
(R-28)
0.7125 (18.10)
0.6969 (17.70)
28
15
1
14
0.1043 (2.65)
0.0926 (2.35)
PIN 1
0.0291 (0.74)
0.0098 (0.25)
؋
45° 0.0500 (1.27)
0.0157 (0.40)
8°
0°
0.0192 (0.49)
0.0138 (0.35)
0.0118 (0.30)
0.0040 (0.10)
0.0500
(1.27)
BSC
SEATING
PLANE
0.0125 (0.32)
0.0091 (0.23)
28-Lead Shrink Small Outline Package
(RS-28)
0.407 (10.34)
0.397 (10.08)
28
15
1
14
0.07 (1.79)
0.066 (1.67)
0.078 (1.98)
0.068 (1.73)
PIN 1
0.03 (0.762)
0.022 (0.558)
8°
0°
0.015 (0.38)
0.010 (0.25)
0.0256
(0.65)
BSC
0.008 (0.203)
0.002 (0.050)
SEATING
PLANE
0.009 (0.229)
0.005 (0.127)
REV. B
–28–
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