AD8330ARQ-REEL7 [ADI]
Low Cost DC-150 MHz Variable Gain Amplifier; 低成本DC - 150 MHz可变增益放大器
| 型号： | AD8330ARQ-REEL7 |
| 厂家： | 
ADI |
| 描述： | Low Cost DC-150 MHz Variable Gain Amplifier |
| 文件： | 总28页 (文件大小：634K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Low Cost DC-150 MHz
Variable Gain Amplifier
AD8330*
FEATURES
FUNCTIONAL BLOCK DIAGRAM
Fully Differential Signal Path
16
15
OFST
14
13
May also be Used with Single-Sided Signals
Inputs from 0.3 mV to 1 V rms, Rail-Rail Outputs
Differential RIN = 1 kꢀ; ROUT (Each Output) 75 ꢀ
Automatic Offset Compensation (Optional)
Linear-in-dB and Linear-in-Magnitude Gain Modes
0 dB to 50 dB, for 0 V < VDBS < 1.5 V (30 mV/dB)
Inverted Gain Mode: 50 dB to 0 dB at –30 mV/dB
ꢁ0.03 to ꢁ10 Nominal Gain for 15 mV < VMAG < 5 V
Constant Bandwidth: 150 MHz at All Gains
Low Noise: 5 nV/√Hz typical at Maximum Gain
Low Distortion: ≤–62 dBc Typ
ENBL
VPOS
CNTR
1
2
3
4
12
11
VPSO
VPSI
CM AND
BIAS AND V
REF
OFFSET
CONTROL
INHI
OPHI
OUTPUT
STAGES
VGA CORE
INLO
OPLO
10
9
Low Power: 20 mA Typ at VS of 2.7 V – 6 V
Available in Space Saving 3 ꢁ 3 LFCSP Package
MODE
OUTPUT
CONTROL
GAIN INTERFACE
CMOP
VMAG
APPLICATIONS
VDBS
CMGN
6
COMM
Pre-ADC Signal Conditioning
75 ꢀ Cable Driving Adjust
AGC Amplifiers
5
7
8
90
70
50
GENERAL DESCRIPTION
The AD8330 is a wideband variable-gain amplifier for use in
applications requiring a fully differential signal path, low noise, well-
defined gain, and moderately low distortion, from dc to 150 MHz.
The input pins can also be driven from a single ended source. The
peak differential input is 2 V, allowing sinewave operation at
1 V rms with generous headroom. The output pins can optionally
drive single-sided loads and each swing essentially rail-to-rail.
The differential output resistance is 150 Ω. The output swing is
a linear function of the voltage applied to the VMAG pin, which
internally defaults to 0.5 V, to provide a peak output of 2 V.
This may be raised to 10 V p-p, limited by the supply voltage.
30
10
–10
–30
–50
100k
1M
10M
FREQUENCY – Hz
100M
300M
The basic gain function is linear-in-dB, controlled by the voltage
applied to pin VDBS. The gain ranges from 0 dB to 50 dB for
control voltages between 0 V and 1.5 V—a slope of 30 mV dB.
The gain linearity is typically within 0.1 dB. By changing the
logic level on pin MODE, the gain will decrease over the same
range, with opposite slope. A second gain control port is provided
at pin VMAG and allows the user to vary the numeric gain from a
factor of 0.03 to 10. All the parameters of the AD8330 have low
sensitivities to temperature and supply voltages.
Figure 1. AC Response over the Extended Gain Range
range, but also when using the linear-in-magnitude function. The
exceptional stability of the HF response over the gain range is of
particular value in those VGA applications where it is essential
to maintain accurate gain law-conformance at high frequencies.
An external capacitor at pin OFST sets the high-pass corner of an
offset reduction loop, whose frequency may be as low as 5 Hz.
When this pin is grounded, the signal path becomes dc-coupled.
When used to drive an ADC, an external common-mode control
voltage at pin CNTR can be driven to within 0.5 V of either
ground or VS to accommodate a wide variety of requirements. By
default, the two outputs are positioned at the mid point of the
supply, VS/2. Other features, such as two levels of power-down
(fully off and a hibernate mode), further extend the practical
value of this exceptionally versatile VGA.
Using VMAG, the basic 0 dB to 50 dB range can be repositioned to any
value from 20 dB higher (that is, 20 dB to 70 dB) to at least 30 dB
lower (that is, –30 dB to +20 dB) to suit the application, providing
an unprecedented gain range of over 100 dB. A unique aspect of
the AD8330 is that its bandwidth and pulse response are essentially
constant for all gains, not only over the basic 50 dB linear-in-dB
*Protected by U.S.Patent No. 5,969,657; other patents pending.
REV. A
The AD8330 is available in a 16-lead LFCSP and 16-lead QSOP
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, norforanyinfringementsofpatentsorotherrightsofthirdpartiesthat
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective companies.
packages and is specified for operation from –40°C to +85°C.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
Fax: 781/326-8703
www.analog.com
© 2003 Analog Devices, Inc. All rights reserved.
(VS = 5 V, TA = 25ꢂC, CL = 12 pF on OPHI and OPLO, RL = 0/C, VDBS = 0.75 V, VMODE = HI,
AD8330–SPECIFICATIONS VMAG = 0/C, VOFST = 0 V, Differential Operation, unless otherwise noted.)
Parameter
Conditions
Min
Typ
Max
Unit
INPUT INTERFACE
Full-Scale Input
Pins INHI, INLO
V
DBS = 0 V, Differential Drive
1.4
4.5
800
2
6.3
1K
4
V
VDBS = 1.5 V
Pin-to-Pin
Either Pin to COMM
f = 1 MHz, VDBS = 1.5 V;
Inputs AC-shorted
mV
Ω
pF
Input Resistance
Input Capacitance
Voltage Noise Spectral Density
1.2K
5
nV/√Hz
Common-Mode Voltage Level
Input Offset
3.0
1
2
V
Pin OFST Connected to COMM
mV rms
µV/°C
V
dB
dB
Drift
Permissible CM Range1
Common-Mode AC Rejection
0
VS
f = 1 MHz, 0.1 V rms
f = 50 MHz
–60
–55
OUTPUT INTERFACE
Small Signal –3dB Bandwidth
Peak Slew Rate
Pins OPHI, OPLO
0 V < VDBS < 1.5 V
VDBS = 0
150
1500
2
4.5
2.5
62
150
–62
–53
MHz
V/µs
V
V
V
Peak-to-Peak Output Swing
1.8
4
2.4
2.2
2.6
VMAG ≥ 2 V (Peaks are Supply Limited)
Pin CNTR O/C
f = 1 MHz, VDBS = 0
Pin-to-Pin
VOUT = 1 V p-p, f = 10 MHz, RL = 1 kΩ
VOUT = 1 V p-p, f = 10 MHz, RL = 1 kΩ
Common-Mode Voltage
Voltage Noise Spectral Density
Differential Output Impedance
HD22
nV/√Hz
120
180
Ω
dBc
dBc
HD32
OUTPUT OFFSET CONTROL
AC-Coupled Offset
Pin OFST
CHPF on Pin OFST (0 V< VDBS < 1.5 V)
HPF = 3.3 nF, from OFST
10
100
mV rms
kHz
High-Pass Corner Frequency
C
to CNTR (Scales as 1/CHPF
)
COMMON-MODE CONTROL
Usable Voltage Range
Input Resistance
Pin CNTR
0.5
4.5
V
kΩ
From Pin CNTR to VS/2
4
DECIBEL GAIN CONTROL
Normal Voltage Range
Elevated Range
Pins VDBS, CMGN, MODE
CMGN Connected to COMM
CMGN O/C (VCMGN Rises to 0.2 V)
Mode HIGH or LOW
0 to 1.5
0.2 to 1.7
30
0.1
0.5
100
100
250
V
V
Gain Scaling
27
–0.35
–2
33
+0.35
+2
mV/dB
dB
dB
nA
MΩ
ns
Gain Linearity Error
Absolute Gain Error
Bias Current
Incremental Resistance
Gain Settling Time to 0.5 dB error
0.3 V ≤ VDBS ≤ 1.2 V
V
DBS = 0
Flows out of pin VDBS
VDBS Stepped from 0.05 V–1.45 V
or 1.45 V–0.05 V
Mode Up/Down
Pin MODE
Mode Up Logic Level
Mode Down Logic Level
Gain Increases with VDBS, MODE = O/C 1.5
Gain Decreases with VDBS
V
V
0.5
4.2
LINEAR GAIN INTERFACE
Pins VMAG, CMGN
Peak Output Scaling, Gain vs. VMAG See Circuit Description Section
Gain Multiplication Factor vs. VMAG Gain is Nominal when VMAG = 0.5 V
Usable Input Range
3.8
4.0
ϫ2
V/V
0
5
V
Default Voltage
Incremental Resistance
Bandwidth
VMAG O/C
0.48
0.5
4
150
0.52
V
kΩ
MHz
For VMAG ≥ 0.1 V
–2–
REV. A
AD8330
Parameter
Conditions
Min
Typ
Max
Unit
CHIP ENABLE
Pin ENBL
Logic Voltage for Full Shutdown
Logic Voltage for Hibernate Mode
Logic Voltage for Full Operation
Current in Full Shutdown
Current in Hibernate Mode
Minimum Time Delay3
0.5
1.7
V
V
V
mA
mA
ms
Output Pins Remain at CNTR
1.3
2.3
1.5
20
1.5
1.7
100
POWER SUPPLY
Pins VPSI, VPOS, VPSO,
COMM, CMOP
Supply Voltage
Quiescent Current
2.7
6
27
V
mA
VDBS = 0.75 V
20
NOTES
1The use of an input common-mode voltage significantly different than the internally set value is not recommended due to its effect on noise performance.
See Figure 13.
2See Typical Performance Characteristics for more detailed information on distortion in a variety of operating conditions.
3For minimum sized coupling capacitors.
REV. A
–3–
AD8330
ABSOLUTE MAXIMUM RATINGS1
Operating Temperature Range . . . . . . . . . . . . –40∞C to +85∞C
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V
Power Dissipation
Lead Temperature (Soldering 60 sec) . . . . . . . . . . . . . . 300∞C
NOTES
1Stresses above those listed under Absolute Maximum Ratings may cause
permanent damage to the device. This is a stress rating only and functional
operation of the device at these or any other conditions above those indicated
in the operational section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
2Four-Layer JEDEC Board (252P).
RQ Package2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0.62 W
CP Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .1.67 W
Input Voltage at Any Pin . . . . . . . . . . . . . . . . . . . VS + 200 mV
Storage Temperature . . . . . . . . . . . . . . . . . . . –65∞C to +105∞C
ORDERING GUIDE
Temperature Range Package Description
Model
Package Outline
AD8330ACP
–40ºC to +85ºC
–40ºC to +85ºC
–40ºC to +85ºC
–40ºC to +85ºC
–40ºC to +85ºC
Chip Scale Package
Chip Scale Package
Chip Scale Package
Thin Shrink SO
Thin Shrink SO
Thin Shrink SO
LFCSP
LFCSP
LFCSP
QSOP
QSOP
QSOP
AD8330ACP-REEL
AD8330ACP-REEL7
AD8330ARQ
AD8330ARQ-REEL
AD8330ARQ-REEL7 –40ºC to +85ºC
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although the
AD8330 features proprietary ESD protection circuitry, permanent damage may occur on devices
subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.
–4–
REV. A
AD8330
PIN CONFIGURATIONS
16-Lead LFCSP
16-Lead QSOP
OFST
ENBL
VPSI
1
2
3
4
5
6
7
8
16 VPOS
15 CNTR
AD8330 14 VPSO
TOP VIEW 13 OPHI
16
15
14
13
INHI
VSPI
INHI
1
2
3
4
12 VPSO
11 OPHI
10 OPLO
(Not to Scale)
INLO
12 OPLO
AD8330
TOP VIEW
MODE
VDBS
CMGN
11 CMOP
10 VMAG
INLO
MODE
(Not to Scale)
9
COMM
9
CMOP
5
6
7
8
PIN FUNCTION DESCRIPTIONS
Pin
16-Lead LFCSP
16-Lead QSOP
Pin
No. Mnemonic Description
No. Mnemonic Description
1
2
3
4
VPSI
Positive Supply for Input Stages
1
2
3
4
5
6
OFST
ENBL
VPSI
Used in Offset Control Modes
INHI
Differential Signal Input, Positive Polarity
Differential Signal Input, Negative Polarity
Power Enable, Active High
INLO
MODE
Positive Supply for Input Stages
Differential Signal Input, Positive Polarity
Differential Signal Input, Negative Polarity
Logic Input: Selects Gain Slope. High =
Gain Up versus VDBS
INHI
INLO
MODE
5
VDBS
Input for Linear-in-dB Gain Control
Voltage, VDBS
Logic Input: Selects Gain Slope. High =
Gain Up versus VDBS
6
7
CMGN
COMM
Common Baseline for Gain Control Interfaces
7
VDBS
Input for Linear-in-dB Gain Control
Voltage, VDBS
Ground for Input and Gain Control Bias
Circuitry
8
9
CMGN
COMM
Common Baseline for Gain Control Interfaces
8
9
VMAG
CMOP
OPLO
OPHI
VPSO
CNTR
VPOS
OFST
ENBL
Input for Gain/Amplitude Control, VMAG
Ground for Output Stages
Ground for Input and Gain Control Bias
Circuitry
10
11
12
13
14
15
16
Differential Signal Output, Negative Polarity
Differential Signal Output, Positive Polarity
Positive Supply for Output Stages
Common-Mode Output Voltage Control
Positive Supply for Inner Stages
Used in Offset Control Modes
10
11
12
13
14
15
16
VMAG
CMOP
OPLO
OPHI
Input for Gain/Amplitude Control, VMAG
Ground for Output Stages
Differential Signal Output, Negative Polarity
Differential Signal Output, Positive Polarity
Positive Supply for Output Stages
Common-Mode Output Voltage Control
Positive Supply for Inner Stages
VPSO
CNTR
VPOS
Power Enable, Active High
REV. A
–5–
AD8330–Typical Performance Characteristics
VS = 5 V, TA = 25ꢂC, CL = 12 pF, VDBS = 0.75 V, VMODE = High (or O/C) VMAG = O/C, RL = O/C, VOFST = 0, Differential Operation, unless otherwise stated.
50
45
40
35
30
25
20
15
10
5
2.0
1.5
NORMALIZED @V
= 0.75V
DBS
LO MODE
HI MODE
1.0
50MHz
100MHz
0.5
10, 50MHz
0
–0.5
–1.0
–1.5
–2.0
1MHz
10MHz
1MHz
100MHz
0
0
0.25
0.50
0.75
–V
1.00
1.25
1.50
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
V
V
–V
DBS
DBS
TPC 1. Gain vs. VDBS
TPC 4. Gain Error vs. VDBS at Various Frequencies
10
9
8
7
6
5
4
3
2
1
0
20
2340 UNITS
MODE = LO
15
10
5
0 –30.6–30.5–30.4–30.3–30.2–30.1–30.0–29.9–29.8–29.7–29.6–29.5–29.4–29.3–29.2–29.1–29.0
20
MODE = HI
15
10
5
0
29.1 29.2 29.3 29.4 29.5 29.6 29.7 29.8 29.9 30.0 30.1 30.2 30.3 30.4 30.5 30.6
0
1
2
3
4
5
GAIN SCALING – mV/dB
V
–V
MAG
TPC 2. Linear Gain Multiplication Factor vs. VMAG
TPC 5. Gain Slope Histogram
60
1.0
0.8
0.6
0.4
V
= 1.5V
DBS
50
40
1.2V
0.9V
0.6V
30
20
0.2
0.3V
0V
T = –40ꢂC
10
0
0
–0.2
–10
–20
–30
–40
–50
T = +85ꢂC
–0.4
T = +25ꢂC
–0.6
–0.8
–1.0
100k
1M
10M
FREQUENCY – Hz
100M
500M
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
V
–V
DBS
TPC 3. Gain Linearity Error Normalized at 25°C vs.
DBS, at Three Temperatures, f = 1 MHz
TPC 6. Frequency Response in 10 dB Steps for
Various Values of VDBS
V
–6–
REV. A
AD8330
50
40
25
20
15
10
5
4.8V
1048 UNITS
ENABLE MODE
1.52V
30
.48V
.15V
20
10
.048V
.015V
0
–10
–20
–30
–40
0
100k
1M
10M
FREQUENCY – Hz
100M
500M
DIFFERENTIAL OFFSET – mV
TPC 7. Frequency Response for Various Values of VMAG
TPC 10. Differential Input Offset Histogram
10
10
0
V
= 0.1V
DBS
8
6
4
2
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
100k
1M
10M
100M
300M
100k
1M
10M
FREQUENCY – Hz
100M
FREQUENCY – Hz
TPC 8. Group Delay vs. Frequency
TPC 11. Output Balance Error vs. Frequency for
a Representative Part
0
200
190
180
170
160
150
140
130
120
110
100
–1
–2
–3
–4
–5
–6
–7
T = –40ꢂC
T = +25ꢂC
T = +85ꢂC
1.2 1.4
0
0.2
0.4
0.6
0.8
1.0
1.6
100k
1M
10M
100M
300M
V
–V
FREQUENCY – Hz
DBS
TPC 9. Differential Output Offset vs. VDBS for
Three Temperatures, for a Representative Part
TPC 12. Output Impedance vs. Frequency
REV. A
–7–
AD8330
90
80
70
60
50
40
30
20
10
0
6000
5000
4000
3000
2000
1000
0
V
= 1.5V
V
= 1.5V
DBS
DBS
OFST: ENABLED
DISABLED
f = 1MHz
V
= .75V
DBS
V
= 0V
DBS
–10
50k 100k
1M
FREQUENCY – Hz
10M
100M
0
0.5
1.0
1.5
2.0
2.5
V
–V
MAG
TPC 13. CMRR vs. Frequency
TPC 16. Output Referred Noise vs. VMAG
1500
1200
900
600
300
0
80
70
60
50
40
30
20
10
0
V
= 0.5V
T = +85ꢂC
T = +25ꢂC
MAG
f = 1MHz
f = 1MHz
T = +85ꢂC
T = +25ꢂC
T = –40ꢂC
T = –40ꢂC
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
FREQUENCY – Hz
V
–V
DBS
TPC 17. Input Referred Noise vs. VDBS for
Three Temperatures
TPC 14. Output Referred Noise vs. VDBS for
Three Temperatures
180
160
140
120
100
80
700
f = 1MHz
f = 1MHz
600
V
= 0.125V
MAG
500
400
300
200
100
0
V
= 0.5V
MAG
60
40
20
V
= 2V
MAG
0
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
0
0.5
1.0
1.5
2.0
2.5
V –V
DBS
V
–V
MAG
TPC 18. Input Referred Noise vs. VDBS for Three
Values of VMAG
TPC 15. Output Referred Noise vs. VMAG
–8–
REV. A
AD8330
7
6
5
4
3
2
1
0
–10
–20
–30
–40
–50
–60
–70
–80
f = 10MHz
V
= 1.5V
DBS
HD3, R = 1kꢀ
L
HD2, R = 1kꢀ
L
0
100k
0
0.3
0.6
V
0.9
–V p-p
1.2
1.5
1M
10M
FREQUENCY – Hz
100M
OUT
TPC 19. Input Referred Noise vs. Frequency
TPC 22. Harmonic Distortion vs.
OUT-DIFFERENTIAL VMAG = 0.5 V
V
0
–10
–20
–30
–40
–50
–60
–70
–80
0
V
= 0.75V
= 1V p-p
= 1kꢀ
DBS
f = 10MHz
V
OUT
–10
–20
–30
–40
–50
–60
–70
–80
R
L
HD2 AND HD3, R = 150ꢀ*
L
HD 3
HD3, R = 1kꢀ
L
HD 2
HD2, R = 1kꢀ
L
*OUTPUT AMPLITUDE HARD LIMITED
0
1
2
3
4
5
100k
1M
10M
FREQUENCY – Hz
100M
V
–V p-p
OUT
TPC 20. Harmonic Distortion vs. Frequency
TPC 23. Harmonic Distortion vs.
OUT-DIFFERENTIAL VMAG = 2.0 V
V
0
0
–10
–20
–30
–40
–50
–60
–70
V
= 0.75V
= 1V p-p
= 1kꢀ
f = 10MHz
– 1V p-p
DBS
V
V
OUT
OUT
–10
–20
–30
–40
–50
–60
–70
–80
R
– 1kꢀ
R
L
L
HD3
HD2
HD 3
HD 2
30
0
10
20
40
50
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
C
– pF
V
–V
LOAD
DBS
TPC 21. Harmonic Distortion vs. CLOAD
TPC 24. Harmonic Distortion vs. VDBS
REV. A
–9–
AD8330
10
23
20
15
10
5
33
28
23
18
13
8
f = 10MHz
f = 10MHz
0
–10
–20
–30
–40
13
3
f = 50MHz
–7
–17
–27
–37
0
–50
0
–5
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
V
–V
V
–V
DBS
DBS
TPC 25. Input Voltage 1 dBV vs. VDBS
TPC 28. Output IP3 vs. VDBS
20
10
33
23
13
3
25
20
15
10
5
38
33
28
23
18
13
8
f = 10MHz
f = 10MHz
0
f = 50MHz
–10
–20
–30
–7
–17
–27
0
–5
–10
–40
0
3
1.6
1
2
3
4
5
6
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
V
–V
V
–V
MAG
MAG
TPC 26. Output Voltage 1 dB vs. VMAG
TPC 29. Output IP3 vs. VMAG
0
1.5
1.0
V
= 0.75V
= 1V p-p
DBS
V
–10
–20
–30
–40
–50
–60
–70
–80
–90
OUT
0.5
V
= 0V
DBS
0
–0.5
–1.0
–1.5
1M
10M
100M
–50
–25
0
25
50
75
100
FREQUENCY – Hz
TIME – ns
TPC 27. IM3 Distortion vs. Frequency
TPC 30. Full-Scale Transient Response, VDBS = 0 V
–10–
REV. A
AD8330
1.5
1.0
1V
0.5
V
= 0.75V
DBS
0
–0.5
–1.0
–1.5
1V
400ns
–50
–25
0
25
50
75
100
TIME – ns
TPC 31. Full-Scale Transient Response,
VDBS = 0.75 V, f = 1 MHz, VOUT = 2 V p-p
TPC 34. VDBS Interface Response
Top: VDBS, Bottom: VOUT
1.5
1.0
0.5
2V
V
= 1.5V
DBS
0
–0.5
–1.0
–1.5
1mV
400ns
–50
–25
0
25
50
75
100
TIME – ns
TPC 32. Full-Scale Transient Response,
TPC 35. VMAG Interface Response
Top: VMAG, Bottom: VOUT
VDBS = 1.5 V, f = 1 MHz, VOUT = 2 V p-p
500mV
1V
V
= 5V
MAG
C
= 12pF
L
V
= 0.5V
MAG
C
= 54pF
L
C
= 24pF
L
V
= 0.05V
MAG
12.5ns
100mV
12.5ns
TPC 33. Transient Response vs. for Various Load
Capacitances, G = 25 dB
TPC 36. Transient Response vs. VMAG
REV. A
–11–
AD8330
26
24
22
20
18
16
14
2.00V
OUTPUT
+85ꢂC
INPUT
+25ꢂC
–40ꢂC
50mV
25ns
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
V
–V
DBS
TPC 37. Overdrive Response, VDBS = 1.5 V,
TPC 40. Supply Current vs. VDBS at Three Temperatures
VMAG = 0.5 V, 18.5 dB Overdrive
2V
3.125V
2.5V
1.875V
3.125V
2.5V
1.875V
1V
400ns
100ns
TPC 38. ENBL Interface Response. Top: VENBL
Bottom: VOUT, f = 10 MHz
;
TPC 41. CNTR Transient Response
Top: Input to CNTR; Bottom, VOUT Single Ended
–10
V
= 0.75V
DBS
–20
–30
–40
–50
–60
–70
–80
–90
–100
–110
V
PSI
V
PSO
V
POS
1M
10M
FREQUENCY – Hz
100M
200M
TPC 39. PSRR vs. Frequency
–12–
REV. A
AD8330
CIRCUIT DESCRIPTION
ENBL
VPSI
OFST
BIAS AND
VPOS
CNTR
Many monolithic variable-gain amplifiers use techniques that share
common principles that are broadly classified as translinear, a
term referring to circuit cells whose functions depend directly on
the very predictable properties of bipolar junction transistors,
notably the linear dependence of their transconductance on collec-
tor current. Since the discovery of these cells in 1967, and their
commercial exploitation in products developed during the early 1970s,
accurate wide bandwidth analog multipliers, dividers, and variable-
gain amplifiers have invariably employed translinear principles.
VPSO
CM MODE AND
OFFSET CONTROL
V
REF
INHI
OPHI
AD8330
OUTPUT
STAGES
INLO
VGA CORE
OPLO
CMOP
While these techniques are well understood, the realization of a
high performance variable-gain amplifier (VGA) requires special
technologies and attention to many subtle details in its design.
The AD8330 is fabricated on a proprietary silicon-on-insulator,
complementary bipolar IC process and draws on decades of
experience in developing many leading-edge products using trans-
linear principles to provide an unprecedented level of versatility.
OUTPUT
CONTROL
MODE
GAIN INTERFACE
CMGN
VDBS
COMM
VMAG
Figure 3. Block Schematic of the AD8330
Overall Structure
Figure 3 shows a block schematic of the AD8330 in which the key
sections are located. More detailed discussions of its structure
and features are provided later; this figure provides a general
overview of its capabilities.
Figure 2 shows a basic representative cell comprising just four
transistors. This, or a very closely related form, is at the heart of most
translinear multipliers, dividers, and VGAs. The key concepts
are as follows: First, the ratio of the currents in the left-hand and
right-hand pairs of transistors are identical; this is represented
by the modulation factor, x, which may have values between –1
and +1. Second, the input signal is arranged to modulate the fixed
tail current ID to cause the variable value of x introduced in the
left-hand pair to be replicated in the right-hand pair, and thus
generate the output by modulating its nominally fixed tail current
IN. Third, the current-gain of this cell is very exactly G = IN/ID
over many decades of variable bias current. In practice, the
realization of the full potential of this circuit involves many other
factors, but these three elementary ideas remain essential.
The VGA core contains a much elaborated version of the cell shown
in Figure 2. The current called ID is controlled exponentially
(linear in decibels) through the decibel gain interface at the pin
VDBS and its local common CMGN. The gain span (that is,
the decibel difference between maximum and minimum values)
provided by this control function is slightly more than 50 dB.
The absolute gain from input to output is a function of source
and load impedance and also depends on the voltage on a second
gain-control pin, VMAG, as will be explained in a moment.
By varying IN, the overall function is that of a two-quadrant
analog multiplier, exhibiting a linear relationship to both the signal
modulation factor x and this numerator current. On the other
hand, by varying ID, a two-quadrant analog divider is realized,
having a hyperbolic gain function with respect to the input
factor x, controlled by this denominator current. The AD8330
exploits both modes of operation. However, since a hyperbolic
gain function is generally of less value than one in which the
decibel gain is a linear function of a control input, a special interface
is included to provide either increasing or decreasing exponential
control of ID.
Normal Operating Conditions
To minimize confusion, we define these normal operating condi-
tions: the input pins are voltage driven (the source impedance is
assumed to be zero); the output pins are open circuited (the load
impedance is assumed to be infinite); pin VMAG is unconnected,
which sets up the output bias current (IN in the four-transistor gain
cell) to its nominal value; pin CMGN is grounded; and MODE
is either tied to a logic high or left unconnected, to set the UP
gain mode. The effects of other operating conditions can then be
considered separately.
Throughout this data sheet, the end-to-end voltage gain for the
normal operating conditions will be referred to as the Basic
Gain. Under these conditions, it runs from 0 dB when VDBS = 0
(where this voltage is more exactly measured with reference to
pin CMGN, which may not necessarily be tied to ground) up to
50 dB for VDBS = 1.5 V. The gain does not “fold-over” when
the VDBS pin is driven below ground or above its nominal full-
scale value.
OUTPUT IS xl
INPUT IS xl
G = I /I
N
N
D
D
LOOP
AMPLIFIER
(1–x) I
2
(1–x) I
2
(1–x) I
2
(1+x) I
2
D
D
N
N
+
–
Q1
Q4
Q2
Q3
The input is accepted at the differential port INHI/INLO. These
pins are internally biased to roughly the midpoint of the supply
VS (it is actually ~2.75 V for VS = 5 V, VDBS = 0, and 1.5 V for VS
= 3 V), but the AD8330 is able to accept a forced common-
mode value, from zero to VS, with certain limitations. This interface
provides good common-mode rejection up to high frequencies
(see TPC 13) and thus can be driven in either a single-sided or
differential manner. However, operation using a differential drive
is preferable, and this is assumed in the specifications, unless
otherwise stated.
DENOMINATOR
BIAS CURRENT
NUMERATOR
BIAS CURRENT
I
I
N
D
Figure 2. The Basic Core of the AD8330
REV. A
–13–
AD8330
The pin-to-pin input resistance is specified as 950 Ω 20%.
The driving-point impedance of the signal source may range
from zero up to values considerably in excess of this resis-
tance, with a corresponding variation in noise figure (see
Figure 10). In most cases, the input will be coupled via two
capacitors, chosen to provide adequate low frequency trans-
mission. This results in the minimum input noise, which is
increased when some other common-mode voltage is forced
onto these pins, as explained later. The short circuit, input-
referred noise at maximum gain is approximately 5 nV/√Hz.
Linear-in-dB Gain Control (VDBS)
A gain control law that is linear in decibels is frequently claimed
for VGAs based more loosely on these principles. However, closer
inspection reveals that their conformance to this ideal gain func-
tion is poor, usually only an approximation over part of the gain
range. Furthermore, the calibration (so many decibels per volt)
is invariably left unspecified, and the resulting gain often varies
wildly with temperature. All Analog Devices VGAs featuring a
linear-in-dB gain law, such as the X-AMP™ family, provide exact,
constant gain scaling over the fully specified gain range, and the
deviation from the ideal response is within a small fraction of a dB.
For the AD8330, the scaling of both its gain interfaces is substan-
tially independent of process, supply voltage, or temperature.
The Basic Gain, GB, is simply:
The output pins OPHI/OPLO operate at a common-mode
voltage at the midpoint of the supply, VS/2, within a few millivolts.
This ensures that an analog-to-digital converter (ADC) attached
to these outputs operates within the often narrow range permit-
ted by their design. When a common-mode voltage other than VS/2
is required at this interface, it can easily be forced by applying an
externally provided voltage to the output centering pin, CNTR.
This voltage may run from zero to the full supply, though it
must be noted that the use of such extreme values would
leave only a small range for the differential output signal swing.
VDBS
30 mV
GB dB =
(
)
(1)
where VDBS is in volts. Alternatively, this can be expressed as a
numerical gain magnitude:
VDBS
GBN = 100.6 V
(2)
The differential impedance measured between OPHI and
OPLO is 150 Ω 20%. It follows that both the gain and the
full-scale voltage swing will depend on the load impedance;
both are nominally halved when this is also 150 Ω. A fixed-
impedance output interface, rather than an op amp style
voltage-mode output, is preferable in high speed applications
since the effects of complex reactive loads on the gain and
phase can be better controlled. The top end of the AD8330’s
ac response is optimally flat for a 12 pF load on each pin, but
this is not critical and the system will remain stable for any
value of load capacitance including zero.
As discussed later, the gain may be increased or decreased by
changing the voltage VMAG applied to the VMAG pin. The internally
set default value of 500 mV is derived from the same band gap
reference that determines the decibel scaling. The tolerance on this
voltage, and mismatches in certain on-chip resistors, cause small
gain errors (see Specifications). While not all applications of VGAs
demand accurate gain calibration, there are many situations in which
it will be a valuable asset, for example, in reducing design tolerances.
Figure 4 shows the core circuit in somewhat more detail. The
range and scaling of VDBS is independent of the supply voltage,
and the gain-control pin, VDBS, presents a high incremental input
resistance (~100 MΩ) with a low bias current (~100 nA), making
the AD8330 easy to drive from a variety of gain-control sources.
Another useful feature of this VGA in connection with the driving
of an ADC is that the peak output magnitude can be precisely
controlled by the voltage on pin VMAG. Usually, this voltage is
internally preset to 500 mV, and the peak differential, unloaded
output swing is 2 V 3%. However, any voltage from zero to
at least 5 V can be applied to this pin to alter the peak output in
an exactly proportional way. Since either output pin can swing
“rail to rail,” which in practice means down to at least 0.35 V
and to within the same voltage below the supply, the peak-to-peak
output between these pins can be as high as 10 V using VS = 6 V.
Inversion of the Gain Slope
The AD8330 supports many new features that further extend
the versatility of this VGA in wide bandwidth, gain-control sys-
tems. For example, the logic pin MODE allows the slope of the
gain function to be inverted, so that the basic gain starts at +50 dB
for a gain voltage VDBS of zero and runs down to 0 dB when this
voltage is at its maximum specified value of 1.5 V. The basic
forms of these two gain control modes are shown in Figure 5.
CM MODE
FEEDBACK
VPSI
VPSO
50
MODE PIN
MODE PIN
LOW, GAIN
DECREASES
HIGH, GAIN
INCREASES
TRANSIMPEDANCE
OUTPUT STAGE
WITH V
WITH V
40
30
20
10
0
DBS
DBS
500ꢀ
500ꢀ
INHI
OPHI
R
=
150ꢀ
OPLO
ꢄV = 0
ꢄV = 0
OUT
INLO
O/P CM-MODE
NORMALLY
AT V /2
P
CNTR
LINEAR-IN-dB
INTERFACE
MAGNITUDE
INTERFACE
100ꢃA
VMAG
MODE
VDBS
V
0
0.25
0.50
0.75
V
1.0
1.25
1.50
MAG
V
DBS
12.65ꢃA–4mA OR
4mA–12.65ꢃA
5kꢀ
COMM
COMM
Figure 5. The Two Gain Directions of the AD8330
Figure 4. Schematic of Key Components
–14–
REV. A
AD8330
V
IN
Gain Magnitude Control (VMAG
)
0.10
0.05
In addition to the basic linear-in-dB control, two more gain-
control features are provided. The voltage applied to pin VMAG
provides accurate linear-in-magnitude gain control with a very
rapid response. The bandwidth of this interface is >100 MHz.
When this pin is unconnected, VMAG assumes its default value
of 500 mV (see Figure 4) to set up the basic 0 dB–50 dB range.
But any voltage from ~15 mV to 5 V may be applied to either
lower the gain by up to 30 dB or to raise it by 20 dB. The com-
bined gain span is thus 100 dB, that is, the 50 dB Basic Gain
span provided by VDBS plus a 60 dB linear-in-magnitude span
provided by VMAG. The latter modifies the basic numerical gain
GBN to generate a total gain, expressed here in magnitude terms:
VMAG
–0.05
–0.05
–0.10
1.2
V
MAG
1.0
0.8
0.6
0.4
0.2
0
V
OUT
2.5
2.0
1.5
1.0
0.5
0
–0.5
–1.0
–1.5
–2.0
GT = GBN
(3)
–400
–300
–200
–100
0
100
200
300
0.5 V
TIME – ns
Using this to calculate the output voltage, we can write:
Figure 7. Using VMAG in Modulation Mode
(4)
VOUT = 2GBN VIN VMAG
Another gain-related feature allows both of the gain control ranges
to be accurately raised by 200 mV. To enable this offset, open-
circuit Pin 6 (CMGN) and add a 0.1 µF capacitor to ground. In
use, the nominal range for VDBS now extends from 0.2 V to 1.7 V
and VMAG from 0.2 V to 5.2 V. These specifications apply for any
supply voltage. This allows the use of DACs whose output range
does not include ground, as sources for the gain control function(s).
from which it is apparent that the AD8330 implements a linear,
two-quadrant multiplier with a bipolar VIN and a unipolar VMAG
.
Since the AD8330 is a dc-coupled system (the management of
dc offsets at high gains is discussed later), it may be used in many
applications where a wideband two-quadrant multiplier function
is required, from dc up to about 100 MHz from either input
(VIN or VMAG).
Note that the 200 mV that appears on this pin will affect the
response to an externally-applied VMAG, but when pin VMAG is
unconnected, the internally set default value of 0.5 V still applies.
Furthermore, the pin CMGN can, if desired, be driven by a user
supplied voltage to reposition the baseline for VDBS (or for an
externally applied VMAG) to any other voltage up to 500 mV. In
all cases, the gain scaling, its law conformance, and temperature
stability are unaffected.
As VMAG is varied, so also is the peak output magnitude, up to a
point where this is limited by the absolute output limit imposed
by the supply voltage. In the absence of the latter effect, the
peak output into an open circuited load is just:
VOUT_PK = 2VMAG
(5)
while for a load resistance of RL directly across OPHI and OPLO, it is:
2VMAGRL
VOUT_PK
=
(6)
Two Classes of Variable-Gain Amplifiers
R + 150
(
)
L
It may be noted at this point that there are two broad classes of
VGA. The first type is designed to cope with a very wide range of
input amplitudes and, by virtue of its gain control function, com-
press this range down to an essentially constant output. This is
the function needed in an AGC system. Such a VGA is called an
IVGA, referring to a structure optimized to address a wide range
of input amplitudes. By contrast, an OVGA is optimized to deliver
a wide range of output values while operating with an essentially
constant input amplitude. This is the function that might be needed,
for example, in providing a variable drive to a power amplifier.
These capabilities are illustrated, first in Figure 6, where
VS = 6 V, RL = O/C, VDBS = 0 V, VIN was swept from –2.5 to
+2.5 VDC and VMAG was set to 0.25 V, 0.5 V, 1 V, and 2 V.
Except for the last value of VMAG, the peak output follows Equa-
tion 5; this exceeds the supply-limited value when VMAG = 2 V
and the peak output is 5.65 V, that is, 6 V–0.35 V. Figure 7
demonstrates the high speed multiplication capability. The
signal input is a 100 MHz, 0.1 V sine wave, VDBS is set to 0.6 V,
and VMAG is a square wave at 5 MHz alternating from 0.25 V to
1 V. The output is ideally a sine wave switching in amplitude
It will be apparent from the foregoing that the AD8330 is both an
IVGA and an OVGA in the one package. This is an unusual and
possibly confusing degree of versatility for a VGA; consequently we
will generally discuss these two distinct control functions at separate
points throughout this data sheet, in explaining the operation and
applications of this product. It is nevertheless useful to briefly
demonstrate the capabilities of these features when used together.
between 0.5 V and 2 V.
8
V
= 2V
1V
MAG
6
4
0.5V
2
0.25V
0
–2
–4
–6
–8
–3
–2
–1
0
1
2
3
V
–V
MAG
Figure 6. Effect of VMAG on Gain and Peak Output
REV. A
–15–
AD8330
Amplitude/Phase Response
The ac response of the AD8330 is remarkably consistent not only
over the full 50 dB of its basic gain range, but also with changes
of gain due to alteration of VMAG, as demonstrated in Figure 8.
This is an overlay of two sets of results: first with a very low VMAG
of 16 mV, which reduces the overall gain by 30 dB [20 ϫ log10
(500 mV/16 mV)]; second, with VMAG = 5 V, which increases the
gain by 20 dB = 20 ϫ log10 (5 V/0.5 V).
80
60
40
20
0
–20
–40
–60
100k
10k
1k
100
10
1
90
70
50
30
10
–10
–30
–50
100k
0
.001
.01
.1
1
10
V
–V
GAIN
1M
10M
100M
300M
Figure 9. Gain Control Function and Input Referred
Noise Spectral Density over a 120 dB Range
G = +70dB
–50
–100
–150
–200
–250
–300
–350
Noise, Input Capacity and Dynamic Range
The design of variable-gain amplifiers invariably incurs some
compromises in noise performance. However, the structure of
the AD8330 is such that this penalty is minimal. Examination of
the simplified schematic (Figure 4) shows that the input voltage
is converted to current-mode form by the two 500 Ω resistors at
pins INHI and INLO, whose combined Johnson noise contributes
4.08 nV/√Hz. The total input noise at full gain, when driven from a
low impedance source, is typically 5 nV/√Hz after accounting for
the voltage and current noise contributions of the loop amplifier.
For a 200 kHz channel bandwidth, this amounts to 2.24 µV rms.
The peak input at full gain is 6.4 mV, or +4.5 mV rms for a
sine wave signal. The signal-to-noise ratio at full input, that is, the
dynamic range, for these conditions is thus 20 log10(4.5 mV/2.24 µV),
or 66 dB. The value of VMAG has essentially no effect on the
input-referred noise, but we assume it to be 0.5 V.
G = –20dB
100M 300M
100k
1M
10M
FREQUENCY – Hz
Figure 8. AC Performance over a 100 dB Gain
Range Obtained by Using Two Values of VMAG
This 50 dB step change in gain produces the two sets of gain
curves, having a total gain span of 100 dB. It is apparent that
the amplitude and phase response are essentially independent of
the gain over this wide range, an aspect of the AD8330’s perfor-
mance potential unprecedented in any prior VGA.
It is unusual for an application to require such a wide range of
gains, of course; and as practical matter, the peak output voltage
for VMAG = 16 mV is reduced by the factor 16/500, compared to
its nominal value of 2 V, to only 64 mV. As already noted,
most applications of VGAs require that they operate in a mode
that is predominantly of either an IVGA or OVGA style, rather
than mixed modes.
Below midgain (25 dB, VDBS = 0.75 V), noise in the output
section dominates, and the total input noise is 11 nV/√Hz, or
4.9 µV rms in a 200 kHz bandwidth, while the peak input is
78 mV rms. Thus, the dynamic range has increased to 84 dB. At
minimum gain, the input noise is up to 120 nV/√Hz, or 53.7 mV rms
in the assumed 200 kHz bandwidth, while the input capacity is
2 V, or +1.414 V rms (sine), a dynamic range of 88.4 dB. In
calculating the dynamic range for other channels bandwidths,
∆f, subtract 10 log10(∆f /200 kHz) from these illustrative values.
A system operating with a 2 MHz bandwidth, for example, will
exhibit dynamic range values that are uniformly 10 dB lower;
used in an audio application with a 20 kHz bandwidth, they will
be 10 dB higher.
With this limitation in mind, and simply in order to illustrate
the unusual possibilities afforded by the AD8330, it is noted
that with appropriate drive to VDBS and VMAG in tandem, the
gain span is a remarkable 120 dB, extending from –50 dB to
+70 dB, as shown in Figure 9 for operation at 1 MHz and
100 MHz. In this case, VDBS and VMAG are driven from a
common control voltage, VGAIN, which is varied from 1.2 mV
to 5 V, with 30% (1.5/5) of VGAIN applied to VDBS, and 100%
Noise figure is a misleading metric for amplifiers that are not
impedance matched at their input, which is the special condition
resulting only when both the voltage and current components of
a signal, that is, the signal power, are used at the input port. When
a source of impedance RS is terminated using a resistor of RS
(a condition that is not to be confused with matching), only one of
these components is used, either the current (as in the AD8330)
or the voltage. Then, even if the amplifier is perfect, the noise
figure cannot be better than 3 dB. The 1 kΩ internal termina-
tion resistance would result in a minimum noise figure of 3 dB for
an RS of 1 kΩ if the amplifier were noise-free. However, this is
not the case and the minimum noise figure will occur at a slightly
different value of RS (see Figure 10 and Using the AD8330).
applied to VMAG
.
The gain varies in a linear-in-dB manner with VDBS, while the
response from VMAG is linear-in-magnitude. Consequently, the
overall numerical gain is the product of these two functions:
VGAIN
GAIN =VGAIN / 0.5 V × 0.3 ×100.6 V
In rare cases where such a wide gain range might be of value, the
calibration will still be accurate and temperature stable.
(7)
–16–
REV. A
AD8330
15
14
3.2
3.1
3.0
2.9
2.8
2.7
2.6
T = +85ꢂC
T = +25ꢂC
13
12
11
10
T = –40ꢂC
9
8
7
6
5
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
10
100
1000
10k
R
–ꢀ
V
ꢅ V
S
DBS
Figure 10. Noise Figure for Source Resistance of 50 Ω to
5 kΩ, at f = 10 MHz (lower) and 100 MHz (Simulation)
Figure 12. Common-Mode Voltage at Input Pins
vs. VDBS, for VS = 5 V, T = –35°C, +25°C, and +85°C
26
24
144
V
= 0.5V
DBS
140
22
20
18
16
14
12
10
CONSTANT 1V rms
OUTPUT, BOTH CASES
136
V
= 0.6V
DBS
X-AMP WITH 40dB
132
OF GAIN AND AN
INPUT NSD
OF nV/ Hz
128
124
V
= 0.75V
DBS
8
6
4
120
116
SIMULATION
V
= 1.5V
DBS
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5
0
0.4 0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0 4.4 4.8
COMMON-MODEVOLTAGE AT INHI, INLO –V
V
– V
DBS
Figure 11. Dynamic Range in dB/ √Hz vs. VDBS (VMAG = 0.5 V,
1 V rms output) Compared with a Representative X-AMP
(Simulation)
Figure 13. Input Noise vs. Common-Mode Input
Voltage for VDBS = 0.5 V, 0.6 V, 0.75 V, and 1.5 V
Output Noise and Peak Swing
Dynamic Range
The output noise of the AD8330 is the input noise multiplied by
the overall gain, which includes any optional change to the voltage
VMAG applied to pin VMAG. The peak output swing is also
proportional to this voltage, which, at low gains and high values
of VMAG, will affect the output noise. The scaling for VDBS = 0 is
as follows:
The ratio of peak output swing, expressed in rms terms, to the
output-referred noise-spectral-density provides a measure of
dynamic range, in dB/√Hz. For a certain class of variable-gain
amplifiers, exemplified by Analog Devices’ X-AMP family, the
dynamic range is essentially independent of the gain setting,
because the peak output swing and noise are both constant. The
AD8330 provides a different dynamic-range profile, since there
is no longer a constant relationship between these two param-
eters. Figure 11 compares the dynamic range of the AD8330 to
a representative X-AMP.
VOUT_PK = 4VMAG
(8)
(9)
VNOISE_OUT = 85 + 70 V
nV / Hz
(
)
MAG
For example, using a reduced value of VMAG = 0.25 V, which
lowers all gain values by 6 dB, the peak output swing is 1 V
(differentially) and the output noise spectral density evaluates to
102.5 nV/√Hz. The peak output swing is no different at full gain,
but the noise is now
Input Common-Mode Range and Rejection Ratio
The inputs INHI and INLO should be ac-coupled in most applica-
tions, to achieve the stated noise performance. When direct coupling
is used, care must be taken in setting the dc voltage level at these
inputs, in general, and particularly when minimizing noise is
critical. This objective is complicated by the fact that the common-
mode level varies with the basic gain voltage VDBS. Figure 12
shows this relationship for a supply voltage of 5 V, for tempera-
tures of –35°C, +25°C and +85°C. Figure 13 shows the input
noise-spectral-density (RS = 0) versus the input common-mode
voltage, for VDBS = 0.5, 0.6 V, 0.75 V, and 1.5 V. It is apparent
that there is a broad range over which the noise is unaffected by
this dc level. The input CMRR is excellent (see TPC 13).
(10)
VNOISE_OUT = 0.1+ 0.32 V
µV / Hz
(
)
MAG
for RS = 0 and VDBS = 1.5 V, assuming an input noise of
5 nV/√Hz. The output noise for very small values of VMAG
(at or below 15 mV) is not precise, partly because the small input
offset associated with this interface has a large effect on the gain.
REV. A
–17–
AD8330
Offset Compensation
from CNTR to ground can lower this voltage, or one to the supply
will raise it. On the other hand, this pin may be driven by an
external voltage source to set the common-mode level, to satisfy
the needs of a following ADC, for example. Any value from 0.5 V
above ground to 0.5 V below the supply is permissible. Of
course, when using an extreme common-mode level, the avail-
able output swing will be limited, and it is recommended that a
value equal or close to the default of VCNTR = VS/2 be used. There
may be a few millivolts of offset between the applied voltage and the
actual common-mode level at the output pins.
The AD8330 includes an offset compensation feature, which is
operational in the default condition (no connection to pin OFST).
This loop introduces a high-pass filter function into the signal
path, whose –3 dB corner frequency is at:
1
fHPF
=
(11)
2π RINT CHP
(
)
where CHP is the external capacitance added from OFST to CNTR,
and RINT is an internal resistance of approximately 480 Ω, having a
maximum uncertainty of about 20%. This evaluates to:
The input common-mode voltage VCMI at pins INHI, INLO is
slaved to the output, but with a shifted value:
330µ
CHP
fHPF
≈
C
in µF
HP
(
)
(12)
(14)
VCMI = 0.757 VCNTR + 1.12 V
for VDBS = 0.75 and T = 25°C. Thus, the default value for VCMI
A small amount of peaking at this corner when using small capacitor
values can be avoided by adding a series resistor. Useful combina-
tions are CHP = 3 nF, RHP = 180 Ω , f = 100 kHz; CHP = 33 nF,
when VS = 5 V is 3.01 V (see Figure 12).
USING THE AD8330
RHP = 10 Ω, f = 10 kHz; CHP = 0.33 µF, RHP = 0 Ω, f = 1 kHz;
There are very few precautions that need to be observed in apply-
ing the AD8330 to a wide variety of circumstances. A selection
of specific applications is presented later. Here we discuss a few
general aspects of utilization.
C
HP = 3.3 µF, RHP = 0 Ω , f = 100 Hz.
The offset compensation feature can be disabled simply by ground-
ing the OFST pin. This provides a dc-coupled signal path, with
no other effects on the overall ac response. Input offsets must be
externally nulled in this mode of operation, as shown in Figure 15.
As in all high frequency circuits, careful observation of the ground
nodes associated with each function is important. Three positive
supply pins are provided. VPSI supports the input circuitry, which
may often be operating at a relatively high sensitivity; VPOS,
which supports general bias sources, needs no decoupling; VPSO
is used to bias the output stage, where decoupling may be useful
in maintaining a glitch-free output. Figure 14 shows the general
case, where VPSI and VPSO are each provided with their own
decoupling network, but this may not be needed in all cases.
Effects of Loading on Gain and AC Response
The differential output impedance RO is 150 Ω and the frequency
response of the output stage is optimized for operation with a certain
load capacitance on each output pin, OPHI and OPLO, to
ground, in combination with a load resistance RL directly across
these pins. In the absence of these capacitances, there will be a
small amount of peaking at the top extremity of the ac response.
Suitable combinations are: RL = ∞, CL = 12 pF; RL = 150 Ω,
CL = 25 pF; RL = 75 Ω, CL = 40 pF; RL = 50 Ω, CL = 50 pF.
Because of the differential nature of the signal path, power-supply
decoupling is in general much less critical than in a single-sided
amplifier, and where the minimization of board-level components
is especially crucial, it may be found that these pins need no
decoupling at all. On the other hand, when the signal source is
The gain calibration is specified for an open-circuited load, such
as the high input resistance of an ADC. When resistively loaded,
all gain values are nominally lowered as follows:
GUNLOADED RL
GLOADED
=
(13)
V
2.7V–6V
CD2
S
150 Ω + R
(
)
L
RD1
CD1
CHPF
RD2
CD3
Thus when RL = 150 Ω, the gain is reduced by 6 dB; for RL = 75 Ω,
the reduction is 9.5 dB; and for RL = 50 Ω, it is 12 dB.
ENBL
VPSI
OFST
VPOS
CNTR
VPSO
Gain Errors Due to On-Chip Resistor Tolerances
In all cases where external resistors are used, keep in mind that all
on-chip resistances, including the RO and the input resistance,
RI, are subject to variances of up to 20%, which will need to
be accounted for when calculating the gain with input and output
loading. This sensitivity can be avoided by adjusting the source
and load resistances to bear an inverse relationship as follows:
If RS = αRI then make RL = RO /α; or, if RL = αRO then make
RS = RI /α. The simplest case is when RS = 1 kΩ and RL = 150 Ω.
Here the gain is 12 dB lower than the basic value. The reduction
of peak swing at the load can be corrected by using VMAG = 1 V,
which also restores 6 dB of gain; using VMAG = 2 V restores the
full basic gain while also doubling the peak available output swing.
BIAS AND
CM MODE AND
OFFSET CONTROL
V-REF
INHI
OPHI
OUTPUT,
ꢆ2V MAX
VGA CORE
INPUT,
0V TO 2V MAX
OUTPUT
STAGES
INLO
OPLO
OUTPUT
MODE
GAIN INTERFACE
CMGN
NC
CONTROL
CMOP
VDBS
COMM
VMAG
Output (Input) Common-Mode Control
The output voltages are nominally positioned at the midpoint of
the supply, VS/2, over the range 2.7 V < VS < 6 V, and this voltage
appears at pin CNTR, which is not normally expected to be
loaded (the source resistance is ~4 kΩ). However, some circum-
stances may require a small change in this voltage, and a resistor
NC
BASIC GAIN BIAS
0V TO 1.5V
V
:
GROUND
DBS
Figure 14. Power Supply Decoupling and Basic Connections
–18– REV. A
AD8330
single-sided extra attention may be needed to the decoupling on pin
VPSI, while if the output is loaded on only one of its two output
pins, this may require care in decoupling the VPSO pin. The
general common COMM and the output stage common CMOP
are usually grounded as shown in the figure; however, the Applica-
tions section shows how a negative supply can optionally be used.
each input pin, their minimum value can be readily found from
this expression:
320µF
fHPF
CIN _CPL
=
(15)
where fHPF is the –3dB frequency expressed in hertz. Thus, for
an fHPF of 10 kHz, 33 nF capacitors would be used.
The AD8330 is enabled by taking the ENBL pin to a logical high
(or, in all cases, the supply). The “UP” gain mode is enabled
either by leaving the MODE pin unconnected or taken to a
logical HI; when the opposite gain direction is needed, this pin
should be grounded or driven to a logical low. The low-pass
corner of the offset loop is determined by the capacitor CHPF;
this is preferably tied to the CNTR pin, which in turn should be
decoupled to ground. The gain-interface common pin CMGN
is grounded, and the output magnitude control pin VMAG is left
unconnected, or may optionally be connected to a 500 mV source
for basic gain calibration.
It may occasionally be possible to avoid the use of coupling
capacitors, when the dc level of the driving source is within a certain
range, as shown in Figure 13. This range extends from 3.5 V to
4.5 V when using a 5 V supply, and at high basic gains, where the
effect of an incorrect dc level would be most troublesome, caus-
ing an increase in noise level due to internal aspects of the input
stage. For example, suppose the driver IC is an LNA having an
output topology in which its load resistors are taken to the supply,
and the output is buffered by emitter-followers. This presents a
source for the AD8330 that could readily be directly coupled.
Connections to the input and output pins are not shown in this
figure because of the many options that are available. When the
AD8330 is used to drive an ADC, pins OPHI and OPLO may be
connected directly to the differential inputs of a suitable converter,
such as an AD9214. If an adjustment is needed to this common-
mode level, it can be introduced by applying that voltage to CNTR,
or, more simply, by using a resistor from this pin to either ground
or the supply (see APPLICATIONS). This pin can also supply the
common-mode voltage to an ADC that supports such a feature.
DC-Coupled Signal Path
In many cases, where the VGA is not required to provide its lowest
noise, the full common-mode input range of zero to VS may be
used without problems, avoiding the need for any ac coupling
means. However, such direct coupling at both the input and output
will not automatically result in a fully dc-coupled signal path.
The internal offset compensation loop must also be disengaged,
by connecting the OFST pin to ground. Keep in mind that at the
maximum basic gain of 50 dB (ϫ316), every millivolt of offset at
the input, arising from whatever source, causes an output offset of
316 mV, which is an appreciable fraction of the peak output swing.
When the loads to be driven introduce a dc resistive path to
ground, coupling capacitors must be used; these should be of
sufficient value to pass the lowest frequency components of the
signal without excessive attenuation. Keep in mind that the voltage
swing on such loads will alternate both above and below ground,
requiring that the subsequent component must be able to cope
with negative signal excursions.
Since the offset correction loop is placed after the front-end
variable-gain sections of the AD8330, the most effective way of deal-
ing with such offsets is at the input pins, as shown in Figure 15.
For example, assume, for illustrative purposes, that the resistances
associated with each side of the source in a certain application
are 50 Ω. If this source has a very low (op amp) output imped-
ance, the extra resistors should be inserted, with a negligible
noise penalty and an attenuation of only 0.83 dB. The resistor
values shown provide a trim range of about 2 mV.
Gain and Swing Adjustments When Loaded
The output can also be coupled to a load via a transformer, in
which case it may be possible to achieve a higher load power by
impedance transformation. For example, using a 2:1 turns ratio,
a 50 Ω final load will present a 200 Ω load on the output. The gain
loss (relative to the basic value with no termination) will be
20 log10{(200+150)/200} or 4.86 dB, which can be restored by
raising the voltage on the VMAG pin by a factor of 104.86/20 or
ϫ1.75, from its basic value of 0.5 V to 0.875 V. This also restores
the peak swing at the 200 Ω level to 2 V, or 1 V into the 50 Ω
final load.
V
2.7V–6V
CD2
S
RD1
RD2
ENBL
VPSI
OFST
VPOS
CNTR
VPSO
CD1
CD3
BIAS AND
V-REF
CM MODE AND
OFFSET CONTROL
Whenever a stable supply voltage is available, the additional voltage
may be provided simply by adding a resistor from this pin to the
supply. The calculation is based on knowing that the internal bias
is delivered via a 5 kΩ source; since an additional 0.375 V is
needed, the current in this external resistor must be 0.375 V/5 kΩ
= 75 µA. Thus, using a 5 V supply, a resistor of 5 V–0.875 V/75 µA
= 55 kΩ would be used. Based on this example, the corrections
for other load conditions should be easy to calculate. If the effects
on gain and peak output swing due to supply variations cannot
be tolerated, VMAG must be driven by an accurate voltage.
INHI
OPHI
R
ASSUMED
S
50kꢀ
TO BE 50ꢀ
ON EACH
SIDE
OUTPUT,
ꢆ2V MAX
VGA CORE
OUTPUT
STAGES
INLO
OPLO
75kꢀ
OUTPUT
CONTROL
MODE
GAIN INTERFACE
CMGN
CMOP
VDBS
COMM
VMAG
Input Coupling
The dc common-mode voltage at the input pins varies with the
supply, the basic gain bias and temperature (see Figure 12); for
this reason, many applications will need to use coupling capacitors
from the source, which should be large enough to support the
lowest frequencies to be transmitted. Using one capacitor at
NC
BASIC GAIN BIAS
: 0V TO 1.5V
V
GROUND
DBS
Figure 15. Input Offset Nulling in a DC-Coupled System
REV. A
–19–
AD8330
90
80
70
60
50
40
30
20
10
0
Adding a dummy 75 Ω to OPLO results in Line 3: the gain is a
further 2.5 dB lower, at about 14 dB. The CM artifacts are no
longer present but there is now a small amount of peaking. If
objectionable, this may be eliminated by raising both of the
capacitors on the output pins to 25 pF, as shown in Line 4.
V
= 1.5V
DBS
OFST: ENABLED
DISABLED
V
= .75V
DBS
The gain reduction incurred both by using only one output and
by the additional effect of loading can be overcome by taking
advantage of the VMAG feature, provided primarily for just such
circumstances. Thus, to restore the basic gain in the first case
(Line 1), a 1 V source should be applied to this pin; to restore
the gain in the second case, this voltage should be raised by a
factor of ϫ1.5, to 1.5 V. In cases 3 and 4, a further factor of
ϫ1.33 is needed to make up the 2.5 dB loss, that is, VMAG should
be raised to 2 V. With the restoration of gain, the peak output
swing at the load is likewise restored to 2 V.
V
= 0V
DBS
–10
50k 100k
1M
FREQUENCY – Hz
10M
100M
Pulse Operation
Figure 16. Input CMRR vs. Frequency for Various
Values of VDBS
When using the AD8330 in applications where its transient response
is of greater interest and the outputs are conveyed to their load via
coaxial cables, the added capacitances may be slightly different in
value, and may be placed either at the sending or load end of the
cables, or divided between these nodes. Figure 18 shows an illustra-
tive example in which dual 1 meter 75 Ω cables are driven through
dc-blocking capacitors and independently terminated at ground level.
Using Single-Sided Sources and Loads
Where the source provides a single-sided output, either INHI or
INLO may be used for the input, with of course a polarity change
when using INLO. The unused pin must be connected either
through a capacitor to ground, or a dc bias point that corresponds
closely to the dc level on the active signal pin. The input CMRR
over the full frequency range is illustrated in Figure 16. In some
cases, an additional element such as a SAW filter (having a single-
sided-balanced configuration) or a flux-coupled transformer may
be interposed. Where this element must be terminated in the
correct impedance, other than 1 kΩ, it will be necessary to add
either shunt or series resistors at this interface.
Because of the considerable variation between applications, only
general recommendations can be made with regard to minimizing
pulse overshoot and droop. The former can be optimized by adding
small load capacitances, if necessary; the latter require the use
of sufficiently large capacitors C1.
V
2.7V–6V
S
30
CD2
RD2
CD3
LINE 1
LINE 3
20
10
ENBL
VPSI
OFST
VPOS
CNTR
0
LINE 4
–10
VPSO
LINE 2
BIAS AND
CM MODE AND
OFFSET CONTROL
–20
V-REF
–30
0
RL1
C1
C1
LINE 2
INHI
OPHI
–100
LINE 3
CL1
–200
VGA CORE
OUTPUT
STAGES
LINE 4
LINE 1
–300
–400
INLO
OPLO
CMOP
–500
–600
CL2
RL2
1M
10M
FREQUENCY – Hz
100M
500M
OUTPUT
CONTROL
MODE
GAIN INTERFACE
CMGN
Figure 17. AC Gain and Phase for Various Loading Conditions
VDBS
COMM
VMAG
NC
When driving a single-sided load, either OPHI or OPLO may be
used. These outputs are very symmetric, so the only effect of
this choice is to select the desired polarity. However, when the
frequency range of interest extends to the upper limits of the
AD8330, a dummy resistor of the same value should be attached
to the unused output. Figure 17 illustrates the ac gain and phase
response for various loads and VDBS = 0.75 V. Line 1 shows the
unloaded (CL = 12 pF) case for reference; the gain is 6 dB lower
(20 dB) using just the single-sided output. Adding a 75 Ω load
just from OPHI to an ac ground results in Line 2. The gain is
now a factor of ϫ1.5 or 3.54 dB lower, but artifacts of the output
common-mode control loop now appear in both the magnitude
and phase response.
Figure 18. Driving Dual Cables with Grounded Loads
–20–
REV. A
AD8330
Table I. Preserving Absolute Gain
1.2
1.0
0.8
0.6
0.4
0.2
0
–0.2
0.2
Uncorrected LossVMAG Required
RS(Ω) RL(Ω)
Factor
dB
to Correct Loss
0
–0.2
–0.4
–0.6
–0.8
–1.0
–1.2
1.2
1.0
0.8
0.6
0.4
0.2
0
–0.2
0.2
0
–0.2
–0.4
–0.6
–0.8
–1.0
–1.2
10
15
20
30
50
15k
10k
7.5k
5.0k
3.0k
2.0k
1.5k
1.0k
750
500
300
200
150
100
75
0.980
0.971
0.961
0.943
0.907
0.865
0.826
0.756
0.694
0.592
0.444
0.327
0.250
0.160
0.111
0.17
0.26
0.34
0.51
0.85
1.26
1.66
2.43
3.17
4.56
7.04
9.72
12.0
15.9
19.1
0.510
0.515
0.520
0.530
0.551
0.578
0.605
0.661
0.720
0.845
1.125
1.531
2.000
3.125
4.500
75
100
150
200
300
500
750
1k
0
5ns
10ns
15ns
20ns
25ns
Figure 19. Typical Pulse Responses for Figure 18
Figure 19 shows typical results for VDBS = 0.24 V, a square wave
input amplitude of 450 mV (the actual combination is not impor-
tant) and a rise time of 2 ns. VMAG raised to 2.0 V is used. In the
upper waveforms the load capacitors are both zero, and a small
amount of overshoot is visible; with 40 pF the response is cleaner.
A shunt capacitance of 20 pF from OPHI to OPLO will have a
similar effect. Coupling capacitors for this demonstration are
sufficiently large to prevent any visible droop over this time scale.
The outputs at the load side will eventually assume a mean value
of zero, with negative and positive excursions depending on the
duty-cycle.
1.5k
2k
Calculation of Noise Figure
The AD8330 noise is a consequence of its intrinsic voltage-noise-
spectral-density (ENSD) and the current-noise-spectral-density
(INSD). Their combined effect generates a net input noise, VNOISE_IN
which is a function of the device’s input resistance, RI, nominally
1 kΩ, and the differential source resistance, RS, as follows:
,
2
2
2
VNOISE_IN
=
ENSD + INSD R + R
(16)
(
)
I
S
{
}
The bandwidth from pin VMAG to these outputs is somewhat
higher than that from the normal input pins. Thus when this pin
is used to rapidly modulate the primary signal, some further
experimentation with response optimization may be required. In
general, the AD8330 is very tolerant of a wide range of loading
conditions.
Note that we assume purely resistive source and input impedances
as a concession to simplicity. A more thorough treatment of noise
mechanisms, for the case where the source is reactive, is beyond the
scope of these brief notes. Also note that VNOISE_IN is the voltage-
noise-spectral-density appearing across the differential input pins,
INHI, INLO. In preparing for the calculation of noise figure,
we will define VSIG as the open-circuit signal voltage across the
source and VIN as the differential input to the AD8330. The
relationship is simply
Preserving Absolute Gain
Although the AD8330 is not laser-trimmed, its absolute gain cali-
bration, being based mainly on ratios, is very good. Full details can
be found in the Specifications and in the typical performance
curves. Nevertheless, having finite input and output impedances,
the gain is necessarily dependent on the source and load condi-
tions. The loss incurred when either of these is finite causes an
error in the absolute gain, which may also be uncertain due to
the approximately 20% tolerance in the absolute value of the
input and output impedances.
VSIGRI
VIN
=
(17)
R + R
(
)
I
S
At maximum gain, ENSD is 4.1 nV/√Hz, and INSD is 3 pA/√Hz.
Thus, the short-circuit voltage noise is:
2
2
VNOISE_IN
=
4.1nV / Hz + 3pA / Hz 1 kΩ + 0 2
(
)
(
)
(
)
Often, such losses and uncertainties can be tolerated and accom-
modated by a correction to the gain control bias. On the other
hand, the error in the loss can be essentially nulled by using
appropriate modifications to either the source impedance (RS)
or the load impedance (RL), or both, in some cases by padding
them with series or shunt components.
(18)
= 5.08 nV / Hz
Next, examine the net noise when RS = RI = 1 kΩ, often incorrectly
called the “matching” condition, rather than “source impedance
termination,” which is the actual situation in this case. Repeating
the procedure:
The formulation for this correction technique was described
previously. However, to simplify its use, Table I is provided, showing
spot values for combinations of RS and RL resulting in an overall
loss that will not be dependent on sample-to-sample variations
in on-chip resistances. Furthermore, this fixed and predictable loss
can be corrected by an adjustment to VMAG, as indicated in Table 1.
2
2
VNOISE_IN
=
4.1nV / Hz + 3pA / Hz 1 kΩ + 1 kΩ 2
(
)
(
)
(
)
(19)
≈ 7.3 nV / Hz
REV. A
–21–
AD8330
The noise figure is just the decibel representation of the noise
P1 dB and V1 dB
factor, NFAC, which is commonly defined as follows:
In addition to the nonlinearities that arise within the core of the
AD8330, at moderate output levels, a further metric that is more
commonly stated for RF components that deliver appreciable
power to a load is the “1 dB compression point.” This is defined
in a very specific manner: it is that point at which, with increas-
ing output level, the power delivered to the load eventually falls
to a value that is 1 dB lower than it would be for a perfectly
linear system. (While this metric is sometimes called the “1 dB
gain-compression point,” it is important to note that this is not
the output level at which the incremental gain has fallen by 1 dB).
Signal-to-noise ratio at input
Signal-to-noise ratio at output
NFAC
=
(20)
However, this is equivalent to
Signal-to-noise ratio at the source
Signal-to-noise ratio at the input pins
NFAC
=
(21)
Let VNSD be the voltage-noise-spectral-density √kTRS due to
the source resistance. Then we have:
As was shown in Figure 6, the output of the AD8330 limits quite
abruptly, and the gain drops sharply above the clipping level.
The output power, on the other hand, using an external resistive
load, RL, continues to increase. In the most extreme case, the
waveform changes from the sinusoidal form of the test signal,
with an amplitude just below the clipping level, say, VCLIP, to a
squarewave of precisely the same amplitude. The change in
power over this range is from (VCLIP/√2)2/RL to (VCLIP)2/RL, that
is, a factor of 2, or 3 dB in power terms. It can be shown that for
an ideal limiting amplifier, the 1 dB compression point occurs
for an overdrive factor of 2 dB.
VSIG R / R + R /V
NSD
(
)
}
RI VNOISE _ IN
RS VNSD
{
I
I
S
NFAC
=
=
(22)
VIN / V
R / R + R
(
)
}
{
NOISE _ IN
S I S
using (17). Thus, using the result (19) for a source resistance of
1 kΩ, having a noise-spectral density of 4.08 nV/√Hz, we have:
1kΩ 7.3 nV / Hz
(
(
)
(
)
NFAC
=
= 1.79
(23)
1kΩ 4.08 nV / Hz
)
(
)
For example, if the AD8330 is driving a 150 Ω load and VMAG
has been set to 2 V, the peak output is nominally 4 V (as noted
above, the actual value when loaded may differ due to the mismatch
between on-chip and external resistors), or 2.83 V rms for a sine
wave output, which corresponds to a power of 53.3 mW, that is,
17.3 dBm in 150 Ω. Thus, the P1dB level, at 2 dB above clipping,
is 19.3 dBm.
Finally, converting this to decibels using:
(24)
NFIG = 10log10
N
FAC
(
)
we find the noise figure in this case to be 5.06 dB, which is some-
what lower than the value shown in Figure 10 for this operating
condition.
Noise as a Function of VDBS
While not involving power transfer, it is sometimes useful to state
the V1dB, which is the output voltage (unloaded or loaded) that
is 2 dB above clipping for a sine waveform. In the above example,
this voltage is still 2.83 V rms, which can be expressed as
9.04 dBV (0 dBV corresponds to a 1 V sine wave). Thus the
V1dB is at 11.04 dBV.
The chief consequence of lowering the basic gain using VDBS is
that the current-noise-spectral-density INSD increases with the
square root of the basic gain magnitude, GBN
:
(25)
INSD = 3 pA / Hz GBN
Thus, at the maximum basic gain of ϫ316, INSD has risen to
53.3 pA/√Hz, and if we recalculate the noise figure using the
procedures just explained, we find it has risen to 17.2 dB.
APPLICATIONS
The AD8330’s versatility, very constant ac response over a wide
range of gains, large signal dynamic range, output swing, single-
supply operation, and low power consumption will commend this
VGA to a diverse variety of applications. Only a few can be described
here, including the most basic uses and some unusual ones.
Distortion Considerations
Continuously variable-gain amplifiers invariably employ nonlinear
circuit elements; consequently it is common for their distortion to
be higher than well-designed fixed-gain amplifiers. The translinear
multiplier principles used in the AD8330 in principle yield extremely
low distortion, a result of the fundamental linearization technique
that is an inherent aspect of these circuits.
ADC Driving
The AD8330 is well-suited to driving a high speed converter.
There are now many available, but to illustrate the general fea-
tures we will use one of the least expensive, the AD9214, which
is available in three grades for operation at 65 MHz, 80 MHz,
and 105 MHz; the AD9214BRS-80 is a good complement to the
general capabilities of this VGA.
In practice, however, the effect of device mismatches and junction
resistances in the core cell, and other mechanisms in its support-
ing circuitry inevitably cause distortion, further aggravated by
other effects in the later output stages. Some of these effects are
very consistent from one sample to the next, while those due to
mismatches (causing predominantly even-order distortion compo-
nents) will be quite variable. Where the highest linearity (and also
lowest noise) is demanded, consider using one of the X-AMP
products such as the AD603 (single-channel), AD604 (dual-channel),
or AD8332 (wideband dual-channel with ultra-low noise LNAs).
–22–
REV. A
AD8330
V , 3.3V
S
3.3ꢀ
3.3ꢀ
0.1ꢃF
0.1ꢃF
OVER-
RANGE
8kꢀ
CHPF
0.1ꢃF
10ꢀ
ENBL
VPSI
OFST
VPOS
CNTR
VPSO
OR
AV
DrV
DD
DD
BIAS AND
CM MODE AND
OFFSET CONTROL
D9
D8
D7
D6
D5
D4
D3
D2
D1
D0
V-REF
PWRDN
DFS/GAIN
0.1ꢃF
A
IN
INHI
OPHI
INPUT,
2V MAX
OUTPUT
STAGES
AD9217BRS-80
VGA CORE
A
IN
INLO
OPLO
REFSENSE
REF
OUTPUT
CONTROL
MODE
GAIN INTERFACE
CMGN
NC
CMOP
AGND
DGND
CLK
VDBS
COMM
VMAG
0.1ꢃF
NC
ANALOG GROUND
GAIN BIAS,
, 0V–1.5V
CLOCK
V
DBS
DIGITAL
GROUND
Figure 20. Driving an Analog-to-Digital Converter (Preliminary)
V , 2.7V–6V
S
Figure 20 shows the connections. A 3.3 V supply is used for
both parts. The ADC requires that its input pins be positioned
at one third of the supply, or 1.1 V. Since the default output
level of the VGA is one half the supply or 1.65 V, a small correc-
tion is introduced by the 8 kΩ resistor from CNTR to ground.
The ADC specifications require that the common-mode input
be within 0.2 V of the nominal 1.1 V; variations of up to 20%
in the AD8330’s on-chip resistors will change this voltage by
only 70 mV. With the connections shown, the AD9214 is able
to receive an input of 2 V p-p; the peak output of the AD8330
can be reduced if desired by adding a resistor from VMAG to
ground. An overrange condition is signaled by a HI state on pin
OR of the AD9214. DFS/GAIN is unconnected in this example;
this produces an offset-binary output. To provide a twos comple-
ment output, it should be connected to the REF pin.
33nF
10ꢀ
4.7ꢀ
ENBL
VPSI
OFST
VPOS
CNTR
VPSO
BIAS AND
CM MODE AND
OFFSET CONTROL
0.1ꢃF
V-REF
0.1ꢃF
INHI
OPHI
OUTPUT,
~1V rms
INPUT,
5mV TO 1V rms
OUTPUT
STAGES
VGA CORE
SEE
TEXT
INLO
OPLO
CMOP
Q2
Q1
OUTPUT
CONTROL
MODE
GAIN INTERFACE
CMGN
VDBS
COMM
VMAG
NC
For ADCs running at sampling rates substantially below the
bandwidth of the AD8330, an intervening noise filter is recom-
mended to limit the noise bandwidth. A one-pole filter can
easily be created with a single differential capacitor between
OPHI and OPLO outputs. For a corner frequency of fc, the
capacitor should have a value of
0.1ꢃF
C1
0.1ꢃF
R1
100kꢀ
0.1ꢃF
Figure 21. Simple AGC Amplifier (Preliminary)
(26)
CFILT = 1/ 942fC
When the loop is settled, the average current in Q1 is VDBS/R1,
which varies from 2 µA at maximum gain (VDBS = 0.2 V) to 17
µA at minimum gain (VDBS = 1.7 V). This change in Q1’s cur-
rent causes an increase of ~0.25 dB over the full gain range in
the differential output of nominally 0.75 dBV at midrange (3.08
V p-p), corresponding to a 200:1 compression ratio. This is
plotted in Figure 22 for a representative 100 kHz input.
For example a 10 MHz corner requires about 100 pF.
Simple AGC Amplifier
Figure 21 illustrates the use of the inverted gain mode and the
offset gain range (0.2 V < VDBS < 1.7 V) in supporting a low cost
AGC loop. Q1 is used as a detector: when OPHI is sufficiently
higher than CNTR, due to the signal swing it conducts and
charges C1; this raises VDBS and rapidly lowers the gain. Note
that MODE is grounded (see Figure 5). The minimum voltage
needed across R1 to set up the full gain is now 0.2 V, since
CMGN is dc open-circuited (this does not alter VMAG), while the
maximum is 1.7 V.
REV. A
–23–
AD8330
1.0
Wide Range True RMS Voltmeter
The AD8362 is an rms-responding detector providing a dynamic
range of 60 dB from low frequencies to 2.7 GHz. This may be
increased to 110 dB using an AD8330 as a preconditioner, pro-
vided the noise bandwidth is limited by an interstage low-pass or
band-pass filter.
0.9
0.8
0.7
0.6
The VGA also provides an input port that is easier to drive
than the 200 Ω input of the AD8362. Figure 24 shows the
general scheme.
Both the AD8330 and AD8362 provide linear-in-decibel control
interfaces. Thus when the output of the latter is used to control
the gain of the former, this functional form is unaffected. The
overall scaling is 33 mV/dB. Figure 25 shows the time domain
response using a loop filter capacitor of 10 nF, for inputs ranging
from 10 µV to 1 V rms, that is, a 100 dB measurement range.
0.5
–50
–40
–30
–20
–10
0
INPUT TO AD8330 – dBV
Figure 22. AGC Output vs. Input Amplitude (Simulation)
1.75
1.50
The upper panel in Figure 23 shows time-domain output for 14
3 dB steps in input amplitude from 5.4 mV to 1.7 V. The wave-
forms in Figure 22 show the AGC voltage VDBS
V
DBS
1.25
1.00
0.75
0.50
0.25
.
This simple detector exhibits a temperature variation in the
differential output amplitude of about 4 mV/°C. It provides a
fast attack time (an increase in the input is quickly leveled to the
nominal output, due to the high peak currents in Q1) and a slow
release time (a decrease in the input is not restored as quickly).
The voltage at the VDBS pin may be used as an RSSI output,
scaled 30 mV/dB. Note that the attack time can be halved by adding
a second transistor as shown in the box (Figure 21). For operation
at lower frequencies, the AGC hold capacitor must be increased.
0
3
2
1
0
–1
–2
–3
–4
OUTPUT
0
10 20 30 40 50 60 70 80 90 100 110 120 130 140 150
TIME – ꢃs
Figure 23. Time Domain Waveforms (Simulation)
(See Text)
5V
3.3ꢀ
3.3ꢀ
3.3ꢀ
AD8362
0.1ꢃF
0.1ꢃF
0.1ꢃF
0.1ꢃF
1
2
3
4
5
6
7
8
16
COMM
CHPF
DECL
INHI
ACOM
10ꢃF
3.6V
15
VREF
ENBL
VPS1
OFST
VPOS
CNTR
VPSO
14
VTGT
INHI
OPHI
OPLO
CMOP
13
VPOS
C
FLT
18nF
3.6V
INPUT
AD8330
INLO
MODE
12
11
10
9
VOUT
INLO
VOUT
VSET
ACOM
CLPF
DECL
PWDN
COMM
0.1ꢃF
VDBS CMGN COMM VMAG
10ꢃF
6.04kꢀ
4.02kꢀ
Figure 24. Wide Range True RMS Voltmeter (Preliminary)
–24–
REV. A
AD8330
4
3
2
1
0
0
0.4 0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0 4.4 4.8
TIME – ms
Figure 25. Time Domain Response of RMS Voltmeter
(Simulation)
REV. A
–25–
AD8330
OUTLINE DIMENSIONS
16-Lead Shrink Small Outline Package [QSOP]
(RQ-16)
Dimensions shown in inches
0.193
BSC
16
1
9
8
0.154
BSC
0.236
BSC
PIN 1
0.069
0.053
0.065
0.049
8ꢂ
0ꢂ
0.010
0.004
0.012
0.008
0.025
BSC
0.050
0.016
SEATING
PLANE
0.010
0.006
COPLANARITY
0.004
COMPLIANT TO JEDEC STANDARDS MO-137AB
16-Lead Lead Frame Chip Scale Package [LFCSP]
3 mm ꢁ 3 mm Body
(CP-16)
Dimensions shown in millimeters
0.50
0.40
0.30
3.00
BSC SQ
0.60 MAX
PIN 1 INDICATOR
1
2
0.45
PIN 1
INDICATOR
1.45
1.30 SQ
1.15
2.75
BSC SQ
TOP
VIEW
BOTTOM
VIEW
0.50
BSC
0.25 MIN
1.50 REF
0.80 MAX
12ꢂ MAX
0.65 NOM
1.00
0.90
0.80
0.05 MAX
0.01 NOM
0.30
0.23
0.18
SEATING
PLANE
0.20 REF
COMPLIANTTO JEDEC STANDARDS MO-220-VEED-2
–26–
REV. A
AD8330
Revision History
Location
Page
4/03—Data Sheet changed from REV. 0 to REV. A.
Update OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
REV. A
–27–
–28–
相关型号：
AD8330ARQZ-REEL
IC SPECIALTY CONSUMER CIRCUIT, PDSO16, LEAD FREE, MO-137AB, QSOP-16, Consumer IC:Other
ADI
AD8330ARQZ-REEL7
IC SPECIALTY CONSUMER CIRCUIT, PDSO16, LEAD FREE, MO-137AB, QSOP-16, Consumer IC:Other
ADI
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