AD8628AUJZ-REEL7 [ADI]
Zero-Drift, Single-Supply, Rail-to-Rail Input/Output Operational Amplifier; 零漂移，单电源，轨到轨输入/输出运算放大器
| 型号： | AD8628AUJZ-REEL7 |
| 厂家： | 
ADI |
| 描述： | Zero-Drift, Single-Supply, Rail-to-Rail Input/Output Operational Amplifier |
| 文件： | 总20页 (文件大小：640K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Zero-Drift, Single-Supply, Rail-to-Rail
Input/Output Operational Amplifier
AD8628/AD8629
FEATURES
PIN CONFIGURATIONS
Lowest auto-zero amplifier noise
Low offset voltage: 1 µV
OUT
V–
1
2
3
5
V+
AD8628
TOP VIEW
Input offset drift: 0.002 µV/°C
Rail-to-rail input and output swing
5 V single-supply operation
High gain, CMRR, and PSRR: 120 dB
Very low input bias current: 100 pA max
Low supply current: 1.0 mA
Overload recovery time: 10 µs
No external components required
(Not to Scale)
+IN
4
–IN
Figure 1. 5-Lead TSOT (UJ-5)
and 5-Lead SOT-23 (RT-5)
NC
–IN
+IN
V–
1
2
3
4
8
7
6
5
NC
V+
AD8628
OUT
NC
TOP VIEW
(Not to Scale)
NC = NO CONNECT
APPLICATIONS
Automotive sensors
Figure 2. 8-Lead SOIC (R-8)
Pressure and position sensors
Strain gage amplifiers
Medical instrumentation
Thermocouple amplifiers
Precision current sensing
Photodiode amplifier
OUT A
–IN A
+IN A
V–
1
2
3
4
8
7
6
5
V+
AD8629
OUT B
–IN B
+IN B
TOP VIEW
(Not to Scale)
Figure 3. 8-Lead SOIC (R-8)
OUT A
–IN A
+IN A
V–
1
2
3
4
8
7
6
5
V+
AD8629
OUT B
–IN B
+IN B
TOP VIEW
(Not to Scale)
Figure 4. 8-Lead MSOP (RM-8)
GENERAL DESCRIPTION
This new breed of amplifier has ultralow offset, drift, and bias
current. The AD8628/AD8629 are wide bandwidth auto-zero
amplifiers featuring rail-to-rail input and output swings and low
noise. Operation is fully specified from 2.7 V to 5 V single
supply ( 1.35 V to 2.5 V dual supply).
With an offset voltage of only 1 µV, drift of less than
0.005 µV/°C, and noise of only 0.5 µV p-p (0 Hz to 10 Hz),
the AD8628/AD8629 are perfectly suited for applications in
which error sources cannot be tolerated. Position and pressure
sensors, medical equipment, and strain gage amplifiers benefit
greatly from nearly zero drift over their operating temperature
range. Many systems can take advantage of the rail-to-rail input
and output swings provided by the AD8628/AD8629 to reduce
input biasing complexity and maximize SNR.
The AD8628/AD8629 provide benefits previously found only in
expensive auto-zeroing or chopper-stabilized amplifiers. Using
Analog Devices’ new topology, these zero-drift amplifiers
combine low cost with high accuracy and low noise. (No exter-
nal capacitor is required.) In addition, the AD8628/AD8629
greatly reduce the digital switching noise found in most
chopper-stabilized amplifiers.
The AD8628/AD8629 are specified for the extended industrial
temperature range (−40°C to +125°C). The AD8628 is available
in tiny TSOT-23, SOT-23, and the popular 8-lead narrow SOIC
plastic packages. The AD8629 is available in the standard 8-lead
narrow SOIC and MSOP plastic packages.
Rev. C
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
or otherwise under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
Fax: 781.326.8703
www.analog.com
© 2004 Analog Devices, Inc. All rights reserved.
AD8628/AD8629
TABLE OF CONTENTS
Specifications..................................................................................... 3
Total Integrated Input-Referred Noise for First-Order Filter15
Input Overvoltage Protection................................................... 16
Output Phase Reversal............................................................... 16
Overload Recovery Time .......................................................... 16
Infrared Sensors.......................................................................... 17
Precision Current Shunts .......................................................... 18
Output Amplifier for High Precision DACs........................... 18
Outline Dimensions....................................................................... 19
Ordering Guide .......................................................................... 20
Electrical Characteristics............................................................. 3
Absolute Maximum Ratings............................................................ 5
ESD Caution.................................................................................. 5
Typical Performance Characteristics ............................................. 6
Functional Description .................................................................. 14
1/f Noise....................................................................................... 14
Peak-to-Peak Noise .................................................................... 15
Noise Behavior with First-Order Low-Pass Filter.................. 15
REVISION HISTORY
10/04—Data Sheet Changed from Rev. B to Rev. C
Updated Formatting...........................................................Universal
Added AD8629....................................................................Universal
Added SOIC and MSOP Pin Configurations ............................... 1
Added Figure 48.............................................................................. 13
Changes to Figure 62...................................................................... 17
Added MSOP Package ................................................................... 19
Changes to Ordering Guide .......................................................... 20
10/03—Data Sheet Changed from Rev. A to Rev. B
Changes to General Description .................................................... 1
Changes to Absolute Maximum Ratings ....................................... 4
Changes to Ordering Guide ............................................................ 4
Added TSOT-23 Package............................................................... 15
6/03—Data Sheet Changed from Rev. 0 to Rev. A
Changes to Specifications................................................................ 3
Changes to Ordering Guide ............................................................ 4
Change to Functional Description............................................... 10
Updated Outline Dimensions....................................................... 15
10/02—Revision 0: Initial Version
Rev. C | Page 2 of 20
AD8628/AD8629
SPECIFICATIONS
ELECTRICAL CHARACTERISTICS
VS = 5.0 V, VCM = 2.5 V, TA = 25°C, unless otherwise noted.
Table 1.
Parameter
Symbol
Conditions
Min
Typ
1
Max
Unit
INPUT CHARACTERISTICS
Offset Voltage
VOS
IB
5
10
µV
µV
pA
nA
pA
pA
V
−40°C ≤ TA ≤ +125°C
−40°C ≤ TA ≤ +125°C
−40°C ≤ TA ≤ +125°C
Input Bias Current
30
50
100
1.5
200
250
5
Input Offset Current
IOS
Input Voltage Range
0
Common-Mode Rejection Ratio
CMRR
AVO
VCM = 0 V to 5 V
120
115
125
120
140
130
145
135
0.002
dB
dB
dB
dB
µV/°C
−40°C ≤ TA ≤ +125°C
RL = 10 kΩ, VO = 0.3 V to 4.7 V
−40°C ≤ TA ≤ +125°C
−40°C ≤ TA ≤ +125°C
Large Signal Voltage Gain1
Offset Voltage Drift
OUTPUT CHARACTERISTICS
Output Voltage High
∆VOS/∆T
VOH
0.02
RL = 100 kΩ to ground
−40°C ≤ TA ≤ +125°C
RL = 10 kΩ to ground
−40°C ≤ TA ≤ +125°C
RL = 100 kΩ to V+
−40°C ≤ TA ≤ +125°C
RL = 10 kΩ to V+
−40°C ≤ TA ≤ +125°C
4.99
4.99
4.95
4.95
4.996
4.995
4.98
4.97
1
2
10
15
50
V
V
V
V
mV
mV
mV
mV
mA
mA
mA
mA
Output Voltage Low
VOL
5
5
20
20
Short-Circuit Limit
Output Current
ISC
IO
25
−40°C ≤ TA ≤ +125°C
−40°C ≤ TA ≤ +125°C
40
30
15
POWER SUPPLY
Power Supply Rejection Ratio
PSRR
ISY
VS = 2.7 V to 5.5 V
−40°C ≤ TA ≤ +125°C
VO = 0 V
115
130
0.85
1.0
dB
mA
mA
Supply Current/Amplifier
1.1
1.2
−40°C ≤ TA ≤ +125°C
INPUT CAPACITANCE
Differential
Common-Mode
CIN
1.5
10
pF
pF
DYNAMIC PERFORMANCE
Slew Rate
Overload Recovery Time
Gain Bandwidth Product
NOISE PERFORMANCE
Voltage Noise
SR
RL = 10 kΩ
1.0
0.05
2.5
V/µs
ms
MHz
GBP
en p-p
en p-p
en
0.1 Hz to 10 Hz
0.1 Hz to 1.0 Hz
f = 1 kHz
0.5
0.16
22
µV p-p
mV p-p
nV/√Hz
fA/√Hz
Voltage Noise Density
Current Noise Density
in
f = 10 Hz
5
1 Gain testing is highly dependent upon test bandwidth.
Rev. C | Page 3 of 20
AD8628/AD8629
VS = 2.7 V, VCM = 1.35 V, VO = 1.4 V, TA = 25°C, unless otherwise noted.
Table 2.
Parameter
Symbol
Conditions
Min
Typ
Max
Unit
INPUT CHARACTERISTICS
Offset Voltage
VOS
IB
1
5
µV
µV
pA
nA
pA
pA
V
−40°C ≤ TA ≤ +125°C
−40°C ≤ TA ≤ +125°C
−40°C ≤ TA ≤ +125°C
10
100
1.5
200
250
5
Input Bias Current
30
1.0
50
Input Offset Current
IOS
Input Voltage Range
0
Common-Mode Rejection Ratio
CMRR
AVO
VCM = 0 V to 2.7 V
115
110
110
105
130
120
140
130
0.002
dB
dB
dB
dB
µV/°C
−40°C ≤ TA ≤ +125°C
RL = 10 kΩ , VO = 0.3 V to 2.4 V
−40°C ≤ TA ≤ +125°C
−40°C ≤ TA ≤ +125°C
Large Signal Voltage Gain
Offset Voltage Drift
OUTPUT CHARACTERISTICS
Output Voltage High
∆VOS/∆T
VOH
0.02
RL = 100 kΩ to ground
−40°C ≤ TA ≤ +125°C
RL = 10 kΩ to ground
−40°C ≤ TA ≤ +125°C
RL = 100 kΩ to V+
−40°C ≤ TA ≤ +125°C
RL = 10 kΩ to V+
−40°C ≤ TA ≤ +125°C
2.68
2.68
2.67
2.67
2.695
2.695
2.68
2.675
1
2
10
15
15
V
V
V
V
mV
mV
mV
mV
mA
mA
mA
mA
Output Voltage Low
VOL
5
5
20
20
Short-Circuit Limit
Output Current
ISC
IO
10
−40°C ≤ TA ≤ +125°C
−40°C ≤ TA ≤ +125°C
10
10
5
POWER SUPPLY
Power Supply Rejection Ratio
PSRR
ISY
VS = 2.7 V to 5.5 V
−40°C ≤ TA ≤ +125°C
VO = 0 V
115
130
0.75
0.9
dB
mA
mA
Supply Current/Amplifier
1.0
1.2
−40°C ≤ TA ≤ +125°C
INPUT CAPACITANCE
Differential
Common-Mode
CIN
1.5
10
pF
pF
DYNAMIC PERFORMANCE
Slew Rate
Overload Recovery Time
Gain Bandwidth Product
NOISE PERFORMANCE
Voltage Noise
SR
RL = 10 kΩ
1
0.05
2
V/µs
ms
MHz
GBP
en p-p
en
in
0.1 Hz to 10 Hz
f = 1 kHz
f = 10 Hz
0.5
22
5
µV p-p
nV/√Hz
fA/√Hz
Voltage Noise Density
Current Noise Density
Rev. C | Page 4 of 20
AD8628/AD8629
ABSOLUTE MAXIMUM RATINGS
Table 3.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those listed in the operational sections
of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Parameters
Ratings
Supply Voltage
6 V
Input Voltage
GND − 0.3 V to VS− + 0.3 V
Differential Input Voltage1
Output Short-Circuit Duration to GND
Storage Temperature Range
R, RM, RT, UJ Packages
Operating Temperature Range
Junction Temperature Range
R, RM, RT, UJ Packages
5.0 V
Indefinite
−65°C to +150°C
−40°C to +125°C
Table 4. Thermal Characteristics
Package Type
1
θJA
θJC
61
146
43
Unit
°C/W
°C/W
°C/W
°C/W
5-Lead TSOT-23 (UJ-5)
5-Lead SOT-23 (RT-5)
8-Lead SOIC (R-8)
207
230
158
190
−65°C to +150°C
300°C
Lead Temperature Range
(Soldering, 60 s)
8-Lead MSOP (RM-8)
44
1 Differential input voltage is limited to 5 V or the supply voltage, whichever
is less.
1 θJA is specified for worst-case conditions, that is, θJA is specified for the device
soldered in a circuit board for surface-mount packages.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. C | Page 5 of 20
AD8628/AD8629
TYPICAL PERFORMANCE CHARACTERISTICS
180
100
V
T
= 2.7V
= 25°C
V
V
= 5V
S
S
90
80
70
60
50
40
30
= 2.5V
160
140
120
100
80
A
CM
T
= 25°C
A
60
40
20
10
20
0
0
–2.5
–2.5
–1.5
–0.5
0.5
1.5
2.5
–1.5
–0.5
0.5
1.5
2.5
10
10
INPUT OFFSET VOLTAGE (µV)
INPUT OFFSET VOLTAGE (µV)
Figure 5. Input Offset Voltage Distribution at 2.7 V
Figure 8. Input Offset Voltage Distribution at 5 V
60
50
40
30
20
7
6
5
4
3
2
V
= 5V
S
V
T
= 5V
S
A
+85°C
= –40°C TO +125°C
+25°C
–40°C
10
0
1
0
0
1
2
3
4
5
6
0
2
4
6
8
INPUT COMMON-MODE VOLTAGE (V)
TCVOS (nV/°C)
Figure 6. Input Bias Current vs. Input Common-Mode Voltage at 5 V
Figure 9. Input Offset Voltage Drift
1500
1k
V
= 5V
V
= 5V
S
150°C
125°C
S
A
T
= 25°C
1000
500
0
100
10
1
SOURCE
SINK
–500
0.1
–1000
–1500
0.01
0.0001
0
1
2
3
4
5
6
0.001
0.01
0.1
1
INPUT COMMON-MODE VOLTAGE (V)
LOAD CURRENT (mA)
Figure 7. Input Bias Current vs. Input Common-Mode Voltage at 5 V
Figure 10. Output Voltage to Supply Rail vs. Load Current at 5 V
Rev. C | Page 6 of 20
AD8628/AD8629
1k
1000
800
600
400
T
= 25°C
A
V
= 2.7V
S
100
10
SOURCE
SINK
1
0.1
200
0
0.01
0.0001
0.001
0.01
0.1
1
10
0
1
2
3
4
5
6
LOAD CURRENT (mA)
SUPPLY VOLTAGE (V)
Figure 11. Output Voltage to Supply Rail vs. Load Current at 2.7 V
Figure 14. Supply Current vs. Supply Voltage
70
60
50
40
30
20
10
0
1500
V
= 2.7V
= 20pF
= ∞
S
V
V
= 5V
S
C
R
φ
L
L
= 2.5V
= –40°C TO +150°C
CM
T
A
= 52.1°
1150
900
M
0
45
90
135
180
225
450
–10
–20
–30
100
0
10k
100k
1M
10M
–50
–25
0
25
50
75
100
125
150
175
FREQUENCY (Hz)
TEMPERATURE (°C)
Figure 15. Open-Loop Gain and Phase vs. Frequency
Figure 12. Input Bias Current vs. Temperature
70
1250
1000
750
V
= 5V
S
T
= 25°C
60
50
40
30
20
10
0
C
R
φ
= 20pF
= ∞
= 52.1°
A
L
L
5V
M
0
2.7V
45
90
135
180
225
500
–10
–20
–30
250
0
10k
100k
1M
10M
–50
0
50
100
150
200
FREQUENCY (Hz)
TEMPERATURE (°C
)
Figure 13. Supply Current vs. Temperature
Figure 16. Open-Loop Gain and Phase vs. Frequency
Rev. C | Page 7 of 20
AD8628/AD8629
70
300
270
240
210
180
150
120
90
V
C
R
= 2.7V
= 20pF
= 2kΩ
V = 5V
S
S
60
50
40
30
20
10
0
L
L
A
= 1
V
A
A
A
= 100
= 10
= 1
V
A
= 100
V
V
V
A = 10
V
–10
–20
–30
60
30
0
1k
10k
100k
1M
10M
100
1k
10k
100k
1M
10M
100M
FREQUENCY (Hz)
FREQUENCY (Hz)
Figure 17. Closed-Loop Gain vs. Frequency at 2.7 V
Figure 20. Output Impedance vs. Frequency at 5 V
70
60
50
40
30
20
10
0
V
C
R
= 5V
= 20pF
= 2kΩ
S
L
L
A
A
= 100
= 10
V
V
= ±1.35V
= 300pF
= ∞
S
C
R
A
L
L
V
V
= 1
A
= 1
V
–10
–20
–30
1k
10k
100k
1M
10M
TIME (4µs/DIV)
FREQUENCY (Hz)
Figure 21. Large Signal Transient Response at 2.7 V
Figure 18. Closed-Loop Gain vs. Frequency at 5 V
300
270
240
210
180
150
120
90
V
= 2.7V
S
A
= 1
V
V
= ±2.5V
= 300pF
= ∞
S
C
R
A
L
L
V
A
= 100
V
= 1
A
= 10
10M
60
30
0
V
100
1k
10k
100k
1M
100M
TIME (5µs/DIV)
FREQUENCY (Hz)
Figure 19. Output Impedance vs. Frequency at 2.7 V
Figure 22. Large Signal Transient Response at 5 V
Rev. C | Page 8 of 20
AD8628/AD8629
80
70
60
50
40
V
= ±1.35V
= 50pF
= ∞
V
R
= ±2.5V
= 2kΩ
= 25°C
S
S
C
R
A
L
L
V
L
T
A
= 1
30
20
OS–
OS+
10
0
1
10
100
1k
TIME (4µs/DIV)
CAPACITIVE LOAD (pF)
Figure 23. Small Signal Transient Response at 2.7 V
Figure 26. Small Signal Overshoot vs. Load Capacitance at 5 V
V
C
R
A
= ±2.5V
= 50pF
= ∞
V
= ±2.5V
= –50
= 10kΩ
= 0
S
S
A
R
C
L
L
V
V
L
L
V
IN
= 1
CH1 = 50mV/DIV
CH2 = 1V/DIV
0V
0V
V
OUT
TIME (4µs/DIV)
TIME (2µs/DIV)
Figure 27. Positive Overvoltage Recovery
Figure 24. Small Signal Transient Response at 5 V
100
90
80
70
60
50
40
30
V
R
= ±1.35V
= 2kΩ
= 25°C
S
0V
L
T
A
V
= ±2.5V
= –50
= 10kΩ
= 0
S
A
R
C
V
L
L
V
IN
CH1 = 50mV/DIV
CH2 = 1V/DIV
OS–
V
OUT
OS+
20
10
0
0V
1
10
100
1k
TIME (10µs/DIV)
CAPACITIVE LOAD (pF)
Figure 28. Negative Overvoltage Recovery
Figure 25. Small Signal Overshoot vs. Load Capacitance at 2.7 V
Rev. C | Page 9 of 20
AD8628/AD8629
140
120
100
80
V
V
C
R
A
= ±2.5V
= 1kHz @ ±3V p-p
= 0pF
= 10kΩ
= 1
S
V
= ±1.35V
IN
S
L
L
V
60
+PSRR
40
20
–PSRR
0
–20
–40
–60
100
1k
10k
100k
1M
10M
10M
1M
TIME (200µs/DIV)
FREQUENCY (Hz)
Figure 32. PSRR vs. Frequency
Figure 29. No Phase Reversal
140
120
100
80
140
120
100
80
V
= ±2.5V
S
V
= 2.7V
S
60
60
+PSRR
40
40
20
–PSRR
20
0
0
–20
–40
–60
–20
–40
–60
100
1k
10k
100k
1M
100
1k
10k
100k
1M
10M
FREQUENCY (Hz)
FREQUENCY (Hz)
Figure 33. PSRR vs. Frequency
Figure 30. CMRR vs. Frequency at 2.7 V
3.0
2.5
2.0
1.5
1.0
140
120
100
80
V
= 5V
S
V
R
= 2.7V
= 10kΩ
= 25°C
= 1
S
L
T
A
A
V
60
40
20
0
–20
–40
–60
0.5
0
100
1k
10k
100k
100
1k
10k
100k
1M
10M
FREQUENCY (Hz)
FREQUENCY (Hz)
Figure 34. Maximum Output Swing vs. Frequency
Figure 31. CMRR vs. Frequency at 5 V
Rev. C | Page 10 of 20
AD8628/AD8629
5.5
5.0
120
105
90
V
= 2.7V
S
NOISE AT 1kHz = 21.3nV
V
R
= 5V
S
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
= 10kΩ
= 25°C
= 1
L
T
A
A
V
75
60
45
30
15
0
0.5
0
100
1k
10k
100k
1M
10
10
0
0.5
1.0
1.5
2.0
2.5
FREQUENCY (Hz)
FREQUENCY (kHz)
Figure 35. Maximum Output Swing vs. Frequency at 5 V
Figure 38. Voltage Noise Density at 2.7 V from 0 Hz to 2.5 kHz
0.60
0.45
0.30
0.15
120
V
= 2.7V
S
V
= 2.7V
S
NOISE AT 10kHz = 42.4nV
105
90
75
0
60
–0.15
–0.30
–0.45
–0.60
45
30
15
0
0
1
2
3
4
5
6
7
8
9
0
5
10
15
20
25
TIME (µs)
FREQUENCY (kHz)
Figure 36. 0.1 Hz to 10 Hz Noise at 2.7 V
Figure 39. Voltage Noise Density at 2.7 V from 0 Hz to 25 kHz
0.60
0.45
0.30
0.15
120
V
= 5V
S
V
= 5V
S
NOISE AT 1kHz = 22.1nV
105
90
75
0
60
–0.15
–0.30
–0.45
–0.60
45
30
15
0
0
1
2
3
4
5
6
7
8
9
0
0.5
1.0
1.5
2.0
2.5
TIME (µs)
FREQUENCY (kHz)
Figure 37. 0.1 Hz to 10 Hz Noise at 5 V
Figure 40. Voltage Noise Density at 5 V from 0 Hz to 2.5 kHz
Rev. C | Page 11 of 20
AD8628/AD8629
120
150
V
T
= 2.7V
= –40°C TO +150°C
S
A
V
= 5V
S
NOISE AT 10kHz = 36.4nV
105
90
100
50
0
75
60
I
–
SC
45
30
15
0
I
+
SC
–50
–100
0
5
10
15
20
25
–50
–25
0
25
50
75
100
C)
125
150
175
175
175
FREQUENCY (kHz)
TEMPERATURE (
°
Figure 41. Voltage Noise Density at 5 V from 0 Hz to 25 kHz
Figure 44. Output Short-Circuit Current vs. Temperature
120
150
V
= 5V
V
T
= 5V
S
S
= –40°C TO +150°C
105
90
A
100
50
0
I
–
SC
75
60
45
30
15
0
–50
I
+
SC
–100
0
5
10
–50
–25
0
25
50
75
100
C)
125
150
FREQUENCY (kHz)
TEMPERATURE (
°
Figure 45. Output Short-Circuit Current vs. Temperature
Figure 42. Voltage Noise
1k
100
10
150
V
= 5V
S
140
130
V
– V @ 1kΩ
OH
CC
V
T
= 2.7V TO 5V
= –40°C TO +125°C
S
A
120
110
100
V
– V @ 1kΩ
EE
OL
V
– V @ 10kΩ
OH
CC
V
– V @ 10kΩ
EE
OL
90
80
70
V
– V @ 100k
Ω
CC
OH
1
V
– V @ 100k
Ω
OL
EE
60
50
0.10
–50
–50
–25
0
25
50
75
100
125
–25
0
25
50
75
100
C)
125
150
TEMPERATURE (°C)
TEMPERATURE (
°
Figure 43. Power Supply Rejection vs. Temperature
Figure 46. Output-to-Rail Voltage vs. Temperature
Rev. C | Page 12 of 20
AD8628/AD8629
1k
100
10
140
V
= 2.7V
V
= ±2.5V
S
SY
120
100
80
V
– V @ 1k
Ω
CC
OH
V
– V @ 1kΩ
EE
OL
V
– V @ 10kΩ
OH
CC
R1
10kΩ
V
– V @ 10kΩ
EE
60
OL
+2.5V
V+
R2
100Ω
V
– V @ 100kΩ
OH
CC
V–
B
40
+
–
V
IN
A
1
28mV p-p
V
V
– V @ 100kΩ
EE
OUT
OL
V–
V+
20
0
–2.5V
0.10
–50
1k
10k
100k
FREQUENCY (Hz)
1M
10M
–25
0
25
50
75
100
C)
125
150
175
TEMPERATURE (
°
Figure 47. Output-to-Rail Voltage vs. Temperature
Figure 48. AD8629 Channel Separation
Rev. C | Page 13 of 20
AD8628/AD8629
FUNCTIONAL DESCRIPTION
The AD8628/AD8629 are single-supply, ultrahigh precision
rail-to-rail input and output operational amplifiers. The typical
offset voltage of less than 1 µV allows these amplifiers to be
easily configured for high gains without risk of excessive
output voltage errors. The extremely small temperature drift of
2 nV/°C ensures a minimum of offset voltage error over their
entire temperature range of −40°C to +125°C, making these
amplifiers ideal for a variety of sensitive measurement
applications in harsh operating environments.
1/F NOISE
1/f noise, also known as pink noise, is a major contributor to
errors in dc-coupled measurements. This 1/f noise error term
can be in the range of several µV or more, and, when amplified
with the closed-loop gain of the circuit, can show up as a large
output offset. For example, when an amplifier with a 5 µV p-p
1/f noise is configured for a gain of 1,000, its output has 5 mV
of error due to the 1/f noise. But the AD8628/AD8629 eliminate
1/f noise internally, and thereby greatly reduce output errors.
The AD8628/AD8629 achieve a high degree of precision
through a patented combination of auto-zeroing and chopping.
This unique topology allows the AD8628/AD8629 to maintain
their low offset voltage over a wide temperature range and over
their operating lifetime. The AD8628/AD8629 also optimize the
noise and bandwidth over previous generations of auto-zero
amplifiers, offering the lowest voltage noise of any auto-zero
amplifier by more than 50%.
The internal elimination of 1/f noise is accomplished as follows.
1/f noise appears as a slowly varying offset to AD8628/AD8629
inputs. Auto-zeroing corrects any dc or low frequency offset.
Therefore, the 1/f noise component is essentially removed,
leaving the AD8628/AD8629 free of 1/f noise.
One of the biggest advantages that the AD8628/AD8629 bring
to systems applications over competitive auto-zero amplifiers is
their very low noise. The comparison shown in Figure 49
indicates an input-referred noise density of 19.4 nV/√Hz at
1 kHz for the AD8628, which is much better than the LTC2050
and LMC2001. The noise is flat from dc to 1.5 kHz, slowly
increasing up to 20 kHz. The lower noise at low frequency is
desirable where auto-zero amplifiers are widely used.
Previous designs used either auto-zeroing or chopping to add
precision to the specifications of an amplifier. Auto-zeroing
results in low noise energy at the auto-zeroing frequency, at the
expense of higher low-frequency noise due to aliasing of
wideband noise into the auto-zeroed frequency band. Chopping
results in lower low-frequency noise at the expense of larger
noise energy at the chopping frequency. The AD8628/AD8629
family use both auto-zeroing and chopping in a patented ping-
pong arrangement to obtain lower low-frequency noise together
with lower energy at the chopping and auto-zeroing
frequencies, maximizing the signal-to-noise ratio (SNR) for the
majority of applications without the need for additional
filtering. The relatively high clock frequency of 15 kHz
simplifies filter requirements for a wide, useful, noise-free
bandwidth.
120
LTC2050
105
(89.7nV/√Hz)
90
75
60
45
30
LMC2001
(31.1nV/√Hz)
The AD8628 is among the few auto-zero amplifiers offered in
the 5-lead TSOT-23 package. This provides a significant
improvement over the ac parameters of the previous auto-zero
amplifiers. The AD8628/AD8629 have low noise over a
relatively wide bandwidth (0 Hz to 10 kHz) and can be used
where the highest dc precision is required. In systems with
signal bandwidths of from 5 kHz to 10 kHz, the AD8628/
AD8629 provide true 16-bit accuracy, making them the best
choice for very high resolution systems.
15
0
AD8628
(19.4nV/√Hz)
MK AT 1kHz FOR ALL 3 GRAPHS
10 12
0
2
4
6
8
FREQUENCY (kHz)
Figure 49. Noise Spectral Density of AD8628 vs. Competition
Rev. C | Page 14 of 20
AD8628/AD8629
50
45
40
35
30
25
20
15
PEAK-TO-PEAK NOISE
Because of the ping-pong action between auto-zeroing and
chopping, the peak-to-peak noise of the AD8628/AD8629 is
much lower than the competition. Figure 50 and Figure 51
show this comparison.
e
p-p = 0.5µV
n
BW = 0.1Hz TO 10Hz
10
5
0
0
10
20
30
40
50
60
70
80
90
100
FREQUENCY (Hz)
Figure 53. Simulation Transfer Function of the Test Circuit
50
45
40
35
30
25
20
15
TIME (1s/DIV)
Figure 50. AD8628 Peak-to-Peak Noise
e
p-p = 2.3µV
n
BW = 0.1Hz TO 10Hz
10
5
0
0
10
20
30
40
50
60
70
80
90
100
FREQUENCY (kHz)
Figure 54. Actual Transfer Function of Test Circuit
The measured noise spectrum of the test circuit shows that
noise between 5 kHz and 45 kHz is successfully rolled off by the
first-order filter.
TOTAL INTEGRATED INPUT-REFERRED NOISE
FOR FIRST-ORDER FILTER
TIME (1s/DIV)
Figure 51. LTC2050 Peak-to-Peak Noise
For a first-order filter, the total integrated noise from the
AD8628 is lower than the LTC2050.
NOISE BEHAVIOR WITH FIRST-ORDER LOW-PASS
FILTER
10
The AD8628 was simulated as a low-pass filter and then
configured as shown in Figure 52. The behavior of the AD8628
matches the simulated data. It was verified that noise is rolled
off by first-order filtering.
LTC2050
AD8551
AD8628
IN
1
OUT
100kΩ
1kΩ
470pF
Figure 52. Test Circuit: First-Order Low-Pass Filter—×101 Gain
and 3 kHz Corner Frequency
0.1
10
100
1k
10k
3dB FILTER BANDWIDTH (Hz)
Figure 55. 3 dB Filter Bandwidth in Hz
Rev. C | Page 15 of 20
AD8628/AD8629
INPUT OVERVOLTAGE PROTECTION
CH1 = 50mV/DIV
CH2 = 1V/DIV
V
IN
A
= –50
Although the AD8628/AD8629 are rail-to-rail input amplifiers,
care should be taken to ensure that the potential difference
between the inputs does not exceed the supply voltage. Under
normal negative feedback operating conditions, the amplifier
corrects its output to ensure that the two inputs are at the same
voltage. However, if either input exceeds either supply rail by
more than 0.3 V, large currents begin to flow through the ESD
protection diodes in the amplifier.
V
0V
0V
These diodes are connected between the inputs and each supply
rail to protect the input transistors against an electrostatic
discharge event and are normally reverse-biased. However, if the
input voltage exceeds the supply voltage, these ESD diodes can
become forward-biased. Without current limiting, excessive
amounts of current could flow through these diodes, causing
permanent damage to the device. If inputs are subject to
overvoltage, appropriate series resistors should be inserted to
limit the diode current to less than 5 mA maximum.
V
OUT
TIME (500µs/DIV)
Figure 56. Positive Input Overload Recovery for the AD8628
CH1 = 50mV/DIV
CH2 = 1V/DIV
V
IN
A
= –50
V
OUTPUT PHASE REVERSAL
0V
0V
Output phase reversal occurs in some amplifiers when the input
common-mode voltage range is exceeded. As common-mode
voltage is moved outside of the common-mode range, the
outputs of these amplifiers can suddenly jump in the opposite
direction to the supply rail. This is the result of the differential
input pair shutting down, causing a radical shifting of internal
voltages that results in the erratic output behavior.
V
OUT
The AD8628/AD8629 amplifiers have been carefully designed
to prevent any output phase reversal, provided that both inputs
are maintained within the supply voltages. If one or both inputs
could exceed either supply voltage, a resistor should be placed in
series with the input to limit the current to less than 5 mA. This
ensures that the output does not reverse its phase.
TIME (500µs/DIV)
Figure 57. Positive Input Overload Recovery for LTC2050
CH1 = 50mV/DIV
CH2 = 1V/DIV
V
IN
A
= –50
V
OVERLOAD RECOVERY TIME
Many auto-zero amplifiers are plagued by a long overload
recovery time, often in ms, due to the complicated settling
behavior of the internal nulling loops after saturation of the
outputs. The AD8628/AD8629 have been designed so that
internal settling occurs within two clock cycles after output
saturation happens. This results in a much shorter recovery
time, less than 10 µs, when compared to other auto-zero
amplifiers. The wide bandwidth of the AD8628/AD8629
enhances performance when they are used to drive loads that
inject transients into the outputs. This is a common situation
when an amplifier is used to drive the input of switched
capacitor ADCs.
0V
0V
V
OUT
TIME (500µs/DIV)
Figure 58. Positive Input Overload Recovery for LMC2001
Rev. C | Page 16 of 20
AD8628/AD8629
The results shown in Figure 56 to Figure 61 are summarized in
Table 5.
0V
CH1 = 50mV/DIV
CH2 = 1V/DIV
Table 5. Overload Recovery Time
A
= –50
V
Positive Overload
Recovery (µs)
Negative Overload
Recovery (µs)
V
IN
Product
AD8628
LTC2050
LMC2001
6
9
650
40,000
25,000
35,000
V
OUT
0V
INFRARED SENSORS
Infrared (IR) sensors, particularly thermopiles, are increasingly
being used in temperature measurement for applications as
wide-ranging as automotive climate control, human ear
thermometers, home insulation analysis, and automotive repair
diagnostics. The relatively small output signal of the sensor
demands high gain with very low offset voltage and drift to
avoid dc errors.
TIME (500µs/DIV)
Figure 59. Negative Input Overload Recovery for the AD8628
0V
CH1 = 50mV/DIV
CH2 = 1V/DIV
A
= –50
V
V
IN
OUT
V
If interstage ac coupling is used (Figure 62), low offset and drift
prevents the input amplifier’s output from drifting close to
saturation. The low input bias currents generate minimal errors
from the sensor’s output impedance. As with pressure sensors,
the very low amplifier drift with time and temperature elimi-
nates additional errors once the temperature measurement has
been calibrated. The low 1/f noise improves SNR for dc
measurements taken over periods often exceeding 1/5 s.
0V
Figure 64 (shows a circuit that can amplify ac signals from
100 µV to 300 µV up to the 1 V to 3 V level, with gain of
10,000 for accurate A/D conversion.
TIME (500µs/DIV)
Figure 60. Negative Input Overload Recovery for LTC2050
10kΩ
100kΩ
100Ω
100kΩ
5V
5V
0V
CH1 = 50mV/DIV
CH2 = 1V/DIV
100µV – 300µV
10µF
A
= –50
V
1/2 AD8629
IR
1/2 AD8629
V
DETECTOR
IN
10kΩ
f
≈ 1.6Hz
C
V
OUT
TO BIAS
VOLTAGE
Figure 62. AD8629 Used as Preamplifier for Thermopile
0V
TIME (500µs/DIV)
Figure 61. Negative Input Overload Recovery for LMC2001
Rev. C | Page 17 of 20
AD8628/AD8629
OUTPUT AMPLIFIER FOR HIGH PRECISION DACs
PRECISION CURRENT SHUNTS
The AD8628/AD8629 are used as output amplifiers for a 16-bit
high precision DAC in a unipolar configuration. In this case, the
selected op amp needs to have very low offset voltage (the DAC
LSB is 38 µV when operated with a 2.5 V reference) to eliminate
the need for output offset trims. Input bias current (typically a
few tens of picoamperes) must also be very low, because it
generates an additional zero code error when multiplied by the
DAC output impedance (approximately 6 kΩ).
A precision shunt current sensor benefits from the unique
attributes of auto-zero amplifiers when used in a differencing
configuration (Figure 63). Shunt current sensors are used in
precision current sources for feedback control systems. They are
also used in a variety of other applications, including battery
fuel gauging, laser diode power measurement and control,
torque feedback controls in electric power steering, and
precision power metering.
Rail-to-rail input and output provide full-scale output with very
little error. Output impedance of the DAC is constant and code-
independent, but the high input impedance of the AD8628/
AD8629 minimizes gain errors. The amplifiers’ wide bandwidth
also serves well in this case. The amplifiers, with settling time of
1 µs, add another time constant to the system, increasing the
settling time of the output. The settling time of the AD5541 is
1 µs. The combined settling time is approximately 1.4 µs, as can
be derived from the following equation:
R
0.1Ω
S
R
SUPPLY
L
I
100kΩ
100Ω
e = 1,000 R
100mV/mA
I
S
C
5V
AD8628
100kΩ
100Ω
2
2
tS
(
TOTAL
)
=
tS DAC
)
+
(
tS AD8628
)
C
Figure 63. Low-Side Current Sensing
In such applications, it is desirable to use a shunt with very low
resistance to minimize the series voltage drop; this minimizes
wasted power and allows the measurement of high currents
without saving power. A typical shunt might be 0.1 Ω. At
measured current values of 1 A, the shunt’s output signal is
hundreds of mV, or even V, and amplifier error sources are not
critical. However, at low measured current values in the 1 mA
range, the 100 µV output voltage of the shunt demands a very
low offset voltage and drift to maintain absolute accuracy. Low
input bias currents are also needed, so that injected bias current
does not become a significant percentage of the measured
current. High open-loop gain, CMRR, and PSRR all help to
maintain the overall circuit accuracy. As long as the rate of
change of the current is not too fast, an auto-zero amplifier can
be used with excellent results.
2.5V
5V
10µF
0.1µF
0.1µF
SERIAL
V
REF(REF*) REFS*
AD5541/AD5542
DD
INTERFACE
CS
DIN
UNIPOLAR
OUTPUT
OUT
SCLK
LDAC*
AD8628
DGND
AGND
*AD5542 ONLY
Figure 64. AD8628 Used as an Output Amplifier
Rev. C | Page 18 of 20
AD8628/AD8629
OUTLINE DIMENSIONS
2.90 BSC
5.00 (0.1968)
4.80 (0.1890)
5
1
4
3
2.80 BSC
1.60 BSC
8
1
5
4
6.20 (0.2440)
5.80 (0.2284)
4.00 (0.1574)
3.80 (0.1497)
2
PIN 1
0.95 BSC
1.27 (0.0500)
BSC
1.90
BSC
0.50 (0.0196)
× 45°
0.90
0.87
0.84
1.75 (0.0688)
1.35 (0.0532)
0.25 (0.0099)
0.25 (0.0098)
0.10 (0.0040)
8°
1.00 MAX
0.51 (0.0201)
0.31 (0.0122)
0° 1.27 (0.0500)
COPLANARITY
0.10
0.25 (0.0098)
0.17 (0.0067)
SEATING
PLANE
0.40 (0.0157)
8°
4°
0.10 MAX
0.60
0.45
0.30
0.50
0.30
SEATING
PLANE
0.20
0.08
COMPLIANT TO JEDEC STANDARDS MS-012AA
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COMPLIANT TO JEDEC STANDARDS MO-193AB
Figure 65. 5-Lead Thin Small Outline Transistor Package [TSOT]
Figure 67. 8-Lead Standard Small Outline Package [SOIC]
Narrow Body (R-8)
(UJ-5)
Dimensions shown in millimeters
Dimensions shown in millimeters and (inches)
2.90 BSC
3.00
BSC
5
4
3
2.80 BSC
1.60 BSC
8
5
4
2
4.90
BSC
3.00
BSC
PIN 1
0.95 BSC
1.90
BSC
PIN 1
1.30
1.15
0.90
0.65 BSC
1.10 MAX
0.15
0.00
1.45 MAX
0.22
0.08
0.80
0.60
0.40
8°
0°
10°
5°
0°
0.38
0.22
COPLANARITY
0.10
0.23
0.08
0.15 MAX
0.50
0.30
0.60
0.45
0.30
SEATING
PLANE
SEATING
PLANE
COMPLIANT TO JEDEC STANDARDS MO-178AA
COMPLIANT TO JEDEC STANDARDS MO-187AA
Figure 66. 5-Lead Small Outline Transistor Package [SOT-23]
Figure 65. 8-Lead Standard Small Outline Package [MSOP]
(RM-8)
(RT-5)
Dimensions shown in millimeters
Dimensions shown in millimeters
Rev. C | Page 19 of 20
AD8628/AD8629
ORDERING GUIDE
Model
AD8628AUJ-R2
Temperature Range
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
Package Description
5-Lead TSOT-23
5-Lead TSOT-23
5-Lead TSOT-23
5-Lead TSOT-23
5-Lead TSOT-23
5-Lead TSOT-23
8-Lead SOIC
8-Lead SOIC
8-Lead SOIC
8-Lead SOIC
8-Lead SOIC
Package Option
UJ-5
UJ-5
UJ-5
UJ-5
UJ-5
UJ-5
R-8
R-8
R-8
R-8
R-8
Branding
AYB
AYB
AYB
AYB
AD8628AUJ-REEL
AD8628AUJ-REEL7
AD8628AUJZ-R21
AD8628AUJZ-REEL1
AD8628AUJZ-REEL71
AD8628AR
AD8628AR-REEL
AD8628AR-REEL7
AD8628ARZ1
AD8628ARZ-REEL1
AD8628ARZ-REEL71
AD8628ART-R2
AD8628ART-REEL7
AD8628ARTZ-R21
AD8628ARTZ-REEL71
AD8629ARZ1
AYB
AYB
8-Lead SOIC
R-8
5-Lead SOT-23
5-Lead SOT-23
5-Lead SOT-23
5-Lead SOT-23
8-Lead SOIC
8-Lead SOIC
8-Lead SOIC
8-Lead MSOP
8-Lead MSOP
RT-5
RT-5
RT-5
RT-5
R-8
R-8
R-8
RM-8
RM-8
AYA
AYA
AYA
AYA
AD8629ARZ-REEL1
AD8629ARZ-REEL71
AD8629ARMZ-R21
AD8629ARMZ-REEL1
A06
A06
1 Z = Pb-free part.
©
2004 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C02735–0–10/04(C)
Rev. C | Page 20 of 20
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