AD9717-EBZ [ADI]
Dual, 8-/10-/12-/14-Bit Low Power Digital-to-Analog Converters; 双通道， 8位/ 10位/ 12位/ 14位低功耗数字 - 模拟转换器
| 型号： | AD9717-EBZ |
| 厂家： | 
ADI |
| 描述： | Dual, 8-/10-/12-/14-Bit Low Power Digital-to-Analog Converters |
| 文件： | 总48页 (文件大小：1840K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Dual, 8-/10-/12-/14-Bit Low
Power Digital-to-Analog Converters
AD9714/AD9715/AD9716/AD9717
FEATURES
GENERAL DESCRIPTION
Power dissipation @ 3.3 V, 2 mA output
37 mW @ 10 MSPS
80 mW @ 125 MSPS
Sleep mode: <3 mW @ 3.3 V
Supply voltage: 1.8 V to 3.3 V
SFDR to Nyquist
84 dBc @ 1 MHz output
75 dBc @ 10 MHz output
AD9717 NSD @ 1 MHz output, 125 MSPS, 2 mA: −151 dBc/Hz
Differential current outputs: 1 mA to 4 mA
Two on-chip auxiliary DACs
CMOS inputs with single-port operation
Output common mode: adjustable 0 V to 1.2 V
Small footprint 40-lead LFCSP Pb-free package
The AD9714/AD9715/AD9716/AD9717 are pin-compatible
dual, 8-/10-/12-/14-bit, low power digital-to-analog converters
(DACs) that provide a sample rate of 125 MSPS. These TxDAC®
converters are optimized for the transmit signal path of commu-
nication systems. All the devices share the same interface, LFCSP
package, and pinout, providing an upward or downward compo-
nent selection path based on performance, resolution, and cost.
The AD9714/AD9715/AD9716/AD9717 offer exceptional ac and
dc performance and support update rates up to 125 MSPS.
The flexible power supply operating range of 1.8 V to 3.3 V and
low power dissipation of the AD9714/AD9715/AD9716/AD9717
make them well-suited for portable and low power applications.
PRODUCT HIGHLIGHTS
APPLICATIONS
1. Low Power.
DACs operate on a single 1.8 V to 3.3 V supply; total power
consumption reduces to 35 mW at 125 MSPS with a 1.8 V
supply. Sleep and power-down modes are provided for low
power idle periods.
Wireless infrastructures
Picocell, femtocell base stations
Medical instrumentation
Ultrasound transducer excitation
Portable instrumentation
2. CMOS Clock Input.
High speed, single-ended CMOS clock input supports
125 MSPS conversion rate.
Signal generators, arbitrary waveform generators
3. Easy Interfacing to Other Components.
Adjustable output common mode from 0 V to 1.2 V allows
for easy interfacing to other components that accept common-
mode levels greater than 0 V.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registeredtrademarks arethe property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
Fax: 781.461.3113
www.analog.com
©2008 Analog Devices, Inc. All rights reserved.
AD9714/AD9715/AD9716/AD9717
TABLE OF CONTENTS
Features .............................................................................................. 1
SPI Register Map ............................................................................ 33
SPI Register Descriptions.............................................................. 34
Digital Interface Operation........................................................... 37
Digital Data Latching and Retimer Section............................ 38
Estimating the Overall DAC Pipeline Delay........................... 39
Self-Calibration........................................................................... 40
Coarse Gain Adjustment........................................................... 41
Using the Internal Termination Resistors............................... 42
Applications Information.............................................................. 43
Output Configurations.............................................................. 43
Differential Coupling Using a Transformer ............................... 43
Single-Ended Buffered Output Using an Op Amp................ 43
Differential Buffered Output Using an Op Amp ................... 44
Auxiliary DACs........................................................................... 44
DAC-to-Modulator Interfacing................................................ 45
Applications....................................................................................... 1
General Description......................................................................... 1
Product Highlights ........................................................................... 1
Revision History ............................................................................... 2
Functional Block Diagram .............................................................. 3
Specifications..................................................................................... 4
DC Specifications ......................................................................... 4
Digital Specifications ................................................................... 6
AC Specifications.......................................................................... 7
Absolute Maximum Ratings............................................................ 8
Thermal Resistance ...................................................................... 8
ESD Caution.................................................................................. 8
Pin Configurations and Function Descriptions ........................... 9
Typical Performance Characteristics ........................................... 17
Terminology .................................................................................... 29
Theory of Operation ...................................................................... 30
Serial Peripheral Interface (SPI) ................................................... 31
General Operation of the Serial Interface............................... 31
Instruction Byte .......................................................................... 31
Serial Interface Port Pin Descriptions ..................................... 31
MSB/LSB Transfers..................................................................... 32
Serial Port Operation................................................................. 32
Pin Mode ..................................................................................... 32
Correcting for Nonideal Performance of Quadrature
Modulators on the IF-to-RF Conversion ................................ 45
I/Q Channel Gain Matching..................................................... 45
LO Feedthrough Compensation .............................................. 46
Results of Gain and Offset Correction .................................... 46
Modifying the Evaluation Board to Use the ADL5370
On-Board Quadrature Modulator ........................................... 47
Outline Dimensions....................................................................... 48
Ordering Guide .......................................................................... 48
REVISION HISTORY
8/08—Revision 0: Initial Version
Rev. 0 | Page 2 of 48
AD9714/AD9715/AD9716/AD9717
FUNCTIONAL BLOCK DIAGRAM
AD9714/AD9715/
AD9716/AD9717
1V
SPI
INTERFACE
DB11
DB10
DB9
R
R
SET
16kΩ
SET
16kΩ
R
CM
1kΩ TO
250Ω
RLIN
500Ω
500Ω
10kΩ
IOUTN
IOUTP
I
I DAC
REF
100µA
DB8
BAND
GAP
RLIP
AUX1DAC
AUX2DAC
DVDDIO
DVSS
DVDD
DB7
AVDD
AVSS
RLQP
1 INTO 2
INTERLEAVED
DATA
I DATA
INTERFACE
500Ω
500Ω
1.8V
LDO
QOUTP
QOUTN
Q DATA
Q DAC
RLQN
CLOCK
DIST
DB6
R
CM
1kΩ TO
250Ω
DB5
Figure 1.
Rev. 0 | Page 3 of 48
AD9714/AD9715/AD9716/AD9717
SPECIFICATIONS
DC SPECIFICATIONS
TMIN to TMAX, AVDD = 3.3 V, DVDD = 3.3 V, DVDDIO = 3.3 V, CVDD = 3.3 V, IOUTFS = 2 mA, maximum sample rate, unless
otherwise noted.
Table 1.
AD9714
AD9715
Typ
AD9716
Typ
AD9717
Typ
Parameter
Min
Typ
Max
Min
Max
Min
Max
Min
Max
Unit
RESOLUTION
8
10
12
14
Bits
ACCURACY @ 3.3 V
Differential Nonlinearity (DNL)
Precalibration
0.02
0.08
0.01
0.4
0.2
1.7
1.0
LSB
LSB
Postcalibration
0.003
Integral Nonlinearity (INL)
Precalibration
0.025
0.01
0.13
0.05
0.4
0.3
1.8
1.3
LSB
LSB
Postcalibration
ACCURACY @ 1.8 V
Differential Nonlinearity (DNL)
Precalibration
0.02
0.08
0.01
0.4
0.2
1.2
1.0
LSB
LSB
Postcalibration
0.005
Integral Nonlinearity (INL)
Precalibration
0.025
0.02
0.12
0.05
0.4
1.5
1.1
LSB
LSB
Postcalibration
0.25
MAIN DAC OUTPUTS
Offset Error
−1
−2
0
+1
+2
−1
−2
0
+1
+2
−1
−2
0
+1
+2
−1
−2
0
+1
+2
mV
Gain Error
Internal Reference
Full-Scale Output Current1
VCC = 3.3 V
% of FSR
1
2
4
1
2
4
1
2
4
1
2
4
mA
mA
V
VCC = 1.8 V
1
2
2.5
+1.2
1
2
2.5
+1.2
1
2
2.5
+1.2
1
2
2.5
+1.2
Output Compliance Range
Output Resistance
Crosstalk, Q DAC to I DAC
fOUT = 30 MHz
−0.5
0
−0.5
0
−0.5
0
−0.5
0
200
200
200
200
MΩ
97
78
97
78
97
78
97
78
dB
dB
fOUT = 60 MHz
MAIN DAC TEMPERATURE DRIFT
Offset
0
0
0
0
ppm/°C
ppm/°C
ppm/°C
Gain
40
25
40
25
40
25
40
25
Reference Voltage
AUXDAC OUTPUTS
Resolution
10
10
10
10
Bits
μA
Full-Scale Output Current
(Current Sourcing Mode)
125
125
125
125
Voltage Output Mode
VSS
VSS
VDD
VSS
VSS
VDD
VSS
VSS
VDD
VSS
VSS
VDD
V
V
Output Compliance Range
(Sourcing 1 mA)
VDD
0.25
−
VDD
0.25
−
VDD
0.25
−
VDD −
0.25
Output Compliance Range
(Sinking 1 mA)
VSS
0.25
+
VDD
VSS
0.25
+
VDD
VSS
0.25
+
VDD
VSS
0.25
+
VDD
V
Output Resistance in Current
Output Mode VSS to +1 V
1
1
1
1
MΩ
Bits
AUX DAC Monotonicity
Guaranteed
10
10
10
10
REFERENCE OUTPUT
Internal Reference Voltage
Output Resistance
0.98
0.1
1.025
10
1.08
1.25
0.98
0.1
1.025
10
1.08
1.25
0.98
0.1
1.025
10
1.08
1.25
0.98
0.1
1.025
10
1.08
1.25
V
kΩ
REFERENCE INPUT
Voltage Compliance
Input Resistance
V
1
1
1
1
MΩ
Rev. 0 | Page 4 of 48
AD9714/AD9715/AD9716/AD9717
AD9714
Typ
AD9715
Typ
AD9716
Typ
AD9717
Typ
Parameter
Min
Max
Min
Max
Min
Max
Min
Max
Unit
DAC MATCHING
Gain Matching
ANALOG SUPPLY VOLTAGES
AVDD
−1
+1
−1
+1
−1
+1
−1
+1
% FSR
1.7
1.7
3.5
3.5
1.7
1.7
3.5
3.5
1.7
1.7
3.5
3.5
1.7
1.7
3.5
3.5
V
V
CVDD
DIGITAL SUPPLY VOLTAGES
DVDD
1.7
1.7
1.9
3.5
1.7
1.7
1.9
3.5
1.7
1.7
1.9
3.5
1.7
1.7
1.9
3.5
V
V
DVDDIO
POWER CONSUMPTION @ 3.3 V
fDAC = 125 MSPS, IF = 12.5 MHz
IAVDD
86
10
0
86
10
0
86
10
0
86
10
0
mW
mA
2
IDVDD
mA
3
IDVDDIO
11
3
11
3
11
3
11
3
mA
ICVDD
mA
Power-Down Mode with Clock
Power-Down Mode No Clock
Power Supply Rejection Ratio
POWER CONSUMPTION @ 1.8 V
fDAC = 125 MSPS, IF = 12.5 MHz
IAVDD
50
1.5
−0.04
50
1.5
−0.04
50
1.5
−0.04
50
1.5
−0.04
mW
mW
% FSR/V
35
35
35
35
mW
mA
10
10
10
10
IDVDD + IDVDDIO
8
8
8
8
mA
ICVDD
1.5
12
1.5
12
1.5
12
1.5
12
mA
Power-Down Mode with Clock
Power-Down Mode No Clock
Power Supply Rejection Ratio
OPERATING RANGE
mW
μW
850
−0.001
+25
850
−0.001
+25
850
−0.001
+25
850
−0.001
+25
% FSR/V
°C
–40
+85
–40
+85
–40
+85
–40
+85
1 Based on a 10 kΩ external resistor.
2 Bypass only.
3 LDO on.
Rev. 0 | Page 5 of 48
AD9714/AD9715/AD9716/AD9717
DIGITAL SPECIFICATIONS
TMIN to TMAX, AVDD = 3.3 V, DVDD = 3.3 V, DVDDIO = 3.3 V, CVDD = 3.3 V, IOUTFS = 2 mA, maximum sample rate, unless
otherwise noted.
Table 2.
Parameter
Min
Typ
Max
Unit
DAC CLOCK INPUT (CLKIN)
VIH
VIL
2.1
3
0
V
V
0.9
Maximum Clock Rate
125
MSPS
SERIAL PERIPHERAL INTERFACE
Maximum Clock Rate (SCLK)
Minimum Pulse Width High
Minimum Pulse Width Low
25
20
20
MHz
ns
ns
INPUT DATA
1.8 V Q-Channel or DCLKIO Falling Edge
Setup
Hold
0.25
1.2
ns
ns
I-Channel or DCLKIO Rising Edge
Setup
Hold
0.13
1.1
ns
ns
3.3 V Q-Channel or DCLKIO Falling Edge
Setup
Hold
−0.2
1.5
ns
ns
I-Channel or DCLKIO Rising Edge
Setup
Hold
VIH
−0.2
1.6
3
ns
ns
V
2.1
VIL
0
0.9
V
Rev. 0 | Page 6 of 48
AD9714/AD9715/AD9716/AD9717
AC SPECIFICATIONS
TMIN to TMAX, AVDD = 3.3 V, DVDD = 3.3 V, DVDDIO = 1.8 V, CVDD = 3.3 V, IOUTFS = 2 mA, maximum sample rate, unless
otherwise noted.
Table 3.
AD9714
Min Typ
AD9715
Max Min Typ
AD9716
Max Min Typ
AD9717
Max Min Typ
Parameter
Max Unit
SPURIOUS FREE DYNAMIC RANGE (SFDR)
fDAC = 125 MSPS, fOUT = 10 MHz
fDAC = 125 MSPS, fOUT = 50 MHz
75
60
82
61
83
62
84
63
dBc
dBc
TWO-TONE INTERMODULATION
DISTORTION (IMD)
fDAC = 125 MSPS, fOUT = 10 MHz
fDAC = 125 MSPS, fOUT = 50 MHz
86
71
87
71
88
71
89
71
dBc
dBc
NOISE SPECTRAL DENSITY (NSD) EIGHT-
TONE, 500 kHz TONE SPACING
fDAC = 125 MSPS, fOUT = 10 MHz
fDAC = 125 MSPS, fOUT = 50 MHz
−129
−123
−141
−135
−149
−137
−152
−141
dBc/Hz
dBc/Hz
W-CDMA ADJACENT CHANNEL LEAKAGE
RATIO (ACLR), SINGLE CARRIER
fDAC = 61.44 MSPS, fOUT = 20 MHz
fDAC = 122.88 MSPS, fOUT = 30 MHz
−71
−72
−71
−72
−71
−72
−71
−72
dBc
dBc
TMIN to TMAX, AVDD = 1.8 V, DVDD = 3.3 V, DVDDIO = 1.8 V, CVDD = 3.3 V, IOUTFS = 2 mA, maximum sample rate, unless
otherwise noted.
Table 4.
AD9714
Min Typ
AD9715
Max Min Typ
AD9716
Max Min Typ
AD9717
Max Min Typ
Parameter
Max Unit
SPURIOUS FREE DYNAMIC RANGE (SFDR)
fDAC = 125 MSPS, fOUT = 10 MHz
fDAC = 125 MSPS, fOUT = 50 MHz
75
55
78
56
79
57
80
58
dBc
dBc
TWO-TONE INTERMODULATION
DISTORTION (IMD)
fDAC = 125 MSPS, fOUT = 10 MHz
fDAC = 125 MSPS, fOUT = 50 MHz
79
53
80
53
84
53
85
53
dBc
dBc
NOISE SPECTRAL DENSITY (NSD) EIGHT-
TONE, 500 kHz TONE SPACING
fDAC = 125 MSPS, fOUT = 10 MHz
fDAC = 125 MSPS, fOUT = 50 MHz
−132
−126
−141
−131
−146
−131
−148
−132
dBc/Hz
dBc/Hz
W-CDMA ADJACENT CHANNEL LEAKAGE
RATIO (ACLR), SINGLE CARRIER
fDAC = 61.44 MSPS, fOUT = 20 MHz
fDAC = 122.88 MSPS, fOUT = 30 MHz
−68
−68
−68
−68
−68
−68
−68
−68
dBc
dBc
Rev. 0 | Page 7 of 48
AD9714/AD9715/AD9716/AD9717
ABSOLUTE MAXIMUM RATINGS
Table 5.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Parameter
Rating
AVDD, DVDDIO, CVDD to AVSS, DVSS, CVSS
DVDD to DVSS
−0.3 V to +3.9 V
−0.3 V to +2.1 V
−0.3 V to +0.3 V
−0.3 V to +0.3 V
−0.3 V to +0.3 V
−0.3 V to AVDD + 0.3 V
−1.0 V to AVDD + 0.3 V
AVSS to DVSS, CVSS
DVSS to AVSS, CVSS
CVSS to AVSS, DVSS
VREF, FSADJQ, FSADJI, CMLQ, CMLI to AVSS
QOUTP, QOUTN, IOUTP, IOUTN, RLQP, RLQN,
RLIP, RLIN to AVSS
THERMAL RESISTANCE
−0.3 V to DVDD + 0.3 V
D13 to D0, CS, SCLK, SDIO, SDO, RESET to DVSS
CLKIN to CVSS
Table 6.
−0.3 V to CVDD + 0.3 V
–0.3 V to DVDD + 0.3 V
Package Type
θJA
Unit
CS, SCLK, SDIO, SDO to DVSS
40-Lead LFCSP (With No Airflow Movement)
29.8
°C/W
Junction Temperature
125°C
Storage Temperature Range
−65°C to +150°C
ESD CAUTION
Rev. 0 | Page 8 of 48
AD9714/AD9715/AD9716/AD9717
PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS
PIN 1
DB11
DB10
DB9
1
2
3
4
5
6
7
8
9
30 RLIN
INDICATOR
29 IOUTN
28 IOUTP
27 RLIP
DB8
AD9717
DVDDIO
DVSS
DVDD
DB7
26 AVDD
25 AVSS
24 RLQP
23 QOUTP
22 QOUTN
21 RLQN
TOP VIEW
(Not to Scale)
DB6
DB5 10
NOTES
1. THE HEAT SINK PAD IS CONNECTED TO AVSS AND
SHOULD BE SOLDERED TO THE GROUND PLANE.
EXPOSED METAL AT PACKAGE CORNERS IS
CONNECTED TO THIS PAD.
Figure 2. AD9717 Pin Configuration
Table 7. AD9717 Pin Function Descriptions
Pin No. Mnemonic
Description
1 to 4
DB[11:8]
DVDDIO
DVSS
Digital Inputs.
Digital I/O Supply Voltage (1.8 V to 3.3 V Nominal).
Digital Common.
Digital Core Supply Voltage (1.8 V). Provides a 1.8 V output when the internal LDO regulator is enabled.
Digital Inputs.
5
6
7
DVDD
8 to 14 DB[7:1]
15
16
17
18
19
20
21
22
23
24
25
26
27
28
29
30
31
32
DB0 (LSB)
DCLKIO
CVDD
CLKIN
CVSS
CMLQ
RLQN
QOUTN
QOUTP
RLQP
AVSS
AVDD
RLIP
IOUTP
IOUTN
RLIN
Digital Input (LSB).
Data Input/Output Clock. Clock used to qualify input data.
Sampling Clock Supply Voltage (1.8 V to 3.3 V). CVDD must be ≥ DVDD.
LVCMOS Sampling Clock Input.
Sampling Clock Supply Voltage Common.
Q DAC Output Common-Mode Level.
Load Resistor (500 Ω) to the CMLQ Pin.
Complementary Q DAC Current Output. Full-scale current is sourced when all data bits are 0s.
Q DAC Current Output. Full-scale current is sourced when all data bits are 1s.
Load Resistor (500 Ω) to the CMLQ Pin.
Analog Common.
Analog Supply Voltage (1.8 V to 3.3 V).
Load Resistor (500 Ω) to the CMLI Pin.
Complementary I DAC Current Output. Full-scale current is sourced when all data bits are 0s.
I DAC Current Output. Full-scale current is sourced when all data bits are 1s.
Load Resistor (500 Ω) to the CMLI Pin.
CMLI
I DAC Output Common-Mode Level.
FSADJQ/AUXQ Full-Scale Current Output Adjust for Q DAC. Connect to AVSS through a resistor.
Auxiliary Q DAC. The pin becomes the output of an optional, serial port driven, auxiliary DAC when the internal
on-chip, RSET, is enabled.
33
34
FSADJI/AUXI
Full-Scale Current Output Adjust for I DAC. Connect to AVSS through a resistor.
Auxiliary I DAC. The pin becomes the output of an optional, serial port driven, auxiliary DAC when the internal
on-chip, RSET, is enabled.
Reference Input/Output. Serves as a reference input when the internal reference is disabled. Provides a 1.0 V
reference output when in internal reference mode (a 0.1 ꢀF capacitor to AVSS is required).
REFIO
Rev. 0 | Page 9 of 48
AD9714/AD9715/AD9716/AD9717
Pin No. Mnemonic
Description
35
RESET/PINMD
Reset. In SPI mode, pulse RESET high to reset SPI registers to default values.
Pin Mode. A constant Logic 1 puts the device into pin mode.
Serial Clock. Clock input for serial port in spi mode
36
SCLK/CLKMD
Clock Mode. In pin mode, CLKMD determines phase of internal retiming clock.
DCLKIO = CLKIN: Tie to 0.
DCLKIO ≠ CLKIN: Pulse 0 to 1 to edge trigger the internal retimer (see the Retimer section).
37
38
SDIO/FORMAT Serial Port Input/Output. Bidirectional data line for serial port in spi mode.
Data Format. In pin mode, FORMAT determines data format of digital data.
CS/PWRDN
Chip Select. Active low chip select in spi mode.
Power Down. In pin mode, PWRDN powers down the device except for the SPI port.
39
40
DB13 (MSB)
DB12
Digital Input (MSB).
Digital Input.
Heat Sink Pad
The heat sink pad is connected to AVSS and should be soldered to the ground plane. Exposed metal at package
corners is connected to this pad.
Rev. 0 | Page 10 of 48
AD9714/AD9715/AD9716/AD9717
PIN 1
DB9
DB8
1
2
3
4
5
6
7
8
9
30 RLIN
INDICATOR
29 IOUTN
28 IOUTP
27 RL2P
DB7
DB6
AD9716
DVDDIO
DVSS
DVDD
DB5
26 AVDD
25 AVSS
24 RLQP
23 QOUTP
22 QOUTN
21 RLQN
TOP VIEW
(Not to Scale)
DB4
DB3 10
NOTES
1. NC = NO CONNECT
2. THE HEAT SINK PAD IS CONNECTED TO AVSS AND
SHOULD BE SOLDERED TO THE GROUND PLANE.
EXPOSED METAL AT PACKAGE CORNERS IS
CONNECTED TO THIS PAD.
Figure 3. AD9716 Pin Configuration
Table 8. AD9716 Pin Function Descriptions
Pin No. Mnemonic
Description
1 to 4
DB[9:6]
DVDDIO
DVSS
Digital Inputs.
Digital I/O Supply Voltage (1.8 V to 3.3 V Nominal).
Digital Common.
Digital Core Supply Voltage (1.8 V). Provides a 1.8 V output when in internal LDO regulator is enabled.
Digital Inputs.
5
6
7
DVDD
8 to 12 DB[5:1]
13
14,15
16
17
18
19
20
21
22
23
24
25
26
27
28
29
30
31
32
DB0 (LSB)
NC
DCLKIO
CVDD
CLKIN
CVSS
CMLQ
RLQN
QOUTN
QOUTP
RLQP
AVSS
AVDD
RL2P
IOUTP
IOUTN
RLIN
Digital Input (LSB).
No Connect. These pins are not connected to the chip.
Data Input Clock. Used to clock data in from digital source.
Sampling Clock Supply Voltage (1.8 V to 3.3 V). CVDD must be ≥ DVDD.
Sampling Clock Input.
Sampling Clock Supply Voltage Common.
Q DAC Output Common-Mode Level.
Load Resistor (500 Ω) to the CMLQ Pin.
Complementary Q DAC Current Output. Full-scale current is sourced when all data bits are 0s.
Q DAC Current Output. Full-scale current is sourced when all data bits are 1s.
Load Resistor (500 Ω) to the CMLQ Pin.
Analog Common.
Analog Supply Voltage (1.8 V to 3.3 V).
Load Resistor (500 Ω) to the CMLI Pin.
Complementary I DAC Current Output. Full-scale current is sourced when all data bits are 0s.
I DAC Current Output. Full-scale current is sourced when all data bits are 1s.
Load Resistor (500 Ω) to the CMLI Pin.
CMLI
I DAC Output Common-Mode Level.
FSADJQ/AUXQ Full-Scale Current Output Adjust for Q DAC. Connect to AVSS through a resistor.
Auxiliary Q DAC. The pin becomes the output of an optional, serial port driven, auxiliary DAC when the internal
on-chip, RSET, is enabled.
33
34
FSADJI/AUXI
Full-Scale Current Output Adjust for I DAC. Connect to AVSS through a resistor.
Auxiliary I DAC. The pin becomes the output of an optional, serial port driven, auxiliary DAC when the internal
on-chip, RSET, is enabled.
Reference Input/Output. Serves as a reference input when the internal reference is disabled. Provides a 1.0 V
reference output when in internal reference mode (a 0.1 ꢀF capacitor to AVSS is required).
REFIO
Rev. 0 | Page 11 of 48
AD9714/AD9715/AD9716/AD9717
Pin No. Mnemonic
Description
35
RESET/PINMD
Reset in SPI Mode. Pulse high to reset SPI registers to default values.
Pin Mode. A constant Logic 1 puts the device into pin mode.
Serial Clock. Clock input for serial port in spi mode.
36
SCLK/CLKMD
Clock Mode. In pin mode, CLKMD determines phase of internal retiming clock.
DCLKIO = CLKIN: Tie to 0.
DCLKIO ≠ CLKIN: Pulse 0 to 1 to edge trigger the internal retimer (see the Retimer section).
37
38
SDIO/FORMAT Serial Port Input/Output. Bidirectional data line for serial port in spi mode.
Data Format. In pin mode, FORMAT determines data format of digital data.
CS/PWRDN
Chip Select. Active low chip select in spi mode.
Power Down. In pin mode, PWRDN powers down the device except for the SPI port.
39
40
DB11 (MSB)
DB10
Digital Input (MSB).
Digital Input.
Heat Sink Pad
The heat sink pad is connected to AVSS and should be soldered to the ground plane. Exposed metal at package
corners is connected to this pad.
Rev. 0 | Page 12 of 48
AD9714/AD9715/AD9716/AD9717
PIN 1
DB7
DB6
1
2
3
4
5
6
7
8
9
30 RLIN
INDICATOR
29 IOUTP
28 IOUTN
27 RL2N
26 AVDD
25 AVSS
24 RL1P
23 QOUTP
22 QOUTN
21 RL1N
DB5
DB4
AD9715
DVDDIO
DVSS
DVDD
DB3
TOP VIEW
(Not to Scale)
DB2
DB1 10
NOTES
1. NC = NO CONNECT
2. THE HEAT SINK PAD IS CONNECTED TO AVSS AND
SHOULD BE SOLDERED TO THE GROUND PLANE.
EXPOSED METAL AT PACKAGE CORNERS IS
CONNECTED TO THIS PAD.
Figure 4. AD9715 Pin Configuration
Table 9. AD9715 Pin Function Descriptions
Pin No.
1 to 4
5
Mnemonic
DB[7:4]
DVDDIO
DVSS
Description
Digital Inputs.
Digital I/O Supply Voltage (1.8 V to 3.3 V Nominal).
Digital Common.
6
7
DVDD
DB[3:1]
DB0 (LSB)
NC
DCLKIO
CVDD
CLKIN
CVSS
Digital Core Supply Voltage (1.8 V). Provides a 1.8 V output when in internal LDO regulator is enabled.
Digital Inputs.
Digital Input (LSB).
No Connect. These pins are not connected to the chip.
Data Input Clock. Used to clock data in from digital source.
Sampling Clock Supply Voltage (1.8 V to 3.3 V). CVDD must be ≥ DVDD.
Sampling Clock Input.
8 to 10
11
12 to 15
16
17
18
19
Sampling Clock Supply Voltage Common.
20
21
CMLQ
RL1N
Q DAC Output Common-Mode Level.
Load Resistor (500 Ω) to the CMLQ Pin.
22
23
24
QOUTN
QOUTP
RL1P
Complementary Q DAC Current Output. Full-scale current is sourced when all data bits are 0s.
Q DAC Current Output. Full-scale current is sourced when all data bits are 1s.
Load Resistor (500 Ω) to the CMLQ Pin.
25
AVSS
Analog Common.
26
27
AVDD
RL2N
Analog Supply Voltage (1.8 V to 3.3 V).
Load Resistor (500 Ω) to the CMLI Pin.
28
29
30
IOUTN
IOUTP
RLIN
Complementary I DAC Current Output. Full-scale current is sourced when all data bits are 0s.
I DAC Current Output. Full-scale current is sourced when all data bits are 1s.
Load Resistor (500 Ω) to the CMLI Pin.
31
CMLI
I DAC Output Common-Mode Level.
32
FSADJQ/AUXQ Full-Scale Current Output Adjust for Q DAC. Connect to AVSS through a resistor.
Auxiliary Q DAC. The pin becomes the output of an optional, serial port driven, auxiliary DAC when the
internal on-chip, RSET, is enabled.
33
34
FSADJI/AUXI
Full-Scale Current Output Adjust for I DAC. Connect to AVSS through a resistor.
Auxiliary I DAC. The pin becomes the output of an optional, serial port driven, auxiliary DAC when the internal
on-chip, RSET, is enabled.
Reference Input/Output. Serves as reference input when the internal reference is disabled. Provides a 1.0 V
reference output when in internal reference mode (a 0.1 μF capacitor to AVSS is required).
REFIO
Rev. 0 | Page 13 of 48
AD9714/AD9715/AD9716/AD9717
Pin No.
Mnemonic
Description
35
RESET/PINMD
Reset in SPI Mode. Pulse high to reset SPI registers to default values.
Pin Mode. A constant Logic 1 puts device into pin mode.
Serial Clock. Clock input for serial port in spi mode.
36
SCLK/CLKMD
Clock Mode. In pin mode, CLKMD determines phase of internal retiming clock.
DCLKIO = CLKIN: Tie to 0.
DCLKIO ≠ CLKIN: Pulse 0 to 1 to edge trigger the internal retimer (see the Retimer section).
Serial Port Input/Output. Bidirectional data line for serial port in spi mode.
Data Format. In pin mode, FORMAT determines data format of digital data.
Chip Select. Active low chip select in spi mode.
37
38
SDIO/FORMAT
CS/PWRDN
Power Down. In pin mode, PWRDN powers down the device except for the SPI port.
Digital Input (MSB).
Digital Input.
39
40
DB9 (MSB)
DB8
Heat Sink Pad
The heat sink pad is connected to AVSS and should be soldered to the ground plane. Exposed metal at
package corners is connected to this pad.
Rev. 0 | Page 14 of 48
AD9714/AD9715/AD9716/AD9717
PIN 1
DB5
DB4
1
2
3
4
5
6
7
8
9
30 RLIN
INDICATOR
29 IOUTP
28 IOUTN
27 RL2N
26 AVDD
25 AVSS
24 RL1P
23 QOUTP
22 QOUTN
21 RL1N
DB3
DB2
AD9714
DVDDIO
DVSS
TOP VIEW
DVDD
DB1
(Not to Scale)
DB0 (LSB)
NC 10
NOTES
1. NC = NO CONNECT
2. THE HEAT SINK PAD IS CONNECTED TO AVSS AND
SHOULD BE SOLDERED TO THE GROUND PLANE.
EXPOSED METAL AT PACKAGE CORNERS IS
CONNECTED TO THIS PAD.
Figure 5. AD9714 Pin Configuration
Table 10. AD9714 Pin Function Descriptions
Pin No.
1 to 4
5
Mnemonic
DB[5:2]
DVDDIO
DVSS
Description
Digital Inputs.
Digital I/O Supply Voltage (1.8 V to 3.3 V Nominal).
Digital Common.
6
7
8
DVDD
DB1
Digital Core Supply Voltage (1.8 V). Provides a 1.8 V output when the internal LDO regulator is enabled.
Digital Inputs.
9
DB0 (LSB)
NC
DCLKIO
CVDD
CLKIN
CVSS
Digital Input (LSB).
10 to 15
16
17
18
19
No Connect. These pins are not connected to the chip.
Data Input Clock. Used to clock data in from digital source.
Sampling Clock Supply Voltage (1.8 V to 3.3 V). CVDD must be ≥ DVDD.
Sampling Clock Input.
Sampling Clock Supply Voltage Common.
20
21
CMLQ
RL1N
Q DAC Output Common-Mode Level.
Load Resistor (500 Ω) to the CMLQ Pin.
22
23
24
QOUTN
QOUTP
RL1P
Complementary Q DAC Current Output. Full-scale current is sourced when all data bits are 0s.
Q DAC Current Output. Full-scale current is sourced when all data bits are 1s.
Load Resistor (500 Ω) to the CMLQ Pin.
25
AVSS
Analog Common.
26
27
AVDD
RL2N
Analog Supply Voltage (1.8 V to 3.3 V).
Load Resistor (500 Ω) to the CMLI Pin.
28
29
30
IOUTN
IOUTP
RLIN
Complementary I DAC Current Output. Full-scale current is sourced when all data bits are 0s.
I DAC Current Output. Full-scale current is sourced when all data bits are 1s.
Load Resistor (500 Ω) to the CMLI Pin.
31
CMLI
I DAC Output Common-Mode Level.
32
FSADJQ/AUXQ Full-Scale Current Output Adjust for Q DAC. Connect to AVSS through a resistor.
Auxiliary Q DAC. The pin becomes the output of an optional, serial port driven, auxiliary DAC when the
internal on-chip, RSET, is enabled.
33
34
FSADJI/AUXI
Full-Scale Current Output Adjust for I DAC. Connect to AVSS through a resistor.
Auxiliary I DAC. The pin becomes the output of an optional, serial port driven, auxiliary DAC when the internal
on-chip, RSET, is enabled.
Reference Input/Output. Serves as reference input when the internal reference is disabled. Provides a 1.0 V
reference output when in internal reference mode (a 0.1 ꢀF capacitor to AVSS is required).
REFIO
Rev. 0 | Page 15 of 48
AD9714/AD9715/AD9716/AD9717
Pin No.
Mnemonic
Description
35
RESET/PINMD
Reset in SPI Mode. Pulse high to reset SPI registers to default values.
Pin Mode. A constant Logic 1 puts device into pin mode.
Serial Clock. Clock input for serial port in spi mode.
36
SCLK/CLKMD
Clock Mode. In pin mode, CLKMD determines phase of internal retiming clock.
DCLKIO = CLKIN: Tie to 0.
DCLKIO ≠ CLKIN: Pulse 0 to 1 to edge trigger the internal retimer (see the Retimer section).
Serial Port Input/Output. Bidirectional data line for serial port in spi mode.
Data Format. In pin mode, FORMAT determines data format of digital data.
Chip Select. Active low chip select in spi mode.
37
38
SDIO/FORMAT
CS/PWRDN
Power Down. In pin mode, PWRDN powers down the device except for the SPI port.
Digital Input (MSB).
Digital Input.
39
40
DB7 (MSB)
DB6
Heat Sink Pad
The heat sink pad is connected to AVSS and should be soldered to the ground plane. Exposed metal at
package corners is connected to this pad.
Rev. 0 | Page 16 of 48
AD9714/AD9715/AD9716/AD9717
TYPICAL PERFORMANCE CHARACTERISTICS
AVDD, DVDD, DVDDIO, CVDD = 1.8 V, IOUTFS = 2 mA, maximum sample rate, unless otherwise noted.
1.5
1.0
0.5
0
1.5
1.0
0.5
0
–0.5
–1.0
–1.5
–0.5
–1.0
–1.5
0
2048
4096
6144
8192 10240 12288 14336 16384
CODE
0
2048
4096
6144
8192 10240 12288 14336 16384
CODE
Figure 6. AD9717 Precalibration INL at 1.8 V
Figure 9. AD9717 Postcalibration INL at 1.8 V
1.5
1.0
0.5
0
1.5
1.0
0.5
0
–0.5
–1.0
–1.5
–0.5
–1.0
–1.5
0
2048
4096
6144
8192 10240 12288 14336 16384
CODE
0
2048
4096
6144
8192 10240 12288 14336 16384
CODE
Figure 7. AD9717 Precalibration DNL at 1.8 V
Figure 10. AD9717 Postcalibration DNL at 1.8 V
1.75
1.25
1.75
1.25
0.75
0.75
0.25
0.25
–0.25
–0.75
–1.25
–1.75
–0.25
–0.75
–1.25
–1.75
0
2048
4096
6144
8192 10240 12288 14336 16384
CODE
0
2048
4096
6144
8192 10240 12288 14336 16384
CODE
Figure 8. AD9717 Precalibration INL at 3.3 V
Figure 11. AD9717 Postcalibration INL at 3.3 V
Rev. 0 | Page 17 of 48
AD9714/AD9715/AD9716/AD9717
1.75
1.75
1.25
1.25
0.75
0.75
0.25
0.25
–0.25
–0.75
–1.25
–1.75
–0.25
–0.75
–1.25
–1.75
0
2048
4096
6144
8192 10240 12288 14336 16384
CODE
0
2048
4096
6144
8192 10240 12288 14336 16384
CODE
Figure 12. AD9717 Precalibration DNL at 3.3 V
Figure 15. AD9717 Postcalibration DNL at 3.3 V
0.4
0.4
0.3
0.2
0.1
0
0.3
0.2
0.1
0
–0.1
–0.2
–0.3
–0.4
–0.1
–0.2
–0.3
–0.4
0
512
1024
1536
2048
2560
3072 3584
4096
0
512
1024
1536
2048
2560
3072 3584
4096
CODE
CODE
Figure 13. AD9716 Precalibration INL at 1.8 V
Figure 16. AD9716 Postcalibration INL at 1.8 V
0.4
0.4
0.3
0.2
0.1
0
0.3
0.2
0.1
0
–0.1
–0.2
–0.3
–0.4
–0.1
–0.2
–0.3
–0.4
0
512
1024
1536
2048
2560
3072 3584
4096
0
512
1024
1536
2048
2560
3072 3584
4096
CODE
CODE
Figure 14. AD9716 Precalibration DNL at 1.8 V
Figure 17. AD9716 Postcalibration DNL at 1.8 V
Rev. 0 | Page 18 of 48
AD9714/AD9715/AD9716/AD9717
0.4
0.4
0.3
0.2
0.1
0
0.3
0.2
0.1
0
–0.1
–0.2
–0.3
–0.4
–0.1
–0.2
–0.3
–0.4
0
512
1024
1536
2048
2560
3072 3584
4096
0
512
1024
1536
2048
2560
3072 3584
4098
CODE
CODE
Figure 18. AD9716 Precalibration INL at 3.3 V
Figure 21. AD9716 Postcalibration INL at 3.3 V
0.4
0.4
0.3
0.2
0.1
0
0.3
0.2
0.1
0
–0.1
–0.2
–0.3
–0.4
–0.1
–0.2
–0.3
–0.4
0
512
1024
1536
2048
2560
3072 3584
4096
0
512
1024
1536
2048
2560
3072 3584
4096
CODE
CODE
Figure 19. AD9716 Precalibration DNL at 3.3 V
Figure 22. AD9716 Postcalibration DNL at 3.3 V
0.13
0.13
0.08
0.08
0.03
0.03
–0.02
–0.07
–0.02
–0.07
–0.12
–0.12
0
128
256
384
512
640
768
896
1024
0
128
256
384
512
640
768
896
1024
CODE
CODE
Figure 20. AD9715 Precalibration INL at 1.8 V
Figure 23. AD9715 Postcalibration INL at 1.8 V
Rev. 0 | Page 19 of 48
AD9714/AD9715/AD9716/AD9717
0.13
0.13
0.08
0.08
0.03
0.03
–0.02
–0.07
–0.12
–0.02
–0.07
–0.12
0
128
256
384
512
640
768
896
1024
0
128
256
384
512
640
768
896
1024
CODE
CODE
Figure 24. AD9715 Precalibration DNL at 1.8 V
Figure 27. AD9715 Postcalibration DNL at 1.8 V
0.13
0.08
0.13
0.08
0.03
0.03
–0.02
–0.07
–0.02
–0.07
–0.12
–0.12
0
128
256
384
512
640
768
896
1024
0
128
256
384
512
640
768
896
1024
CODE
CODE
Figure 25. AD9715 Precalibration INL at 3.3 V
Figure 28. AD9715 Postcalibration INL at 3.3 V
0.13
0.08
0.13
0.08
0.03
0.03
–0.02
–0.07
–0.02
–0.07
–0.12
–0.12
0
128
256
384
512
640
768
896
1024
0
128
256
384
512
640
768
896
1024
CODE
CODE
Figure 26. AD9715 Precalibration DNL at 3.3 V
Figure 29. AD9715 Postcalibration DNL at 3.3 V
Rev. 0 | Page 20 of 48
AD9714/AD9715/AD9716/AD9717
0.025
0.020
0.015
0.010
0.005
0
0.025
0.020
0.015
0.010
0.005
0
–0.005
–0.010
–0.015
–0.020
–0.025
–0.005
–0.010
–0.015
–0.020
–0.025
0
16 32 48 64 80 96 112 128 144 160 176 192 208 224 240 256
CODE
0
16 32 48 64 80 96 112 128 144 160 176 192 208 224 240 256
CODE
Figure 30. AD9714 Precalibration INL at 1.8 V
Figure 33. AD9714 Postcalibration INL at 1.8 V
0.025
0.020
0.015
0.010
0.005
0
0.025
0.020
0.015
0.010
0.005
0
–0.005
–0.010
–0.015
–0.020
–0.025
–0.005
–0.010
–0.015
–0.020
–0.025
0
16 32 48 64 80 96 112 128 144 160 176 192 208 224 240 256
CODE
0
16 32 48 64 80 96 112 128 144 160 176 192 208 224 240 256
CODE
Figure 31. AD9714 Precalibration DNL at 1.8 V
Figure 34. AD9714 Postcalibration DNL at 1.8 V
0.025
0.020
0.015
0.010
0.005
0
0.025
0.020
0.015
0.010
0.005
0
–0.005
–0.010
–0.015
–0.020
–0.025
–0.005
–0.010
–0.015
–0.020
–0.025
0
16 32 48 64 80 96 112 128 144 160 176 192 208 224 240 256
CODE
0
16 32 48 64 80 96 112 128 144 160 176 192 208 224 240 256
CODE
Figure 32. AD9714 Precalibration INL at 3.3 V
Figure 35. AD9714 Postcalibration INL at 3.3 V
Rev. 0 | Page 21 of 48
AD9714/AD9715/AD9716/AD9717
0.025
0.020
0.015
0.010
0.005
0
0.025
0.020
0.015
0.010
0.005
0
–0.005
–0.010
–0.015
–0.020
–0.025
–0.005
–0.010
–0.015
–0.020
–0.025
0
16 32 48 64 80 96 112 128 144 160 176 192 208 224 240 256
CODE
0
16 32 48 64 80 96 112 128 144 160 176 192 208 224 240 256
CODE
Figure 36. AD9714 Precalibration DNL at 3.3 V
Figure 39. AD9714 Postcalibration DNL at 3.3 V
–126
–126
AD9714
AD9714
–129
–132
–135
–138
–132
–138
–144
–150
AD9715
AD9715
–141
–144
AD9716
AD9717
–147
–150
–153
AD9716
AD9717
–156
–156
0
5
10
15
20
25
30
35
40
45
50
55
0
5
10
15
20
25
30
35
40
45
50
55
fOUT (MHz)
fOUT (MHz)
Figure 37. Noise Spectral Density at 3.3 V
Figure 40. AD9714/AD9715/AD9716/AD9717 Noise Spectral Density at 1.8 V
93
86
AD9715
90
87
84
80
74
AD9715
81
78
75
72
69
66
AD9716
68
AD9714
AD9717
62
56
AD9714
AD9716
AD9717
50
0
5
10 15 20 25 30 35 40 45 50 55 60
fOUT (MHz)
0
5
10 15 20 25 30 35 40 45 50 55 60
fOUT (MHz)
Figure 38. AD9714/AD9715/AD9716/AD9717 SFDR at 3.3 V
Figure 41. AD9714/AD9715/AD9716/AD9717 SFDR at 1.8 V
Rev. 0 | Page 22 of 48
AD9714/AD9715/AD9716/AD9717
–61
–64
100
1mA
AD9717
94
88
AD9716
2mA
AD9715
–67
–70
–73
82
AD9714
10
76
70
15
20
25
30
35
40
45
5
15
20
25
30
35
40
45
50
fOUT (MHz)
fOUT (MHz)
Figure 42. AD9714/AD9715/AD9716/AD9717 IMD at 3.3 V
Figure 45. 1 Carrier ACLR First Adjacent Channel at 1.8 V
–60
88
–63
–66
–69
–72
–75
82
76
70
64
58
52
1mA
2mA
AD9714
AD9715
AD9716
AD9717
4mA
–78
15
20
25
30
fOUT (MHz)
35
40
45
5
10
15
20
25
30
35
40
45
50
fOUT (MHz)
Figure 46. 1 Carrier ACLR First Adjacent Channel at 3.3 V
Figure 43. AD9714/AD9715/AD9716/AD9717 IMD at 1.8 V
25
TOTAL CURRENT @ 1mA OUT
TOTAL CURRENT @ 2mA OUT
30
20
20
15
10
5
TOTAL CURRENT @ 4mA OUT
TOTAL CURRENT @ 2mA OUT
AVDD @ 4mA OUT
AVDD @ 2mA OUT
TOTAL CURRENT @ 1mA OUT
AVDD @ 2mA OUT
AVDD @ 1mA OUT
DVDD
10
AVDD @ 1mA OUT
DVDD
CVDD
CVDD
0
0
0
20
40
60
80
100
120
140
0
20
40
60
80
100
120
140
fOUT (MHz)
fOUT (MHz)
Figure 44. Supply Current vs. Frequency at 1.8 V
Figure 47. Supply Current vs. Frequency at 3.3 V
Rev. 0 | Page 23 of 48
AD9714/AD9715/AD9716/AD9717
–60
–60
–65
1mA PRECAL
1mA PRECAL
–65
2mA POSTCAL
2mA POSTCAL
1mA POSTCAL
1mA POSTCAL
–70
2mA PRECAL
4mA POSTCAL
–75
–70
–75
2mA PRECAL
4mA PRECAL
–80
15
25
35
45
20
30
fOUT (MHz)
40
fOUT (MHz)
Figure 48. AD9717 1-Carrier W-CDMA First Adjacent Channel ACLR 3.3 V
Figure 51. AD9717 1-Carrier W-CDMA Third Adjacent Channel ACLR 1.8 V
–55
–60
1mA PRECAL
1mA PRECAL
1mA POSTCAL
1mA POSTCAL
–65
2mA PRECAL
–60
2mA POSTCAL
–70
2mA POSTCAL
–65
–70
2mA PRECAL
–75
15
15
20
25
30
35
40
25
35
45
fOUT (MHz)
fOUT (MHz)
Figure 52. AD9717 1-Carrier W-CDMA First Adjacent Channel ACLR 1.8 V
Figure 49. AD9717 1-Carrier W-CDMA First Adjacent Channel ACLR 1.8 V
–60
–60
1mA PRECAL
1mA PRECAL
–65
–65
2mA PRECAL
1mA POSTCAL
–70
1mA POSTCAL
2mA PRECAL
–70
2mA POSTCAL
4mA PRECAL
–75
2mA POSTCAL
4mA POSTCAL
–80
15
–75
15
25
35
45
25
35
45
fOUT (MHz)
fOUT (MHz)
Figure 53. AD9717 1-Carrier W-CDMA Second Adjacent Channel ACLR 3.3 V
Figure 50. AD9717 1-Carrier W-CDMA Second Adjacent Channel ACLR 1.8 V
Rev. 0 | Page 24 of 48
AD9714/AD9715/AD9716/AD9717
–60
–55
–60
1mA PRECAL
1mA PRECAL
–65
–70
1mA POSTCAL
2mA PRECAL
1mA POSTCAL
2mA PRECAL
4mA PRECAL
–65
–70
2mA POSTCAL
–75
–80
2mA POSTCAL
4mA POSTCAL
20
30
fOUT (MHz)
40
20
25
30
fOUT (MHz)
35
40
Figure 54. AD9717 1-Carrier W-CDMA Third Adjacent Channel ACLR 3.3 V
Figure 57. AD9717 2-Carrier W-CDMA Third Adjacent Channel ACLR 1.8 V
–55
1mA POSTCAL
1mA PRECAL
–60
2mA PRECAL
–65
2mA POSTCAL
4mA PRECAL
–70
4mA POSTCAL
–75
CENTER 22.90MHz
SPAN 38.84MHz
15
20
25
30
35
40
VBW 300kHz
fOUT (MHz)
Figure 55. AD9717 1-Carrier W-CDMA First Adjacent Channel ACLR 3.3 V
Figure 58. AD9717 ACLR 1-Carrier 1.8 V
–55
–55
–60
–65
–70
–75
1mA PRECAL
1mA PRECAL
–60
1mA POSTCAL
2mA PRECAL
1mA POSTCAL
2mA PRECAL
–65
2mA POSTCAL
2mA POSTCAL
4mA POSTCAL
35
4mA PRECAL
25
–70
15
20
25
30
35
40
15
20
30
40
fOUT (MHz)
fOUT (MHz)
Figure 56. AD9717 2-Carrier W-CDMA Second Adjacent Channel ACLR 1.8 V
Figure 59. AD9717 2-Carrier W-CDMA Second Adjacent Channel ACLR 3.3 V
Rev. 0 | Page 25 of 48
AD9714/AD9715/AD9716/AD9717
–55
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
–110
1mA PRECAL
–60
1mA POSTCAL
2mA PRECAL
–65
2mA POSTCAL
–70
4mA POSTCAL
4mA PRECAL
–75
20
25
30
35
40
fOUT (MHz)
START 1MHz
1.5MHz/DIV
STOP 16MHz
Figure 60. AD9717 2-Carrier W-CDMA Third Adjacent Channel ACLR 3.3 V
Figure 63. AD9717 Single Tone 1.8 V
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
–110
CENTER 22.90MHz
SPAN 38.84MHz
START 1MHz
1.5MHz/DIV
STOP 16MHz
VBW 300kHz
Figure 61. AD9717 ACLR 1-Carrier 3.3 V
Figure 64. AD9717 Two Tone 1.8 V
CENTER 22.90MHz
SPAN 38.84MHz
CENTER 22.90MHz
SPAN 38.84MHz
VBW 300kHz
VBW 300kHz
Figure 62.AD9717 ACLR 2-Carrier 1.8 V
Figure 65. AD9717 ACLR 2-Carrier 3.3 V
Rev. 0 | Page 26 of 48
AD9714/AD9715/AD9716/AD9717
91
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
–110
88
85
82
0dB
–3dB
–6dB
79
76
5
10
15
20
25
30
35
40
45
50
fIN (MHz)
START 1MHz
1.4MHz/DIV
STOP 15MHz
Figure 66. AD9717 Single Tone, 3.3 V
Figure 69. IMD vs. Digital Input Level 3.3 V
90
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
–110
85
80
75
70
–6dB
–3dB
65
60
55
50
0dB
0
10
20
30
40
50
60
START 1MHz
1.4MHz/DIV
STOP 15MHz
fIN (MHz)
Figure 67. AD9717 Two Tone, 3.3 V
Figure 70. SFDR vs. Digital Input Level 3.3 V
88
82
90
85
80
75
70
–6dB
76
70
–3dB
–6dB
0dB
–3dB
65
60
55
50
64
0dB
58
52
5
10
15
20
25
F
30
35
40
45
50
0
10
20
30
40
50
60
(MHz)
fIN (MHz)
IN
Figure 68. IMD vs. Digital Input Level 1.8 V
Figure 71. SFDR vs. Digital Input Level 1.8 V
Rev. 0 | Page 27 of 48
AD9714/AD9715/AD9716/AD9717
108
90
4mA
102
85
80
75
70
+85°C
96
+25°C
90
–40°C
2mA
84
78
72
66
65
60
55
50
1mA
60
54
0
5
10 15 20 25
30 35 40 45 50 55 60
0
10
20
30
40
50
60
fOUT (MHz)
fOUT (MHz)
Figure 72. SFDR Over Temperature at 3.3 V
Figure 74. SFDR vs. fOUT
90
85
1.0
INL
0.8
0.6
0.4
80
75
4mA
DNL
0.2
0
1mA
70
65
60
–0.2
–0.4
5
10
15
20
25
30
35
40
45
50
0
128
256
384
512
640
768
896
1024
fOUT (MHz)
CODE
Figure 73. IMD vs. fOUT, FCLK = 125 MHz
Figure 75. AUXDAC DNL and INL
Rev. 0 | Page 28 of 48
AD9714/AD9715/AD9716/AD9717
TERMINOLOGY
Linearity Error or Integral Nonlinearity (INL)
Linearity error is defined as the maximum deviation of the
actual analog output from the ideal output, determined by
a straight line drawn from zero scale to full scale.
Power Supply Rejection
Power supply rejection is the maximum change in the full-scale
output as the supplies are varied from minimum to maximum
specified voltages.
Differential Nonlinearity (DNL)
DNL is the measure of the variation in analog value, normalized
to full scale, associated with a 1 LSB change in digital input code.
Settling Time
Settling time is the time required for the output to reach and
remain within a specified error band around its final value,
measured from the start of the output transition.
Monotonicity
A DAC is monotonic if the output either increases or remains
constant as the digital input increases.
Spurious Free Dynamic Range (SFDR)
SFDR is the difference, in decibels, between the peak amplitude
of the output signal and the peak spurious signal between dc
and the frequency equal to half the input data rate.
Offset Error
Offset error is the deviation of the output current from the
ideal of zero. For IOUTP, 0 mA output is expected when the
inputs are all 0. For IOUTN, 0 mA output is expected when all
inputs are set to 1.
Total Harmonic Distortion (THD)
THD is the ratio of the rms sum of the first six harmonic
components to the rms value of the measured fundamental.
It is expressed as a percentage or in decibels.
Gain Error
Gain error is the difference between the actual and ideal output
span. The actual span is determined by the difference between
the output when all inputs are set to 1 and the output when all
inputs are set to 0.
Signal-to-Noise Ratio (SNR)
SNR is the ratio of the rms value of the measured output signal
to the rms sum of all other spectral components below the
Nyquist frequency, excluding the first six harmonics and dc.
The value for SNR is expressed in decibels.
Output Compliance Range
Output compliant range is the range of allowable voltage at
the output of a current-output DAC. Operation beyond the
maximum compliance limits can cause either output stage
saturation or breakdown, resulting in nonlinear performance.
Adjacent Channel Leakage Ratio (ACLR)
ACLR is the ratio in dBc between the measured power within
a channel relative to its adjacent channel.
Complex Image Rejection
Temperature Drift
Temperature drift is specified as the maximum change from
the ambient value (25°C) to the value at either TMIN or TMAX
For offset and gain drift, the drift is reported in ppm of full-
scale range per degree Celsius (ppm FSR/°C). For reference
drift, the drift is reported in parts per million per degree
Celsius (ppm/°C).
In a traditional two-part upconversion, two images are created
around the second IF frequency. These images have the effect of
wasting transmitter power and system bandwidth. By placing
the real part of a second complex modulator in series with the
first complex modulator, either the upper or lower frequency
image near the second IF can be rejected.
.
Rev. 0 | Page 29 of 48
AD9714/AD9715/AD9716/AD9717
THEORY OF OPERATION
AD9714/AD9715/
AD9716/AD9717
1V
SPI
INTERFACE
DB11
R
R
SET
SET
16kΩ
16kΩ
R
CM
1kΩ TO
250Ω
DB10
RLIN
500Ω
500Ω
10kΩ
DB9
DB8
IOUTN
IOUTP
I
I DAC
REF
100µA
BAND
GAP
RLIP
AUX1DAC
AUX2DAC
DVDDIO
1 INTO 2
AVDD
AVSS
RLQP
INTERLEAVED
DATA
INTERFACE
I DATA
DVSS
500Ω
500Ω
DVDD
DB7
1.8V
LDO
QOUTP
QOUTN
Q DATA
Q DAC
RLQN
CLOCK
DIST
DB6
R
CM
1kΩ TO
250Ω
DB5
Figure 76. Simplified Block Diagram
Figure 76 shows a simplified block diagram of the AD9714/
AD9715/AD9716/AD9717 that consists of two DACs, digital
control logic, and a full-scale output current control. The DAC
contains a PMOS current source array capable of providing a
nominal full-scale current (IOUTFS) of 2 mA and a maximum
of 4 mA. The array is divided into 31 equal currents that make
up the five most significant bits (MSBs). The next four bits, or
middle bits, consist of 15 equal current sources whose value is
1/16 of an MSB current source. The remaining LSBs are binary
weighted fractions of the current sources of the middle bits.
Implementing the middle and lower bits with current sources,
instead of an R-2R ladder, enhances its dynamic performance
for multitone or low amplitude signals and helps maintain
the high output impedance of the DAC (that is, >200 MΩ).
range. The core digital section, which is powered optionally by
either the on-chip LDO or through DVDD (Pin 7), is capable of
operating at a rate of up to 125 MSPS. It consists of edge-triggered
latches and segment decoding logic circuitry. The analog section
includes the PMOS current sources, the associated differential
switches, a 1.0 V band gap voltage reference, and a reference
control amplifier.
Each DAC full-scale output current is regulated by the reference
control amplifier and can be set from 1 mA to 4 mA via an external
resistor, RSET, connected to its full-scale adjust pin (FSADJ).
The external resistor, in combination with both the reference
control amplifier and voltage reference, VREFIO, sets the reference
current, IREF, which is replicated to the segmented current sources
with the proper scaling factor. The full-scale current, IOUTFS, is
All of these current sources are switched to one or the other
of the two output nodes (IOUTP or IOUTN) via PMOS differential
current switches. The switches are based on the architecture
that was pioneered in the AD976x family, with further refine-
ments to reduce distortion contributed by the switching transient.
This switch architecture also reduces various timing errors and
provides matching complementary drive signals to the inputs of
the differential current switches.
32 × IREF
.
Optional on-chip RSET resistors are provided that can be pro-
grammed between a nominal value of 8 kΩ to 32 kΩ (4 mA to
1 mA IOUTFS ).
The AD9714/AD9715/AD9716/AD9717 provide the option of
setting the output common mode to a value other than ACOM
via the output common-mode pins (CMLI and CMLQ). This
facilitates directly interfacing the output of the AD9714/AD9715/
AD9716/AD9717 to components that require common-mode
levels greater than 0 V.
The analog and digital I/O sections of the AD9714/AD9715/
AD9716/AD9717 have separate power supply inputs (AVDD and
DVDDIO) that can operate independently over a 1.7 V to 3.5 V
Rev. 0 | Page 30 of 48
AD9714/AD9715/AD9716/AD9717
SERIAL PERIPHERAL INTERFACE (SPI)
The serial port of the AD9714/AD9715/AD9716/AD9717 is a
flexible, synchronous serial communications port allowing easy
interfacing to many industry-standard microcontrollers and
microprocessors. The serial I/O is compatible with most synchron-
ous transfer formats, including both the Motorola SPI® and Intel®
SSR protocols. The interface allows read/write access to all registers
that configure the AD9714/AD9715/AD9716/AD9717. Single or
multiple byte transfers are supported, as well as MSB first or
LSB first transfer formats. The serial interface port of the AD9714/
AD9715/AD9716/AD9717 is configured as a single I/O pin on
the SDIO pin.
INSTRUCTION BYTE
The instruction byte contains the information shown in Table 11.
Table 11.
MSB
DB7
R/W
LSB
DB6
DB5
DB4 DB3 DB2 DB1 DB0
N1
N0
A4
A3
A2
A1
A0
W
R/ (Bit 7 of the instruction byte) determines whether a read or a
write data transfer occurs after the instruction byte write. Logic 1
indicates a read operation. Logic 0 indicates a write operation.
N1 and N0 (Bit 6 and Bit 5 of the instruction byte) determine the
number of bytes to be transferred during the data transfer cycle.
The bit decodes are shown in Table 12.
GENERAL OPERATION OF THE SERIAL INTERFACE
There are two phases to a communications cycle on the AD9714/
AD9715/AD9716/AD9717. Phase 1 is the instruction cycle, which
is the writing of an instruction byte into the AD9714/AD9715/
AD9716/AD9717, coinciding with the first eight SCLK rising
edges. In Phase 2, the instruction byte provides the serial port
controller of the AD9714/AD9715/AD9716/AD9717 with infor-
mation regarding the data transfer cycle. The Phase 1 instruction
byte defines whether the upcoming data transfer is a read or write,
the number of bytes in the data transfer, and the starting register
address for the first byte of the data transfer. The first eight SCLK
rising edges of each communication cycle are used to write the
instruction byte into the AD9714/AD9715/AD9716/AD9717.
Table 12. Byte Transfer Count
N1
N0
Description
0
0
Transfer 1 byte
Transfer 2 bytes
Transfer 3 bytes
Transfer 4 bytes
0
1
1
0
1
1
A4, A3, A2, A1, and A0 (Bit 4, Bit 3, Bit 2, Bit 1, and Bit 0 of the
instruction byte) determine which register is accessed during the
data transfer portion of the communications cycle. For multi-
byte transfers, this address is the starting byte address. The
following register addresses are generated internally, based
on the LSBFIRST bit (Register 0x00, Bit 6).
A Logic 1 on Pin 35 (RESET/PINMD), followed by a Logic 0,
resets the SPI port timing to the initial state of the instruction
cycle. This is true regardless of the present state of the internal
registers or the other signal levels present at the inputs to the
SPI port. If the SPI port is in the midst of an instruction cycle
or a data transfer cycle, none of the present data is written.
SERIAL INTERFACE PORT PIN DESCRIPTIONS
SCLK—Serial Clock
The serial clock pin is used to synchronize data to and from the
AD9714/AD9715/AD9716/AD9717 and to run the internal state
machines. The SCLK maximum frequency is 20 MHz. All data
input to the AD9714/AD9715/AD9716/AD9717 is registered on
the rising edge of SCLK. All data is driven out of the AD9714/
AD9715/AD9716/AD9717 on the falling edge of SCLK.
The remaining SCLK edges are for Phase 2 of the communication
cycle. Phase 2 is the actual data transfer between the AD9714/
AD9715/AD9716/AD9717 and the system controller. Phase 2 of
the communication cycle is a transfer of one, two, three, or four
data bytes, as determined by the instruction byte. Using one multi-
byte transfer is the preferred method. Single byte data transfers
are useful to reduce CPU overhead when register access requires
one byte only. Registers change immediately upon writing to the
last bit of each transfer byte.
CS
—Chip Select
An active low input starts and gates a communications cycle.
It allows more than one device to be used on the same serial
communications lines. The SDIO/FORMAT pin reaches a
high impedance state when this input is high. Chip select
should stay low during the entire communications cycle.
SDIO—Serial Data I/O
The SDIO pin is used as a bidirectional data line to transmit
and receive data.
Rev. 0 | Page 31 of 48
AD9714/AD9715/AD9716/AD9717
MSB/LSB TRANSFERS
SERIAL PORT OPERATION
The serial port of the AD9714/AD9715/AD9716/AD9717 can
support both most significant bit (MSB) first or least significant
bit (LSB) first data formats. This functionality is controlled by
the LSBFIRST bit (Register 0x00, Bit 6). The default is MSB first
(LSBFIRST = 0).
The serial port configuration of the AD9714/AD9715/AD9716/
AD9717 is controlled by Register 0x00. It is important to note
that the configuration changes immediately upon writing to the
last bit of the register. For multibyte transfers, writing to this
register can occur during the middle of the communications
cycle. Care must be taken to compensate for this new configu-
ration for the remaining bytes of the current communications cycle.
When LSBFIRST = 0 (MSB first), the instruction and data bytes
must be written from the most significant bit to the least significant
bit. Multibyte data transfers in MSB first format start with an
instruction byte that includes the register address of the most
significant data byte. Subsequent data bytes should follow in
order from a high address to a low address. In MSB first mode,
the serial port internal byte address generator decrements for
each data byte of the multibyte communications cycle.
The same considerations apply to setting the software reset,
RESET (Register 0x00, Bit 5). All registers are set to their default
values except Register 0x00, which remains unchanged.
Use of single-byte transfers or initiating a software reset is
recommended when changing serial port configurations to
prevent unexpected device behavior.
When LSBFIRST = 1 (LSB first), the instruction and data bytes
must be written from the least significant bit to the most signifi-
cant bit. Multibyte data transfers in LSB first format start with
an instruction byte that includes the register address of the least
significant data byte followed by multiple data bytes. The serial
port internal byte address generator increments for each byte
of the multibyte communication cycle.
PIN MODE
The AD9714/AD9715/AD9716/AD9717 can also be operated
without ever writing to the serial port. With RESET/PINMD
pin tied high, the SCLK pin becomes CLKMD to provide for
clock mode control (see the Retimer section), the former SDIO
pin selects the input data format, and the
power down the device.
CS
pin serves to
The serial port controller data address of the AD9714/AD9715/
AD9716/AD9717 decrements from the data address written
toward 0x00 for multibyte I/O operations if the MSB first mode
is active. The serial port controller address increments from the
data address written toward 0x1F for multibyte I/O operations
if the LSB first mode is active.
Operation is otherwise exactly as defined by the default register
values in Table 12, therefore external resistors at FSADJI and
FSADJQ are needed to set the DAC currents, and both DACs
are active. This is also a convenient quick checkout mode.
DAC currents can be externally adjusted in pin mode by sourcing
or sinking currents at the FSADJI/AUXI and FSADJQ/AUXQ
pins as desired with the fixed resistors installed. An op amp
output with appropriate series resistance would be one of
many possibilities. This has the same effect as changing the
resistor value. Place at least 10 kΩ resistors in series right at
the DAC to guard against accidental short circuits and noise
modulation. The REFIO pin can be adjusted 25% in a similar
manner, if desired.
Rev. 0 | Page 32 of 48
AD9714/AD9715/AD9716/AD9717
SPI REGISTER MAP
Table 13.
Name
Addr Default Bit 7
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
SPI Control
Power Down
Data Control
I DAC Gain
IRSET
0x00 0x00
0x01 0x40
0x02 0x34
0x03 0x00
0x04 0x00
0x05 0x00
0x06 0x00
0x07 0x00
0x08 0x00
0x09 0x00
0x0A 0x00
0x0B 0x00
0x0C 0x00
0x0D 0x00
LSBFIRST RESET
LDOSTAT PWRDN
IFIRST
LNGINS
Q DACOFF
IRISING
LDOOFF
TWOS
I DACOFF
SIMULBIT
QCLKOFF
DCI_EN
ICLKOFF EXTREF
DCOSGL DCODBL
I DACGAIN[5:0]
IRSETEN
IRSET[5:0]
IRCML[5:0]
IRCML
IRCMLEN
Q DAC Gain
QRSET
Q DACGAIN[5:0]
QRSET[5:0]
QRSETEN
QRCMLEN
QRCML
QRCML[5:0]
AUXDAC Q
AUX CTLQ
AUXDAC I
AUX CTLI
QAUXDAC[7:0]
QAUXOFS[2:0]
IAUXDAC[7:0]
IAUXOFS[2:0]
RREF[5:0]
QAUXEN
IAUXEN
QAUXRNG[1:0]
IAUXRNG[1:0]
QAUXDAC[9:8]
IAUXDAC[9:8]
Reference
Resistor
Cal Control
Cal Memory
0x0E 0x00
0x0F 0x00
PRELDQ
PRELDI
CALSELQ CALSELI
CALCLK
DIVSEL[2:0]
CALSTATQ CALSTATI
CALMEMQ[1:0]
MEMADDR[5:0]
MEMDATA[5:0]
CALMEMI[1:0]
Memory Address 0x10 0x00
Memory Data
Memory R/W
CLKMODE
Version
0x11 0x34
0x12 0x00
0x14 0x00
0x1F N/A
CALRSTQ
CALRSTI
CALEN
SMEMWR
SMEMRD
UNCALQ UNCALI
CLKMODEQ[1:0]
SEARCHING REACQUIRE CLKMODEN CLKMODEI[1:0]
VERSION[7:0]
Rev. 0 | Page 33 of 48
AD9714/AD9715/AD9716/AD9717
SPI REGISTER DESCRIPTIONS
Reading these registers returns previously written values for all defined register bits, unless otherwise noted.
Table 14.
Register
Address Bit Name
Function
SPI Control
0x00
6
LSBFIRST
0: MSB first, per SPI standard
1: LSB first, per SPI standard
Note that the user must always change the LSB/MSB order in single-byte instructions
to avoid erratic behavior due to bit order errors
5
4
RESET
Execute software reset of SPI and controllers, reload default register values except
Register 0x00
1: Set software reset; write 0 on the next (or any following) cycle to release the reset
0: The SPI instruction word utilizes a 5-bit address
LNGINS
1: The SPI instruction word utilizes a 13-bit address
0x01
Power Down
7
6
LDOOFF
LDOSTAT
1: turn core LDO voltage regulator off
0: Indicates core LDO voltage regulator is off
1: Indicates core LDO voltage regulator is on
1: Powers down all analog and digital circuitry except for SPI logic
1: Turns off Q DAC output current
1: Turns off I DAC output current
1: Turns off Q DAC clock
5
4
3
2
1
0
7
PWRDN
Q DACOFF
I DACOFF
QCLKOFF
ICLKOFF
EXTREF
1: Turns off I DAC clock
1: Powers down internal voltage reference (external reference required)
Data Control 0x02
TWOS
0: Unsigned binary input data format
1: Twos complement input data format
5
4
3
2
IFIRST
0: Pairing of data—Q first of pair on data input pads
1: Pairing of data—I first of pair on data input pads (default)
0: Q data latched on DCLKIO rising edge
1: I data latched on DCLKIO falling edge (default)
0: Allows simultaneous input and output enable on DCLKIO
1: Disallows simultaneous input and output enable on DCLKIO
Controls the use of DCLKIO pad for data clock input
0: Data clock input disabled
IRISING
SIMULBIT
DCI_EN
1: Data clock input enabled (default)
1
0
DCOSGL
DCODBL
Controls the use of DCLKIO pad for data clock output
0: Data clock output disabled
1: Data clock output enabled; regular strength driver
Controls the use of DCLKIO pad for data clock output
0: DCOBL data clock output disabled
1: DCOBL data clock output enabled; paralleled with DCOSGL for 2× drive current
I DAC Gain
IRSET
0x03
0x04
5:0 I DACGAIN[5:0]
IRSETEN
5:0 IRSET[5:0]
DAC I fine gain adjustment; alters the full-scale current as shown in Figure 86
1: Enables the on-chip RSET value to be changed
Changes the value of the on-chip RSET resistor; this scales the full-scale current of the
7
DAC in ~0.25 dB steps (nonlinear); see Figure 85
000000: RSET = 8 kΩ
100000: RSET = 16 kΩ
111111: RSET = 32 kΩ
Rev. 0 | Page 34 of 48
AD9714/AD9715/AD9716/AD9717
Register
Address Bit Name
Function
IRCML
0x05
7
IRCMLEN
1: Enables on-chip RCML adjustment
5:0 IRCML[5:0]
Changes the value of the on-chip RCML resistor; this adjusts the common-mode level of
the DAC output stage
000000: RSET = 250 Ω
100000: RSET = 625 Ω
111111: RSET = 1 kΩ
Q DAC Gain
QRSET
0x06
0x07
5:0 Q DACGAIN[5:0]
DAC Q fine gain adjustment; alters the full-scale current as shown in Figure 86
1: Enables on-chip RCML adjustment
7
QRSETEN
5:0 QRSET[5:0]
Changes the value of the on-chip RSET resistor; this scales the full-scale current of the
DAC in ~0.25 dB steps (nonlinear), see Figure 85
000000: RSET = 8 kΩ
100000: RSET = 16 kΩ
111111: RSET = 32 kΩ
QRCML
0x08
7
QRCMLEN
1: Enables on-chip RCML adjustment
5:0 QRCML[5:0]
Changes the value of the on-chip RCML resistor; this adjusts the common-mode level of
the DAC output stage
000000, RSET = 250 Ω
100000, RSET = 625 Ω
111111, RSET = 1 kΩ
AUXDAC Q
AUX CTLQ
0x09
0x0A
7:0 QAUXDAC[7:0]
AUXDAC Q output voltage adjustment word LSBs
0x3FF: Sets AUXDAC Q output to full scale
0x200: Sets AUXDAC Q output to midscale
0x000: Sets AUXDAC Q output to bottom of scale
7
QAUXEN
1: enables AUXDAC Q
6:5 QAUXRNG[1:0]
00: Sets AUXDAC Q output voltage range to 2 V
01: Sets AUXDAC Q output voltage range to 1.5 V
10: Sets AUXDAC Q output voltage range to 1.0 V
11: Sets AUXDAC Q output voltage range to 0.5 V
000: Sets AUXDAC Q top of range to 1.0 V
001: Sets AUXDAC Q top of range to 1.5 V
010: Sets AUXDAC Q top of range to 2.0 V
011: Sets AUXDAC Q top of range to 2.5 V
100: Sets AUXDAC Q top of range to 2.9 V
AUXDAC Q output voltage adjustment word MSBs
4:2 QAUXOFS[2:0]
1:0 QAUXDAC[9:8]
7:0 IAUXDAC[7:0]
AUXDAC I
AUX CTLI
0x0B
0x0C
AUXDAC I output voltage adjustment word LSBs
0x3FF: Sets AUXDAC I output to full scale
0x200: Sets AUXDAC I output to midscale
0x000: Sets AUXDAC I output to bottom of scale
7
IAUXEN
1: enables AUXDAC I
6:5 IAUXRNG[1:0]
00: Sets AUXDAC I output voltage range to 2 V
01: Sets AUXDAC I output voltage range to 1.5 V
10: Sets AUXDAC I output voltage range to 1.0 V
11: Sets AUXDAC I output voltage range to 0.5 V
000: Sets AUXDAC I top of range to 1.0 V
001: Sets AUXDAC I top of range to 1.5 V
010: Sets AUXDAC I top of range to 2.0 V
011: Sets AUXDAC I top of range to 2.5 V
100: Sets AUXDAC I top of range to 2.9 V
AUXDAC I output voltage adjustment word MSBs
4:2 IAUXOFS[2:0]
1:0 IAUXDAC[9:8]
5:0 RREF[5:0]
Reference
Resistor
0x0D
Permits an adjustment of the on-chip reference voltage and output at REFIO (see Figure 84)
000000: Sets the value of RREF to 8 kΩ, VREF = 0.8 V
100000: Sets the value of RREF to 10 kΩ, VREF = 1.0 V
111111: Sets the value of RREF to 12 kΩ, VREF = 1.2 V
Rev. 0 | Page 35 of 48
AD9714/AD9715/AD9716/AD9717
Register
Address Bit Name
Function
Cal Control
0x0E
7
PRELDQ
0: Preload Q DAC calibration reference set to 32
1: Preload Q DAC calibration reference set by user (Cal Address 1)
0: Preload I DAC calibration reference set to 32
1: Preload I DAC calibration reference set by user (Cal Address 1)
1: Select Q DAC self-calibration
6
PRELDI
5
4
3
CALSELQ
CALSELI
CALCLK
1: Select I DAC self-calibration
1: Calibration clock enabled
2:0 DIVSEL[2:0]
Calibration clock divide ratio from DAC clock rate
000 = divide by 256; 001 = divide by 128 … 110 = divide by 4; 111 = divide by 2
Cal Memory 0x0F
7
6
CALSTATQ
CALSTATI
1: Calibration of Q DAC complete
1: Calibration of I DAC complete
Status of Q DAC calibration memory
00: Uncalibrated
3:2 CALMEMQ[1:0]
01: Self-calibrated
10: User calibrated
1:0 CALMEMI[1:0]
Status of I DAC calibration memory
00: Uncalibrated
01: Self-calibrated
10: User calibrated
Memory
Address
0x10
0x11
0x12
5:0 MEMADDR[5:0]
5:0 MEMDATA[5:0]
Address of static memory to be accessed
Memory
Data
Data for static memory access
Memory
R/W
7
6
4
3
2
1
0
CALRSTQ
CALRSTI
CALEN
SMEMWR
SMEMRD
UNCALQ
UNCALI
1: Clear CALSTATQ
1: Clear CALSTATI
1: Initiate device self-calibration
1: Write to static memory (calibration coefficients)
1: Read from static memory (calibration coefficients)
1: Reset Q DAC calibration coefficients to default (uncalibrated)
1: Reset I DAC calibration coefficients to default (uncalibrated)
Q datapath retimer clock select output (that is, readback after Q retimer acquires)
CLKMODE
0x14
7:6 CLKMODEQ[1:0]
4
SEARCHING
High indicates internal data path retimer is searching for clock relationship (device
output is not usable while this bit is high)
3
2
REACQUIRE
CLKMODEN
Edge triggered, 0 to 1 causes the retimer to reacquire the clock relationship
0: CLKMODEI/Q values computed by the two retimers and read back in CLKMODEI[1:0]
and CLKMODEQ[1:0]
1: CLKMODE values set in CLKMODEI[1:0] over-ride both I and Q retimers
1:0 CLKMODEI[1:0]
7:0 VERSION[7:0]
0: CLKMODEN, read only; clock phase chosen by retimer
1: CLKMODEN, read/write; value in this register sets I and Q clock phases
Version
0x1F
Hardware version of the device
Rev. 0 | Page 36 of 48
AD9714/AD9715/AD9716/AD9717
DIGITAL INTERFACE OPERATION
Digital data for the I and Q DACs is supplied over a single
parallel bus (DB[MSB:0]) accompanied by a qualifying clock
(DCLKIO). The I and Q data is provided to the chip in an
interleaved double data rate (DDR) format. The maximum
guaranteed data rate is 250 MSPS with a 125 MHz clock. The
order of data pairing and the sampling edge selection is user
programmable using the IFIRST and IRISING configuration
bits, resulting in four possible timing diagrams. These are
shown in Figure 77, Figure 78, Figure 79, and Figure 80.
DCLKIO
DB[13:0]
I DATA
Z
A
B
C
D
E
F
G
H
Z
B
D
F
Q DATA
A
C
E
G
Figure 79. Timing Diagram with IFIRST = 1, IRISING = 0
DCLKIO
DCLKIO
DB[13:0]
I DATA
Z
A
B
C
D
E
F
G
H
DB[13:0]
I DATA
Z
A
B
C
D
E
F
G
H
Z
B
D
F
E
Y
A
C
E
F
Q DATA
Y
A
C
Figure 77. Timing Diagram with IFIRST = 0, IRISING = 0
Q DATA
Z
B
D
Figure 80. Timing Diagram with IFIRST = 1, IRISING = 1
DCLKIO
Ideally, the rising and falling edges of the clock fall in the center
of the keep-in-window formed by the set-up and hold times, tS
and tH. A detailed timing diagram is shown in Figure 81.
DB[13:0]
I DATA
Z
A
B
C
D
E
F
G
H
DCLKIO
Y
A
C
E
D
tS tH
tS tH
Q DATA
X
Z
B
Figure 78. Timing Diagram with IFIRST = 0, IRISING = 1
DB[13:0]
Figure 81. Set-Up and Hold Times for All Input Modes
In addition to the different timing modes listed in Table 2, the
input data can also be presented to the device in either unsigned
binary or twos complement format. The format type is chosen
via the TWOS configuration bit.
Rev. 0 | Page 37 of 48
AD9714/AD9715/AD9716/AD9717
OR
RETIMER-CLK
D-FF
D-FF
D-FF
1
D-FF
2
D-FF
4
0
3
TO DAC CORE
DCLKIO-INT
I
OUT
DATA DB[13:0]
(INPUT)
DCLKIO-INT
I
OUT
NOTES
D-FFs:
0: RISING OR FALLING EDGE
TRIGGERED FOR I OR Q DATA.
1, 2, 3, 4: RISING EDGE TRIGGERED.
IE
IE
OE
DELAY2
DCLKIO
(INPUT/OUTPUT)
CLKIN
(INPUT)
Figure 82. Simplified Diagram of AD9714/AD9715/AD9716/AD9717 Timing
Settling OE high in the SPI allows the user to get a DCLKIO
DIGITAL DATA LATCHING AND RETIMER SECTION
output from the CLKIN input for use in the user’s PCB system.
It is strongly recommended that IE = OE = high not be used
even though the device may appear to function correctly.
The AD9714/AD9715/AD9716/AD9717 have two clock inputs,
DCLKIO and CLKIN. The CLKIN is the analog clock whose
jitter affects DAC performance and the DCLKIO is a digital
clock, probably from an FPGA that needs to have a fixed rela-
tionship with the input data to ensure that the data is picked
up correctly by the flip-flops on the pads.
Retimer
The AD9714/AD9715/AD9716/AD9717 have an internal data
retimer circuit that compares the CLKIN-INT and DCLKIO-INT
clocks and, depending on their phase relationship, selects a
retimer clock (RETIMER-CLK) to safely transfer data from
the DCLKIO used at the chip’s input interface to the CLKIN
used to clock the analog DAC cores (D-FF (4)).
Figure 82 is a simplified diagram of the entire data capture system
in the AD9714/AD9715/AD9716/AD9717. The double data rate
input data, DB[13:0], is latched at the pads/pins either on the
rising edge or the falling edge of the DCLKIO-INT clock, as deter-
mined by IRISING, the SPI bit. IFIRST, the SPI bit determines
which channel data is latched first (that is, I or Q). The captured
data is then retimed to the internal clock (CLKIN-INT) in the
retimer block before being sent to the final analog DAC core
(D-FF (4)), which controls the current steering output switches.
All delay blocks depicted in Figure 82 are noninverting, and any
wires without an explicit delay block can be assumed to have no
delay for the purpose of understanding.
The retimer selects one of the three phases shown in Figure 83.
The retimer is controlled by the SPI bits is shown in Table 15.
RETIMER-CLKs
1/2 PERIOD
DATA
CLOCK
180°
90°
270°
1/4 PERIOD 1/2 PERIOD
Figure 83. RETIMER-CLK Phases
Only one channel is shown in Figure 82 with the data pads
(DB[13:0]) serving as double data rate pads for both channels.
Note that in most cases, more than one retimer phase works,
and in such cases, the retimer arbitrarily picks one phase that
works. The retimer cannot pick the best or safest phase. If the
user has a working knowledge of the exact phase relationship
between DCLKIO and CLKIN (and thus DCLKIO-INT and
CLKIN-INT, because the delay is approximately the same for
both clocks and equal to DELAY1), then the retimer can be
forced to this phase with CLKMODEN = 1 as described in
Table 15 and the following paragraphs.
The default PINMD and SPI settings are IE = high (closed)
and OE = low (open). These settings are enabled when RESET/
PINMD (Pin 35) is held high. In this mode, the user has to supply
both DCLKIO and CLKIN. In PINMD, it is also recommended
that the DCLKIO and the CLKIN be in-phase for proper func-
tioning of the DAC, which can easily be ensured by tying the
pins together on the PCB. If the user can access the SPI, settling
IE
IE low (that is, is high) causes the CLKIN to be used as the
DCLKIO also.
Rev. 0 | Page 38 of 48
AD9714/AD9715/AD9716/AD9717
Table 15. Timer Register List
Bit Name
Description
CLKMODEQ[1:0]
Searching
Reacquire
Q datapath retimer clock selected output. Valid after SEARCHING goes low.
High indicates the internal data path retimer is searching for clock relationship (DAC is not usable until it is low again).
Changing this bit from 0 to 1 causes the data path retimer circuit to reacquire the clock relationship.
0: Uses CLKMODEI/CLKMODEQ values (as computed by the two internal retimers) for I and Q clocking.
1: Uses CLKMODE value set in CLKMODEI[1:0] for both I and Q retimers (that is, force the retimer).
I datapath retimer clock selected output. Valid after searching goes low.
CLKMODEN
CLKMODEI[1:0]
If CLKMODEN = 1, a value written to this register overrides both I and Q automatic retimer values.
Table 16. CLKMODE Details
CLKMODEI[1:0]/CLKMODEQ[1:0] DCLKIO-to-CLKIN Phase Relationship
RETIMER-CLK Selected
Phase 2
Phase 3
Phase 3
Phase 1
00
01
10
11
0° to 90°
90° to 180°
180° to 270°
270° to 360°
When reset is pulsed high and then returns low (the part is in
SPI mode), the retimer runs and automatically selects a suitable
clock phase for the RETIMER-CLK within 128 clock cycles. The
SPI searching bit returns to low, indicating that the retimer has
locked and the part is ready for use. The reacquire bit can be
used to reinitiate phase detection in the I and Q retimers at
any time. CLKMODEQ[1:0] and CLKMODEI[1:0] provide
readback for the values picked by the internal phase detectors
in the retimer (see Table 16).
causes the internal clock detector to use the phase detector
output to determine which clock to use in the retimer (that is,
select a suitable RETIMER-CLK phase). The action of taking
SCLK high causes the internal phase detector to reexamine the
two clocks and determine the relative phase. Whenever the user
wants to reevaluate the relative phase of the two clocks, the
SCLK pin can be taken low and then high again.
ESTIMATING THE OVERALL DAC PIPELINE DELAY
DAC pipeline latency is affected by the phase of the RETIMER-
CLK that is selected. If latency is critical to the system and needs
to be constant, the retimer should be forced to a particular phase
and not be allowed to automatically select a phase each time.
To force the two retimers (I and Q) to pick a particular phase
for the retimer clock (they must both be forced to the same
value), CLKMODEN should be set high and the required phase
value is written into CLKMODEI[1:0] and CLKMODEQ[1:0].
For example, if the DCLKIO and the CLKIN are in phase to first
order, the user could safely force the retimers to pick Phase 2 for
the RETIMER-CLK. This forcing function may be useful for
synchronizing multiple devices.
Consider the case when DCLKIO = CLKIN (that is, in phase),
and the RETIMER-CLK is forced to Phase 2. Assume that
IRISING is 1 (that is, I data is latched on the rising edge and
Q data on the falling edge). Then the latency to the output for
the I-channel is 3 clock cycles (D-FF (1), D-FF (3), and D-FF
(4), but not D-FF (2) because it is latched on the half clock
cycle or 180°). The latency to the output for the Q-channel
In pin mode, it is expected that the user tie CLKIN and DCLKIO
together. The device has a small amount of programmable
functionality using the now unused SPI pins (SCLK, SDIO, and
from the time the falling edge latches it at the pads in D-FF (0)
is 2.5 clock cycles (½ clock cycle to D-FF (1), 1 clock cycle to
D-FF (3), and 1 clock cycle to D-FF (4)). This latency for the
AD9714/AD9715/ AD9716/AD9717 is case specific and needs
to be calculated based on the RETIMER-CLK phase that is
automatically selected or manually forced.
CS
). If the two chip clocks are tied together, the SCLK pin can
be tied to ground and the chip uses a clock for the retimer that
is 180° out of phase with the two input clocks (that is, Phase 2,
which is the safest or best option). The chip has an additional
option in pin mode when the redefined SCLK pin is high. Use
this mode if utilizing pin mode, but CLKIN and DCLKIO are
not tied together (that is, not in phase). Holding SCLK high
Rev. 0 | Page 39 of 48
AD9714/AD9715/AD9716/AD9717
The AD9714/AD9715/AD9716/AD9717 allow reading and
writing of the calibration coefficients. There are 32 coefficients
in total. The read/write feature of the coefficients can be useful
for improving the results of the self-calibration routine by
averaging the results of several self-calibration cycles and
loading the averaged results back into the device.
SELF-CALIBRATION
The AD9714/AD9715/AD9716/AD9717 have a self-calibration
feature that improves the DNL of the device. Performing a self-
calibration on the device improves device performance in low
frequency applications. The device performance in applications
where the analog output frequencies are above 5 MHz are generally
influenced more by dynamic device behavior than by DNL, and
in these cases, self-calibration is unlikely to provide much benefit.
The calibration clock frequency is equal to the DAC clock
divided by the division factor chosen by the DIVSEL value. Each
calibration clock cycle is between 32 and 2048 DAC input clock
cycles, depending on the value of DIVSEL[2:0] (Register 0x0E,
Bits[2:0]). The frequency of the calibration clock should be
between 0.5 MHz and 4 MHz for reliable calibrations. Best
results are obtained by setting DIVSEL[2:0] (Register 0x0E,
Bits[2:0]) to produce a calibration clock frequency between
these values. Separate self-calibration hardware is included
for each DAC. The DACs can be self-calibrated individually or
simultaneously.
To read the calibration coefficients, use the following steps:
1. Select which DAC core to read by setting either Bit 4
(CALSELI) for the I DAC or Bit 5 (CALSELQ) for the
Q DAC in Register 0x0E. Write the address of the first
coefficient (0x01) to Register 0x10.
2. Set the SMEMRD bit (Register 0x12, Bit 2 ) by writing 0x04
to Register 0x12.
3. Read the 6-bit value of the first coefficient by reading the
contents of Register 0x11.
4. Clear the SMEMRD bit by writing 0x00 to Register 0x12.
5. Repeat Step 2 through Step 4 for each of the remaining 31
coefficients by incrementing the address by one for each read.
6. Deselect the DAC core by clearing either Bit 4 (CALSELI)
for the I DAC or Bit 5 (CALSELQ) for the Q DAC in
Register 0x0E.
To perform a device self-calibration, the following procedure
can be used:
To write the calibration coefficients to the device, use the
following steps:
1. Write 0x00 to Register 0x12. This ensures that the
UNCALI and UNCALQ bits are reset.
2. Set up a calibration clock between 0.5 MHz and 4 MHz
using DIVSEL[2:0] and then enable the calibration clock
by setting the CALCLK bit (Register 0x0E, Bit 3).
3. Select the DAC(s) to self-calibrate by setting either Bit 4
(CALSELI) for the I DAC and/or Bit 5 (CALSELQ) for the
Q DAC in Register 0x0E. Note that each DAC contains
independent calibration hardware so they can be calibrated
simultaneously.
4. Start self-calibration by setting Bit 4 in Register 0x12. Wait
approximately 300 calibration clock cycles.
5. Check if the self-calibration has completed by reading
the CALSTATI bit (Bit 6) and CALSTATQ bit (Bit 7) in
Register 0x0F. Logic 1 indicates the calibration has
completed.
1. Select which DAC core to write by setting either Bit 4
(CALSELI) for the I DAC or Bit 5 (CALSELQ) for the Q
DAC in Register 0x0E.
2. Set the SMEMWR bit (Register 0x12, Bit 3) by writing 0x08
to Register 0x12.
3. Write the address of the first coefficient (0x01) to
Register 0x10.
4. Write the value of the first coefficient to Register 0x11.
5. Repeat Step 2 through Step 4 for each of the remaining 31
coefficients by incrementing the address by one for each
write.
6. Clear the SMEMWR bit by writing 0x00 to Register 0x12.
7. Deselect the DAC core by clearing either Bit 4 (CALSELI)
for the I DAC or Bit 5 (CALSELQ) for the Q DAC in
Register 0x0E.
6. When the self-calibration has completed, write 0x00 to
Register 0x12.
7. Disable the calibration clock by clearing the CALCLK bit
(Register 0x0E, Bit 3).
Rev. 0 | Page 40 of 48
AD9714/AD9715/AD9716/AD9717
Option 3
COARSE GAIN ADJUSTMENT
Even when the device is in pin mode, full-scale values can be
adjusted by sourcing or sinking current from the FSADJ pins.
Any noise injected here appears as amplitude modulation of the
output. Thus, a portion of the required series resistance (at least
20 kΩ) must be installed right at the pin. A range of 10% is
quite practical using this method.
Option 1
A coarse full-scale output current adjustment can be achieved
using the lower six bits in Register 0x0D. This adds or subtracts
up to 20% from the band gap voltage on Pin 34 (REFIO), and
the voltage on the FSADJx resistors tracks this change. As a
result, the DAC full-scale current varies the same amount.
A secondary effect to changing the REFIO voltage is that the
full-scale voltage in the AUXDAC also changes by the same
magnitude. The register uses twos complement format, in
which 011111 maximizes the voltage on the REFIO node
and 100000 minimizes the voltage.
Option 4
As in Option 3, when the device is in pin mode both full-scale
values can be adjusted by sourcing or sinking current from the
REFIO pin. Noise injected here appears as amplitude modulation
of the output, so a portion of the required series resistance (at
least 10 kΩ) must be installed at the pin. A range of 25% is
quite practical when using this method.
1.30
1.25
1.20
1.15
1.10
1.05
1.00
0.95
0.90
0.85
0.80
Fine Gain
Each main DAC has independent fine gain control using the
lower six bits in Register 0x03 (I DAC gain) and Register 0x06
(Q DAC gain). Unlike Coarse Gain Option 1, this impacts only
the main DAC full-scale output current. This register uses straight
binary format. One application where this straight binary for-
mat is critical is for side-band suppression while using a quadrature
modulator. This is described in more detail in the Applications
Information section.
2.22
0
8
16
24
32
40
48
56
3.3V DAC1
CODE
3.3V DAC2
1.8V DAC1
2.20
Figure 84. Typical VREF Voltage vs. Code
1.8V DAC2
Option 2
2.18
While utilizing the internal FSADJx resistors, each main DAC
can achieve independently controlled coarse gain using the
lower six bits of Register 0x04 (IRSET[5:0]) and Register 0x07
(QRSET[5:0]). Unlike Coarse Gain Option 1, this impacts only
the main DAC full-scale output current. The register uses twos
complement format and allows the output current to be changed
in approximately 0.25 dB steps.
2.16
2.14
2.12
2.10
4.0
0
8
16
24
32
40
48
56
64
GAIN DAC CODE
3.5
3.0
Figure 86. Typical DAC Gain Characteristics
V
_Q OR V _I
OUT OUT
2.5
2.0
1.5
1.0
0.5
0
0
10
20
30
RSET CODE
40
50
60
Figure 85. Effect of RSET Code
Rev. 0 | Page 41 of 48
AD9714/AD9715/AD9716/AD9717
1200
1100
1000
900
800
700
600
500
400
300
200
USING THE INTERNAL TERMINATION RESISTORS
The AD9717/AD9716/AD9715/AD9714 have four 500 Ω
termination internal resistors (two for each DAC output).
To use these resistors to convert the DAC output current to a
voltage, connect each DAC output pin to the adjacent load pin.
For example, on the I DAC, IOUTP must be shorted to RLIP
and IOUTN must be shorted to RLIN. In addition, the CMLI
or CMLQ pin must be connected to ground directly or through
a resistor. If the output current is at the nominal 2 mA and the
CMLI or CMLQ pin is tied directly to ground, this produces a
dc common-mode bias voltage on the DAC output equal to 0.5 V.
If the DAC dc bias needs to be higher than 0.5 V, an external
resistor can be connected between the CMLI or CMLQ pin and
ground. This part also has an internal common-mode resistor
that can be enabled. This is explained in the Using the Internal
Common-Mode Resistor section.
0
8
16
24
32
40
48
56
CODE
Figure 88. Typical CML Resistor Value vs. Register Code
Using the CMLx Pins for Optimal Performance
CML
The CMLx pins also serve to change the DAC bias voltages
in the parts allowing them to run at higher dc output bias
voltages. When running the bias voltage below 0.9 V and an
AVDD of 3.3 V, the parts perform optimally when the CMLx
pins are tied to ground. When the dc bias increases above 0.9 V,
set the CMLx pins at 0.5 V for optimal performance. The maxi-
mum dc bias on the DAC output should be kept at or below 1.2 V
when the supply is 3.3 V. When the supply is 1.8 V, keep the dc
bias close to 0 V and connect the CMLx pins directly to ground.
RCM
RLIN
500Ω
IOUTN
IOUTP
RLIP
I DAC
OR
Q DAC
500Ω
Figure 87. Simplified Internal Load Options
Using the Internal Common-Mode Resistor
These devices contain an adjustable internal common-mode
resistor that can be used to increase the dc bias of the DAC
outputs. By default, the common-mode resistor is not con-
nected. When enabled, it can be adjusted from ~250 Ω to
~1 kΩ. Each main DAC has an independent adjustment
using the lower six bits in Register 0x05 (IRCML[5:0]) and
Register 0x08 (QRCML[5:0]).
Rev. 0 | Page 42 of 48
AD9714/AD9715/AD9716/AD9717
APPLICATIONS INFORMATION
A differential resistor, RDIFF, can be inserted in applications
where the output of the transformer is connected to the load,
OUTPUT CONFIGURATIONS
The following sections illustrate some typical output confi-
gurations for the AD9714/AD9715/AD9716/AD9717. Unless
otherwise noted, it is assumed that IOUTFS is set to a nominal
2 mA. For applications requiring the optimum dynamic perfor-
mance, a differential output configuration is suggested. A
differential output configuration can consist of either an
RF transformer or a differential op amp configuration. The
transformer configuration provides the optimum high fre-
quency performance and is recommended for any application
that allows ac coupling. The differential op amp configuration
is suitable for applications requiring dc coupling, signal gain,
and/or a low output impedance.
RLOAD, via a passive reconstruction filter or cable. RDIFF, as
reflected by the transformer, is chosen to provide a source
termination that results in a low VSWR. Note that approx-
imately half the signal power is dissipated across RDIFF
.
SINGLE-ENDED BUFFERED OUTPUT USING
AN OP AMP
An op amp such as the ADA4899-1 can be used to perform
a single-ended current-to-voltage conversion, as shown in
Figure 90. The AD9714/AD9715/AD9716/AD9717 are config-
ured with a pair of series resistors, RS, off each output. For best
distortion performance, RS should be set to 0 Ω. The feedback
resistor, RFB, determines the peak-to-peak signal swing by the
formula
A single-ended output is suitable for applications where low
cost and low power consumption are primary concerns.
V
OUT = RFB × IFS
DIFFERENTIAL COUPLING USING A TRANSFORMER
An RF transformer can be used to perform a differential-to-
single-ended signal conversion, as shown in Figure 89. The
distortion performance of a transformer typically exceeds
that available from standard op amps, particularly at higher
frequencies. Transformer coupling provides excellent rejection
of common-mode distortion (that is, even-order harmonics)
over a wide frequency range. It also provides electrical isolation
and can deliver voltage gain without adding noise. Transformers
with different impedance ratios can also be used for impedance
matching purposes. The main disadvantages of transformer
coupling are low frequency roll-off, lack-of-power gain, and
high output impedance.
The common-mode voltage of the output is determined by the
formula
⎛
⎜
⎝
⎞
⎟
⎟
⎠
RFB
RB
RFB × IFS
⎜
VCM = VREF × 1+
−
2
The maximum and minimum voltages out of the amplifier are,
respectively,
⎛
⎜
⎝
⎞
⎟
⎟
⎠
RFB
RB
⎜
VMAX = VREF × 1+
V
MIN = VMAX – IFS × RFB
C
F
R
R
FB
B
29
IOUTN
+5V
AD9714/AD9715/
AD9716/AD9717
IOUTP
AD9714/AD9715/
AD9716/AD9717
R
S
R
LOAD
28
34
–
ADA4899-1
+
V
OUT
28
IOUTP
REFIO
OPTIONAL R
DIFF
C
R
–5V
S
29
25
IOUTN
AVSS
Figure 89. Differential Output Using a Transformer
The center tap on the primary side of the transformer must be
connected to a voltage that keeps the voltages on IOUTP and
IOUTN within the output common-mode voltage range of the
device. Note that the dc component of the DAC output current
is equal to IOUTFS and flows out of both IOUTP and IOUTN. The
center tap of the transformer should provide a path for this dc
current. In most applications, AGND provides the most conve-
nient voltage for the transformer center tap. The complementary
voltages appearing at IOUTP and IOUTN (that is, VIOUTP and
Figure 90. Single-Supply Single-Ended Buffer
VIOUTN) swing symmetrically around AGND and should be
maintained with the specified output compliance range of the
AD9714/AD9715/AD9716/AD9717.
Rev. 0 | Page 43 of 48
AD9714/AD9715/AD9716/AD9717
To keep the pin count reasonable, these auxiliary DACs each
share a pin with the corresponding FSADJx resistor. They are,
therefore, usable only when enabled and when that DAC is
operated on its internal full-scale resistors. A simple I-to-V
converter is implemented on chip with selectable shunt resistors
(3.2 kΩ to 16 kΩ) such that if REFIO is set to exactly 1 V, REFIO/2
equals 0.5 V and the following equation describes the no load
output voltage:
DIFFERENTIAL BUFFERED OUTPUT
USING AN OP AMP
A dual op amp (see the circuit shown in Figure 91) can be used
in a differential version of the single-ended buffer shown in
Figure 90. The same R-C network is used to form a one-pole
differential, low-pass filter to isolate the op amp inputs from
the high frequency images produced by the DAC outputs. The
feedback resistors, RFB, determine the differential peak-to-peak
signal swing by the formula
⎛
⎞
⎟
1.5
⎜
⎟
16 k Ω
VOUT = 0.5 V − IDAC
−
⎜
⎝
RS
V
OUT = 2 × RFB × IFS
⎠
The maximum and minimum single-ended voltages out of the
amplifier are, respectively,
Figure 92 illustrates the function of all the SPI bits controlling
these DACs with the exception of the QAUXEN and IAUXEN
bits and gating to prohibit RS < 3.2 kΩ.
⎛
⎜
⎝
⎞
⎟
⎟
⎠
RFB
RB
⎜
VMAX =VREF × 1+
AVDD
RNG0
RNG1
RNG: 00 = > 125µA fS
01 = > 62µA fS
V
MIN = VMAX − RFB × IFS
AUXDAC
[9:0]
10 = > 31µA fS
11 = > 16µA fS
The common-mode voltage of the differential output is
determined by the formula
(OFS > 4 = 4)
OFS2
OFS1
OFS0
V
CM = VMAX – RFB × IFS
16kΩ
AUX
PIN
C
F
4kΩ 8kΩ 16kΩ 16kΩ
–
R
R
FB
B
OP AMP
+
AD9714/AD9715/
AD9716/AD9717
IOUTP
R
S
REFIO
2
28
34
–
ADA4841-2
+
REFIO
Figure 92. AUXDAC Simplified Circuit Diagram
V
C
OUT
AVSS
25
29
The SPI speed limits the update rate of the auxiliary DACs. The
data is inverted such that IAUXDAC is full scale at 0x000 and zero
at 0x1FF, as shown in Figure 93.
+
R
S
IOUTN
ADA4841-2
–
3.0
OP AMP OUTPUT VOLTAGE vs.
CHANGES IN R_OFFSET AND IDAC
C
F
2.8
2.6
R
R
FB
B
R_OFFSET = 3.3kΩ
2.4
R_OFFSET = 4kΩ
R_OFFSET = 5.3kΩ
R_OFFSET = 8kΩ
2.2
2.0
Figure 91. Single-Supply Differential Buffer
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
R_OFFSET = 16kΩ
AUXILIARY DACs
The DACs of the AD9714/AD9715/AD9716/AD9717 feature
two versatile and independent 10-bit auxiliary DACs suitable
for dc offset correction and similar tasks.
Because the AUXDACs are driven through the SPI port, they
should never be used in timing-critical applications, such
as inside analog feedback loops.
0
10 20 30 40 50 60 70 80 90 100 110 120 130
I
(µA)
AUXDAC
Figure 93. AUXDAC Op Amp Output vs. Current, AVDD = 3.3 V No Load,
AUXDAC 0x1FF to 0x000
Rev. 0 | Page 44 of 48
AD9714/AD9715/AD9716/AD9717
Two registers are assigned to each DAC with 10 bits for the
actual DAC current to be generated, a 3-bit offset (and gain)
adjustment, a 2-bit current range adjustment, and an enable/
disable bit. Setting the QAUXOFS and IAUXOFS bits to all 1s
disables the respective op amp and routes the DAC current
directly to their respective FSADJI/ AUXI or FSADJQ/AUXQ
pins. This is especially useful where the loads to be driven are
beyond the limited capability of the on-chip amplifier. The
DAC output will open circuit when not enabled (QAUXEN
or IAUXEN = 0).
OPTIONAL
PASSIVE
FILTERING
AD9714/AD9715/
AD9716/AD9717
ADL537x FAMILY
I OR Q INPUTS
1kΩ
I OR Q DAC
AD9714/AD9715/
AD9716/AD9717
AUXDAC
100kΩ
Figure 95. Typical Use of Auxiliary DACs When DC Coupling to Quadrature
Modulator ADL537x Family
CORRECTING FOR NONIDEAL PERFORMANCE OF
QUADRATURE MODULATORS ON THE IF-TO-RF
CONVERSION
DAC-TO-MODULATOR INTERFACING
The auxiliary DACs can be used for local oscillator (LO) cancella-
tion when the DAC output is followed by a quadrature modulator.
This LO feedthrough is caused by the input referred dc offset
voltage of the quadrature modulator (and the DAC output offset
voltage mismatch) and can degrade system performance. Typical
DAC-to-quadrature modulator interfaces are shown in Figure 94
and Figure 95. Often, the input common-mode voltage for the
modulator is much higher than the output compliance range of
the DAC, so that ac coupling or a dc level shift is necessary. If the
required common-mode input voltage on the quadrature modu-
lator matches that of the DAC, the dc blocking capacitors in
Figure 94 can be removed. A low-pass or band-pass passive filter
is recommended when spurious signals from the DAC (distortion
and DAC images) at the quadrature modulator inputs can affect
the system performance. Placing the filter at the location shown
in Figure 94 and Figure 95 allows easy design of the filter, because
the source and load impedances can easily be designed close to
500 ꢀ for a 2 mA full-scale output.
Analog quadrature modulators make it very easy to realize
single sideband radios. However, there are several nonideal
aspects of quadrature modulator performance. Among these
analog degradations are gain mismatch and LO feedthrough.
Gain Mismatch
The gain in the real and imaginary signal paths of the quad-
rature modulator may not be matched perfectly. This leads
to less than optimal image rejection because the cancellation of
the negative frequency image is less than perfect.
LO Feedthrough
The quadrature modulator has a finite dc referred offset, as well
as coupling from its LO port to the signal inputs. These can lead
to a significant spectral spur at the frequency of the Quadrature
Modulator LO.
The AD9714/AD9715/AD9716/AD9717 have the capability
to correct for both of these analog degradations. However,
understand that these degradations drift over temperature;
therefore, if close to optimal single sideband performance
is desired, a scheme for sensing these degradations over
temperature and correcting them may be necessary.
MODULATOR V+
OPTIONAL
AD9714/AD9715/
AD9716/AD9717
I DAC
QUADRATURE
MODULATOR
I INPUTS
PASSIVE
FILTERING
I/Q CHANNEL GAIN MATCHING
AD9714/AD9715/
AD9716/AD9717
AUXDAC1
Fine gain matching is achieved by adjusting the values in the
DAC fine gain adjustment registers. For the I DAC, these values
are in the I DAC gain register (Register 0x03). For the Q DAC,
these values are in the Q DAC gain register (Register 0x06). These
are 6-bit values that cover 2% of full scale. To perform gain
compensation starting from the default values of zero, raise the
value of one of these registers a few steps until it can be deter-
mined if the amplitude of the unwanted image is increased or
decreased. If the unwanted image increased in amplitude, remove
the step and try the same adjustment on the other DAC control
register. Iterate register changes until the rejection cannot be
improved further. If the fine gain adjustment range is not sufficient
to find a null (that is, the register goes full scale with no null
apparent) adjust the course gain settings of the two DACs
accordingly and try again. Variations on this simple method
are possible.
100kΩ
MODULATOR V+
OPTIONAL
PASSIVE
FILTERING
AD9714/AD9715/
AD9716/AD9717
Q DAC
QUADRATURE
MODULATOR
Q INPUTS
AD9714/AD9715/
AD9716/AD9717
AUX2DAC
100kΩ
Figure 94. Typical Use of Auxiliary DACs and On-Chip Resistors for Direct
Coupling to Quadrature Modulators
Rev. 0 | Page 45 of 48
AD9714/AD9715/AD9716/AD9717
5
0
–5
Note that LO feedthrough compensation is independent of
phase compensation. However, gain compensation can affect
the LO compensation because the gain compensation may
change the common-mode level of the signal. The dc offset of
some modulators is common-mode level dependent. Therefore,
it is recommended that the gain adjustment be performed prior
to LO compensation.
–10
–15
–20
–25
–30
–35
–40
–45
–50
–55
–60
–65
–70
–75
–80
–85
–90
–95
LO FEEDTHROUGH COMPENSATION
To achieve LO feedthrough compensation in a circuit, each
output of the two AUXDACs must be connected through a
100 kΩ resistor to one side of the differential DAC output. See
the Auxiliary DACs section for details of how to use AUXDACs.
The purpose of these connections is to drive a very small amount
of current into the nodes at the quadrature modulator inputs,
therefore adding a slight dc bias to one or the other of the
quadrature modulator signal inputs.
447.5
449.0
450.0
451.0
452.5
FREQUENCY (MHz)
Figure 96. AD9714/AD9715/AD9716/AD9717 and ADL5370 with a Single-
Tone Signal at 450 MHz, No Gain or LO Compensation
5
0
To achieve LO feedthrough compensation, the user should
start with the default conditions of the AUXDAC registers,
then increment the magnitude of one or the other AUXDAC
output voltages. While this is being done, the amplitude of the
LO feedthrough at the quadrature modulator output should be
sensed. If the LO feedthrough amplitude increases, try either
decreasing the output voltage of the AUXDAC being adjusted,
or try adjusting the output voltage of the other AUXDAC. It
may take practice before an effective algorithm is achieved.
Using the AD9714/AD9715/AD9716/AD9717 evaluation
board, the LO feedthrough can typically be adjusted down to
the noise floor, although this is not stable over temperature.
–5
–10
–15
–20
–25
–30
–35
–40
–45
–50
–55
–60
–65
–70
–75
–80
–85
–90
–95
447.5
449.0
450.0
451.0
452.5
FREQUENCY (MHz)
RESULTS OF GAIN AND OFFSET CORRECTION
Figure 97. AD9714/AD9715/AD9716/AD9717 and ADL5370 with a Single-
Tone Signal at 450 MHz, Gain and LO Compensation Optimized
The results of gain and offset correction can be seen in Figure 96
and Figure 97. Figure 96 shows the output spectrum of the
quadrature demodulator before gain and offset correction.
Figure 97 shows the output spectrum after correction. The
LO feedthrough spur at 450 MHz has been suppressed to the
noise level. This result can be achieved by applying the correc-
tion, but the correction needs to be repeated after a large change
in temperature.
Note that gain matching improves the negative frequency image
rejection, but it is also related to the phase mismatch in the
quadrature modulator. It can be improved by adjusting the
relative phase between the two quadrature signals at the digital
side or properly designing the low-pass filter between the DACs
and quadrature modulators. Phase mismatch is frequency depen-
dent, so routines have to be developed to adjust it if wideband
signals are desired.
Rev. 0 | Page 46 of 48
AD9714/AD9715/AD9716/AD9717
To evaluate the ADL5370 on this board, the population of these
same components should be reversed so that they are in the
following positions:
MODIFYING THE EVALUATION BOARD TO
USE THE ADL5370 ON-BOARD QUADRATURE
MODULATOR
•
•
•
JP55, JP56, JP76, JP82—soldered
R13, R14, R52, R53—populated
R50, R57, T1, T2—unpopulated
The evaluation board contains an Analog Devices, Inc.,
ADL5370 quadrature modulator. The AD9714/AD9715/
AD9716/AD9717 and the ADL5370 provide an easy-to-
interface DAC/modulator combination that can be easily
characterized on the evaluation board. Solderable jumpers
can be configured to evaluate the single-ended or differential
outputs of the AD9714/ AD9715/AD9716/AD9717. This is
the default configuration from the factory and consists of
the following population of the components:
The AUXDAC outputs can be connected to Test Point TP44 and
Test Point TP45 if LO feedthrough compensation is necessary.
•
•
•
JP55, JP56, JP76, JP82—unsoldered
R13, R14, R52, R53—unpopulated
R50, R57, T1, T2—populated
Rev. 0 | Page 47 of 48
AD9714/AD9715/AD9716/AD9717
OUTLINE DIMENSIONS
6.00
0.60 MAX
BSC SQ
0.60 MAX
PIN 1
INDICATOR
31
40
1
30
PIN 1
INDICATOR
0.50
BSC
TOP
VIEW
4.25
4.10 SQ
3.95
5.75
BSC SQ
EXPOSED
PAD
(BOT TOM VIEW)
0.50
0.40
0.30
21
10
20
11
0.25 MIN
4.50
REF
12° MAX
0.80 MAX
0.65 TYP
FOR PROPER CONNECTION OF
THE EXPOSED PAD, REFER TO
THE PIN CONFIGURATION AND
FUNCTION DESCRIPTIONS
0.05 MAX
0.02 NOM
1.00
0.85
0.80
0.30
0.23
0.18
COPLANARITY
0.08
0.20 REF
SECTION OF THIS DATA SHEET.
SEATING
PLANE
COMPLIANT TO JEDEC STANDARDS MO-220-VJJD-2
Figure 98. 40-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
6 mm × 6 mm, Very Thin Quad
(CP-40-1)
Dimensions shown in millimeters
ORDERING GUIDE
Model
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Package Description
40-Lead LFCSP_VQ
40-Lead LFCSP_VQ
40-Lead LFCSP_VQ
40-Lead LFCSP_VQ
40-Lead LFCSP_VQ
40-Lead LFCSP_VQ
40-Lead LFCSP_VQ
40-Lead LFCSP_VQ
Evaluation Board
Evaluation Board
Evaluation Board
Evaluation Board
Package Option
CP-40-1
CP-40-1
AD9714BCPZ1
AD9714BCPZRL71
AD9715BCPZ1
AD9715BCPZRL71
AD9716BCPZ1
AD9716BCPZRL71
AD9717BCPZ1
AD9717BCPZRL71
AD9714-EBZ1
CP-40-1
CP-40-1
CP-40-1
CP-40-1
CP-40-1
CP-40-1
AD9715-EBZ1
AD9716-EBZ1
AD9717-EBZ1
1 Z = RoHS Compliant Part.
©2008 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D07265-0-8/08(0)
Rev. 0 | Page 48 of 48
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