AD9777AST_技术文档

16-Bit, 160 MSPS 2x/4x/8x

Interpolating Dual TxDAC+^®D/A Converter

AD9777

Versatile input data interface

FEATURES

Twos complement/straight binary data coding

Dual-port or single-port interleaved input data

Single 3.3 V supply operation

16-bit resolution, 160 MSPS/400 MSPS input/output

data rate

Selectable 2×/4×/8× interpolating filter

Programmable channel gain and offset adjustment

f_S/4, f_S/8 digital quadrature modulation capability

Direct IF transmission mode for 70 MHz + IFs

Enables image rejection architecture

Fully compatible SPI® port

Power dissipation: typical 1.2 W @ 3.3 V

On-chip 1.2 V reference

80-lead thin quad flat package, exposed pad (TQFP_EP)

APPLICATIONS

Communications

Excellent ac performance

Analog quadrature modulation architecture

3G, multicarrier GSM, TDMA, CDMA systems

Broadband wireless, point-to-point microwave radios

Instrumentation/ATE

SFDR −73 dBc @ 2 MHz to 35 MHz

WCDMA ACPR 71 dB @ IF = 19.2 MHz

Internal PLL clock multiplier

Selectable internal clock divider

Versatile clock input

Differential/single-ended sine wave or

TTL/CMOS/LVPECL compatible

FUNCTIONAL BLOCK DIAGRAM

IDAC

COS

AD9777

HALF-

BAND

FILTER1*

HALF-

BAND

HALF-

BAND

GAIN

DAC

OFFSET

DAC

FILTER2* FILTER3*

DATA

SIN

f_DAC/2, 4, 8

SIN

ASSEMBLER

IMAGE

REJECTION/

DUAL DAC

MODE

BYPASS

MUX

16

I

I/Q DAC

GAIN/OFFSET

REGISTERS

LATCH

I AND Q

NONINTERLEAVED

OR INTERLEAVED

DATA

16

Q

LATCH

16

FILTER

BYPASS

MUX

COS

WRITE

MUX

CONTROL

I

IDAC

SELECT

OUT

/2

(f_DAC)

CLOCK OUT

/2

SPI INTERFACE AND

CONTROL REGISTERS

PRESCALER

DIFFERENTIAL

CLK

PHASE DETECTOR

AND VCO

* HALF-BAND FILTERS ALSO CAN BE

CONFIGURED FOR ZERO STUFFING ONLY

PLL CLOCK MULTIPLIER AND CLOCK DIVIDER

Figure 1.

Rev. C

Information furnished by Analog Devices is believed to be accurate and reliable. However, no

responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other

rights of third parties that may result from its use. Specifications subject to change without notice. No

license is granted by implication or otherwise under any patent or patent rights of Analog Devices.

Trademarks and registeredtrademarks arethe property of their respective owners.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.

Tel: 781.329.4700

Fax: 781.461.3113

www.analog.com

AD9777

TABLE OF CONTENTS

Features .............................................................................................. 1

Sleep/Power-Down Modes........................................................ 29

Two Port Data Input Mode ....................................................... 29

PLL Enabled, Two-Port Mode.................................................. 30

DATACLK Inversion.................................................................. 30

DATACLK Driver Strength....................................................... 30

PLL Enabled, One-Port Mode .................................................. 30

ONEPORTCLK Inversion......................................................... 31

ONEPORTCLK Driver Strength.............................................. 31

IQ Pairing.................................................................................... 31

PLL Disabled, Two-Port Mode................................................. 31

PLL Disabled, One-Port Mode................................................. 32

Digital Filter Modes ................................................................... 32

Amplitude Modulation.............................................................. 32

Modulation, No Interpolation.................................................. 34

Modulation, Interpolation = 2× ............................................... 35

Modulation, Intermodulation = 4× ......................................... 36

Modulation, Intermodulation = 8× ......................................... 37

Zero Stuffing ............................................................................... 38

Interpolating (Complex Mix Mode)........................................ 38

Operations on Complex Signals............................................... 38

Applications....................................................................................... 1

General Description......................................................................... 4

Product Highlights....................................................................... 4

Specifications..................................................................................... 5

DC Specifications ......................................................................... 5

Dynamic Specifications ............................................................... 6

Digital Specifications ................................................................... 7

Digital Filter Specifications......................................................... 8

Absolute Maximum Ratings............................................................ 9

Thermal Characteristics .............................................................. 9

ESD Caution.................................................................................. 9

Pin Configuration and Function Descriptions........................... 10

Terminology .................................................................................... 12

Typical Performance Characteristics ........................................... 13

Mode Control (via SPI Port)..................................................... 18

Register Description................................................................... 20

Functional Description.................................................................. 22

Serial Interface for Register Control........................................ 22

General Operation of the Serial Interface............................... 22

Instruction Byte .......................................................................... 23

R/W .............................................................................................. 23

N1, N0.......................................................................................... 23

A4, A3, A2, A1, A0..................................................................... 23

Serial Interface Port Pin Descriptions ..................................... 23

MSB/LSB Transfers..................................................................... 23

Notes on Serial Port Operation ................................................ 25

DAC Operation........................................................................... 25

1R/2R Mode ................................................................................ 26

CLOCK Input Configuration ................................................... 26

Programmable PLL .................................................................... 27

Power Dissipation....................................................................... 29

Complex Modulation and Image Rejection of Baseband

Signals .......................................................................................... 39

Image Rejection and Sideband Suppressions of Modulated

Carriers........................................................................................ 41

Applying the Output Configurations........................................... 46

Unbuffered Differential Output, Equivalent Circuit ............. 46

Differential Coupling Using a Transformer............................ 46

Differential Coupling Using an Op Amp................................ 47

Interfacing with the AD8345 Quadrature Modulator........... 47

Evaluation Board ............................................................................ 48

Outline Dimensions....................................................................... 58

Ordering Guide .......................................................................... 58

Rev. C | Page 2 of 60

AD9777

REVISION HISTORY

Edits to Register Description—Address 07h, 0Bh Section...... 16

Edits to Equation 1........................................................................ 16

Edits to MSB/LSB Transfers Section........................................... 18

Changes to Figure 8 ...................................................................... 20

Edits to Programmable PLL Section........................................... 21

Added new Figure 14.................................................................... 22

Renumbered Figures 15 to 69...................................................... 22

Added Two-Port Data Input Mode Section............................... 23

Edits to PLL Enabled, Two-Port Mode Section ........................ 24

Edits to Figure 19 ......................................................................... 24

Edits to Figure 21 .......................................................................... 25

Edits to PLL Disabled, Two-Port Mode Section ....................... 25

Edits to Figure 22 .......................................................................... 25

Edits to Figure 23 .......................................................................... 26

Edits to Figure 26a ........................................................................ 27

Changes to Figures 53 and 54...................................................... 38

Edits to Evaluation Board Section .............................................. 39

Changes to Figures 56 to 59......................................................... 40

Replaced Figures 60 to 69 ............................................................ 42

Updated Outline Dimensions...................................................... 49

1/06—Rev. B to Rev. C

Updated Formatting .........................................................Universal

Changes to Figure 32 .................................................................... 22

Changes to Figure 108.................................................................. 54

Updated Outline Dimensions ..................................................... 58

Changes to Ordering Guide......................................................... 58

6/04—Data Sheet Changed from Rev. A to Rev. B.

Changes to DC Specifications ....................................................... 5

Changes to Absolute Maximum Ratings...................................... 8

Changes to DAC Operation Section........................................... 25

Changes to Figure 49, Figure 50, and Figure 51........................ 29

Changes to the PLL Enabled, One-Port Mode Section............ 30

Changes to the PLL Disabled, One-Port Mode Section........... 32

Changes to the Ordering Guide .................................................. 57

Updated the Outline Dimensions ............................................... 57

3/03—Data Sheet Changed from Rev. 0 to Rev. A.

Edits to Features .............................................................................. 1

Edits to DC Specifications ............................................................. 3

Edits to Dynamic Specifications.................................................... 4

Edits to Pin Function Descriptions............................................... 7

Edits to Table I ............................................................................... 14

Edits to Register Description—Address 02h Section............... 15

Edits to Register Description—Address 03h Section............... 16

7/02—Revision 0: Initial Version

Rev. C | Page 3 of 60

AD9777

GENERAL DESCRIPTION

The AD9777¹is the 16-bit member of the AD977x pin

compatible, high performance, programmable 2×/4×/8×

interpolating TxDAC+ family. The AD977x family features a

serial port interface (SPI) that provides a high level of

programmability, thus allowing for enhanced system level

options. These options include selectable 2×/4×/8× interpola-

tion filters; f_S/2, f_S/4, or f_S/8 digital quadrature modulation with

image rejection; a direct IF mode; programmable channel gain

and offset control; programmable internal clock divider;

straight binary or twos complement data interface; and a single-

port or dual-port data interface.

Targeted at wide dynamic range, multicarrier, and

multistandard systems, the superb baseband performance of the

AD9777 is ideal for wideband CDMA, multicarrier CDMA,

multicarrier TDMA, multicarrier GSM, and high performance

systems employing high-order QAM modulation schemes. The

image rejection feature simplifies and can help to reduce the

number of signal band filters needed in a transmit signal chain.

The direct IF mode helps to eliminate a costly mixer stage for a

variety of communications systems.

PRODUCT HIGHLIGHTS

1. The AD9777 is the 16-bit member of the AD977x pin

compatible, high performance, programmable 2×/4×/8×

interpolating TxDAC+ family.

The selectable 2×/4×/8× interpolation filters simplify the

requirements of the reconstruction filters while simultaneously

enhancing the TxDAC+ family’s pass-band noise/distortion

performance. The independent channel gain and offset adjust

registers allow the user to calibrate LO feedthrough and sideband

suppression errors associated with analog quadrature modulators.

The 6 dB of gain adjustment range can also be used to control the

output power level of each DAC.

2. Direct IF transmission is possible for 70 MHz + IFs

through a novel digital mixing process.

3. f_S/2, f_S/4, and f_S/8 digital quadrature modulation and user

selectable image rejection simplify/remove cascaded SAW

filter stages.

4. A 2×/4×/8× user selectable interpolating filter eases data

rate and output signal reconstruction filter requirements.

The AD9777 features the ability to perform f_S/2, f_S/4, and f_S/8

digital modulation and image rejection when combined with an

analog quadrature modulator. In this mode, the AD9777 accepts I

and Q complex data (representing a single or multicarrier wave-

form), generates a quadrature modulated IF signal along with its

orthogonal representation via its dual DACs, and presents these

two reconstructed orthogonal IF carriers to an analog quadrature

modulator to complete the image rejection upconversion process.

Another digital modulation mode (that is, the direct IF mode)

allows the original baseband signal representation to be fre-

quency translated such that pairs of images fall at multiples of

one-half the DAC update rate.

5. User selectable twos complement/straight binary data

coding.

6. User programmable channel gain control over 1 dB range

in 0.01 dB increments.

7. User programmable channel offset control 10ꢀ over the

FSR.

8. Ultrahigh speed 400 MSPS DAC conversion rate.

9. Internal clock divider provides data rate clock for easy

interfacing.

The AD977x family includes a flexible clock interface accepting

differential or single-ended sine wave or digital logic inputs. An

internal PLL clock multiplier is included and generates the

necessary on-chip high frequency clocks. It can also be disabled

to allow the use of a higher performance external clock source.

An internal programmable divider simplifies clock generation

in the converter when using an external clock source. A flexible

data input interface allows for straight binary or twos comple-

ment formats and supports single-port interleaved or dual-port

data.

10. Flexible clock input with single-ended or differential input,

CMOS, or 1 V p-p LO sine wave input capability.

11. Low power: Complete CMOS DAC operates on 1.2 W

from a 3.1 V to 3.5 V single supply. The 20 mA full-scale

current can be reduced for lower power operation, and

several sleep functions are provided to reduce power

during idle periods.

12. On-chip voltage reference: The AD9777 includes a 1.20 V

temperature compensated band gap voltage reference.

Dual high performance DAC outputs provide a differential

current output programmable over a 2 mA to 20 mA range. The

AD9777 is manufactured on an advanced 0.35 micron CMOS

process, operates from a single-supply of 3.1 V to 3.5 V, and

consumes 1.2 W of power.

13. An 80-lead thin quad flat package, exposed pad

(TQFP_EP).

¹Protected by U.S. Patent Numbers, 5,568,145; 5,689,257; and 5,703,519.

Other patents pending.

Rev. C | Page 4 of 60

AD9777

SPECIFICATIONS

DC SPECIFICATIONS

T_MINto T_MAX, AVDD = 3.3 V, CLKVDD = 3.3 V, DVDD = 3.3 V, PLLVDD = 3.3 V, I_OUTFS= 20 mA, unless otherwise noted.

Table 1.

Parameter

Min

Typ

Max

Unit

RESOLUTION

DC Accuracy¹

16

Bits

Integral Nonlinearity

6

3

LSB

Differential Nonlinearity

ANALOG OUTPUT (for 1R and 2R Gain Setting Modes)

Offset Error

Gain Error (with Internal Reference)

Gain Matching

Full-Scale Output Current²

Output Compliance Range

Output Resistance

−6.5

+6.5

−0.025

−1.0

−1

2

−1.0

0.01

+0.025

+1.0

+1

20

+1.25

% of FSR

mA

V

kΩ

0.1

200

3

Output Capacitance

pF

Gain, Offset Cal DACs, Monotonicity Guaranteed

REFERENCE OUTPUT

Reference Voltage

1.14

0.1

1.20

100

1.26

1.25

V

nA

Reference Output Current³

REFERENCE INPUT

Input Compliance Range

Reference Input Resistance

Small Signal Bandwidth

TEMPERATURE COEFFICIENTS

Offset Drift

Gain Drift (with Internal Reference)

Reference Voltage Drift

POWER SUPPLY

V

kΩ

MHz

7

0.5

0

50

ppm of FSR/°C

ppm/°C

AVDD

Voltage Range

3.1

3.3

72.5

23.3

3.5

76

26

V

mA

4

Analog Supply Current (I_AVDD

I_AVDDin SLEEP Mode

CLKVDD (PLL OFF)

)

Voltage Range

3.3

8.5

3.5

10.0

V

mA

4

Clock Supply Current (I_CLKVDD

CLKVDD (PLL ON)

Clock Supply Current (I_CLKVDD

DVDD

)

23.5

mA

Voltage Range

3.1

3.3

34

380

1.75

6.0

0.4

3.5

41

410

V

mA

mW

W

mW

% of FSR/V

°C

4

Digital Supply Current (I_DVDD

)

Nominal Power Dissipation⁴

5

P_DIS

P_DISin PWDN

Power Supply Rejection Ratio—AVDD

OPERATING RANGE

−40

+85

¹Measured at I_OUTAdriving a virtual ground.

²Nominal full-scale current, I_OUTFS, is 32× the I_REFcurrent.

³Use an external amplifier to drive any external load.

⁴100 MSPS f_DACwith f_OUT= 1 MHz, all supplies = 3.3 V, no interpolation, no modulation.

⁵400 MSPS f_DAC, f_DATA= 50 MSPS, f_S/2 modulation, PLL enabled.

Rev. C | Page 5 of 60

AD9777

DYNAMIC SPECIFICATIONS

T_MINto T_MAX, AVDD = 3.3 V, CLKVDD = 3.3 V, DVDD = 3.3 V, PLLVDD = 0 V, I_OUTFS= 20 mA, Interpolation = 2×, differential

transformer-coupled output, 50 Ω doubly terminated, unless otherwise noted.

Table 2.

Parameter

Min

Typ

Max

Unit

DYNAMIC PERFORMANCE

Maximum DAC Output Update Rate (f_DAC

Output Settling Time (t_ST) (to 0.025%)

Output Rise Time (10% to 90%)¹

Output Fall Time (10% to 90%)¹

Output Noise (I_OUTFS= 20 mA)

AC LINEARITY—BASEBAND MODE

)

400

MSPS

ns

pA/√Hz

11

0.8

50

Spurious-Free Dynamic Range (SFDR) to Nyquist (f_OUT= 0 dBFS)

f_DATA= 100 MSPS, f_OUT= 1 MHz

f_DATA= 65 MSPS, f_OUT= 1 MHz

f_DATA= 65 MSPS, f_OUT= 15 MHz

f_DATA= 78 MSPS, f_OUT= 1 MHz

f_DATA= 78 MSPS, f_OUT= 15 MHz

f_DATA= 160 MSPS, f_OUT= 1 MHz

f_DATA= 160 MSPS, f_OUT= 15 MHz

71

85

84

85

83

85

83

dBc

Spurious-Free Dynamic Range within a 1 MHz Window

f_OUT= 0 dBFS, f_DATA= 100 MSPS, f_OUT= 1 MHz

Two-Tone Intermodulation (IMD) to Nyquist (f_OUT1= f_OUT2= −6 dBFS)

f_DATA= 65 MSPS, f_OUT1= 10 MHz; f_OUT2= 11 MHz

f_DATA= 65 MSPS, f_OUT1= 20 MHz; f_OUT2= 21 MHz

f_DATA= 78 MSPS, f_OUT1= 10 MHz; f_OUT2= 11 MHz

f_DATA= 78 MSPS, f_OUT1= 20 MHz; f_OUT2= 21 MHz

f_DATA= 160 MSPS, f_OUT1= 10 MHz; f_OUT2= 11 MHz

f_DATA= 160 MSPS, f_OUT1= 20 MHz; f_OUT2= 21 MHz

Total Harmonic Distortion (THD)

73

99.1

dBc

85

78

85

78

85

84

dBc

f_DATA= 100 MSPS, f_OUT= 1 MHz; 0 dBFS

−71

−83

dB

Signal-to-Noise Ratio (SNR)

f_DATA= 78 MSPS, f_OUT= 5 MHz; 0 dBFS

f_DATA= 160 MSPS, f_OUT= 5 MHz; 0 dBFS

79

75

dB

Adjacent Channel Power Ratio (ACPR)

WCDMA with 3.84 MHz BW, 5 MHz Channel Spacing

IF = Baseband, f_DATA= 76.8 MSPS

IF = 19.2 MHz, f_DATA= 76.8 MSPS

73

dBc

Four-Tone Intermodulation

21 MHz, 22 MHz, 23 MHz, and 24 MHz at −12 dBFS (f_DATA= MSPS, Missing Center)

AC LINEARITY—IF MODE

Four-Tone Intermodulation at IF = 200 MHz

201 MHz, 202 MHz, 203 MHz, and 204 MHz at −12 dBFS (f_DATA= 160 MSPS, f_DAC= 320 MHz)

76

72

dBFS

¹Measured single-ended into 50 Ω load.

Rev. C | Page 6 of 60

AD9777

DIGITAL SPECIFICATIONS

T_MINto T_MAX, AVDD = 3.3 V, CLKVDD = 3.3 V, PLLVDD = 0 V, DVDD = 3.3 V, I_OUTFS= 20 mA, unless otherwise noted.

Table 3.

Parameter

Min

Typ

Max

Unit

DIGITAL INPUTS

Logic 1 Voltage

Logic 0 Voltage

2.1

3

0

V

0.9

Logic 1 Current

Logic 0 Current

Input Capacitance

CLOCK INPUTS

−10

+10

µA

pF

5

Input Voltage Range

Common-Mode Voltage

Differential Voltage

SERIAL CONTROL BUS

Maximum SCLK Frequency (f_SLCK

0

0.75

0.5

3

2.25

V

1.5

)

15

30

MHz

ns

Mimimum Clock Pulse Width High (t_PWH

Mimimum Clock Pulse Width Low (t_PWL

)

Maximum Clock Rise/Fall Time

1

ms

ns

Minimum Data/Chip Select Setup Time (t_DS)

Minimum Data Hold Time (t_DH)

Maximum Data Valid Time (t_DV)

RESET Pulse Width

25

0

30

1.5

2.1

Inputs (SDI, SDIO, SCLK, CSB)

Logic 1 Voltage

Logic 0 Voltage

3

0

V

0.9

Logic 1 Current

Logic 0 Current

Input Capacitance

−10

+10

µA

pF

5

SDIO Output

Logic 1 Voltage

DRVDD − 0.6

V

Logic 0 Voltage

0.4

V

Logic 1 Current

Logic 0 Current

30

50

mA

Rev. C | Page 7 of 60

AD9777

DIGITAL FILTER SPECIFICATIONS

20

0

Table 4. Half-Band Filter No. 1 (43 Coefficients)

Tap

Coefficient

1, 43

2, 42

3, 41

4, 40

5, 39

6, 38

7, 37

8, 36

8

0

−29

0

67

0

−134

0

244

0

−414

0

673

0

−1,079

0

1,772

0

−3,280

0

10,364

16,384

–20

–40

–60

–80

–100

–120

9, 35

10, 34

11, 33

12, 32

13, 31

14, 30

15, 29

16, 28

17, 27

18, 26

19, 25

20, 24

21, 23

22

0

0.5

1.0

1.5

2.0

8

f_OUT(NORMALIZED TO INPUT DATA RATE)

Figure 2. 2× Interpolating Filter Response

20

0

–20

–40

–60

–80

–100

–120

Table 5. Half-Band Filter No. 2 (19 Coefficients)

Tap

Coefficient

1, 19

2, 18

3, 17

4, 16

5, 15

6, 14

7, 13

8, 12

9, 11

10

19

0

−120

0

438

0

−1,288

0

5,047

8,192

0.5

1.0

1.5

f_OUT(NORMALIZED TO INPUT DATA RATE)

Figure 3. 4× Interpolating Filter Response

20

0

–20

–40

–60

–80

–100

–120

Table 6. Half-Band Filter No. 3 (11 Coefficients)

Tap

1, 11

2, 10

3, 9

4, 8

5, 7

6

Coefficient

7

0

−53

0

302

512

2

4

6

f_OUT(NORMALIZED TO INPUT DATA RATE)

Figure 4. 8× Interpolating Filter Response

Rev. C | Page 8 of 60

AD9777

ABSOLUTE MAXIMUM RATINGS

Table 7.

Parameter

With Respect To

AGND, DGND, CLKGND

AVDD, DVDD, CLKVDD

AGND, DGND, CLKGND

AGND

DGND

CLKGND

Min

−0.3

−4.0

−0.3

−1.0

−0.3

Max

Unit

V

°C

AVDD, DVDD, CLKVDD

AGND, DGND, CLKGND

REFIO, FSADJ1/FSADJ2

I_OUTA, I_OUTB

P1B15 to P1B0, P2B15 to P2B0, RESET

DATACLK/PLL_LOCK

CLK+, CLK−

+4.0

+0.3

AVDD + 0.3

DVDD + 0.3

CLKVDD + 0.3

DVDD + 0.3

125

LPF

CLKGND

DGND

SPI_CSB, SPI_CLK, SPI_SDIO, SPI_SDO

Junction Temperature

Storage Temperature

Lead Temperature (10 sec)

−65

+150

300

Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only;

functional operation of the device at these or any other conditions above those indicated in the operational sections of this specification is

not implied. Exposure to absolute maximum ratings for extended periods may affect device reliability.

THERMAL CHARACTERISTICS

Thermal Resistance

80-lead thin quad flat package, exposed pad [TQFP_EP]

θ_JA= 23.5°C/W (With thermal pad soldered to PCB)

ESD CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the

human body and test equipment and can discharge without detection. Although this product features

proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy

electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance

degradation or loss of functionality.

Rev. C | Page 9 of 60

AD9777

PIN CONFIGURATION AND FUNCTION DESCRIPTIONS

80 79 78 77 76 75 74 73 72 71 70 69 68 67 66 65 64 63 62 61

1

60

59

58

57

56

55

54

53

52

51

50

49

48

47

46

45

44

43

42

41

FSADJ1

FSADJ2

REFIO

RESET

SPI_CSB

SPI_CLK

SPI_SDIO

SPI_SDO

DGND

CLKVDD

LPF

PIN 1

2

3

CLKVDD

CLKGND

CLK+

4

5

6

CLK–

7

CLKGND

DATACLK/PLL_LOCK

DGND

AD9777

TxDAC+

TOP VIEW

(Not to Scale)

8

9

10

11

12

13

14

15

16

17

18

19

20

DVDD

P2B0 (LSB)

P2B1

P1B15 (MSB)

P1B14

P2B2

P1B13

P2B3

P1B12

P2B4

P1B11

P1B10

P2B5

DGND

DVDD

P1B9

P2B6

P1B8

P2B7

21 22 23 24 25 26 27 28 29 30 31 32 33 34 35 36 37 38 39 40

NC = NO CONNECT

Figure 5. Pin Configuration

Rev. C | Page 10 of 60

AD9777

Table 8. Pin Function Description

Pin No.

Mnemonic

CLKVDD

LPF

Description

1, 3

2

Clock Supply Voltage.

PLL Loop Filter.

4, 7

5

6

CLKGND

CLK+

CLK−

Clock Supply Common.

Differential Clock Input.

8

DATACLK/PLL_LOCK

With the PLL enabled, this pin indicates the state of the PLL. A read of a Logic 1 indicates the

PLL is in the locked state. Logic 0 indicates the PLL has not achieved lock. This pin may also be

programmed to act as either an input or output (Address 02h, Bit 3) DATACLK signal running

at the input data rate.

9, 17, 25,

35, 44, 52

DGND

Digital Common.

10, 18, 26,

36, 43, 51

11 to 16, 19

to 24, 27 to

30

DVDD

Digital Supply Voltage.

Port 1 Data Inputs.

P1B15 (MSB) to P1B0 (LSB)

31

IQSEL/P2B15 (MSB)

ONEPORTCLK/P2B14

In one-port mode, IQSEL = 1 followed by a rising edge of the differential input clock latches

the data into the I channel input register. IQSEL = 0 latches the data into the Q channel input

register. In two-port mode, this pin becomes the Port 2 MSB.

With the PLL disabled and the AD9777 in one-port mode, this pin becomes a clock output

that runs at twice the input data rate of the I and Q channels. This allows the AD9777 to

accept and demux interleaved I and Q data to the I and Q input registers.

32

33, 34, 37 to

42, 45 to 50

53

P2B13 to P2B0 (LSB)

SPI_SDO

Port 2 Data Inputs.

In the case where SDIO is an input, SDO acts as an output. When SDIO becomes an output,

SDO enters a High-Z state. This pin can also be used as an output for the data rate clock. For

more information, see the Two Port Data Input Mode section.

54

55

56

57

SPI_SDIO

SPI_CLK

SPI_CSB

RESET

Bidirectional Data Pin. Data direction is controlled by Bit 7 of Register Address 00h. The

default setting for this bit is 0, which sets SDIO as an input.

Data input to the SPI port is registered on the rising edge of SPI_CLK. Data output on the SPI

port is registered on the falling edge.

Chip Select/SPI Data Synchronization. On momentary logic high, resets SPI port logic and

initializes instruction cycle.

Logic 1 resets all of the SPI port registers, including Address 00h, to their default values. A

software reset can also be done by writing a Logic 1 to SPI Register 00h, Bit 5. However, the

software reset has no effect on the bits in Address 00h.

58

59

60

REFIO

Reference Output, 1.2 V Nominal.

Full-Scale Current Adjust, Q Channel.

Full-Scale Current Adjust, I Channel.

Analog Supply Voltage.

FSADJ2

FSADJ1

AVDD

61, 63, 65,

76, 78, 80

62, 64, 66, 67, AGND

70, 71, 74,

Analog Common.

75, 77, 79

68, 69

72, 73

I_OUTB2, I_OUTA2

I_OUTB1, I_OUTA1

Differential DAC Current Outputs, Q Channel.

Differential DAC Current Outputs, I Channel.

Rev. C | Page 11 of 60

AD9777

TERMINOLOGY

Adjacent Channel Power Ratio (ACPR)

A ratio in dBc between the measured power within a channel

relative to its adjacent channel.

Monotonicity

A DAC is monotonic if the output either increases or remains

constant as the digital input increases.

Complex Image Rejection

Offset Error

In a traditional two-part upconversion, two images are created

around the second IF frequency. These images are redundant and

have the effect of wasting transmitter power and system bandwidth.

By placing the real part of a second complex modulator in series

with the first complex modulator, either the upper or lower

frequency image near the second IF can be rejected.

The deviation of the output current from the ideal of 0 is called

offset error. For I_OUTA, 0 mA output is expected when the inputs

are all 0. For I_OUTB, 0 mA output is expected when all inputs are

set to 1.

Output Compliance Range

The range of allowable voltage at the output of a current-output

DAC. Operation beyond the maximum compliance limits may

cause either output stage saturation or breakdown, resulting in

nonlinear performance.

Complex Modulation

The process of passing the real and imaginary components of a

signal through a complex modulator (transfer function = e^jωt

=

cosωt + jsinωt) and realizing real and imaginary components

on the modulator output.

Pass Band

Frequency band in which any input applied therein passes

unattenuated to the DAC output.

Differential Nonlinearity (DNL)

DNL is the measure of the variation in analog value, normalized to

full scale, associated with a 1 LSB change in digital input code.

Power Supply Rejection

The maximum change in the full-scale output as the supplies

are varied from minimum to maximum specified voltages.

Gain Error

The difference between the actual and ideal output span. The

actual span is determined by the output when all inputs are set

to 1 minus the output when all inputs are set to 0.

Signal-to-Noise Ratio (SNR)

SNR is the ratio of the rms value of the measured output signal

to the rms sum of all other spectral components below the

Nyquist frequency, excluding the first six harmonics and dc.

The value for SNR is expressed in decibels.

Glitch Impulse

Asymmetrical switching times in a DAC give rise to undesired

output transients that are quantified by a glitch impulse. It is

specified as the net area of the glitch in pV-s.

Spurious-Free Dynamic Range

The difference, in dB, between the rms amplitude of the

output signal and the peak spurious signal over the specified

bandwidth.

Group Delay

Number of input clocks between an impulse applied at the

device input and the peak DAC output current. A half-band FIR

filter has constant group delay over its entire frequency range.

Settling Time

The time required for the output to reach and remain within a

specified error band about its final value, measured from the

start of the output transition.

Impulse Response

Response of the device to an impulse applied to the input.

Interpolation Filter

Stop-Band Rejection

If the digital inputs to the DAC are sampled at a multiple rate of

f_DATA(interpolation rate), a digital filter can be constructed that

has a sharp transition band near f_DATA/2. Images that would

typically appear around f_DAC(output data rate) can be greatly

suppressed.

The amount of attenuation of a frequency outside the pass band

applied to the DAC, relative to a full-scale signal applied at the

DAC input within the pass band.

Temperature Drift

It is specified as the maximum change from the ambient (25°C)

value to the value at either T_MINor T_MAX. For offset and gain

drift, the drift is reported in ppm of full-scale range (FSR) per

°C. For reference drift, the drift is reported in ppm per °C.

Linearity Error

(Also called integral nonlinearity or INL) Linearity error is

defined as the maximum deviation of the actual analog output

from the ideal output, determined by a straight line drawn from

zero to full scale.

Total Harmonic Distortion (THD)

THD is the ratio of the rms sum of the first six harmonic

components to the rms value of the measured fundamental. It is

expressed as a percentage or in decibels (dB).

Rev. C | Page 12 of 60

AD9777

TYPICAL PERFORMANCE CHARACTERISTICS

T = 25°C, AVDD = 3.3 V, CLKVDD = 3.3 V, DVDD = 3.3 V, I_OUTFS= 20 mA, Interpolation = 2×, differential transformer-coupled output,

50 Ω doubly terminated, unless otherwise noted.

10

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

–10

–20

–30

–40

–50

–60

–70

–80

–90

0

65

130

0

50

100

150

FREQUENCY (MHz)

Figure 6. Single-Tone Spectrum @ f_DATA= 65 MSPS with f_OUT= f_DATA/3

Figure 9. Single-Tone Spectrum @ f_DATA= 78 MSPS with f_OUT= f_DATA/3

90

0dBFS

85

80

75

–6dBFS

80

75

70

–6dBFS

70

–12dBFS

65

–12dBFS

65

60

55

50

60

55

50

0

5

10

15

20

25

30

0

5

10

15

20

25

30

FREQUENCY (MHz)

Figure 7. In-Band SFDR vs. f_OUT@ f_DATA= 65 MSPS

Figure 10. In-Band SFDR vs. f_OUT@ f_DATA= 78 MSPS

90

85

80

75

70

65

60

55

50

90

85

80

75

70

65

60

55

50

0dBFS

–12dBFS

–12dBFS 0dBFS

–6dBFS

0

5

10

15

20

25

30

0

5

10

15

20

25

30

FREQUENCY (MHz)

Figure 8. Out-of-Band SFDR vs. f_OUT@ f_DATA= 65 MSPS

Figure 11. Out-of-Band SFDR vs. f_OUT@ f_DATA= 78 MSPS

Rev. C | Page 13 of 60

AD9777

10

0

90

85

80

75

70

65

60

55

50

0dBFS

–10

–20

–30

–40

–50

–60

–70

–80

–3dBFS

–6dBFS

–90

0

100

200

300

0

5

10

15

20

25

30

FREQUENCY (MHz)

Figure 12. Single-Tone Spectrum @ f_DATA= 160 MSPS with f_OUT= f_DATA/3

Figure 15. Third-Order IMD Products vs. Two-Tone f_OUT@ f_DATA= 65 MSPS

90

0dBFS

85

80

75

80

75

70

–6dBFS

–3dBFS

70

65

60

55

50

–6dBFS

–12dBFS

65

60

55

50

0

10

20

30

40

50

0

5

10

15

20

25

30

FREQUENCY (MHz)

Figure 13. In-Band SFDR vs. f_OUT@ f_DATA= 160 MSPS

Figure 16. Third-Order IMD Products vs. Two-Tone f_OUT@ f_DATA= 78 MSPS

90

85

80

75

70

65

60

55

50

90

0dBFS

85

80

–3dBFS

75

–6dBFS

–12dBFS

–6dBFS

70

65

60

55

50

0dBFS

0

10

20

30

40

50

0

10

20

30

40

50

60

FREQUENCY (MHz)

Figure 14. Out-of-Band SFDR vs. f_OUT@ f_DATA= 160 MSPS

Figure 17. Third-Order IMD Products vs. Two-Tone f_OUT@ f_DATA= 160 MSPS

Rev. C | Page 14 of 60

AD9777

90

85

80

75

70

65

60

55

50

90

85

80

75

70

65

60

55

50

4

×

0dBFS

–3dBFS

–6dBFS

8×

1

×

2×

0

10

20

30

40

50

60

3.1

3.2

3.3

3.4

3.5

FREQUENCY (MHz)

AVDD (V)

Figure 18. Third-Order IMD Products vs. Two-Tone f_OUTand Interpolation

Rate, 1× f_DATA= 160 MSPS, 2× f_DATA= 160 MSPS,

Figure 21. Third-Order IMD Products vs. AVDD @ f_OUT= 10 MHz,

f

_DAC= 320 MSPS, f_DATA= 160 MSPS

4× f_DATA= 80 MSPS, 8× f_DATA= 50 MSPS

90

85

80

75

70

65

60

55

50

8×

85

PLL OFF

80

4×

75

1×

70

2×

PLL ON

65

60

55

50

–15

–10

–5

0

50

100

150

A

(dBFS)

INPUT DATA RATE (MSPS)

OUT

Figure 19. Third-Order IMD Products vs. Two-Tone A_OUTand Interpolation

Rate, f_DATA= 50 MSPS in All Cases, 1× f_DAC= 50 MSPS, 2× f_DAC= 100 MSPS,

4× f_DAC= 200 MSPS, 8× f_DAC= 400 MSPS

Figure 22. SNR vs. Data Rate for f_OUT= 5 MHz

90

85

80

75

70

65

60

55

50

78MSPS

85

80

75

70

65

60

55

50

0dBFS

–6dBFS

f_DATA= 65MSPS

160MSPS

–12dBFS

3.1

3.2

3.3

AVDD (V)

3.4

3.5

–50

0

50

100

TEMPERATURE (°C)

Figure 20. SFDR vs. AVDD f_OUT= 10 MHz, f_DAC= 320 MSPS, f_DATA= 160 MSPS

Figure 23. SFDR vs. Temperature @ f_OUT= f_DATA/11

Rev. C | Page 15 of 60

AD9777

0

–20

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

–40

–60

–80

–100

0

–100

0

5

10

15

20

25

30

35

40

45

50

100

150

FREQUENCY (MHz)

Figure 27. Two-Tone IMD Performance, f_DATA= 90 MSPS, Interpolation = 4×

Figure 24. Single-Tone Spurious Performance, f_OUT= 10 MHz,

f

_DATA= 150 MSPS, No Interpolation

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

0

–20

–40

–60

–80

–100

0

50

100

150

200

250

300

0

10

20

30

40

50

FREQUENCY (MHz)

Figure 25. Two-Tone IMD Performance,

_DATA= 150 MSPS, No Interpolation

Figure 28. Single-Tone Spurious Performance, f_OUT= 10 MHz,

_DATA= 80 MSPS, Interpolation = 4×

f

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

0

–20

–40

–60

–80

–100

0

50

100

150

200

250

300

0

5

10

15

20

25

FREQUENCY (MHz)

Figure 26. Single-Tone Spurious Performance, f_OUT= 10 MHz,

_DATA= 150 MSPS, Interpolation = 2×

Figure 29. Two-Tone IMD Performance, f_OUT= 10 MHz,

f_DATA= 50 MSPS, Interpolation = 8×

f

Rev. C | Page 16 of 60

AD9777

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

0

100

200

300

400

0

20

40

60

80

FREQUENCY (MHz)

Figure 30. Single-Tone Spurious Performance, f_OUT= 10 MHz,

_DATA= 50 MSPS, Interpolation = 8×

Figure 31. Eight-Tone IMD Performance, f_DATA= 160 MSPS, Interpolation = 8x

f

Rev. C | Page 17 of 60

AD9777

MODE CONTROL (VIA SPI PORT)

Table 9. Mode Control via SPI Port¹

Address Bit 7

Bit 6

Bit 5

Bit 4

Bit 3

Bit 2

Bit 1

Bit 0

00h

01h

02h

SDIO

LSB, MSB First

0 = MSB

1 = LSB

Software

Reset on

Logic 1

Sleep Mode

Logic 1

Shuts Down

the DAC

Output

Currents

Power-Down

Mode Logic 1

Shuts Down All Current Set by

Digital and

Analog

Functions

1R/2R Mode

DAC Output

PLL_LOCK

Indicator

Bidirectional

0 = Input

1 = I/O

One or Two

External

Resistors

0 = 2R, 1 = 1R

Filter

Modulation Modulation

Mode

(None, f_S/2,

f_S/4, f_S/8)

0 = No Zero

Stuffing on

Interpolation

Filters, Logic 1

Enables Zero

Stuffing

1 = Real Mix

Mode

0 = Complex

Mix Mode

0 = e^−jωt

1 = e^+jωt

DATACLK/

PLL_LOCK²

Select

0 =

PLL_LOCK

Interpolation Interpolation

Rate

(1×, 2×,

4×, 8×)

Mode

(None, f_S/2,

f_S/4, f_S/8)

Rate

(1×, 2×,

4×, 8×)

1 =

DATACLK

0 = Signed

Input Data

1 =

0 = Two-Port

Mode

1 = One-Port

Mode

DATACLK

Driver

Strength

DATACLK

Invert

0 = No

Invert

1 = Invert

ONEPORTCLK

Invert

0 = No Invert

1 = Invert

IQSEL

Invert

0 = No

Invert

1 = Invert

Q First

0 = I First

1 = Q First

Unsigned

03h

04h

Data Rate²

Clock

Output

PLL Divide

(Prescaler)

Ratio

PLL Divide

(Prescaler)

Ratio

0 = PLL

0 = Automatic

Charge Pump

Control

PLL

OFF²

Charge

Pump

Control

Charge

Pump

Control

Charge

Pump

Control

1 = PLL ON

1 =

Programmable

05h

06h

07h

08h

IDAC

Fine Gain

Adjustment

IDAC

Fine Gain

Adjustment

IDAC

Fine Gain

Adjustment Adjustment

IDAC

Fine Gain

IDAC

Fine Gain

Adjustment

IDAC

Fine Gain

Adjustment

IDAC

Fine Gain

Adjustment Adjustment

IDAC

Fine Gain

IDAC

Coarse Gain

Adjustment

IDAC

Coarse Gain

Adjustment

IDAC IDAC

Coarse Gain Coarse Gain

Adjustment Adjustment

IDAC Offset

Adjustment

Bit 9

IDAC Offset

Adjustment

Bit 8

IDAC Offset IDAC Offset

Adjustment Adjustment

Bit 7

IDAC Offset

Adjustment

Bit 5

IDAC Offset

Adjustment

Bit 4

IDAC Offset IDAC Offset

Adjustment Adjustment

Bit 3

Bit 6

Bit 2

IDAC I_OFFSET

Direction

0 = I_OFFSETon

I_OUTA

IDAC Offset IDAC Offset

Adjustment Adjustment

Bit 1

Bit 0

1 = I_OFFSETon

I

_OUTB

09h

QDAC

Fine Gain

Adjustment

Fine Gain

Adjustment

Fine Gain

Adjustment Adjustment

Fine Gain

Adjustment

Fine Gain

Adjustment

Fine Gain

Adjustment Adjustment

Fine Gain

Rev. C | Page 18 of 60

AD9777

Address Bit 7

Bit 6

Bit 5

Bit 4

Bit 3

Bit 2

Bit 1

Bit 0

0Ah

QDAC

Coarse Gain

Adjustment

Coarse Gain

Adjustment

Coarse Gain Coarse Gain

Adjustment Adjustment

0Bh

0Ch

QDAC Offset QDAC Offset

QDAC

Offset

Adjustment Bit 6

Bit 7

QDAC Offset QDAC Offset

QDAC Offset

Adjustment

Bit 4

QDAC

Offset

QDAC

Offset

Adjustment

Bit 9

Adjustment

Bit 8

Adjustment

Bit 5

Adjustment Adjustment

Bit 3

Bit 2

QDAC I_OFFSET

Direction

0 = I_OFFSETon

I_OUTA

QDAC

Offset

QDAC

Offset

Adjustment Adjustment

Bit 1

Bit 0

1 = I_OFFSETon

I_OUTB

0Dh

Version

Register

Version

Register

Version

Register

Version

Register

¹Default values are shown in bold.

²For more information, see the Two Port Data Input Mode section.

Rev. C | Page 19 of 60

AD9777

Bit 3: Logic 1 enables zero stuffing mode for interpolation filters.

REGISTER DESCRIPTION

Address 00h

Bit 2: Default (1) enables the real mix mode. The I and Q data

channels are individually modulated by f_S/2, f_S/4, or f_S/8 after

the interpolation filters. However, no complex modulation is

done. In the complex mix mode (Logic 0), the digital

modulators on the I and Q data channels are coupled to create a

digital complex modulator. When the AD9777 is applied in

conjunction with an external quadrature modulator, rejection

can be achieved of either the higher or lower frequency image

around the second IF frequency (that is, the LO of the analog

quadrature modulator external to the AD9777) according to the

bit value of Register 01h, Bit 1.

Bit 7: Logic 0 (default) causes the SPI_SDIO pin to act as an

input during the data transfer (Phase 2) of the communications

cycle. When set to 1, SPI_SDIO can act as an input or output,

depending on Bit 7 of the instruction byte.

Bit 6: Logic 0 (default). Determines the direction (LSB/MSB

first) of the communications and data transfer communications

cycles. Refer to the MSB/LSB Transfers section for more details.

Bit 5: Writing a 1 to this bit resets the registers to their default

values and restarts the chip. The RESET bit always reads back 0.

Register Address 00h bits are not cleared by this software reset.

However, a high level at the RESET pin forces all registers,

including those in Address 00h, to their default state.

Bit 1: Logic 0 (default) causes the complex modulation to be of

the form e^−jωt, resulting in the rejection of the higher frequency

image when the AD9777 is used with an external quadrature

modulator. A Logic 1 causes the modulation to be of the form

e^+jωt, which causes rejection of the lower frequency image.

Bit 4: Sleep Mode. A Logic 1 to this bit shuts down the DAC

output currents.

Bit 0: In two-port mode, a Logic 0 (default) causes Pin 8 to act

as a lock indicator for the internal PLL. A Logic 1 in this register

causes Pin 8 to act as a DATACLK. For more information, see

the Two Port Data Input Mode section.

Bit 3: Power-Down. Logic 1 shuts down all analog and digital

functions except for the SPI port.

Bit 2: 1R/2R Mode. The default (0) places the AD9777 in two

resistor mode. In this mode, the I_REFcurrents for the I and Q

DAC references are set separately by the R_SETresistors on

FSADJ2 and FSADJ1 (Pins 59 and 60). In 2R mode, assuming

the coarse gain setting is full scale and the fine gain setting is 0,

Address 02h

Bit 7: Logic 0 (default) causes data to be accepted on the inputs

as twos complement binary. Logic 1 causes data to be accepted

as straight binary.

I

_FULLSCALE1= 32 × V_REF/FSADJ1 and I_FULLSCALE2= 32 ×

_REF/FSADJ2. With this bit set to 1, the reference currents for

both I and Q DACs are controlled by a single resistor on Pin 60.

_FULLSCALEin one resistor mode for both the I and Q DACs is half

V

Bit 6: Logic 0 (default) places the AD9777 in two-port mode. I

and Q data enters the AD9777 via Ports 1 and 2, respectively. A

Logic 1 places the AD9777 in one-port mode in which

interleaved I and Q data is applied to Port 1. See Table 8 for

detailed information on how to use the DATACLK/PLL_LOCK,

IQSEL, and ONEPORTCLK modes.

I

of what it would be in 2R mode, assuming all other conditions

(R_SET, register settings) remain unchanged. The full-scale

current of each DAC can still be set to 20 mA by choosing a

resistor of half the value of the R_SETvalue used in 2R mode.

Bit 5: DATACLK Driver Strength. With the internal PLL

disabled and this bit set to Logic 0, it is recommended that

DATACLK be buffered. When this bit is set to Logic 1,

DATACLK acts as a stronger driver capable of driving small

capacitive loads.

Bit 1: PLL_LOCK Indicator. When the PLL is enabled, reading

this bit gives the status of the PLL. A Logic 1 indicates the PLL

is locked. A Logic 0 indicates an unlocked state.

Address 01h

Bit 7, Bit 6: This is the filter interpolation rate according to the

following table.

Bit 4: Logic 0 (default). A value of 1 inverts DATACLK at Pin 8.

Bit 2: Logic 0 (default). A value of 1 inverts ONEPORTCLK

at Pin 32.

00

01

10

11

1×

2×

4×

8×

Bit 1: Logic 0 (default) causes IQSEL = 0 to direct input data to

the I channel, while IQSEL = 1 directs input data to the Q

channel.

Bit 5 and Bit 4: This is the modulation mode according to the

following table.

Bit 0: Logic 0 (default) defines IQ pairing as IQ, IQ…, while

programming a Logic 1 causes the pair ordering to

be QI, QI…

00

01

10

11

none

f_S/2

f_S/4

f_S/8

Rev. C | Page 20 of 60

AD9777

Address 03h

Address 05h, 09h

Bit 7: This allows the data rate clock (divided down from the

DAC clock) to be output at either the DATACLK/PLL_LOCK

pin (Pin 8) or at the SPI_SDO pin (Pin 53). The default of 0 in

this register enables the data rate clock at DATACLK/

PLL_LOCK, while a 1 in this register causes the data rate clock

to be output at SPI_SDO. For more information, see the Two

Port Data Input Mode section.

Bit 7, Bit 6, Bit 5, Bit 4, Bit 3, Bit 2, Bit 1, and Bit 0: These bits

represent an 8-bit binary number (Bit 7 MSB) that defines the

fine gain adjustment of the I (05h) and Q (09h) DAC according

to Equation 1.

Address 06h, 0Ah

Bit 3, Bit 2, Bit 1, and Bit 0: These bits represent a 4-bit binary

number (Bit 3 MSB) that defines the coarse gain adjustment of

the I (06h) and Q (0Ah) DACs according to Equation 1.

Bit 1, Bit 0: Setting this divide ratio to a higher number allows

the VCO in the PLL to run at a high rate (for best performance)

while the DAC input and output clocks run substantially slower.

The divider ratio is set according to the following table.

Address 07h, 0Bh

Bit 7, Bit 6, Bit 5, Bit 4, Bit 3, Bit 2, Bit 1, and Bit 0: These bits

are used in conjunction with Address 08h, 0Ch, Bits 1, 0.

00

÷1

÷2

÷4

÷8

01

Address 08h, 0Ch

10

Bit 1 and Bit 0: The 10 bits from these two address pairs (07h,

08h and 0Bh, 0Ch) represent a 10-bit binary number that

defines the offset adjustment of the I and Q DACs according to

Equation 1. (07h, 0Bh–Bit 7 MSB/08h, 0Ch–Bit 0 LSB).

11

Address 04h

Bit 7: Logic 0 (default) disables the internal PLL. Logic 1

Address 08h, 0Ch

enables the PLL.

Bit 7: This bit determines the direction of the offset of the I

(08h) and Q (0Ch) DACs. A Logic 0 applies a positive offset

current to I_OUTA, while a Logic 1 applies a positive offset current

to I_OUTB. The magnitude of the offset current is defined by the

bits in Addresses 07h, 0Bh, 08h, 0Ch according to Equation 1.

Equation 1 shows I_OUTAand I_OUTBas a function of fine gain,

coarse gain, and offset adjustment when using 2R mode. In 1R

mode, the current I_REFis created by a single FSADJ resistor

(Pin 60). This current is divided equally into each channel so that

a scaling factor of one-half must be added to these equations for

full-scale currents for both DACs and the offset.

Bit 6: Logic 0 (default) sets the charge pump control to

automatic. In this mode, the charge pump bias current is

controlled by the divider ratio defined in Address 03h, Bits 1

and 0. Logic 1 allows the user to manually define the charge

pump bias current using Address 04h, Bits 2, 1, and 0. Adjusting

the charge pump bias current allows the user to optimize the

noise/settling performance of the PLL.

Bit 2, Bit 1, Bit 0: With the charge pump control set to manual,

these bits define the charge pump bias current according to the

following table.

000

001

010

011

111

50 µA

100 µA

200 µA

400 µA

800 µA

6×I

COARSE +1

3×I

32

FINE

256

1024 DATA

⎡

⎢

⎤

⎡

⎤

⎥

⎛

⎜

⎝

⎞

⎟

⎠

⎛

⎞

⎟

⎛

⎜

⎝

⎞

⎟

⎠

⎛

⎜

⎝

⎞⎛

⎟⎜

⎠⎝

⎞

⎟

⎠

⎛

⎜

⎝

⎞

⎟

⎠

REF

I_OUTA

=

−

×

(A)

⎜

2¹⁶

⎥

⎦

⎢

⎣

8

16

24

⎝

⎠

⎦ ⎣

16

⎡

⎤

⎥

⎛

⎞

⎜

⎟

⎜

⎠

⎝

⎞

⎟

⎠

6×I

COARSE +1

3×I

32

FINE

256

1024

24

2

− DATA −1

⎡

⎛

⎜

⎝

⎞⎛

⎞

⎟

⎠

⎛

⎜

⎝

⎞⎛

⎟⎜

⎠⎝

⎞

⎟

⎠

⎛

⎢⎜

⎝

REF

I_OUTB

×

⎥

(A)

(1)

⎜

⎟

2¹⁶

⎢

⎣

8

16

⎠⎝

⎦ ⎢

⎥

⎦

⎣

OFFSET

1024

⎛

⎜

⎞

⎠

I_OFFSET= 4×I_REF

(A)

⎟

⎝

Rev. C | Page 21 of 60

AD9777

FUNCTIONAL DESCRIPTION

The AD9777 dual interpolating DAC consists of two data

channels that can be operated independently or coupled to form

a complex modulator in an image reject transmit architecture.

Each channel includes three FIR filters, making the AD9777

capable of 2×, 4×, or 8× interpolation. High speed input and

output data rates can be achieved within the following

limitations.

SDO (PIN 53)

SDIO (PIN 54)

AD9777 SPI PORT

INTERFACE

SPI_CLK (PIN 55)

CSB (PIN 56)

Figure 32. SPI Port Interface

Interpolation Rate

(MSPS)

Input Data Rate

(MSPS)

DAC Sample Rate

(MSPS)

SERIAL INTERFACE FOR REGISTER CONTROL

The AD9777 serial port is a flexible, synchronous serial

communications port that allows easy interface to many

industry-standard microcontrollers and microprocessors. The

serial I/O is compatible with most synchronous transfer

formats, including both the Motorola SPI® and Intel® SSR

protocols. The interface allows read/write access to all registers

that configure the AD9777. Single- or multiple-byte transfers

are supported as well as MSB first or LSB first transfer formats.

The AD9777’s serial interface port can be configured as a single

pin I/O (SDIO) or two unidirectional pins for I/O (SDIO/SDO).

1×

2×

4×

8×

160

100

50

160

320

400

Both data channels contain a digital modulator capable of

mixing the data stream with an LO of f_DAC/2, f_DAC/4, or f_DAC/8,

where f_DACis the output data rate of the DAC. A zero stuffing

feature is also included and can be used to improve pass-band

flatness for signals being attenuated by the SIN(x)/x

characteristic of the DAC output. The speed of the AD9777,

combined with its digital modulation capability, enables direct

IF conversion architectures at 70 MHz and higher.

GENERAL OPERATION OF THE SERIAL INTERFACE

There are two phases to a communication cycle with the

AD9777. Phase 1 is the instruction cycle, which is the writing of

an instruction byte into the AD9777 coincident with the first

eight SCLK rising edges. The instruction byte provides the

AD9777 serial port controller with information regarding the

data transfer cycle, which is Phase 2 of the communication

cycle. The Phase 1 instruction byte defines whether the upcom-

ing data transfer is read or write, the number of bytes in the

data transfer, and the starting register address for the first byte

of the data transfer. The first eight SCLK rising edges of each

communication cycle are used to write the instruction byte into

the AD9777.

The digital modulators on the AD9777 can be coupled to form

a complex modulator. By using this feature with an external

analog quadrature modulator, such as the Analog Devices

AD8345, an image rejection architecture can be enabled. To

optimize the image rejection capability, as well as LO feed-

through in this architecture, the AD9777 offers programmable

(via the SPI port) gain and offset adjust for each DAC.

Also included on the AD9777 are a phase-locked loop (PLL)

clock multiplier and a 1.20 V band gap voltage reference. With

the PLL enabled, a clock applied to the CLK+/CLK− inputs is

frequency multiplied internally and generates all necessary

internal synchronization clocks. Each 16-bit DAC provides two

complementary current outputs whose full-scale currents can

be determined either from a single external resistor or inde-

pendently from two separate resistors (see the 1R/2R Mode

section). The AD9777 features a low jitter, differential clock

input that provides excellent noise rejection while accepting a

sine or square wave input. Separate voltage supply inputs are

provided for each functional block to ensure optimum noise

and distortion performance.

A Logic 1 on the SPI_CSB pin, followed by a logic low, resets

the SPI port timing to the initial state of the instruction cycle.

This is true regardless of the present state of the internal

registers or the other signal levels present at the inputs to the

SPI port. If the SPI port is in the middle of an instruction cycle

or a data transfer cycle, none of the present data is written.

The remaining SCLK edges are for Phase 2 of the

communication cycle. Phase 2 is the actual data transfer

between the AD9777 and the system controller. Phase 2 of the

communication cycle is a transfer of one to four data bytes, as

determined by the instruction byte. Normally, using one multi-

byte transfer is the preferred method. However, single byte data

transfers are useful to reduce CPU overhead when register

access requires one byte only. Registers change immediately

upon writing to the last bit of each transfer byte.

Sleep and power-down modes can be used to turn off the DAC

output current (sleep) or the entire digital and analog sections

(power-down) of the chip. A SPI-compliant serial port is used

to program the many features of the AD9777. Note that in

power-down mode, the SPI port is the only section of the chip

still active.

Rev. C | Page 22 of 60

AD9777

SPI_CSB (Pin 56)—Chip Select

INSTRUCTION BYTE

The instruction byte contains the information shown below.

Active low input starts and gates a communication cycle. It

allows more than one device to be used on the same serial

communications lines. The SPI_SDO and SPI_SDIO pins go to

a high impedance state when this input is high. Chip select

should stay low during the entire communication cycle.

N1

0

N0

0

Description

Transfer 1 Byte

Transfer 2 Bytes

Transfer 3 Bytes

Transfer 4 Bytes

0

1

0

1

SPI_SDIO (Pin 54)—Serial Data I/O

R/W

Data is always written into the AD9777 on this pin. However,

this pin can be used as a bidirectional data line. Bit 7 of Register

Address 00h controls the configuration of this pin. The default

is Logic 0, which configures the SPI_SDIO pin as

unidirectional.

Bit 7 of the instruction byte determines whether a read or a

write data transfer occurs after the instruction byte write. Logic

1 indicates read operation. Logic 0 indicates a write operation.

N1, N0

SPI_SDO (Pin 53)—Serial Data Out

Bit 6 and Bit 5 of the instruction byte determine the number of

bytes to be transferred during the data transfer cycle. The bit

decodes are shown in the following table.

Data is read from this pin for protocols that use separate lines

for transmitting and receiving data. In the case where the

AD9777 operates in a single bidirectional I/O mode, this pin

does not output data and is set to a high impedance state.

MSB

I7

LSB

I0

I6

I5

I4

I3

I2

I1

R/W

N1

N0

A4

A3

A2

A1

A0

MSB/LSB TRANSFERS

A4, A3, A2, A1, A0

The AD9777 serial port can support both most significant bit

(MSB) first or least significant bit (LSB) first data formats. This

functionality is controlled by the LSB first bit in Register 0. The

default is MSB first.

Bit 4, Bit 3, Bit 2, Bit 1, and Bit 0 of the instruction byte

determine which register is accessed during the data transfer

portion of the communications cycle. For multibyte transfers,

this address is the starting byte address. The remaining register

addresses are generated by the AD9777.

When this bit is set active high, the AD9777 serial port is in LSB

first format. In LSB first mode, the instruction byte and data

bytes must be written from LSB to MSB. In LSB first mode, the

serial port internal byte address generator increments for each

byte of the multibyte communication cycle.

SERIAL INTERFACE PORT PIN DESCRIPTIONS

SPI_CLK (Pin 55)—Serial Clock

The serial clock pin is used to synchronize data to and from the

AD9777 and to run the internal state machines. SPI_CLK

maximum frequency is 15 MHz. All data input to the AD9777

is registered on the rising edge of SPI_CLK. All data is driven

out of the AD9777 on the falling edge of SCLK.

When this bit is set default low, the AD9777 serial port is in

MSB first format. In MSB first mode, the instruction byte and

data bytes must be written from MSB to LSB. In MSB first

mode, the serial port internal byte address generator decre-

ments for each byte of the multibyte communication cycle.

When incrementing from 1Fh, the address generator changes to

00h. When decrementing from 00h, the address generator

changes to 1Fh.

Rev. C | Page 23 of 60

AD9777

INSTRUCTION CYCLE

DATA TRANSFER CYCLE

CS

SCLK

SDIO

R/W

I6

I5

I4

I3

I2

I1

I0

D7

D6

D2

D1

D0

(N)

N

0

SDO

0

N

Figure 33. Serial Register Interface Timing MSB First

INSTRUCTION CYCLE

DATA TRANSFER CYCLE

CS

SCLK

SDIO

I0

I1

I2

I3

I4

I5

I6

R/W

D0

D1

D2

D6

D7

(N)

0

N

SDO

D2

D6

D7

Figure 34. Serial Register Interface Timing LSB First

t_SCLK

t_DS

CS

t_PWH

t_PWL

SCLK

SDIO

t_DS

t_DH

INSTRUCTION BIT 7

INSTRUCTION BIT 6

Figure 35. Timing Diagram for Register Write to AD9777

CS

SCLK

t_DV

SDIO

SDO

DATA BIT N

DATA BIT N–1

Figure 36. Timing Diagram for Register Read from AD9777

Rev. C | Page 24 of 60

AD9777

The offset control defines a small current that can be added to

I_OUTAor I_OUTB(not both) on the IDAC and QDAC. The selection

of which I_OUTthis offset current is directed toward is program-

mable via Register 08h, Bit 7 (IDAC) and Register 0Ch, Bit 7

(QDAC). Figure 42 shows the scale of the offset current that can

be added to one of the complementary outputs on the IDAC

and QDAC. Offset control can be used for suppression of LO

leakage resulting from modulation of dc signal components. If

the AD9777 is dc-coupled to an external modulator, this feature

can be used to cancel the output offset on the AD9777 as well as

the input offset on the modulator. Figure 42 shows a typical

example of the effect that the offset control has on LO

suppression.

NOTES ON SERIAL PORT OPERATION

The AD9777 serial port configuration bits reside in Bit 6 and

Bit 7 of Register Address 00h. It is important to note that the

configuration changes immediately upon writing to the last bit

of the register. For multibyte transfers, writing to this register

may occur during the middle of the communication cycle. Care

must be taken to compensate for this new configuration for the

remaining bytes of the current communication cycle.

The same considerations apply to setting the reset bit in

Register Address 00h. All other registers are set to their default

values, but the software reset does not affect the bits in Register

Address 00h.

OFFSET

CONTROL

REGISTERS

It is recommended to use only single byte transfers when

changing serial port configurations or initiating a software

reset.

OFFSET

DAC

FINE

GAIN

DAC

GAIN

CONTROL

REGISTERS

FINE

GAIN

DAC

I

IDAC

OUTA1

A write to Bit 1, Bit 2, and Bit 3 of Address 00h with the same

logic levels as for Bit 7, Bit 6, and Bit 5 (bit pattern: XY1001YX

binary) allows the user to reprogram a lost serial port configu-

ration and to reset the registers to their default values. A second

write to Address 00h with reset bit low and serial port configu-

ration as specified above (XY) reprograms the OSC IN

1.2VREF

REFIO

0.1µF

OUTB1

COARSE COARSE

QDAC

I

OUTA2

GAIN

DAC

GAIN

DAC

OUTB2

FSADJ1

RSET1

OFFSET

CONTROL

REGISTERS

FSADJ2

OFFSET

DAC

multiplier setting. A changed f_SYSCLKfrequency is stable after a

maximum of 200 f_MCLKcycles (equals wake-up time).

GAIN

CONTROL

RSET2

REGISTERS

Figure 37. DAC Outputs, Reference Current Scaling, and Gain/Offset Adjust

DAC OPERATION

The dual 16-bit DAC output of the AD9777, along with the

reference circuitry, gain, and offset registers, is shown in Figure

37 and Figure 38. Note that an external reference can be used by

simply overdriving the internal reference with the external

reference. Referring to the transfer functions in Equation 1, a

reference current is set by the internal 1.2 V reference, the

external R_SETresistor, and the values in the coarse gain register.

The fine gain DAC subtracts a small amount from this and the

result is input to IDAC and QDAC, where it is scaled by an

amount equal to 1024/24. Figure 39 and Figure 40 show the

scaling effect of the coarse and fine adjust DACs. IDAC and

QDAC are PMOS current source arrays, segmented in a 5-4-7

configuration. The five MSB control an array of 31 current

sources. The next four bits consist of 15 current sources whose

values are all equal to 1/16 of an MSB current source. The 7

LSBs are binary weighted fractions of the middle bits’ current

AVDD

84µA

REFIO

7kΩ

0.7V

Figure 38. Internal Reference Equivalent Circuit

25

20

2R MODE

15

sources. All current sources are switched to either I_OUTAor I_OUTB

,

depending on the input code.

10

The fine adjustment of the gain of each channel allows for im-

proved balance of QAM modulated signals, resulting in improved

modulation accuracy and image rejection. In the Interfacing with

the AD8345 Quadrature Modulator section, the performance

data shows to what degree image rejection can be improved when

the AD9777 is used with an AD8345 quadrature modulator from

Analog Devices, Inc.

1R MODE

5

0

5

10

15

20

COARSE GAIN REGISTER CODE

(ASSUMING RSET1, RSET2 = 1.9kΩ)

Figure 39. Coarse Gain Effect on I_FULLSCALE

Rev. C | Page 25 of 60

AD9777

0

–0.5

–1.0

–1.5

–2.0

–2.5

0

–10

–20

–30

–40

–50

–60

–70

–80

OFFSET REGISTER 1 ADJUSTED

1R MODE

2R MODE

OFFSET REGISTER 2

ADJUSTED, WITH OFFSET

REGISTER 1 SET

TO OPTIMIZED VALUE

–3.0

0

200

400

600

800

1000

–1024 –768

–512

–256

0

256

512

768

1024

FINE GAIN REGISTER CODE

(ASSUMING RSET1, RSET2 = 1.9k

DAC1, DAC2 (OFFSET REGISTER CODES)

Ω

)

Figure 42. Offset Adjust Control, Effect on LO Suppression

Figure 40. Fine Gain Effect on I_FULLSCALE

1R/2R MODE

In Figure 42, the negative scale represents an offset added to

_OUTB, while the positive scale represents an offset added to I_OUTA

I

In 2R mode, the reference current for each channel is set

independently by the FSADJ resistor on that channel. The

AD9777 can be programmed to derive its reference current

from a single resistor on Pin 60 by putting the part into 1R

mode. The transfer functions in Equation 1 are valid for 2R

mode. In 1R mode, the current developed in the single FSADJ

resistor is split equally between the two channels. The result is

that in 1R mode, a scale factor of 1/2 must be applied to the

formulas in Equation 1. The full-scale DAC current in 1R mode

can still be set to as high as 20 mA by using the internal 1.2 V

reference and a 950 Ω resistor instead of the 1.9 kΩ resistor

typically used in 2R mode.

of the respective DAC. Offset Register 1 corresponds to IDAC,

while Offset Register 2 corresponds to QDAC. Figure 42

represents the AD9777 synthesizing a complex signal that is

then dc-coupled to an AD8345 quadrature modulator with an

LO of 800 MHz. The dc coupling allows the input offset of the

AD8345 to be calibrated out as well. The LO suppression at the

AD8345 output was optimized first by adjusting Offset

Register 1 in the AD9777. When an optimal point was found

(roughly Code 54), this code was held in Offset Register 1, and

Offset Register 2 was adjusted. The resulting LO suppression is

70 dBFS. These are typical numbers, and the specific code for

optimization varies from part to part.

CLOCK INPUT CONFIGURATION

5

4

3

The clock inputs to the AD9777 can be driven differentially or

single-ended. The internal clock circuitry has supply and

ground (CLKVDD, CLKGND) separate from the other supplies

on the chip to minimize jitter from internal noise sources.

Figure 43 shows the AD9777 driven from a single-ended clock

source. The CLK+/CLK− pins form a differential input

(CLKIN) so that the statically terminated input must be dc-

biased to the midswing voltage level of the clock driven input.

2R MODE

2

1R MODE

1

AD9777

R

SERIES

0

CLK+

0

200

400

600

800

1000

COARSE GAIN REGISTER CODE

(ASSUMING RSET1, RSET2 = 1.9kΩ)

CLKVDD

CLK–

Figure 41. DAC Output Offset Current

V

THRESHOLD

0.1µF

CLKGND

Figure 43. Single-Ended Clock Driving Clock Inputs

Rev. C | Page 26 of 60

AD9777

A configuration for differentially driving the clock inputs is

given in Figure 44. DC-blocking capacitors can be used to

couple a clock driver output whose voltage swings exceed

CLKVDD or CLKGND. If the driver voltage swings are within

the supply range of the AD9777, the dc-blocking capacitors and

bias resistors are not necessary.

PROGRAMMABLE PLL

CLKIN can function either as an input data rate clock (PLL

enabled) or as a DAC data rate clock (PLL disabled) according

to the state of Address 02h, Bit 7 in the SPI port register. The

internal operation of the AD9777 clock circuitry in these two

modes is illustrated in Figure 45 and Figure 46.

AD9777

The PLL clock multiplier and distribution circuitry produce the

necessary internal synchronized 1×, 2×, 4×, and 8× clocks for

the rising edge triggered latches, interpolation filters,

modulators, and DACs. This circuitry consists of a phase

detector, charge pump, voltage controlled oscillator (VCO),

prescaler, clock distribution, and SPI port control. The charge

pump, VCO, differential clock input buffer, phase detector,

prescaler, and clock distribution are all powered from

CLKVDD. PLL lock status is indicated by the logic signal at the

PLL_LOCK pin, as well as by the status of Bit 1, Register 00h.

To ensure optimum phase noise performance from the PLL

clock multiplier and distribution, CLKVDD should originate

from a clean analog supply. Table 10 defines the minimum

input data rates versus the interpolation and PLL divider

setting. If the input data rate drops below the defined minimum

under these conditions, VCO phase noise can increase

significantly. The VCO speed is a function of the input data

rate, the interpolation rate, and the VCO prescaler, according to

the following function:

1kΩ

0.1µF

CLK+

1kΩ

0.1µF

ECL/PECL

CLKVDD

CLK–

1kΩ

CLKGND

Figure 44. Differential Clock Driving Clock Inputs

A transformer, such as the T1-1T from Mini-Circuits, can also

be used to convert a single-ended clock to differential. This

method is used on the AD9777 evaluation board so that an

external sine wave with no dc offset can be used as a differential

clock.

PECL/ECL drivers require varying termination networks, the

details of which are left out of Figure 43 and Figure 44 but can

be found in application notes such as AND8020/D from On

Semiconductor. These networks depend on the assumed

transmission line impedance and power supply voltage of the

clock driver. Optimum performance of the AD9777 is achieved

when the driver is placed very close to the AD9777 clock inputs,

thereby negating any transmission line effects such as reflec-

tions due to mismatch.

VCO Speed (MHz) =

Input Data Rate (MHz)× Interpolation Rate × Prescaler

CLK+ CLK–

PLLVDD

PLL_LOCK

1 = LOCK

0 = NO LOCK

AD9777

The quality of the clock and data input signals is important in

achieving optimum performance. The external clock driver

circuitry should provide the AD9777 with a low jitter clock

input that meets the minimum/maximum logic levels while

providing fast edges. Although fast clock edges help minimize

any jitter that manifests itself as phase noise on a reconstructed

waveform, the high gain bandwidth product of the AD9777’s

clock input comparator can tolerate differential sine wave inputs

as low as 0.5 V p-p with minimal degradation of the output

noise floor.

INTERPOLATION

FILTERS,

MODULATORS,

AND DACS

PHASE

DETECTOR

CHARGE

PUMP

LPF

2

4

8

1

CLOCK

DISTRIBUTION

CIRCUITRY

PRESCALER

VCO

INPUT

DATA

LATCHES

PLL DIVIDER

(PRESCALER)

CONTROL

INTERNAL SPI

CONTROL

REGISTERS

INTERPOLATION

RATE

PLL

CONTROL

(PLL ON)

MODULATION

CONTROL

RATE

SPI PORT

CONTROL

Figure 45. PLL and Clock Circuitry with PLL Enabled

Rev. C | Page 27 of 60

AD9777

Table 11. Required PLL Prescaler Ration vs. f_DATA

CLK+ CLK–

f_DATA(MSPS)

PLL

Prescaler Ratio

PLL_LOCK

1 = LOCK

0 = NO LOCK

125

100

75

Disabled

Enabled

AD9777

Div 1

Div 2

Div 4

INTERPOLATION

FILTERS,

MODULATORS,

AND DACS

PHASE

DETECTOR

CHARGE

PUMP

50

2

4

8

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

1

CLOCK

DISTRIBUTION

CIRCUITRY

PRESCALER

VCO

INPUT

DATA

LATCHES

PLL DIVIDER

(PRESCALER)

CONTROL

INTERNAL SPI

CONTROL

REGISTERS

INTERPOLATION

RATE

PLL

CONTROL

(PLL ON)

MODULATION

RATE

CONTROL

SPI PORT

Figure 46. PLL and Clock Circuitry with PLL Disabled

In addition, if the zero stuffing option is enabled, the VCO doubles its

speed again. Phase noise can be slightly higher with the PLL enabled.

Figure 47 illustrates typical phase noise performance of the AD9777

with 2× interpolation and various input data rates. The signal

synthesized for the phase noise measurement was a single carrier at a

frequency of f_DATA/4. The repetitive nature of this signal eliminates

quantization noise and distortion spurs as a factor in the measure-

ment. Although the curves blend in Figure 47, the different

conditions are given for clarity in the table preceding Figure 47.

Figure 47 also contains a table detailing PLL divider settings vs.

interpolation rate and maximum and minimum f_DATArates. Note that

maximum f_DATArates of 160 MSPS are due to the maximum input

data rate of the AD9777. However, maximum rates of less than 160

MSPS and all minimum f_DATArates are due to maximum and mini-

mum speeds of the internal PLL VCO. Figure 48 shows typical

performance of the PLL lock signal (Pin 8 or Pin 53) when the PLL is

in the process of locking.

–110

0

1

2

3

4

5

FREQUENCY OFFSET (MHz)

Figure 47. Phase Noise Performance

Table 10. PLL Optimization

Interpolation

Rate

Divider

Setting

Minimum

f_DATA

Maximum

f_DATA

Figure 48. PLL_LOCK Output Signal (Pin 8) in the Process of Locking

(Typical Lock Time)

1

2

4

8

1

2

4

8

1

2

4

8

1

2

4

8

1

2

4

8

32

16

8

160

112

56

160

112

56

28

100

56

28

14

50

28

14

7

It is important to note that the resistor/capacitor needed for the

PLL loop filter is internal on the AD9777. This suffices unless

the input data rate is below 10 MHz, in which case an external

series RC is required between the LPF and CLKVDD pins.

4

24

12

6

3

24

12

6

3

24

12

6

3

Rev. C | Page 28 of 60

AD9777

35

30

25

20

15

10

5

POWER DISSIPATION

8×

The AD9777 has three voltage supplies: DVDD, AVDD, and

CLKVDD. Figure 49, Figure 50, and Figure 51 show the current

required from each of these supplies when each is set to the

3.3 V nominal specified for the AD9777. Power dissipation (P_D)

can easily be extracted by multiplying the given curves by 3.3.

As Figure 49 shows, I_DVDDis very dependent on the input data

rate, the interpolation rate, and the activation of the internal

digital modulator. I_DVDD, however, is relatively insensitive to the

modulation rate by itself. In Figure 50, I_AVDDshows the same

type of sensitivity to data, interpolation rate, and the modulator

function but to a much lesser degree (<10ꢀ). In Figure 51,

I_CLKVDDvaries over a wide range yet is responsible for only a

small percentage of the overall AD9777 supply current

requirement.

4×

2×

1×

0

50

100

150

200

f_DATA(MHz)

Figure 51. I_CLKVDDvs. f_DATAvs. Interpolation Rate, PLL Disabled

400

SLEEP/POWER-DOWN MODES

8×, (MOD. ON)

2×, (MOD. ON)

350

300

250

200

150

100

50

(Control Register 00h, Bit 3 and Bit 4)

4×, (MOD. ON)

The AD9777 provides two methods for programmable

reduction in power savings. The sleep mode, when activated,

turns off the DAC output currents but the rest of the chip

remains functioning. When coming out of sleep mode, the

AD9777 immediately returns to full operation. Power-down

mode, on the other hand, turns off all analog and digital

circuitry in the AD9777 except for the SPI port. When

returning from power-down mode, enough clock cycles must

be allowed to flush the digital filters of random data acquired

during the power-down cycle. Note that optimal performance

with the PLL enabled is achieved with the UCO in the PLL

control loop running at 450 MHz to 550 MHz.

8×

4×

2×

1×

0

50

100

150

200

f_DATA(MHz)

Figure 49. I_DVDDvs. f_DATAvs. Interpolation Rate, PLL Disabled

TWO PORT DATA INPUT MODE

76.0

75.5

75.0

74.5

74.0

73.5

73.0

72.5

72.0

4×, (MOD. ON)

8×, (MOD. ON)

The digital data input ports can be configured as two

independent ports or as a single (one-port mode) port. In the

two-port mode, data at the two input ports is latched into the

AD9777 on every rising edge of the data rate clock (DATACLK).

In addition, in the two-port mode, the AD9777 can be

programmed to generate an externally available DATACLK for

the purpose of data synchronization. This data rate clock can be

programmed to be available at either Pin 8 (DATACLK/

PLL_LOCK) or Pin 53 (SPI_SDO). Because Pin 8 can also

function as a PLL lock indicator when the PLL is enabled, there

are several options for configuring Pin 8 and Pin 53. The

following information describes these options.

2×, (MOD. ON)

4×

8×

2×

1×

0

50

100

f_DATA(MHz)

150

200

PLL Off (Register 4, Bit 7 = 0)

Figure 50. I_AVDDvs. f_DATAvs. Interpolation Rate, PLL Disabled

Register 3, Bit 7 = 0; DATACLK out of Pin 8.

Register 3, Bit 7 = 1; DATACLK out of Pin 53.

PLL On (Register 4, Bit 7 = 1)

Register 3, Bit 7 = 0, Register 1, Bit 0 = 0; PLL lock indicator out

of Pin 8.

Rev. C | Page 29 of 60

AD9777

t_OD

Register 3, Bit 7 = 1, Register 1, Bit 0 = 0; PLL lock indicator out

of Pin 53.

Register 3, Bit 7 = 0, Register 1, Bit 0 = 1; DATACLK out of Pin 8.

Register 3, Bit 7 = 1, Register 1, Bit 0 = 1; DATACLK out of Pin 53.

CLKIN

In one-port mode, P2B14 and P2B15 from input data port two

are redefined as IQSEL and ONEPORTCLK, respectively. The

input data in one-port mode is steered to one of the two inter-

nal data channels based on the logic level of IQSEL. A clock

signal, ONEPORTCLK, is generated by the AD9777 in this

mode for the purpose of data synchronization. ONEPORTCLK

runs at the input interleaved data rate, which is 2× the data rate

at the internal input to either channel.

DATACLK

DATA AT PORTS

1 AND 2

t_S= 0.0ns (MAX)

t_H= 2.5ns (MAX)

Test configurations showing the various clocks that are required

and generated by the AD9777 with the PLL enabled/disabled

and in the one-port/two-port modes are given in Figure 101 to

Figure 104. Jumper positions needed to operate the AD9777

evaluation board in these modes are given as well.

t_St_H

Figure 52. Timing Requirements in Two-Port Input Mode with PLL Enabled

DATACLK DRIVER STRENGTH

(Control Register 02h, Bit 5)

PLL ENABLED, TWO-PORT MODE

The DATACLK output driver strength is capable of driving

>10 mA into a 330 Ω load while providing a rise time of 3 ns.

Figure 53 shows DATACLK driving a 330 Ω resistive load at a

frequency of 50 MHz. By enabling the drive strength option

(Control Register 02h, Bit 5), the amplitude of DATACLK under

these conditions increases by approximately 200 mV.

(Control Register 02h, Bits 6 to 0 and 04h, Bits 7 to 1)

With the phase-locked loop (PLL) enabled and the AD9777 in

two-port mode, the speed of CLKIN is inherently that of the

input data rate. In two-port mode, Pin 8 (DATACLK/PLL_

LOCK) can be programmed (Control Register 01h, Bit 0) to

function as either a lock indicator for the internal PLL or as a

clock running at the input data rate. When Pin 8 is used as a

clock output (DATACLK), its frequency is equal to that of

CLKIN. Data at the input ports is latched into the AD9777 on

3.0

2.5

2.0

1.5

1.0

0.5

the rising edge of the CLKIN. Figure 52 shows the delay, t_OD

,

inherent between the rising edge of CLKIN and the rising edge

of DATACLK, as well as the setup and hold requirements for

the data at Ports 1 and 2. The setup and hold times given in Figure

52 are the input data transitions with respect to CLKIN. Note

that in two-port mode (PLL enabled or disabled), the data rate

at the interpolation filter inputs is the same as the input data

rate at Port 1 and Port 2.

0

DELTA APPROX. 2.8ns

–0.5

The DAC output sample rate in two-port mode is equal to the

clock input rate multiplied by the interpolation rate. If zero

stuffing is used, another factor of 2 must be included to

calculate the DAC sample rate.

0

10

20

30

40

50

TIME (ns)

Figure 53. DATACLK Driver Capability into 330 Ω at 50 MHz

PLL ENABLED, ONE-PORT MODE

DATACLK INVERSION

(Control Register 02h, Bits 6 to 1 and 04h, Bits 7 to 1)

(Control Register 02h, Bit 4)

In one-port mode, the I and Q channels receive their data from

an interleaved stream at digital input Port 1. The function of

Pin 32 is defined as an output (ONEPORTCLK) that generates a

clock at the interleaved data rate, which is 2× the internal input

data rate of the I and Q channels. The frequency of CLKIN is

equal to the internal input data rate of the I and Q channels.

By programming this bit, the DATACLK signal shown in Figure 53

can be inverted. With inversion enabled, t_ODrefers to the time

between the rising edge of CLKIN and the falling edge of

DATACLK. No other effect on timing occurs.

Rev. C | Page 30 of 60

AD9777

IQ PAIRING

The selection of the data for the I or Q channel is determined by

the state of the logic level at Pin 31 (IQSEL when the AD9777 is

in one-port mode) on the rising edge of ONEPORTCLK.

(Control Register 02h, Bit 0)

In one-port mode, the interleaved data is latched into the

AD9777 internal I and Q channels in pairs. The order of how

the pairs are latched internally is defined by this control register.

The following is an example of the effect this has on incoming

interleaved data.

Under these conditions, IQSEL = 0 latches the data into the I

channel on the clock rising edge, while IQSEL = 1 latches the

data into the Q channel. It is possible to invert the I and Q

selection by setting Control Register 02h, Bit 1 to the invert

state (Logic 1). Figure 54 illustrates the timing requirements for

the data inputs as well as the IQSEL input. Note that the 1×

interpolation rate is not available in one-port mode.

Given the following interleaved data stream, where the data

indicates the value with respect to full scale:

I

Q

0.5

I

1

Q

1

I

Q

0.5

I

0

Q

0

I

Q

0.5

The DAC output sample rate in one-port mode is equal to

CLKIN multiplied by the interpolation rate. If zero stuffing is

used, another factor of 2 must be included to calculate the DAC

sample rate.

0.5

With the control register set to 0 (I first), the data appears at the

internal channel inputs in the following order in time:

I Channel

Q Channel

0.5

1

0.5

0

0.5

ONEPORTCLK INVERSION

(Control Register 02h, Bit 2)

With the control register set to 1 (Q first), the data appears at

the internal channel inputs in the following order in time:

By programming this bit, the ONEPORTCLK signal shown in

Figure 54 can be inverted. With inversion enabled, t_ODrefers to

the delay between the rising edge of the external clock and the

falling edge of ONEPORTCLK. The setup and hold times, t_S

and t_H, are with respect to the falling edge of ONEPORTCLK.

There is no other effect on timing.

I Channel

Q Channel

0.5

y

1

0.5

1

0

0.5

0

x

0.5

The values x and y represent the next I value and the previous Q

value in the series.

PLL DISABLED, TWO-PORT MODE

ONEPORTCLK DRIVER STRENGTH

With the PLL disabled, a clock at the DAC output rate must be

applied to CLKIN. Internal clock dividers in the AD9777 syn-

thesize the DATACLK signal at Pin 8, which runs at the input

data rate and can be used to synchronize the input data. Data is

latched into input Port 1 and Port 2 of the AD9777 on the rising

edge of DATACLK. DATACLK speed is defined as the speed of

CLKIN divided by the interpolation rate. With zero stuffing en-

abled, this division increases by a factor of 2. Figure 55 illustrates

the delay between the rising edge of CLKIN and the rising edge

of DATACLK, as well as t_Sand t_Hin this mode.

The drive capability of ONEPORTCLK is identical to that of

DATACLK in the two-port mode. Refer to Figure 53 for

performance under load conditions.

t_OD

t_OD= 4.0ns (MIN)

TO 5.5ns (MAX)

CLKIN

t_S= 3.0ns (MAX)

t_H= –0.5ns (MAX)

t_IQS= 3.5ns (MAX)

t_IQH= –1.5ns (MAX

ONEPORTCLK

The programmable modes DATACLK inversion and DATACLK

driver strength described in the PLL Enabled, Two-Port Mode

section have identical functionality with the PLL disabled.

The data rate CLK created by dividing down the DAC clock in

this mode can be programmed (via Register x03h, Bit 7) to be

output from the SPI_SDO pin, rather than the DATACLK pin.

In some applications, this may improve complex image rejec-

tion. t_ODincreases by 1.6 ns when SPI_SDO is used as data rate

clock out.

I AND Q INTERLEAVED

INPUT DATA AT PORT 1

t_St_H

IQSEL

t_IQS

t_IQH

Figure 54. Timing Requirements in One-Port

Input Mode, with the PLL Enabled

Rev. C | Page 31 of 60

AD9777

t_OD

CLKIN

DATACLK

ONEPORTCLK

DATA AT PORTS

1 AND 2

I AND Q INTERLEAVED

INPUT DATA AT PORT 1

t_OD= 6.5ns (MIN) TO 8.0ns (MAX)

t_S= 5.0ns (MAX)

t_H= –3.2ns (MAX)

t_S

t_H

Figure 55. Timing Requirements in Two-Port Input Mode, with PLL Disabled

t_St_H

PLL DISABLED, ONE-PORT MODE

IQSEL

t_OD= 4.0ns (MIN)

TO 5.5ns (MAX)

t_OD= 4.7ns (MAX)

t_S= 3.0ns (MAX)

t_H= –1.0ns (MAX)

t_IQS= 3.5ns (MAX)

t_IQH= –1.5ns (MAX)

(TYP SPECS)

In one-port mode, data is received into the AD9777 as an

interleaved stream on Port 1. A clock signal (ONEPORTCLK),

running at the interleaved data rate, which is 2× the input data

rate of the internal I and Q channels, is available for data

synchronization at Pin 32.

t_IQS

t_IQH

Figure 56. Timing Requirements in One-Port Input Mode, with PLL Disabled

With PLL disabled, a clock at the DAC output rate must be applied

to CLKIN. Internal dividers synthesize the ONEPORTCLK signal

at Pin 32. The selection of the data for the I or Q channel is deter-

mined by the state of the logic level applied to Pin 31 (IQSEL when

the AD9777 is in one-port mode) on the rising edge of

ONEPORTCLK. Under these conditions, IQSEL = 0 latches the

data into the I channel on the clock rising edge, while IQSEL = 1

latches the data into the Q channel. It is possible to invert the I and

Q selection by setting Control Register 02h, Bit 1 to the invert state

(Logic 1). Figure 56 illustrates the timing requirements for the data

inputs as well as the IQSEL input. Note that the 1× interpolation

rate is not available in the one-port mode.

DIGITAL FILTER MODES

The I and Q data paths of the AD9777 have their own

independent half-band FIR filters. Each data path consists of

three FIR filters, providing up to 8× interpolation for each

channel. The rate of interpolation is determined by the state of

Control Register 01h, Bit 7 and Bit 6. Figure 2 to Figure 4 show

the response of the digital filters when the AD9777 is set to 2×,

4×, and 8× modes. The frequency axes of these graphs have

been normalized to the input data rate of the DAC. As the

graphs show, the digital filters can provide greater than 75 dB of

out-of-band rejection.

An online tool is available for quick and easy analysis of the

AD9777 interpolation filters in the various modes. The link can be

accessed at http://www.analog.com/Analog_Root/static/

techsupport/designtools/interactiveTools/dac/ad9777image.html.

One-port mode is very useful when interfacing with devices,

such as the Analog Devices AD6622 or AD6623 transmit signal

processors, in which two digital data channels have been inter-

leaved (multiplexed). The programmable modes’ ONEPORTCLK

inversion, ONEPORTCLK driver strength, and IQ pairing

described in the PLL Enabled, One-Port Mode section have

identical functionality with the PLL disable.

AMPLITUDE MODULATION

Given two sine waves at the same frequency but with a 90°

phase difference, a point of view in time can be taken such that

the waveform that leads in phase is cosinusoidal and the

waveform that lags is sinusoidal. Analysis of complex variables

states that the cosine waveform can be defined as having real

positive and negative frequency components, while the sine

waveform consists of imaginary positive and negative frequency

images. This is shown graphically in the frequency domain in

Figure 57.

Rev. C | Page 32 of 60

AD9777

–jωt

e /2j

The phase relationship of the modulated signals is dependent

on whether the modulating carrier is sinusoidal or cosinusoidal,

again with respect to the reference point of the viewer.

SINE

DC

–jωt

e

/2j

Examples of sine and cosine modulation are given in Figure 58.

–jωt

Ae

/2j

–jωt

/2

–jωt

e

/2

SINUSOIDAL

MODULATION

COSINE

DC

–jωt

Figure 57. Real and Imaginary Components of

Sinusoidal and Cosinusoidal Waveforms

Ae

/2j

/2

–jωt

Ae

/2

COSINUSOIDAL

MODULATION

Amplitude modulating a baseband signal with a sine or a cosine

convolves the baseband signal with the modulating carrier in

the frequency domain. Amplitude scaling of the modulated

signal reduces the positive and negative frequency images by a

factor of 2. This scaling is very important in the discussion of

the various modulation modes.

DC

Figure 58. Baseband Signal, Amplitude Modulated with Sine and Cosine Carriers

Rev. C | Page 33 of 60

AD9777

By comparing the digital domain spectrum to the DAC

SIN(x)/x roll-off, an estimate can be made for the characteris-

tics required for the DAC reconstruction filter. Note also, per

the previous discussion on amplitude modulation, that the

spectral components (where modulation is set to f_S/4 or f_S/8) are

scaled by a factor of 2. In the situation where the modulation is

f_S/2, the modulated spectral components add constructively,

and there is no scaling effect.

MODULATION, NO INTERPOLATION

With Control Register 01h, Bit 7 and Bit 6 set to 00, the

interpolation function on the AD9777 is disabled. Figure 59 to

Figure 62 show the DAC output spectral characteristics of the

AD9777 in the various modulation modes, all with the

interpolation filters disabled. The modulation frequency is

determined by the state of Control Register 01h, Bits 5 and 4.

The tall rectangles represent the digital domain spectrum of a

baseband signal of narrow bandwidth.

The Effects of Digital Modulation on DAC Output Spectrum, Interpolation Disabled

0

–20

–40

–60

–80

–40

–60

–80

–100

0

0.2

0.4

0.6

0.8

1.0

0

0.2

0.4

0.6

0.8

1.0

f_OUT

(×

f_DATA

)

f_OUT

(×f_DATA)

Figure 59. No Interpolation, Modulation Disabled

Figure 61. No Interpolation, Modulation = f_DAC/4

0

–20

–40

0

–20

–40

–60

–80

–100

0

0.2

0.4

0.6

0.8

1.0

0

0.2

0.4

0.6

0.8

1.0

f_OUT

(×f_DATA)

f_OUT

(×f_DATA)

Figure 60. No Interpolation, Modulation = f_DAC/2

Figure 62. No Interpolation, Modulation = f_DAC/8

Rev. C | Page 34 of 60

AD9777

For this reason, as Figure 63 to Figure 66 show, the pass band of

the interpolation filters can be frequency shifted, giving the

equivalent of a high-pass digital filter.

MODULATION, INTERPOLATION = 2×

With Control Register 01h, Bit 7 and Bit 6 set to 01, the

interpolation rate of the AD9777 is 2×. Modulation is achieved

by multiplying successive samples at the interpolation filter

output by the sequence (+1, −1). Figure 63 to Figure 66

represent the spectral response of the AD9777 DAC output with

2× interpolation in the various modulation modes to a narrow

band baseband signal (again, the tall rectangles in the graphic).

The advantage of interpolation becomes clear in Figure 63 to

Figure 66, where it can be seen that the images that would

normally appear in the spectrum around the significant point is

that the interpolation filtering is done prior to the digital

modulator.

Note that when using the f_S/4 modulation mode, there is no

true stop band as the band edges coincide with each other. In

the f_S/8 modulation mode, amplitude scaling occurs over only a

portion of the digital filter pass band due to constructive

addition over just that section of the band

The Effects of Digital Modulation on DAC Output Spectrum, Interpolation = 2x

0

–20

–40

–60

–80

–40

–60

–80

–100

0

0.5

1.0

1.5

2.0

0

0.5

1.0

1.5

2.0

f_OUT

(×

f_DATA

)

f_OUT

(×f_DATA)

Figure 63. 2× Interpolation, Modulation = Disabled

Figure 65. 2× Interpolation, Modulation = f_DAC/4

0

–20

–40

–60

–80

–40

–60

–80

–100

0

0.5

1.0

1.5

2.0

0

0.5

1.0

1.5

2.0

f_OUT

(×f_DATA)

f_OUT

(×f_DATA)

Figure 64. 2× Interpolation, Modulation = f_DAC/2

Figure 66. 2× Interpolation, Modulation = f_DAC/8

Rev. C | Page 35 of 60

AD9777

Figure 67 to Figure 70 represent the spectral response of the

AD9777 DAC output with 4× interpolation in the various

modulation modes to a narrow band baseband signal.

MODULATION, INTERMODULATION = 4×

With Control Register 01h, Bit 7 and Bit 6 set to 10, the

interpolation rate of the AD9777 is 4×. Modulation is achieved

by multiplying successive samples at the interpolation filter

output by the sequence (0, +1, 0, −1).

The Effects of Digital Modulation on DAC Output Spectrum, Interpolation = 4x

0

–20

–40

–60

–80

–40

–60

–80

–100

0

1

2

3

4

0

1

2

3

4

f_OUT

(×

f_DATA

)

f_OUT

(×f_DATA)

Figure 67. 4x Interpolation, Modulation Disabled

Figure 69. 4x Interpolation, Modulation = f_DAC/4

0

–20

–40

–60

–80

–40

–60

–80

–100

0

1

2

3

4

0

1

2

3

4

f_OUT

(×

f_DATA

)

f_OUT

(×f_DATA)

Figure 68. 4x Interpolation, Modulation = f_DAC/2

Figure 70. 4x Interpolation, Modulation = f_DAC/8

Rev. C | Page 36 of 60

AD9777

Looking at Figure 59 to Figure 75, the user can see how higher

interpolation rates reduce the complexity of the reconstruction

filter needed at the DAC output. It also becomes apparent that

the ability to modulate by f_S/2, f_S/4, or f_S/8 adds a degree of

flexibility in frequency planning

MODULATION, INTERMODULATION = 8×

With Control Register 01h, Bits 7 and 6, set to 11, the

interpolation rate of the AD9777 is 8×. Modulation is achieved

by multiplying successive samples at the interpolation filter

output by the sequence (0, +0.707, +1, +0.707, 0, –0.707, −1,

+0.707). Figure 71 to Figure 74 represent the spectral response

of the AD9777 DAC output with 8× interpolation in the various

modulation modes to a narrow band baseband signal.

The Effects of the Digital Modulation on the DAC Output Spectrum, Interpolation = 8×

0

–20

–40

–60

–40

–60

–80

–100

0

1

2

3

4

5

6

7

8

0

1

2

3

4

f_OUT(×f_DATA

)

f_OUT

(×f_DATA)

Figure 73. 8x Interpolation, Modulation = f_DAC/4

Figure 71. 8× Interpolation, Modulation Disabled

0

–20

–40

–60

–40

–60

–80

–100

0

1

2

3

4

5

6

7

8

0

1

2

3

4

f_OUT

(×f_DATA)

f_OUT

(×f_DATA)

Figure 74. 8x Interpolation, Modulation = f_DAC/8

Figure 72. 8x Interpolation, Modulation = f_DAC/2

Rev. C | Page 37 of 60

AD9777

The net effect is to increase the DAC output sample rate by a

factor of 2× with the 0 in the SIN(x)/x DAC transfer function

occurring at twice the original frequency. A 6 dB loss in

amplitude at low frequencies is also evident, as can be seen in

Figure 76.

ZERO STUFFING

(Control Register 01h, Bit 3)

As shown in Figure 75, a 0 or null in the output frequency

response of the DAC (after interpolation, modulation, and DAC

reconstruction) occurs at the final DAC sample rate (f_DAC). This

is due to the inherent SIN(x)/x roll-off response in the digital-

to-analog conversion. In applications where the desired fre-

quency content is below f_DAC/2, this may not be a problem. Note

that at f_DAC/2 the loss due to SIN(x)/x is 4 dB. In direct RF appli-

cations, this roll-off may be problematic due to the increased

pass-band amplitude variation as well as the reduced amplitude

of the desired signal.

It is important to realize that the zero stuffing option by itself

does not change the location of the images but rather their

amplitude, pass-band flatness, and relative weighting. For

instance, in the previous example, the pass-band amplitude

flatness of the image at 3× f_DATA/4 is now improved to 0.59 dB

while the signal level has increased slightly from −10.5 dBFS to

–8.1 dBFS.

INTERPOLATING (COMPLEX MIX MODE)

Consider an application where the digital data into the AD9777

represents a baseband signal around f_DAC/4 with a pass band of

f_DAC/10. The reconstructed signal out of the AD9777 would

experience only a 0.75 dB amplitude variation over its pass

band. However, the image of the same signal occurring at 3×

f_DAC/4 suffers from a pass-band flatness variation of 3.93 dB.

This image may be the desired signal in an IF application using

one of the various modulation modes in the AD9777. This roll-

off of image frequencies can be seen in Figure 59 to Figure 74,

where the effect of the interpolation and modulation rate is

apparent as well.

(Control Register 01h, Bit 2)

In the complex mix mode, the two digital modulators on the

AD9777 are coupled to provide a complex modulation function.

In conjunction with an external quadrature modulator, this

complex modulation can be used to realize a transmit image

rejection architecture. The complex modulation function can be

programmed for e^+jωtor e^−jωtto give upper or lower image

rejection. As in the real modulation mode, the modulation

frequency ω can be programmed via the SPI port for f_DAC/2,

f_DAC/4, and f_DAC/8, where f_DACrepresents the DAC output rate.

10

OPERATIONS ON COMPLEX SIGNALS

ZERO STUFFING

ENABLED

Truly complex signals cannot be realized outside of a computer

simulation. However, two data channels, both consisting of real

data, can be defined as the real and imaginary components of a

complex signal. I (real) and Q (imaginary) data paths are often

defined this way. By using the architecture defined in Figure 76,

a system that operates on complex signals can be realized,

giving a complex (real and imaginary) output.

0

–10

–20

ZERO STUFFING

–30

DISABLED

If a complex modulation function (e^+jωt) is desired, the real and

imaginary components of the system correspond to the real and

imaginary components of e^+jωtor cosωt and sinωt. As Figure 77

shows, the complex modulation function can be realized by

applying these components to the structure of the complex

system defined in Figure 76.

–40

–50

0

0.5

1.0

1.5

2.0

f_OUT, NORMALIZED TO f_DATAWITH ZERO STUFFING DISABLED (Hz)

Figure 75. Effect of Zero Stuffing on DAC’s SIN(x)/x Response

To improve upon the pass-band flatness of the desired image,

the zero stuffing mode can be enabled by setting the control

register bit to a Logic 1. This option increases the ratio of

f_DAC/f_DATAby a factor of 2 by doubling the DAC sample rate and

inserting a midscale sample (that is, 1000 0000 0000 0000) after

every data sample originating from the interpolation filter. This

is important as it affects the PLL divider ratio needed to keep

the VCO within its optimum speed range. Note that the zero

stuffing takes place in the digital signal chain at the output of

the digital modulator, before the DAC.

a(t)

c(t) × b(t) + d × b(t)

INPUT

OUTPUT

COMPLEX FILTER

= (c + jd)

IMAGINARY

INPUT

b(t)

OUTPUT

b(t) × a(t) + c × b(t)

Figure 76. Realization of a Complex System

Rev. C | Page 38 of 60

AD9777

INPUT

(REAL)

OUTPUT

(REAL)

OUTPUT

INPUT

(IMAGINARY)

INPUT

(IMAGINARY)

SINωt

90°

COSωt

Figure 78. Quadrature Modulation

OUTPUT

(IMAGINARY)

The entire upconversion from baseband to transmit frequency

is represented graphically in Figure 79. The resulting spectrum

shown in Figure 79 represents the complex data consisting of

the baseband real and imaginary channels, now modulated onto

orthogonal (cosine and negative sine) carriers at the transmit

frequency. It is important to remember that in this application

(two baseband data channels), the image rejection is not

dependent on the data at either of the AD9777 input channels.

–jωt

e

= COSωt + jSINωt

Figure 77. Implementation of a Complex Modulator

COMPLEX MODULATION AND IMAGE REJECTION

OF BASEBAND SIGNALS

In traditional transmit applications, a two-step upconversion is

done in which a baseband signal is modulated by one carrier to

an IF (intermediate frequency) and then modulated a second

time to the transmit frequency. Although this approach has

several benefits, a major drawback is that two images are created

near the transmit frequency. Only one image is needed, the other

being an exact duplicate. Unless the unwanted image is filtered,

typically with analog components, transmit power is wasted and

the usable bandwidth available in the system is reduced.

In fact, image rejection still occurs with either one or both of

the AD9777 input channels active. Note that by changing the

sign of the sinusoidal multiplying term in the complex

modulator, the upper sideband image could have been

suppressed while passing the lower one. This is easily done in

the AD9777 by selecting the e^+jωtbit (Register 01h, Bit 1). In

purely complex terms, Figure 79 represents the two-stage

upconversion from complex baseband to carrier.

A more efficient method of suppressing the unwanted image

can be achieved by using a complex modulator followed by a

quadrature modulator. Figure 78 is a block diagram of a

quadrature modulator. Note that it is in fact the real output half

of a complex modulator. The complete upconversion can

actually be referred to as two complex upconversion stages, the

real output of which becomes the transmitted signal.

Rev. C | Page 39 of 60

AD9777

REAL CHANNEL (OUT)

A/2 A/2

1

–F

F

C

REAL CHANNEL (IN)

A

–B/2J

B/2J

DC

–F

C

F

C

COMPLEX

MODULATOR

TO QUADRATURE

MODULATOR

IMAGINARY CHANNEL (OUT)

–A/2J A/2J

IMAGINARY CHANNEL (IN)

–F

C

B

DC

B/2

–F

F

C

A/4 + B/4J A/4 – B/4J

2

–F

F

Q

–F – F

–F + F

F

– F

F + F

Q C

Q

C

Q

C

Q

C

OUT

REAL

–A/4 – B/4J A/4 – B/4J

A/4 + B/4J –A/4 + B/4J

QUADRATURE

MODULATOR

–F

Q

F

Q

IMAGINARY

REJECTED IMAGES

A/2 + B/2J

A/2 – B/2J

–F

Q

F

Q

1

2

F

= COMPLEX MODULATION FREQUENCY

= QUADRATURE MODULATION FREQUENCY

C

Q

Figure 79. Two-Stage Upconversion and Resulting Image Rejection

Rev. C | Page 40 of 60

AD9777

COMPLEX BASEBAND

SIGNAL

A system in which multiple baseband signals are complex

modulated and then applied to the AD9777 real and imaginary

inputs, followed by a quadrature modulator, is shown in Figure 82,

which also describes the transfer function of this system and the

spectral output. Note the similarity of the transfer functions

given in Figure 82 and Figure 80. Figure 82 adds an additional

complex modulator stage for summing multiple carriers at the

AD9777 inputs. In addition, as in Figure 79, the image rejection

is not dependent on the real or imaginary baseband data on any

channel. Image rejection on a channel occurs if either the real

or imaginary data, or both, is present on the baseband channel.

1

j(ω1 + ω2)t

×

OUTPUT = REAL

1/2

e

1/2

= REAL

–ω1 – ω2

DC

ω1 + ω2

FREQUENCY

Figure 80. Two-Stage Complex Upconversion

IMAGE REJECTION AND SIDEBAND

SUPPRESSIONS OF MODULATED CARRIERS

It is important to remember that the magnitude of a complex

signal can be 1.414× the magnitude of its real or imaginary

components. Due to this 3 dB increase in signal amplitude, the

real and imaginary inputs to the AD9777 must be kept at least

3 dB below full scale when operating with the complex

modulator. Overranging in the complex modulator results in

severe distortion at the DAC output.

As shown in Figure 79, image rejection can be achieved by

applying baseband data to the AD9777 and following the

AD9777 with a quadrature modulator. To process multiple

carriers while still maintaining image reject capability, each

carrier must be complex modulated. As Figure 80 shows, single

or multiple complex modulators can be used to synthesize

complex carriers. These complex carriers are then summed and

applied to the real and imaginary inputs of the AD9777.

BASEBAND CHANNEL 1

R(1)

REAL INPUT

COMPLEX

MULTICARRIER

REAL OUTPUT =

R(1) + R(2) + . . .R(N)

(TO REAL INPUT OF AD9777)

MODULATOR 1

R(1)

IMAGINARY INPUT

BASEBAND CHANNEL 2

REAL INPUT

R(2)

COMPLEX

MODULATOR 2

MULTICARRIER

IMAGINARY OUTPUT =

I(1) + I(2) + . . .I(N)

IMAGINARY INPUT

(TO IMAGINARY INPUT OF AD9777)

R(N) = REAL OUTPUT OF N

I(N) = IMAGINARY OUTPUT OF N

BASEBAND CHANNEL N

REAL INPUT

R(N)

COMPLEX

MODULATOR N

IMAGINARY INPUT

Figure 81. Synthesis of Multicarrier Complex Signal

MULTIPLE

BASEBAND

CHANNELS

REAL

MULTIPLE

COMPLEX

MODULATORS

AD9777

COMPLEX

MODULATOR

FREQUENCY = ω

QUADRATURE

MODULATOR

FREQUENCY = ω

IMAGINARY

Q

FREQUENCY = ω , ω ...ω

1

2

N

C

COMPLEX BASEBAND

SIGNAL

×

OUTPUT = REAL

j(ω + ω + ω )t

e

N

C

Q

–ω – ω – ω

ω

+ ω + ω

Q

DC

REJECTED IMAGES

1

C

Q

1

C

Figure 82. Image Rejection with Multicarrier Signals

Rev. C | Page 41 of 60

AD9777

The complex carrier synthesized in the AD9777 digital

Region C

modulator is accomplished by creating two real digital carriers

in quadrature. Carriers in quadrature cannot be created with

the modulator running at f_DAC/2. As a result, complex modula-

tion only functions with modulation rates of f_DAC/4 and f_DAC/8.

Region C is most accurately described as a down conversion, as

the modulating carrier is −e^jωt. If viewed as a complex signal,

only the images in Region C remains. This image appears on

the real and imaginary outputs of the AD9777, as well as on the

output of the quadrature modulator, where the center of the

spectral plot now represents the quadrature modulator LO and

the horizontal scale represents the frequency offset from this

LO.

Region A and Region B of Figure 83 to Figure 88 are the result

of the complex signal described previously, when complex

modulated in the AD9777 by +e^jωt. Region C and Region D are

the result of the complex signal described previously, again with

positive frequency components only, modulated in the AD9777

by −e^jωt. The analog quadrature modulator after the AD9777

inherently modulates by +e^jωt.

Region D

Region D is the image (complex conjugate) of Region C. If a

spectrum analyzer is used to view the real or imaginary DAC

outputs of the AD9777, Region D appears in the spectrum.

However, on the output of the quadrature modulator, Region D

is rejected.

Region A

Region A is a direct result of the upconversion of the complex

signal near baseband. If viewed as a complex signal, only the

images in Region A remains. The complex Signal A, consisting

of positive frequency components only in the digital domain,

has images in the positive odd Nyquist zones (1, 3, 5, and so

on), as well as images in the negative even Nyquist zones. The

appearance and rejection of images in every other Nyquist zone

becomes more apparent at the output of the quadrature

modulator. The A images appear on the real and the imaginary

outputs of the AD9777, as well as on the output of the

Figure 89 to Figure 96 show the measured response of the

AD9777 and AD8345 given the complex input signal to the

AD9777 in Figure 89. The data in these graphs was taken with a

data rate of 12.5 MSPS at the AD9777 inputs. The interpolation

rate of 4× or 8× gives a DAC output data rate of 50 MSPS or

100 MSPS. As a result, the high end of the DAC output

spectrum in these graphs is the first null point for the SIN(x)/x

roll-off, and the asymmetry of the DAC output images is

representative of the SIN(x)/x roll-off over the spectrum. The

internal PLL was enabled for these results. In addition, a

35 MHz third-order low-pass filter was used at the AD9777/

AD8345 interface to suppress DAC images.

quadrature modulator, where the center of the spectral plot now

represents the quadrature modulator LO and the horizontal scale

now represents the frequency offset from this LO.

Region B

Region B is the image (complex conjugate) of Region A. If a

spectrum analyzer is used to view the real or imaginary DAC

outputs of the AD9777, Region B appears in the spectrum.

However, on the output of the quadrature modulator, Region B

is rejected.

An important point can be made by looking at Figure 91 and

Figure 93. Figure 91 represents a group of positive frequencies

modulated by complex +f_DAC/4, while Figure 93 represents a

group of negative frequencies modulated by complex −f_DAC/4.

When looking at the real or imaginary outputs of the AD9777,

as shown in Figure 91 and Figure 93, the results look identical.

However, the spectrum analyzer cannot show the phase

relationship of these signals. The difference in phase between

the two signals becomes apparent when they are applied to the

AD8345 quadrature modulator, with the results shown in Figure

92 and Figure 94.

Rev. C | Page 42 of 60

AD9777

0

–20

–40

–60

–80

0

–20

–40

–60

–80

D

A

B

C

D

A

B

C

D

A

B

C D

A

B

C

–100

–2.0

–1.5

–1.0

–0.5

0

0.5

1.0

1.5

2.0

–2.0

–1.5

–1.0

–0.5

0

0.5

1.0

1.5

2.0

(LO)

f_OUT(×f_DATA

)

Figure 83. 2× Interpolation, Complex f_DAC/4 Modulation

Figure 86. 2× Interpolation, Complex f_DAC/8 Modulation

0

–20

–40

–60

–80

0

–20

–40

–60

–80

D

A

B

C

D

A

B

C

D

A

B

C

D

A

B

C

–100

–4.0

–3.0

–2.0

–1.0

0

1.0

2.0

3.0

4.0

–4.0

–3.0

–2.0

–1.0

0

1.0

2.0

3.0

4.0

(LO)

f_OUT(×f_DATA

)

Figure 84. 4× Interpolation, Complex f_DAC/4 Modulation

Figure 87. 4× Interpolation, Complex f_DAC/8 Modulation

0

–20

–40

–60

–80

0

–20

–40

–60

–80

D

A

B

C

D

A

B

C

D A

B C

D A

B C

–100

–8.0

–6.0

–4.0

–2.0

0

2.0

4.0

6.0

8.0

–8.0

–6.0

–4.0

–2.0

0

2.0

4.0

6.0

8.0

(LO)

f_OUT(×f_DATA

)

f_OUT(×f_DATA

)

Figure 85. 8× Interpolation, Complex f_DAC/4 Modulation

Figure 88. 8× Interpolation, Complex f_DAC/8 Modulation

Rev. C | Page 43 of 60

AD9777

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

0

10

20

30

40

50

750 760 770 780 790 800 810 820 830 840 850

FREQUENCY (MHz)

Figure 89. AD9777, Real DAC Output of Complex Input Signal Near Baseband

(Positive Frequencies Only), Interpolation = 4×, No Modulation in AD9777

Figure 92. AD9777 Complex Output from Figure 91, Now Quadrature

Modulated by AD8345 (LO = 800 MHz)

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

0

10

20

30

40

50

–100

750 760 770 780 790 800 810 820 830 840 850

FREQUENCY (MHz)

Figure 93. AD9777, Real DAC Output of Complex Input Signal Near

Baseband (Negative Frequencies Only), Interpolation = 4×,

Complex Modulation in AD9777 = −f_DAC/4

Figure 90. AD9777 Complex Output from Figure 89, Now Quadrature

Modulated by AD8345 (LO = 800 MHz)

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

750 760 770 780 790 800 810 820 830 840 850

0

10

20

30

40

50

FREQUENCY (MHz)

Figure 91. AD9777, Real DAC Output of Complex Input Signal Near

Baseband (Positive Frequencies Only), Interpolation = 4×,

Complex Modulation in AD9777 = +f_DAC/4

Figure 94. AD9777 Complex Output from Figure 93, Now Quadrature

Modulated by AD8345 (LO = 800 MHz)

Rev. C | Page 44 of 60

AD9777

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

0

–20

–40

–60

–80

–100

700 720 740 760 780 800 820 840 860 880 900

0

20

40

60

80

100

FREQUENCY (MHz)

Figure 95. AD9777, Real DAC Output of Complex Input Signal Near

Baseband (Positive Frequencies Only), Interpolation = 8×,

Complex Modulation in AD9777 = +f_DAC/8

Figure 96. AD9777 Complex Output from Figure 95, Now Quadrature

Modulated by AD8345 (LO = 800 MHz)

Rev. C | Page 45 of 60

AD9777

APPLYING THE OUTPUT CONFIGURATIONS

The following sections illustrate typical output configurations

for the AD9777. Unless otherwise noted, it is assumed that

I_OUTFSis set to a nominal 20 mA. For applications requiring

optimum dynamic performance, a differential output configu-

ration is suggested. A simple differential output may be

achieved by converting I_OUTAand I_OUTBto a voltage output by

terminating them to AGND via equal value resistors. This type

of configuration may be useful when driving a differential

voltage input device such as a modulator. If a conversion to a

single-ended signal is desired and the application allows for ac

coupling, an RF transformer may be useful; if power gain is

required, an op amp may be used. The transformer configura-

tion provides optimum high frequency noise and distortion

performance. The differential op amp configuration is suitable

for applications requiring dc coupling, signal gain, and/or level

shifting within the bandwidth of the chosen op amp.

For the typical situation, where I_OUTFS= 20 mA and R_Aand R_B

both equal 50 Ω, the equivalent circuit values become

V_SOURCE= 2 V_P-P

R_OUT= 100 Ω

Note that the output impedance of the AD9777 DAC itself is

greater than 100 kΩ and typically has no effect on the

impedance of the equivalent output circuit.

DIFFERENTIAL COUPLING USING A

TRANSFORMER

An RF transformer can be used to perform a differential-to-

single-ended signal conversion, as shown in Figure 98. A

differentially coupled transformer output provides the optimum

distortion performance for output signals whose spectral

content lies within the transformer’s pass band. An RF

transformer, such as the Mini-Circuits T1-1T, provides excellent

rejection of common-mode distortion (that is, even-order

harmonics) and noise over a wide frequency range. It also

provides electrical isolation and the ability to deliver twice the

power to the load. Transformers with different impedance ratios

may also be used for impedance matching purposes.

A single-ended output is suitable for applications requiring a

unipolar voltage output. A positive unipolar output voltage

results if I_OUTAand/or I_OUTBis connected to a load resistor, R_LOAD

referred to AGND. This configuration is most suitable for a

single-supply system requiring a dc-coupled, ground referred

output voltage. Alternatively, an amplifier could be configured

as an I-V converter, thus converting I_OUTAor I_OUTBinto a

negative unipolar voltage. This configuration provides the best

DAC dc linearity as I_OUTAor I_OUTBare maintained at ground or

virtual ground.

,

MINI-CIRCUITS

T1-1T

I

OUTA

DAC

R

LOAD

I

OUTB

Figure 98. Transformer-Coupled Output Circuit

UNBUFFERED DIFFERENTIAL OUTPUT,

EQUIVALENT CIRCUIT

The center tap on the primary side of the transformer must be

connected to AGND to provide the necessary dc current path

for both I_OUTAand I_OUTB. The complementary voltages appearing

at I_OUTAand I_OUTB(that is, V_OUTAand V_OUTB) swing symmetrically

around AGND and should be maintained within the specified

output compliance range of the AD9777. A differential resistor,

R_DIFF, can be inserted in applications where the output of the

transformer is connected to the load, R_LOAD, via a passive

reconstruction filter or cable. R_DIFFis determined by the

transformer’s impedance ratio and provides the proper source

termination that results in a low VSWR. Note that approxi-

In many applications, it may be necessary to understand the

equivalent DAC output circuit. This is especially useful when

designing output filters or when driving inputs with finite input

impedances. Figure 97 illustrates the output of the AD9777 and

the equivalent circuit. A typical application where this

information may be useful is when designing an interface filter

between the AD9777 and the Analog Devices AD8345

quadrature modulator.

V

+

–

I

OUT

OUTA

OUTB

mately half the signal power dissipates across R_DIFF

.

V

R

+ R

B

A

V

=

)

p-p

SOURCE

× (R + R

V

OUT

(DIFFERENTIAL)

I

OUTFS

A

B

Figure 97. DAC Output Equivalent Circuit

Rev. C | Page 46 of 60

AD9777

Gain/Offset Adjust

DIFFERENTIAL COUPLING USING AN OP AMP

The matching of the DAC output to the common-mode input

of the AD8345 allows the two components to be dc-coupled,

with no level shifting necessary. The combined voltage offset of

the two parts can therefore be compensated via the AD9777

programmable offset adjust. This allows excellent LO

cancellation at the AD8345 output. The programmable gain

adjust allows for optimal image rejection as well.

An op amp can also be used to perform a differential-to-single

ended conversion, as shown in Figure 99. This has the added

benefit of providing signal gain as well. In Figure 99, the

AD9777 is configured with two equal load resistors, R_LOAD, of

25 Ω. The differential voltage developed across I_OUTAand I_OUTBis

converted to a single-ended signal via the differential op amp

configuration. An optional capacitor can be installed across

I_OUTAand I_OUTB, forming a real pole in a low-pass filter. The

addition of this capacitor also enhances the op amp’s distortion

performance by preventing the DAC’s fast slewing output from

overloading the input of the op amp.

The AD9777 evaluation board includes an AD8345 and

recommended interface (Figure 105 and Figure 106). On the

output of the AD9777, R9 and R10 convert the DAC output

current to a voltage. R16 can be used to do a slight common-

mode shift if necessary. The (now voltage) signal is applied to a

low-pass reconstruction filter to reject DAC images. The

components installed on the AD9777 provide a 35 MHz cutoff

but can be changed to fit the application. A balun (Mini-

Circuits ADTL1-12) is used to cross the ground plane boundary

to the AD8345. Another balun (Mini-Circuits ETC1-1-13) is

used to couple the LO input of the AD8345. The interface

requires a low ac impedance return path from the AD8345,

therefore a single connection between the AD9777 and AD8345

ground planes is recommended.

500Ω

225Ω

I

OUTA

AD8021

DAC

I

OUTB

C

OPT

225Ω

500Ω

AVDD

R

OPT

225Ω

25Ω

Figure 99. Op Amp-Coupled Output Circuit

The common-mode (and second-order distortion) rejection of

this configuration is typically determined by the resistor

matching. The op amp used must operate from a dual supply

since its output is approximately 1.0 V. A high speed amplifier,

such as the AD8021, capable of preserving the differential

performance of the AD9777 while meeting other system level

objectives (for example, cost, power) is recommended. The op

amp’s differential gain, gain setting resistor values, and full-scale

output swing capabilities should all be considered when

optimizing this circuit. R_OPTis necessary only if level shifting is

required on the op amp output. In Figure 99, AVDD, which is

the positive analog supply for both the AD9777 and the op amp,

is also used to level shift the differential output of the AD9777

to midsupply (that is, AVDD/2).

The performance of the AD9777 and AD8345 in an image

reject transmitter, reconstructing three WCDMA carriers, can

be seen in Figure 100. The LO of the AD8345 in this application

is 800 MHz. Image rejection (50 dB) and LO feedthrough

(−78 dBFS) have been optimized with the programmable

features of the AD9777. The average output power of the digital

waveform for this test was set to −15 dBFS to account for the

peak-to-average ratio of the WCDMA signal.

0

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

INTERFACING WITH THE AD8345 QUADRATURE

MODULATOR

The AD9777 architecture was defined to operate in a transmit

signal chain using an image reject architecture. A quadrature

modulator is also required in this application and should be

designed to meet the output characteristics of the DAC as much

as possible. The AD8345 from Analog Devices meets many of

the requirements for interfacing with the AD9777. As with any

DAC output interface, there are a number of issues that have to

be resolved. The following sections list some of the major issues.

762.5

782.5

802.5

822.5

842.5

FREQUENCY (MHz)

Figure 100. AD9777/AD8345 Synthesizing a

Three-Carrier WCDMA Signal at an LO of 800 MHz

DAC Compliance Voltage/Input Common-Mode Range

The dynamic range of the AD9777 is optimal when the DAC

outputs swing between 1.0 V. The input common-mode range

of the AD8345, at 0.7 V, allows optimum dynamic range to be

achieved in both components.

Rev. C | Page 47 of 60

AD9777

EVALUATION BOARD

DAC Differential Outputs

The AD9777 evaluation board allows easy configuration of the

various modes, programmable via the SPI port. Software is

available for programming the SPI port from Windows® 95,

Windows 98, or Windows NT®/2000. The evaluation board also

contains an AD8345 quadrature modulator and support

circuitry that allows the user to optimally configure the AD9777

in an image reject transmit signal chain.

Transformers T2 and T3 should be in place. Note that the lower

band of operation for these transformers is 300 kHz to 500 kHz.

Jumpers 4, 8, 13 to 17, and 28 to 30 should remain unsoldered.

The outputs are taken from S3 and S4.

Using the AD8345

Remove Transformers T2 and T3. Jumpers JP4 and 28 to 30

should remain unsoldered. Jumpers 13 to 16 should be

soldered. The desired components for the low-pass interface

filters L6, L7, C55, and C81 should be in place. The LO drive is

connected to the AD8345 via J10 and the balun T4, and the

AD8345 output is taken from J9.

Figure 101 to Figure 104 describe how to configure the

evaluation board in the one-port and two-port input modes with

the PLL enabled and disabled. Refer to Figure 105 to Figure 114,

the schematics, and the layout for the AD9777 evaluation board

for the jumper locations described below. The AD9777 outputs

can be configured for various applications by referring to the

following instructions.

DAC Single-Ended Outputs

Remove Transformers T2 and T3. Solder jumper link JP4 or

JP28 to look at the DAC1 outputs. Solder jumper link JP29 or

JP30 to look at the DAC2 outputs. Jumpers 8 and 13 to 17

should remain unsoldered. Jumpers JP35 to JP38 may be used

to ground one of the DAC outputs while the other is measured

single-ended. Optimum single-ended distortion performance is

typically achieved in this manner. The outputs are taken from

S3 and S4.

Rev. C | Page 48 of 60

AD9777

LECROY

PULSE

GENERATOR

SIGNAL GENERATOR

TRIG

INP

DATACLK

CLK+/CLK–

INPUT CLOCK

AWG2021

OR

DG2020

40-PIN RIBBON CABLE

DAC1, DB15–DB0

DAC2, DB15–DB0

AD9777

JUMPER CONFIGURATION FOR TWO PORT MODE PLL ON

SOLDERED/IN UNSOLDERED/OUT

JP1 –

JP2 –

×

JP3 –

JP5 –

×

JP6 –

×

JP12 –

JP24 –

JP25 –

JP26 –

JP27 –

JP31 –

JP32 –

JP33 –

×

NOTES

1. TO USE PECL CLOCK DRIVER (U8), SOLDER JP41 AND JP42 AND REMOVE TRANSFORMER T1.

2. IN TWO-PORT MODE, IF DATACLK/PLL_LOCK IS PROGRAMMED TO OUTPUT PIN 8, JP25 AND

JP39 SHOULD BE SOLDERED. IF DATACLK/PLL_LOCK IS PROGRAMMED TO OUTPUT PIN 53,

JP46 AND JP47 SHOULD BE SOLDERED. SEE THE TWO PORT DATA INPUT MODE SECTION

FOR MORE INFORMATION.

Figure 101. Test Configuration for AD9777 in Two-Port Mode with PLL Enabled Signal Generator Frequency = Input Data Rate,

DAC Output Data Rate = Signal Generator Frequency × Interpolation Rate

LECROY

PULSE

GENERATOR

SIGNAL GENERATOR

TRIG

INP

ONEPORTCLK

CLK+/CLK–

INPUT CLOCK

AWG2021

OR

DG2020

DAC1, DB15–DB0

DAC2, DB15–DB0

AD9777

JUMPER CONFIGURATION FOR ONE PORT MODE PLL ON

SOLDERED/IN UNSOLDERED/OUT

JP1 –

JP2 –

×

JP3 –

×

JP5 –

JP6 –

×

JP12 –

JP24 –

JP25 –

JP26 –

JP27 –

JP31 –

JP32 –

JP33 –

×

NOTES

1. TO USE PECL CLOCK DRIVER (U8), SOLDER JP41 AND JP42 AND REMOVE TRANSFORMER T1.

Figure 102. Test Configuration for AD9777 in One-Port Mode with PLL Enabled, Signal Generator Frequency = One-Half Interleaved Input Data Rate,

ONEPORTCLK = Interleaved Input Data Rate, DAC Output Data Rate = Signal Generator Frequency × Interpolation Rate

Rev. C | Page 49 of 60

AD9777

LECROY

PULSE

GENERATOR

SIGNAL GENERATOR

TRIG

INP

DATACLK

CLK+/CLK–

INPUT CLOCK

AWG2021

OR

DG2020

40-PIN RIBBON CABLE

DAC1, DB15–DB0

DAC2, DB15–DB0

AD9777

JUMPER CONFIGURATION FOR TWO PORT MODE PLL OFF

SOLDERED/IN UNSOLDERED/OUT

JP1 –

JP2 –

×

JP3 –

JP5 –

×

JP6 –

×

JP12 –

JP24 –

JP25 –

JP26 –

JP27 –

JP31 –

JP32 –

JP33 –

×

NOTES

1. TO USE PECL CLOCK DRIVER (U8), SOLDER JP41 AND JP42 AND REMOVE TRANSFORMER T1.

2. IN TWO-PORT MODE, IF DATACLK/PLL_LOCK IS PROGRAMMED TO OUTPUT PIN 8, JP25 AND

JP39 SHOULD BE SOLDERED. IF DATACLK/PLL_LOCK IS PROGRAMMED TO OUTPUT PIN 53,

JP46 AND JP47 SHOULD BE SOLDERED. SEE THE TWO PORT DATA INPUT MODE SECTION

FOR MORE INFORMATION.

Figure 103. Test Configuration for AD9777 in Two-Port Mode with PLL Disabled, DAC Output Data Rate = Signal Generator Frequency,

DATACLK = Signal Generator Frequency/Interpolation Rate

LECROY

PULSE

GENERATOR

SIGNAL GENERATOR

TRIG

INP

ONEPORTCLK

CLK+/CLK–

INPUT CLOCK

AWG2021

OR

DG2020

DAC1, DB15–DB0

DAC2, DB15–DB0

AD9777

JUMPER CONFIGURATION FOR ONE PORT MODE PLL OFF

SOLDERED/IN UNSOLDERED/OUT

JP1 –

JP2 –

×

JP3 –

×

JP5 –

JP6 –

×

JP12 –

JP24 –

JP25 –

JP26 –

JP27 –

JP31 –

JP32 –

JP33 –

×

NOTES

1. TO USE PECL CLOCK DRIVER (U8), SOLDER JP41 AND JP42 AND REMOVE TRANSFORMER T1.

Figure 104. Test Configuration for AD9777 in One-Port Mode with PLL Disabled, DAC Output Data Rate = Signal Generator Frequency,

ONEPORTCLK = Interleaved Input Data Rate = 2× Signal Generator Frequency/Interpolation Rate.

Rev. C | Page 50 of 60

AD9777

0 6 C 0 R 3

G 2

G 3

N E B L

V P S 1

T

V O U

L O I P

L O I N

G 1 B

G 1 A

V P S 2

G 4 A

G 4 B

Q B B N I B B N

Q B B P I B B P

A D T L 1 - 1 2

0 6 C 0 C 3

0 8 C 0 C 5

Figure 105. AD8345 Circuitry on AD9777 Evaluation Board

Rev. C | Page 51 of 60

AD9777

C C 0 6 0 3

R C 1 2 0 6

C C 0 6 0 3

R C 0 6 0 3

C C 0 6 0 5

C C 0 6 0 3

C C 0 8 0 5

Figure 106. AD9777 Clock, Power Supplies, and Output Circuitry

Rev. C | Page 52 of 60

AD9777

Figure 107. AD9777 Evaluation Board Input (A Channel) and Clock Buffer Circuitry

Rev. C | Page 53 of 60

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Figure 108. AD9777 Evaluation Board Input (B Channel) and SPI Port Circuitry

Rev. C | Page 54 of 60

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Figure 109. AD9777 Evaluation Board Components, Top Side

Figure 110. AD9777 Evaluation Board Components, Bottom Side

Rev. C | Page 55 of 60

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Figure 111. AD9777 Evaluation Board Layout, Layer One (Top)

Figure 112. AD9777 Evaluation Board Layout, Layer Two (Ground Plane)

Rev. C | Page 56 of 60

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Figure 113. AD9777 Evaluation Board Layout, Layer Three (Power Plane)

Figure 114. AD9777 Evaluation Board Layout, Layer Four (Bottom)

Rev. C | Page 57 of 60

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OUTLINE DIMENSIONS

14.20

14.00 SQ

13.80

12.20

1.20

MAX

12.00 SQ

11.80

0.75

0.60

0.45

61

80

61

1

60

1

60

PIN 1

EXPOSED

PAD

6.00

BSC SQ

TOP VIEW

(PINS DOWN)

BOTTOM VIEW

(PINS UP)

0° MIN

1.05

1.00

0.95

0.20

0.09

7°

3.5°

0°

20

41

20

40

21

VIEW A

0.15

0.05

0.50 BSC

0.27

0.22

0.17

SEATING

PLANE

LEAD PITCH

0.08 MAX

COPLANARITY

VIEW A

ROTATED 90° CCW

COMPLIANT TO JEDEC STANDARDS MS-026-ADD-HD

Figure 115. 80-Lead Thin Quad Flat Package, Exposed Pad [TQFP_EP]

(SV-80-1)

Dimensions shown in millimeters

ORDERING GUIDE

Model

Temperature Range

−40°C to +85°C

Package Description

Package Option

AD9777BSV

AD9777BSVRL

AD9777BSVZ¹

80-Lead Thin Quad Flat Package, Exposed Pad [TQFP_EP]

Evaluation Board

SV-80-1

AD9777BSZVRL¹−40°C to +85°C

AD9777-EB

¹Z = Pb-free part.

Rev. C | Page 58 of 60

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NOTES

Rev. C | Page 59 of 60

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NOTES

©

registered trademarks are the property of their respective owners.

C02706-0-1/06(C)

Rev. C | Page 60 of 60


型号：	AD9777AST
厂家：	ADI
描述：	IC SERIAL INPUT LOADING, 16-BIT DAC, PQFP80, 1.40 MM HEIGHT, PLASTIC, LQFP-80, Digital to Analog Converter 输入元件转换器
文件：	总60页 (文件大小：1451K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

AD9777AST [ADI]

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AD9777BSV

AD9777BSVRL

AD9777BSVZ

AD9777BSVZRL

AD9777BSZVRL

AD9777EB

AD9778

AD9778-EB

AD9778A

AD9778A-EBZ

AD9778ABSVZ

AD9778ABSVZRL