ADP2107ACPZ-R7 [ADI]
1 Amp/1.5 Amp/2 Amp Synchronous, Step-Down DC-to-DC Converters; 1安培/ 1.5安培/ 2安培同步，降压型DC- DC转换器
| 型号： | ADP2107ACPZ-R7 |
| 厂家： | 
ADI |
| 描述： | 1 Amp/1.5 Amp/2 Amp Synchronous, Step-Down DC-to-DC Converters |
| 文件： | 总32页 (文件大小：1006K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
1 Amp/1.5 Amp/2 Amp Synchronous,
Step-Down DC-to-DC Converters
ADP2105/ADP2106/ADP2107
GENERAL DESCRIPTION
FEATURES
Extremely high 97% efficiency
Ultralow quiescent current: 20 μA
1.2 MHz switching frequency
0.1 μA shutdown supply current
Maximum load current:
ADP2105: 1 A
ADP2106: 1.5 A
ADP2107: 2 A
Input voltage: 2.7 V to 5.5 V
Output voltage: 0.8 V to VIN
Maximum duty cycle: 100%
Smoothly transitions into low dropout (LDO) mode
Internal synchronous rectifier
Small 16-lead 4 mm × 4 mm LFCSP_VQ package
Optimized for small ceramic output capacitors
Enable/Shutdown logic input
Undervoltage lockout
The ADP2105/ADP2106/ADP2107 are low quiescent current,
synchronous, step-down dc-to-dc converters in a compact 4 mm ×
4 mm LFCSP_VQ package. At medium-to-high load currents,
these devices use a current-mode, constant-frequency pulse
width modulation (PWM) control scheme for excellent stability
and transient response. To ensure the longest battery life in
portable applications, the ADP2105/ADP2106/ADP2107 use a
pulse frequency modulation (PFM) control scheme under light
load conditions that reduces switching frequency to save power.
The ADP2105/ADP2106/ADP2107 run from input voltages of
2.7 V to 5.5 V, allowing single Li+/Li− polymer cell, multiple
alkaline/NiMH cells, PCMCIA, and other standard power sources.
The output voltage of ADP2105/ADP2106/ADP2107-ADJ is
adjustable from 0.8 V to the input voltage, while the ADP2105/
ADP2106/ADP2107-xx are available in preset output voltage
options of 3.3 V, 1.8 V, 1.5 V, and 1.2 V. Each of these variations is
available in three maximum current levels, 1 A (ADP2105), 1.5 A
(ADP2106), and 2 A (ADP2107). The power switch and synchro-
nous rectifier are integrated for minimal external part count
and high efficiency. During logic-controlled shutdown, the
input is disconnected from the output, and it draws less than
0.1 μA from the input source. Other key features include
undervoltage lockout to prevent deep-battery discharge and
programmable soft start to limit inrush current at startup.
Soft start
APPLICATIONS
Mobile handsets
PDAs and palmtop computers
Telecommunication/Networking equipment
Set top boxes
Audio/Video consumer electronics
TYPICAL PERFORMANCE CHARACTERISTICS
TYPICAL OPERATING CIRCUIT
0.1μF
V
INPUT VOLTAGE = 2.7V TO 5.5V
10Ω
IN
100
10μF
V
= 2.5V
OUT
V
= 3.6V
IN
V
= 3.3V
FB
IN
16
15
14
IN PWIN1
LX2 12
13
95
90
85
80
75
FB GND
ON
OUTPUT VOLTAGE = 2.5V
1
2
3
4
EN
OFF
2μH
PGND
11
10
9
GND
GND
GND
10μF 4.7μF
85kΩ
ADP2107-ADJ
V
= 5V
IN
LX1
FB
V
IN
40kΩ
PWIN2
LOAD
COMP SS AGND NC
5
0A TO 2A
10μF
6
7
8
1nF
70kΩ
120pF
0
200 400 600 800 1000 1200 1400 1600 1800 2000
LOAD CURRENT (mA)
NC = NO CONNECT
Figure 2. Circuit Configuration of ADP2107 with VOUT = 2.5 V
Figure 1. Efficiency vs. Load Current for the ADP2107 with VOUT = 2.5 V
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registeredtrademarks arethe property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
Fax: 781.461.3113
www.analog.com
©2006 Analog Devices, Inc. All rights reserved.
ADP2105/ADP2106/ADP2107
TABLE OF CONTENTS
Features .............................................................................................. 1
Setting the Output Voltage........................................................ 15
Inductor Selection...................................................................... 16
Output Capacitor Selection....................................................... 17
Input Capacitor Selection.......................................................... 17
Input Filter................................................................................... 18
Soft Start ...................................................................................... 18
Loop Compensation .................................................................. 18
Bode Plots.................................................................................... 19
Load Transient Response .......................................................... 20
Efficiency Considerations ......................................................... 21
Thermal Considerations............................................................ 21
Design Example.......................................................................... 22
External Component Recommendations.................................... 24
Circuit Board Layout Recommendations ................................... 26
Evaluation Board ............................................................................ 27
Evaluation Board Schematic (ADP2107-1.8V)...................... 27
Applications....................................................................................... 1
General Description......................................................................... 1
Typical Performance Characteristics ............................................. 1
Typical Operating Circuit................................................................ 1
Revision History ............................................................................... 2
Specifications..................................................................................... 3
Absolute Maximum Ratings............................................................ 5
Thermal Resistance ...................................................................... 5
Boundary Condition.................................................................... 5
ESD Caution.................................................................................. 5
Pin Configuration and Function Descriptions............................. 6
Typical Performance Characteristics ............................................. 7
Theory of Operation ...................................................................... 12
Control Scheme .......................................................................... 12
PWM Mode Operation.............................................................. 12
PFM Mode Operation................................................................ 12
Pulse-Skipping Threshold ......................................................... 12
100% Duty Cycle Operation (LDO Mode)............................. 12
Slope Compensation .................................................................. 13
Features........................................................................................ 13
Applications Information .............................................................. 15
External Component Selection................................................. 15
Recommended PCB Board Layout
(Evaluation Board Layout)........................................................ 27
Application Circuits ....................................................................... 29
Outline Dimensions....................................................................... 31
Ordering Guide .......................................................................... 31
REVISION HISTORY
7/06—Revision 0: Initial Version
Rev. 0 | Page 2 of 32
ADP2105/ADP2106/ADP2107
SPECIFICATIONS
VIN = 3.6 V @ TA = 25°C, unless otherwise noted.1 Bold values indicate −40°C ≤ TJ ≤ +125°C.
Table 1.
Parameter
Conditions
Min
Typ
Max
Unit
INPUT CHARACTERISTICS
Input Voltage Range
Undervoltage Lockout Threshold
V
V
V
mV
2.7
2.2
2.0
5.5
2.6
2.5
VIN rising
VIN falling
2.4
2.2
200
Undervoltage Lockout Hysteresis2
OUTPUT CHARACTERISTICS
Output Regulation Voltage
ADP210x-3.3, load = 10 mA
ADP210x-3.3, VIN = 3.5 V to 5.5 V, no load to full load
ADP210x-1.8, load = 10 mA
ADP210x-1.8, VIN = 2.7 V to 5.5 V, no load to full load
ADP210x-1.5, load = 10 mA
ADP210x-1.5, VIN = 2.7 V to 5.5 V, no load to full load
ADP210x-1.2, load = 10 mA
ADP210x-1.2, VIN = 2.7 V to 5.5 V, no load to full load
ADP2105
ADP2106
ADP2107
Measured in servo loop
ADP210x-ADJ
3.267 3.3
3.3
1.782 1.8
1.8
1.485 1.5
1.5
1.188 1.2
3.333
3.399
1.818
1.854
1.515
1.545
1.212
1.236
V
V
V
V
V
V
V
V
%/A
%/A
%/A
%/V
V
3.201
1.746
1.455
1.2
0.4
0.5
0.6
0.1
1.164
Load Regulation
Line Regulation3
0.3
VIN
Output Voltage Range
FEEDBACK CHARACTERISTICS
OUT_SENSE Bias Current
0.8
ADP210x-1.2
ADP210x-1.5
ADP210x-1.8
ADP210x-3.3
ADP210x-ADJ
ADP210x-ADJ
3
6
μA
μA
μA
μA
V
4
8
5
10
10
0.8
20
FB Regulation Voltage
FB Bias Current
0.784
−0.1
0.816
+0.1
μA
INPUT CURRENT CHARACTERISTICS
IN Operating Current
ADP210x-ADJ, VFB = 0.9 V
20
20
0.1
30
30
15
μA
μA
μA
ADP210x-xx, output voltage 10% above regulation voltage
VEN = 0 V
IN Shutdown Current
LX (SWITCH NODE) CHARACTERISTICS
LX On Resistance4
P-channel switch
100
90
165
140
15
mΩ
mΩ
μA
A
A
A
N-channel synchronous rectifier
VIN = 5.5 V, VLX = 0 V, 5.5 V
LX Leakage Current4
LX Peak Current Limit4
0.1
2.9
2.25
1.5
P-channel switch, ADP2107
P-channel switch, ADP2106
P-channel switch, ADP2105
In PWM mode of operation, VIN = 5.5 V
2.6
2.0
1.3
3.3
2.6
1.8
100
LX Minimum On-Time4
ENABLE CHARACTERISTICS
EN Input High Voltage
EN Input Low Voltage
ns
VIN = 2.7 V to 5.5 V
VIN = 2.7 V to 5.5 V
VIN = 5.5 V, VEN = 0 V, 5.5 V
V
V
2
0.4
+1
EN Input Leakage Current
−1
1
−0.1
1.2
μA
MHz
OSCILLATOR FREQUENCY
SOFT START PERIOD
VIN = 2.7 V to 5.5 V
CSS = 1 nF
1.4
750
1000
1200
μs
Rev. 0 | Page 3 of 32
ADP2105/ADP2106/ADP2107
Parameter
Conditions
Min
Typ
Max
Unit
THERMAL CHARACTERISTICS
Thermal Shutdown Threshold
Thermal Shutdown Hysteresis
140
40
°C
°C
COMPENSATOR TRANSCONDUCTANCE (Gm)
CURRENT SENSE AMPLIFIER GAIN (GCS)2
50
μA/V
A/V
A/V
A/V
ADP2105
ADP2106
ADP2107
1.875
2.8125
3.625
1 All limits at temperature extremes are guaranteed via correlation using standard statistical quality control (SQC). Typical values are at TA = 25°C.
2 Guaranteed by design.
3 The ADP2015/ADP2106/ADP2107 line regulation was measured in a servo loop on the ATE that adjusts the feedback voltage to achieve a specific comp voltage.
4 All LX (switch node) characteristics are guaranteed only when the LX1 and LX2 pins are tied together.
5 These specifications are guaranteed from −40°C to +85°C.
Rev. 0 | Page 4 of 32
ADP2105/ADP2106/ADP2107
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter
THERMAL RESISTANCE
Rating
θJA is specified for the worst-case conditions, that is, a device
soldered in a circuit board for surface-mount packages.
IN, EN, SS, COMP, OUT_SENSE/FB to
AGND
−0.3 V to +6 V
LX1, LX2 to PGND
PWIN1, PWIN2 to PGND
PGND to AGND
GND to AGND
PWIN1, PWIN2 to IN
−0.3 V to (VIN + 0.3 V)
−0.3 V to +6 V
−0.3 V to +0.3 V
−0.3 V to +0.3 V
−0.3 V to +0.3 V
Table 3. Thermal Resistance
Package Type
16-Lead LFCSP_VQ/QFN
Maximum Power Dissipation
1
θJA
40
1
Unit
°C/W
W
Operating Junction Temperature Range −40°C to +125°C
1 θJA is specified for the worst-case conditions; that is, θJA is specified for device
soldered in circuit board for surface mount packages.
Storage Temperature Range
Soldering Conditions
−65°C to +150°C
JEDEC J-STD-020
BOUNDARY CONDITION
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Natural convection, 4-layer board, exposed pad soldered to
the PCB.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the
human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. 0 | Page 5 of 32
ADP2105/ADP2106/ADP2107
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
PIN 1
INDICATOR
12 LX2
11 PGND
10 LX1
EN
GND
GND
GND
1
2
3
4
ADP2105/
ADP2106/
ADP2107
TOP VIEW
(Not to Scale)
9
PWIN2
NC = NO CONNECT
Figure 3. Pin Configuration
Table 4. Pin Function Descriptions
Mnemonic
Pin No. ADP210x-xx ADP210x-ADJ Description
1
EN
EN
Enable Input. Drive EN high to turn on the ADP2105/ADP2106/ADP2107. Drive EN low to turn
it off and reduce the input current to 0.1 μA.
2, 3, 4,
15
GND
GND
Test Pins. These pins are used by Analog Devices, Inc. for internal testing and are not ground
return pins. Tie these pins to the AGND plane as close to the ADP2105/ADP2106/ADP2107 as
possible.
5
COMP
COMP
Feedback Loop Compensation Node. COMP is the output of the internal transconductance
error amplifier. Place a series RC network from COMP to AGND to compensate the converter.
See the Loop Compensation section.
6
7
SS
SS
Soft Start Input. Place a capacitor from SS to AGND to set the soft start period. A 1 nF capacitor
sets a 1 ms soft start period.
Analog Ground. Connect the ground of the compensation components, soft start capacitor,
and the voltage divider on the FB pin to the AGND pin as close as possible to the ADP2105/
ADP2106/ADP2107. Also connect AGND to the exposed pad of ADP2105/ADP2106/ADP2107.
AGND
AGND
8
NC
NC
No Connect. Not internally connected. Can be connected to other pins or left unconnected.
9, 13
PWIN2,
PWIN1
PWIN2, PWIN1 Power Source Inputs. The source of the PFET high-side switch. Bypass each PWIN pin to the nearest
PGND plane with a 4.7 μF or greater capacitor as close as possible to the ADP2105/ADP2106/
ADP2107. See the Input Capacitor Selection section.
10, 12
11
LX1, LX2
PGND
IN
LX1, LX2
PGND
IN
Switch Outputs. The drain of the P-channel power switch and N-channel synchronous rectifier.
Tie the two LX pins together and connect the output LC filter between LX and the output
voltage.
Power Ground. Connect the ground return of all input and output capacitors to PGND pin,
using a power ground plane as close as possible to the ADP2105/ADP2106/ADP2107. Also
connect PGND to the exposed pad of the ADP2105/ADP2106/ADP2107.
ADP2105/ADP2106/ADP2107 Power Input. The power source for the ADP2105/ADP2106/
ADP2107 internal circuitry. Connect IN and PWIN1 with a 10 Ω resistor as close as possible to
the ADP2105/ADP2106/ADP2107. Bypass IN to AGND with a 0.1 μF or greater capacitor. See
the Input Filter section.
14
16
OUT_SENSE
FB
Output Voltage Sense or Feedback Input. For fixed output versions, connect OUT_SENSE to the
output voltage. For adjustable versions, FB is the input to the error amplifier. Drive FB through
a resistive voltage divider to set the output voltage. The FB regulation voltage is 0.8 V.
Rev. 0 | Page 6 of 32
ADP2105/ADP2106/ADP2107
TYPICAL PERFORMANCE CHARACTERISTICS
100
100
95
90
85
80
75
70
65
60
55
50
95
V
= 2.7V
IN
90
85
80
75
70
65
60
55
50
V
= 2.7V
IN
V
= 3.6V
IN
V
= 3.6V
IN
V
= 5.5V
IN
V
= 4.2V
IN
V
= 4.2V
IN
V
= 5.5V
IN
INDUCTOR: SD14, 2.5µH
INDUCTOR: SD3814, 3.3µH
DCR: 60mꢀ
A
DCR: 93mꢀ
T
= 25°C
T
= 25°C
A
1
1
1
10
100
1000
1
1
1
1000
10
100
LOAD CURRENT (mA)
LOAD CURRENT (mA)
Figure 4. Efficiency—ADP2105 (1.2 V Output)
Figure 7. Efficiency—ADP2105 (1.8 V Output)
100
95
90
85
80
75
70
65
60
55
50
100
95
90
85
80
75
70
65
60
55
50
V
= 3.6V
IN
V
= 2.7V
IN
V
= 3.6V
IN
V
= 4.2V
V
= 5.5V
IN
IN
V
= 4.2V
IN
V
= 5.5V
IN
INDUCTOR: D62LCB, 2µH
DCR: 28mꢀ
= 25°C
INDUCTOR: CDRH5D18, 4.1μH
DCR: 43mꢀ
T
T
= 25°C
A
A
10
100
LOAD CURRENT (mA)
1000
10000
1000
10
100
LOAD CURRENT (mA)
Figure 5. Efficiency—ADP2105 (3.3 V Output)
Figure 8. Efficiency—ADP2106 (1.2 V Output)
100
95
90
85
80
75
70
65
60
55
50
100
95
90
85
80
75
70
65
60
55
50
V
= 3.6V
IN
V
= 2.7V
IN
V
= 5.5V
IN
V
= 4.2V
IN
V
= 4.2V
IN
V
= 5.5V
IN
V
= 3.6V
10
IN
INDUCTOR: D62LCB, 2µH
INDUCTOR: D62LCB, 3.3µH
DCR: 28mꢀ
DCR: 47mꢀ
T = 25°C
T
= 25°C
A
A
10000
10000
10
100
LOAD CURRENT (mA)
1000
100
LOAD CURRENT (mA)
1000
Figure 6. Efficiency—ADP2106 (1.8 V Output)
Figure 9. Efficiency—ADP2106 (3.3 V Output)
Rev. 0 | Page 7 of 32
ADP2105/ADP2106/ADP2107
100
100
95
90
85
80
75
70
65
60
55
50
V
= 3.6V
IN
95
V
= 2.7V
IN
90
85
80
75
70
65
60
55
50
V
= 2.7V
IN
V
= 3.6V
IN
V
= 4.2V
IN
V
= 5.5V
IN
V
= 4.2V
IN
V
= 5.5V
IN
INDUCTOR: SD12, 1.2µH
DCR: 37mꢀ
= 25°C
INDUCTOR: D62LCB, 1.5µH
DCR: 21mꢀ
T
T
= 25°C
A
A
1
10
100
LOAD CURRENT (mA)
1000
10000
1
10000
10000
10000
10
100
LOAD CURRENT (mA)
1000
Figure 10. Efficiency—ADP2107 (1.2 V)
Figure 13. Efficiency—ADP2107 (1.8 V)
100
95
90
85
80
75
70
65
60
55
50
1.23
1.22
1.21
1.20
1.19
1.18
1.17
2.7V, –40°C
3.6V, –40°C
5.5V, –40°C
2.7V, +25°C
3.6V, +25°C
5.5V, +25°C
2.7V, +125°C
3.6V, +125°C
5.5V, +125°C
V
= 5.5V
IN
V
= 4.2V
IN
V
= 3.6V
IN
INDUCTOR: CDRH5D28, 2.5µH
DCR: 13mꢀ
A
T
= 25°C
1
10000
0.01
0.1
1
10
100
1000
10
100
1000
LOAD CURRENT (mA)
LOAD CURRENT (mA)
Figure 11. Efficiency—ADP2107 (3.3 V)
Figure 14. Output Voltage Accuracy—ADP2107 (1.2 V)
1.85
1.83
1.81
1.79
1.77
1.75
3.38
3.36
3.34
3.32
3.30
3.28
3.26
3.24
3.22
3.6V, –40°C
5.5V, –40°C
3.6V, +25°C
5.5V, +25°C
3.6V, +125°C
5.5V, +125°C
2.7V, –40°C
3.6V, –40°C
5.5V, –40°C
2.7V, +25°C
3.6V, +25°C
5.5V, +25°C
2.7V, +125°C
3.6V, +125°C
5.5V, +125°C
0.1
1
10
100
1000
10000
0.01
0.1
1
10
100
1000
LOAD CURRENT (mA)
LOAD CURRENT (mA)
Figure 12. Output Voltage Accuracy—ADP2107 (1.8 V)
Figure 15. Output Voltage Accuracy—ADP2107 (3.3 V)
Rev. 0 | Page 8 of 32
ADP2105/ADP2106/ADP2107
10000
1000
100
10
120
100
80
60
40
20
0
PMOS POWER SWITCH
+25°C
–40°C
NMOS SYNCHRONOUS RECTIFIER
+125°C
T
= 25°C
A
1
0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0 4.4 4.8 5.2
2.7
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Figure 16. Quiescent Current vs. Input Voltage
Figure 19. Switch On Resistance vs. Input Voltage
0.802
1260
1250
1240
1230
1220
1210
1200
1190
0.801
0.800
0.799
0.798
0.797
0.796
0.795
+125°C
+25°C
–40°C
2.7
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
–40
–20
0
20
40
60
80
100
120125
INPUT VOLTAGE (V)
TEMPERATURE (°C)
Figure 20. Switching Frequency vs. Input Voltage
Figure 17. Feedback Voltage vs. Temperature
2.35
2.30
2.25
2.20
2.15
2.10
2.05
2.00
1.95
1.90
1.85
1.75
1.70
1.65
1.60
1.55
1.50
1.45
1.40
1.35
1.30
1.25
ADP2106 (1.5A)
ADP2105 (1A)
T
A
= 25°C
5.4 5.7
T
= 25°C
A
2.7
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
2.7
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
5.7
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Figure 21. Peak Current Limit of ADP2106
Figure 18. Peak Current Limit of ADP2105
Rev. 0 | Page 9 of 32
ADP2105/ADP2106/ADP2107
3.00
2.95
2.90
135
120
105
90
2.85
ADP2107 (2A)
V
= 1.2V
OUT
2.80
75
2.75
2.70
2.65
2.60
2.55
2.50
60
45
V
= 1.8V
OUT
V
= 2.5V
OUT
30
15
T
= 25°C
T
= 25°C
A
A
0
2.7
2.7
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
5.7
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
5.7
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Figure 22. Peak Current Limit of ADP2107
Figure 25. Pulse Skipping Threshold vs. Input Voltage for ADP2105
150
195
180
135
120
105
90
V
= 1.2V
165
150
135
120
105
90
OUT
V
V
= 1.8V
= 2.5V
OUT
OUT
V
= 1.2V
OUT
75
60
75
V
= 2.5V
OUT
V
= 1.8V
OUT
60
45
45
30
30
15
15
T
= 25°C
T
= 25°C
A
A
0
2.7
0
2.7
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
5.7
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
5.7
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Figure 23. Pulse Skipping Threshold vs. Input Voltage for ADP2106
Figure 26. Pulse Skipping Threshold vs. Input Voltage for ADP2107
140
LX NODE (SWITCH NODE)
120
3
PMOS POWER SWITCH
100
INDUCTOR CURRENT
Δ: 260mV
80
@: 3.26V
NMOS SYNCHRONOUS RECTIFIER
60
40
20
0
1
OUTPUT VOLTAGE
4
CH1 1V
CH3 5V
M
T
10µs
45.8%
A
CH1
1.78V
–40
–20
0
20
40
60
80
100
120
CH4 1Aꢀ
JUNCTION TEMPERATURE (°C)
Figure 24. Short Circuit Response at Output
Figure 27. Switch On Resistance vs. Temperature
Rev. 0 | Page 10 of 32
ADP2105/ADP2106/ADP2107
LX NODE (SWITCH NODE)
LX NODE
(SWITCH NODE)
3
1
3
1
OUTPUT VOLTAGE (AC-COUPLED)
INDUCTOR CURRENT
OUTPUT VOLTAGE (AC-COUPLED)
INDUCTOR CURRENT
4
4
CH1 50mV
CH3 2V
M
T
2µs
6%
A
CH3
3.88V
CH1 20mV
CH3 2V
M
T
1µs
17.4%
A
CH3
3.88V
CH4 200mAꢀ
CH4 1Aꢀ
Figure 28. PFM Mode of Operation at Very Light Load (10 mA)
Figure 31. PWM Mode of Operation at Medium/Heavy Load (1.5 A)
LX NODE (SWITCH NODE)
3
CHANNEL 3
Δ: 2.86A
3
FREQUENCY
@: 2.86A
= 336.6kHz
LX NODE (SWITCH NODE)
1
4
INDUCTOR CURRENT
OUTPUT VOLTAGE (AC-COUPLED)
OUTPUT VOLTAGE
1
4
INDUCTOR CURRENT
CH1 50mV
CH3 2V
M
T
400ns
17.4%
A
CH3
3.88V
CH1 1V
CH3 5V
M
T
4µs
45%
A
CH3
1.8V
CH4 200mAꢀ
CH4 1Aꢀ
Figure 29. DCM Mode of Operation at Light Load (100 mA)
Figure 32. Current Limit Behavior of ADP2107 (Frequency Foldback)
LX NODE (SWITCH NODE)
ENABLE VOLTAGE
3
OUTPUT VOLTAGE
3
1
1
INDUCTOR CURRENT
OUTPUT VOLTAGE (AC-COUPLED)
INDUCTOR CURRENT
4
4
CH1 20mV
CH3 2V
M
T
2µs
13.4%
A
CH3
1.84V
CH1 1V
CH3 5V
M
T
400µs
20.2%
A
CH1
1.84V
CH4 1Aꢀ
CH4 500mAꢀ
Figure 30. Minimum Off Time Control at Dropout
Figure 33. Startup and Shutdown Waveform (CSS = 1 nF → SS Time = 1 ms)
Rev. 0 | Page 11 of 32
ADP2105/ADP2106/ADP2107
THEORY OF OPERATION
The ADP2105/ADP2106/ADP2107 are step-down, dc-to-dc
converters that use a fixed frequency, peak current-mode
architecture with an integrated high-side switch and low-side
synchronous rectifier. The high 1.2 MHz switching frequency
and tiny 16-lead, 4 mm × 4 mm LFCSP_VQ package allow for
a small step-down dc-to-dc converter solution. The integrated
high-side switch (P-channel MOSFET) and synchronous rectifier
(N-channel MOSFET) yield high efficiency at medium-to-
heavy loads. Light load efficiency is improved by smoothly
transitioning to variable frequency PFM mode.
PFM MODE OPERATION
The ADP2105/ADP2106/ADP2107 smoothly transition to the
variable frequency PFM mode of operation when the load current
decreases below the pulse-skipping threshold current, switching
only as necessary to maintain the output voltage within regulation.
When the output voltage dips below regulation, the ADP2105/
ADP2106/ADP2107 enter PWM mode for a few oscillator cycles
to increase the output voltage back to regulation. During the wait
time between bursts, both power switches are off, and the output
capacitor supplies all the load current. Because the output voltage
dips and recovers occasionally, the output voltage ripple in this
mode is larger than the ripple in the PWM mode of operation.
The ADP2105/ADP2106/ADP2107-ADJ operate with an input
voltage from 2.7 V to 5.5 V and regulate an output voltage down
to 0.8 V. The ADP2105/ADP2106/ADP2107 are also available with
preset output voltage options of 3.3 V, 1.8 V, 1.5 V, and 1.2 V.
PULSE-SKIPPING THRESHOLD
The output current at which the ADP2105/ADP2106/ADP2107
transition from variable frequency PFM control to fixed frequency
PWM control is called the pulse-skipping threshold. The pulse-
skipping threshold has been optimized for excellent efficiency
over all load currents. The variation of pulse-skipping threshold
with input voltage and output voltage is shown in Figure 23,
Figure 25, and Figure 26.
CONTROL SCHEME
The ADP2105/ADP2106/ADP2107 operate with a fixed
frequency, peak current-mode PWM control architecture at
medium-to-high loads for high efficiency, but shift to a variable
frequency PFM control scheme at light loads for lower quies-
cent current. When operating in fixed frequency PWM mode,
the duty cycle of the integrated switches is adjusted to regulate
the output voltage, but when operating in PFM mode at light
loads, the switching frequency is adjusted to regulate the output
voltage.
100% DUTY CYCLE OPERATION (LDO MODE)
As the input voltage drops, approaching the output voltage,
the ADP2105/ADP2106/ADP2107 smoothly transition to 100%
duty cycle, maintaining the P-channel MOSFET switch on continu-
ously. This allows the ADP2105/ADP2106/ADP2107 to regulate
the output voltage until the drop in input voltage forces the
P-channel MOSFET switch to enter dropout, as shown in the
following equation:
The ADP2105/ADP2106/ADP2107 operate in the PWM mode
only when the load current is greater than the pulse-skipping
threshold current. At load currents below this value, the converter
smoothly transitions to the PFM mode of operation.
V
IN(MIN) = IOUT × (RDS(ON) − P + DCRIND) + VOUT(NOM)
PWM MODE OPERATION
In PWM mode, the ADP2105/ADP2106/ADP2107 operate at
a fixed frequency of 1.2 MHz set by an internal oscillator. At the
start of each oscillator cycle, the P-channel MOSFET switch is
turned on, putting a positive voltage across the inductor. Current
in the inductor increases until the current sense signal crosses
the peak inductor current level that turns off the P-channel
MOSFET switch and turns on the N-channel MOSFET synchro-
nous rectifier. This puts a negative voltage across the inductor,
causing the inductor current to decrease. The synchronous
rectifier stays on for the rest of the cycle, unless the inductor
current reaches zero, which causes the zero-crossing comparator
to turn off the N-channel MOSFET, as well. The peak inductor
current is set by the voltage on the COMP pin. The COMP pin
is the output of a transconductance error amplifier that compares
the feedback voltage with an internal 0.8 V reference.
The ADP2105/ADP2106/ADP2107 achieve 100% duty cycle
operation by stretching the P-channel MOSFET switch on-time
if the inductor current does not reach the peak inductor current
level by the end of the clock cycle. Once this happens, the oscil-
lator remains off until the inductor current reaches the peak
inductor current level, at which time the switch is turned off and
the synchronous rectifier is turned on for a fixed off-time. At
the end of the fixed off-time, another cycle is initiated. As the
ADP2105/ADP2106/ADP2107 approach dropout, the switching
frequency decreases gradually to smoothly transition to 100%
duty cycle operation.
Rev. 0 | Page 12 of 32
ADP2105/ADP2106/ADP2107
Short Circuit Protection
SLOPE COMPENSATION
The ADP2105/ADP2106/ADP2107 include frequency foldback
to prevent output current run-away on a hard short. When the
voltage at the feedback pin falls below 0.3 V, indicating the possi-
bility of a hard short at the output, the switching frequency is
reduced to 1/4 of the internal oscillator frequency. The reduction
in the switching frequency gives more time for the inductor to
discharge, preventing a runaway of output current.
Slope compensation stabilizes the internal current control loop
of the ADP2105/ADP2106/ADP2107 when operating beyond
50% duty cycle to prevent sub-harmonic oscillations. It is imple-
mented by summing a fixed scaled voltage ramp to the current
sense signal during the on-time of the P-channel MOSFET switch.
The slope compensation ramp value determines the minimum
inductor that can be used to prevent sub-harmonic oscillations
at a given output voltage. The slope compensation ramp values
for ADP2105/ADP2106/ADP2107 follow. For more information,
see the Inductor Selection section.
Undervoltage Lockout (UVLO)
To protect against deep battery discharge, undervoltage lockout
circuitry is integrated on the ADP2105/ADP2106/ADP2107.
If the input voltage drops below the 2.2 V UVLO threshold, the
ADP2105/ADP2106/ADP2107 shut down, and both the power
switch and synchronous rectifier turn off. Once the voltage rises
again above the UVLO threshold, the soft start period is initiated,
and the part is enabled.
For the ADP2105:
Slope Compensation Ramp Value = 0.72 A/ꢀs
For the ADP2106:
Slope Compensation Ramp Value = 1.07 A/ꢀs
For the ADP2107:
Thermal Protection
In the event that the ADP2105/ADP2106/ADP2107 junction
temperatures rise above 140°C, the thermal shutdown circuit turns
off the converter. Extreme junction temperatures can be the
result of high current operation, poor circuit board design, and/or
high ambient temperature. A 40°C hysteresis is included so that
when thermal shutdown occurs, the ADP2105/ADP2106/
ADP2107 do not return to operation until the on-chip tempera-
ture drops below 100°C. When coming out of thermal
shutdown, soft start is initiated.
Slope Compensation Ramp Value = 1.38 A/ꢀs
FEATURES
Enable/Shutdown
Drive EN high to turn on the ADP2105/ADP2106/ADP2107.
Drive EN low to turn off the ADP2105/ADP2106/ADP2107,
reducing input current below 0.1 μA. To force the ADP2105/
ADP2106/ADP2107 to automatically start when input power
is applied, connect EN to IN. When shut down, the ADP2105/
ADP2106/ADP2107 discharge the soft start capacitor, causing
a new soft start cycle every time they are re-enabled.
Soft Start
The ADP2105/ADP2106/ADP2107 include soft start circuitry
to limit the output voltage rise time to reduce inrush current at
startup. To set the soft start period, connect the soft start
capacitor (CSS) from SS to AGND. When the ADP2105/ADP2106/
ADP2107 are disabled, or if the input voltage is below the under-
voltage lockout threshold, CSS is internally discharged. When the
ADP2105/ADP2106/ADP2107 are enabled, CSS is charged through
an internal 0.8 ꢀA current source, causing the voltage at SS to rise
linearly. The output voltage rises linearly with the voltage at SS.
Synchronous Rectification
In addition to the P-channel MOSFET switch, the ADP2105/
ADP2106/ADP2107 include an integrated N-channel MOSFET
synchronous rectifier. The synchronous rectifier improves
efficiency, especially at low output voltage, and reduces cost and
board space by eliminating the need for an external rectifier.
Current Limit
The ADP2105/ADP2106/ADP2107 have protection circuitry to
limit the direction and amount of current flowing through the
power switch and synchronous rectifier. The positive current
limit on the power switch limits the amount of current that can
flow from the input to the output, while the negative current
limit on the synchronous rectifier prevents the inductor current
from reversing direction and flowing out of the load.
Rev. 0 | Page 13 of 32
ADP2105/ADP2106/ADP2107
5
14
COMP
IN
9
PWIN2
PWIN1
SOFT
START
REFERENCE
0.8V
6
16
16
7
SS
1
CURRENT SENSE
AMPLIFIER
13
FB
1
OUT_SENSE
GM ERROR
AMP
CURRENT
LIMIT
PWM/
PFM
CONTROL
AGND
FOR PRESET
VOLTAGES
OPTIONS ONLY
DRIVER
AND
ANTI-
2
3
GND
GND
GND
NC
10
12
LX1
LX2
SHOOT
THROUGH
SLOPE
COMPENSATION
4
8
15
GND
OSCILLATOR
ZERO CROSS
COMPARATOR
11
PGND
THERMAL
SHUTDOWN
1
EN
1
FB FOR ADP210x-ADJ (ADJUSTABLE VERSION) AND OUT_SENSE FOR ADP210x-xx (FIXED VERSION).
Figure 34. Block Diagram of the ADP2105/ADP2106/ADP2107
Rev. 0 | Page 14 of 32
ADP2105/ADP2106/ADP2107
APPLICATIONS INFORMATION
into account when calculating resistor values. The FB bias
current can be ignored for a higher divider string current, but
this degrades efficiency at very light loads.
EXTERNAL COMPONENT SELECTION
The external component selection for the ADP2105/ADP2106/
ADP2107 application circuits shown in Figure 35 and Figure 36
depend on input voltage, output voltage, and load current
requirements. Additionally, tradeoffs between performance
parameters like efficiency and transient response can be made
by varying the choice of external components.
To limit output voltage accuracy degradation due to FB bias
current to less than 0.05% (0.5% maximum), ensure that the
divider string current is greater than 20 ꢀA. To calculate the
desired resistor values, first determine the value of the bottom
divider string resistor, RBOT, by
SETTING THE OUTPUT VOLTAGE
VFB
ISTRING
RBOT
=
The output voltage of ADP2105/ADP2106/ADP2107-ADJ is
externally set by a resistive voltage divider from the output
voltage to FB. The ratio of the resistive voltage divider sets the
output voltage, while the absolute value of those resistors sets
the divider string current. For lower divider string currents, the
small 10 nA (0.1 ꢀA maximum) FB bias current should be taken
where:
FB = 0.8 V, the internal reference.
V
ISTRING is the resistor divider string current.
0.1μF
V
INPUT VOLTAGE = 2.7V TO 5.5V
10ꢀ
IN
C
IN1
V
OUT
16
15
14
GND IN PWIN1
LX2
13
OUT_SENSE
ON
1
2
3
4
12
EN
OFF
OUTPUT VOLTAGE = 1.2V, 1.5V, 1.8V, 3.3V
L
V
OUT
GND
PGND 11
LX1 10
ADP2105/
ADP2106/
ADP2107
C
OUT
LOAD
GND
V
IN
GND
PWIN2
9
C
IN2
COMP SS AGND NC
5
6
7
8
C
SS
R
COMP
C
COMP
NC = NO CONNECT
Figure 35. Typical Applications Circuit for Fixed Output Voltage Options (ADP2105/ADP2106/ADP2107-xx)
0.1μF
V
INPUT VOLTAGE = 2.7V TO 5.5V
C
10ꢀ
IN
IN1
FB
16
15
14
13
FB GND
IN PWIN1
ON
1
2
3
4
EN
LX2 12
OFF
OUTPUT VOLTAGE
= 0.8V TO V
L
IN
GND
GND
GND
PGND 11
ADP2105/
ADP2106/
ADP2107
C
OUT
R
R
TOP
LOAD
LX1
10
9
V
FB
IN
PWIN2
C
IN2
BOT
COMP SS AGND NC
5
6
7
8
C
SS
R
COMP
C
COMP
NC = NO CONNECT
Figure 36. Typical Applications Circuit for Adjustable Output Voltage Option (ADP2105/ADP2106/ADP2107-ADJ)
Rev. 0 | Page 15 of 32
ADP2105/ADP2106/ADP2107
Ensure that the maximum rms current of the inductor is greater
than the maximum load current, and the saturation current of
the inductor is greater than the peak current limit of the converter
used in the application.
Once RBOT is determined, calculate the value of the top resistor,
RTOP, by
⎡
⎤
VOUT −VFB
RTOP = RBOT
⎢
⎣
⎥
⎦
VFB
Table 5. Minimum Inductor Value for Common Output
The ADP2105/ADP2106/ADP2107-xx (where xx represents
the fixed output voltage) include the resistive voltage divider
internally, reducing the external circuitry required. Connect the
OUT_SENSE to the output voltage as close as possible to the
load for improved load regulation.
Voltage Options for the ADP2105 (1 A)
VIN
VOUT
2.7 V
3.6 V
4.2 V
5.5 V
1.67 μH
1.68 μH
2.02 μH
2.80 μH
3.70 μH
2.00 μH
2.19 μH
2.25 μH
2.80 μH
3.70 μH
2.14 μH
2.41 μH
2.57 μH
2.80 μH
3.70 μH
2.35 μH
2.73 μH
3.03 μH
3.41 μH
3.70 μH
1.2 V
1.5 V
1.8 V
2.5 V
3.3 V
INDUCTOR SELECTION
The high switching frequency of ADP2105/ADP2106/ADP2107
allows for minimal output voltage ripple even with small inductors.
The sizing of the inductor is a trade-off between efficiency and
transient response. A small inductor leads to larger inductor
current ripple that provides excellent transient response but
degrades efficiency. Due to the high switching frequency of
ADP2105/ADP2106/ADP2107, shielded ferrite core inductors
are recommended for their low core losses and low EMI.
Table 6. Minimum Inductor Value for Common Output
Voltage Options for the ADP2106 (1.5 A)
VIN
VOUT
2.7 V
3.6 V
4.2 V
5.5 V
1.11 μH
1.25 μH
1.49 μH
2.08 μH
2.74 μH
2.33 μH
1.46 μH
1.50 μH
2.08 μH
2.74 μH
2.43 μH
1.61 μH
1.71 μH
2.08 μH
2.74 μH
1.56 μH
1.82 μH
2.02 μH
2.27 μH
2.74 μH
1.2 V
1.5 V
1.8 V
2.5 V
3.3 V
As a guideline, the inductor peak-to-peak current ripple, ΔIL,
is typically set to 1/3 of the maximum load current for optimal
transient response and efficiency.
ILOAD(MAX)
VOUT × (VIN − VOUT
VIN × fSW × L
)
ΔIL
=
≈
3
Table 7. Minimum Inductor Value for Common Output
Voltage Options for the ADP2107 (2 A)
2.5×VOUT ×(VIN −VOUT
)
⇒ LIDEAL
=
ꢀH
VIN
VIN × ILOAD
(MAX)
VOUT
2.7 V
3.6 V
4.2 V
5.5 V
where fSW is the switching frequency (1.2 MHz).
0.83 μH
0.99 μH
1.19 μH
1.65 μH
2.18 μH
1.00 μH
1.09 μH
1.19 μH
1.65 μH
2.18 μH
1.07 μH
1.21 μH
1.29 μH
1.65 μH
2.18 μH
1.17 μH
1.36 μH
1.51 μH
1.70 μH
2.18 μH
1.2 V
1.5 V
1.8 V
2.5 V
3.3 V
The ADP2105/ADP2106/ADP2107 use slope compensation in
the current control loop to prevent subharmonic oscillations
when operating beyond 50% duty cycle. The fixed slope compen-
sation limits the minimum inductor value as a function of
output voltage.
Table 8. Inductor Recommendations for the ADP2105/
ADP2106/ADP2107
For the ADP2105:
L > (1.12 ꢀH/V) × VOUT
For the ADP2106:
Small-Sized Inductors Large-Sized Inductors
Vendor
( < 5 mm × 5 mm)
( > 5 mm × 5 mm)
Sumida
CDRH2D14, 3D16,
3D28
CDRH4D18, 4D22,
4D28, 5D18, 6D12
L > (0.83 ꢀH/V) × VOUT
For the ADP2107:
Toko
1069AS-DB3018,
1098AS-DE2812,
1070AS-DB3020
D52LC, D518LC,
D62LCB
L > (0.66 ꢀH/V) × VOUT
Also, 4.7 ꢀH or larger inductors are not recommended because
they may cause instability in discontinuous conduction mode
under light load conditions.
Coilcraft
LPS3015, LPS4012,
DO3314
DO1605T
Cooper
Bussmann
SD3110, SD3112,
SD3114, SD3118,
SD3812, SD3814
SD10, SD12, SD14, SD52
Finally, it is important that the inductor be capable of handling
the maximum peak inductor current, IPK, determined by the
following equation:
ΔI
2
⎛
⎜
⎝
⎞
⎟
⎠
L
IPK = ILOAD(MAX)
+
Rev. 0 | Page 16 of 32
ADP2105/ADP2106/ADP2107
It is also important, while choosing output capacitors, to
account for the loss of capacitance due to output voltage dc bias.
Figure 38 shows the loss of capacitance due to output voltage dc
bias for a few X5R MLCC capacitors from Murata.
OUTPUT CAPACITOR SELECTION
The output capacitor selection affects both the output voltage
ripple and the loop dynamics of the converter. For a given loop
crossover frequency (the frequency at which the loop gain
drops to 0 dB), the maximum voltage transient excursion
(overshoot) is inversely proportional to the value of the output
capacitor. Therefore, larger output capacitors result in improved
load transient response. To minimize the effects of the dc-to-dc
converter switching, the crossover frequency of the compensation
loop should be less than 1/10 of the switching frequency. Higher
crossover frequency leads to faster settling time for a load transient
response, but it can also cause ringing due to poor phase
margin. Lower crossover frequency helps to provide stable
operation but needs large output capacitors to achieve competitive
overshoot specifications. Therefore, the optimal crossover
frequency for the control loop of ADP2105/ADP2106/ADP2107
is 80 kHz, 1/15 of the switching frequency. For a crossover
frequency of 80 kHz, Figure 37 shows the maximum output
voltage excursion during a 1A load transient, as the product of
the output voltage and the output capacitor is varied. Choose
the output capacitor based on the desired load transient
response and target output voltage.
20
0
–20
1
–40
2
3
–60
–80
1
4.7µF 0805 X5R MURATA GRM21BR61A475K
2
10µF 0805 X5R MURATA GRM21BR61A106K
3
22µF 0805 X5R MURATA GRM21BR60J226M
–100
0
2
4
6
VOLTAGE (V
DC
)
Figure 38. % Drop-In Capacitance vs. DC Bias for Ceramic Capacitors
(Information Provided by Murata Corporation)
For example, to get 20 ꢀF output capacitance at an output voltage
of 2.5 V, based on Figure 38, as well as giving some margin for
temperature variance, it is suggested that a 22 ꢀF and a 10 ꢀF
capacitor be used in parallel to ensure that the output capacitance
is sufficient under all conditions for stable behavior.
18
17
16
15
14
13
12
11
10
9
Table 9. Recommended Input and Output Capacitor Selection
for the ADP2105/ADP2106/ADP2107
Vendor
8
Capacitor
Murata
Taiyo Yuden
7
6
5
4
4.7 μF 10 V
X5R 0805
GRM21BR61A475K
LMK212BJ475KG
3
2
1
0
10 μF 10 V
X5R 0805
GRM21BR61A106K
GRM21BR60J226M
LMK212BJ106KG
JMK212BJ226MG
22 μF 6.3 V
X5R 0805
15
20
25
30
35
40
45
50
55
60
65
70
OUTPUT CAPACITOR × OUTPUT VOLTAGE (μC)
INPUT CAPACITOR SELECTION
Figure 37. % Overshoot for a 1 A Load Transient Response vs.
Output Capacitor × Output Voltage
The input capacitor reduces input voltage ripple caused by the
switch currents on the PWIN pins. Place the input capacitors as
close as possible to the PWIN pins. Select an input capacitor
capable of withstanding the rms input current for the maximum
load current in your application.
For example, if the desired 1A load transient response (overshoot)
is 5% for an output voltage of 2.5 V, then from Figure 37
Output Capacitor × Output Voltage = 50 ꢀC
50 ꢀC
2.5
For the ADP2105, it is recommended that each PWIN pin be
bypassed with a 4.7 ꢀF or larger input capacitor. For the ADP2106,
bypass the PWIN pins with a 10 ꢀF and a 4.7 ꢀF capacitor, and
for the ADP2107, bypass each PWIN pin with a 10 ꢀF capacitor.
⇒ Output Capacitor =
≈ 20 ꢀF
The ADP2105/ADP2106/ADP2107 have been designed for
operation with small ceramic output capacitors that have low
ESR and ESL, thus comfortably able to meet tight output voltage
ripple specifications. X5R or X7R dialectrics are recommended
with a voltage rating of 6.3 V or 10 V. Y5V and Z5U dialectrics
are not recommended, due to their poor temperature and dc
bias characteristics. Table 9 shows a list of recommended MLCC
capacitors from Murata and Taiyo Yuden.
As with the output capacitor, a low ESR ceramic capacitor is
recommended to minimize input voltage ripple. X5R or X7R
dialectrics are recommended, with a voltage rating of 6.3 V or
10 V. Y5V and Z5U dialectrics are not recommended, due to
their poor temperature and dc bias characteristics. Refer to
Table 9 for input capacitor recommendations.
Rev. 0 | Page 17 of 32
ADP2105/ADP2106/ADP2107
The transconductance error amplifier drives the compensation
INPUT FILTER
network that consists of a resistor (RCOMP) and capacitor (CCOMP
connected in series to form a pole and a zero, as shown in the
following equation:
)
The IN pin is the power source for the ADP2105/ADP2106/
ADP2107 internal circuitry, including the voltage reference and
current sense amplifier that are sensitive to power supply noise.
To prevent high frequency switching noise on the PWIN pins from
corrupting the internal circuitry of the ADP2105/ADP2106/
ADP2107, a low-pass RC filter should be placed between the IN
pin and the PWIN1 pin. The suggested input filter consists of
a small 0.1 ꢀF ceramic capacitor placed between IN and AGND
and a 10 Ω resistor placed between IN and PWIN1. This forms
a 150 kHz low-pass filter between PWIN1 and IN that prevents
any high frequency noise on PWIN1 from coupling into the
IN pin.
⎛
⎞
⎟
⎟
⎠
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
1 + sRCOMPCCOMP
1
⎜
Z
COMP (s) = RCOMP
+
=
⎜
sCCOMP
sCCOMP
⎝
At the crossover frequency, the gain of the open loop transfer
function is unity. This yields the following equation for the
compensation network impedance at the crossover frequency:
⎛
⎜
⎜
⎝
⎞
⎛
⎟
⎜
⎜
⎟
⎝
⎠
⎞
⎟
⎟
⎠
(2π )FCROSS
GmGCS
COUTVOUT
VREF
Z
COMP (FCROSS ) =
where:
SOFT START
The ADP2105/ADP2106/ADP2107 include soft start circuitry
to limit the output voltage rise time to reduce inrush current at
startup. To set the soft start period, connect a soft start capacitor
(CSS) from SS to AGND. The soft start period varies linearly
with the size of the soft start capacitor, as shown in the
following equation:
F
CROSS = 80 kHz, the crossover frequency of the loop.
COUT OUT is determined from the Output Capacitor Selection
section.
V
To ensure that there is sufficient phase margin at the crossover
frequency, place the Compensator Zero at 1/4 of the crossover
frequency, as shown in the following equation:
T
SS = CSS × 109 ms
F
⎛
⎝
⎞
⎟
⎠
CROSS
(2π)
RCOMPCCOMP = 1
⎜
To get a soft start period of 1 ms, a 1 nF capacitor must be
connected between SS and AGND.
4
Solving the above two simultaneous equations yields the value
for the compensation resistor and compensation capacitor, as
shown in the following equation:
LOOP COMPENSATION
The ADP2105/ADP2106/ADP2107 utilize a transconductance
error amplifier to compensate the external voltage loop. The
open loop transfer function at angular frequency, s, is given by
⎛
⎜
⎜
⎝
⎞⎛
⎞
⎟
⎟
⎠
(2π)FCROSS COUTVOUT
⎟⎜
⎟⎜
RCOMP = 0.8
GmGCS
VREF
⎠⎝
⎛
⎜
⎜
⎝
⎞⎛
⎞
⎟
⎟
⎠
Z
sCOUT
COMP (s) VREF
2
⎟⎜
⎟⎜
H(s) = GmGCS
where:
CCOMP
=
VOUT
⎠⎝
πFCROSS RCOMP
V
V
REF is the internal reference voltage (0.8 V).
OUT is the nominal output voltage.
Z
COMP(s) is the impedance of the compensation network at the
angular frequency, s.
OUT is the output capacitor.
C
Gm is the transconductance of the error amplifier (50 ꢀA/V
nominal).
G
CS is the effective transconductance of the current loop.
G
G
G
CS = 1.875 A/V for the ADP2105.
CS = 2.8125 A/V for the ADP2106.
CS = 3.625 A/V for the ADP2107.
Rev. 0 | Page 18 of 32
ADP2105/ADP2106/ADP2107
BODE PLOTS
60
60
50
ADP2106
ADP2105
50
40
30
20
LOOP GAIN
0
40
0
LOOP GAIN
45
90
135
180
30
45
90
135
180
PHASE
MARGIN = 48°
PHASE
MARGIN = 49°
20
LOOP PHASE
10
0
10
LOOP PHASE
0
CROSSOVER
FREQUENCY = 87kHz
CROSSOVER
FREQUENCY = 79kHz
OUTPUT VOLTAGE = 1.8V
INPUT VOLTAGE = 5.5V
LOAD CURRENT = 1A
OUTPUT VOLTAGE = 1.2V
INPUT VOLTAGE = 5.5V
LOAD CURRENT = 1A
–10
–20
–30
–40
–10
–20
–30
–40
INDUCTOR = 2.2µH (LPS4012)
INDUCTOR = 3.3µH (SD3814)
OUTPUT CAPACITOR = 22µF + 22µF
COMPENSATION RESISTOR = 180kꢀ
COMPENSATION CAPACITOR = 56pF
OUTPUT CAPACITOR = 22µF + 22µF + 4.7µF
COMPENSATION RESISTOR = 267kꢀ
COMPENSATION CAPACITOR = 39pF
1
10
100
300
1
10
100
300
(kHz)
(kHz)
NOTES
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LOAD TRANSIENT.
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LOAD TRANSIENT.
Figure 39. ADP2106 Bode Plot at VIN = 5.5 V, VOUT = 1.8 V and Load = 1 A
Figure 42. ADP2105 Bode Plot at VIN = 5.5 V, VOUT = 1.2 V and Load = 1 A
60
60
ADP2106
ADP2107
50
50
40
30
0
40
30
0
LOOP GAIN
LOOP GAIN
45
90
135
180
45
90
135
180
PHASE
MARGIN = 65°
PHASE
MARGIN = 52°
20
20
LOOP PHASE
10
10
LOOP PHASE
0
0
CROSSOVER
FREQUENCY = 83kHz
CROSSOVER
FREQUENCY = 76kHz
OUTPUT VOLTAGE = 1.8V
INPUT VOLTAGE = 3.6V
LOAD CURRENT = 1A
OUTPUT VOLTAGE = 2.5V
INPUT VOLTAGE = 5V
LOAD CURRENT = 1A
–10
–20
–30
–40
–10
–20
–30
–40
INDUCTOR = 2.2µH (LPS4012)
INDUCTOR = 2µH (D62LCB)
OUTPUT CAPACITOR = 22µF + 22µF
COMPENSATION RESISTOR = 180kꢀ
COMPENSATION CAPACITOR = 56pF
OUTPUT CAPACITOR = 10µF + 4.7µF
COMPENSATION RESISTOR = 70kꢀ
COMPENSATION CAPACITOR = 120pF
1
10
100
300
1
10
100
300
(kHz)
(kHz)
NOTES
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LOAD TRANSIENT.
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
10% OVERSHOOT FOR A 1A LOAD TRANSIENT.
Figure 40. ADP2106 Bode Plot at VIN = 3.6 V, VOUT = 1.8 V, and Load = 1 A
Figure 43. ADP2107 Bode Plot at VIN = 5 V, VOUT = 2.5 V and Load = 1 A
60
60
ADP2105
ADP2107
50
50
LOOP GAIN
40
30
0
40
30
0
LOOP GAIN
45
90
135
180
45
90
135
180
PHASE
MARGIN = 70°
PHASE
MARGIN = 51°
20
20
LOOP PHASE
10
10
LOOP PHASE
0
0
CROSSOVER
FREQUENCY = 71kHz
CROSSOVER
FREQUENCY = 67kHz
OUTPUT VOLTAGE = 3.3V
INPUT VOLTAGE = 5V
LOAD CURRENT = 1A
OUTPUT VOLTAGE = 1.2V
INPUT VOLTAGE = 3.6V
LOAD CURRENT = 1A
–10
–20
–30
–40
–10
–20
–30
–40
INDUCTOR = 2.5µH (CDRH5D28)
INDUCTOR = 3.3µH (SD3814)
OUTPUT CAPACITOR = 10µF + 4.7µF
COMPENSATION RESISTOR = 70kꢀ
COMPENSATION CAPACITOR = 120pF
OUTPUT CAPACITOR = 22µF + 22µF + 4.7µF
COMPENSATION RESISTOR = 267kꢀ
COMPENSATION CAPACITOR = 39pF
1
10
100
300
1
10
100
300
(kHz)
(kHz)
NOTES
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LOAD TRANSIENT.
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
10% OVERSHOOT FOR A 1A LOAD TRANSIENT.
Figure 41. ADP2105 Bode Plot at VIN = 3.6 V, VOUT = 1.2 V, and Load = 1 A
Figure 44. ADP2107 Bode Plot at VIN = 5 V, VOUT = 3.3 V, and Load = 1 A
Rev. 0 | Page 19 of 32
ADP2105/ADP2106/ADP2107
LOAD TRANSIENT RESPONSE
OUTPUT CURRENT
3
OUTPUT CURRENT
3
2
CH2 LOW
–51mV
CH2 LOW
–93mV
OUTPUT VOLTAGE (AC-COUPLED)
OUTPUT VOLTAGE (AC-COUPLED)
2
1
1
LX NODE (SWITCH NODE)
LX NODE (SWITCH NODE)
CH2 50mV~ 10µs CH3 1A
CH1 2V
CH2 50mV~
M
10µs CH3 1A
A
CH3
0.5A
CH1 2V
M
A
CH3
0.5A
OUTPUT CAPACITOR: 22µF + 22µF + 4.7µF
INDUCTOR: SD14, 2.5µH
COMPENSATION RESISTOR: 270kꢀ
COMPENSATION CAPACITOR: 39pF
OUTPUT CAPACITOR: 22µF + 4.7µF
INDUCTOR: SD14, 2.5µH
COMPENSATION RESISTOR: 135kꢀ
COMPENSATION CAPACITOR: 82pF
Figure 45. 1 A Load Transient Response for ADP2105-1.2
with External Components Chosen for 5% Overshoot
Figure 48. 1 A Load Transient Response for ADP2105-1.2
with External Components Chosen for 10% Overshoot
OUTPUT CURRENT
OUTPUT CURRENT
3
2
3
CH2 LOW
–164mV
CH2 LOW
–112mV
2
OUTPUT VOLTAGE (AC-COUPLED)
OUTPUT VOLTAGE (AC-COUPLED)
1
1
LX NODE (SWITCH NODE)
CH2 100mV~ 10µs CH3 1A
LX NODE (SWITCH NODE)
CH2 100mV~ 10µs CH3 1A
CH1 2V
M
A
CH3
0.5A
CH1 2V
M
A
CH3
0.5A
OUTPUT CAPACITOR: 10µF + 10µF
INDUCTOR: SD3814, 3.3µH
COMPENSATION RESISTOR: 135kꢀ
COMPENSATION CAPACITOR: 82pF
OUTPUT CAPACITOR: 22µF + 22µF
INDUCTOR: SD3814, 3.3µH
COMPENSATION RESISTOR: 270kꢀ
COMPENSATION CAPACITOR: 39pF
Figure 49. 1 A Load Transient Response for ADP2105-1.8
with External Components Chosen for 10% Overshoot
Figure 46. 1 A Load Transient Response for ADP2105-1.8
with External Components Chosen for 5% Overshoot
OUTPUT CURRENT
OUTPUT CURRENT
3
2
3
2
CH2 LOW
–178mV
OUTPUT VOLTAGE (AC-COUPLED)
CH2 LOW
OUTPUT VOLTAGE (AC-COUPLED)
–308mV
1
1
LX NODE (SWITCH NODE)
LX NODE (SWITCH NODE)
CH1 2V
CH2 100mV~
M
10µs CH3 1A
A
CH3
0.5A
CH1 2V
CH2 200mV~
M
10µs CH3 1A
A
CH3
0.5A
OUTPUT CAPACITOR: 22µF + 4.7µF
INDUCTOR: CDRH5D18, 4.1µH
COMPENSATION RESISTOR: 270kꢀ
COMPENSATION CAPACITOR: 39pF
OUTPUT CAPACITOR: 10µF + 4.7µF
INDUCTOR: CDRH5D18, 4.1µH
COMPENSATION RESISTOR: 135kꢀ
COMPENSATION CAPACITOR: 82pF
Figure 47. 1 A Load Transient Response for ADP2105-3.3
with External Components Chosen for 5% Overshoot
Figure 50. 1 A Load Transient Response for ADP2105-3.3
with External Components Chosen for 10% Overshoot
Rev. 0 | Page 20 of 32
ADP2105/ADP2106/ADP2107
EFFICIENCY CONSIDERATIONS
The amount of power loss can by calculated by
SW = (CGATE − P + CGATE − N) × VIN2 × fSW
where:
(CGATE − P + CGATE − N) ~ 600 pF.
Efficiency is defined as the ratio of output power to input power.
The high efficiency of the ADP2105/ADP2106/ADP2107 has
two distinct advantages. First, only a small amount of power is
lost in the dc-to-dc converter package that reduces thermal
constraints. In addition, high efficiency delivers the maximum
output power for the given input power, extending battery life
in portable applications.
P
f
SW = 1.2 MHz, the switching frequency.
Transition Losses
There are four major sources of power loss in dc-to-dc
converters like the ADP2105/ADP2106/ADP2107.
Transition losses occur because the P-channel MOSFET power
switch cannot turn on or turn off instantaneously. At the middle of
a LX node transition, the power switch is providing all the inductor
current, while the source to drain voltage of the power switch is
half the input voltage, resulting in power loss. Transition losses
increase with load current and input voltage and occur twice for
each switching cycle.
•
•
•
•
Power switch conduction losses
Inductor losses
Switching losses
Transition losses
Power Switch Conduction Losses
Power switch conduction losses are caused by the flow of output
current through the P-channel power switch and the N-channel
The amount of power loss can be calculated by
VIN
P
=
× IOUT × (tON + tOFF) × fSW
TRAN
2
synchronous rectifier, which have internal resistances (RDS(ON)
associated with them. The amount of power loss can be approxi-
mated by
)
where tON and tOFF are the rise time and fall time of the LX node,
which are approximately 3 ns.
THERMAL CONSIDERATIONS
2
P
SW − COND = [RDS(ON) − P × D + RDS(ON) − N × (1 − D)] × IOUT
In most applications, the ADP2105/ADP2106/ADP2107 do not
dissipate a lot of heat due to their high efficiency. However, in
applications with high ambient temperature, low supply voltage,
and high duty cycle, the heat dissipated in the package is large
enough that it can cause the junction temperature of the die to
exceed the maximum junction temperature of 125°C. Once the
junction temperature exceeds 140°C, the converter goes into
thermal shutdown. It recovers only after the junction temperature
has decreased below 100°C to prevent any permanent damage.
Therefore, thermal analysis for the chosen application solution
is very important to guarantee reliable performance over all
conditions.
where D = VOUT/VIN.
The internal resistance of the power switches increases with
temperature but decreases with higher input voltage. Figure 19
in the Typical Performance Characteristics section shows the
change in RDS(ON) vs. input voltage, while Figure 27 in the
Typical Performance Characteristics section shows the change
in RDS(ON) vs. temperature for both power devices.
Inductor Losses
Inductor conduction losses are caused by the flow of current
through the inductor, which has an internal resistance (DCR)
associated with it. Larger sized inductors have smaller DCR,
which can improve inductor conduction losses.
The junction temperature of the die is the sum of the ambient
temperature of the environment and the temperature rise of the
package due to the power dissipation, as shown in the following
equation:
Inductor core losses are related to the magnetic permeability of
the core material. Because the ADP2105/ADP2106/ADP2107
are high switching frequency dc-to-dc converters, shielded ferrite
core material is recommended for its low core losses and low EMI.
TJ = TA + TR
where:
The total amount of inductor power loss can be calculated by
PL = DCR × IOUT2 + Core Losses
TJ is the junction temperature.
TA is the ambient temperature.
TR is the rise in temperature of the package due to power
dissipation in it.
Switching Losses
Switching losses are associated with the current drawn by the
driver to turn on and turn off the power devices at the
switching frequency. Each time a power device gate is turned on
and turned off, the driver transfers a charge ΔQ from the input
supply to the gate and then from the gate to ground.
Rev. 0 | Page 21 of 32
ADP2105/ADP2106/ADP2107
2. See whether the output voltage desired is available as a
fixed output voltage option. Because 2 V is not one of the
fixed output voltage options available, choose the adjustable
version of ADP2106.
The rise in temperature of the package is directly proportional
to the power dissipation in the package. The proportionality
constant for this relationship is defined as the thermal
resistance from the junction of the die to the ambient
temperature, as shown in the following equation:
3. The first step in external component selection for an
adjustable version converter is to calculate the resistance of
the resistive voltage divider that sets the output voltage.
TR = θJA × PD
where:
0.8 V
VFB
RBOT
=
=
= 40 kΩ
TR is the rise in temperature of the package.
PD is the power dissipation in the package.
θJA is the thermal resistance from the junction of the die to the
ambient temperature of the package.
ISTRING 20 ꢀA
⎡
⎢
⎤
⎥
⎡
⎢
⎣
⎤
⎥
⎦
2 V −0.8 V
VOUT −VFB
RTOP = RBOT
= 40 kΩ ×
= 60 kΩ
VFB
0.8 V
⎢
⎣
⎥
⎦
For example, consider an application where the ADP2107-1.8
is used with an input voltage of 3.6 V and a load current of 2 A.
Also, assume that the maximum ambient temperature is 85°C.
At a load current of 2 A, the most significant contributor of
power dissipation in the dc-to-dc converter package is the
conduction loss of the power switches. Using the graph of
switch resistance vs. temperature (see Figure 27), as well as the
equation of power loss given in the Power Switch Conduction
Losses section, the power dissipation in the package can be
calculated by
4. Calculate the minimum inductor value as follows:
For the ADP2106:
L > (0.83 ꢀH/V) × VOUT
Ö L > 0.83 ꢀH/V × 2 V
Ö L > 1.66 ꢀH
Next, calculate the ideal inductor value that sets the
inductor peak-to-peak current ripple, ΔIL, to1/3 of the
maximum load current at the maximum input voltage.
2
P
SW − COND = [RDS(ON) − P × D + RDS(ON) − N × (1 − D)] × IOUT
=
[109 mΩ × 0.5 + 90 mΩ × 0.5] × (2 A)2 ~ 400 mW
The θJA for the LFCSP_VQ package is 40°C/W, as shown in
Table 3. Thus, the rise in temperature of the package due to
power dissipation is
2.5×VOUT ×(VIN −VOUT
)
LIDEAL
=
ꢀH =
VIN × ILOAD
(MAX )
2.5×2 ×(4.2 − 2)
4.2×1.2
TR = θJA × PD = 40°C/W × 0.40 W = 16°C
The junction temperature of the converter is
TJ = TA + TR = 85°C + 16°C = 101°C
ꢀH = 2.18 ꢀH
The closest standard inductor value is 2.2 ꢀH. The
maximum rms current of the inductor should be greater
than 1.2 A, and the saturation current of the inductor
should be greater than 2 A. One inductor that meets these
criteria is the LPS4012-2.2 ꢀH from Coilcraft.
which is below the maximum junction temperature of 125°C.
Thus, this application operates reliably from a thermal point
of view.
5. Choose the output capacitor based on the transient
response requirements. The worst-case load transient is
1.2 A, for which the overshoot must be less than 100 mV,
which is 5% of the output voltage. Therefore, for a 1 A load
transient, the overshoot must be less than 4% of the output
voltage. For these conditions, Figure 37 gives
DESIGN EXAMPLE
Consider an application with the following specifications:
Input Voltage = 3.6 V to 4.2 V.
Output Voltage = 2 V.
Typical Output Current = 600 mA.
Maximum Output Current = 1.2 A.
Soft Start Time = 2 ms.
Output Capacitor × Output Voltage = 60 ꢀC
60 ꢀC
⇒ Output Capacitor =
≈ 30 ꢀF
Overshoot ≤ 100 mV under all load transient conditions.
2.0 V
1. Choose the dc-to-dc converter that satisfies the maximum
output current requirement. Because the maximum output
current for this application is 1.2 A, the ADP2106 with a
maximum output current of 1.5 A is ideal for this
application.
Next, taking into account the loss of capacitance due to dc
bias, as shown in Figure 38, two 22 ꢀF X5R MLCC capacitors
from Murata (GRM21BR60J226M) are sufficient for this
application.
Rev. 0 | Page 22 of 32
ADP2105/ADP2106/ADP2107
6. Because the ADP2106 is being used in this application, the
input capacitors are 10 ꢀF and 4.7 ꢀF X5R Murata capacitors
(GRM21BR61A106K and GRM21BR61A475K).
9. Finally, the compensation resistor and capacitor can be
calculated as
⎛
⎜
⎜
⎝
⎞⎛
⎞
⎟
⎟
⎠
(2π)FCROSS COUTVOUT
7. The input filter consists of a small 0.1 ꢀF ceramic capacitor
placed between IN and AGND and a 10 Ω resistor placed
between IN and PWIN1.
⎟⎜
⎟⎜
RCOMP = 0.8
GmGCS
VREF
⎠⎝
⎛
⎞⎛
⎟⎜
⎟⎜
⎠⎝
⎞
⎟
⎟
⎠
(2π) × 80 kHz
30 ꢀF × 2 V
⎜
⎜
⎝
= 0.8
= 215 kΩ
8. Choose a soft start capacitor of 2 nF to achieve a soft start
time of 2 ms.
50 ꢀA / V × 2.8125 A / V
0.8 V
2
2
CCOMP
=
=
= 39 pF
πFCROSSRCOMP π × 80 kHz × 215 kΩ
Rev. 0 | Page 23 of 32
ADP2105/ADP2106/ADP2107
EXTERNAL COMPONENT RECOMMENDATIONS
Table 10. Recommended External Components for Popular Output Voltage Options at 80 kHz Crossover Frequency with
10% Overshoot for a 1 A Load Transient (Refer to Figure 35 and Figure 36)
1
2
3
4
5
Part
VOUT (V)
0.9
1.2
1.5
1.8
2.5
3.3
0.9
1.2
1.5
1.8
2.5
3.3
0.9
1.2
1.5
1.8
2.5
3.3
1.2
1.5
1.8
3.3
1.2
1.5
1.8
3.3
1.2
1.5
1.8
3.3
CIN1 (μF)
CIN2 (μF)
COUT (μF)
L (μH)
2.0
2.5
3.0
3.3
3.6
4.1
1.5
1.8
2.0
2.2
2.5
3.0
1.2
1.5
1.5
1.8
1.8
2.5
2.5
3.0
3.3
4.1
1.8
2.0
2.2
3.0
1.5
1.5
1.8
2.5
RCOMP (kΩ)
135
135
135
135
135
135
90
CCOMP (pF)
82
RTOP (kΩ) RBOT (kΩ)
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2105-1.2
ADP2105-1.5
ADP2105-1.8
ADP2105-3.3
ADP2106-1.2
ADP2106-1.5
ADP2106-1.8
ADP2106-3.3
ADP2107-1.2
ADP2107-1.5
ADP2107-1.8
ADP2107-3.3
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
10
4.7
4.7
4.7
4.7
4.7
4.7
10
10
10
10
10
10
10
10
10
10
10
10
4.7
4.7
4.7
4.7
10
10
10
10
10
10
10
10
22 + 10
22 + 4.7
10 + 10
10 + 10
10 + 4.7
10 + 4.7
22 + 10
22 + 4.7
10 + 10
10 + 10
10 + 4.7
10 + 4.7
22 + 10
22 + 4.7
10 + 10
10 + 10
10 + 4.7
10 + 4.7
22 + 4.7
10 + 10
10 + 10
10 + 4.7
22 + 4.7
10 + 10
10 + 10
10 + 4.7
22 + 4.7
10 + 10
10 + 10
10 + 4.7
5
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
-
82
82
82
82
20
35
50
85
125
5
20
35
50
85
125
5
20
35
50
85
125
-
82
100
100
100
100
100
100
120
120
120
120
120
120
82
90
90
90
90
90
70
70
70
70
70
70
10
10
10
10
10
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
10
135
135
135
135
90
90
90
90
82
82
82
-
-
-
-
-
-
100
100
100
100
120
120
120
120
-
-
-
-
-
-
-
-
70
70
70
70
-
-
-
-
-
-
-
-
10
10
10
1 4.7 μF 0805 X5R 10 V Murata–GRM21BR61A475KA73L.
10 μF 0805 X5R 10 V Murata–GRM21BR61A106KE19L.
2 4.7 μF 0805 X5R 10 V Murata–GRM21BR61A475KA73L.
10 μF 0805 X5R 10 V Murata–GRM21BR61A106KE19L.
3 4.7 μF 0805 X5R 10 V Murata–GRM21BR61A475KA73L.
10 μF 0805 X5R 10 V Murata–GRM21BR61A106KE19L.
22 μF 0805 X5R 6.3 V Murata–GRM21BR60J226ME39L.
4 0.5% accuracy resistor.
5 0.5% accuracy resistor.
Rev. 0 | Page 24 of 32
ADP2105/ADP2106/ADP2107
Table 11. Recommended External Components for Popular Output Voltage Options at 80 kHz Crossover Frequency with
5% Overshoot for a 1 A Load Transient (Refer to Figure 35 and Figure 36)
2
CIN2
1
3
4
5
Part
VOUT (V)
0.9
1.2
1.5
1.8
2.5
3.3
0.9
1.2
1.5
1.8
2.5
3.3
0.9
1.2
1.5
1.8
2.5
3.3
1.2
1.5
1.8
3.3
1.2
1.5
1.8
3.3
1.2
1.5
1.8
3.3
CIN1 (μF)
(μF)
4.7
4.7
4.7
4.7
4.7
4.7
10
10
10
10
10
10
10
10
10
10
10
10
4.7
4.7
4.7
4.7
10
10
10
10
10
10
10
10
COUT (μF)
L (μH)
2.0
2.5
3.0
3.3
3.6
4.1
1.5
1.8
2.0
2.2
2.5
3.0
1.2
1.5
1.5
1.8
1.8
2.5
2.5
3.0
3.3
4.1
1.8
2.0
2.2
3.0
1.5
1.5
1.8
2.5
RCOMP (kΩ) CCOMP (pF)
RTOP (kΩ)
RBOT (kΩ)
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2105-1.2
ADP2105-1.5
ADP2105-1.8
ADP2105-3.3
ADP2106-1.2
ADP2106-1.5
ADP2106-1.8
ADP2106-3.3
ADP2107-1.2
ADP2107-1.5
ADP2107-1.8
ADP2107-3.3
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
10
22 + 22 + 22
22 + 22 + 4.7
22 + 22
22 + 22
22 + 10
270
270
270
270
270
270
180
180
180
180
180
180
140
140
140
140
140
140
270
270
270
270
180
180
180
180
140
140
140
140
39
39
39
39
39
39
56
56
56
56
56
56
68
68
68
68
68
68
39
39
39
39
56
56
56
56
68
68
68
68
5
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
-
20
35
50
85
125
5
20
35
50
85
125
5
20
35
50
85
125
-
22 + 4.7
22 + 22 + 22
22 + 22 + 4.7
22 + 22
22 + 22
22 + 10
22 + 4.7
22 + 22 + 22
22 + 22 + 4.7
22 + 22
22 + 22
22 + 10
10
10
10
10
10
22 + 4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
10
22 + 22 + 4.7
22 + 22
22 + 22
-
-
-
-
-
-
22 + 4.7
22 + 22 + 4.7
22 + 22
22 + 22
-
-
-
-
-
-
-
-
22 + 4.7
22 + 22 + 4.7
22 + 22
22 + 22
-
-
-
-
-
-
-
-
10
10
10
22 + 4.7
1 4.7μF 0805 X5R 10V Murata – GRM21BR61A475KA73L
10μF 0805 X5R 10V Murata – GRM21BR61A106KE19L
2 4.7μF 0805 X5R 10V Murata – GRM21BR61A475KA73L
10μF 0805 X5R 10V Murata – GRM21BR61A106KE19L
3 4.7μF 0805 X5R 10V Murata – GRM21BR61A475KA73L
10μF 0805 X5R 10V Murata – GRM21BR61A106KE19L
22μF 0805 X5R 6.3V Murata – GRM21BR60J226ME39L
4 0.5% Accuracy Resistor
5 0.5% Accuracy Resistor
Rev. 0 | Page 25 of 32
ADP2105/ADP2106/ADP2107
CIRCUIT BOARD LAYOUT RECOMMENDATIONS
Good circuit board layout is essential in obtaining the best
performance from the ADP2105/ADP2106/ADP2107. Poor
circuit layout degrades the output ripple, as well as the
electromagnetic interference (EMI) and electromagnetic
compatibility (EMC) performance.
Also, make the high current path from PGND pin of the
ADP2105/ADP2106/ADP2107 through L and COUT back
to the PGND plane as short as possible. To do this, ensure
that the PGND pin of the ADP2105/ADP2106/ADP2107
is tied to the PGND plane as close as possible to the input
and output capacitors.
Figure 52 and Figure 53 show the ideal circuit board layout for
the ADP2105/ADP2106/ADP2107. Use this layout to achieve
the highest performance. Refer to the following guidelines if
adjustments to the suggested layout are needed.
•
Place the feedback resistor divider network as close as
possible to the FB pin to prevent noise pickup. Try to
minimize the length of trace connecting the top of the
feedback resistor divider to the output while keeping away
from the high current traces and the switch node (LX) that
can lead to noise pickup. To reduce noise pickup, place an
analog ground plane on either side of the FB trace. For the
low fixed voltage options (1.2 V and 1.5 V), poor routing
of the OUT_SENSE trace can lead to noise pickup, adversely
affecting load regulation. This can be fixed by placing a 1 nF
bypass capacitor close to the OUT_SENSE pin.
•
Use separate analog and power ground planes. Connect
the ground reference of sensitive analog circuitry (such as
compensation and output voltage divider components) to
analog ground; connect the ground reference of power
components (such as input and output capacitors) to power
ground. In addition, connect both the ground planes to the
exposed pad of the ADP2105/ADP2106/ADP2107.
•
For each PWIN pin, place an input capacitor as close to the
PWIN pin as possible and connect the other end to the closest
power ground plane.
•
The placement and routing of the compensation components
are critical for proper behavior of the ADP2105/ADP2106/
ADP2107. The compensation components should be placed
as close to the COMP pin as possible. It is advisable to use
0402-sized compensation components for closer placement,
leading to smaller parasitics. Surround the compensation
components with analog ground plane to prevent noise
pickup. Also, ensure that the metal layer under the
•
•
Place the 0.1 ꢀF, 10 Ω low-pass input filter between the IN
pin and the PWIN1 pin, as close to the IN pin as possible.
Ensure that the high current loops are as short and as wide
as possible. Make the high current path from CIN through
L, COUT, and the PGND plane back to CIN as short as possible.
To accomplish this, ensure that the input and output
capacitors share a common PGND plane.
compensation components is the analog ground plane.
Rev. 0 | Page 26 of 32
ADP2105/ADP2106/ADP2107
EVALUATION BOARD
EVALUATION BOARD SCHEMATIC (ADP2107-1.8)
C7
0.1µF
VCC
R3
10ꢀ
INPUT VOLTAGE = 2.7V TO 5.5V
VIN
VCC
C1
1
10µF
OUT
16
GND
J1
U1
15
14
GND IN PWIN1
LX2
13
OUT_SENSE
EN
1
2
3
4
12
11
10
9
EN
R2
100kꢀ
GND
PGND
LX1
2
L1
ADP2107-1.8
2µH
OUTPUT VOLTAGE = 1.8V, 2A
1
2
GND
GND
V
OUT
VCC
R4
0ꢀ
C4
22µF
C3
1
1
22µF
PWIN2
OUT
C2
10µF
GND
COMP SS AGND PADDLE NC
17
1
5
6
7
8
R5
NS
R1
140kꢀ
1
2
MURATA X5R 0805
C6
68pF
C5
1nF
10μF: GRM21BR61A106KE19L
22μF: GRM21BR60J226ME39L
2μH INDUCTOR D62LCB TOKO
NC = NO CONNECT
Figure 51. Evaluation Board Schematic of the ADP2107-1.8 (Bold Traces Are High Current Paths)
RECOMMENDED PCB BOARD LAYOUT (EVALUATION BOARD LAYOUT)
JUMPER TO ENABLE
ENABLE
GROUND
V
IN
100kꢀ PULL-DOWN
GROUND
INPUT
INPUT CAPACITOR
CONNECT THE GROUND RETURN OF
ALL POWER COMPONENTS SUCH AS
INPUT AND OUTPUT CAPACITORS TO
THE POWER GROUND PLANE.
POWER GROUND
PLANE
PLACE THE FEEDBACK RESISTORS AS
CLOSE TO THE FB PIN AS POSSIBLE.
OUTPUT CAPACITOR
C
C
R
R
IN
OUT
TOP BOT
LX
OUTPUT
INDUCTOR (L)
PGND
LX
V
ADP2105/ADP2106/ADP2107
OUT
R
COMP
C
C
C
OUT
COMP
IN
OUTPUT CAPACITOR
C
SS
PLACE THE COMPENSATION
COMPONENTS AS CLOSE TO
THE COMP PIN AS POSSIBLE.
ANALOG GROUND PLANE
POWER GROUND
CONNECT THE GROUND RETURN OF ALL
SENSITIVE ANALOG CIRCUITRY SUCH AS
COMPENSATION AND OUTPUT VOLTAGE
DIVIDER TO THE ANALOG GROUND PLANE.
INPUT CAPACITOR
Figure 52. Recommended Layout of Top Layer of ADP2105/ADP2106/ADP2107
Rev. 0 | Page 27 of 32
ADP2105/ADP2106/ADP2107
ENABLE
V
GND
IN
GND
ANALOG GROUND PLANE
POWER GROUND PLANE
INPUT VOLTAGE PLANE
CONNECTING THE TWO
PWIN PINS AS CLOSE
AS POSSIBLE.
V
IN
V
OUT
CONNECT THE EXPOSED PAD OF
THE ADP2105/ADP2106/ADP2107
TO A LARGE GROUND PLANE TO
AID POWER DISSIPATION.
CONNECT THE PGND PIN
TO THE POWER GROUND
PLANE AS CLOSE TO THE
ADP2105/ADP2106/ADP2107
AS POSSIBLE.
FEEDBACK TRACE: THIS TRACE CONNECTS THE TOP OF THE
RESISTIVE VOLTAGE DIVIDER ON THE FB PIN TO THE OUTPUT.
PLACE THIS TRACE AS FAR AWAY FROM THE LX NODE AND HIGH
CURRENT TRACES AS POSSIBLE TO PREVENT NOISE PICKUP.
Figure 53. Recommended Layout of Bottom Layer of ADP2105/ADP2106/ADP2107
Rev. 0 | Page 28 of 32
ADP2105/ADP2106/ADP2107
APPLICATION CIRCUITS
0.1μF
V
INPUT VOLTAGE = 5V
10ꢀ
IN
1
10μF
V
OUT
16
15
14
GND IN PWIN1
LX2
13
OUT_SENSE
ON
1
2
3
4
12
EN
OFF
2
2.5μH
V
OUTPUT VOLTAGE = 3.3V
OUT
GND
PGND
LX1
11
10
9
1
1
10μF
4.7μF
ADP2107-3.3
LOAD
0A TO 2A
GND
V
IN
GND
PWIN2
1
1
2
10μF
MURATA X5R 0805
COMP SS AGND NC
10μF: GRM21BR61A106KE19L
4.7μF: GRM21BR61A475KA73L
SUMIDA CDRH5D28: 2.5μH
5
6
7
8
1nF
70kꢀ
120pF
NOTES
1. NC = NO CONNECT.
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 10% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
Figure 54. Application Circuit—VIN = 5 V, VOUT = 3.3 V, LOAD = 0 A to 2 A
0.1μF
V
INPUT VOLTAGE = 3.6V
10ꢀ
IN
1
10μF
V
OUT
16
15
14
GND IN PWIN1
LX2 12
13
OUT_SENSE
ON
1
2
3
4
EN
OFF
2
1.5μH
V
OUTPUT VOLTAGE = 1.5V
1
OUT
PGND
LX1
11
10
9
GND
1
22μF
22μF
ADP2107-1.5
LOAD
0A TO 2A
GND
V
IN
GND
PWIN2
1
1
2
10μF
MURATA X5R 0805
10μF: GRM21BR61A106KE19L
22μF: GRM21BR60J226ME39L
COMP SS AGND NC
5
6
7
8
TOKO D62LCB OR COILCRAFT LPS4012
1nF
140kꢀ
NOTES
1. NC = NO CONNECT.
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 5% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
68pF
Figure 55. Application Circuit—VIN = 3.6 V, VOUT = 1.5 V, LOAD = 0 A to 2 A
0.1μF
V
INPUT VOLTAGE = 2.7V TO 4.2V
10ꢀ
IN
1
4.7μF
V
OUT
16
15
14
GND IN PWIN1
LX2
13
OUT_SENSE
ON
1
2
3
4
12
EN
OFF
2
2.7μH
V
OUTPUT VOLTAGE = 1.8V
1
OUT
GND
PGND
LX1
11
10
9
1
22μF
22μF
ADP2105-1.8
LOAD
0A TO 1A
GND
V
IN
GND
PWIN2
1
1
2
4.7μF
MURATA X5R 0805
COMP SS AGND NC
4.7μF: GRM21BR61A475KA73L
22μF: GRM21BR60J226ME39L
TOKO 1098AS-DE2812: 2.7μH
5
6
7
8
1nF
270kꢀ
NOTES
1. NC = NO CONNECT.
39pF
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 5% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
Figure 56. Application Circuit—VIN = Li-Ion Battery, VOUT = 1.8 V, LOAD = 0 A to 1 A
Rev. 0 | Page 29 of 32
ADP2105/ADP2106/ADP2107
0.1μF
V
INPUT VOLTAGE = 2.7V TO 4.2V
10ꢀ
IN
1
4.7μF
V
OUT
16
15
14
GND IN PWIN1
LX2
13
OUT_SENSE
ON
1
2
3
4
12
EN
OFF
2
2.4μH
V
OUTPUT VOLTAGE = 1.2V
OUT
GND
PGND
11
1
1
22μF
4.7μF
ADP2105-1.2
LOAD
0A TO 1A
GND
GND
LX1 10
V
IN
PWIN2
9
1
1
2
4.7μF
MURATA X5R 0805
COMP SS AGND NC
5
4.7μF: GRM21BR61A475KA73L
22μF: GRM21BR60J226ME39L
TOKO 1069AS-DB3018HCT OR
TOKO 1070AS-DB3020HCT
6
7
8
1nF
135kꢀ
82pF
NOTES
1. NC = NO CONNECT.
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 10% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
Figure 57. Application Circuit—VIN = Li-Ion Battery, VOUT = 1.2 V, LOAD = 0 A to 1 A
0.1μF
V
INPUT VOLTAGE = 5V
10ꢀ
IN
1
10μF
FB
16
15
14
IN PWIN1
LX2 12
13
FB GND
ON
1
2
3
4
EN
OFF
2
2.5μH
OUTPUT VOLTAGE = 2.5V
PGND
11
10
9
GND
GND
GND
1 1
10μF 22μF
85kꢀ
ADP2106-ADJ
LOAD
0A TO 1.5A
LX1
FB
V
IN
40kꢀ
PWIN2
1
4.7μF
COMP SS AGND NC
5
6
7
8
1
2
MURATA X5R 0805
1nF
4.7μF: GRM21BR61A475KA73L
10μF: GRM21BR61A106KE19L
22μF: GRM21BR60J226ME39L
COILTRONICS SD14: 2.5μH
180kꢀ
56pF
NOTES
1. NC = NO CONNECT.
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 5% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
Figure 58. Application Circuit—VIN = 5 V, VOUT = 2.5 V, LOAD = 0 A to 1.5 A
Rev. 0 | Page 30 of 32
ADP2105/ADP2106/ADP2107
OUTLINE DIMENSIONS
4.00
0.60 MAX
(BOTTOM VIEW)
BSC SQ
0.60 MAX
0.65 BSC
PIN 1
INDICATOR
13
16
1
4
12
PIN 1
INDICATOR
2.25
2.10 SQ
1.95
TOP
VIEW
3.75
BSC SQ
EXPOSED
PAD
0.75
0.60
0.50
9
8
5
0.25 MIN
1.95 BSC
0.80 MAX
0.65 TYP
12° MAX
0.05 MAX
0.02 NOM
1.00
0.85
0.80
0.35
0.30
0.25
0.20 REF
COPLANARITY
0.08
SEATING
PLANE
COMPLIANT TO JEDEC STANDARDS MO-220-VGGC
Figure 59. 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
4 mm × 4 mm Body, Very Thin Quad
(CP-16-4)
Dimensions shown in millimeters
ORDERING GUIDE
Junction
Output
Current
Temperature
Range
Model
Output Voltage
Package Description
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
Evaluation Board
Evaluation Board
Evaluation Board
Evaluation Board
Evaluation Board
Evaluation Board
Package Option
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
ADP2105ACPZ-1.2-R71
ADP2105ACPZ-1.5-R71
ADP2105ACPZ-1.8-R71
ADP2105ACPZ-3.3-R71
ADP2105ACPZ-R71
ADP2106ACPZ-1.2-R71
ADP2106ACPZ-1.5-R71
ADP2106ACPZ-1.8-R71
ADP2106ACPZ-3.3-R71
ADP2106ACPZ-R71
ADP2107ACPZ-1.2-R71
ADP2107ACPZ-1.5-R71
ADP2107ACPZ-1.8-R71
ADP2107ACPZ-3.3-R71
ADP2107ACPZ-R71
ADP2105-1.8-EVAL
ADP2105-EVAL
1 A
1 A
1 A
1 A
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
1.2 V
1.5 V
1.8 V
3.3 V
1 A
ADJ
1.5 A
1.5 A
1.5 A
1.5 A
1.5 A
2 A
2 A
2 A
2 A
2 A
1.2 V
1.5 V
1.8 V
3.3 V
ADJ
1.2 V
1.5 V
1.8 V
3.3 V
ADJ
1.8 V
Adjustable, but set to 2.5 V
1.8 V
Adjustable, but set to 2.5 V
1.8 V
ADP2106-1.8-EVAL
ADP2106-EVAL
ADP2107-1.8-EVAL
ADP2107-EVAL
Adjustable, but set to 2.5 V
1 Z = Pb-free part.
Rev. 0 | Page 31 of 32
ADP2105/ADP2106/ADP2107
NOTES
©2006 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D06079-0-7/06(0)
Rev. 0 | Page 32 of 32
相关型号：
SI9130DB
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