LT8301 [ADI]
42VIN Micropower No-Opto Isolated Flyback Converter with 65V/3.6A Switch;
| 型号： | LT8301 |
| 厂家： | 
ADI |
| 描述： | 42VIN Micropower No-Opto Isolated Flyback Converter with 65V/3.6A Switch |
| 文件： | 总26页 (文件大小：1546K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LT8302/LT8302-3
42V Micropower No-Opto
IN
Isolated Flyback Converter
with 65V/3.6A Switch
FEATURES
DESCRIPTION
The LT®8302/LT8302-3 is a monolithic micropower iso-
lated flyback converter. By sampling the isolated output
voltage directly from the primary-side flyback waveform,
the part requires no third winding or opto-isolator for
regulation. The output voltage is programmed with two
external resistors and a third optional temperature com-
pensation resistor. Boundary mode operation provides a
small magnetic solution with excellent load regulation.
Low ripple Burst Mode operation maintains high efficiency
at light load while minimizing the output voltage ripple. A
3.6A, 65V DMOS power switch is integrated along with all
the high voltage circuitry and control logic into a thermally
enhanced 8-lead SO package.
n
3V to 42V Input Voltage Range
n
3.6A, 65V Internal DMOS Power Switch
n
Low Quiescent Current:
n
106µA in Sleep Mode
380µA in Active Mode
n
n
Quasi-Resonant Boundary Mode Operation at
Heavy Load
Low Ripple Burst Mode® Operation at Light Load
n
n
Minimum Load < 0.5% (Typ) of Full Output
n
No Transformer Third Winding or Opto-Isolator
Required for Output Voltage Regulation
n
Accurate EN/UVLO Threshold and Hysteresis
n
Internal Compensation and Soft-Start
n
Temperature Compensation for Output Diode
The LT8302/LT8302-3 operates from an input voltage
range of 3V to 42V and delivers up to 18W of isolated
output power. The high level of integration and the use of
boundary and low ripple burst modes result in a simple to
use, low component count, and high efficiency application
solution for isolated power delivery.
n
Output Short-Circuit Protection
n
Thermally Enhanced 8-Lead SO Package
n
AEC-Q100 Qualified for Automotive Applications
APPLICATIONS
n
All registered trademarks and trademarks are the property of their respective owners. Protected
by U.S. patents, including 5438499, 7463497, 7471522.
Isolated Automotive, Industrial, Medical
Power Supplies
n
Isolated Auxiliary/Housekeeping Power Supplies
TYPICAL APPLICATION
3V to 32VIN/5VOUT Isolated Flyback Converter
Efficiency vs Load Current
ꢒꢋ
+
V
V
IN
OUT
3V TO 32V
3:1
5V
470pF
9µH
ꢐꢑ
•
220µF
1µH
39Ω
V
IN
•
ꢐꢋ
10µF
–
V
EN/UVLO
SW
OUT
LT8302/LT8302-3
154k
ꢗꢑ
GND
INTV
R
FB
10mA TO 1.1A (V = 5V)
IN
10mA TO 2.0A (V = 12V)
IN
R
REF
ꢗꢋ
ꢘꢑ
ꢘꢋ
CC
10mA TO 2.9A (V = 24V)
IN
1µF
115k
10k
ꢙ
ꢙ
ꢙ
ꢚ ꢑꢙ
ꢚ ꢓꢔꢙ
ꢚ ꢔꢛꢙ
ꢍꢇ
ꢍꢇ
ꢍꢇ
TC
8302 TA01a
ꢔ.ꢋ
ꢀꢁꢂꢃ ꢄꢅRRꢆꢇꢈ ꢉꢂꢊ
ꢕ.ꢋ
ꢋ
ꢋ.ꢑ
ꢓ.ꢋ
ꢓ.ꢑ
ꢔ.ꢑ
ꢐꢕꢋꢔ ꢈꢂꢋꢓꢖ
Rev. G
1
Document Feedback
For more information www.analog.com
LT8302/LT8302-3
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
SW (Note 2)..............................................................65V
IN
ꢈꢉꢊ ꢋꢌꢍꢎ
V ............................................................................42V
EN/UVLO....................................................................V
ꢍꢓꢔꢕꢋꢖꢉ
ꢌꢓꢈꢋ
ꢀ
ꢁ
ꢂ
ꢃ
ꢄ
ꢅ
ꢆ
ꢇ
ꢈꢏ
IN
IN
R ........................................................V – 0.5V to V
R
Rꢍꢐ
R
ꢐꢑ
ꢏꢏ
FB
IN
ꢠ
Gꢓꢗ
Current Into R ....................................................200µA
ꢋ
ꢌꢓ
FB
INTV , R , TC.........................................................4V
Gꢓꢗ
ꢒꢎ
CC REF
Operating Junction Temperature Range (Notes 3, 4)
LT8302E, LT8302E-3 ......................... –40°C to 125°C
LT8302I, LT8302I-3 ........................... –40°C to 125°C
LT8302J, LT8302J-3.......................... –40°C to 150°C
LT8302H, LT8302H-3 ........................ –40°C to 150°C
LT8302MP ......................................... –55°C to 150°C
Storage Temperature Range .................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec)...................300°C
ꢒꢄꢍ ꢊꢘꢏꢙꢘGꢍ
ꢄꢚꢖꢍꢘꢗ ꢊꢖꢘꢒꢈꢌꢏ ꢒꢉ
θ
ꢛꢘ
ꢜ ꢂꢂꢝꢏꢔꢎ
ꢍꢞꢊꢉꢒꢍꢗ ꢊꢘꢗ ꢟꢊꢌꢓ ꢠꢡ ꢌꢒ Gꢓꢗꢢ ꢣꢕꢒꢈ ꢑꢍ ꢒꢉꢖꢗꢍRꢍꢗ ꢈꢉ ꢊꢏꢑ
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING* PACKAGE DESCRIPTION
TEMPERATURE RANGE
–40°C to 125°C
LT8302ES8E#PBF
LT8302ES8E#TRPBF
LT8302IS8E#TRPBF
LT8302JS8E#TRPBF
LT8302HS8E#TRPBF
LT8302MPS8E#TRPBF
LT8302ES8E-3#TRPBF
LT8302IS8E-3#TRPBF
LT8302JS8E-3#TRPBF
LT8302HS8E-3#TRPBF
8302
8-Lead Plastic SO
8-Lead Plastic SO
8-Lead Plastic SO
8-Lead Plastic SO
8-Lead Plastic SO
8-Lead Plastic SO
8-Lead Plastic SO
8-Lead Plastic SO
8-Lead Plastic SO
LT8302IS8E#PBF
8302
–40°C to 125°C
–40°C to 150°C
–40°C to 150°C
–55°C to 150°C
–40°C to 125°C
–40°C to 125°C
–40°C to 150°C
–40°C to 150°C
LT8302JS8E#PBF
8302
LT8302HS8E#PBF
8302
LT8302MPS8E#PBF
LT8302ES8E-3#PBF
LT8302IS8E-3#PBF
LT8302JS8E-3#PBF
LT8302HS8E-3#PBF
AUTOMOTIVE PRODUCTS**
LT8302ES8E#WPBF
LT8302IS8E#WPBF
LT8302JS8E#WPBF
LT8302HS8E#WPBF
LT8302ES8E-3#WPBF
LT8302IS8E-3#WPBF
LT8302JS8E-3#WPBF
LT8302HS8E-3#WPBF
8302
83023
83023
83023
83023
LT8302ES8E#WTRPBF
LT8302IS8E#WTRPBF
LT8302JS8E#WTRPBF
LT8302HS8E#WTRPBF
LT8302ES8E-3#WTRPBF
LT8302IS8E-3#WTRPBF
LT8302JS8E-3#WTRPBF
LT8302HS8E-3#WTRPBF
8302
8-Lead Plastic SO
8-Lead Plastic SO
8-Lead Plastic SO
8-Lead Plastic SO
8-Lead Plastic SO
8-Lead Plastic SO
8-Lead Plastic SO
8-Lead Plastic SO
–40°C to 125°C
–40°C to 125°C
–40°C to 150°C
–40°C to 150°C
–40°C to 125°C
–40°C to 125°C
–40°C to 150°C
–40°C to 150°C
8302
8302
8302
83023
83023
83023
83023
Contact the factory for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Tape and reel specifications. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix.
**Versions of this part are available with controlled manufacturing to support the quality and reliability requirements of automotive applications. These
models are designated with a #W suffix. Only the automotive grade products shown are available for use in automotive applications. Contact your
local Analog Devices account representative for specific product ordering information and to obtain the specific Automotive Reliability reports for
these models.
Rev. G
2
For more information www.analog.com
LT8302/LT8302-3
ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, VEN/UVLO = VIN, CINTVCC = 1µF to GND, unless otherwise noted.
SYMBOL PARAMETER
CONDITIONS
MIN
TYP
MAX
42
UNIT
l
V
IN
V
V
Voltage Range
3
V
IN
IN
I
Q
Quiescent Current
V
V
= 0.2V
= 1.1V
0.5
53
106
380
2
µA
µA
µA
µA
EN/UVLO
EN/UVLO
Sleep Mode (Switch Off)
Active Mode (Switch On)
l
l
l
EN/UVLO Shutdown Threshold
EN/UVLO Enable Threshold
EN/UVLO Enable Threshold
EN/UVLO Enable Hysteresis
EN/UVLO Hysteresis Current
For Lowest Off I
0.2
0.75
1.214
1.214
14
V
V
Q
Falling (E, I, H, MP Grades)
Falling (J Grade Only)
1.178
1.160
1.250
1.268
V
mV
I
V
V
V
= 0.3V
= 1.1V
= 1.3V
–0.1
2.3
–0.1
0
2.5
0
0.1
2.7
0.1
µA
µA
µA
HYS
EN/UVLO
EN/UVLO
EN/UVLO
V
INTV Regulation Voltage
I
= 0mA to 10mA
= 2.8V
INTVCC
2.85
10
3
3.1
20
V
mA
V
INTVCC
CC
INTVCC
I
INTV Current Limit
V
13
INTVCC
CC
INTV UVLO Threshold
Falling
2.39
2.47
105
2.55
CC
INTV UVLO Hysteresis
mV
mV
V
CC
(R – V ) Voltage
I = 75µA to 125µA
RFB
–50
0.98
50
FB
REF
REF
IN
l
R
R
Regulation Voltage
1.00
0
1.02
0.01
Regulation Voltage Line Regulation
3V ≤ V ≤ 42V
–0.01
%/V
V
IN
V
TC
TC Pin Voltage
TC Pin Current
1.00
I
TC
V
V
V
= 1.2V (LT8302)
= 1.2V (LT8302-3)
= 0.8V
12
7
15
10
–200
18
13
µA
µA
µA
TC
TC
TC
f
t
t
I
I
Minimum Switching Frequency
Minimum Switch-On Time
Maximum Switch-Off Time
Maximum Switch Current Limit
Minimum Switch Current Limit
Switch On-Resistance
11.3
12
160
170
4.5
0.87
80
12.7
kHz
ns
MIN
ON(MIN)
OFF(MAX)
SW(MAX)
SW(MIN)
Backup Timer
µs
3.6
5.4
A
0.70
1.04
A
R
I
SW
= 1.5A
= 65V
mΩ
µA
ms
DS(ON)
LKG
I
t
Switch Leakage Current
Soft-Start Timer
V
SW
0.1
11
0.5
SS
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
controls. The LT8302I/LT8302I-3 is guaranteed over the full –40°C to
125°C operating junction temperature range. The LT8302J/LT8302J-3
and LT8302H/LT8302H-3 are guaranteed over the full –40°C to 150°C
operating junction temperature range. The LT8302MP is guaranteed
over the full –55°C to 150°C operating junction temperature range. High
junction temperatures degrade operating lifetimes. Operating lifetime is
derated at junction temperature greater than 125°C.
Note 4: The LT8302/LT8302-3 includes overtemperature protection that
is intended to protect the devices during momentary overload conditions.
Junction temperature will exceed 150°C when overtemperature protection
is active. Continuous operation above the specified maximum operating
junction temperature may impair device reliability.
Note 2: The SW pin is rated to 65V for transients. Depending on the
leakage inductance voltage spike, operating waveforms of the SW pin
should be derated to keep the flyback voltage spike below 65V as shown
in Figure 5.
Note 3: The LT8302E/LT8302E-3 is guaranteed to meet performance
specifications from 0°C to 125°C junction temperature. Specifications
over the –40°C to 125°C operating junction temperature range are
assured by design, characterization and correlation with statistical process
Rev. G
3
For more information www.analog.com
LT8302/LT8302-3
TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C, unless otherwise noted.
Switching Frequency
vs Load Current
Output Load and Line Regulation
Output Temperature Variation
ꢎ.ꢒꢋ
ꢋ.ꢒ
ꢋ.ꢑ
ꢋ.ꢐ
ꢋ.ꢌ
ꢌꢋꢋ
ꢗꢋꢋ
ꢖꢋꢋ
ꢓꢋꢋ
ꢔꢋꢋ
ꢋ
ꢗRꢍꢘꢀ ꢃꢄGꢁ ꢄꢃꢃꢏꢙꢈꢄꢀꢙꢍꢘ
ꢍRꢁꢇꢈ ꢛꢂGꢆ ꢂꢛꢛꢀꢙꢄꢂꢈꢙꢁꢇ
ꢎ
ꢚ ꢐꢑꢎ
ꢎ.ꢏꢎ
ꢎ.ꢏꢋ
ꢙꢘ
ꢍꢅꢀ
ꢙ
ꢚ ꢐꢄ
ꢎ.ꢋꢎ
ꢎ.ꢋꢋ
ꢓ.ꢔꢎ
ꢓ.ꢔꢋ
ꢓ.ꢐꢎ
R
ꢚ ꢐꢐꢋꢛ
ꢀꢈ
R
ꢀꢈ
ꢚ ꢍꢃꢁꢘ
ꢕ.ꢖ
ꢕ.ꢔ
ꢕ.ꢓ
ꢍ
ꢍ
ꢍ
ꢖ ꢎꢍ
ꢖ ꢏꢒꢍ
ꢖ ꢒꢓꢍ
ꢘ
ꢙꢇ
ꢘ
ꢙꢇ
ꢘ
ꢙꢇ
ꢚ ꢌꢘ
ꢚ ꢔꢓꢘ
ꢚ ꢓꢗꢘ
ꢕꢇ
ꢕꢇ
ꢕꢇ
ꢓ.ꢐꢋ
ꢋ.ꢎ
ꢏ.ꢋ
ꢒ.ꢋ
ꢒ.ꢎ
ꢑ.ꢋ
ꢋꢌ
ꢐꢌꢌ ꢐꢑꢋ ꢐꢋꢌ
ꢋ
ꢏ.ꢎ
ꢊꢋꢌ ꢊꢑꢋ
ꢌ
ꢑꢋ
ꢓꢋ
ꢋ
ꢋ.ꢌ
ꢔ.ꢋ
ꢔ.ꢌ
ꢓ.ꢋ
ꢓ.ꢌ
ꢖ.ꢋ
ꢀꢁꢂꢃ ꢄꢅRRꢆꢇꢈ ꢉꢂꢊ
ꢀꢁꢂꢃꢁRꢄꢀꢅRꢁ ꢆꢇꢈꢉ
ꢀꢁꢂꢃ ꢄꢅRRꢆꢇꢈ ꢉꢂꢊ
ꢐꢑꢋꢒ Gꢋꢏ
ꢔꢒꢌꢑ Gꢌꢑ
ꢕꢖꢋꢓ Gꢋꢖ
Boundary Mode Waveforms
Discontinuous Mode Waveforms
Burst Mode Waveforms
ꢀ
ꢁꢂ
ꢀ
ꢁꢂ
ꢃꢄꢀꢅꢆꢇꢀ
ꢀ
ꢁꢂ
ꢃꢄꢀꢅꢆꢇꢀ
ꢃꢄꢀꢅꢆꢇꢀ
ꢀ
ꢀ
ꢈꢉꢊ
ꢀ
ꢈꢉꢊ
ꢈꢉꢊ
ꢋꢄꢌꢀꢅꢆꢇꢀ
ꢋꢄꢌꢀꢅꢆꢇꢀ
ꢋꢄꢌꢀꢅꢆꢇꢀ
ꢘꢙꢄꢃ Gꢄꢚ
ꢘꢙꢄꢃ Gꢄꢋ
ꢘꢙꢄꢃ Gꢄꢚ
ꢃꢍꢎꢅꢆꢇꢀ
ꢃꢍꢎꢅꢆꢇꢀ
ꢃꢄꢍꢎꢅꢆꢇꢀ
ꢏRꢈꢐꢊ ꢑꢒGꢓ ꢒꢑꢑꢔꢇꢕꢒꢊꢇꢈꢐ
ꢏRꢈꢐꢊ ꢑꢒGꢓ ꢒꢑꢑꢔꢇꢕꢒꢊꢇꢈꢐ
ꢏRꢈꢐꢊ ꢑꢒGꢓ ꢒꢑꢑꢔꢇꢕꢒꢊꢇꢈꢐ
ꢀ
ꢖ ꢗꢃꢀ
ꢀ
ꢖ ꢗꢃꢀ
ꢀ
ꢖ ꢗꢃꢀ
ꢇꢐ
ꢇꢐ
ꢇꢐ
ꢇ
ꢖ ꢃꢒ
ꢇ
ꢖ ꢄ.ꢋꢒ
ꢇ
ꢖ ꢗꢄꢌꢒ
ꢈꢉꢊ
ꢈꢉꢊ
ꢈꢉꢊ
VIN Quiescent Current,
Sleep Mode
VIN Quiescent Current,
Active Mode
VIN Shutdown Current
ꢋꢎꢅ
ꢋꢍꢅ
ꢋꢌꢅ
ꢋꢅꢅ
ꢉꢊꢅ
ꢉꢎꢅ
ꢉꢋꢅ
ꢉꢌꢅ
ꢋꢅ
ꢊ
ꢑ
ꢑ
ꢑ
ꢓ ꢋꢐꢅꢔꢕ
ꢒ
ꢒ
ꢒ
ꢓ ꢎꢐꢔꢕ
ꢑ
ꢓ ꢏꢐꢅꢔꢕ
ꢒ
ꢐ
ꢒ ꢋꢏꢅꢓꢔ
ꢑ
ꢓ ꢖꢐꢐꢔꢕ
ꢋꢌꢅ
ꢋꢋꢅ
ꢑ
ꢓ ꢌꢐꢔꢕ
ꢒ
ꢉ
ꢐ
ꢒ ꢌꢏꢓꢔ
ꢑ
ꢑ
ꢓ ꢖꢐꢐꢔꢕ
ꢒ
ꢌ
ꢐ
ꢒ ꢕꢏꢏꢓꢔ
ꢑ
ꢋꢅꢅ
ꢊꢅ
ꢎ
ꢆꢅ
ꢅ
ꢅ
ꢋꢅ
ꢌꢅ
ꢍꢅ
ꢎꢅ
ꢏꢅ
ꢅ
ꢏꢅ
ꢌꢅ
ꢉꢅ
ꢋꢅ
ꢐꢅ
ꢅ
ꢋꢅ
ꢎꢅ
ꢍꢅ
ꢃꢀꢄ
ꢌꢅ
ꢐꢅ
ꢀ
ꢃꢀꢄ
ꢀ
ꢃꢀꢄ
ꢀ
ꢁꢂ
ꢁꢂ
ꢁꢂ
ꢆꢍꢅꢌ Gꢅꢆ
ꢊꢉꢅꢌ Gꢅꢍ
ꢊꢍꢅꢎ Gꢅꢏ
Rev. G
4
For more information www.analog.com
LT8302/LT8302-3
TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C, unless otherwise noted.
EN/UVLO Enable Threshold
EN/UVLO Hysteresis Current
INTVCC Voltage vs Temperature
ꢒ.ꢓꢗꢌ
ꢒ.ꢓꢔꢋ
ꢒ.ꢓꢔꢌ
ꢒ.ꢓꢓꢋ
ꢒ.ꢓꢓꢌ
ꢒ.ꢓꢒꢋ
ꢒ.ꢓꢒꢌ
ꢒ.ꢓꢌꢋ
ꢒ.ꢓꢌꢌ
ꢋ
ꢓ.ꢔꢌ
ꢓ.ꢌꢋ
ꢓ.ꢌꢌ
ꢍ.ꢒꢋ
ꢍ.ꢒꢌ
ꢍ.ꢎꢋ
ꢍ.ꢎꢌ
RꢘꢙꢘꢎG
ꢓ
ꢒ
ꢐ
ꢖ ꢌꢗꢄ
ꢐꢑꢀꢏꢈꢈ
ꢐ
ꢖ ꢔꢌꢗꢄ
ꢐꢑꢀꢏꢈꢈ
ꢚꢄꢐꢐꢘꢎG
ꢔ
ꢖ
ꢌ
ꢋꢌ ꢖꢋ
ꢀꢁꢂꢃꢁRꢄꢀꢅRꢁ ꢆꢇꢈꢉ
ꢊꢋꢌ ꢊꢓꢋ
ꢌ
ꢓꢋ
ꢒꢌꢌ ꢒꢓꢋ ꢒꢋꢌ
ꢕꢋ ꢔꢌꢌ
ꢔꢍꢋ ꢔꢋꢌ
ꢋꢌ
ꢀꢁꢂꢃꢁRꢄꢀꢅRꢁ ꢆꢇꢈꢉ
ꢊꢋꢌ ꢊꢍꢋ
ꢌ
ꢍꢋ ꢋꢌ
ꢊꢋꢌ ꢊꢔꢋ
ꢌ
ꢔꢋ
ꢕꢋ ꢖꢌꢌ ꢖꢔꢋ ꢖꢋꢌ
ꢀꢁꢂꢃꢁRꢄꢀꢅRꢁ ꢆꢇꢈꢉ
ꢕꢔꢌꢓ Gꢒꢌ
ꢗꢒꢌꢔ Gꢖꢖ
ꢎꢓꢌꢍ Gꢔꢍ
INTVCC Voltage vs VIN
INTVCC UVLO Threshold
(RFB-VIN) Voltage
ꢊ.ꢎꢋ
ꢊ.ꢋꢅ
ꢊ.ꢋꢋ
ꢈ.ꢉꢅ
ꢈ.ꢉꢋ
ꢈ.ꢍꢅ
ꢈ.ꢍꢋ
ꢍ.ꢕ
ꢍ.ꢖ
ꢍ.ꢓ
ꢍ.ꢋ
ꢍ.ꢒ
ꢍ.ꢑ
ꢍ.ꢍ
ꢕꢌ
ꢔꢌ
ꢗ
ꢚ ꢒꢑꢋꢛꢄ
Rꢘꢙ
ꢑꢌ
ꢁ
ꢏ ꢋꢐꢑ
RꢏꢙꢏꢐG
ꢁꢂꢆꢀꢇꢇ
ꢒꢌ
ꢗ
ꢚ ꢒꢌꢌꢛꢄ
Rꢘꢙ
ꢌ
ꢗꢄꢘꢘꢏꢐG
ꢁ
ꢏ ꢎꢋꢐꢑ
ꢁꢂꢆꢀꢇꢇ
ꢊꢒꢌ
ꢊꢑꢌ
ꢊꢔꢌ
ꢊꢕꢌ
ꢗ
ꢚ ꢖꢋꢛꢄ
Rꢘꢙ
ꢖꢋ ꢔꢌꢌ
ꢎꢋ ꢎꢅ
ꢌꢅ
ꢊꢋꢌ ꢊꢍꢋ
ꢌ
ꢍꢋ ꢋꢌ
ꢔꢍꢋ ꢔꢋꢌ
ꢋꢌ ꢖꢋ
ꢅ
ꢈꢋ ꢈꢅ ꢊꢋ ꢊꢅ ꢌꢋ
ꢃꢀꢄ
ꢊꢋꢌ ꢊꢑꢋ
ꢌ
ꢑꢋ
ꢒꢌꢌ ꢒꢑꢋ
ꢒꢋꢌ
ꢀꢁꢂꢃꢁRꢄꢀꢅRꢁ ꢆꢇꢈꢉ
ꢀ
ꢀꢁꢂꢃꢁRꢄꢀꢅRꢁ ꢆꢇꢈꢉ
ꢁꢂ
ꢍꢊꢋꢈ Gꢎꢊ
ꢕꢑꢌꢍ Gꢔꢒ
ꢓꢔꢌꢑ Gꢒꢋ
RREF Regulation Voltage
RREF Line Regulation
TC Pin Voltage
ꢓ.ꢌꢓꢌ
ꢓ.ꢌꢌꢒ
ꢓ.ꢌꢌꢑ
ꢓ.ꢌꢌꢔ
ꢓ.ꢌꢌꢐ
ꢓ.ꢌꢌꢌ
ꢌ.ꢍꢍꢒ
ꢌ.ꢍꢍꢑ
ꢌ.ꢍꢍꢔ
ꢌ.ꢍꢍꢐ
ꢌ.ꢍꢍꢌ
ꢈ.ꢅꢈꢅ
ꢈ.ꢅꢅꢌ
ꢈ.ꢅꢅꢊ
ꢈ.ꢅꢅꢋ
ꢈ.ꢅꢅꢉ
ꢈ.ꢅꢅꢅ
ꢅ.ꢏꢏꢌ
ꢅ.ꢏꢏꢊ
ꢅ.ꢏꢏꢋ
ꢅ.ꢏꢏꢉ
ꢅ.ꢏꢏꢅ
ꢎ.ꢋ
ꢎ.ꢔ
ꢎ.ꢏ
ꢎ.ꢑ
ꢎ.ꢎ
ꢎ.ꢌ
ꢌ.ꢒ
ꢌ.ꢐ
ꢌ.ꢓ
ꢊꢋꢌ
ꢋꢌ
ꢓꢌꢌ ꢓꢐꢋ
ꢅ
ꢈꢅ
ꢍꢅ
ꢃꢀꢄ
ꢋꢅ
ꢐꢅ
ꢋꢌ ꢓꢋ
ꢀꢁꢂꢃꢁRꢄꢀꢅRꢁ ꢆꢇꢈꢉ
ꢊꢐꢋ
ꢌ
ꢐꢋ
ꢕꢋ
ꢓꢋꢌ
ꢉꢅ
ꢊꢋꢌ ꢊꢑꢋ
ꢌ
ꢑꢋ
ꢎꢌꢌ ꢎꢑꢋ ꢎꢋꢌ
ꢀꢁꢂꢃꢁRꢄꢀꢅRꢁ ꢆꢇꢈꢉ
ꢀ
ꢁꢂ
ꢒꢖꢌꢐ Gꢓꢑ
ꢌꢍꢅꢉ Gꢈꢎ
ꢐꢏꢌꢑ Gꢎꢐ
Rev. G
5
For more information www.analog.com
LT8302/LT8302-3
TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C, unless otherwise noted.
RDS(ON)
Switch Current Limit
Maximum Switching Frequency
200
ꢋꢌꢌ
ꢋ
ꢂꢄꢖꢍꢂꢅꢂ ꢈꢅRRꢁꢗꢀ ꢘꢍꢂꢍꢀ
160
120
ꢕꢌꢌ
ꢔꢌꢌ
ꢑ
ꢐ
80
40
0
ꢖꢌꢌ
ꢘꢌꢌ
ꢌ
ꢒ
ꢔ
ꢌ
ꢂꢍꢗꢍꢂꢅꢂ ꢈꢅRRꢁꢗꢀ ꢘꢍꢂꢍꢀ
50
TEMPERATURE (°C)
ꢋꢌ
ꢀꢁꢂꢃꢁRꢄꢀꢅRꢁ ꢆꢇꢈꢉ
–50 –25
0
25
75 100 125 150
ꢋꢌ
ꢀꢁꢂꢃꢁRꢄꢀꢅRꢁ ꢆꢇꢈꢉ
ꢊꢋꢌ ꢊꢖꢋ
ꢌ
ꢖꢋ
ꢗꢋ ꢘꢌꢌ ꢘꢖꢋ ꢘꢋꢌ
ꢊꢋꢌ ꢊꢒꢋ
ꢌ
ꢒꢋ
ꢓꢋ ꢔꢌꢌ ꢔꢒꢋ ꢔꢋꢌ
8302 G19
ꢙꢔꢌꢖ Gꢖꢘ
ꢕꢐꢌꢒ Gꢒꢌ
Minimum Switching Frequency
Minimum Switch-On Time
Minimum Switch-Off Time
ꢕꢌ
400
300
200
100
400
300
200
100
ꢔꢖ
ꢔꢕ
ꢘ
ꢚ
ꢌ
0
0
ꢋꢌ
ꢀꢁꢂꢃꢁRꢄꢀꢅRꢁ ꢆꢇꢈꢉ
ꢊꢋꢌ ꢊꢕꢋ
ꢌ
ꢕꢋ
ꢗꢋ ꢔꢌꢌ ꢔꢕꢋ ꢔꢋꢌ
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
ꢘꢙꢌꢕ Gꢕꢕ
8302 G23
8302 G24
Rev. G
6
For more information www.analog.com
LT8302/LT8302-3
PIN FUNCTIONS
EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The
EN/UVLO pin is used to enable the LT8302/LT8302-3.
Pull the pin below 0.3V to shut down the LT8302/LT8302-
3. This pin has an accurate 1.214V threshold and can
SW (Pin 5): Drain of the Internal DMOS Power Switch.
Minimize trace area at this pin to reduce EMI and voltage
spikes.
RFB (Pin 6): Input Pin for External Feedback Resistor.
be used to program a V undervoltage lockout (UVLO)
IN
Connect a resistor from this pin to the transformer pri-
threshold using a resistor divider from V to ground. A
IN
mary SW pin. The ratio of the R resistor to the R
FB
REF
2.5µA current hysteresis allows the programming of V
IN
resistor, times the internal voltage reference, determines
the output voltage (plus the effect of any non-unity trans-
former turns ratio). Minimize trace area at this pin.
UVLO hysteresis. If neither function is used, tie this pin
directly to V .
IN
INTV (Pin 2): Internal 3V Linear Regulator Output. The
CC
RREF (Pin 7): Input Pin for External Ground Referred
Reference Resistor. The resistor at this pin should be in
the range of 10k, but for convenience in selecting a resis-
tor divider ratio, the value may range from 9.09k to 11.0k.
INTV pin is supplied from VIN and powers the inter-
CC
nal control circuitry and gate driver. Do not overdrive the
INTV pin with any external supply, such as a third wind-
CC
ing supply. Locally bypass this pin to ground with a mini-
TC (Pin 8): Output Voltage Temperature Compensation.
The voltage at this pin is proportional to absolute tem-
perature (PTAT) with temperature coefficient equal to
3.35mV/°K, i.e., equal to 1V at room temperature 25°C.
The TC pin voltage can be used to estimate the LT8302/
LT8302-3 junction temperature. Connect a resistor from
mum 1µF ceramic capacitor.
V (Pin 3): Input Supply. The V pin supplies current to
IN
IN
the internal circuitry and serves as a reference voltage for
the feedback circuitry connected to the R pin. Locally
FB
bypass this pin to ground with a capacitor.
GND (Pin 4, Exposed Pad Pin 9): Ground. The exposed
pad provides both electrical contact to ground and good
thermal contact to the printed circuit board. Solder the
exposed pad directly to the ground plane.
this pin to the R
pin to compensate the output diode
REF
temperature coefficient.
Rev. G
7
For more information www.analog.com
LT8302/LT8302-3
BLOCK DIAGRAM
ꢅ
ꢗꢙꢊ
ꢊꢌ
ꢉꢍꢌ
ꢁ
ꢀ
ꢇ
ꢗꢙꢊ
ꢋ
ꢗꢙꢊ
ꢇ
ꢗꢙꢊ
ꢇ
ꢆꢉ
ꢋ
ꢆꢉ
ꢧ
ꢘꢌꢎ
ꢘꢌꢑ
ꢧ
R
ꢒꢑ
ꢂ
ꢡ
ꢢ
ꢇ
R
ꢏꢣ
ꢆꢉ
ꢒꢑ
ꢆꢉꢊꢇ
ꢋꢋ
ꢇ
ꢆꢉ
ꢃ
ꢌ
ꢘꢅꢗ
ꢌꢍꢕ
ꢋ
ꢆꢉꢊꢇꢋꢋ
ꢖꢂ
ꢖꢃ
ꢏꢊꢎRꢊꢛꢙ ꢝ
RꢈꢒꢈRꢈꢉꢋꢈꢝ
ꢋꢗꢉꢊRꢗꢘ
ꢑꢗꢙꢉꢅꢎRꢚ
ꢅꢈꢊꢈꢋꢊꢗR
ꢃꢢꢤꢎ
R
ꢗꢏꢋꢆꢘꢘꢎꢊꢗR
ꢈꢉꢌ
ꢈꢉꢃ
ꢆꢉꢊꢇ
ꢋꢋ
ꢀ
ꢁ
ꢈꢉꢥꢙꢇꢘꢗ
ꢌ.ꢃꢌꢕꢇ
ꢀ
ꢁ
R
ꢀ
ꢓ
ꢔ
ꢏ
ꢁ
R
ꢠ
ꢖꢌ
ꢎꢂ
ꢌꢇ
ꢅRꢆꢇꢈR
ꢎꢌ
ꢃ.ꢢꢤꢎ
ꢁ
ꢀ
ꢜꢊꢎꢊ
ꢇꢗꢘꢊꢎGꢈ
ꢖꢕ
ꢎꢃ
R
ꢏꢈꢉꢏꢈ
Gꢉꢅ
ꢕꢝ ꢈꢞꢜꢗꢏꢈꢅ ꢜꢎꢅ ꢜꢆꢉ ꢟ
R
ꢊꢋ
ꢄ
Rꢈꢒ
ꢦ
R
ꢊꢋ
ꢄꢂꢐꢃ ꢑꢅ
R
Rꢈꢒ
Rev. G
8
For more information www.analog.com
LT8302/LT8302-3
OPERATION
The LT8302/LT8302-3 is a current mode switching regula-
tor IC designed specially for the isolated flyback topology.
The key problem in isolated topologies is how to commu-
nicate the output voltage information from the isolated
secondary side of the transformer to the primary side
for regulation. Historically, opto-isolators or extra trans-
former windings communicate this information across
the isolation boundary. Opto-isolator circuits waste output
power, and the extra components increase the cost and
physical size of the power supply. Opto-isolators can also
cause system issues due to limited dynamic response,
nonlinearity, unit-to-unit variation and aging over life-
time. Circuits employing extra transformer windings also
exhibit deficiencies, as using an extra winding adds to
the transformer’s physical size and cost, and dynamic
response is often mediocre.
provides easy output diode temperature compensation.
By integrating the loop compensation and soft-start
inside, the part reduces the number of external compo-
nents. As shown in the Block Diagram, many of the blocks
are similar to those found in traditional switching reg-
ulators including reference, regulators, oscillator, logic,
current amplifier, current comparator, driver, and power
switch. The novel sections include a flyback pulse sense
circuit, a sample-and-hold error amplifier, and a boundary
mode detector, as well as the additional logic for boundary
conduction mode, discontinuous conduction mode, and
low ripple Burst Mode operation.
Quasi-Resonant Boundary Mode Operation
The LT8302/LT8302-3 features quasi-resonant bound-
ary conduction mode operation at heavy load, where
the chip turns on the primary power switch when the
secondary current is zero and the SW rings to its valley.
Boundary conduction mode is a variable frequency, vari-
able peak-current switching scheme. The power switch
turns on and the transformer primary current increases
until an internally controlled peak current limit. After the
power switch turns off, the voltage on the SW pin rises to
the output voltage multiplied by the primary-to-secondary
transformer turns ratio plus the input voltage. When the
secondary current through the output diode falls to zero,
the SW pin voltage collapses and rings around VIN. A
boundary mode detector senses this event and turns the
power switch back on at its valley.
The LT8302/LT8302-3 samples the isolated output voltage
through the primary-side flyback pulse waveform. In this
manner, neither opto-isolator nor extra transformer wind-
ing is required for regulation. Since the LT8302/LT8302-3
operates in either boundary conduction mode or discon-
tinuous conduction mode, the output voltage is always
sampled on the SW pin when the secondary current is
zero. This method improves load regulation without the
need of external load compensation components.
The LT8302/LT8302-3 is a simple to use micropower iso-
lated flyback converter housed in a thermally enhanced
8-lead SO package. The output voltage is programmed
with two external resistors. An optional TC resistor
Rev. G
9
For more information www.analog.com
LT8302/LT8302-3
OPERATION
Boundary conduction mode returns the secondary current
to zero every cycle, so parasitic resistive voltage drops
do not cause load regulation errors. Boundary conduc-
tion mode also allows the use of smaller transformers
compared to continuous conduction mode and does not
exhibit subharmonic oscillation.
minimum switch current limit and minimum switch-off
time are necessary to guarantee the correct operation of
specific applications.
As the load gets very light, the LT8302/LT8302-3 starts to
fold back the switching frequency while keeping the min-
imum switch current limit. So the load current is able to
decrease while still allowing minimum switch-off time for
the sample-and-hold error amplifier. Meanwhile, the part
switches between sleep mode and active mode, thereby
reducing the effective quiescent current to improve light
load efficiency. In this condition, the LT8302/LT8302-3
runs in low ripple Burst Mode operation. The typical
12kHz minimum switching frequency determines how
often the output voltage is sampled and also the mini-
mum load requirement.
Discontinuous Conduction Mode Operation
As the load gets lighter, boundary conduction mode
increases the switching frequency and decreases the
switch peak current at the same ratio. Running at a higher
switching frequency up to several MHz increases switch-
ing and gate charge losses. To avoid this scenario, the
LT8302/LT8302-3 has an additional internal oscillator,
which clamps the maximum switching frequency to be
less than 380kHz. Once the switching frequency hits the
internal frequency clamp, the part starts to delay the switch
turn-on and operates in discontinuous conduction mode.
Difference Between LT8302 and LT8302-3
The difference between LT8302 and LT8302-3 is
the boundary detection method. The LT8302 is using the
dv/dt slope on R pin, while the LT8302-3 is using the
Low Ripple Burst Mode Operation
voltage level onRREF pin. For good transformers with
Unlike traditional flyback converters, the LT8302/
LT8302-3 has to turn on and off at least for a minimum
amount of time and with a minimum frequency to allow
accurate sampling of the output voltage. The inherent
REF
low leakage inductance, both the LT8302 and LT8302-3
are behaving the same. The LT8302-3 is recommended
for multiple-winding output applications due to its lower
sensitivity to the noise on R pin.
REF
Rev. G
10
For more information www.analog.com
LT8302/LT8302-3
APPLICATIONS INFORMATION
Output Voltage
Combination with the previous V
equation yields an
FLBK
equation for V , in terms of the R and R resistors,
OUT
FB
REF
The RFB and RREF resistors as depicted in the Block
Diagram are external resistors used to program the out-
put voltage. The LT8302/LT8302-3 operates similar to
traditional current mode switchers, except in the use of a
unique flyback pulse sense circuit and a sample-and-hold
error amplifier, which sample and therefore regulate the
isolated output voltage from the flyback pulse.
transformer turns ratio, and diode forward voltage:
⎛
⎜
⎝
⎞
⎟
⎠
⎛
⎜
⎝
⎞
⎟
⎠
RFB
1
N
PS
VOUT = VREF
•
•
– V
F
R
REF
Output Temperature Compensation
The first term in the V
equation does not have tem-
perature dependence,OUbTut the output diode forward
Operation is as follows: when the power switch M1 turns
off, the SW pin voltage rises above the V supply. The
IN
voltage, V , has a significant negative temperature coef-
F
amplitude of the flyback pulse, i.e., the difference between
ficient (–1mV/°C to –2mV/°C). Such a negative tem-
perature coefficient produces approximately 200mV to
300mV voltage variation on the output voltage across
temperature.
the SW pin voltage and V supply, is given as:
IN
V
FLBK
= (V
+ V + I
• ESR) • N
SEC PS
OUT
F
V = Output diode forward voltage
F
I
= Transformer secondary current
For higher voltage outputs, such as 12V and 24V, the
output diode temperature coefficient has a negligible
effect on the output voltage regulation. For lower voltage
outputs, such as 3.3V and 5V, however, the output diode
temperature coefficient does count for an extra 2% to 5%
output voltage regulation.
SEC
ESR = Total impedance of secondary circuit
NPS = Transformer effective primary-to-secondary
turns ratio
The flyback voltage is then converted to a current, I
by the RFB resistor and the flyback pulse sense cirRcFuBit
(M2 and M3). This current, I , also flows through the
REF
resulting voltage feeds to the inverting input of the sam-
ple-and-hold error amplifier. Since the sample-and-hold
error amplifier samples the voltage when the secondary
current is zero, the (ISEC • ESR) term in the VFLBK equation
can be assumed to be zero.
,
The LT8302/LT8302-3 junction temperature usually tracks
the output diode junction temperature to the first order.
To compensate the negative temperature coefficient of the
RFB
R
resistor to generate a ground-referred voltage. The
output diode, a resistor, R , connected between the TC
TC
and R pins generates a proportional-to-absolute-tem-
REF
perature (PTAT) current. The PTAT current is zero at 25°C,
flows into the R pin at hot temperature, and flows out
REF
of the R pin at cold temperature. With the R resistor
REF
TC
The internal reference voltage, V , 1.00V, feeds to the
in place, the output voltage equation is revised as follows:
REF
noninverting input of the sample-and-hold error amplifier.
RFB
RREF
1
NPS
The relatively high gain in the overall loop causes the
VOUT = VREF
•
•
– V TO – V / T •
) (
(
)
F
TC
voltage at the R
pin to be nearly equal to the internal
REF
reference voltage V . The resulting relationship between
REF
RFB
RTC
1
NPS
V
and V can be expressed as:
T–TO •
•
– V / T • T–TO
F
FLBK
REF
(
)
(
(
)
)
⎛
⎜
⎝
⎞
VFLBK
RFB
•RREF = VREF or
⎟
TO=Room temperature 25°C
⎠
V / T =Output diode forward voltage
(
)
F
temperature coefficient
⎛
⎜
⎝
⎞
⎟
⎠
RFB
V
FLBK = VREF
•
°
V / T = 3.35mV/ C
R
TC
REF
V
REF
= Internal reference voltage 1.00V
Rev. G
11
For more information www.analog.com
LT8302/LT8302-3
APPLICATIONS INFORMATION
To cancel the output diode temperature coefficient, the
following two equations should be satisfied:
First, build and power up the application with the starting
, R values (no R resistor yet) and other com-
R
REF
FB
TC
ponents connected, and measure the regulated output
RFB
RREF
1
NPS
voltage, V
to:
. The new R value can be adjusted
VOUT = VREF
V / T •
•
•
– V TO
F
OUT(MEAS)
FB
(
)
VOUT
VOUT(MEAS)
RFB
RTC
1
RFB(NEW)
=
•RFB
•
= – V / T
(
)
(
)
F
TC
NPS
Second, with a new RFB resistor value selected, the output
diode temperature coefficient in the application can be
Selecting Actual R , R , R Resistor Values
REF FB TC
The LT8302/LT8302-3 uses a unique sampling scheme
to regulate the isolated output voltage. Due to the sam-
pling nature, the scheme contains repeatable delays and
error sources, which will affect the output voltage and
force a re-evaluation of the R and R resistor values.
tested to determine the R value. Still without the R
TC
TC
resistor, the V
should be measured over temperature
OUT
at a desired target output load. It is very important for this
evaluation that uniform temperature be applied to both the
output diode and the LT8302/LT8302-3. If freeze spray or
a heat gun is used, there can be a significant mismatch
in temperature between the two devices that causes sig-
nificant error. Attempting to extrapolate the data from a
diode data sheet is another option if there is no method
to apply uniform heating or cooling such as an oven. With
at least two data points spreading across the operating
temperature range, the output diode temperature coeffi-
cient can be determined by:
FB
TC
Therefore, a simple 2-step sequential process is recom-
mended for selecting resistor values.
Rearrangement of the expression for VOUT in the previous
sections yields the starting value for R :
FB
RREF •N • V + V TO
( )
(
)
F
PS
OUT
RFB =
VREF
V
= Output voltage
OUT
V
T1 – V
T1–T2
T2
OUT ( ) OUT ( )
– δV /δT =
(
)
VF (TO) = Output diode forward voltage at 25°C = ~0.3V
F
NPS = Transformer effective primary-to-secondary
turns ratio
Using the measured output diode temperature coefficient,
an exact RTC value can be selected with the following
equation:
The equation shows that the R resistor value is indepen-
dent of the R resistor value. FABny R resistor connected
TC
between the TTCC and R pins has no effect on the output
(
)
)
RFB
⎛
⎜
⎝
⎞
⎟
⎠
δV /δT
TC
REF
RTC
=
•
voltage setting at 25°C because the TC pin voltage is equal
– δV /δT
N
(
F
PS
to the R regulation voltage at 25°C.
REF
Once the R , R , and R values are selected, the reg-
REF FB
TC
The RREF resistor value should be approximately 10k
because the LT8302/LT8302-3 is trimmed and specified
ulation accuracy from board to board for a given appli-
cation will be very consistent, typically under 5% when
including device variation of all the components in the
system (assuming resistor tolerances and transformer
windings matching within 1%). However, if the trans-
former or the output diode is changed, or the layout is
using this value. If the R resistor value varies consid-
REF
erably from 10k, additional errors will result. However, a
variation in R
up to 10% is acceptable. This yields a
REF
bit of freedom in selecting standard 1% resistor values
to yield nominal R /R ratios.
FB REF
dramatically altered, there may be some change in V
.
OUT
Rev. G
12
For more information www.analog.com
LT8302/LT8302-3
APPLICATIONS INFORMATION
Output Power
5V, 12V, and 24V. The maximum output power curve is
the calculated output power if the switch voltage is 50V
during the switch-off time. 15V of margin is left for leak-
age inductance voltage spike. To achieve this power level
at a given input, a winding ratio value must be calculated
to stress the switch to 50V, resulting in some odd ratio
values. The curves below the maximum output power
curve are examples of common winding ratio values and
the amount of output power at given input voltages.
A flyback converter has a complicated relationship between
the input and output currents compared to a buck or a
boost converter. A boost converter has a relatively constant
maximum input current regardless of input voltage and a
buck converter has a relatively constant maximum out-
put current regardless of input voltage. This is due to the
continuous non-switching behavior of the two currents. A
flyback converter has both discontinuous input and out-
put currents which make it similar to a nonisolated buck-
boost converter. The duty cycle will affect the input and
output currents, making it hard to predict output power. In
addition, the winding ratio can be changed to multiply the
output current at the expense of a higher switch voltage.
One design example would be a 5V output converter with
a minimum input voltage of 8V and a maximum input
voltage of 32V. A three-to-one winding ratio fits this
design example perfectly and outputs equal to 15.3W at
32V but lowers to 7.7W at 8V.
The graphs in Figure 1 to Figure 4 show the typical maxi-
mum output power possible for the output voltages 3.3V,
ꢓꢌ
ꢓꢌ
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ꢆꢃꢄꢂꢃꢄ ꢂꢆꢍꢉR
ꢘꢈꢙꢀꢘꢃꢘ
ꢆꢃꢄꢂꢃꢄ ꢂꢆꢍꢉR
ꢁ ꢕ ꢓꢖꢎ
ꢎꢏ
ꢎꢏ
ꢎꢌ
ꢏ
ꢁ ꢕ ꢖꢗꢎ
ꢁ ꢕ ꢒꢖꢓ
ꢁ ꢕ ꢎꢖꢎ
ꢁ ꢕ ꢎꢖꢓ
ꢁ ꢕ ꢐꢗꢎ
ꢁ ꢕ ꢒꢗꢎ
ꢎꢌ
ꢏ
ꢁ ꢕ ꢓꢗꢎ
ꢌ
ꢌ
ꢌ
ꢎꢌ
ꢓꢌ
ꢒꢌ
ꢐꢌ
ꢎꢌ
ꢓꢌ
ꢀꢁꢂꢃꢄ ꢅꢆꢇꢄꢈGꢉ ꢊꢅꢋ
ꢒꢌ
ꢌ
ꢐꢌ
ꢀꢁꢂꢃꢄ ꢅꢆꢇꢄꢈGꢉ ꢊꢅꢋ
ꢑꢒꢌꢓ ꢔꢌꢎ
ꢑꢒꢌꢓ ꢔꢌꢒ
Figure 1. Output Power for 3.3V Output
Figure 3. Output Power for 12V Output
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ꢓꢌ
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ꢆꢃꢄꢂꢃꢄ ꢂꢆꢍꢉR
ꢗꢈꢘꢀꢗꢃꢗ
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ꢁ ꢕ ꢎꢖꢎ
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ꢁ ꢗ ꢒꢘꢎ
ꢁ ꢗ ꢓꢘꢎ
ꢎꢏ
ꢎꢌ
ꢎꢏ
ꢎꢌ
ꢏ
ꢁ ꢕ ꢓꢖꢒ
ꢁ ꢕ ꢎꢖꢓ
ꢁ ꢕ ꢎꢖꢒ
ꢁ ꢗ ꢎꢘꢎ
ꢏ
ꢌ
ꢌ
ꢌ
ꢎꢌ
ꢓꢌ
ꢀꢁꢂꢃꢄ ꢅꢆꢇꢄꢈGꢉ ꢊꢅꢋ
ꢒꢌ
ꢌ
ꢎꢌ
ꢓꢌ
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ꢒꢌ
ꢐꢌ
ꢐꢌ
ꢑꢒꢌꢓ ꢔꢌꢓ
ꢑꢒꢌꢓ ꢔꢌꢐ
Figure 2. Output Power for 5V Output
Figure 4. Output Power for 24V Output
Rev. G
13
For more information www.analog.com
LT8302/LT8302-3
APPLICATIONS INFORMATION
The equations below calculate output power:
the power switch shorter than approximately 160ns. This
minimum switch-on time is mainly for leading-edge blank-
ing the initial switch turn-on current spike. If the inductor
current exceeds the desired current limit during that time,
oscillation may occur at the output as the current control
loop will lose its ability to regulate. Therefore, the following
equation relating to maximum input voltage must also be
followed in selecting primary-side magnetizing inductance:
P
OUT
= η • V • D • I
• 0.5
IN
SW(MAX)
ꢀ η = Efficiency = ~85%
V
+V •N
)
F
PS
(
OUT
D=Duty Cycle=
V
+V •N +V
)
F IN
PS
(
OUT
ISW(MAX) = Maximum switch current limit = 3.6A (MIN)
tON(MIN) •V
IN(MAX)
LPRI
≥
Primary Inductance Requirement
ISW(MIN)
The LT8302/LT8302-3 obtains output voltage information
from the reflected output voltage on the SW pin. The con-
duction of secondary current reflects the output voltage on
the primary SW pin. The sample-and-hold error amplifier
needs a minimum 350ns to settle and sample the reflected
output voltage. In order to ensure proper sampling, the
secondary winding needs to conduct current for a mini-
mum of 350ns. The following equation gives the minimum
value for primary-side magnetizing inductance:
t
= Minimum switch-on time = 160ns (TYP)
ON(MIN)
In general, choose a transformer with its primary mag-
netizing inductance about 40% to 60% larger than the
minimum values calculated above. A transformer with
much larger inductance will have a bigger physical size
and may cause instability at light load.
Selecting a Transformer
Transformer specification and design is perhaps the most
critical part of successfully applying the LT8302/LT8302-
3. In addition to the usual list of guidelines dealing with
high frequency isolated power supply transformer design,
the following information should be carefully considered.
tOFF(MIN) •N • V +V
(
)
F
PS
OUT
LPRI
≥
ISW(MIN)
t
= Minimum switch-off time = 350ns (TYP)
= Minimum switch current limit = 0.87A (TYP)
OFF(MIN)
I
SW(MIN)
Analog Devices has worked with several leading magnetic
component manufacturers to produce pre-designed fly-
back transformers for use with the LT8302/LT8302-3.
Table 1 shows the details of these transformers.
In addition to the primary inductance requirement for the
minimum switch-off time, the LT8302/LT8302-3 has mini-
mum switch-on time that prevents the chip from turning on
Table 1. Predesigned Transformers–Typical Specifications
TARGET APPLICATION
TRANSFORMER
PART NUMBER
DIMENSIONS
L
L
R
R
SEC
PRI
LKG
PRI
(W × L × H) (mm)
(µH)
(µH)
0.35
0.12
0.6
N :N
(mΩ) (mΩ) VENDOR
V
(V)
V
(V)
I
(A)
OUT
P
S
IN
OUT
750311625
750311564
750313441
750311624
12387-TO79
750313445
750313457
750313460
750311342
750313439
750313442
17.75 × 13.46 × 12.70
17.75 × 13.46 × 12.70
15.24 × 13.34 x 11.43
17.75 × 13.46 × 12.70
15.5 × 12.5 × 11.5
15.24 × 13.34 × 11.43
15.24 × 13.34 × 11.43
15.24 × 13.34 × 11.43
15.24 × 13.34 × 11.43
15.24 × 13.34 × 11.43
15.24 × 13.34 × 11.43
9
4:1
3:1
2:1
3:2
43
36
6
Wurth Elektronik
Wurth Elektronik
Wurth Elektronik
Wurth Elektronik
Sumida
8 to 32
8 to 32
8 to 32
8 to 32
8 to 36
8 to 36
8 to 36
4 to 18
4 to 18
18 to 42
18 to 42
3.3
2.1
9
7
5
5
8
1.5
1.3
0.9
0.3
0.3
0.15
0.9
0.4
2.1
1.6
9
75
18
21
90
9
0.18
0.5
34
9
1:1:1
1:2
1:4
4:1
2:1
2:1
3:2
55
12
24
48
5
9
0.25
0.25
0.7
85
190 Wurth Elektronik
770 Wurth Elektronik
9
85
12
15
12
12
85
11
22
28
53
Wurth Elektronik
Wurth Elektronik
Wurth Elektronik
Wurth Elektronik
0.44
0.6
85
12
3.3
5
115
150
0.75
Rev. G
14
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LT8302/LT8302-3
APPLICATIONS INFORMATION
Turns Ratio
The turns ratio is an important element in the isolated
feedback scheme, and directly affects the output voltage
accuracy. Make sure the transformer manufacturer spec-
ifies turns ratio accuracy within 1%.
Note that when choosing an R /R resistor ratio to set
FB REF
output voltage, the user has relative freedom in selecting
a transformer turns ratio to suit a given application. In
contrast, the use of simple ratios of small integers, e.g.,
3:1, 2:1, 1:1, etc., provides more freedom in settling total
turns and mutual inductance.
Saturation Current
The current in the transformer windings should not exceed
its rated saturation current. Energy injected once the core
is saturated will not be transferred to the secondary and
will instead be dissipated in the core. When designing
custom transformers to be used with the LT8302/LT8302-
3, the saturation current should always be specified by
the transformer manufacturers.
Typically, choose the transformer turns ratio to maximize
available output power. For low output voltages (3.3V
or 5V), a N:1 turns ratio can be used with multiple pri-
mary windings relative to the secondary to maximize the
transformer’s current gain (and output power). However,
remember that the SW pin sees a voltage that is equal
to the maximum input supply voltage plus the output
voltage multiplied by the turns ratio. In addition, leakage
inductance will cause a voltage spike (VLEAKAGE) on top of
this reflected voltage. This total quantity needs to remain
below the 65V absolute maximum rating of the SW pin to
prevent breakdown of the internal power switch. Together
these conditions place an upper limit on the turns ratio,
NPS, for a given application. Choose a turns ratio low
enough to ensure
Winding Resistance
Resistance in either the primary or secondary windings
will reduce overall power efficiency. Good output voltage
regulation will be maintained independent of winding
resistance due to the boundary/discontinuous conduction
mode operation of the LT8302/LT8302-3.
Leakage Inductance and Snubbers
Transformer leakage inductance on either the primary
or secondary causes a voltage spike to appear on the
primary after the power switch turns off. This spike is
increasingly prominent at higher load currents where
more stored energy must be dissipated. It is very import-
ant to minimize transformer leakage inductance.
65V – VIN(MAX) – VLEAKAGE
NPS <
VOUT +VF
For larger N:1 values, choose a transformer with a larger
physical size to deliver additional current. In addition,
choose a large enough inductance value to ensure that
the switch-off time is long enough to accurately sample
the output voltage.
When designing an application, adequate margin should
be kept for the worst-case leakage voltage spikes even
under overload conditions. In most cases shown in
Figure 5, the reflected output voltage on the primary plus
For lower output power levels, choose a 1:1 or 1:N trans-
former for the absolute smallest transformer size. A 1:N
transformer will minimize the magnetizing inductance
(and minimize size), but will also limit the available output
power. A higher 1:N turns ratio makes it possible to have
very high output voltages without exceeding the break-
down voltage of the internal power switch.
V
should be kept below 50V. This leaves at least 15V
IN
margin for the leakage spike across line and load condi-
tions. A larger voltage margin will be required for poorly
wound transformers or for excessive leakage inductance.
Rev. G
15
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LT8302/LT8302-3
APPLICATIONS INFORMATION
then add capacitance until the period of the ringing is 1.5
to 2 times longer. The change in period determines the
value of the parasitic capacitance, from which the para-
sitic inductance can be also determined from the initial
period. Once the value of the SW node capacitance and
inductance is known, a series resistor can be added to
the snubber capacitance to dissipate power and critically
damp the ringing. The equation for deriving the optimal
series resistance using the observed periods ( tPERIOD and
tPERIOD(SNUBBED)) and snubber capacitance (CSNUBBER) is:
ꢐꢑꢅꢉ
ꢉ
ꢊꢋꢌꢍꢌGꢋ
ꢐꢅꢆꢉ
ꢉ
ꢎꢏ
ꢀ
ꢃ ꢄꢅꢆꢇꢈ
ꢁꢂꢂ
ꢀ
ꢎꢗ
ꢐ ꢖꢅꢆꢇꢈ
ꢕꢄꢆꢖ ꢂꢆꢅ
ꢒꢓꢔꢋ
CSNUBBER
Figure 5. Maximum Voltages for SW Pin Flyback Waveform
CPAR
=
2
⎛
⎜
⎝
⎞
⎟
⎠
tPERIOD(SNUBBED)
tPERIOD
–1
In addition to the voltage spikes, the leakage inductance
also causes the SW pin ringing for a while after the power
switch turns off. To prevent the voltage ringing falsely trig-
ger boundary mode detector, the LT8302/LT8302-3 inter-
nally blanks the boundary mode detector for approximately
250ns. Any remaining voltage ringing after 250ns may turn
the power switch back on again before the secondary cur-
rent falls to zero. In this case, the LT8302/LT8302-3 enters
continuous conduction mode. So the leakage inductance
spike ringing should be limited to less than 250ns.
2
tPERIOD
LPAR
=
CPAR •4π2
LPAR
CPAR
RSNUBBER
=
Note that energy absorbed by the RC snubber will be con-
verted to heat and will not be delivered to the load. In high
voltage or high current applications, the snubber needs
to be sized for thermal dissipation. A 470pF capacitor in
series with a 39Ω resistor is a good starting point.
To clamp and damp the leakage voltage spikes, a
(RC + DZ) snubber circuit in Figure 6 is recommended.
The RC (resistor-capacitor) snubber quickly damps the
voltage spike ringing and provides great load regulation
and EMI performance. And the DZ (diode-Zener) ensures
well defined and consistent clamping voltage to protect
SW pin from exceeding its 65V absolute maximum rating.
For the DZ snubber, proper care should be taken when
choosing both the diode and the Zener diode. Schottky
diodes are typically the best choice, but some PN diodes
can be used if they turn on fast enough to limit the leakage
inductance spike. Choose a diode that has a reverse-volt-
age rating higher than the maximum SW pin voltage. The
Zener diode breakdown voltage should be chosen to bal-
ance power loss and switch voltage protection. The best
compromise is to choose the largest voltage breakdown
with 5V margin. Use the following equation to make the
proper choice:
L
ℓ
•
Z
C
R
D
•
8302 F06
V
≤ 60V – V
IN(MAX)
ZENNER(MAX)
Figure 6. (RC + DZ) Snubber Circuit
For an application with a maximum input voltage of 32V,
choose a 24V Zener diode, the V of which is
around 26V and below the 28V maximum. The power loss
in the DZ snubber determines the power rating of the Zener
diode. A 1.5W Zener diode is typically recommended.
Rev. G
ZENER(MAX)
The recommended approach for designing an RC snubber
is to measure the period of the ringing on the SW pin
when the power switch turns off without the snubber and
16
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LT8302/LT8302-3
APPLICATIONS INFORMATION
Undervoltage Lockout (UVLO)
minimum amount of energy even during light load con-
ditions to ensure accurate output voltage information.
The minimum energy delivery creates a minimum load
requirement, which can be approximately estimated as:
A resistive divider from V to the EN/UVLO pin imple-
IN
ments undervoltage lockout (UVLO). The EN/UVLO enable
falling threshold is set at 1.214V with 14mV hysteresis. In
addition, the EN/UVLO pin sinks 2.5µA when the voltage
on the pin is below 1.214V. This current provides user
programmable hysteresis based on the value of R1. The
programmable UVLO thresholds are:
LPRI •ISW(MIN)2 •fMIN
ILOAD(MIN)
=
2•VOUT
L
PRI
= Transformer primary inductance
1.228V • R1+R2
ISW(MIN) = Minimum switch current limit = 1.04A (MAX)
= Minimum switching frequency = 12.7kHz (MAX)
(
)
+
V
=
=
+2.5µA •R1
IN(UVLO )
R2
f
MIN
The LT8302/LT8302-3 typically needs less than 0.5% of
its full output power as minimum load. Alternatively, a
Zener diode with its breakdown of 10% higher than the
output voltage can serve as a minimum load if pre-loading
is not acceptable. For a 5V output, use a 5.6V Zener with
cathode connected to the output.
1.214V • R1+R2
(
)
–
V
IN(UVLO )
R2
Figure 7 shows the implementation of external shutdown
control while still using the UVLO function. The NMOS
grounds the EN/UVLO pin when turned on, and puts the
Output Short Protection
ꢌ
ꢓꢈ
When the output is heavily overloaded or shorted to
ground, the reflected SW pin waveform rings longer
than the internal blanking time. After the 350ns minimum
switch-off time, the excessive ringing falsely triggers the
boundary mode detector and turns the power switch back
on again before the secondary current falls to zero. Under
this condition, the LT8302/LT8302-3 runs into continu-
ous conduction mode at 380kHz maximum switching fre-
Rꢎ
Rꢅ
ꢊꢈꢆꢋꢌꢀꢍ
ꢀꢁꢂꢃꢄꢅꢆꢀꢁꢂꢃꢄꢅꢇꢃ
Gꢈꢉ
RUNꢆꢏꢁꢍꢐ
ꢑꢍꢈꢁRꢍꢀ
ꢒꢍꢐꢁꢓꢍꢈꢔꢀꢕ
ꢂꢃꢄꢅ ꢖꢄꢗ
Figure 7. Undervoltage Lockout (UVLO)
quency. If the sampled R voltage is still less than 0.6V
REF
after 11ms (typ) soft-start timer, the LT8302/LT8302-3
initiates a new soft-start cycle. If the sampled RREF voltage
is larger than 0.6V after 11ms, the switch current may
run away and exceed the 4.5A maximum current limit.
Once the switch current hits 7.2A over current limit, the
LT8302/LT8302-3 also initiates a new soft-start cycle.
Under either condition, the new soft-start cycle throttles
back both the switch current limit and switch frequency.
The output short-circuit protection prevents the switch
current from running away and limits the average output
diode current.
LT8302/LT8302-3 in shutdown with quiescent current
less than 2µA.
Minimum Load Requirement
The LT8302/LT8302-3 samples the isolated output
voltage from the primary-side flyback pulse waveform.
The flyback pulse occurs once the primary switch turns
off and the secondary winding conducts current. In order
to sample the output voltage, the LT8302/LT8302-3 has
to turn on and off for a minimum amount of time and with
a minimum frequency. The LT8302/LT8302-3 delivers a
Rev. G
17
For more information www.analog.com
LT8302/LT8302-3
APPLICATIONS INFORMATION
Design Example
Step 2: Determine the primary inductance.
Use the following design example as a guide to design-
ing applications for the LT8302/LT8302-3. The design
example involves designing a 5V output with a 1.5A load
current and an input range from 8V to 32V.
Primary inductance for the transformer must be set above
a minimum value to satisfy the minimum switch-off and
switch-on time requirements:
tOFF(MIN) •N • V +V
(
)
F
PS
OUT
LPRI
≥
VIN(MIN) = 8V, VIN(NOM) = 12V, VIN(MAX) = 32V,
ISW(MIN)
V
= 5V, I
= 1.5A
OUT
OUT
tON(MIN) •V
Step 1: Select the transformer turns ratio.
IN(MAX)
LPRI
≥
ISW(MIN)
65V – VIN(MAX) – VLEAKAGE
NPS <
VOUT +VF
t
t
I
= 350ns
= 160ns
= 0.87A
OFF(MIN)
ON(MIN)
SW(MIN)
VLEAKAGE = Margin for transformer leakage spike = 15V
V = Output diode forward voltage = ~0.3V
F
Example:
Example:
350ns•3• 5V+0.3V
65V –32V –15V
(
)
NPS <
=3.4
LPRI
≥
≥
=6.4µH
5V+0.3V
0.87A
160ns•32V
The choice of transformer turns ratio is critical in deter-
mining output current capability of the converter. Table 2
shows the switch voltage stress and output current capa-
bility at different transformer turns ratio.
LPRI
=5.9µH
0.87A
Most transformers specify primary inductance with a tol-
erance of 20%. With other component tolerance consid-
ered, choose a transformer with its primary inductance
40% to 60% larger than the minimum values calculated
Table 2. Switch Voltage Stress and Output Current Capability vs
Turns Ratio
V
V
at
I
at
(A)
SW(MAX)
IN(MAX)
OUT(MAX)
IN(MIN)
NPS
1:1
2:1
3:1
(V)
V
DUTY CYCLE (%)
14-40
above. L = 9µH is then chosen in this example.
PRI
37.3
0.92
Once the primary inductance has been determined, the
maximum load switching frequency can be calculated as:
42.6
47.9
1.31
1.53
25-57
33-67
1
1
fSW
=
=
Clearly, only NPS = 3 can meet the 1.5A output current
LPRI •ISW
LPRI •ISW
t
ON +tOFF
+
requirement, so N = 3 is chosen as the turns ratio in
PS
V
N • V +V
(
)
F
IN
PS
OUT
this example.
V
OUT •IOUT •2
ISW
=
η•V •D
IN
Rev. G
18
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LT8302/LT8302-3
APPLICATIONS INFORMATION
Example:
and cost of a larger capacitor. Use the following equation
to calculate the output capacitance:
5V + 0.3V • 3
(
)
D =
= 0.57
2
LPRI •ISW
5V + 0.3V • 3+12V
(
)
COUT
=
2•VOUT •ΔVOUT
5V •1.5A • 2
ISW
=
0.8 •12V • 0.57
Example:
fSW = 277kHz
Design for output voltage ripple less than 1% of V
i.e., 100mV.
,
OUT
The transformer also needs to be rated for the correct
saturation current level across line and load conditions.
A saturation current rating larger than 7A is necessary to
work with the LT8302/LT8302-3. The 750311564 from
Wurth is chosen as the flyback transformer.
2
9µH• 4.5A
(
)
COUT
=
=182µF
2•5V •0.1V
Remember ceramic capacitors lose capacitance with
applied voltage. The capacitance can drop to 40% of
quoted capacitance at the maximum voltage rating. So
a 220µF, 6.3V rating X5R or X7R ceramic capacitor is
chosen.
Step 3: Choose the output diode.
Two main criteria for choosing the output diode include
forward current rating and reverse-voltage rating. The
maximum load requirement is a good first-order guess
at the average current requirement for the output diode.
Under output short-circuit condition, the output diode
needs to conduct much higher current. Therefore, a con-
servative metric is 60% of the maximum switch current
limit multiplied by the turns ratio:
Step 5: Design snubber circuit.
The snubber circuit protects the power switch from leak-
age inductance voltage spike. A (RC + DZ) snubber is
recommended for this application. A 470pF capacitor in
series with a 39Ω resistor is chosen as the RC snubber.
I
= 0.6 • I • N
SW(MAX) PS
DIODE(MAX)
The maximum Zener breakdown voltage is set according
to the maximum V :
Example:
IN
V
≤ 60V – V
IN(MAX)
I
= 8.1A
ZENNER(MAX)
DIODE(MAX)
Example:
Next calculate reverse voltage requirement using maxi-
mum V :
IN
V
≤ 60V – 32V = 28V
ZENNER(MAX)
V
IN(MAX)
A 24V Zener with a maximum of 26V will provide optimal
protection and minimize power loss. So a 24V, 1.5W Zener
from Central Semiconductor (CMZ5934B) is chosen.
V
REVERSE = VOUT
+
NPS
Example:
Choose a diode that is fast and has sufficient reverse
voltage breakdown:
32V
3
VREVERSE =5V+
=15.7V
V
V
> V
SW(MAX)
REVERSE
SW(MAX)
The PDS835L (8A, 35V diode) from Diodes Inc. is chosen.
= V
+ V
ZENNER(MAX)
IN(MAX)
Step 4: Choose the output capacitor.
Example:
The output capacitor should be chosen to minimize the
output voltage ripple while considering the increase in size
V
> 60V
REVERSE
A 100V, 1A diode from Diodes Inc. (DFLS1100) is chosen.
Rev. G
19
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LT8302/LT8302-3
APPLICATIONS INFORMATION
Step 6: Select the R
and R resistors.
Example:
REF
FB
Use the following equation to calculate the starting values
5.189V –5.041V
– δV /δT =
=1.48mV /°C
(
)
F
for R and R :
REF
FB
100°C– 0°C
(
)
R
REF •N • V
+ V TO
(
)
)
(
F
PS
OUT
⎛
⎞
⎟
⎠
3.35mV/°C 154
RFB =
RTC
=
•
=115k
⎜
VREF
⎝
1.48mV/°C
3
RREF = 10k
Step 9: Select the EN/UVLO resistors.
Example:
Determine the amount of hysteresis required and calcu-
late R1 resistor value:
10k •3• 5V+0.3V
(
)
RFB =
=159k
1.00V
V
= 2.5µA • R1
IN(HYS)
For 1% standard values, a 158k resistor is chosen.
Example:
Step 7: Adjust R resistor based on output voltage.
FB
Choose 2V of hysteresis, R1 = 806k
Build and power up the application with application com-
ponents and measure the regulated output voltage. Adjust
FB
Determine the UVLO thresholds and calculate R2 resistor
value:
R
resistor based on the measured output voltage:
1.228V • R1+R2
(
)
+ 2.5µA •R1
V
=
VOUT
VOUT(MEASURED)
IN(UVLO+)
R2
RFB(NEW)
=
•RFB
Example:
Example:
Set V UVLO rising threshold to 7.5V:
IN
5V
5.14V
R2 = 232k
RFB =
•158k =154k
V
V
+ = 7.5V
IN(UVLO )
Step 8: Select RTC resistor based on output voltage tem-
perature variation.
– = 5.5V
IN(UNLO )
Step 10: Ensure minimum load.
Measure output voltage in a controlled temperature envi-
ronment like an oven to determine the output temperature
coefficient. Measure output voltage at a consistent load
current and input voltage, across the operating tempera-
ture range.
The theoretical minimum load can be approximately esti-
mated as:
9µH• 1.04A 2 •12.7kHz
(
)
ILOAD(MIN)
=
=12.4mA
2 • 5V
Calculate the temperature coefficient of V :
F
Remember to check the minimum load requirement in
real application. The minimum load occurs at the point
where the output voltage begins to climb up as the con-
verter delivers more energy than what is consumed at
the output. The real minimum load for this application is
about 10mA. In this example, a 500Ω resistor is selected
as the minimum load.
V
T1 – V
T1–T2
T2
OUT ( ) OUT ( )
– δV /δT =
(
)
F
⎛
⎞
⎟
⎠
3.35mV/°C RFB
RTC
=
•
⎜
– δV /δT
N
PS
(
)
⎝
F
Rev. G
20
For more information www.analog.com
LT8302/LT8302-3
TYPICAL APPLICATIONS
8V to 32VIN/12VOUT Isolated Flyback Converter
+
V
D2
T1
OUT
12V
V
IN
8V TO 32V
1:1
5mA TO 0.8A (V = 12V)
5mA TO 1.1A (V = 24V)
IN
IN
C3
Z1
470pF
9µH
•
C4
47µF
9µH
R3
R1
D1
V
C1
10µF
IN
39Ω
806k
•
–
V
EN/UVLO
SW
OUT
R2
232k
R4
121k
LT8302/LT8302-3
GND
INTV
R
FB
D1: DIODES DFLS1100
D2: DIODES PDS360
T1: SUMIDA 12387-TO79
Z1: CENTRAL CMZ5934B
R
REF
CC
C2
1µF
R5
R6
OPEN
10k
TC
8302 TA02a
Efficiency vs Load Current
Load and Line Regulation
ꢍꢎ.ꢑ
ꢍꢎ.ꢎ
ꢕꢎ
ꢕꢌ
ꢍꢎ.ꢌ
ꢍꢍ.ꢓ
ꢔꢎ
ꢔꢌ
ꢍꢍ.ꢒ
ꢍꢍ.ꢑ
ꢍꢍ.ꢎ
ꢓꢎ
ꢓꢌ
ꢍꢎ
ꢛ
ꢛ
ꢜ ꢘꢖꢛ
ꢜ ꢖꢗꢛ
ꢐꢇ
ꢐꢇ
ꢐ
ꢐ
ꢗ ꢍꢎꢐ
ꢗ ꢎꢑꢐ
ꢖꢇ
ꢖꢇ
ꢌ
ꢎꢌꢌ
ꢑꢌꢌ
ꢒꢌꢌ
ꢓꢌꢌ ꢍꢌꢌꢌ ꢍꢎꢌꢌ
ꢌ
ꢖꢌꢌ
ꢗꢌꢌ
ꢍꢌꢌ
ꢔꢌꢌ ꢘꢌꢌꢌ ꢘꢖꢌꢌ
ꢀꢁꢂꢃ ꢄꢅRRꢆꢇꢈ ꢉꢊꢂꢋ
ꢀꢁꢂꢃ ꢄꢅRRꢆꢇꢈ ꢉꢊꢂꢋ
ꢓꢔꢌꢎ ꢈꢂꢌꢎꢕ
ꢔꢙꢌꢖ ꢈꢂꢌꢖꢚ
8V to 32VIN/3.3VOUT Isolated Flyback Converter
Output Temperature Variation
ꢓ.ꢎꢏ
ꢓ.ꢔꢎ
ꢓ.ꢔꢏ
ꢓ.ꢓꢎ
ꢓ.ꢓꢏ
ꢓ.ꢗꢎ
ꢓ.ꢗꢏ
ꢓ.ꢕꢎ
ꢓ.ꢕꢏ
+
ꢑ
ꢐꢈꢆ
ꢚ ꢕꢗꢑ
ꢃꢅ
V
D2
T1
OUT
3.3V
ꢃ
ꢚ ꢕꢀ
V
IN
8V TO 32V
20mA TO 2.7A (V = 12V)
20mA TO 3.8A (V = 24V)
IN
4:1
IN
C3
Z1
470pF
9µH
•
C4
0.56µH
R3
R1
470µF
D1
V
C1
10µF
IN
39Ω
806k
•
R
ꢆꢋ
ꢚ ꢕꢏꢎꢛ
–
V
SW
EN/UVLO
LT8302/LT8302-3
OUT
R2
232k
R4
140k
R
ꢆꢋ
ꢚ ꢐꢇꢄꢅ
R
GND
INTV
FB
D1: DIODES DFLS1100
D2: DIODES PDS1040L
T1: WURTH 750311625
R
REF
CC
C2
1µF
R5
R6
105k
Z1: CENTRAL CMZ5934B
10k
TC
8302 TA03
ꢎꢏ ꢙꢎ
ꢏ
ꢀꢁꢂꢃꢄꢅꢆ ꢆꢄꢁꢇꢄRꢀꢆꢈRꢄ ꢉꢊꢋꢌ
ꢍꢎꢏ ꢍꢗꢎ
ꢗꢎ
ꢕꢏꢏ ꢕꢗꢎ ꢕꢎꢏ
ꢖꢓꢏꢗ ꢆꢀꢏꢓꢘ
Rev. G
21
For more information www.analog.com
LT8302/LT8302-3
TYPICAL APPLICATIONS
8V to 36VIN/ 12VOUT Isolated Flyback Converter
+
T1
1:1:1
V
D2
OUT1
12V
V
IN
8V TO 36V
5mA TO 0.4A (V = 12V)
IN
C3
5mA TO 0.55A (V = 24V)
IN
Z1
470pF
•
C4
9µH
9µH
D3
R3
39Ω
R1
22µF
D1
V
C1
10µF
IN
806k
•
–
V
V
12V
EN/UVLO
SW
OUT2
+
R2
232k
R4
121k
OUT2
LT8302/LT8302-3
GND
INTV
R
5mA TO 0.4A (V = 12V)
FB
IN
5mA TO 0.55A (V = 24V)
IN
•
R
REF
C5
22µF
CC
9µH
C2
1µF
R5
R6
OPEN
10k
–
V
TC
OUT2
8302 TA04
D1: DIODES DFLS1100
D2, D3: DIODES PDS360
T1: SUMIDA 12387-TO79
Z1: CENTRAL CMZ5934B
8V to 36VIN/24VOUT Isolated Flyback Converter
+
V
D2
T1
OUT
24V
V
IN
8V TO 36V
1:2
2.5mA TO 0.4A (V = 12V)
2.5mA TO 0.55A (V = 24V)
IN
IN
C3
Z1
470pF
9µH
•
C4
10µF
36µH
R3
R1
D1
V
C1
10µF
IN
39Ω
806k
•
–
V
EN/UVLO
SW
OUT
R2
232k
R4
121k
LT8302/LT8302-3
GND
INTV
R
FB
D1: DIODES DFLS1100
D2: DIODES SBR2U150SA
T1: WURTH 750313445
Z1: CENTRAL CMZ5934B
R
REF
CC
C2
1µF
R5
R6
OPEN
10k
TC
8302 TA05
8V to 36VIN/48VOUT Isolated Flyback Converter
+
V
D2
T1
OUT
48V
V
IN
8V TO 36V
1:4
1.2mA TO 0.2A (V = 12V)
1.2mA TO 0.27A (V = 24V)
IN
IN
C3
Z1
470pF
9µH
•
C4
144µH
R3
R1
2.2µF
D1
V
C1
10µF
IN
39Ω
806k
•
–
EN/UVLO
V
SW
OUT
R2
232k
R4
121k
LT8302/LT8302-3
GND
INTV
R
FB
D1: DIODES DFLS1100
D2: DIODES SBR1U200P1
T1: WURTH 750313457
Z1: CENTRAL CMZ5934B
R
REF
CC
C2
1µF
R5
R6
OPEN
10k
TC
8302 TA06
Rev. G
22
For more information www.analog.com
LT8302/LT8302-3
TYPICAL APPLICATIONS
8V to 32VIN/5VOUT Isolated Flyback Converter with LT8309
Efficiency vs Load Current
+
V
OUT
T1
V
IN
8V TO 32V
5V/2.0A (V = 12V)
5V/2.9A (V = 24V)
IN
ꢒꢑ
IN
3:1
C3
470pF
Z1
•
9µH
1µH
ꢒꢋ
ꢐꢑ
ꢐꢋ
C4
R3
39Ω
R1
R7
D1
D2
V
C1
10µF
220µF
IN
806k
5Ω
•
EN/UVLO
SW
C4
R2
232k
R4
10µF
LT8302/LT8302-3
154k
R8
2.1k
GND
INTV
R
FB
V
CC
R
REF
DRAIN
CC
C2
1µF
R5
10k
R6
OPEN
LT8309
GATE INTV
GND
ꢖꢑ
ꢖꢋ
ꢘꢑ
M1
CC
TC
C5
4.7µF
D1: DIODES DFLS1100
–
D2: CENTRAL CMMSH1-60
M1: INFINEON BSC059N04LS
T1: WURTH 750311564
V
OUT
8302 TA07
ꢋ
ꢋ.ꢑ
ꢓ.ꢋ
ꢓ.ꢑ
ꢔ.ꢋ
ꢔ.ꢑ
ꢕ.ꢋ
Z1: CENTRAL CMZ5934B
ꢀꢁꢂꢃ ꢄꢅRRꢆꢇꢈ ꢉꢂꢊ
ꢐꢕꢋꢔ ꢈꢂꢋꢖꢗ
–4V to –42VIN/12VOUT Buck-Boost Converter
Efficiency vs Load Current
ꢅꢍ
ꢀ
ꢕꢎ
ꢕꢌ
ꢖꢕꢆ
ꢍꢊꢀꢋꢉ.ꢘꢟꢞ ꢣꢀ ꢤ ꢠꢟꢀꢥ
ꢐꢍ
ꢍꢊꢎꢏ
ꢁꢂ
ꢍꢊꢀꢋꢉ.ꢇꢞ ꢣꢀ ꢤ ꢠꢍꢊꢀꢥ
ꢁꢂ
ꢍꢊꢀꢋꢍ.ꢍꢞ ꢣꢀ ꢤ ꢠꢊꢘꢀꢥ
ꢁꢂ
ꢗꢈ
ꢘꢙꢎꢒ
Rꢘ
ꢑꢍ
ꢍꢊꢀꢋꢍ.ꢈꢞ ꢣꢀ ꢤ ꢠꢘꢊꢀꢥ
ꢁꢂ
ꢀ
ꢃꢄ
ꢁꢂ
ꢔꢂꢋꢕꢀꢅꢖ
ꢅꢆꢇꢈꢉꢊꢋꢅꢆꢇꢈꢉꢊꢌꢈ
ꢍꢍꢇꢡ
ꢔꢎ
ꢔꢌ
R
ꢒꢓ
ꢗꢍ
ꢍꢉꢎꢒ
R
Rꢔꢒ
ꢁꢂꢆꢀ
ꢐꢍꢚ ꢐꢁꢖꢐꢔꢃ ꢛꢜꢔGꢝꢉꢈꢉꢔꢛ
ꢅꢍꢚ ꢄꢕRꢆꢏ ꢙꢘꢘꢙꢙꢉꢍꢍꢊ
ꢑꢍꢚ ꢗꢔꢂꢆRꢞꢅ ꢗꢜꢏꢑꢟꢊꢘꢈꢓ
ꢗꢗ
Rꢟ
ꢍꢉꢡ
ꢗꢊ
ꢍꢎꢒ
ꢓꢎ
ꢓꢌ
ꢍꢎ
Gꢂꢐ
ꢛ
ꢐꢇ
ꢛ
ꢐꢇ
ꢛ
ꢐꢇ
ꢛ
ꢐꢇ
ꢜ ꢝꢎꢛ
ꢀ
ꢁꢂ
ꢇꢈꢉꢊ ꢆꢞꢉꢇꢢ
ꢜ ꢝꢘꢖꢛ
ꢜ ꢝꢖꢗꢛ
ꢜ ꢝꢗꢖꢛ
ꢠꢘꢀ ꢆꢖ ꢠꢘꢊꢀ
ꢌ
ꢖꢌꢌ ꢗꢌꢌ ꢍꢌꢌ ꢔꢌꢌ ꢘꢌꢌꢌ ꢘꢖꢌꢌ ꢘꢗꢌꢌ
ꢀꢁꢂꢃ ꢄꢅRRꢆꢇꢈ ꢉꢊꢂꢋ
ꢔꢙꢌꢖ ꢈꢂꢌꢔꢚ
–18V to –42VIN/–12VOUT Negative Buck Converter
Efficiency vs Load Current
ꢕꢌꢌ
ꢟꢆ
ꢞꢝꢌꢗ
ꢔꢒ
ꢓꢋ
ꢒꢋ
ꢀ
ꢐꢋꢈꢀ
ꢋ.ꢅꢑ
ꢎꢏꢄ
ꢔꢌ
ꢓꢒ
ꢃꢋ
ꢋꢈꢌꢍ
Rꢋ
ꢅꢇꢜꢡ
ꢀ
ꢁꢂ
ꢖꢂꢉꢏꢀꢃꢎ
ꢔꢕ
ꢟꢋ
ꢋꢇꢌꢗ
Rꢈ
ꢈꢆꢈꢡ
Rꢞ
ꢋꢋꢅꢡ
ꢃꢄꢅꢆꢇꢈꢉꢃꢄꢅꢆꢇꢈꢊꢆ
ꢓꢌ
ꢍꢒ
ꢍꢌ
ꢒꢋꢙ ꢒꢁꢎꢒꢖꢔ ꢚꢛꢖGꢜꢇꢆꢇꢖꢚ
ꢃꢋꢙ ꢕꢏRꢄꢍ ꢝꢞꢞꢝꢝꢇꢋꢋꢈ
ꢓꢋꢙ ꢟꢖꢂꢄRꢑꢃ ꢟꢛꢍꢓꢠꢈꢞꢆꢘ
ꢖꢂꢉꢏꢀꢃꢎ
R
ꢗꢘ
ꢙ
ꢙ
ꢙ
ꢚ ꢛꢕꢓꢙ
ꢚ ꢛꢖꢜꢙ
ꢚ ꢛꢜꢖꢙ
ꢏꢇ
ꢏꢇ
ꢏꢇ
R
ꢁꢂꢄꢀ
ꢟꢟ
Rꢖꢗ
Rꢠ
ꢟꢈ
ꢋꢌꢗ
ꢋꢇꢡ
ꢀ
ꢁꢂ
ꢌ
ꢒꢌꢌ
ꢕꢌꢌꢌ
ꢕꢒꢌꢌ
ꢖꢌꢌꢌ
ꢅꢆꢇꢈ ꢄꢑꢇꢢꢣ
ꢐꢋꢅꢀ ꢄꢎ ꢐꢞꢈꢀ
ꢀꢁꢂꢃ ꢄꢅRRꢆꢇꢈ ꢉꢊꢂꢋ
ꢓꢗꢌꢖ ꢈꢂꢌꢔꢘ
Rev. G
23
For more information www.analog.com
LT8302/LT8302-3
PACKAGE DESCRIPTION
S8E Package
8-Lead Plastic SOIC (Narrow .150 Inch) Exposed Pad
ꢅReꢧeꢨeꢩꢪe ꢜꢋꢒ ꢛꢡG ꢫ ꢀꢄꢬꢀꢊꢬꢁꢊꢄꢈ Rev ꢒꢉ
.ꢁꢊꢓ ꢃ .ꢁꢓꢈ
ꢅꢆ.ꢊꢀꢁ ꢃ ꢄ.ꢀꢀꢆꢉ
.ꢀꢄꢀ
ꢅꢁ.ꢇꢈꢉ
ꢔꢏꢒ
ꢕꢖꢋꢐ ꢎ
.ꢀꢆꢄ ±.ꢀꢀꢄ
ꢅꢁ.ꢁꢆꢎ ±ꢀ.ꢁꢇꢈꢉ
.ꢀꢀꢄ ꢅꢀ.ꢁꢎꢉ ꢗꢘꢙ
ꢈ
ꢄ
ꢊ
ꢂ
.ꢁꢄꢀ ꢃ .ꢁꢄꢈ
ꢅꢎ.ꢊꢁꢀ ꢃ ꢎ.ꢓꢊꢊꢉ
ꢕꢖꢋꢐ ꢎ
.ꢀꢊꢓ
ꢅꢇ.ꢇꢂꢉ
Rꢐꢚ
.ꢇꢆꢄ
ꢅꢂ.ꢇꢇꢉ
ꢗꢞꢕ
.ꢀꢊꢀ ꢃ .ꢀꢓꢓ
ꢅꢇ.ꢀꢎꢇ ꢃ ꢇ.ꢄꢎꢀꢉ
.ꢁꢂꢀ ±.ꢀꢀꢄ
ꢅꢆ.ꢀꢂ ±ꢀ.ꢁꢇꢈꢉ
.ꢇꢇꢊ ꢃ .ꢇꢆꢆ
ꢅꢄ.ꢈꢓꢁ ꢃ ꢂ.ꢁꢓꢈꢉ
ꢁ
ꢎ
ꢆ
ꢇ
.ꢀꢎꢀ ±.ꢀꢀꢄ
.ꢁꢁꢊ ꢃ .ꢁꢎꢓ
ꢅꢇ.ꢓꢓꢈ ꢃ ꢎ.ꢄꢄꢀꢉ
ꢅꢀ.ꢈꢂ ±ꢀ.ꢁꢇꢈꢉ
.ꢁꢁꢊ
ꢅꢇ.ꢓꢓꢉ
Rꢐꢚ
ꢋꢌꢍ
Rꢐꢒꢖꢗꢗꢐꢕꢛꢐꢛ ꢏꢖꢜꢛꢐR ꢍꢘꢛ ꢜꢘꢌꢖꢝꢋ
.ꢀꢁꢀ ꢃ .ꢀꢇꢀ
ꢅꢀ.ꢇꢄꢆ ꢃ ꢀ.ꢄꢀꢊꢉ
× ꢆꢄ°
.ꢀꢄꢎ ꢃ .ꢀꢂꢓ
ꢅꢁ.ꢎꢆꢂ ꢃ ꢁ.ꢈꢄꢇꢉ
ꢆ
ꢄ
.ꢀꢀꢆ ꢃ .ꢀꢁꢀ
ꢅꢀ.ꢁꢀꢁ ꢃ ꢀ.ꢇꢄꢆꢉ ꢅꢀ.ꢀ ꢃ ꢀ.ꢁꢎꢀꢉ
ꢀ.ꢀ ꢃ ꢀ.ꢀꢀꢄ
.ꢀꢀꢊ ꢃ .ꢀꢁꢀ
ꢀ°ꢃ ꢊ° ꢋꢌꢍ
ꢅꢀ.ꢇꢀꢎ ꢃ ꢀ.ꢇꢄꢆꢉ
.ꢀꢁꢂ ꢃ .ꢀꢄꢀ
ꢅꢀ.ꢆꢀꢂ ꢃ ꢁ.ꢇꢈꢀꢉ
.ꢀꢄꢀ
ꢅꢁ.ꢇꢈꢀꢉ
ꢔꢏꢒ
.ꢀꢁꢆ ꢃ .ꢀꢁꢓ
ꢅꢀ.ꢎꢄꢄ ꢃ ꢀ.ꢆꢊꢎꢉ
ꢋꢌꢍ
ꢕꢖꢋꢐꢠ
ꢁ. ꢛꢞꢗꢐꢕꢏꢞꢖꢕꢏ ꢞꢕ
ꢏꢊꢐ ꢁꢀꢁꢄ Rꢐꢑ ꢒ
ꢞꢕꢒꢟꢐꢏ
ꢅꢗꢞꢜꢜꢞꢗꢐꢋꢐRꢏꢉ
ꢇ. ꢛRꢘꢡꢞꢕG ꢕꢖꢋ ꢋꢖ ꢏꢒꢘꢜꢐ
ꢎ. ꢋꢟꢐꢏꢐ ꢛꢞꢗꢐꢕꢏꢞꢖꢕꢏ ꢛꢖ ꢕꢖꢋ ꢞꢕꢒꢜꢝꢛꢐ ꢗꢖꢜꢛ ꢚꢜꢘꢏꢟ ꢖR ꢍRꢖꢋRꢝꢏꢞꢖꢕꢏ.
ꢗꢖꢜꢛ ꢚꢜꢘꢏꢟ ꢖR ꢍRꢖꢋRꢝꢏꢞꢖꢕꢏ ꢏꢟꢘꢜꢜ ꢕꢖꢋ ꢐꢙꢒꢐꢐꢛ .ꢀꢁꢀꢢ ꢅꢀ.ꢇꢄꢆꢣꢣꢉ
ꢆ. ꢏꢋꢘꢕꢛꢘRꢛ ꢜꢐꢘꢛ ꢏꢋꢘꢕꢛꢖꢚꢚ ꢞꢏ ꢆꢣꢤꢥꢦ ꢋꢖ ꢁꢀꢣꢤꢥꢦ ꢅꢛꢘꢋꢐ ꢒꢖꢛꢐ ꢔꢐꢚꢖRꢐ ꢄꢆꢇꢉ
ꢄ. ꢜꢖꢡꢐR ꢜꢐꢘꢛ ꢏꢋꢘꢕꢛꢖꢚꢚ ꢞꢏ ꢀꢣꢤꢥꢦ ꢋꢖ ꢄꢣꢤꢥꢦ ꢅꢛꢘꢋꢐ ꢒꢖꢛꢐ ꢘꢚꢋꢐR ꢄꢆꢇꢉ
Rev. G
24
For more information www.analog.com
LT8302/LT8302-3
REVISION HISTORY
REV
DATE
DESCRIPTION
PAGE NUMBER
A
11/14 Modified I and I
conditions.
HYS
3
Q
Modified L equation.
14
23
26
24
PRI
Modified schematic.
Updated Related Parts.
B
C
11/15 Revised package drawing.
9/16
5/19
7/19
Reduced EN/UVLO shutdown threshold.
3
3
Increased I
max current limit.
current limit range.
INTVCC
Changed I
3
SW(MIN)
Corrected I
equation.
20
LOAD(MIN)
D
Changed V minimum from 2.8V to 3V.
1, 3
14
IN
Table 1, Line 5: Replaced Wurth predesigned transformer with Sumida equivalent.
Table 1 Sumida transformer used in 12V
and 12V
Typical Application circuits.
21, 22
OUT
OUT
5V/1.1A (V = 5V) output capability line removed from LT8302/LT8302-3/LT8309 Typical Application circuit.
23
2
IN
E
F
Added AEC-Q100 automotive models.
12/19 Added LT8302-3 Models
All
2, 3
G
04/20 Added J grade option and specifications
Rev. G
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog
Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications
subject to change without notice. No license i rantedbyica r otheris ndany patent or patent rights of Analog Devices.
25
s
g
impltionoweuer
LT8302/LT8302-3
TYPICAL APPLICATION
4V to 42VIN/48VOUT Boost Converter
ꢅꢍ
ꢀ
ꢖꢕꢆ
ꢐꢍ
ꢊꢊꢎꢏ
ꢝꢇꢀꢋꢍ.ꢝꢞ ꢢꢀ ꢣ ꢝꢊꢀꢤ
ꢁꢂ
ꢀ
ꢁꢂ
ꢝꢇꢀꢋꢉ.ꢇꢞ ꢢꢀ ꢣ ꢊꢝꢀꢤ
ꢁꢂ
ꢝꢀ ꢆꢖ ꢝꢊꢀ
ꢝꢇꢀꢋꢉ.ꢝꢞ ꢢꢀ ꢣ ꢍꢊꢀꢤ
ꢁꢂ
ꢝꢇꢀꢋꢉ.ꢍꢚꢞ ꢢꢀ ꢣ ꢚꢀꢤ
ꢁꢂ
ꢀ
ꢃꢄ
ꢁꢂ
ꢔꢂꢋꢕꢀꢅꢖ
R
ꢒꢓ
ꢗꢈ
ꢍꢉꢎꢒ
Rꢈ
ꢍꢟ
Rꢝ
ꢝꢛꢝꢠ
ꢗꢍ
ꢍꢉꢎꢒ
ꢑꢍ
ꢅꢆꢇꢈꢉꢊꢋꢅꢆꢇꢈꢉꢊꢌꢈ
R
Rꢔꢒ
ꢁꢂꢆꢀ
ꢗꢗ
ꢐꢍꢘ ꢐꢁꢖꢐꢔꢃ ꢙꢐꢃꢚꢛꢉ
ꢅꢍꢘ ꢄꢕRꢆꢏ ꢜꢝꢝꢈꢚꢚꢍꢊꢊꢍ
ꢑꢍꢘ ꢗꢔꢂꢆRꢞꢅ ꢗꢟꢏꢑꢚꢊꢛꢊꢓ
Rꢚ
ꢍꢉꢠ
ꢗꢊ
ꢍꢎꢒ
Gꢂꢐ
ꢇꢈꢉꢊ ꢆꢞꢍꢉꢡ
Efficiency vs Load Current
ꢕꢌꢌ
ꢔꢒ
ꢔꢌ
ꢓꢒ
ꢓꢌ
ꢍꢒ
ꢍꢌ
ꢙ
ꢙ
ꢙ
ꢙ
ꢚ ꢒꢙ
ꢏꢇ
ꢏꢇ
ꢏꢇ
ꢏꢇ
ꢚ ꢕꢖꢙ
ꢚ ꢖꢛꢙ
ꢚ ꢛꢖꢙ
ꢌ
ꢖꢒꢌ
ꢒꢌꢌ
ꢍꢒꢌ ꢕꢌꢌꢌ ꢕꢖꢒꢌ ꢕꢒꢌꢌ
ꢀꢁꢂꢃ ꢄꢅRRꢆꢇꢈ ꢉꢊꢂꢋ
ꢓꢗꢌꢖ ꢈꢂꢕꢌꢘ
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT8301
42V Micropower Isolated Flyback Converter with 65V/1.2A
Low I Monolithic No-Opto Flyback 5-Lead TSOT-23
Q
IN
Switch
LT8300
LT8309
100V Micropower Isolated Flyback Converter with
Low I Monolithic No-Opto Flyback, 5-Lead TSOT-23
Q
IN
150V/260mA Switch
Secondary-Side Synchronous Rectifier Driver
40V Isolated Flyback Converters
4.5V ≤ V ≤ 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23
CC
LT3573/LT3574
LT3575
Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A
Switch
LT3511/LT3512
100V Isolated Flyback Converters
Monolithic No-Opto Flybacks with Integrated 240mA/420mA
Switch, MSOP-16(12)
LT3748
LT3798
100V Isolated Flyback Controller
5V ≤ V ≤ 100V, No-Opto Flyback, MSOP-16(12)
IN
Off-Line Isolated No-Opto Flyback Controller with Active PFC
40V/100V Flyback/Boost Controllers
V
IN
and V
Limited Only by External Components
OUT
LT3757A/LT3759
LT3758
Universal Controllers with Small Package and Powerful Gate Drive
LT3957/LT3958
40V/80V Boost/Flyback Converters
Monolithic with Integrated 5A/3.3A Switch
Rev. G
04/20
www.analog.com
ANALOG DEVICES, INC. 2013-2020
26
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