SSM2475S [ADI]
Rail-to-Rail Output Audio Amplifiers; 轨至轨输出音频放大器
| 型号： | SSM2475S |
| 厂家： | 
ADI |
| 描述： | Rail-to-Rail Output Audio Amplifiers |
| 文件： | 总16页 (文件大小：194K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Rail-to-Rail Output
Audio Amplifiers
a
SSM2275/SSM2475*
FEATURES
PIN CONFIGURATIONS
Single or Dual-Supply Operation
Excellent Sonic Characteristics
Low Noise: 7 nV/√Hz
8-Lead Narrow Body SOIC 14-Lead Narrow Body SOIC
(SO-8)
(R-14)
Low THD: 0.0006%
Rail-to-Rail Output
1
2
3
4
8
7
6
5
14
13
12
1
2
3
4
5
6
7
OUT A
–IN A
OUT D
–IN D
+IN D
V+
OUT A
–IN A
+IN A
V+
High Output Current: ؎50 mA
Low Supply Current: 1.7 mA/Amplifier
Wide Bandwidth: 8 MHz
High Slew Rate: 12 V/s
No Phase Reversal
OUT B
–IN B
SSM2275
(Not to Scale)
+IN A
V–
SSM2475
(Not to Scale)
11 V–
+IN B
10
9
+IN C
–IN C
OUT C
+IN B
–IN B
OUT B
Unity Gain Stable
Stable Parameters Over Temperature
8
APPLICATIONS
Multimedia Audio
Professional Audio Systems
High Performance Consumer Audio
Microphone Preamplifier
MIDI Instruments
8-Lead microSOIC
(RM-8)
14-Lead TSSOP
(RU-14)
OUT A
–IN A
+IN A
V+
+IN B
–IN B
OUT B
OUT D
1
14
OUT A
–IN A
+IN A
V–
V+
1
8
–IN D
+IN D
V–
OUT B
–IN B
+IN B
SSM2275
SSM2475
4
5
+IN C
–IN C
OUT C
GENERAL DESCRIPTION
The SSM2275 and SSM2475 use the Butler Amplifier front
end, which combines both bipolar and FET transistors to offer
the accuracy and low noise performance of bipolar transistors
and the slew rates and sound quality of FETs. This product
family includes dual and quad rail-to-rail output audio amplifi-
ers that achieve lower production costs than the industry stan-
dard OP275 (the first Butler Amplifier offered by Analog
Devices). This lower cost amplifier also offers operation from a
single 5 V supply, in addition to conventional ±15 V supplies.
The ac performance meets the needs of the most demanding au-
dio applications, with 8 MHz bandwidth, 12 V/µs slew rate and
extremely low distortion.
7
8
8-Lead Plastic DIP
(N-8)
OUT A
–IN A
+IN A
V–
1
2
3
4
8
7
6
5
V+
(Not to
Scale)
OUT B
–IN B
+IN B
SSM2275
The SSM2275 and SSM2475 are ideal for application in high
performance audio amplifiers, recording equipment, synthesiz-
ers, MIDI instruments and computer sound cards. Where cas-
caded stages demand low noise and predictable performance,
SSM2275 and SSM2475 are a cost effective solution. Both are
stable even when driving capacitive loads.
The ability to swing rail-to-rail at the outputs (see Applications sec-
tion) and operate from low supply voltages enables designers to at-
tain high quality audio performance, even in single supply systems.
The SSM2275 and SSM2475 are specified over the extended
industrial (–40°C to +85°C) temperature range. The SSM2275 is
available in 8-lead plastic DIPs, SOICs, and microSOIC surface-
mount packages. The SSM2475 is available in narrow body
SOICs and thin shrink small outline (TSSOP) surface-mount
packages.
*Protected by U.S. Patent No. 5,101,126.
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood. MA 02062-9106, U.S.A.
Tel: 781/329-4700
Fax: 781/326-8703
World Wide Web Site: http://www.analog.com
© Analog Devices, Inc., 1999
SSM2275/SSM2475–SPECIFICATIONS
ELECTRICAL CHARACTERISTICS
(VS = ؎15 V, TA = ؉25؇C, VCM = 0 V unless otherwise noted)
Parameter
Symbol Conditions
Min
Typ Max
Units
INPUT CHARACTERISTICS
Offset Voltage
VOS
1
1
250
300
5
4
6
400
500
75
125
+14
mV
mV
nA
nA
nA
nA
V
–40°C ≤ TA ≤ +85°C
Input Bias Current
Input Offset Current
IB
–40°C ≤ TA ≤ +85°C
IOS
–40°C ≤ TA ≤ +85°C
VS = ±15 V
–12.5 V ≤ VCM ≤ +12.5 V
–40°C ≤ TA ≤ +85°C,
–12.5 V ≤ VCM ≤ +12.5 V
RL = 2 kΩ, –12 V ≤ VO ≤ +12 V
–40°C ≤ TA ≤ +85°C
15
Input Voltage Range
Common-Mode Rejection Ratio
VIN
CMRR
–14
80
100
dB
80
100
80
100
240
120
V/mV
V/mV
V/mV
AVO
OUTPUT CHARACTERISTICS
Output Voltage, High
VOH
VOL
IL ≤ 20 mA
–40°C ≤ TA ≤ +85°C
IL = 20 mA
14
14.5
14.5
14.7
–14
V
V
V
Output Voltage, Low
–13.5
IL = 10 mA
IL = 10 mA, –40°C ≤ TA ≤ +85°C
–14.6 –14.4
–14.3 –13.9
V
V
Output Short Circuit Current Limit
ISC
±25
±17
±50
±40
±75
±80
mA
mA
–40°C ≤ TA ≤ +85°C
POWER SUPPLY
Power Supply Rejection Ratio
PSRR
ISY
±2.5 V ≤ VS ≤ ±18 V
–40°C ≤ TA ≤ +85°C
VO = 0 V
85
80
110
105
1.7
dB
dB
mA
mA
Supply Current/Amplifier
2.9
–40°C ≤ TA ≤ +85°C
1.75 3.0
DYNAMIC PERFORMANCE
Total Harmonic Distortion
Slew Rate
Gain Bandwidth Product
Channel Separation
THD
SR
GBW
CS
RL = 10 kΩ, f = 1 kHz, VO = 1 V rms
RL = 2 kΩʈ50 pF
0.0006
12
8
%
9
V/µs
MHz
dB
RL = 2 kΩ, f =1 kHz
128
NOISE PERFORMANCE
Voltage Noise Spectral Density
Current Noise Spectral Density
en
in
f > 1 kHz
f > 1 kHz
8
< 1
nV/√Hz
pA/√Hz
Specifications subject to change without notice.
–2–
REV. A
SSM2275/SSM2475
(VS = ؉5 V, TA = ؉25؇C, VCM = 2.5 V unless otherwise noted)
ELECTRICAL CHARACTERISTICS
Parameter
Symbol
Conditions
Min
Typ Max
Units
INPUT CHARACTERISTICS
Offset Voltage
VOS
IB
1
1
250
300
5
4
6
400
500
75
125
4.7
mV
mV
nA
nA
nA
nA
V
–40°C ≤ TA ≤ +85°C
–40°C ≤ TA ≤ +85°C
–40°C ≤ TA ≤ +85°C
Input Bias Current
Input Offset Current
IOS
15
Input Voltage Range
VIN
0.3
Common-Mode Rejection Ratio
CMRR
+0.8 V ≤ VCM ≤ +2 V
–40°C ≤ TA ≤ +85°C
RL = 2 kΩ, –0.5 V ≤ VO ≤ +4.5 V
–40°C ≤ TA ≤ +85°C
85
80
60
50
dB
dB
V/mV
V/mV
AVO
25
20
OUTPUT CHARACTERISTICS
Output Voltage, High
VOH
VOL
IL ≤ –15 mA
IL ≤ –10 mA, –40°C ≤ TA ≤ +85°C
IL ≤ –15 mA
IL ≤ –10 mA
IL ≤ –10 mA, –40°C ≤ TA ≤ +85°C
–40°C ≤ TA ≤ +85°C
4.2
4.5
4.5
4.8
0.6
0.3
0.7
40
V
V
V
V
V
mA
Output Voltage, Low
1.0
0.5
1.1
Output Short Circuit Current Limit
ISC
POWER SUPPLY
Supply Current/Amplifier
ISY
VO = 0 V
–40°C ≤ TA ≤ +85°C
1.7
1.75 3.0
2.9
mA
mA
DYNAMIC PERFORMANCE
Total Harmonic Distortion
Slew Rate
Gain Bandwidth Product
Channel Separation
THD
SR
GBW
CS
RL = 10 kΩ, f = 1 kHz, VO = 1 V rms
RL = 2 kΩʈ50 pF
0.0006
12
6
%
V/µs
MHz
dB
RL = 2 kΩʈ10 pF
RL = 2 kΩ, f =1 kHz
128
NOISE PERFORMANCE
Voltage Noise Spectral Density
Current Noise Spectral Density
en
in
f > 1 kHz
f > 1 kHz
8
< 1
nV/√Hz
pA/√Hz
Specifications subject to change without notice.
REV. A
–3–
SSM2275/SSM2475
ABSOLUTE MAXIMUM RATINGS1
Package Type
JA*
Units
JC
Supply Voltage (VS) . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±18 V
Input Voltage (VIN) . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±15 V
Differential Input Voltage2 . . . . . . . . . . . . . . . . . . . . . . . ±15 V
Storage Temperature Range . . . . . . . . . . . . Ϫ65°C to ϩ150°C
Operating Temperature Range . . . . . . . . . . . Ϫ40°C to ϩ85°C
Junction Temperature Range . . . . . . . . . . . . Ϫ65°C to ϩ150°C
Lead Temperature Range (Soldering, 60 sec) . . . . . . . ϩ300°C
ESD Susceptibility. . . . . . . . . . . . . . . . . . . . . . . . . . . . 2,000 V
8-Lead Plastic DIP
8-Lead SOIC
8-Lead microSOIC
14-Lead SOIC
103
158
206
120
180
43
43
43
36
35
°C/W
°C/W
°C/W
°C/W
°C/W
14-Lead TSSOP
*θJA is specified for the worst case conditions, i.e., for device in socket for DIP
packages and soldered onto a circuit board for surface mount packages.
NOTES
1Stresses above those listed under Absolute Maximum Ratings may cause perma-
nent damage to the device. This is a stress rating only; the functional operation of
the device at these or any other conditions above those indicated in the operational
sections of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2For supplies less than ± 15 V, the input voltage and differential input voltage
must be less than ±15 V.
ORDERING GUIDE
Temperature
Range
Package
Description
Package
Options
Model
SSM2275P
SSM2275S
SSM2275RM
SSM2475S
SSM2475RU
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
8-Lead PDIP
8-Lead SOIC
8-Lead microSOIC
14-Lead SOIC
14-Lead TSSOP
N-8
SO-8
RM-8
R-14
RU-14
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the SSM2275/SSM2475 features proprietary ESD protection circuitry, permanent
damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper
ESD precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
100
225
180
135
90
100
80
225
180
135
90
V
R
C
= ؎2.5V
= 600⍀
= 10pF
S
V
R
C
= ؎2.5V
= 2k⍀
= 10pF
S
L
L
L
L
80
60
60
40
40
20
45
20
45
0
0
0
0
–20
–40
–45
–90
–20
–40
–45
–90
10
100
1k
10k
100k
1M
10M 40M
10
100
1k
10k
100k
1M
10M 40M
FREQUENCY – Hz
FREQUENCY – Hz
Figure 1. Phase/Gain vs. Frequency
Figure 2. Phase/Gain vs. Frequency
–4–
REV. A
Typical Characteristics–SSM2275/SSM2475
100
80
225
180
135
90
60
V
R
C
= ؎15V
= 2k⍀
= 10pF
V
T
= ؎15V
= ؉25؇C
S
S
L
L
A
50
40
30
20
10
0
60
40
20
45
0
0
–20
–40
–45
–90
10
100
1k
10k
100k
1M
10M 40M
10
100
1k
10k
100k
FREQUENCY – Hz
FREQUENCY – Hz
Figure 3. Phase/Gain vs. Frequency
Figure 6. SSM2275 Voltage Noise Density (Typical)
100
80
225
180
135
90
140
V
T
= ؎15V
= ؉25؇C
S
A
V
R
C
= ؎15V
= 600⍀
= 10pF
S
L
L
120
100
80
60
40
20
0
60
40
20
45
0
0
–20
–40
–45
–90
10
100
1k
10k
100k
1M
10M 40M
100
1k
10k
1M
10M
30M
FREQUENCY – Hz
FREQUENCY – Hz
Figure 4. Phase/Gain vs. Frequency
Figure 7. Common-Mode Rejection vs. Frequency
2.0
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
140
V
T
= ؎15V
= ؉25؇C
V
T
= ؎15V
= ؉25؇C
S
S
A
A
120
100
80
60
40
20
0
10
100
1k
FREQUENCY – Hz
10k
100
1k
10k
1M
10M
FREQUENCY – Hz
Figure 5. SSM2275 Current Noise Density vs. Frequency
Figure 8. Power Supply Rejection vs. Frequency
REV. A
–5–
SSM2275/SSM2475–Typical Characteristics
1.0
0.9
0.8
0.7
0.6
0.5
–100
V
A
R
= +5V
= +1
= 100k⍀
= 0dBV
SY
V
L
–110
–120
–130
–140
–150
–160
V
IN
V
– (V+)
OUT
0.4
0.3
V
– (V–)
OUT
0.2
0.1
0
5
10
15
20
0
2
4
6
8
10
12
14
16
18
20 22
LOAD CURRENT – mA
FREQUENCY – kHz
Figure 12. Headroom (VOH and VOL-to-Rails), TA = +25°C
Figure 9. THD vs. Frequency (FFT)
29.5mV
11.0mV
0
0
0
0
20mV
200ns
20mV
200ns
Figure 13. Small Signal Response; RL = 600 Ω, CL = 200 pF,
VS = ±2.5 V, AV = +1, VIN = 100 mV p-p
Figure 10. Small Signal Response; RL = 600 Ω, CL = 0 pF,
VS = ±2.5 V, AV = +1, VIN = 100 mV p-p
35.5mV
22.5mV
0
0
0
0
20mV
200ns
20mV
200ns
Figure 11. Small Signal Response; RL = 600 Ω, CL = 100 pF,
VS = ±2.5 V, AV = +1, VIN = 100 mV p-p
Figure 14. Small Signal Response; RL = 600 Ω, CL = 300 pF,
VS = ±2.5 V, AV = +1, VIN = 100 mV p-p
–6–
REV. A
SSM2275/SSM2475
43.0mV
17.5mV
0
0
0
0
20mV
20mV
200ns
200ns
Figure 15. Small Signal Response; RL = 2 kΩ, CL = 0 pF,
VS = ±2.5 V, AV = +1, VIN = 100 mV p-p
Figure 18. Small Signal Response; RL = 2 kΩ, CL = 300 pF,
VS = ±2.5 V, AV = +1, VIN = 100 mV p-p
10.5mV
31.0mV
0
0
0
0
20mV
20mV
200ns
100ns
Figure 16. Small Signal Response; RL = 2 kΩ, CL = 100 pF,
VS = ±2.5 V, AV = +1, VIN = 100 mV p-p
Figure 19. Small Signal Response; RL = 600 Ω, CL = 0 pF,
VS = ±15 V, AV = +1, VIN = 100 mV p-p
38.0mV
22.5mV
0
0
0
0
20mV
20mV
200ns
100ns
Figure 17. Small Signal Response; RL = 2 kΩ, CL = 200 pF,
VS = ±2.5 V, AV = +1, VIN = 100 mV p-p
Figure 20. Small Signal Response; RL = 600 Ω, CL = 100 pF,
VS = ±15 V, AV = +1, VIN = 100 mV p-p
REV. A
–7–
SSM2275/SSM2475–Typical Characteristics
28.0mV
29.0mV
0
0
0
0
20mV
20mV
200ns
100ns
Figure 21. Small Signal Response; RL = 600 Ω, CL = 200 pF,
VS = ±15 V, AV = +1, VIN = 100 mV p-p
Figure 24. Small Signal Response; RL = 2 kΩ, CL = 100 pF,
VS = ±15 V, AV = +1, VIN = 100 mV p-p
36.5mV
35.5mV
0
0
0
0
20mV
20mV
200ns
100ns
Figure 22. Small Signal Response; RL = 600 Ω, CL = 300 pF,
VS = ±15 V, AV = +1, VIN = 100 mV p-p
Figure 25. Small Signal Response; RL = 2 kΩ, CL = 200 pF,
VS = ±15 V, AV = +1, VIN = 100 mV p-p
13.0mV
42.0mV
0
0
0
0
20mV
20mV
100ns
200ns
Figure 23. Small Signal Response; RL = 2 kΩ, CL = 0 pF,
VS = ±15 V, AV = +1, VIN = 100 mV p-p
Figure 26. Small Signal Response; RL = 2 kΩ, CL = 300 pF,
VS = ±15 V, AV = +1, VIN = 100 mV p-p
–8–
REV. A
SSM2275/SSM2475
±7 V, then the input current should be limited to less than
±5 mA. This can be easily done by placing a resistor in series
with both inputs. The minimum value of the resistor can be
determined by:
THEORY OF OPERATION
The SSM2275 and SSM2475 are low noise and low distortion
rail-to-rail output amplifiers that are excellent for audio applica-
tions. Based on the OP275 audiophile amplifier, the SSM2275/
SSM2475 offers many similar performance characteristics with
the advantage of a rail-to-rail output from a single supply
source. Its low input voltage noise figure of 7 nV/√Hz allows the
device to be used in applications requiring high gain, such as
microphone preamplifiers. Its 11 V/µs slew rate also allows the
SSM2275/SSM2475 to produce wide output voltage swings
while maintaining low distortion. In addition, its low harmonic
distortion figure of 0.0006% makes the SSM2275 and
SSM2475 ideal for high quality audio applications.
VDIFF, MAX − 7
RIN
=
(1)
0.01
There are also ESD protection diodes that are connected from
each input to each power supply rail. These diodes are normally
reversed biased, but will turn on if either input voltage exceeds
either supply rail by more than 0.6 V. Again, should this condi-
tion occur the input current should be limited to less than
±5 mA. The minimum resistor value should then be:
Figure 27 shows the simplified schematic for a single amplifier.
The amplifier contains a Butler Amplifier at the input. This
front-end design uses both bipolar and MOSFET transistors in
the differential input stage. The bipolar devices, Q1 and Q2,
improve the offset voltage and achieve the low noise perfor-
mance, while the MOS devices, M1 and M2, are used to obtain
higher slew rates. The bipolar differential pair is biased with a
proportional-to-absolute-temperature (PTAT) bias source, IB1,
while the MOS differential pair is biased with a non-PTAT
source, IB2. This results in the amplifier having a constant gain-
bandwidth product and a constant slew rate over temperature.
VIN , MAX
RIN
=
(2)
5mA
In practice, RIN should be placed in series with both inputs to
reduce offset voltages caused by input bias current. This is
shown in Figure 28.
V+
R
IN
The amplifier also contains a rail-to-rail output stage that can
sink or source up to 50 mA of current. As with any rail-to-rail
output amplifier the gain of the output stage, and consequently
the open loop gain of the amplifier, is proportional to the load
resistance. With a load resistance of 50 kΩ, the dc gain of the
amplifier is over 110 dB. At load currents less than 1 mA, the
output of the amplifier can swing to within 30 mV of either sup-
ply rail. As load current increases, the maximum voltage swing
of the output will decrease. This is due to the collector to emit-
ter saturation voltage of the output transistors increasing with an
increasing collector current.
R
IN
V–
Figure 28. Using Resistors for Input Overcurrent Protection
Output Voltage Phase Reversal
The SSM2275/SSM2475 was designed to have a wide common-
mode range and is immune to output voltage phase reversal with
an input voltage within the supply voltages of the device. How-
ever, if either of the device’s inputs exceeds 0.6 V above the posi-
tive voltage supply, the output could exhibit phase reversal.
This is due to the input transistor’s B–C junction becoming for-
ward biased, causing the polarity of the input terminals of the
device to switch.
Input Overvoltage Protection
The maximum input differential voltage that can be applied to
the SSM2275/SSM2475 is ±7 V. A pair of internal back-to-back
Zener diodes are connected across the input terminals. This
prevents emitter-base junction breakdown from occurring to the
input transistors, Q1 and Q2, when very large differential volt-
ages are applied. If the device’s differential voltage could exceed
V
CC
IB2
Q2
IN+
IN–
Q1
OUT
CFI
M2
M1
IB1
V
EE
Figure 27. Simplified Schematic
–9–
REV. A
SSM2275/SSM2475
This phase reversal can be prevented by limiting the input cur-
rent to +1 mA. This can be done by placing a resistor in series
with the input terminal that is expected to be overdriven. The
series resistance should be at least:
For example, with the 8-lead SOIC, the calculation gives a
maximum internal power dissipation (for all amplifiers, worst
case) of PMAX = (150°C – 85°C)/158°C/W = 0.41 W. For the
DIP package, a similar calculation indicates that 0.63 W (ap-
proximately 50% more) can be safely dissipated. Note that am-
bient temperature is defined as the temperature of the PC board
to which the device is connected (in the absence of radiated or
convected heat loss). It is good practice to place higher power
devices away from the more sensitive circuits. When in doubt,
measure the temperature in the vicinity of the SSM2275 with a
thermocouple thermometer.
VIN , MAX − 0.6
RIN
=
(3)
1 mA
An equivalent resistor should be placed in series with both in-
puts to prevent offset voltages due to input bias currents, as
shown in Figure 28.
Output Short Circuit Protection
Maximizing Low Distortion Performance
To achieve high quality rail-to-rail performance, the output of
the SSM2275/SSM2475 is not short-circuit protected. Shorting
the output may damage or destroy the device when excessive
voltages or currents are applied. To protect the output stage, the
maximum output current should be limited to ±40 mA. Placing
a resistor in series with the output of the amplifier as shown in
Figure 29, the output current can be limited. The minimum
value for RX can be found from Equation 4.
Because the SSM2275/SSM2475 is a very low distortion amplifier,
careful attention should be given to the use of the device to prevent
inadvertently introducing distortion. Source impedances seen by
both inputs should be made equal, as shown in Figure 28, with
RB = R1ʈRF for minimum distortion. This eliminates any offset
voltages due to varying bias currents. Proper power supply
decoupling reduces distortion due to power supply variations.
Because the open loop gain of the amplifier is directly dependent
on the load resistance, loads of less than 10 kΩ will increase the
distortion of the amplifier. This is a trait of any rail-to-rail op
amp. Increasing load capacitance will also increase distortion.
VSY
RX
=
(4)
40mA
It is recommended that any unused amplifiers be configured as a
unity gain follower with the noninverting input tied to ground.
This minimizes the power dissipation and any potential crosstalk
from the unused amplifier.
For a +5 V single supply application, RX should be at least
125 Ω. Because RX is inside the feedback loop, VOUT is not
affected. The trade off in using RX is a slight reduction in output
voltage swing under heavy output current loads. RX will also
increase the effective output impedance of the amplifier to
RO + RX, where RO is the output impedance of the device.
As with many FET-type amplifiers, the PMOS devices in the
input stage exhibit a gate-to-source capacitance that varies with
the common mode voltage. In an inverting configuration, the in-
verting input is held at a virtual ground and the common-mode
voltage does not vary. This eliminates distortion due to input
capacitance modulation. In noninverting applications, the gate-
to-source voltage is not constant, and the resulting capacitance
modulation can cause a slight increase in distortion.
R
FB
FEEDBACK
V
RX
125⍀
A1
OUT
A1 = 1/2 SSM2275
Figure 30 shows a unity gain inverter and a unity gain follower
configuration. Figure 31 shows an FFT of the outputs of these
amplifiers with a 1 kHz sine wave. Notice how the largest har-
monic amplitude (2nd harmonic) is –120 dB below the funda-
mental (0.0001%) in the inverting configuration.
Figure 29. Output Short Circuit Protection Configuration
Power Dissipation Considerations
While many designers are constrained to use very small and low
profile packages, reliable operation demands that the maximum
junction temperatures not be exceeded. A simple calculation
will ensure that your equipment will enjoy reliable operation
over a long lifetime. Modern IC design allows dual and quad
amplifiers to be packaged in SOIC and microSOIC packages,
but it is the responsibility of the designer to determine what the
actual junction temperature will be, and prevent it from exceed-
ing the 150°C. Note that while the θJC is similar between pack-
age options, the θJA for the SOIC and TSSOP are nearly double
the PDIP. The calculation of maximum ambient temperature is
relatively simple to make.
V+
V+
10F
10F
0.1F
0.1F
R
R
FB
FB
R1
R1
SSM2275
SSM2275
V
IN
R
R
V
V
OUT
B
B
OUT
V
IN
R
R
L
L
0.1F
0.1F
10F
10F
TI , MAX −TA
V–
V–
PMAX
=
(5)
Figure 30. Basic Inverting and Noninverting Amplifiers
θJA
–10–
REV. A
SSM2275/SSM2475
10
8
–100
–110
–120
–130
–140
–150
–160
V
A
R
= ؎5V
= ؉1
= 100k⍀
= 0dBV
SY
V
L
؉0.1%
؉0.01%
6
V
IN
4
2
0
0
10
FREQUENCY – kHz
20 22
–2
–4
–100
–110
–120
–130
–140
–150
–160
V
= ؉5V
= –1
= 100k⍀
= 0dBV
SY
A
R
V
L
–6
–8
–0.01%
V
IN
–10
400
600
800
1000
1200
SETTLING TIME – ns
Figure 33. Settling Time vs. Step Size
Overdrive Recovery
0
10
FREQUENCY – kHz
20 22
The overdrive, or overload, recovery time of an amplifier is the time
required for the output voltage to return to a rated output voltage
from a saturated condition. This recovery time can be important in
applications where the amplifier must recover quickly after a large
transient event, or overload. The circuit in Figure 34 was used to
evaluate the recovery time for the SSM2275/SSM2475. Also shown
are the input and output voltages. It takes approximately 0.5 µs for
the device to recover from output overload.
Figure 31. Spectral Graph of Amplifier Outputs
Settling Time
The high slew rate and wide gain-bandwidth product of the
SSM2275 and SSM2475 amplifiers result in fast settling times
(tS < 1 µs) that are suitable for 16- and 20-bit applications. The
test circuit used to measure the settling time of the SSM2275/
SSM2475 is shown in Figure 32. This test method has advan-
tages over false-sum node techniques of measuring settling times
in that the actual output of the amplifier is measured, instead of
an error voltage at the sum node. Common-mode settling ef-
fects are also taken into account in this circuit in addition to
slew rate and bandwidth factors.
+5V
R
S
909⍀
V
V
OUT
IN
R
L
2V p-p
10kHz
R1
1k⍀
10k⍀
The output waveform of the device under test is clamped by
Schottky diodes and buffered by the JFET source follower. The
signal is amplified by a factor of ten by the OP260 current feed-
back amplifier and then Schottky-clamped at the output to the
oscilloscope. The 2N2222 transistor sets up the bias current for
the JFET and the OP41 is configured as a fast integrator, pro-
viding overall dc offset nulling at the output.
–5V
R
F
10k⍀
Figure 34. Overload Recovery Time Test Circuit
9V–15V
–
+
+15V
1k⍀
OUTPUT
(TO SCOPE)
0.1F
R
L
D3
D4
V+
1k⍀
2N4416
DUT
V–
1/2 OP260AJ
1F
D1
D2
0.1F
R
F
2k⍀
10k⍀
10k⍀
+
–
IC2
9V–15V
R
G
222⍀
؎5V
2N2222A
750⍀
1N4148
15k⍀
SCHOTTKY DIODES D1–D4 ARE
HEWLETT-PACKARD HP5082-2835
IC1 IS 1/2 OP260AJ
–15V
IC2 IS PMI OP41EJ
Figure 32. Settling Time Test Fixture
REV. A
–11–
SSM2275/SSM2475
Capacitive Loading
For the values given in Figure 36, RIN = 5 kΩ. With C1 omitted
the circuit will provide a balanced output down to dc, otherwise
the –3 dB corner for the input frequency is set by:
The output of the SSM2275/SSM2475 can tolerate a degree of
capacitive loading. However, under certain conditions, a heavy
capacitive load could create excess phase shift at the output and
put the device into oscillation. The degree of capacitive loading
is dependent on the gain of the amplifier. At unity gain, the am-
plifier could become unstable at loads greater than 600 pF. At
gain greater than unity, the amplifier can handle a higher degree
of capacitive load without oscillating. Figure 35 shows how to
configure the device to prevent oscillations from occurring.
1
f−3dB
=
(8)
2π RINCL
The circuit can also be configured to provide additional gain if
desired. The gain of the circuit is:
VOUT 2(R2)
A =
V
=
C
C
FB
(9)
FB
VIN
R1
R
R
FB
FB
where VOUT = VO1 – VO2, R1 = R3 = R5 = R7 and,
R2 = R4 = R6 = R8
R
R
R
R
I
I
V
IN
V
V
OUT
OUT
Figure 37 shows the THD+N versus frequency response of the
circuit while driving a 600 Ω load at 1 V rms.
V
IN
C
C
L
B
L
B
SSM2275
SSM2275
50k⍀
50k⍀
C3
33pF
INVERTING GAIN AMPLIFIER
NONINVERTING GAIN AMPLIFIER
Figure 35. Configurations for Driving Heavy Capacitive
Loads
R1
10k⍀
R2
10k⍀
RB should be at least 50 kΩ. To minimize offset voltage, the
parallel combination of RFB and RI should be equal to RB. Set-
ting a minimum CF of 15 pF bandlimits the amplifier enough to
eliminate any oscillation problems from any sized capacitive
load. The low-pass frequency is determined by:
+12V
R9
50⍀
C1*
10F
R5
10k⍀
V
01
R11
10k⍀
R12
10k⍀
V
IN
SSM2475-A
+12V
R13
+5V
1
C4
33pF
f−3dB
=
(6)
R6
10k⍀
2π RFBCF
100k⍀
R7
10k⍀
R8
10k⍀
C2
10F
R14
100k⍀
With RFB = 50 kΩ and CF = 15 pF, this results in an amplifier
with a 210 kHz bandwidth that can be used with any capacitive
load. If the amplifier is being used in a noninverting unity gain
configuration and RI is omitted, CFB should be at least 100 pF.
If the offset voltage can be tolerated at the output, RFB can be
replaced by a short and CFB can be removed entirely. With the
typical input bias current of 200 nA and RB = 50 kΩ, the in-
crease in offset voltage would be 10 mV. This configuration will
stabilize the amplifier under all capacitive loads.
C3
10F
SSM2475-C
+12V
R10
50⍀
R3
10k⍀
V
02
C1* IS OPTIONAL
SSM2475-B
R4
C4
10F
10k⍀
Single Supply Differential Line Driver
Figure 36. A Low Noise, Single Supply Differential
Line Driver
Figure 36 shows a single supply differential line driver circuit
that can drive a 600 Ω load with less than 0.001% distortion.
The design mimics the performance of a fully balanced trans-
former based solution. However, this design occupies much less
board space while maintaining low distortion and can operate
down to dc. Like the transformer based design, either output
can be shorted to ground for unbalanced line driver applications
without changing the circuit gain of 1.
0.1
V
= 12V
SY
R
= 600⍀
L
0.01
0.001
R13 and R14 set up the common-mode output voltage equal to
half of the supply voltage. C1 is used to couple the input signal
and can be omitted if the input’s dc voltage is equal to half of
the supply voltage. The minimum input impedance of the cir-
cuit as seen from VIN is:
RIN = R1+ R5 || R3+ R7 || R11
(7)
(
) (
)
0.0001
20
100
1k
10k 20k
FREQUENCY – Hz
Figure 37. THD+N vs. Frequency of Differential Line Driver
–12–
REV. A
SSM2275/SSM2475
Multimedia Soundcard Microphone Preamplifier
The AD1862 has a built in 3 kΩ resistor that is connected from
the inverting input to the output of the amplifier. The full-scale
output current of the AD1862 is ±1 mA, resulting in a maximum
output voltage of ±3 V. Additional feedback resistance can be
added in the feedback loop to increase the output voltage. With
RFB connected the maximum output voltage will be:
The low distortion and low noise figures of the SSM2275 make
it an excellent device for amplifying low level audio signals. Fig-
ure 38 shows how the SSM2275 can be configured as a stereo
microphone preamplifier driving the input to a multimedia
sound codec, the AD1848. The SSM2275 can be powered from
the same +5 V single supply as the AD1848. The VREF pin on
the AD1848 provides a bias voltage of 2.25 V for the SSM2275.
This voltage can also be used to provide phantom power to a
condenser microphone through a 2N4124 transistor buffer and
2 kΩ resistors. The phantom power circuitry can be omitted for
dynamic microphones. The gain of SSM2275 amplifiers is set
by R2/R1 which is 100 (40 dB) as shown. Figure 39 shows the
device’s THD+N performance with a 1 VRMS output.
VOUT ,MAX =1mA× 3kΩ + R
(10)
(
)
FB
+12V
16
V
CC
AD1862
12
11
SSM2275-A
ACOM
TO LPF
100pF
I
OUT
R2
10k⍀
10
R
F
R
FB
+5V
10F
(OPTIONAL)
R1
100⍀
L CHANNEL
MIC IN
2
3
NOTE: ADDITIONAL PIN CONNECTIONS OMITTED FOR CLARITY
8
1
29
10F
LMIC
Figure 40. A High Performance I-V Converter for a 20-Bit DAC
1/2
35/36
SSM2275
4
+5V
V
CC
10k⍀
2k⍀
In Figure 41, the SSM2275 is used as a low-pass filter for one
channel of the AD1855, a 24-bit 96 kHz stereo sigma-delta
DAC, which uses a complementary voltage output. The filter is
configured as a second order low-pass Bessel filter with a cutoff
frequency of 50 kHz. This provides a phase linear response from
dc to 24 kHz, which is ideal for high quality audio applications.
The SSM2275 can be connected to the same +5 V power sup-
ply source, that the AD1855 is connected to, eliminating the
need for extra power circuitry. The FILT output (Pin 14) from
the AD1855 provides a common reference voltage equal to half
of the supply voltage for the SSM2275.
0.1F
+5V
34/37
32
GND
2N4124
V
REF
R CHANNEL
MIC IN
0.1F
10F
AD1848
RMIC
2k⍀
10k⍀
5
6
7
28
10F
1/2
R1
100⍀
SSM2275
R2
10k⍀
Figure 38. Low Noise Microphone Preamplifier for
Multimedia Soundcard Codec
Amplifier A1 is used as a unity-gain inverter for the positive out-
put of the AD1855. The output of A1 is combined with a nega-
tive output of the AD1855 into the active low pass filter around
A2. The output impedance of each output of the AD1855 is
100 Ω which must be taken into account to achieve proper dc
gain, which in Figure 41 is unity gain. In this configuration the
SSM2275 can drive reasonable capacitive loads, making the de-
vice suitable for the RCA jack line outputs found in most con-
sumer audio equipment.
1
A
= +40dB
V
V
V
R
B
= ±2.5V
= –40dBV
> 10k⍀
SY
IN
L
= 22kHz
W
0.1
0.01
+5V
28
V
DD
18
4.7nF
A2
V
562⍀
DD
237⍀
10F
1.05k⍀
1.15k⍀
13 OR 18
12 OR 17
14
OUT–
OUT
0.001
AD1855
OUT+
20
100
1k
10k 20k
1.15k⍀
1.05k⍀
FREQUENCY – Hz
A1
Figure 39. THD+N vs. Frequency (VSY = +5 V, AV = 40 dB,
OUT = 1 V rms)
V
FILT
High Performance I-V Converters and Filters for 20-Bit DACs
Because of the increasing resolution and lower harmonic distor-
tions required by more audio applications, the need for high
quality amplifiers at the output of D/A converters becomes criti-
cal. The SSM2275 and SSM2475 can be used as current-to-
voltage converters and smoothing filters for 18- and 20-bit
DACs, achieving 0.0006% THD+N figures while running from
the same +5 V or +12 V source used to power the D/A con-
verter. Figure 40 shows how the SSM2275 can be used with the
AD1862, a current output 20-bit DAC.
0.1F
10F
1
A1 AND A2 ARE SSM2275
OR 1/2 SSM2475
GND
GND
15
NOTE: ADDITIONAL PIN CONNECTIONS
OMITTED FOR CLARITY
+5V
Figure 41. Low-Pass Filter for a 24-Bit Stereo Sigma-
Delta DAC
REV. A
–13–
SSM2275/SSM2475
SPICE Macro-model
A secondary pole section is also set up to vary the gain band-
width product and phase margin of the model based on the
supply voltage. The H1 and VR1 sources set up an equivalent
resistor that is linearly varied with supply voltage. This equiva-
lent resistance, in parallel with C2, creates the secondary pole.
G2 is also linearly varied to increase the GBW at higher supply
voltages. With a supply voltage of 5 V, the gain bandwidth
product is 6.3 MHz with a 47 degree phase margin. At a 30 V
supply voltage, the GBW product moves out to 7.5 MHz with
48° phase margin.
The SPICE macro-model for the SSM2275 is shown in Listing
1 on the following page. This model is based on typical values
for the device and can be downloaded from Analog Devices’
Internet site at www.analog.com. The model uses a common
emitter output stage to provide rail-to-rail performance. A resis-
tor and dc voltage source, in series with the collector, accurately
portray output dropout voltage versus output current. The
VCMH and VCML sources set the upper and lower limits of
the input common mode voltage range. Both are set up as a
function of the supply voltage to mimic the varying common
mode range with supply voltage. The EOS voltage source estab-
lishes the offset voltage and is also used to create the common-
mode rejection and power supply rejection characteristics for
the model.
The broadband input referred voltage noise for the model is
6.8 nV/√Hz. Flicker noise characteristics are also accurately
modeled with the 1/f corner frequency set through the KF and
AF terms in the input stage transistors. Finally, a voltage-con-
trolled current source, GSY, is used to model the amplifier’s
supply current versus supply voltage characteristics.
+5V SUPPLY
1/2
SSM2275
18-BIT
VBL
DAC
1
2
3
4
5
6
7
8
VL
16
15
8
LL
220F
47kΩ
3
2
LEFT
CHANNEL
OUTPUT
1
18-BIT
SERIAL
REG.
7.68k⍀
9.76k⍀
DL
CK
4
14
13
VOL
330pF
VREF
7.68k⍀
DR
LR
100pF
AD1868
AGND 12
11
18-BIT
SERIAL
REG.
VREF
7.68k⍀
VOR
10
DGND
VBR
100pF
7.68k⍀
9.76k⍀
18-BIT
DAC
6
9
220F
47k⍀
VS
RIGHT
CHANNEL
OUTPUT
1/2
SSM2275
7
330pF
5
Figure 42. A Smoothing Filter for an 18-Bit Stereo DAC
–14–
REV. A
SSM2275/SSM2475
*ADAPTIVEPOLEANDGAINSTAGE
*ATVsy=5,fp=12.50MHz,Av=1
*ATVsy=30,fp=18.75MHz,Av=1.16
Listing 1: SSM2275 SPICE Macro-Model
*SSM2275SPICEMacro-ModelTypicalValues
* 8/97, Ver. 1
* TAM / ADSC
*
*Nodeassignments
*
*
*
*
*
*
G2 98 20 POLY(2) (4,6) (99,50) 0 80.3E-6 0 0 2.79E-6
VR1 20 21 DC 0
H1 21 98 POLY(2) VR1 VSN1 0 11.317E3 0 0 -28.29E6
C2 20 98 1.2E-12
non-invertinginput
|
|
|
|
|
|
1
invertinginput
*
|
|
|
|
|
2
positivesupply
* POLE AT 90MHz
*
G39823(20,98)565.5E-6
R5 23 98 1.768E3
C3 23 98 1E-12
*
*GAINSTAGE
*
G19830(23,98)733.3E-6
R1 30 98 9.993E3
CF 30 45 200E-12
D5 31 99 DX
D6 50 32 DX
V1 31 30 0.6
V2 30 32 0.6
*
|
negativesupply
|
|
output
|
|
45
*
*
|
|
|
|
.SUBCKTSSM2275
*
*INPUTSTAGE
*
99
50
Q1
Q2
4
6
3
2
5 QNIX
7 QNIX
RC1 99 11 15E3
RC2 99 12 15E3
RE1
RE2
EOS
IOS
5
7
3
8 1E3
8 1E3
1 POLY(2) (61,98) (73,98) 1.5E-3 1.78E-5 1
2 5E-9
1
*OUTPUTSTAGE
ECMH1 4 11 POLY(1) (99,50) 0.9 -30E-3
ECMH2 6 12 POLY(1) (99,50) 0.9 -30E-3
ECML1 9 50 POLY(1) (99,50) 0.1 30E-3
ECML21050POLY(1)(99,50)0.130E-3
*
Q3 46 42 99 QPOX
Q4 47 44 50 QNOX
RO1 46 48 30
RO2 47 49 30
VO1 45 48 15E-3
VO2 49 45 10E-3
RB1 41 42 200
RB2 43 44 200
EO1 99 41 POLY(1) (98,30) 0.7528 1
EO2 43 50 POLY(1) (30,98) 0.7528 1
*
*MODELS
*
.MODELQNIXNPN(IS=1E-16,BF=400,KF=1.96E-14,AF=1)
.MODELQNOXNPN(IS=1E-16,BF=100,VAF=130)
.MODELQPOXPNP(IS=1E-16,BF=100,VAF=130)
.MODELDXD(IS=1E-16)
.MODELDZD(IS=1E-14,BV=6.6)
.ENDSSSM2275
D1
D2
D3
D4
9
5 DX
10 7 DX
13 1 DZ
2 13 DZ
IBIAS 8 50 200E-6
*
* CMRR=115 dB, ZERO AT 1kHz, POLE AT 10kHz
*
ECM1 60 98 POLY(2) (1,98) (2,98) 0 .5 .5
RCM16061159.2E3
RCM2619817.66E3
CCM1 60 61 1E-9
*
*PSRR=120dB, ZEROAT1kHz
*
RPS1 70 0 1E6
RPS2 71 0 1E6
CPS1 99 70 1E-5
CPS2 50 71 1E-5
EPSY 98 72 POLY(2) (70,0) (0,71) 0 1 1
RPS3 72 73 1.59E6
CPS3 72 73 1E-10
RPS4 73 98 1.59
*
*INTERNALVOLTAGEREFERENCE
*
RSY1 99 91 100E3
RSY2 50 90 100E3
VSN1 91 90 DC 0
EREF 98 0 (90,0) 1
GSY 99 50 POLY(1) (99,50) 0.97E-3 -7E-6
*
REV. A
–15–
SSM2275/SSM2475
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
8-Lead SOIC
14-Lead SOIC
(R-14)
(SO-8)
0.1968 (5.00)
0.1890 (4.80)
0.3444 (8.75)
0.3367 (8.55)
8
1
5
4
14
1
8
7
0.1574 (4.00)
0.1497 (3.80)
0.1574 (4.00)
0.1497 (3.80)
0.2440 (6.20)
0.2284 (5.80)
0.2440 (6.20)
0.2284 (5.80)
PIN 1
0.0688 (1.75)
0.0532 (1.35)
0.0688 (1.75)
0.0532 (1.35)
PIN 1
0.0196 (0.50)
0.0196 (0.50)
0.0099 (0.25)
x 45°
x 45°
0.0098 (0.25)
0.0040 (0.10)
0.0098 (0.25)
0.0040 (0.10)
0.0099 (0.25)
8°
0°
8°
0°
0.0500
(1.27)
BSC
0.0500
(1.27)
BSC
0.0192 (0.49)
0.0138 (0.35)
0.0192 (0.49)
0.0138 (0.35)
SEATING
PLANE
SEATING
PLANE
0.0098 (0.25)
0.0075 (0.19)
0.0500 (1.27)
0.0160 (0.41)
0.0500 (1.27)
0.0160 (0.41)
0.0099 (0.25)
0.0075 (0.19)
8-Lead Plastic DIP
(N-8)
14-Lead TSSOP
(RU-14)
0.430 (10.92)
0.348 (8.84)
0.201 (5.10)
0.193 (4.90)
8
5
0.280 (7.11)
0.240 (6.10)
14
8
7
1
4
0.325 (8.25)
0.300 (7.62)
0.060 (1.52)
0.015 (0.38)
PIN 1
0.195 (4.95)
0.115 (2.93)
0.210 (5.33)
MAX
0.130
(3.30)
1
0.160 (4.06)
0.115 (2.93)
MIN
0.015 (0.381)
SEATING
PLANE
0.100
(2.54)
BSC
0.022 (0.558)
0.014 (0.356)
0.070 (1.77)
0.045 (1.15)
0.008 (0.204)
PIN 1
0.006 (0.15)
0.002 (0.05)
0.0433
(1.10)
MAX
0.028 (0.70)
0.020 (0.50)
8°
0°
8-Lead microSOIC
(RM-8)
0.0118 (0.30)
0.0075 (0.19)
0.0256
(0.65)
BSC
SEATING
PLANE
0.0079 (0.20)
0.0035 (0.090)
0.122 (3.10)
0.114 (2.90)
5
4
8
1
0.199 (5.05)
0.187 (4.75)
0.122 (3.10)
0.114 (2.90)
PIN 1
0.0256 (0.65) BSC
0.120 (3.05)
0.112 (2.84)
0.120 (3.05)
0.112 (2.84)
0.043 (1.09)
0.037 (0.94)
0.006 (0.15)
0.002 (0.05)
33°
27°
0.018 (0.46)
0.011 (0.28)
0.003 (0.08)
0.028 (0.71)
0.016 (0.41)
SEATING
PLANE
0.008 (0.20)
–16–
REV. A
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