B82791G2301N001 [DIODES]
High Power Factor LED Replacement T8 Fluorescent Tube; 高功率因数LED替代T8荧光灯管
| 型号： | B82791G2301N001 |
| 厂家： | 
DIODES INCORPORATED |
| 描述： | High Power Factor LED Replacement T8 Fluorescent Tube |
| 文件： | 总12页 (文件大小：552K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
AN75
AN75
High Power Factor LED Replacement T8 Fluorescent Tube
using the AL9910 High Voltage LED Controller
Yong Ang, Diodes Inc.
Introduction
This application note describes the principles and design equations required for the design of a high
brightness LED lamp using the AL9910. The equations are then used to demonstrate the design of a
universal, offline, high power factor (PF), 13W LED lamp suitable for use as the replacement for T8
fluorescent tube. A complete design including the electrical diagram, component list and performance
measurements are provided.
AL9910 high power factor buck LED driver
Figure 1 Electrical schematic of a high power factor 13W LED lamp
Figure 1 shows the electrical diagram of an offline 13W LED driver.
On the input side, CX1, CX2, CX3, CX4, L1 and L2 provide sufficient filtering for both differential mode
and common mode EMI noise which are generated by the switching converter circuit.
The rectified AC line voltage from the bridge rectifier DB1 is then fed into a passive power factor
correction or valley fill circuit which consists of 3 diodes and 2 capacitors. D1, D2, D3, C1, C2 improve
the input line current distortion in order to achieve PF greater than 0.9 for the AC line input.
The constant current regulator section consists of a buck converter driven by the AL9910. Normally,
the buck regulator is used in fixed frequency mode but its duty cycle limitation of 50% is not practical
for offline lamp. This problem can be overcome by changing the control method to a fixed off-time
operation.
The design of the internal oscillator in the AL9910 allows the IC to be configured for either fixed
frequency or fixed off-time based on how resistor RT is connected. For fixed off-time operation, the
resistor RT is connected between the Gate and ROSC pins, as shown in Figure 1. This converter has
now a constant off-time when the power MOSFET is turned off. The on-time is based on the current
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sense signal and the switching adjusts to be the sum of the on- and off-time. This change allows the
converter to work with duty cycles greater than 50%.
Design Guide – High power factor offline LED driver
In this section the design procedure is outlined according to the schematic shown in Figure 1. First,
the guideline for selecting the components for valley fill power factor correction stage and fixed off-
time buck converter is shown. The power inductor calculation is then demonstrated and finally, the
power losses within MOSFET and free-wheel diode are assessed.
The specifications for the system are:
V
V
AC = 230Vac
AC(min) = 85Vac
VAC(max)= 264Vac
LED(nom) = 240mA
I
VLED(nom) = 54V
VLED(min) = 42V
VLED(max) = 59V
POUT = 12.96W
fswi(nom) = 55kHz
Passive factor correction stage design
The purpose of the valley fill circuit (see Figure 2) is to allow the buck converter to pull power directly
off the AC line when the line voltage is greater than 50% of its peak voltage.
Figure 2 Valley-fill PFC stage and operating waveforms (Green: VIN to LED driver; Orange:
AL9910’s gate voltage)
The maximum bus voltage at the input of the buck converter is,
V
= 2 × Vac(max) = 2 × 264Vac = 373V
IN(max)
During this time, capacitors within the valley fill circuit (C1 and C2) are in series and charged via D2
and R1. If the capacitors have identical capacitance value, the peak voltage across C1 and C2
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is V
2 = 186V . Often a 20% difference in capacitance could be observed between like
IN(max)
capacitors. Therefore a voltage rating margin of 25% should be considered.
Once the line drops below 50% of its peak voltage, the two capacitors are essentially placed in
parallel. The bus voltage VIN(min) is the lowest voltage value at the input of the buck converter. VIN(min)
at the minimum AC line voltage Vac(min) is,
V
= 2 × Vac(min) 2 = 2 ×85Vac 2 = 60V
IN(min)
At 60Hz, the total time of a half AC line cycle is 8.33ms. The power to the buck converter is derived
from the valley-fill capacitors when the AC line voltage is equal to or less than 50% of its peak voltage.
The hold up time for the capacitors equates to tHOLD = 1 3× 8.33ms = 2.77ms . The valley-fill capacitor
value can then be calculated,
Pout
× tHOLD
12.96W
× 2.77ms
V
IN(min)
60V
20V
CTOTAL
=
=
= 30μF
VDROOP
Therefore, C1= C2 = 15μF . VDROOP is the voltage droop on the capacitors when they are delivering full
power to the buck converter. Ideally VDROOP should be set to less than VDROOP = V − VLED(max) in
IN(min)
order to ensure continuous LED conduction at low line voltage. Nevertheless, VDROOP is set to be 20V
in the design example to avoid the need for very large valley-fill electrolytic capacitor.
A 20V VDROOP implies that the bus voltage VIN at the input of buck converter will drop to 40V during
part of the AC line cycle. As the buck regulator requires VIN to be greater than the LED stack voltage
(VLED(max)=59V) for regulation, the LED will be off during part of the AC line cycle. This has the effect of
reducing the actual output LED current at low AC input voltage. In the design example, the LED
current drops by approximately 20% from its nominal value at 85Vac (see Figure 4).
Setting the fixed off-time and switching frequency range
For fixed off-time operation, the switching frequency will vary subjected to the actual input voltage and
output LED conditions.
A nominal switching frequency fswi(nom) should be chosen. A high nominal switching frequency will
result in smaller inductor size, but could lead to increased switching losses in the circuit. A good
design practice is to choose a nominal switching frequency knowing that the switching frequency will
decrease as the line voltage drops and increases as the line voltage increases.
The fixed off-time tOFF can be computed as,
VLED(nom)
54V
1-
1-
Vac(nom)
230V
55kHz
toff
=
=
= 13.9μs
fswi(nom)
The off-time is programmed by timing resistor RT as shown in Figure 1. The value of RT is given by,
RT kΩ = tOFF μs × 25 − 22 = 13.9× 25 − 22 = 326kΩ
(
)
(
)
A 330kΩ is selected for RT. Next, the two extremes of the variable switching frequency can be
approximated as,
1− VLED(max)
V
IN(min)
1− 59V 69V
13.9μs
fswi(min)
=
=
= 10kHz
tOFF
1− VLED(min)
V
IN(max)
1− 42V 373V
13.9μs
fswi(max)
=
=
= 63.8kHz
tOFF
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It is advisable to keep below the maximum switching frequency fswi(max) below 150kHz to avoid
excessive switching loss.
Inductor selection and setting the LED current
The fixed off-time architecture of the AL9910 regulates the average current through the inductor LBUCK
.
The value of LBUCK depends on the desirable peak-to-peak ripple ΔIL in the output LED current. LBUCK
can be set with the following equation,
VLED(nom) × tOFF
54V ×13.9μs
LBUCK
=
=
= 6.6mH
ΔIL
115mA
Due to diameter limitation of the T8 tube, LBUCK is made up of L3 and L4 as shown in Figure 1.
The AL9910 constant off-time control loop regulates the peak inductor current Ipk. As the average
inductor current equals the average LED current, the average LED current can be regulated by
controlling Ipk.
Given a fixed inductor value, the change in the inductor current over time is proportional to the voltage
applied across the inductor. During the off-time, the voltage seen by the inductor is the LED stack
voltage. So, the peak inductor current should be regulated to,
0.5× VLED(nom) × tOFF
0.5× 54V ×13.9μs
Ipk = ILED(nom)
+
= 240mA +
= 297mA
LBUCK
6.6mH
The peak current is constant and set by the sense resistor RSENSE. If the LD pin is tied to the VDD pin,
the value of RSENSE can be easily calculated because the voltage threshold on the CS pin is 0.25V,
0.25
RSENSE
=
= 0.84Ω
297mA
In the circuit shown in Figure 1, RSENSE consists of R5, R6 and R7.
The peak current rating of the LBUCK should be greater than Ipk and the RMS current rating of the
inductor should be at least 110% of ILED(nom)
.
Although the described solution, working in fixed off-time and Continuous Conduction Mode (CCM),
works as a constant current source, a limitation to the output LED current accuracy is its dependency
on the number of LEDs and overall LED chain voltage. The best result can be achieved using a fixed
number of LEDs. A variable number of LEDs results in reduced current precision.
The two extremes of the output LED current can be approximated as,
0.5× VLED(max) × tOFF
0.5× 59V ×13.9μs
ILED(min) = Ipk
-
= 297mA -
= 297mA -
= 234mA
= 253mA
LBUCK
6.6mH
0.5× VLED(min) × tOFF
0.5× 42V ×13.9μs
ILED(max) = Ipk
-
LBUCK
6.6mH
The above equation shows that the precision of the LED current also depends on the tolerance of
practical inductor LBUCK. Inductor with tolerance rating equal or less than 10% should be chosen to
ensure good LED current precision at mass production.
Power MOSFET calculation
The power MOSFET is chosen based on maximum voltage stress, peak MOSFET current, total power
losses, maximum allowable working temperature and the gate driver capability of the AL9910.
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Maximum drain-source voltage stress on the power MOSFET for this converter is equal to the input
voltage. However, a typical voltage safety margin for the MOSFET defines the maximum reverse
voltage as follows,
VDSS = 1.3× V
= 1.3×373V = 485V
IN(max)
which implies that a common 500V MOSFET is suitable.
The power MOSFET losses will be dominated by switching loss. The switching loss depends on the
switching time, frequency, MOSFET drain current and drain-source voltage. The switching rise time
tRISE and fall time tFALL is a function of the MOSFET’s gate capacitance, the gate driver capability of the
AL9910 and layout design. The worse case switching power losses occurs at VLED(min) and VIN(max)
The switching loss is approximately,
.
VLED(min) OFF
t
⎛
⎞
⎜
⎟
⎟
V
IN(max) × Ipk
−
× tRISE × fswi(max)
⎜
V
IN(max) ×Ipk × tFALL × fswi(max)
LBUCK
2
297mA − 88mA
⎝
⎠
PSW
=
=
+
2
373V ×
(
)
× 65ns× 63.8kHz 373V × 65ns× 63.8kHz
+
2
2
= 455mW
where the switching time tRISE and tFALL are measured to be 65ns with the 600V MOSFET
SPB03N60S5 as the power MOSFET. As shown in Figure 1, R10 is a series gate resistor that slows
down the MOSFET switching and reduces EMI emission.
The RMS current through the MOSFET at VLED(min) and VIN(max) is given by,
VLED(min)
VLED(min) × tOFF LBUCK
⎛
⎜
⎝
⎞
⎟
⎟
⎠
⎜
ID(RMS)
=
=
× ILED(nom)
+
V
12
IN(max)
⎛
⎞
42V
42V ×13.9μs 6.6mH
⎜
⎟
⎟
× 240mA +
⎜
373V
12
⎝
⎠
= 89mA
The power MOSFET conduction loss depends on its static drain-source resistance RDS(ON) at the
MOSFET working temperature. It is possible to calculate the continuous conduction loss:
PCOND = ID2(RMS) ×RDS(ON)
The total power MOSFET loss is:
=
(
89mA
2 × 2.5Ω = 19mW
)
PTOT = PSW + PCOND = 455mW +19mW = 474mW
Total MOSFET power loss is dissipated from the SMD package into the PC Board. So it is possible to
calculate the MOSFET working junction temperature can be calculated if the package junction-to-
ambient thermal resistance RthJA is known. The calculated MOSFET junction temperature, TJ, must be
lower then the maximum allowable junction temperature TJ(MAX)
:
TJ = PTOT × θthJA + TAMB = 474mW × 62 oC W + 80oC = 109.4o C
The internal ambient temperature within the LED converter, TAMB, is assumed to be 80ºC. θthJA
=
62 oC W is the thermal resistance for TO-263 with minimum copper area. For practical design, it is
recommended to keep the junction temperature below 110ºC to avoid temperature stress on the
device.
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Free-wheel diode calculation
The free-wheel diode DF shown in Figure 1 is chosen based on its maximum stress voltage and total
power loss. The maximum stress voltage rating of the free-wheel diode is the same as the MOSFET. It
is advisable to use ultra-low reverse recovery time TRR (<35ns) diode as DF to reduce the MOSFET’s
switching ON loss. In the design example, 1A 600V rectifier, MUR160, is selected.
The worst case average current through the diode occurs at VLED(max) and VIN(min)
.
⎛
⎜
⎝
⎞
⎟
VLED(min)
42V
⎛
⎞
⎟
⎜
ID(avg) = ILED(nom) × 1−
= 240mA × 1−
= 202mA
⎜
⎟
⎠
V
373V
⎝
⎠
IN(max)
Assuming a constant forward voltage drop VF across the diode, the conduction power loss can be
calculated,
PD _ COND = ID(avg) × VF = 202mA ×1.1V = 222mW
Finally, the diode junction temperature without using the heat sink can be calculated from,
Tj = PD_COND × θthJA + TAMB = 222mW × 32 oC W + 80o C = 87o C
The internal ambient temperature within the LED converter, TAMB, is assumed to be 80ºC. θthJA
=
32 o C W is the thermal resistance for DO-201 package. For practical design, it is recommended to
keep the junction temperature below 110ºC to avoid temperature stress on the device.
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The BOM in table 1 and the PCB layout in Figure 2 complete the tools needed to design a high power
factor LED driver using the AL9910. Figure 3 shows the picture of the completed LED driver designed
with a footprint to fit inside the T8 LED Fluorescent replacement lamp tube.
Table 1 BOM
Ref.
Descriptions
Part number
Package
Mfr.
U1
Universal high brightness
LED driver
AL9910
SO8
Diodes Inc.
D1, D2, D3 1A, 1kV diode tRR = 1.8μs
S1M-13-F
MUR160
SMA
DO201AD
Diodes Inc.
Diodes Inc.
D4
Ultra-fast-recovery diode
1A, 600V, tRR = 35ns
DB1
C1, C2
1A, 600V bridge rectifier
15μF, 450V electrolytic
capacitor +/-20% 1000hrs 400KXW27M10X30
DF06S
EEUED2W150
DF-S
5mm pitch
Diodes Inc.
Panasonic
Rubicon
@ 105ºC
UCY2G150MPD
ECE-A1HKG4R7
Nichicon
Panasonic
C4
C5
4.7μF, 50V electrolytic
capacitor +/-20% 1000hrs
@ 105ºC
10μF 450V electrolytic
capacitor +/-20% 1000hrs
@ 105ºC, 10mm diameter
1.5mm pitch
5mm pitch
EEUEE2W100U
Panasonic
CX1, CX2,
CX3, CX4
F1
100nF, 275VAC, Film, X2 ECQU2A104ML
17.5mm pitch Panasonic
10Ohm 1W fusible
resistor +/-200ppm
6.8mH inductor +/-10%
290mA radial
NFR0100001009JR500
Through-hole Vishay
axial
L1
19R685C
5mm pitch
10mm pitch
6mm pitch
Murata
EPCOS
Murata
L2
30mH common-mode
inductor, 8mm height
3.3mH inductor +/-10%
420mA radial
B82791G2301N001
19R335C
L3, L4
MOV1
Q1
275V, 21J, 9mm, Radial
N-ch MOSFET 600V,
3.2A, Qg(max) = 16nC
10R 3W wire wound
resistor, 50ppm/ºC, +/-1%
3k 0.25W resistor +/-5%
1R2 0.25W +/-1%
2R7 0.25W +/-1%
100R 0.25W +/-1%
330k 0.125W resistor +/-
1%
B72207S0271K101
SPB03N60S5
5mm pitch
TO263
EPCOS
Infineon
R1
UB3C-10RF1
Through-hole Riedon
axial
R2
R5
R6
R7
RT
Any
Any
Any
Any
Any
1206
1206
1206
1206
1206
Any
Any
Any
Any
Any
R10
10R 0.25W +/-5%
Any
1206
Any
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Figure 2 Top layer and bottom layer layout
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Figure 3 Picture of the LED T8 Fluorescent replacement lamp driver
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Results
The performance of the system is outlined in Figures 4, and 5.
They display a level of system efficiency higher than 87% when driving 18 LEDs. The system
efficiency reduces with decreasing number of LEDs but 83% can still be achieved when driving
14LEDs at 264Vac input.
When driving 18 LEDs, a current regulation of around 3% is achieved between the input voltages of
110Vac to 264Vac. The LED current drops to 190mA at 85Vac as the minimum bus voltage VIN(min)
falls below the LED stack voltage (VLED(max)) during part of the AC line cycle, driving the LED off.
Figure 6 shows the power factor across the line voltage range. Power factor greater than 0.9 can be
achieved at 85Vac.
Figure 4 LED driver system efficiency
Figure 5 LED driver current regulation
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Figure 6 LED driver power factor
Conclusion
This application note provides a simple tool to design an offline LED driver using the AL9910 high
voltage LED controller. It provides a high level of efficiency as well as LED current control over a wide
range of input voltages. Moreover the document explains how to design a system with passive power
factor correction to achieve PF greater than 0.7, allowing compliant with emergent international solid
state lighting standards.
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