AB-179 [ETC]

AB-179 - VIDEO OPERATIONAL AMPLIFIER ; AB - 179 - 视频运算放大器\n


型号：	AB-179
厂家：	ETC
描述：	AB-179 - VIDEO OPERATIONAL AMPLIFIER AB - 179 - 视频运算放大器\n 运算放大器
文件：	总8页 (文件大小：125K)
中文：	中文翻译
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VIDEO OPERATIONAL AMPLIFIER

By Klaus Lehmann, Burr-Brown International GmbH

CURRENT OR VOLTAGE FEEDBACK?

THAT’S THE QUESTION HERE.

In analog technology, important advances are being made in

video operational amplifiers, both in processing and in

circuitry. The complementary bipolar processes used by

“Bell Laboratories” should be emphasized, which have ver-

tically structured and approximately equal electric NPN and

PNP transistors on one substrate. These processes help to

create the basis for an effective complementary-symmetric

circuit technique, which is currently obtaining its best results

with the so-called “Diamond” structure. This structure plays

a key role in transimpedance or current-feedback amplifiers.

The following article is concerned exclusively with explain-

ing these key topics and the new circuit designs.

+V_IN

V_OUT

R₂

R₂₃

FIGURE 2. OPA as Series Connection between TA and B.

V_CC

The following discussion will deal first with practical cir-

cuits and then with theoretical models. It is based on publi-

cations [5] and [6], and essays [15] and [16], written at the

Institute for Semiconductor Technology of the Darmstadt

Technical College, with the assistance of Diplom Engineer

R.B. Steck.

+V_IN

V_OUT

CONVENTIONAL CIRCUIT TECHNIQUES

Figure 1 illustrates the conventional structure of a feedback

amplifier, including the wide-band operational amplifier

OPA discussed here. As can be seen in Figure 2, this kind of

OPA consists of a differential amplifier TA with high-

impedance output (7) and an impedance converter B inserted

afterwards. The elements R and C function between them,

determining, among other things, the open-loop gain G_OL,

slew rate, and bandwidth f_–3dB. The circuit techniques of B

are not discussed here, but it could have a structure such as

the push-pull one shown in Figure 3.

V_EE

FIGURE 3. Buffer with Push-pull Structure.

+SR_MAX= I/C

V_CC

–SR_MAX= I/C

Figure 4 illustrates what is probably the simplest and most

well-known structure of the differential amplifier TA. It can

be seen in Figure 4 that the capacitor C can be charged with

R₂₃

V_IN

+V_IN

V_OUT

OPA

V_OUT

R₂

R₂₃

V_EE

FIGURE 4. Conventional Structure of a Differential

Amplifier.

FIGURE 1. Voltage-Feedback Operational Amplifier.

AB-179

Printed in U.S.A. May, 1993

bias current I for rising and falling signals. The following

equations result from the positive slew rate SR + max. or

negative rate SR – max:

current-feedback amplifier. Differing input impedances can

result in adverse performance under certain conditions, but

these are not discussed here. The structure’s slew rate

performance shown in Figure 7 is worth noting. In theory,

the transfer current of C is not limited for a rising signal. In

practice, however, a current limitation appears at 10-20-fold

bias current, dependent upon the dimensioning in the current

mirror D1/Tr.4.

∆V

∆t

SR + max =

SR – max =

_C[_s]

∆V

∆t

_C[_s]

For sinusiodal signals, the largest signal change occurs at the

zero point. The following equation results:

V_CC

+SR_MAX= I/C

–SR_MAX= I/C

SRmax

ˆ ˆ

[Hz]; VpO = V

f_–3dB

) )

( (_VpO

2π

√

Because SR + max. ≠ SR – max, a asymmetric limited

frequency response results during large signal modulation.

The cascaded circuit varitations also have current-limited

modulation behavior as shown in Figure 5.

R₂₃

+V_IN

V_OUT

R₂

V_CC

V_EE

+SR_MAX= I/C

–SR_MAX= I/C

FIGURE 6. Differential Amplifier with Signal Current

Mirror.

V_CONST.

+SR_MAX≈ 10I/C

V_CC

–SR_MAX= I/C

R₂₃

+V_IN

V_OUT

R₂

R₂₃

+V_IN

V_EE

V_OUT

R₂

FIGURE 5. Cascaded Differential Amplifier.

V_EE

CURRENT FEEDBACK CONCEPT

FIGURE 7. The Current-Feedback Concept.

The introduction of the signal current mirror diode D1/

Transistor Tr.4, as shown in Figure 6, does not yet result in

improvement of the SR behavior, but will prove to be useful

later on. V_BEis compensated by Tr.1 according to the

differential amplifier principle with transistor Tr.2. The V_BE

compensation of Tr.1 is illustrated in Figure 7, along with

the previously inserted complementary emitter follower Tr.5.

The feedback loop can now be directly connected to the

emitter of Tr.1. The adaptors 1 and 2 are still the inputs of

the differential amplifier. Its impedance has now been

changed.

DIAMOND STRUCTURE

Asymmetric SR performance is caused by the asymmetric

circuit structure. A complementary-symmetric structure, as

shown in Figure 8, attains symmetric SR values at least 10

times higher than those of conventional structures with the

same quiescent current. The analogies to CMOS technology

are worth noting.

Parameters such as frequency response, modulation capabil-

ity, and distortion can be attained with drastically reduced

quiescent current. This advantage plays an important role in

portable devices, simplifies power supply equipment, and

reduces inhouse heating, especially with respect to the de-

velopment of increasingly small devices. The circuit shown

in Figure 8 without B, R₂, and R₂₃, is called a Diamond

1 = high impedance

2 = low impedance

The conventional voltage feedback has now become current

feedback. This structure is designated a transimpedance or

circuit, and is typical for a current-feedback amplifier. Pro-

cesses such as correct transmission of coded color television

signals require low phase delays dependent upon the signal

modulation. Such phase delays can arise in circuits such as

the one shown in Figure 7 through the voltage-dependent

collector substrate capacity of the Tr.4.

The circuit in Figure 10, for example, delivers the following

as voltage feedback:

G_C100kHz= –32dB; G_OL100kHz= +52dB

As current feedback, it delivers:

G_C100kHz= +5dB; G_OL100kHz= +58dB

In the circuit shown in Figure 8, the varicap of the Tr.4 is

largely compensated through the varicap of the complemen-

tary Tr.8. The advantages of the current-feedback concepts,

however, are counterbalanced by several disadvantages.

Another problem is the size of the input voltage offset.

±1mV is typical in circuits with voltage-feedback differen-

tial amplifiers (Figure 6). Current-feedback circuits like the

ones shown in Figures 7 and 8 are typical at ±50mV.

Currently, there is no better serial “matching” between NPN

and PNP transistors. Various suggestions have attained small

offset voltages with circuit variations, but these resulted in

poorer transmission characteristics. Differing early voltages

in NPN and PNP transistors cause different mirroring in D1/

Tr.4 and D2/Tr.8 and thus bring about an output bias current

at point 7. This effect is a general weakness of the Diamond

structure.

Generally stated, applications do exist in which two high-

impedance differential inputs are necessary. Poorer com-

mon-mode gain is a direct result of the unequal inputs.

+SR_MAX≈ 10I/C

V_CC

–SR_MAX≈ 10I/C

VOLTAGE FEEDBACK

WITH THE DIAMOND STRUCTURE

In order to make the best of the advantages of both ideas, it

now makes sense to combine the Diamond structure and

voltage feedback together in one structure. The current

feedback shown in Figure 8 can be transferred to voltage

feedback illustrated in Figure 9 by inserting the buffer B2.

The desired combination would be attainable with a buffer

which was “ideal” in respect to its amplitude and phase

behavior. The amplitude behavior of the buffer shown in

Figure 3 (with real current source I) reaches this condition

with f_–3dB≈ 3.0 GHz (I = 1.9 mA). The phase delay continues

to be damaged inside the control loop through the lengthen-

ing of the signal delay time by approximately T_DB2≈ 25ps.

The principle illustrated in Figure 9 is shown in more detail

in Figure 10. The contrasting results of PSPICE simulations

R₂₃

V_OUT

+V_IN

R₂

V_EE

FIGURE 8. Current-Feedback Amplifier with Diamond

Structure.

V_CC

+SR_MAX≈ 10I/C

–SR_MAX≈ 10I/C

R₈₉

R₂₃

+V_IN

V_OUT

R₂

V_EE

FIGURE 9. Voltage-Feedback Amplifier with Diamond Structure.

conducted with current and voltage feedback are summa-

rized in Figure 11. All simulations were carried out with

fully equipped current sources. To give the user an over-

view, the f_–3dBfrequencies are illustrated in Figure 12 depen-

dent upon the closed-loop gain for both feedback types. The

relatively low frequency response differences shown here do

not concur with the assertions in many publications about

current-feedback amplifiers. This discrepancy can be ex-

plained by the fact that these articles place an unrealistic

emphasis on the ability of the current-feedback concept—

they maintain that this concept alone brought about the

improved bandwidth. Improved technology and the Dia-

mond structure play a more decisive role than the current-

feedback concept.

The first section of this article deals with the basics of

current and voltage-feedback video operational amplifiers

and their corresponding circuits. These practically oriented

considerations are then followed by extensive theoretical

analysis. The point of departure for the following models is

the performance of Diamond structure with open loop gain

G_OL⁺as shown in Figures 8 and 10. The measurement circuit

shown in Figure 13 is used for analysis. The results are

presented in Figure 14. The curves follow a simple model up

to f ≈ 300 MHz, formed by the parameters gm, R2, R, and

C. Simple modelling of the amplitude response with R and

C is not enough to describe the phase delay. Through a row

of additional, partially very small RC parts inside the control

loop, more phase delays result than would occur alone with

Buffer B₂

V_CC

R₈₉

+V_IN

V_OUT

V_EE

R₂

R₂₃

FIGURE 10. Developed Voltage-Feedback Amplifier.

f_–3dB

(GHz)

1.0

0.8

0.6

0.4

0.2

–C_GL

(dB)

+C_GL

(dB)

–20

–40

24 18 12

12 18 24

FIGURE 12. Bandwidth with Current and Voltage Feedback.

100

300M

1.0G

3.0G

10G

Frequency (Hz)

FIGURE 11. Closed Gain Progress with Current and Volt-

age Feedback.

R and C. An additional phase delay with variation of the

amplitude response is attained through insertion of the delay

time T_D. The following equation applies to the delay time of

a RC part:

to form the new current source gm’ (Figure 16). The volt-

age-feedback shown in Figure 16 varies from the current-

feedback model (Figure 17) only in the use (or non-use) of

the buffer in the feedback loop.

T = [arc tg (ωRC)]/ω.

For a better overview, the output conditions, various equa-

tions, and optimal operating conditions are summarized in

Table I. The derivations are described in detail [5].

As already mentioned, if ωRC remains sufficiently small, it

will be T ≈ RC. All of these small time constants are

summarized in the following model to the total delay time

T_D. To be able to imagine this concretely, it is important to

remember that small time constants always turn the phase,

but do not necessarily damage the amplitude response con-

sidered here.

T_D

–1

V_OUT

R₂₃

V_IN

+V_IN

R₈₉

R₂

V_OUT

C₂

R₂₃

R₂

FIGURE 15. Model of a Differential Amplifier Type.

10kΩ

gm = 2I/V_T

gm’ = 1/(R₂– 1/gm)

G⁺_OL= V_OUT/V⁺_IN= gm’ + R

FIGURE 13. Circuitry for Adjustment of G⁺

T_D

–1

V_OUT

R₂₃

0.01

+V_IN

g'm

100

300

R₂

R₈₉

–20

–40

FIGURE 16. Model of the Voltage Feedback.

100

10M

100M

1.0G

10G

Frequency (Hz)

T_D

FIGURE 14. Open-Loop Frequency Response.

–1

V_OUT

R₂₃

DELAY TIME MODEL

+V_IN

gm'

The circuits with voltage feedback as shown in Figures 4, 5,

6, and 9 are described with the model shown in Figure 15.

The signal inversion of the current mirror in Figures 6 and

9 is viewed in the output buffer with G = –1. The delay time

T_Dis inserted here. The dependence of the transconductance

gm on the quiescent current I varies according to circuit

structure. In the case shown in Figure 4, for example, gm =

I/2V_Τ, while in the more interesting Figures 8 and 9, gm =

2I/V_Τ. In the next formal step, the two controlled current

sources gm and the resistor R₈₉(see Figure 15) are combined

R₂

FIGURE 17. Model of the Current Feedback.

OPTIMAL FREQUENCY

DIFFERENCES BETWEEN

RESPONSE ADJUSTMENT

CURRENT AND VOLTAGE FEEDBACK

The equation for calculation of |A_N| and the graphic values

in Figure 18 describe the various operating conditions of the

four amplifier variations. Normally, the optimal adjustment

is installed. This has the following conditions:

G_CLMAXis reached for R₈₉→ 0 or R₂|| R₂₃→ 0, which is

practically unattainable with current feedback. Current feed-

back usually obtains lower values for G_CLMAXwith otherwise

comparable parameters. G_OL0is dependent upon G_CLO. Am-

plifiers with adjustable G_CLO,such as those for gain control,

simultaneously require correction of G_OLO. Figure 19 shows

the frequency response differences between current and

voltage feedback according to the delay time model. In the

case of voltage feedback, a change in R₂means a change in

C/gm = 2 k_OT_D

Internally compensated amplifiers use an integrated com-

pensation capacitor C corresponding to the already men-

tioned condition for the poorest case i.e. the smallest closed

loop gain, for example k_O= 1. This method is not well-suited

to the wide-band amplifiers discussed here. The slew rate

and large-signal frequency response are significantly re-

duced through the compensation capacitor. From this adjust-

ment, deviating, i.e. larger, closed-loop gain leads to re-

duced frequency response. Amplifiers adjustable externally

with C allow optimal adjustment with k_O, but do not signifi-

cantly improve large-signal performance. Optimal adjust-

ment with gm avoids these disadvantages. One possible

compromise is to insert amplifiers compensated internally

₈₉as well. With current feedback, the gains G_CLOand G_OLO

are simultaneously adjusted to one another during changes

in R₂, which can be an important advantage. Feedback

buffers (G_CL= +1) have no inverting input. While the

disadvantage ^Oof the low impedance inverting input of the

current-feedback concept is not significant, here the low

circuit expenditure and quiescent current are important ad-

vantages.

over gm for various closed-loop gain ranges (e.g. G_CL

1-3, 3-10, 10-30). Making the open-loop gain G_OLexternally

adjustable over gm for optimal frequency response is the

most effective solution. This can be adjusted with R₈₉using

voltage feedback and with R₂/R₂₃₆using current feedback.

R₂||R₂₃

R₈₉

0.01

5.9

f_–3dB

–12

–4

11.2

25.6

21.5

–2

–1

47.13

45.1

–10

–20

–30

780MHz

640MHz

10M

30M

100M

300M

1.0G

3.0G

10G

R₈₉

|A_N|opt.

12dB/Okt.

Frequency (Hz)

–10

–20

–30

with internal

C-compensation

5dB/Okt.

FIGURE 19. Model Frequency Response with Current and

Voltage Feedback.

163

470

10M

30M

100M

300M

1.0G

3.0G

10G

Frequency (Hz)

FIGURE 18. Adjustment of the Frequency Response.

DIMENSIONING

Inside the optimal adjustment range, the four amplifier

variations have a bandwidth f_–3dB, which is dependent only

upon the additional delay time T_D. The bandwidth corre-

sponds to the model and is independent of the closed-loop

gain G_CL(Figure 19). The deviations of the model frequency

response (Figure 19) from the simulated frequency response

(Figure 11) result from simplified modelling (see Figure 14).

This performance is not limited to the current feedback and

has become practically visible through technological ad-

vances. The first priority goes to achieving the shortest

possible delay time T_Dinside the control loop. The second

priority goes to small capacity C due to high slew rate and

wide large-signal frequency response, as well as higher, if

possible closed-loop gain G_CLMAX

FEEDBACK MODE

OPERATION MODE

MODEL CIRCUITRY

VOLTAGE FEEDBACK

NON-INVERTING INVERTING

CURRENT FEEDBACK

NON-INVERTING INVERTING

T_D

V_OUT

–1

+V_IN

R₂₃

gm'

R₂

–V_IN

G_C⁺_LO= 1 / k₀⁺= 1+ R₂₃/ R₂

G_C⁺_L= V_OUT/ V_I⁺_N

G_C⁺_LO= 1 / k₀⁺= 1+ R₂₃/ R₂

G_C⁺_L= V_OUT/ V_I⁺_N

G_C^–_LO= 1 / k₀^–= 1+ R₂₃/ R₂

G_C^–_L= V_OUT/ V_I^–_N

G_C^–_LO= 1 / k₀^–= 1+ R₂₃/ R₂

G_C^–_L= V_OUT/ V_I^–_N

CLOSED-LOOP GAIN DC

CLOSED-LOOP GAIN AC

G_O^–_L0= g′ R; G_OL= g′ / ωC | f >1MHz

OPEN-LOOP GAIN DC & AC

TRANSCONDUCTANCE

g_m= 2 I / V_T; T_C= C / g′

ABBREVIATIONS

OPERATING

TRANSCONDUCTANCE

g′ = 1 / (R₂/ /R₂₃+1 / g_m

)

g′ = 1 / (R₈₉+ 2 / g′ )

V_IN= V_I⁺_N; G_CL= G_C⁺_L; G_CLO= G_C⁺_LO= G_C^–_LO+1; k₀= k₀⁺= k₀^–/ (k₀^–+1)

AGREEMENTS

(V_IN– V₂) / V₇= jωC / g′ ; V₂= k₀V_OUT; V_OUT= V₇exp(– jωT_D

)

OUTPUT EQUATIONS

G_CL= [k₀– ωT_Csin(ωT_D) + jωT_Ccos(ωT_D)]^–1/2

CLOSED-LOOP

GAIN COMPLEX

CLOSED-LOOP

GAIN (Amount)

G_CL= [k²₀+ (ωT_C

)

– 2k₀ωT_Csin(ωT_D)]^–1/2

sin(ωT_D) ≈ ωT_D– (ωT_D

G_CL= [k²₀+ T_C(T_C– 2k₀T_D)ω²+ k₀T_CT³_Dω⁴/ 3]^–1/2

)

³/ 6

APPROXIMATION

AMOUNT OF NORMALIZED

CLOSED-LOOP GAIN

–1/2

T_CT³_D

3k₀

G_CL

T_C

k²₀

A_N

(T_C– 2k₀T_D)ω²

ω⁴

G_CLO

CONDITIONS FOR OPTIMIZED

FREQUENCY RESPONSE

T_C– 2k₀T_D= 0; T_D

g′

4πf G_OL

AMOUNT OF OPTIMIZED

CLOSED-LOOP GAIN

G_CL

= [k²₀+ k₀T_CT³_Dω⁴/ 3]^–1/2

OPT

–1/2

AMOUNT OF NORMALIZED

OPTIMIZED CLOSED-LOOP

GAIN

G_CL

C / g′

T³_Dω⁴

1+ T_D⁴ω⁴

OPT

A_N

OPT

G_CLO

3k₀

AMPLIFIER

CHARACTERISTIC VALUES

I; R; C; T_D

R₈₉

BLOCK DIAGRAM

+V_IN

V_OUT

VFA

CFA

R₂₃

CFA

R₂

R₂₃

R₂

R₂₃

R₂

–V_IN

R₂₃

–V_IN

OPEN-LOOP GAIN DC FOR

OPTIMIZED FREQUENCY

RESPONSE

G_OL0= G_C⁺_L0

G_OL0= (G_C^–_L0+1)

G_OL0= G_C⁺_L0

G_OL0= (G_C^–_L0+1)

2T_D

V_T

2T_D

V_T

– (G_C^–_LO+1)

2T_D

V_T

2T_D

V_T

ADJUSTMENT FOR OPTIMAL

FREQUENCY RESPONSE

R₂₃

≈

– G_C⁺_LO

R₂₃

≈

R₈₉

≈

–

R₈₉

≈

–

(G_C^–_LO+1)C

G_C⁺_LO

2T_D

G_C⁺_LO

2T_D

(G_C^–_LO+1)C

2T_D

METHOD OF APPROXIMATION

FOR G_CLO➞ 1

2I T_D

4I T_D

G_C^–_{LO max}

= –1

MAX ATTAINABLE OPTIMIZED

CLOSED-LOOP GAIN

G_C^–_{LO max}

–1

G_C⁺_{LO max}

V_T

V_TC

V_T

V_TC

BANDWIDTH OF THE

OPTIMIZED RESPONSE

f_–3dB= 0.176/T_D

TABLE I. Summary of the Basis and Result of the Delay Time Model.

LITERATURE

[9] Nelson, D.;

US-Patent, No. 4 502 020; Feb. 26, ’85

[1] Friedrich, H.;

Elektronik 87,vol. 26

Das Zauberwort heißt Transimpedanz-Verstärker

(The Magic Word is Transimpedance Amplifiers)

[10] Nelson, D.;

US-Patent, No. 4 628 279, Dec. 9, ’86

[11] Nelson, D.;

US-Patent, No. 4 713 628; Dec. 15, ’87

[2] Goodenough, F.;

Electronic Design 87, April 16, pp. 59…

[12] Palmer, W.;

Electronics 88, Jan. 7,pp. 151…

Transimpedance amps: fast yet accurate

[13] Saller, K.;

US-Patent, No. 4 639 685, Jan. 27, ’87

[14] Shattock, R.;

[3] Goodenough, F.;

Electronic Design 87, Oct. 29, pp. 67…

Exotic ICs put 200MHz signals, 15ns settling in

everyday jobs

[4] Goodenough, F.;

Electronic Design 88, vol. 2, pp. 29…

A slew of new high performance op amps shatters

speed limits

Electronic Engineering 86, Mar., pp. 59…

A novel design approach to high frequency op amps

[15] Schwehr, S.; Sibrai, A.;

[5] Lehmann, K.;

Studienarbeit p. 48, TH Darmstadt 88, Inst. f.

Halbleitertechnik Entwurf und Layouterstellung eines

Video-Operationsverstärkers in komplementärer

Bipolartechnologie. (Design and Layout of a Video

Operational Amplifier in Complementary Bipolar

Technology)

Elektronik 80, vol. 24, pp. 101…

Berechnung des Frequenzganges von Video-

Operationsverstärkern (Calculation of the Frequency

Response of Video Operational Amplifiers)

[6] Lehmann, K.;

Elektronik 80, vol. 26, pp. 81…

Schaltungstechnik von Video Operationsverstärkern

(Circuit Techniques with Video Operational

Amplifiers)

[16] Sibrai, A.;

Diplomarbeit D56, TH-Darmstadt 1/89,

Inst. f. Halbleitertechnik

Makromodellierung von Video-Operationsverstärkern

in komplementärer Bipolartechnologie (Macro Model

of Video Operational Amplifiers in Complementary

Bipolar Techonolgy)

[7] Moser, K.D.;

eee-Bauelemente 87,vol. 23, pp. 22…

Neuartige Operationsverstärker (New Kinds of

Operational Amplifiers)

[17] Yee, S.;

[8] Nelson, D.;

US-Patent, No. 3 418; Dec. 24, ’68

US-Patent, No. 4 358 739, Nov. 9,’82

AB-179 [ETC]

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