IRS29831 [INFINEON]

Primary constant power control;


型号：	IRS29831
厂家：	Infineon
描述：	Primary constant power control
文件：	总17页 (文件大小：384K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

IRS29831PBF

LED FLYBACK CONTROL IC

Product Summary

Features

Topology

Flyback

700 V

0.65 A

25 W

•

Flyback LED Driver

Integrated 700 V MOSFET

Critical-conduction / Transition mode operation

Primary constant power control

Burst mode operation at light load

Over-current protection

Drain Source Voltage

Max Drain Current

Max Converter Power

Micro power startup (150 A)

Low quiescent current (2.5 mA)

Latch immunity and ESD protection

Open load / Over voltage protection

Compatible with Triac Dimmers

High Power Factor / Low THD

Package

Typical Applications

•

LED Drivers

8-Lead DIP

Ordering Information

Base Part Number

IRS29831PBF

Standard Pack

Package Type

Complete Part Number

Form

Tube/Bulk

Quantity

DIP8

IRS29831PBF

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September 10, 2014

IRS29831PBF

Table of Contents

Page

Ordering Information

Description

Absolute Maximum Ratings

Recommended Operating Conditions

Electrical Characteristics

Functional Block Diagram

State Diagram

Input/Output Pin Equivalent Circuit Diagram

Lead Definitions

Lead Assignments

Application Information and Additional Details

Package Details

Part Marking Information

Qualification Information

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September 10, 2014

IRS29831PBF

Description

The IRS29831 is an integrated LED driver control IC and power MOSFET designed to drive Flyback and Buck-

Boost converter based LED drivers. The IRS29831 includes primary side power regulation allowing a low cost

isolated or non-isolated LED driver to be implemented without the need for an opto-isolator for a fixed LED load.

The IRS29831 is also compatible with converters that include secondary feedback circuitry. Other features of the

IRS29831 include a high voltage startup enabling VCC supply to be derived initially from the high voltage DC bus

until the auxiliary Flyback inductor winding takes over for rapid startup. The IRS29831 typically operates in critical

conduction (CrCM) with full protection against open and short circuit as well as inductor saturation. The IRS29831

may be used in single stage LED drivers with no DC bus smoothing capacitor enabling high power factor and low

THD with minimal component count.

Typical Connection Diagram (non-dimming)

DFB

CSNUB RSNUB

DSNUB

DVCC

RIN

CIN

RZX1

RVCC

IC1

COMP

VDC

BR1

CVOUT

ROUT

Line

Input

VCC

DRN

COM

CCOMP

RDC

CDC

CVCC

RZX2

DZVCC

RCS

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IRS29831PBF

Absolute Maximum Ratings

Absolute maximum ratings indicate sustained limits beyond which damage to the device may occur. All voltage

parameters are absolute voltages referenced to COM; all currents are defined positive into any lead. The thermal

resistance and power dissipation ratings are measured under board mounted and still air conditions.

Symbol

V_DS

I_DS

Definition

Min.

---

Max.

700.0

0.65

0.4

Units

Drain-Source Voltage

Drain Current at 25ºC^†

Drain Current at 100ºC^†

IC Low Voltage Supply^††

VCC current

---

I_DS

---

V_CC

I_CC

-0.3

20.8

25.0

600

V_HV

I_HV

HV Pin Voltage

-0.3

HV Pin Current

5.0

V_DS

V_COMP

V_DC

V_ZX

V_CS

I_COMP

I_DC

Drain Pin Voltage

COMP Pin Voltage

VDC Pin Voltage

ZX Pin Voltage

-0.3

700

V_CC+0.3

-0.3

VCS Pin Voltage

COMP Pin Current

VDC Pin Current

ZX Pin Current

-5

I_ZX

I_CS

CS Pin Current

Package Power Dissipation @ T_A≤ +25ºC

P_D= (T_JMAX-T_A)/R_θJA

P_D

(8-Pin DIP)

---

R_θJA

T_J

Thermal Resistance, Junction to Ambient

Junction Temperature

---

-55

---

125

300

ºC/W

T_S

Storage Temperature

ºC

T_L

Lead Temperature (soldering, 10 seconds)

Recommended Operating Conditions

For proper operation the device should be used within recommended conditions.

Symbol

V_CC

Definition

Min.

Max.

Units

V_CCUV+

Note 2

Supply Voltage

V_CCSupply Current

CS Pin Current

V_DCPin Current

ZX Pin Current

COMP Pin Current

V_DCPin Voltage

V_CSPin Voltage

Junction Temperature

I_CC

I_CS

I_DC

I_ZX

-1

I_COMP

V_DC

V_CS

T_J

6.0

2.0

0.1

-25

100

ºC

†: The MOSFET device used in this product is rated to 4A. It has been de-rated to conform to the thermal limits of the DIP8 package

assuming no heat sink is attached.

††

: This IC contains a zener clamp structure between the chip VCC and COM which has a nominal breakdown voltage of 20V. This supply pin

should not be driven by a DC, low impedance power source greater than the VCLAMP specified in the Electrical Characteristics section.

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September 10, 2014

IRS29831PBF

Electrical Characteristics

V_CC= V_BIAS=14V +/- 0.25V, C_OUT= 1000pF,

V_COMP= V_OC= V_DC= V_ZX= 0V, T_A=25ºC unless otherwise specified

Symbol

MOSFET Characteristics

Definition

Min

Typ

Max Units Test Conditions

V_DSMAX

I_DMAX

Maximum Drain-Source Voltage

700.0

---

Maximum Continuous Drain Current

0.65

Source-Drain Diode Characteristics

V_SD

T_rr

Diode Forward Voltage

Reverse Recovery Time

Reverse Recovery Charge

---

437.0

2.2

1.4

---

μC

Q_rr

---

Supply Characteristics

V_CCSupply Under Voltage Positive Going

V_CCUV

11.5

9.5

12.5

10.5

2.0

13.5

11.5

3.0

Threshold

V_CCSupply Under Voltage Negative Going

Threshold

V_CCUV

V_CCSupply Under Voltage Lockout

Hysteresis

V_UVHYS

1.5

I_QCCUV

I_CC

UVLO Mode V_CCQuiescent Current

---

150

2.5

---

5.0

V_CC= 6V

μA

V_CCSupply Current

Zener Clamp Voltage

V_CC

V_CLAMP

I_CC= 10mA

19.8

20.8

21.8

High Voltage Startup Characteristics

V_HVSMIN

Minimum Startup Voltage

30.0

---

V <

V_CCUV-

I_{HV_CHARGE}

V_CCCharge Current

HV=100V~400V

High Voltage Start-Up Circuit OFF State

Leakage Current

Error Amplifier Characteristics

I_{HVS_OFF}

---

HV=400V

μA

COMP Pin Error Amplifier Output Current

Sourcing

COMP Pin Error Amplifier Output Current

Sinking

Error Amplifier Output Voltage Swing (high

state)

Error Amplifier Output Voltage Swing (low

state)

I_COMPSOURCE

I_COMPSINK

V_COMPOH

---

13.5

2.5

V_COMPOL

GBD

V_COMPFLT

I_VDC

Error Amplifier Output Voltage in Fault Mode

Input bias current

---

-1

V_DC=0 to 3V

μA

Control Characteristics

V_ZX+

V_ZX-

ZX Pin Threshold Voltage (Arm)

ZX Pin Threshold Voltage (Trigger)

Power Regulation Reference

Multiplier Gain

1.40

0.52

---

1.54

0.6

1.68

0.68

---

V_PREF

K_MULT

1.00

2.00

1.90

2.10

V_CS=0.5V

V_DC=2.5V

COMP=4.0V

CS=1.5V

t_BLANK

OC pin current-sensing blank time

160

200

264

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IRS29831PBF

Electrical Characteristics (cont’d)

V_CC= V_BIAS=14V +/- 0.25V, C_OUT= 1000pF,

V_COMP= V_OC= V_DC= V_ZX= 0V, T_A=25ºC unless otherwise specified.

ZX = 0

μs

T_WD

PFC Watch-dog Pulse Interval

PWM Minimum ON time

100

280

135

340

COMP = 4.0V

††

t_ONMIN

220

ZX = 0

t_ONMAX

= 13V

PWM Maximum ON Time

COMP

††

μs

††

t_OFFMIN

V_DCMAX

PWM Minimum OFF Time

2.7

---

3.0

---

3.3

7.0

Maximum voltage for multiplier input^†

GBD

Signal is

averaged before

entering multiplier

input.

Maximum peak voltage for multiplier input^†

V_CSPKMAX

---

1.0

GBD

Protection Circuitry Characteristics

V_CSTH

CS Pin Over-current Sense Threshold

Cut off voltage below which gate drive output

is disabled

1.19

1.12

---

1.25

1.40

1.31

1.68

---

V_COMPOFF

V_{COMPOFF_HYS}Cut off voltage hysteresis

V_OVTH

ZX Pin Over-voltage Comparator Threshold

ZX Pin Over-voltage Comparator Hysteresis

4.90

---

5.15

200

5.40

---

V_OVHYS

†:

††:

GBD:

Multiplier operates accurately from zero to the maximum input specified.

Measured at the Drain with MOSFET switching delay also included.

Guaranteed by design.

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IRS29831PBF

Functional Block Diagram

STARTUP

VCC

VCLAMP

UVLO

DRN

VCC

VDC

Set

dominant

VPREF

K_MULT

Blank Time

Watchdog

Timer

COM

COMP

AVERAGER

VCSTH

VOVTH

VZX

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IRS29831PBF

State Diagram

Power Turned On

UVLO Mode

ICC = IQCCUV

HVREG = On

OUT = Low

COMP = Held Low

VZX < VOVTH-

VOVHYS

Delay tWD

VCC > VCCUV+

VCS < VCSTH

-VOVHYS

Fault Mode

OUT = Low

COMP = Held Low

VZX > VOVTH

Current Limit

Startup Mode

OUT = Switching

OUT = Low

VCS > VCSTH

VMULT > VPREF

VCS < VCSTH

-VOVHYS

Regulating Mode

HVREG = Off

VZX > VOVTH

OUT = Switching

VCS > VCSTH

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September 10, 2014

IRS29831PBF

Input/Output Pin Equivalent Circuit Diagrams

VCC

COMP,

ESD

VDC,

Diode

CS,

VCLAMP

ESD

Diode

COM

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IRS29831PBF

Lead Definitions

Symbol

COMP

Description

Compensation and averaging capacitor input

PFC Zero-Crossing & Over-Voltage Detection

PFC Current Sensing Input

IC Power & Signal Ground

MOSFET Drain

COM

DRN

VCC

Logic & Low-Side Gate Driver Supply

High Voltage Startup Input

DC Bus Voltage Input

VDC

Lead Assignments

COMP

VDC

VCC

DRN

COM

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IRS29831PBF

LEDs, the IRS29831 is capable of regulating the

output current indirectly by calculating and

controlling the input power of the converter. Since

an LED load has an approximately fixed voltage

the power consumed is proportional to the current.

In practice there are variations in LED forward

voltage drop due to tolerance and temperature,

however perfect accuracy is not usually required in

such applications. For a fixed number of LEDs the

current will be approximately proportional to the

input power allowing output current regulation of

+/-5% over line input from 120VAC to 230VAC.

Power regulation has been shown to provide

slightly less Lumen output variation than current

regulation.

Application Information and Additional

Details

The IRS29831 is a switched mode controller IC

with an integrated high voltage MOSFET designed

for use in Flyback and Buck-Boost converters. An

internal high voltage regulator is included to supply

the IC low voltage VCC supply allowing operation

directly from a DC input voltage up to 600V with

rapid startup at low and high AC line inputs.

Internal high voltage startup

In order to begin operating, the IRS29831 requires

its VCC supply to be raised above the under

voltage lockout positive threshold (VCCUV+) and

to continue operating requires VCC to be

maintained above the under voltage lockout

negative threshold (VCCUV-).

The IRS29831 senses input voltage and current

then averages and multiplies these quantities to

determine the input power. This is then regulated

against an accurate fixed reference to control the

LED current.

The internal high voltage start up circuit provides

the initial VCC voltage until an auxiliary winding

from the converter transformer takes over. A

series resistor RVCC and 18V zener clamp

DZVCC or other voltage limiting scheme is

necessary in line with VCC to prevent damage to

the IRS29831 if the auxiliary winding voltage

exceeds the internal clamp voltage (VCLAMP).

The HV regulator enables the IRS29831 based

LED driver to start up very rapidly and deliver light

within 0.5s of switch on at any line input voltage.

When the converter reaches steady state and

VCC can be supplied through the auxiliary winding

the HV regulator switches off for zero power

dissipation.

The line input voltage is sensed through a resistor

divider (RIN and RDC) to provide a voltage within

the range from 0V to VDCMAX. Primary current is

sensed through shunt resistor (RCS) connected

from the source of the Flyback MOSFET switch to

the DC bus return. This waveform is a high

frequency ramp rising from zero at the beginning

of each switching cycle to reach a peak level at

the point the MOSFET is switched off, remaining

at zero during the off time.

The IRS29831 is primarily targeted at LED driver

applications up to 25W using isolated or non-

isolated Flyback converter or Buck-Boost

topologies. The auxiliary winding is also used to

detect output voltage and zero-crossing point. In

the event of a short circuit at the output, the VCC

supply from the auxiliary winding collapses

causing the IRS29831 to enter under voltage

lockout and shut down. The startup sequence is

then re-initialized continuing in “hiccup” mode until

the short circuit is removed. Short circuit protection

is therefore auto-recovering enabling the driver to

tolerate the condition without damage to the

components. A capacitor in the order of 10pF at

the ZX may be required for correct operation.

V(t)

VOUT(t)

VCS(t)

Figure 1: Current sense waveform

A transconductance error amplifier (OTA) uses an

external capacitor (CCOMP) referenced to 0V to

realize an integrator that provides a stable error

voltage used to control the PWM on time. A

response time of several AC line cycles is

normally used as in typical power factor correction

circuits to enable high power factor and low THD.

Primary power regulation

To eliminate feedback circuitry and opto-isolators

where the load consists of a fixed number of

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IRS29831PBF

LED output current typically increases gradually as

the input voltage is increased. At light loads the

IRS29831 operates in burst mode to minimize

losses and maintain a stable output when the

COMP output voltage drops below the

VCOMPOFF threshold.

The pulse appearing at ZX has an amplitude

proportional to the secondary output voltage and

therefore the DC output voltage:

N_A⋅ RZX 2⋅VOUT

N_S⋅(RZX1+ RZX 2)

VZX =

[1]

Primary current limiting

Where,

At low line input voltages the power regulation

loop demands a high peak current which can

cause saturation of the primary inductor. In order

to prevent this from occurring, the IRS29831

includes cycle by cycle primary current limiting

with a fixed threshold VCSTH at the CS pin input.

Under low line or fault conditions where the

MOSFET current is abnormally high, the gate

drive switches off as the current ramps up above

VCSTH with a leading edge blanking period of

tBLANK. Leading edge blanking avoids false

tripping due to the fast high current switch on

transient caused by parasitic capacitances in the

internal MOSFET. This transient is also blanked

from the averaging input that feeds the power

regulation multiplier to prevent inaccuracies.

The IRS29831 normally operates in critical

conduction mode (CrCM), also known as transition

or boundary mode. The Flyback transformer

auxiliary winding used to supply VCC is also used

to provide the zero crossing or demagnetization

signal to the IRS29831. This indicates when all of

the energy stored in the inductor has been

transferred to the output to trigger the next

switching cycle.

N_A= Number of turns on the auxiliary winding

N_S= Number of turns on the secondary winding

VOUT = DC Output Voltage (LED voltage)

When the IRS29831 integrated MOSFET switches

off the voltage VZX transitions high. The values of

RZX1 and RZX2 must be selected so that this

voltage always exceeds the VZX+ threshold.

It should be noted that if the IRS29831 is used in a

converter that is required to drive loads with

different numbers of LEDs with a range of voltage,

an additional feedback circuit is needed to

regulate the output current. In this case the VZX

voltage needs to exceed VZX+ at the minimum

load voltage. If VZX does not exceed VZX+ the

IRS29831 will operate in discontinuous mode

(DCM) with a fixed time of tWD.

When the voltage at VZX exceeds VZX+ the

IRS29831 is armed. It then waits until VZX drops

below VZX- again to trigger the next switching

cycle.

The IRS29831 includes a minimum off time

function so that if the ZX pin input transitions high

and low before tOFFMIN the gate drive output will

not go high again until after this period. This

prevents false tripping at the ZX input and also

limits the maximum switching frequency of the

converter by entering discontinuous mode (DCM)

under conditions where the off time would

otherwise be very short. This reduces switching

losses and prevents transformer overheating.

The auxiliary winding voltage is divided through

RZX1 and RZX2 to provide the ZX pin input signal.

V(t)

VOUT(t)

VCSTH

VCS(t)

VZX(t)

V(t)

VOUT(t)

Figure 2: Cycle by cycle current limiting

Figure 3: Zero crossing detection

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IRS29831PBF

Integrated MOSFET

The IRS29831 includes an internal 700V rated

MOSFET with low RDSon to enable driver operation

up to 25W with minimal temperature rise. A snubber

network consisting of DSNUB, CSNUB and RSNUB

is required in Flyback converters to limit the peak

ringing transient. In a typical circuit with appropriate

snubber values the transient will not exceed 650V at

maximum line input.

VCOMP(t)

VZX(t)

VOUT(t)

V(t)

twd

Over voltage protection

Figure 4: Overvoltage protection

The ZX input is also used for output over voltage

protection. If the load becomes disconnected the

output voltage could potentially rise very high

damaging components as well as presenting an

electric shock hazard. In order to protect against this

the IRS29831 is able to detect the output voltage

indirectly through the proportional ZX input. If the ZX

input voltage exceeds VOVTH when the MOSFET

switches off, the gate drive switches off and remains

off for a period of tWD before beginning the next

cycle irrespective of when the ZX voltage transitions

low. In this case the IRS29831 discharges the

COMP capacitor so that the next cycle will begin at

reduced duty cycle. When the open circuit is

removed the converter recovers with a soft start.

This protection scheme allows the LED load to be

“hot” connected and disconnected from the

converter output without risk of damaging the circuit

or of high voltages appearing at the output.

Operating with a secondary feedback circuit

In applications where more accurate current

regulation over a wide input and/or output voltage

range required the IRS29831 can be used in

conjunction with a secondary sensing and feedback

circuit. This technique is also applied in designs

where dimming to low levels is required for example

in a 0-10V controlled dimmable LED driver.

The feedback circuit can be fed through an opto-

isolator or from the output of an operational

amplifier if isolation is not required.

DVCC

RPU

IC1

RZX1

COMP

VDC

The overvoltage threshold is set by choosing the

values of RZX1 and RZX2 appropriately, according

to the formula:

VCC

DRN

COM

VOVTH ⋅ N_S⋅(RZX1+ RZX 2)

CVCC

VOUT_OV

[2]

N_A⋅ RZX 2

Secondary error

feedback

RCS

The recommended over voltage threshold is 20-

25% above the normal operating voltage of the

LED load. This is important since re-connecting an

LED load with the output capacitor charged to a

higher voltage causes a high current discharge

that can cause severe damage to LEDs. A bleed

resistor is also recommended to discharge the

output capacitor.

RZX2

Figure 5: Secondary feedback circuit

A simple output voltage feedback scheme is

shown in figure 5 to demonstrate how the opto-

isolator can be connected to the IRS29831 to

create a feedback circuit. The VDC input is tied to

COM leaving the multiplier output at zero with the

COMP output pulled up to maximum by the

internal error amplifier. The opto-isolator feedback

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IRS29831PBF

pulls down on the COMP voltage to reduce the on

time as the opto-diode current is increased driven

by a secondary error amplifier circuit. A pull up

resistor (RPU) may be added to improve stability.

filter. Typical values for RPD and CPD are 470Ohms

(1W) and 100nF to 220nF (250V).

The active damping circuit based around a low

voltage MOSFET MD is used to limit the inrush

current when the triac fires. At the start of each line

cycle the triac is in the off state and during this time

MD is turned off while QD holds the gate low. This

means that when the triac fires the series damping

resistor RD limits the input current for a period of

time until MD switches on. Zener diode DZ2

normally rated at 10V prevents MD from switching

on due to residual leakage voltages that appear on

the DC bus during the period before the triac fires.

After the triac fires a voltage appears at DZ1 which

is typically rated at 16V. This causes QD to switch

off and CD to charge through RP1, RP2 and DD.

After the delay determined by these components,

MD switches on shorting out RD to remove the

damping resistance when it is no longer necessary.

Typical values for RP1, RP2 are 680K and CD is

4.7nF. DD can be a typical small diode such as a

1N4148.

The active circuit is designed to provide input

resistance to damp the circuit at the firing point

without incurring unnecessary power losses in the

damping resistor RD. A typical value for RD is

100Ohms rated at 2W.

If some power loss can be tolerated in the interests

of saving component cost RD can be reduced in

value and MD and the components driving its gate

can be removed. This low cost approach can still

provide acceptable dimming performance with a

small loss in converter efficiency.

Triac Dimming

triac dimming LED driver can be easily

implemented with the IRS29831 using a small

number of additional components. The dimming

design should be optimized to work in either the

120VAC or 220VAC range. It is not practical to

create a design with good dimming performance and

efficiency for both input voltages.

The dimming driver implementation consists of a

single stage high power factor converter previously

described with anti-ringing and active damping

circuits added to the input to provide triac stability.

The COMP output is also clamped to a maximum

level by adding an external zener diode DZ3

referenced to COM. This prevents the primary side

regulation circuit from attempting to compensate for

the reduced AC input voltage detected during

dimming. During dimming the converter operates

with the on time determined by the voltage at which

COMP is limited by the zener diode. A value of 6.8V

typically provides good results for a 120VAC system.

The schematic below shows a full implementation of

a 120VAC triac dimmable LED driver.

RPD and CPD for the anti-ringing network to

suppress high frequency oscillations that typically

occur when the dimmer triac fires caused by the

interaction of the dimmer with the LED driver input

Triac Dimmable LED Driver

DFB

CSNUB RSNUB

DSNUB

DVCC

RIN

CIN

RZX1

IC1

BR1

COMP

VDC

RVCC

CVOUT

ROUT

Line

Input

VCC

CPD

RPD

RP1

RP2

DRN

COM

CCOMP

DZ3

DZ1

DZ2

RDC

CDC

CVCC

RZX2

DZVCC

RCS

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IRS29831PBF

Package Details

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IRS29831PBF

Part Marking Information

Part number

Date code

IRS29831

YWW ?

IR logo

Pin 1

Identifier

? XXXX

Lot Code

(Prod mode –

4 digit SPN code)

MARKING CODE

Lead Free Released

Assembly site code

Per SCOP 200-002

Non-Lead Free Released

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IRS29831PBF

Qualification Information^†

Industrial^††

Comments: This family of ICs has passed JEDEC’s Industrial

qualification. IR’s Consumer qualification level is granted by

Qualification Level

extension of the higher Industrial level.

Class B

Machine Model

Human Body Model

(per JEDEC standard JESD22-A115)

ESD

Class 1C

(per ANSI/ESDA/JEDEC standard JS-001-2012)

Class I, Level A

(per JESD78)

Yes

IC Latch-Up Test

RoHS Compliant

†

Qualification standards can be found at International Rectifier’s web site http://www.irf.com/

†† Higher qualification ratings may be available should the user have such requirements. Please contact

your International Rectifier sales representative for further information.

††† Higher MSL ratings may be available for the specific package types listed here. Please contact your

International Rectifier sales representative for further information.

The information provided in this document is believed to be accurate and reliable. However, International Rectifier assumes no

responsibility for the consequences of the use of this information. International Rectifier assumes no responsibility for any infringement

of patents or of other rights of third parties which may result from the use of this information. No license is granted by implication or

otherwise under any patent or patent rights of International Rectifier. The specifications mentioned in this document are subject to

change without notice. This document supersedes and replaces all information previously supplied.

For technical support, please contact IR’s Technical Assistance Center

http://www.irf.com/technical-info/

WORLD HEADQUARTERS:

233 Kansas St., El Segundo, California 90245

Tel: (310) 252-7105

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IRS29831 [INFINEON]

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