ISL6531CBZ-T [INTERSIL]
Dual 5V Synchronous Buck Pulse-Width Modulator (PWM) Controller for DDRAM Memory VDDQ and VTT Termination; 5V双路同步降压型脉宽调制器(PWM )控制器,用于DDRAM内存VDDQ和VTT终端型号: | ISL6531CBZ-T |
厂家: | Intersil |
描述: | Dual 5V Synchronous Buck Pulse-Width Modulator (PWM) Controller for DDRAM Memory VDDQ and VTT Termination |
文件: | 总17页 (文件大小:459K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
ISL6531
®
Data Sheet
August 11, 2005
FN9053.2
Dual 5V Synchronous Buck Pulse-Width
Modulator (PWM) Controller for DDRAM
Features
• Provides V
channel DDRAM memory systems
, V
, and V voltages for one- and two-
TT
DDQ REF
Memory V
and V Termination
DDQ
TT
The ISL6531 provides complete control and protection for
dual DC-DC converters optimized for high-performance
DDRAM memory applications. It is designed to drive low
cost N-channel MOSFETs in synchronous-rectified buck
• Excellent voltage regulation
- V
- V
= 2.5V ±2% over full operating range
DDQ
1
--
=
±1% over full operating range
⋅ V
REF
DDQ
2
- V = V
TT
± 30mV
REF
topology to efficiently generate 2.5V V
for powering
DDQ
for DDRAM differential signalling,
• Supports ‘S3’ sleep mode
1
DDRAM memory, V
REF
--
and V for signal termination. The ISL6531 integrates all of
- V is held at
via a low power window
DDQ
⋅ V
TT
TT
2
regulator to minimize wake-up time
the control, output adjustment, monitoring and protection
functions into a single package.
• Fast transient response
- Full 0% to 100% duty ratio
The V
DDQ
output of the converter is maintained at 2.5V
through an integrated precision voltage reference. The V
output is precisely regulated to 1/2 the memory power
supply, with a maximum tolerance of ±1% over temperature
REF
• Operates from +5V Input
• V regulator internally compensated
TT
and line voltage variations. V accurately tracks V
.
• Overcurrent fault monitor on VDD
TT REF
During V2_SD sleep mode, the V output is maintained by
a low power window regulator.
TT
- Does not require extra current sensing element
- Uses MOSFET’s r
DS(ON)
The ISL6531 provides simple, single feedback loop, voltage-
• Drives inexpensive N-Channel MOSFETs
mode control with fast transient response for the V
DDQ
• Small converter size
regulator. The V regulator features internal compensation
TT
- 300kHz fixed frequency oscillator
that eases the design. It includes two phase-locked 300kHz
o
triangle-wave oscillators which are displaced 90 to minimize
• 24 Lead, SOIC or 32 Lead, 5mm×5mm QFN
interference between the two PWM regulators. The
regulators feature error amplifiers with a 15MHz gain-
bandwidth product and 6V/µs slew rate which enables high
converter bandwidth for fast transient performance. The
resulting PWM duty ratio ranges from 0% to 100%.
• Pb-Free Plus Anneal Available (RoHS Compliant)
Applications
• V
, V , and VREF regulation for DDRAM memory
DDQ TT
systems
The ISL6531 protects against overcurrent conditions by
inhibiting PWM operation. The ISL6531 monitors the current
- Main memory in AMD® Athlon™ and K8™, Pentium®
III, Pentium IV, Transmeta, PowerPC™, AlphaPC™, and
UltraSparc® based computer systems
in the V
regulator by using the r
of the upper
DDQ
DS(ON)
MOSFET which eliminates the need for a current sensing
resistor.
• High-power tracking DC-DC regulators
Ordering Information
TEMP
o
PART NUMBER RANGE( C)
PACKAGE
PKG. DWG. #
M24.3
ISL6531CB
0 to 70
0 to 70
24 Lead SOIC
ISL6531CBZ
(See Note)
24 Lead SOIC
(Pb-free)
M24.3
ISL6531CR
0 to 70
32 Lead 5x5 QFN L32.5x5
ISL6530/31EVAL1 Evaluation Board
Add “-T” suffix for tape and reel.
NOTE: Intersil Pb-free plus anneal products employ special Pb-free material
sets; molding compounds/die attach materials and 100% matte tin plate
termination finish, which are RoHS compliant and compatible with both SnPb
and Pb-free soldering operations. Intersil Pb-free products are MSL classified
at Pb-free peak reflow temperatures that meet or exceed the Pb-free
requirements of IPC/JEDEC J STD-020.
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2003, 2005. All Rights Reserved.
All other trademarks mentioned are the property of their respective owners.
ISL6531
Pinouts
24 LEAD (SOIC)
32 LEAD 5X5 (QFN)
TOP VIEW
TOP VIEW
24
1
2
PGND1
LGATE1
PVCC1
OCSET/SD
V2_SD
PGOOD
N/C
UGATE1
BOOT1
PHASE1
VREF
23
22
21
20
19
18
17
16
15
14
13
3
32 31 30 29 28 27 26 25
4
PHASE 1
PVCC1
OCSET/SD
V2_SD
PGOOD
N/C
1
2
3
4
5
6
7
8
24
23
22
21
20
19
18
17
FB1
5
VREF
FB1
6
COMP1
SENSE1
VREF_IN
GNDA
7
SENSE2
N/C
8
COMP1
SENSE1
VREF_IN
GNDA
9
10
11
VCC
PHASE2
BOOT2
LGATE2
PGND2
SENSE2
N/C
UGATE2 12
GNDA
VCC
9
10 11 12 13 14 15 16
2
ISL6531
Block Diagram
OCSET/SD
VCC
PGOOD
POWER-ON
RESET (POR)
+
-
SOFT-
START
BOOT1
40µA
OVER-
CURRENT
UGATE1
PHASE1
PWM
ERROR
AMP
COMPARATOR
GATE
CONTROL
LOGIC
INHIBIT
PWM
+
-
+
-
FB1
PVCC1
COMP1
0.8V
REFERENCE
LGATE1
SENSE1
VREF_IN
OSCILLATOR
PGND1
BOOT2
+
-
VREF
o
90 Phase
Shift
ERROR
AMP
UGATE2
+
-
Z
f
PWM
GATE
CONTROL
LOGIC
-
+
Z
PHASE2
c
INHIBIT
PWM
COMPARATOR
VCC
WINDOW
REGULATOR
SENSE2
V2_SD
LGATE2
PGND2
GND
3
ISL6531
Typical Application
+5V
PGOOD
R
OCSET
D
BOOT1
VCC
PGOOD
BOOT1
OCSET/SD
Q
RESET
1
UGATE1
C
GNDA
BOOT1
+5V
V
DDQ
PHASE1
L
OUT1
C
OUT1
PVCC1
Q
V2_SD
SLEEP
2
LGATE1
ISL6531
VREF_IN
PGND1
V
REF
(.5xV
)
DDQ
VREF
D
BOOT2
COMP1
BOOT2
Q
Q
3
UGATE2
C
BOOT2
PHASE2
V
TT
L
OUT2
4
C
LGATE2
PGND2
OUT2
FB1
SENSE1
SENSE2
R
FB1
FIGURE 1. TYPICAL APPLICATION FOR ISL6531
4
ISL6531
Absolute Maximum Ratings
Thermal Information
o
o
Supply Voltage, V
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +7.0V
Thermal Resistance
θ
( C/W)
θ
( C/W)
CC
JA
JC
Boot Voltage, V
Input, Output or I/O Voltage . . . . . . . . . . . .GND -0.3V to V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2
- V
. . . . . . . . . . . . . . . . . . . . . . +7.0V
+0.3V
BOOTn
PHASEn
SOIC Package (Note 1) . . . . . . . . . . . .
QFN Package (Note 2). . . . . . . . . . . . .
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . 150 C
Maximum Storage Temperature Range. . . . . . . . . . -65 C to 150 C
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300 C
65
33
N/A
4
CC
o
o
o
o
Operating Conditions
(SOIC - Lead tips only)
For Recommended soldering conditions see Tech Brief TB389.
Supply Voltage, V
CC
Ambient Temperature Range. . . . . . . . . . . . . . . . . . . . . 0 C to 70 C
Junction Temperature Range. . . . . . . . . . . . . . . . . . . . 0 C to 125 C
. . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V ±10%
o
o
o
o
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE:
1. θ is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
JA
2. θ is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. θ
the
JA
JC,
“case temp” is measured at the center of the exposed metal pad on the package underside. See Tech Brief TB379.
Electrical Specifications Recommended Operating Conditions with Vcc = 5V, unless otherwise noted.
PARAMETER
VCC SUPPLY CURRENT
Nominal Supply
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
I
OCSET/SD=V ; UGATE1, UGATE2,
CC
LGATE1, and LGATE2 Open
-
-
5
3
-
-
mA
mA
CC
Shutdown Supply
OCSET/SD=0V
POWER-ON RESET
Rising V
Threshold
Threshold
V
V
=4.5V
=4.5V
4.25
3.75
-
-
4.5
4.0
V
V
CC
OCSET/SD
Falling V
CC
OCSET/SD
OSCILLATOR
Free Running Frequency
REFERENCES
V
=5
CC
275
300
325
kHz
%SENSE1
Reference Voltage
(V2 Error Amp Reference)
V
SENSE1=2.5V
49.5
50.0
-
50.5
VREF
%
V
V1 Error Amp Reference Voltage
Tolerance
-
-
2
-
V1 Error Amp Reference
ERROR AMPLIFIERS
DC Gain
V
V
=5
CC
0.8
REF
-
-
-
82
15
6
-
-
-
dB
Gain-Bandwidth Product
Slew Rate
GBW
SR
MHz
V/µs
COMP=10pF
WINDOW REGULATOR
Load Current
-
-
±10
±7
-
-
mA
%
Output Voltage Error
V2_SD=VCC; ±10mA load on V2
GATE DRIVERS
Upper Gate Source (UGATE1 and 2)
Upper Gate Sink (UGATE1 and 2)
Lower Gate Source (LGATE1 and 2)
Lower Gate Sink (LGATE1 and 2)
PROTECTION
I
V
V
V
V
=5V, V
CC
=2.5V
UGATE
-
-
-
-
-1
1
-
-
-
-
A
A
A
A
UGATE
I
=2.5V
UGATE
UGATE-PHASE
=5V, V
I
=2.5V
-1
2
LGATE
CC
LGATE
=2.5V
I
LGATE
LGATE
OCSET/SD Current Source
OCSET/SD Disable Voltage
I
V
=4.5VDC
34
-
40
46
-
µA
OCSET
OCSET
V
0.8
V
RESET
5
ISL6531
Functional Pin Description
24 LEAD (SOIC)
32 LEAD 5X5 (QFN)
TOP VIEW
TOP VIEW
24
23
22
21
20
19
18
17
16
15
14
13
1
2
PGND1
LGATE1
PVCC1
OCSET/SD
V2_SD
PGOOD
N/C
UGATE1
BOOT1
PHASE1
VREF
3
32 31 30 29 28 27 26 25
4
PHASE 1
PVCC1
OCSET/SD
V2_SD
PGOOD
N/C
1
2
3
4
5
6
7
8
24
23
22
21
20
19
18
17
FB1
5
VREF
FB1
6
COMP1
SENSE1
VREF_IN
GNDA
7
SENSE2
N/C
8
COMP1
SENSE1
VREF_IN
GNDA
9
10
11
VCC
PHASE2
BOOT2
LGATE2
PGND2
SENSE2
N/C
UGATE2 12
GNDA
VCC
9
10 11 12 13 14 15 16
(r
) set the V
converter overcurrent (OC) trip point
DDQ
BOOT1 and BOOT2
DS(ON)
according to the following equation:
These pins provide bias voltage to the upper MOSFET
drivers. A single capacitor bootstrap circuit may be used to
create a BOOT voltage suitable to drive a standard N-
Channel MOSFET.
I
• R
OCSET
OCS
I
= -------------------------------------------
PEAK
r
DS(ON)
UGATE1 and UGATE2
An overcurrent trip cycles the soft-start function.
Connect UGATE1 and UGATE2 to the corresponding upper
MOSFET gate. These pins provide the gate drive for the
upper MOSFETs. UGATE2 is also monitored by the adaptive
shoot through protection to determine when the upper FET
Pulling the OCSET/SD pin to ground resets the ISL6531 and
all external MOSFETS are turned off allowing the two output
voltage power rails to float.
of the V regulator has turned off.
TT
PGOOD
A high level on this open-drain output indicates that both the
LGATE1 and LGATE2
V
and V regulators are within normal operating
DDQ
TT
Connect LGATE1 and LGATE2 to the corresponding lower
MOSFET gate. These pins provide the gate drive for the
lower MOSFETs. These pins are monitored by the adaptive
shoot through protection circuitry to determine when the
lower FET has turned off.
voltage ranges.
GNDA
Signal ground for the IC. Tie this pin to the ground plane
through the lowest impedence connection available.
PGND1 and PGND2
VCC
These are the power ground connections for the gate drivers
of the PWM controllers. Tie these pins to the ground plane
through the lowest impedence connection available.
The 5V bias supply for the chip is connected to this pin. This
pin is also the positive supply for the lower gate driver,
LGATE2. Connect a well decoupled 5V supply to this pin.
OCSET/SD
V2_SD
A resistor (R
) connected from this pin to the drain of
the upper MOSFET of the V regulator sets the
OCSET
A high level on the V2_SD input places the V controller
TT
into “sleep” mode. In sleep mode, both UGATE2 and
LGATE2 are driven low, effectively floating the V supply.
TT
DDQ
, an internal 40µA current
), and the upper MOSFET on-resistance
overcurrent trip point. R
OCSET
source (I
OCS
6
ISL6531
While the V supply “floats”, it is held to about 50% of
TT
300kHz clocks. The clocks are phase locked and displaced
90 to minimize noise coupling between the controllers.
o
V
via a low current window regulator which drives V
DDQ
via the SENSE2 pin. The window regulator can overcome up
to at least ±10mA of leakage on V
TT
The first regulator includes a precision 0.8V reference and is
.
TT
intended to provide the proper V
system. The V
DDQ
to a DDRAM memory
controller implements overcurrent
DDQ
While V2_SD is high, PGOOD is low.
protection utilizing the r
Following a fault condition, the V
via a digital soft-start circuit.
of the upper MOSFET.
DS(ON)
PHASE1 and PHASE2
regulator is softstarted
DDQ
Connect PHASE1 and PHASE2 to the corresponding upper
MOSFET source. This pin is used as part of the upper
MOSFET bootstrapped drives. PHASE1 is used to monitor
the voltage drop across the upper MOSFET of the V
regulator for overcurrent protection. The PHASE1 pin is
monitored by the adaptive shoot through protection circuitry
Included in the ISL6531 is a precision V
reference
REF
1
--
output. V
is a buffered representation of
. V
⋅ V
REF
REF
DDQ
DDQ
2
is derived via a precision internal resistor divider connected
to the SENSE1 terminal.
to determine when the upper FET of the V
turned off.
supply has
DDQ
The second PWM regulator is designed to provide V
TT
termination for the DDRAM signal lines. The reference to the
V
regulator is V
. Thus the V regulator provides a
TT
termination voltage equal to
REF
TT
FB1, COMP1
1
--
DDQ. The drain of the
⋅ V
2
COMP1 and FB1 are the available external pins of the error
amplifier for the V regulator. The FB1 pin is the inverting
inputs of the error amplifier and the COMP1 pin is the
associated output. An appropriate AC network across these
pins is used to compensate the voltage-controlled feedback
upper MOSFET of the V supply is connected to the
regulated V
enable both sinking and sourcing current on the V rail.
TT
TT
DDQ
voltage. The V controller is designed to
DDQ
TT
Two benefits result from the ISL6531 dual controller
1
--
loop of the V
DDQ
converter.
topology. First, as VREF is always
, the V supply
TT
⋅ V
DDQ
2
will track the V
the overcurrent protection incorporated into the V
DDQ
will simultaneously protect the V supply.
supply during soft-start cycles. Second,
DDQ
VREF and VREF_IN
supply
VREF produces a voltage equal to one half of the voltage on
SENSE1. This low current output is connected to the VREF
input of the DDRAM devices being powered. This same
TT
Initialization
voltage is used as the reference input of the V error
The ISL6531 automatically initializes upon application of
input power. Special sequencing of the input supplies is not
necessary. The Power-On Reset (POR) function continually
monitors the input bias supply voltage at the VCC pin. The
POR function initiates soft-start operation after the 5V bias
supply voltage exceeds its POR threshold.
TT
amplifier. Thus V is controlled to 50% of V
TT
.
DDQ
VREF_IN is used as an option to overdrive the internal
resistor divider network that sets the voltage for both
VREF_OUT and the reference voltage for the V supply. A
TT
100pF capacitor between VREF_IN and ground is
recommended for proper operation.
Soft-Start
The POR function initiates the digital soft start sequence. The
PWM error amplifier reference input for the VDDQ regulator is
clamped to a level proportional to the soft-start voltage. As the
soft-start voltage slews up, the PWM comparator generates
PHASE pulses of increasing width that charge the output
capacitor(s). This method provides a rapid and controlled
output voltage rise. The soft-start sequence typically takes
about 7ms.
PVCC1
This is the positive supply for the lower gate driver, LGATE1.
PVCC1 is connected to a well decoupled 5V.
SENSE1 and SENSE2
Both SENSE1 and SENSE2 are connected directly to the
regulated outputs of the V
respectively. SENSE1 is used as an input to create the
and V supplies,
DDQ
TT
1
2
voltage at VREF_OUT and the reference voltage for the V
TT
--
With the V regulator reference held at
TT
it will
⋅ V
DDQ
supply. SENSE2 is used as the feedback pin of the V
TT
automatically track the ramp of the V
softstart, thus
DDQ
regulator and as the regulation point for the window regulator
that is enabled in V2_SD mode.
enabling a soft-start for V
.
TT
Figure 2 shows the soft-start sequence for a typical application.
At T0, the +5V VCC bias voltage starts to ramp. Once the
voltage on VCC crosses the POR threshold at time T1, both
outputs begin their soft-start sequence. The triangle waveforms
from the PWM oscillators are compared to the rising error
amplifier output voltage. As the error amplifier voltage
increases, the pulse-widths on the UGATE pins increase to
reach their steady-state duty cycle at time t2.
Functional Description
Overview
The ISL6531 contains control and drive circuitry for two
synchronous buck PWM voltage regulators. Both regulators
utilize 5V bootstrapped output topology to allow use of low
cost N-Channel MOSFETs. The regulators are driven by
7
ISL6531
When the V2_SD input of the ISL6531 is driven high, the
regulator is placed into a “sleep” state. In the sleep state
V
TT
the main V regulator is disabled, with both the upper and
TT
VCC (5V)
lower MOSFETs being turned off. The V bus is maintained
TT
via a low current window regulator
(1V/DIV)
1
2
--
at close to
⋅ V
DDQ
which drives V via the SENSE2 pin. Maintaining V at
TT
TT
1
--
consumes negligible power and enables rapid
wake-up from sleep mode without the need of softstarting
⋅ V
DDQ
2
V
(2.5V)
DDQ
the V regulator. During this power down mode, PGOOD is
TT
held LOW.
V
(1.25V)
TT
Output Voltage Selection
0V
The output voltage of the V
regulator can be
DDQ
programmed to any level between V (i.e. +5V) and the
IN
T2
T0
T1
internal reference, 0.8V. An external resistor divider is used
to scale the output voltage relative to the reference voltage
and feed it back to the inverting input of the error amplifier,
see Figure 3.F However, since the value of R1 affects the
values of the rest of the compensation components, it is
advisable to keep its value less than 5kΩ. R4 can be
calculated based on the following equation:
TIME
FIGURE 2. SOFT-START INTERVAL
Shoot-Through Protection
A shoot-through condition occurs when both the upper
MOSFET and lower MOSFET are turned on simultaneously,
effectively shorting the input voltage to ground. To protect
the regulators from a shoot-through condition, the ISL6531
incorporates specialized circuitry which insures that
complementary MOSFETs are not ON simultaneously.
R1 × 0.8V
R4 = -------------------------------------
V
– 0.8V
OUT1
If the output voltage desired is 0.8V, simply route V
to the FB pin through R1, but do not populate R4.
back
DDQ
The adaptive shoot-through protection utilized by the V
DDQ
regulator looks at the lower gate drive pin, LGATE1, and the
phase node, PHASE1, to determine whether a MOSFET is
ON or OFF. If PHASE1 is below 0.8V, the upper gate is
defined as being OFF. Similarly, if LGATE1 is below 0.8V, the
lower MOSFET is defined as being OFF. This method of
+5V
D1
VCC
BOOT1
shoot-through protection allows the V
source current only.
regulator to
DDQ
C4
Q1
Due to the necessity of sinking current, the V regulator
TT
employs a modified protection scheme from that of the
UGATE1
L
OUT1
V
DDQ
PHASE1
ISL6531
V
regulator. If the voltage from UGATE2 or from
DDQ
Q2
LGATE2 to GND is less than 0.8V, then the respective
MOSFET is defined as being OFF and the other MOSFET is
turned ON.
LGATE1
+
C
OUT1
FB1
Since the voltage of the lower MOSFET gates and the upper
C1
R1
C3
COMP1
MOSFET gate of the V supply are being measured to
R3
TT
determine the state of the MOSFET, the designer is
encouraged to consider the repercussions of introducing
external components between the gate drivers and their
respective MOSFET gates before actually implementing
such measures. Doing so may interfere with the shoot-
through protection.
C2
R2
R4
FIGURE 3. OUTPUT VOLTAGE SELECTION OF V
DDQ
Power Down Mode
DDRAM systems include a sleep state in which the V
voltage to the memories is maintained, but signaling is
V
Reference Overdrive
TT
The ISL6531 allows the designer to bypass the internal 50%
tracking of V that is used as the reference for V . The
DDQ
DDQ TT
suspended. During this mode the V termination voltage is
ISL6531 was designed to divide down the V voltage by
TT
DDQ
no longer needed. The only load placed on the V bus is
TT
50% through two internal matched resistances. These
the leakage of the associated signal pins of the DDRAM and
resistances are typically 200kΩ.
memory controller ICs.
8
ISL6531
One method that may be employed to bypass the internal
reference generation is to supply an external reference
directly to the VREF_IN pin. When doing this the SENSE1
pin must remain unconnected. Caution must be exercised
programs the overcurrent trip level (see Figure 1). An internal
V
40µA (typical) current sink develops a voltage across R
TT
OCSET
that is referenced to V . When the voltage across the upper
IN
MOSFET of V
voltage across R
(also referenced to V ) exceeds the
DDQ
IN
, the overcurrent function initiates a
when using this method as the V regulator does not
TT
OCSET
employ a soft start of its own.
soft-start sequence.
Figure 5 illustrates the protection feature responding to an
overcurrent event on V . At time t0, an overcurrent
A second method would be to overdrive the internal
resistors. Figure 3 shows how to implement this method. The
external resistors used to overdrive the internal resistors
should be less than 2kΩ and have a tolerance of 1% or
better. This method still supplies a buffer between the
resistor network and any loading on the VREF pin. If there is
no loading on the VREF pin, then no buffering is necessary
and the reference voltage created by the resistor network
can be tied directly to VREF.
DDQ
condition is sensed across the upper MOSFET of the V
DDQ
regulator. As a result, both regulators are quickly shutdown
and the internal soft-start function begins producing soft-
start ramps. The delay interval seen by the output is
equivalent to three soft-start cycles. The fourth internal soft-
start cycle initiates a normal soft-start ramp of the output, at
time t1. Both outputs are brought back into regulation by time
t2, as long as the overcurrent event has cleared.
V
DDQ
ISL6531
Had the cause of the overcurrent still been present after the
delay interval, the overcurrent condition would be sensed
and both regulators would be shut down again for another
delay interval of three soft start cycles. The resulting hiccup
mode style of protection would continue to repeat
indefinitely.
SENSE1
VREF_IN
R
R
A
VREF
+
-
B
V
(2.5V)
DDQ
TO ERROR
AMPLIFIER
V
(1.25V)
TT
FIGURE 4. V REFERENCE OVERDRIVE
TT
Converter Shutdown
0V
Pulling and holding the OCSET/SD pin below 0.8V will
shutdown both regulators. During this state, PGOOD will be
held LOW. Upon release of the OCSET/SD pin, the IC enters
into a soft start cycle which brings both outputs back into
regulation.
INTERNAL SOFT-START FUNCTION
DELAY INTERVAL
Voltage Monitoring
The ISL6531 offers a PGOOD signal that will communicate
whether the regulation of both V
and V are within
DDQ
TT
±15% of regulation, the V2_SD pin is held low and the bias
voltage of the IC is above the POR level. If all the criteria
above are true, the PGOOD pin will be at a high impedence
level. When one or more of the criteria listed above are false,
the PGOOD pin will be held low.
T1
T0
T2
TIME
FIGURE 5. OVERCURRENT PROTECTION RESPONSE
Overcurrent Protection
The overcurrent function will trip at a peak inductor current
(I
determined by:
The overcurrent function protects the converter from a shorted
PEAK)
I
x R
OCSET
output by using the upper MOSFET on-resistance, r
, of
DS(ON)
OCSET
I
= ----------------------------------------------------
PEAK
V
to monitor the current. This method enhances the
r
DDQ
DS(ON)
converter’s efficiency and reduces cost by eliminating a
current sensing resistor.
where I
OCSET
is the internal OCSET current source (40µA
typical). The OC trip point varies mainly due to the MOSFET
variations. To avoid overcurrent tripping in the
The overcurrent function cycles the soft-start function in a
r
DS(ON)
hiccup mode to provide fault protection. A resistor (R
)
OCSET
9
ISL6531
normal operating load range, find the R
the equation above with:
resistor from
OCSET
+5V
1. The maximum r
temperature.
at the highest junction
ISL6531
DS(ON)
UGATE1
PHASE1
V
DDQ
2. The minimum I
3. Determine I
from the specification table.
OCSET
LGATE1
(∆I)
I
> I
+ ----------
,
OUT(MAX)
for
PEAK
PEAK
2
DDR
SDRAM
where ∆I is the output inductor ripple current.
UGATE2
PHASE2
For an equation for the ripple current see the section under
component guidelines titled Output Inductor Selection.
+
-
V
TT
R
T
LGATE2
V
REF
A small ceramic capacitor should be placed in parallel with
R
to smooth the voltage across
R
in the
OCSET
OCSET
presence of switching noise on the input voltage.
FIGURE 6. V CURRENT SINKING LOOP
TT
Current Sinking
Application Guidelines
The ISL6531 V regulator incorporates a MOSFET shoot-
TT
Layout Considerations
through protection method which allows the converter to sink
current as well as source current. Care should be exercised
when designing a converter with the ISL6531 when it is
known that the converter may sink current.
Layout is very important in high frequency switching
converter design. With power devices switching efficiently at
300kHz, the resulting current transitions from one device to
another cause voltage spikes across the interconnecting
impedances and parasitic circuit elements. These voltage
spikes can degrade efficiency, radiate noise into the circuit,
and lead to device over-voltage stress. Careful component
layout and printed circuit board design minimizes the voltage
spikes in the converters.
When the converter is sinking current, it is behaving as a
boost converter that is regulating its input voltage. This
means that the converter is boosting current into the input
rail of the regulator. If there is nowhere for this current to go,
such as to other distributed loads on the rail or through a
voltage limiting protection device, the capacitance on this rail
will absorb the current. This situation will the allow voltage
level of the input rail to increase. If the voltage level of the rail
is boosted to a level that exceeds the maximum voltage
rating of any components attached to the input rail, then
those components may experience an irreversible failure or
experience stress that may shorten their lifespan. Ensuring
that there is a path for the current to flow other than the
capacitance on the rail will prevent this failure mode.
As an example, consider the turn-off transition of the PWM
MOSFET. Prior to turn-off, the MOSFET is carrying the full
load current. During turn-off, current stops flowing in the
MOSFET and is picked up by the lower MOSFET. Any
parasitic inductance in the switched current path generates a
large voltage spike during the switching interval. Careful
component selection, tight layout of the critical components,
and short, wide traces minimizes the magnitude of voltage
spikes.
To insure that the current does not boost up the input rail
voltage of the V regulator, it is recommended that the
There are two sets of critical components in a DC-DC
converter using the ISL6531. The switching components are
the most critical because they switch large amounts o
energy, and therefore tend to generate large amounts of
noise. Next are the small signal components which connect
to sensitive nodes or supply critical bypass current and
signal coupling.
TT
input rail of the V regulator be the output of the V
TT
DDQ
regulator. The current being sunk by the V regulator will
TT
be fed into the V
DDQ
rail and then drawn into the DDR
SDRAM memory module and back into the V regulator.
TT
Figure 6 shows the recommended configuration and the
resulting current loop.
A multi-layer printed circuit board is recommended. Figure 7
shows the connections of the critical components in the
converter. Note that capacitors C and C
could each
IN OUT
represent numerous physical capacitors. Dedicate one solid
layer, usually a middle layer of the PC board, for a ground
plane and make all critical component ground connections
with vias to this layer. Dedicate another solid layer as a
power plane and break this plane into smaller islands of
common voltage levels. Keep the metal runs from the
PHASE terminals to the output inductor short. The power
plane should support the input power and output power
10
ISL6531
nodes. Use copper filled polygons on the top and bottom
The critical small signal components include any bypass
capacitors, feedback components, and compensation
circuit layers for the phase nodes. Use the remaining printed
circuit layers for small signal wiring. The wiring traces from
the GATE pins to the MOSFET gates should be kept short
and wide enough to easily handle the 1A of drive current.f
components. Position the bypass capacitor, C , close to the
BP
VCC pin with a via directly to the ground plane. Place the
PWM converter compensation components close to the FB
and COMP pins. The feedback resistors for both regulators
should also be located as close as possible to the relevant
FB pin with vias tied straight to the ground plane as required.
+5V V
IN
ISL6531
VCC
GND
V
Feedback Compensation
DDQ
This section discusses the feedback compensation of the
regulator. Figure 8 highlights the voltage-mode
C
BP
C
IN
D1
V
DDQ
control loop for a synchronous-rectified buck converter. The
output voltage (V ) is regulated to the Reference voltage
BOOT1
C
BOOT1
OUT
Q1
Q2
level. The error amplifier (error amp) output (V ) is
compared with the oscillator (OSC) triangular wave to
provide a pulse-width modulated (PWM) wave with an
E/A
UGATE1
PHASE1
L
OUT1
V
DDQ
PHASE1
amplitude of V at the PHASE node. The PWM wave is
IN
C
LGATE1
PGND1
COMP1
OUT1
smoothed by the output filter (L and C ).
O
O
The modulator transfer function is the small-signal transfer
function of V /V . This function is dominated by a DC
OUT E/A
gain and the output filter (L and C ), with a double pole
C
2A
C
2A
O
O
1A
R
break frequency at F and a zero at F
. The DC gain of
R
1A
LC
ESR
FB1
the modulator is simply the input voltage (V ) divided by the
IN
C
R
3A
3A
R4
peak-to-peak oscillator voltage ∆V
.
OSC
V
SENSE1
DRIVER
IN
OSC
PWM
+5V V
IN
L
COMPARATOR
O
V
OUT
DV
OSC
V
D2
DDQ
DRIVER
-
PHASE
+
C
O
BOOT2
C
BOOT2
ESR
(PARASITIC)
Q3
UGATE2
PHASE2
L
Z
OUT2
FB
V
TT
PHASE2
V
E/A
Z
-
IN
+
Q4
C
LGATE2
PGND2
OUT2
REFERENCE
ERROR
AMP
DETAILED COMPENSATION COMPONENTS
SENSE2
Z
FB
V
OUT
C
1
Z
KEY
IN
C
ISLAND ON POWER PLANE LAYER
ISLAND ON CIRCUIT PLANE LAYER
C
R
R
3
2
3
2
R
1
COMP
VIA CONNECTION TO GROUND PLANE
FB
-
FIGURE 7. PRINTED CIRCUIT BOARD POWER PLANES
AND ISLANDS
+
ISL6531
REFERENCE
The switching components should be placed close to the
ISL6531 first. Minimize the length of the connections
FIGURE 8. VOLTAGE-MODE BUCK CONVERTER
COMPENSATION DESIGN
between the input capacitors, C , and the power switches
IN
by placing them nearby. Position both the ceramic and bulk
input capacitors as close to the upper MOSFET drain as
possible. Position the output inductor and output capacitors
between the upper MOSFET and lower diode and the load.
Modulator Break Frequency Equations
1
1
F
= -----------------------------------------
F
= ------------------------------------------
LC
ESR
2π x ESR x C
2π x
L
x C
O O
O
11
ISL6531
The compensation network consists of the error amplifier
(internal to the ISL6531) and the impedance networks Z
The compensation gain uses external impedance networks
and Z to provide a stable, high bandwidth (BW) overall
Z
IN
FB
IN
and Z . The goal of the compensation network is to provide
a closed loop transfer function with the highest 0dB crossing
loop. A stable control loop has a gain crossing with
-20dB/decade slope and a phase margin greater than 45
degrees. Include worst case component variations when
determining phase margin
FB
frequency (f
) and adequate phase margin. Phase margin
is the difference between the closed loop phase at f and
0dB
0dB
180 degrees. The equations below relate the compensation
network’s poles, zeros and gain to the components (R , R ,
V
Feedback Compensation
TT
To ease design and reduce the number of small-signal
1
2
R , C , C , and C ) in Figure 7. Use these guidelines for
3
1
2
3
components required, the V regulator is internally
TT
locating the poles and zeros of the compensation network:
compensated. The only stability criteria that needs to be
met relates the minimum value of the inductor to the
equivalent ESR of the output capacitor bank as shown in
the following equation:
1. Pick gain (R /R ) for desired converter bandwidth.
2
1
2. Place first zero below filter’s double pole (~75% F ).
LC
3. Place second zero at filter’s double pole.
4. Place first pole at the ESR zero.
–6
L
≥ 20 ⋅ (10 ) × ESR
× V
OUT IN
OUT(MIN)
5. Place second pole at half the switching frequency.
6. Check gain against error amplifier’s open-loop gain.
7. Estimate phase margin - repeat if necessary.
where
L
= minimum output inductor value at full output
= equivalent ESR of the output capacitor bank
OUT(MIN)
Compensation Break Frequency Equations
current
1
1
ESR
F
F
= ----------------------------------
F
F
= --------------------------------------------------------
OUT
Z1
Z2
P1
P2
2π × R × C
C
x C
2
2
1
2
---------------------
2π x R
x
V
= Input voltage of the converter
2
IN
C
+ C
2
1
The design procedure for this output should follow the
following steps:
1
1
= ------------------------------------------------------
2π x (R + R ) x C
= -----------------------------------
2π x R x C
3
1
3
3
3
1. Choose the number and type of output capacitors to meet
the output transient requirements based on the dynamic
loading characteristics of the output.
Figure 9 shows an asymptotic plot of the DC-DC converter’s
gain vs frequency. The actual Modulator Gain has a high gain
peak due to the high Q factor of the output filter and is not
shown in Figure 9. Using the above guidelines should give a
Compensation Gain similar to the curve plotted. The open
loop error amplifier gain bounds the compensation gain.
2. Determine the equivalent ESR of the output capacitor
bank and calculate the minimum output inductor value.
3. Verify that the chosen inductor meets this minimum value
criteria at full output load. It is recommended that the
chosen inductor be no more than 30% saturated at full
output load.
Check the compensation gain at F with the capabilities of
P2
the error amplifier. The Closed Loop Gain is constructed on
the graph of Figure 9 by adding the Modulator Gain (in dB) to
the Compensation Gain (in dB). This is equivalent to
multiplying the modulator transfer function to the
Component Selection Guidelines
Output Capacitor Selection
An output capacitor is required to filter the output and supply
the load transient current. The filtering requirements are a
function of the switching frequency and the ripple current.
The load transient requirements are a function of the slew
rate (di/dt) and the magnitude of the transient load current.
These requirements are generally met with a mix of
capacitors and careful layout.
compensation transfer function and plotting the gain..
OPEN LOOP
ERROR AMP GAIN
F
F
F
P1
F
Z1
Z2
P2
100
80
V
IN
---------------
20log
V
OSC
60
40
COMPENSATION
GAIN
Modern digital ICs can produce high transient load slew
rates. High frequency capacitors initially supply the transient
and slow the current load rate seen by the bulk capacitors.
The bulk filter capacitor values are generally determined by
the ESR (effective series resistance) and voltage rating
requirements rather than actual capacitance requirements.
20
0
R2
-------
20log
R1
-20
-40
-60
MODULATOR
GAIN
LOOP GAIN
10M
F
F
ESR
LC
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
10
100
1K
10K
100K
1M
FREQUENCY (Hz)
FIGURE 9. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
12
ISL6531
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.
response time to the removal of load. The worst case
response time can be either at the application or removal of
load. Be sure to check both of these equations at the
minimum and maximum output levels for the worst case
response time.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor’s ESR will determine the output ripple voltage
and the initial voltage drop after a high slew-rate transient. An
aluminum electrolytic capacitor’s ESR value is related to the
case size with lower ESR available in larger case sizes.
However, the equivalent series inductance (ESL) of these
capacitors increases with case size and can reduce the
usefulness of the capacitor to high slew-rate transient loading.
Unfortunately, ESL is not a specified parameter. Work with
your capacitor supplier and measure the capacitor’s
Input Capacitor Selection
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic
capacitors for high frequency decoupling and bulk capacitors
to supply the current needed each time Q turns on. Place the
1
small ceramic capacitors physically close to the MOSFETs
and between the drain of Q and the source of Q .
1
2
The important parameters for the bulk input capacitor are the
voltage rating and the RMS current rating. For reliable
operation, select the bulk capacitor with voltage and current
ratings above the maximum input voltage and largest RMS
current required by the circuit. The capacitor voltage rating
should be at least 1.25 times greater than the maximum
input voltage and a voltage rating of 1.5 times is a
conservative guideline. The RMS current rating requirement
for the input capacitor of a buck regulator is approximately
1/2 the DC load current.
impedance with frequency to select a suitable component. In
most cases, multiple electrolytic capacitors of small case size
perform better than a single large case capacitor.
Output Inductor Selection
The output inductor is selected to meet the output voltage
ripple requirements and minimize the converter’s response
time to the load transient. Additionally, the output inductor for
the V regulator has to meet the minimum value criteria for
TT
loop stability as described in the V Feedback
TT
The maximum RMS current required by the regulator may be
closely approximated through the following equation:
Compensation section. The inductor value determines the
converter’s ripple current and the ripple voltage is a function
of the ripple current. The ripple voltage and current are
approximated by the following equations:
2
V
V
– V
V
OUT
V
IN
2
1
12
OUT
IN
OUT
---------------
------
------------------------------- ---------------
I
=
× I
+
×
×
RMS
OUT
V
L × f
MAX
MAX
IN
s
V
- V
OUT
V
OUT
IN
f x L
∆V
OUT
= ∆I x ESR
∆I =
x
For a through hole design, several electrolytic capacitors may
be needed. For surface mount designs, solid tantalum
capacitors can be used, but caution must be exercised with
regard to the capacitor surge currentrating. These capacitors
must be capable of handling the surge-current at power-up.
Some capacitor series available from reputable manufacturers
are surge current tested.
V
s
IN
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to
a load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
ISL6531 will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
Minimizing the response time can minimize the output
capacitance required.
MOSFET Selection/Considerations
The ISL6531 requires two N-Channel power MOSFETs for
each PWM regulator. These should be selected based upon
r
, gate supply requirements, and thermal management
DS(ON)
requirements.
In high-current applications, the MOSFET power dissipation,
package selection and heatsink are the dominant design
factors. The power dissipation includes two loss components;
conduction loss and switching loss. The conduction losses are
the largest component of power dissipation for both the upper
and the lower MOSFETs. These losses are distributed between
the two MOSFETs according to duty factor. The switching
losses seen when sourcing current will be different from the
The response time to a transient is different for the
application of load and the removal of load. The following
equations give the approximate response time interval for
application and removal of a transient load:
switching losses seen when sinking current. The V
regulator will only source current while the V regulator can
TT
L x I
L x I
V
DDQ
TRAN
- V
OUT
TRAN
t
=
t
=
FALL
RISE
V
IN
OUT
sink and source. When sourcing current, the upper MOSFET
realizes most of the switching losses. The lower switch realizes
most of the switching losses when the converter is sinking
where: I
is the transient load current step, t
is the
TRAN
RISE
is the
response time to the application of load, and t
FALL
13
ISL6531
current (see the equations below). These equations assume
linear voltage-current transitions and do not adequately model
power loss due the reverse-recovery of the upper and lower
MOSFET’s body diode. The gate-charge losses are dissipated
by the ISL6531 and don't heat the MOSFETs. However, large
VCC
BOOT
D
+
V
IN
V
D
-
BOOTn
C
gate-charge increases the switching interval, t
increases the MOSFET switching losses.
which
ISL6531
BOOT
SW
UGATEn
PHASEn
Q
Q
UPPER
LOSSES WHILE SOURCING CURRENT
NOTE:
≈ V
V
-V
G-S
CC
D
2
1
2
--
× D + ⋅ Io × V × t
P
= Io × r
× f
UPPER
DS(ON)
IN
SW
s
2
LGATEn
LOWER
-
P
= Io x r
x (1 - D)
DS(ON)
LOWER
+
NOTE:
≈ V
LOSSES WHILE SINKING CURRENT
2
V
G-S
CC
P
P
= Io x r x D
DS(ON)
GND
UPPER
2
1
2
--
× (1 – D) + ⋅ Io × V × t
= Io × r
× f
s
LOWER
DS(ON)
IN
SW
FIGURE 10. UPPER GATE DRIVE BOOTSTRAP
Where: D is the duty cycle = V
OUT
/ V ,
IN
is the combined switch ON and OFF time, and
t
SW
f is the switching frequency.
where Q
is the maximum total gate charge of the upper
GATE
MOSFET, C
s
is the bootstrap capacitance, V
is
BOOT
BOOT1
the bootstrap voltage immediately before turn-on, and
is the bootstrap voltage immediately after turn-on.
Ensure that both MOSFETs are within their maximum junction
temperature at high ambient temperature by calculating the
temperature rise according to package thermal-resistance
specifications. A separate heat sink may be necessary
depending upon MOSFET power, package type, ambient
temperature and air flow.
V
BOOT2
The bootstrap capacitor begins its refresh cycle when the
gate drive begins to turn-off the upper MOSFET. A refresh
cycle ends when the upper MOSFET is turned on again,
which varies depending on the switching frequency and
duty cycle.
Given the reduced available gate bias voltage (5V), logic-
level or sub-logic-level transistors should be used for both N-
MOSFETs. Caution should be exercised when using devices
The minimum bootstrap capacitance can be calculated by
rearranging the previous equation and solving for CBOOT.
with very low gate thresholds (V ). The shoot-through
protection circuitry may be circumvented by these
MOSFETs. Very high dv/dt transitions on the phase node
may cause the Miller capacitance to couple the lower gate
with the phase node and cause an undesireable turn on of
the lower MOSFET while the upper MOSFET is on.
TH
Q
GATE
----------------------------------------------------
C
≥
BOOT
V
– V
BOOT1
BOOT2
Typical gate charge values for MOSFETs considered in
these types of applications range from 20 to 100nC. Since
the voltage drop across Q
is negligible, V is
LOWER
BOOT1
Bootstrap Component Selection
simply VCC - V . A schottky diode is recommended to
D
External bootstrap components, a diode and capacitor, are
required to provide sufficient gate enhancement to the upper
MOSFET. The internal MOSFET gate driver is supplied by
the external bootstrap circuitry as shown in Figure 10. The
minimize the voltage drop across the bootstrap capacitor
during the on-time of the upper MOSFET. Initial calculations
with V
no less than 4V will quickly help narrow the
BOOT2
bootstrap capacitor range.
boot capacitor, C
, develops a floating supply voltage
BOOT
For example, consider an upper MOSFET is chosen with a
referenced to the PHASE pin. This supply is refreshed each
cycle, when D conducts, to a voltage of VCC less the
maximum gate charge, Q , of 100nC. Limiting the voltage
g
BOOT
boot diode drop, V , plus the voltage rise across Q
drop across the bootstrap capacitor to 1V results in a value
of no less than 0.1µF. The tolerance of the ceramic capacitor
should also be considered when selecting the final bootstrap
capacitance value.
.
D
LOWER
Just after the PWM switching cycle begins and the charge
transfer from the bootstrap capacitor to the gate capacitance
is complete, the voltage on the bootstrap capacitor is at its
lowest point during the switching cycle. The charge lost on
the bootstrap capacitor will be equal to the charge
transferred to the equivalent gate-source capacitance of the
upper MOSFET as shown:
A fast recovery diode is recommended when selecting a
bootstrap diode to reduce the impact of reverse recovery
charge loss. Otherwise, the recovery charge, Q , would
RR
have to be added to the gate charge of the MOSFET and
taken into consideration when calculating the minimum
bootstrap capacitance.
Q
= C
× (V
– V
)
BOOT2
GATE
BOOT
BOOT1
14
ISL6531
ISL6531 DC-DC Converter Application Circuit
Figure 11 shows an application circuit for a DDR SDRAM
power supply, including V (+2.5V) and V (+1.25V).
Detailed information on the circuit, including a complete
Billof-Materials and circuit board description, can be found
inApplication Note AN9993-of-Materials and circuit board
description, can be found in Application Note AN9993
Application Note AN9993-of-Materials and circuit board
description, can be found in Application Note AN9993.
DDQ TT
+5V
R
C
2
1
C
1
3.48kΩ
0.1µF
1000pF
D
1
C
4,5
150µF(x2)
C
3
1.0µF
OCSET/SD
VCC
BOOT1
V2_SD
Q
1
PGOOD
UGATE1
PHASE1
C
VREF
6
V
0.1µF
DDQ
@10A
L
1
1µH
VREF_IN
PVCC1
C
7,8,9,10
150µF(x4)
Q
C
2
30
100pF
LGATE1
GNDA
C
15
0.1µF
PGND1
ISL6531
D
2
C
26
5600pF
COMP1
BOOT2
C
17
1.0µF
C
27
100pF
R
26
UGATE2
6.34kΩ
C
16
0.1µF
V
TT
@5A
L
PHASE2
LGATE2
2
1µH
R
C
20
18,19
150µF(x2)
FB1
1.43kΩ
Q
3
PGND2
SENSE1
SENSE2
R
19
3.01kΩ
C
25
15000pF
R
25
100Ω
L1,2 - Each 1mH Inductor, Panasonic P/N ETQ-P6F1ROSFA
Q1,2 - Each Fairchild MOSFET; ITF86130DK8
Q3 - Fairchild MOSFET; ITF86110DK8
Component Selection Notes:
C4,5,7,8,9,10,18,19 - Each 150mF, Panasonic EEF-UE0J151R
D1,2 - Each 30mA Schottky Diode, MA732
FIGURE 11. DDR SDRAM VOLTAGE REGULATOR
15
ISL6531
Small Outline Plastic Packages (SOIC)
M24.3 (JEDEC MS-013-AD ISSUE C)
24 LEAD WIDE BODY SMALL OUTLINE PLASTIC PACKAGE
INCHES
MILLIMETERS
N
INDEX
SYMBOL
MIN
MAX
MIN
2.35
0.10
0.33
0.23
MAX
2.65
0.30
0.51
0.32
15.60
7.60
NOTES
0.25(0.010)
M
B M
H
AREA
E
A
A1
B
C
D
E
e
0.0926
0.0040
0.013
0.1043
0.0118
0.020
-
-
-B-
9
1
2
3
L
0.0091
0.5985
0.2914
0.0125
-
0.6141 15.20
3
SEATING PLANE
A
0.2992
7.40
4
-A-
o
h x 45
D
0.05 BSC
1.27 BSC
-
H
h
0.394
0.010
0.016
0.419
0.029
0.050
10.00
0.25
0.40
10.65
0.75
1.27
-
-C-
5
α
µ
e
L
6
A1
C
B
N
α
24
24
7
0.10(0.004)
o
o
o
o
0
8
0
8
-
0.25(0.010) M
C A M B S
Rev. 0 12/93
NOTES:
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of
Publication Number 95.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate burrs.
Mold flash, protrusion and gate burrs shall not exceed 0.15mm
(0.006 inch) per side.
4. Dimension “E” does not include interlead flash or protrusions. Inter-
lead flash and protrusions shall not exceed 0.25mm (0.010 inch) per
side.
5. The chamfer on the body is optional. If it is not present, a visual index
feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater
above the seating plane, shall not exceed a maximum value of
0.61mm (0.024 inch)
10. Controlling dimension: MILLIMETER. Converted inch dimensions
are not necessarily exact.
16
ISL6531
Quad Flat No-Lead Plastic Package (QFN)
Micro Lead Frame Plastic Package (MLFP)
L32.5x5
32 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-220VHHD-2 ISSUE C
MILLIMETERS
SYMBOL
MIN
NOMINAL
MAX
1.00
0.05
1.00
NOTES
A
A1
A2
A3
b
0.80
0.90
-
-
-
-
-
-
9
0.20 REF
9
0.18
2.95
2.95
0.23
0.30
3.25
3.25
5,8
D
5.00 BSC
-
D1
D2
E
4.75 BSC
9
3.10
7,8
5.00 BSC
-
E1
E2
e
4.75 BSC
9
3.10
7,8
0.50 BSC
-
k
0.25
0.30
-
-
-
-
L
0.40
0.50
0.15
8
L1
N
-
32
8
8
-
10
2
Nd
Ne
P
3
8
-
3
0.60
12
9
θ
-
-
9
Rev. 1 10/02
NOTES:
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
2. N is the number of terminals.
3. Nd and Ne refer to the number of terminals on each D and E.
4. All dimensions are in millimeters. Angles are in degrees.
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
8. Nominal dimensionsare provided toassistwith PCBLandPattern
Design efforts, see Intersil Technical Brief TB389.
9. Features and dimensions A2, A3, D1, E1, P & θ are present when
Anvil singulation method is used and not present for saw
singulation.
10. Depending on the method of lead termination at the edge of the
package, a maximum 0.15mm pull back (L1) maybe present. L
minus L1 to be equal to or greater than 0.3mm.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
17
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