LTC1539CGW_技术文档

LTC1538-AUX/ LTC1539

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FEATURES

DESCRIPTION

TheLTC^®1538-AUX/LTC1539aredual,synchronous step-

■

Maintains Constant Frequency at Low Output Currents

Dual N-Channel MOSFET Synchronous Drive

Programmable Fixed Frequency (PLL Lockable)

down switching regulator controllers which drive external

N-channel power MOSFETs in a phase-lockable fixed

frequencyarchitecture.TheAdaptivePower^TMoutputstage

selectively drives two N-channel MOSFETs at frequencies

up to 400kHz while reducing switching losses to maintain

high efficiencies at low output currents.

Wide V Range: 3.5V to 36V Operation

IN

Ultrahigh Efficiency

Very Low Dropout Operation: 99% Duty Cycle

Low Dropout, 0.5A Linear Regulator for VPP

Generation or Low Noise Audio Supply

Built-In Power-On Reset Timer

Programmable Soft Start

Low-Battery Detector

Remote Output Voltage Sense

Foldback Current Limiting (Optional)

Pin Selectable Output Voltage

5V Standby Regulator Active in Shutdown: I_Q< 200µA

Output Voltages from 1.19V to 9V

Available in 28- and 36-Lead SSOP Packages

An auxiliary 0.5A linear regulator using an external PNP

pass device provides a low noise, low dropout voltage

source. A secondary winding feedback control pin (SFB1)

guarantees regulation regardless of load on the main

output by forcing continuous operation.

■

A 5V/20mA regulator, internal 1.19V reference and an

uncommitted comparator remain active when both con-

trollers are shut down. A power-on reset timer (POR) is

included which generates a signal delayed by 65536/f_CLK

(typ 300ms) after the controller’s output is within 5% of

the regulated first voltage. Internal resistive dividers pro-

vide pin selectable output voltages with remote sense

capability on one of the two outputs.

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APPLICATIONS

■

Notebook and Palmtop Computers, PDAs

Portable Instruments

Battery-Operated Devices

DC Power Distribution Systems

■

The operating current levels are user-programmable via

external current sense resistors. Wide input supply range

allows operation from 3.5V to 30V (36V maximum).

■

, LTC and LT are registered trademarks of Linear Technology Corporation.

Adaptive Power is a trademark of Linear Technology Corporation.

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5V STANDBY

TYPICAL APPLICATION

V

IN

5.2V TO 28V

C

IN

+

D

, CMDSH-3

D , CMDSH-3

B2

B1

4.7µF

22µF

35V

× 4

16V

V

PROG1

V

IN

INTV

CC

BOOST 1

TGL1

BOOST 2

TGL2

M1

M4

L2

10µH

TGS1

TGS2

M6*

M3*

L1

10µH

C

0.1µF

C

, 0.1µF

B1

B2

SW2

BG2

SW1

BG1

D2

MBR140T3

M5

D1

LTC1539

M2

MBR140T3

+

SENSE

2

+

1000pF

SENSE

1

R

–

SENSE2

SENSE

R

0.03Ω

SENSE1

0.03Ω

1000pF

V

OSENSE2

–

V

SENSE 1

V

3.3V

3.5A

OUT1

5V

OUT2

I

C

TH1

TH2

C1

C2

3.5A

+

1000pF

C

220µF

10V

RUN/SS1

C

DSC

V

SGND PGND RUN/SS2

OUT1

PROG2

+

C

OUT

R

C

SS1

0.1µF

C

SS2

0.1µF

R

C

C2A

470pF

C1

10k

C1A

OSC

56pF

C2

10k

220µF

10V

220pF

1538 F01

M1, M2, M4, M5: Si4412DY

M3, M6: IRLML2803

*NOT REQUIRED FOR LTC1538-AUX BOLD LINES INDICATE HIGH CURRENT PATHS

Figure 1. High Efficiency Dual 5V/3V Step-Down Converter

1

LTC1538-AUX/ LTC1539

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ABSOLUTE MAXIMUM RATINGS

AUXON, PLLIN, SFB1,

Input Supply Voltage (V )....................... 36V to –0.3V

IN

RUN/SS1, RUN/SS2, LBI, Voltages ......... 10V to –0.3V

Peak Output Current < 10µs (TGL1, 2, BG1, 2)......... 2A

Peak Output Current < 10µs (TGS1, 2) .............. 250mA

INTV_CCOutput Current ........................................ 50mA

Operating Temperature Range

LTC1538-AUXCG/LTC1539CGW............ O°C to 70°C

LTC1538-AUXIG/LTC1539IGW .......... –40°C to 85°C

Junction Temperature (Note 1)............................125°C

Storage Temperature Range ................ –65°C to 150°C

Lead Temperature (Soldering, 10 sec).................300°C

Topside Driver Voltage (BOOST 1, 2) ...... 42V to –0.3V

Peak Switch Voltage > 10µs (SW 1, 2) ... V + 5V to – 5V

IN

EXTV_CCVoltage........................................ 10V to –0.3V

POR1, LBO Voltages ................................ 12V to –0.3V

AUXFB Voltage ........................................ 20V to –0.3V

AUXDR Voltage........................................ 28V to –0.3V

SENSE⁺1, SENSE⁺2, SENSE^–1, SENSE^–2,

V_OSENSE2Voltages ................... INTV + 0.3V to –0.3V

CC

V

_PROG1, V_PROG2Voltages .................... INTV to –0.3V

CC

PLL LPF, I_TH1, I_TH2Voltages ................... 2.7V to –0.3V

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PACKAGE/ORDER INFORMATION

TOP VIEW

ORDER

PART NUMBER

TOP VIEW

1

2

PLL LPF

PLLIN

36

35

34

RUN/SS1

+

SENSE 1

1

2

TGL1

SW1

28

27

26

25

24

23

22

21

20

19

18

17

16

15

BOOST 1

–

3

BOOST 1

SENSE 1

RUN/SS1

+

LTC1538CG-AUX

LTC1538IG-AUX

LTC1539CGW

LTC1539IGW

4

33 TGL1

32 SW1

V

PROG1

3

V

IN

SENSE

1

–

5

I

TH1

4

BG1

SENSE

V

6

TGS1

31

30

29

28

27

26

25

POR1

5

INTV /5V

CC

PROG1

7

V

IN

C

OSC

6

PGND

BG2

I

TH1

8

BG1

SGND

LBI

7

C

OSC

9

INTV /5V

CC

8

EXTV

CC

SGND

SFB1

10

11

12

13

14

15

16

17

18

PGND

BG2

LBO

SFB1

9

SW2

10

11

12

13

14

TGL2

I

TH2

EXTV

CC

I

TH2

BOOST 2

AUXON

AUXFB

AUXDR

V

OSENSE2

–

24 TGS2

23 SW2

V

PROG2

SENSE

2

+

V

OSENSE2

SENSE

–

TGL2

22

21

20

19

SENSE 2

RUN/SS2

+

BOOST 2

AUXON

AUXFB

SENSE 2

G PACKAGE

28-LEAD PLASTIC SSOP

RUN/SS2

AUXDR

T_JMAX= 125°C, θ_JA= 95°C/ W

GW PACKAGE

36-LEAD PLASTIC SSOP

_JMAX= 125°C, θ_JA= 85°C/ W

T

Consult factory for Military grade parts.

2

LTC1538-AUX/ LTC1539

ELECTRICAL CHARACTERISTICS ^T_A^{= 25°C, V = 15V, V}_RUN/SS1,2= 5V unless otherwise noted.

IN

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

Main Control Loops

I

V

Feedback Current

V

Pins Open (Note 2)

10

50

nA

IN OSENSE2

PROG1, PROG2

V

OUT1,2

Regulated Output Voltage

1.19V (Adjustable) Selected

3.3V Selected

(Note 2)

V

Pins Open

= 0V

●

1.178

3.220

4.900

1.19

3.30

5.00

1.202

3.380

5.100

V

PROG1, PROG2

V

PROG1, PROG2

5V Selected

V

= INTV

PROG1, PROG2 CC

V

Reference Voltage Line Regulation

Output Voltage Load Regulation

V = 3.6V to 20V (Note 2), V Pins Open

PROG1,2

0.002

0.01

%/V

LINEREG1,2

IN

V

I

Sinking 5µA (Note 2)

Sourcing 5µA

●

0.5

–0.5

0.8

–0.8

%

LOADREG1,2

TH1,2

I

TH1,2

V

Secondary Feedback Threshold

Secondary Feedback Current

Output Overvoltage Lockout

V

Ramping Negative

= 1.5V

●

1.16

1.24

1.19

–1

1.22

–2

V

µA

V

SFB1

I

V

SFB1

–

V

OVL

V

Pin Open, SENSE 1 and V Pins

OSENSE2

1.28

1.32

PROG1,2

I

V

Input Current

0.5V > V

PROG1,2

–3

3

–6

6

µA

PROG1,2

INTV – 0.5V < V

< INTV

CC

PROG1,2

I

Q

Input DC Supply Current

Normal Mode

EXTV = 5V (Note 3)

CC

3.6V < V < 30V, V

= 0V

320

70

µA

IN

AUXON

Shutdown

V

= 0V, 3.6V < V < 15V

200

2

RUN/SS1,2

IN

V

Run Pin Threshold

●

0.8

1.5

1.3

3

V

µA

RUN/SS1,2

I

Soft Start Current Source

Maximum Current Sense Threshold

V

= 0V

4.5

180

RUN/SS1,2

∆V

V

= 0V, 5V V = Pins Open

PROG1,2

130

150

mV

SENSE(MAX)

OSENSE1,2

TGL1, 2 t , t

TGL1, TGL2 Transition Time

Rise Time

r

f

C

LOAD

= 3000pF

50

150

ns

LOAD

Fall Time

TGS1, 2 t , t

TGS1, TGS2 Transition Time

Rise Time

r

f

C

LOAD

= 500pF

100

50

150

ns

LOAD

Fall Time

BG1, 2 t , t

BG1, BG2 Transition Time

Rise Time

r

f

C

LOAD

= 3000pF

50

150

ns

LOAD

Fall Time

Internal V Regulator –5V Standby

CC

V

Internal V Voltage

6V < V < 30V, V = 4V

EXTVCC

●

4.8

4.5

5.0

–0.2

170

4.7

5.2

–1

V

%

INTVCC

CC

IN

V

LDO

INT

INTV Load Regulation

INTV = 20mA, V

EXTVCC

= 4V

= 5V

CC

V

LDO

EXT

EXTV Voltage Drop

INTV = 20mA, V

EXTVCC

300

mV

V

CC

V

EXTVCC

EXTV Switchover Voltage

INTV = 20mA, EXTV Ramping Positive

CC CC

CC

Oscillator and Phase-Locked Loop

f

Oscillator Frequency

VCO High

C

= 100pF, LTC1539: PLL LPF = 0V (Note 4)

112

200

125

240

138

kHz

OSC

LTC1539, V

= 2.4V

PLLLPF

R

PLLIN

PLLIN Input Resistance

50

kΩ

I

Phase Detector Output Current

Sinking Capability

LTC1539

< f

PLLLPF

f

10

15

20

µA

PLLIN OSC

Sourcing Capability

f

> f

PLLIN OSC

Power-On Reset

V

POR1 Saturation Voltage

I

V

= 1.6mA, V

= 1V,

0.6

0.2

1

V

SATPOR1

POR1

OSENSE1

Pins Open

PROG1

I

POR1 Leakage

V

= 12V, V

= 1.19V, V Pin Open

PROG1

µA

LPOR1

POR1

OSENSE1

V

THPOR1

POR1 Trip Voltage

V

Pin Open % of V

PROG1 REF

V

Ramping Negative

Pin Open

PROG1

–11

–7.5

–4

%

OSENSE1

t

POR1 Delay

V

65536

Cycles

DPOR1

3

LTC1538-AUX/ LTC1539

ELECTRICAL CHARACTERISTICS ^T_A^{= 25°C, V = 15V, V}_RUN/SS1,2= 5V unless otherwise noted.

IN

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

Low-Battery Comparator

V

LBO Saturation Voltage

LBO Leakage

I

= 1.6mA, V = 1.1V

0.6

0.01

1.19

1

V

SATLBO

LBO

LBI

I

V

= 12V, V = 1.4V

●

µA

V

LLBO

LBO

LBI

V

THLB1

LBI Trip Voltage

LBI Input Current

LBO Hysteresis

High to Low Transition on LBO

= 1.19V

1.16

1.22

50

I

V

nA

mV

INLB1

LBI

V

HYSLBO

20

Auxiliary Regulator/Comparator

I

AUXDR Current

V

EXTVCC

= 0V

AUXDR

Max Current Sinking Capability

Control Current

Leakage when OFF

V

= 4V, V

= 1.0V, V

AUXON

= 1.5V, V

AUXON

= 5V

10

15

1

0.01

mA

µA

AUXDR

AUXFB

V

= 5V, V

5

1

AUXDR

AUXFB

V

AUXDR

= 24V, V

= 1.5V, V = 0V

AUXON

AUXFB

I

AUXFB Input Current

AUXON Input Current

AUXON Trip Voltage

AUXDR Saturation Voltage

AUXFB Voltage

V

= 1.19V, V = 5V

AUXON

0.01

1.19

0.4

1

µA

V

INAUXFB

AUXFB

I

V

AUXON

= 5V

1

INAUXON

V

AUXDR

= 4V, V = 1V

AUXFB

1.0

1.4

0.8

THAUXON

V

I

= 1.6mA, V

= 1V, V = 5V

AUXON

V

SATAUXDR

AUXDR

AUXFB

V

= 5V, 11V < V < 24V (Note 5)

AUXDR

●

11.5

1.14

12.00

1.19

12.5

1.24

V

AUXFB

AUXON

V

AUXON

= 5V, 3V < V

< 7V

AUXDR

V

AUXFB Divider Disconnect Voltage

V

AUXON

= 5V (Note 5); Ramping Negative

7.5

8.5

9.5

V

THAUXDR

The

● denotes specifications which apply over the full operating

Note 3: Dynamic supply current is higher due to the gate charge being

temperature range.

delivered at the switching frequency. See Applications Information.

Note 1: T is calculated from the ambient temperature T and power

Note 4: Oscillator frequency is tested by measuring the C charge and

J

A

OSC

dissipation P according to the following formulas:

discharge current (I ) and applying the formula:

D

OSC

8

-1

–1

f

(kHz) = 8.4(10 )[C (pF) + 11] (1/I

+ 1/I

)

LTC1538CG-AUX: T = T + (P )(95°C/W)

OSC

CHG

DISC

J

A

D

LTC1539CGW: T = T + (P )(85°C/W)

Note 5: The auxiliary regulator is tested in a feedback loop which servos

to the balance point for the error amplifier. For applications with

J

A

D

V

AUXFB

Note 2: The LTC1538-AUX and LTC1539 are tested in a feedback loop

which servos V to the balance point for the error amplifier

V

AUXDR

> 9.5V, V

uses an internal resistive divider. See Applications

AUXFB

OSENSE1,2

Information section.

(V

= 1.19V).

ITH1,2

4

LTC1538-AUX/ LTC1539

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TYPICAL PERFORMANCE CHARACTERISTICS

Efficiency vs Input Voltage:

V_OUT= 3.3V

Efficiency vs Input Voltage:

_OUT= 5V

V

Efficiency vs Load Current

100

95

90

85

80

75

70

100

95

90

85

80

75

70

V

IN

= 10V

V

OUT

= 3.3V

V

OUT

= 5V

95

90

85

80

75

70

65

60

55

50

V

OUT

= 5V

R

SENSE

= 0.33Ω

I

= 1A

LOAD

I

= 1A

LOAD

CONTINUOUS

MODE

I

= 100mA

LOAD

Burst Mode^TM

OPERATION

I

= 100mA

LOAD

Adaptive Power^TM

MODE

0.001

0.01

0.1

1

10

0

10

15

20

25

30

0

10

15

20

25

30

5

LOAD CURRENT (A)

INPUT VOLTAGE (V)

1538/39 • G03

1538/39 • G02

1538/39 • G01

V – V_OUTDropout Voltage vs

Load Current

IN

Load Regulation

V_ITHPin Voltage vs Output Current

0

0.5

0.4

0.3

0.2

0.1

3.0

2.5

R

V

OUT

= 0.033Ω

DROP OF 5%

R

= 0.033Ω

SENSE

–0.25

–0.50

–0.75

–1.00

–1.25

–1.50

2.0

1.5

1.0

0.5

0

Burst Mode

OPERATION

CONTINUOUS/

Adaptive Power

MODE

0

0.5

1.0

1.5

2.0

2.5

3.0

0

1.0

1.5

2.0

2.5

3.0

0.5

0

10 20 30 40 50 60 70 80 90 100

OUTPUT CURRENT (%)

LOAD CURRENT (A)

1538/39 • G04

1538/39 • G05

1538/39 • G06

Input Supply Current

vs Input Voltage

INTV_CCRegulation

vs INTV_CCLoad Current

EXTV_CCSwitch Drop

vs INTV_CCLoad Current

300

200

100

0

2.5

2.0

1.5

1.0

100

80

2

EXTV = 0V

CC

70°C

25°C

1

0

5V, 3.3V OFF

5V STANDBY

70°C

25°C

60

–45°C

5V, 3.3V ON

40

5V OFF, 3.3V ON

–1

–2

0.5

0

20

0

5V ON, 3.3V OFF

0

5

10

15

20

25

30

0

10

15

20

25

30

20

5

0

30

40

50

10

INPUT VOLTAGE (V)

INTV LOAD CURRENT (mA)

CC

1538/39 • G09

LTC1538/39 • TPC07

1538/39 • G08

Adaptive Power and Burst Mode are trademarks of Linear Technology Corporation.

5

LTC1538-AUX/ LTC1539

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TYPICAL PERFORMANCE CHARACTERISTICS

Normalized Oscillator Frequency

vs Temperature

RUN/SS Pin Current vs

Temperature

SFB1 Pin Current vs

Temperature

10

5

0

–0.25

–1.50

–0.75

4

3

2

1

f

O

–1.00

–1.25

–1.50

–5

0

–10

60

TEMPERATURE (°C)

110 135

–40 –15 10

35

60

85 110 135

60

TEMPERATURE (°C)

110 135

–40 –15

10

35

85

–40 –15

10

35

85

TEMPERATURE (°C)

1538/39 • G11

1538/39 • G10

1538/39 • G12

Maximum Current Comparator

Threshold Voltage vs Temp

Transient Response

154

152

150

148

V

V_OUT

50mV/DIV

OUT

50mV/DIV

1538/39 • G14

1538/39 • G15

I_LOAD= 50mA to 1A

I_LOAD= 1A to 3A

146

–40 –15 10

35

60

85 110 135

TEMPERATURE (°C)

1538/39 • G13

Auxiliary Regulator Load

Regulation

Burst Mode Operation

Soft Start: Load Current vs Time

12.2

12.1

12.0

EXTERNAL PNP: 2N2907A

V_OUT

20mV/DIV

RUN/SS

5V/DIV

INDUCTOR

CURRENT

1A/DIV

V_OUT

200mV/DIV

11.9

11.8

11.7

1538/39 • G16

1538/39 • G17

I_LOAD= 50mA

0

40

80

120

160

200

AUXILIARY LOAD CURRENT (mA)

1538/39 • G18

6

LTC1538-AUX/ LTC1539

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TYPICAL PERFORMANCE CHARACTERISTICS

Auxiliary Regulator Sink

Current Available

Auxiliary Regulator PSRR

100

90

80

70

60

50

40

30

20

10

0

20

I

= 10mA

L

15

10

5

I

= 100mA

L

0

2

4

6

8

10 12 14 16

10

100

1000

AUX DR VOLTAGE (V)

FREQUENCY (kHz)

1538/39 • G20

1538/39 • G19

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PIN FUNCTIONS

V : Main Supply Pin. Must be closely decoupled to the

IC’s signal ground pin.

SGND:SmallSignalGround.Commontobothcontrollers,

must be routed separately from high current grounds to

the (–) terminals of the C_OUTcapacitors.

IN

INTV /5V STANDBY: Output of the Internal 5V Regulator

CC

and the EXTV_CCSwitch. The driver and control circuits are

powered from this voltage. Must be closely decoupled to

power ground with a minimum of 2.2µF tantalum or

electrolytic capacitor. The INTV_CCregulator remains on

when both RUN/SS1 and RUN/SS2 are low. Refer to the

LTC1438/LTC1439 for applications which do not require a

5V standby regulator.

PGND: Driver Power Ground. Connects to sources of

bottom N-channel MOSFETs and the (–) terminals of C .

IN

SENSE^–1, SENSE^–2: Connects to the (–) input for the

current comparators. SENSE^–1 is internally connected to

the first controllers V_OUTsensing point preventing true

remote output voltage sensing operation. The first con-

troller can only be used as a 3.3V or 5.0V regulator

controlledbytheV_PROG1pin. Thesecondcontrollercanbe

set to a 3.3V, 5.0V or an adjustable regulator controlled by

the V_PROG2pin (see Table 1).

EXTV : External Power Input to an Internal Switch. This

CC

switch closes and supplies INTV_CC,bypassing the internal

low dropout regulator whenever EXTV is higher than 4.8V.

CC

Connect this pin to V_OUTof the controller with the higher

Table 1. Output Voltage Table

output voltage. Do not exceed 10V on this pin. See EXTV

CC

LTC1538-AUX

LTC1539

connection in Applications Information section.

CONTROLLER 1

CONTROLLER 2

5V or 3.3V Only, Secondary Feedback Loop

Adjustable Only

Remote Sensing

5V/3.3V/Adjustable

Remote Sensing

POR1 Output

BOOST1, BOOST2:Supplies totheTopsideFloatingDrivers.

The bootstrap capacitors are returned to these pins. Voltage

swing at these pins is from INTV to V + INTV .

CC

IN

CC

SW1, SW2: Switch Node Connections to Inductors. Volt-

age swing at these pins is from a Schottky diode (external)

voltage drop below ground to V .

IN

7

LTC1538-AUX/ LTC1539

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PIN FUNCTIONS

SENSE⁺1, SENSE⁺2: The (+) Input to Each Current

Comparator. Built-in offsets between SENSE^–1 and

SENSE⁺1 pins in conjunction with R_SENSE1set the current

trip threshold (same for second controller).

BG1, BG2: High Current Gate Drive Outputs for Bottom N-

Channel MOSFETs. Voltage swing at these pins is from

ground to INTV .

CC

SFB1: SecondaryWindingFeedbackInput. This inputacts

only on the first controller and is normally connected to a

feedback resistive divider from the secondary winding.

Pulling this pin below 1.19V will force continuous syn-

chronous operation forthe first controller. This pinshould

V

_OSENSE2:Receives theremotelysensedfeedbackvoltagefor

the second controller either from the output directly or from

an external resistive divider across the output . The V

PROG2

pin determines which point. V_OSENSE2must connect to.

be tied to: ground to force continuous operation; INTV

CC

V

_PROG1, V_PROG2: Programs Internal Voltage Attenuators

in applications that don’t use a secondary winding; and a

resistive divider from the output in applications using a

secondary winding.

for Output Voltage Sensing. The voltage sensing for the

first controller is internally connected to SENSE^–1 while

the V_OSENSE2pin allows for remote sensing for the second

controller. For V_PROG1, V_PROG2< V_INTVCC/3, the divider is

POR1: This output is a drain of an N-channel pull-down.

This pin sinks current when the output voltage of the first

controller drops 7.5% below its regulated voltage and re-

leases 65536 oscillator cycles after the output voltage of the

firstcontrollerrisestowithin–5%valueofitsregulatedvalue.

The POR1 output is asserted when RUN/SS1 and RUN/SS2

set for an output voltage of 3.3V. With V_PROG1

,

V

> V_INTVCC/1.5 the divider is set for an output

PROG2

voltageof5V. LeavingV_PROG2open(DC)allows theoutput

voltage of the second controller to be set by an external

resistive divider connected to V_OSENSE2

.

are both low, independent of the V

.

OUT1

C_OSC: External capacitor C_OSCfrom this pin to ground sets

the operating frequency.

LBO:This outputis adrainofanN-channelpull-down.This

pin will sink current when the LBI pin goes below 1.19V

irrespective of the RUN/SS pin voltage.

I_TH1, I_TH2: Error Amplifier Compensation Point. Each as-

sociated current comparator threshold increases with this

control voltage.

LBI: The (+) input of a comparator which can be used as

a low-battery voltage detector irrespective of the RUN/SS

pin voltage. The (–) input is connected to the 1.19V

internal reference.

RUN/SS1, RUN/SS2: Combination of Soft Start and RUN

Control Inputs. A capacitor to ground at each of these pins

sets the ramp time to full current output. The time is

approximately 0.5s/µF. Forcing either of these pins below

1.3V causes the IC to shut down the circuitry required for

that particular controller. Forcing both of these pins below

1.3V causes the device to shut down both controllers,

leaving the 5V standby regulator, internal reference and a

comparator active. Refer to the LTC1438/LTC1439 for appli-

cations which do not require a 5V standby regulator.

PLLIN: External Synchronizing Input to Phase Detector.

This pin is internally terminated to SGND with 50kΩ. Tie

this pin to SGND in applications which do not use the

phase-locked loop.

PLL LPF: Output of Phase Detector and Control Input of

Oscillator. Normally a series RC lowpass filter network is

connected from this pin to ground. Tie this pin to SGND in

applications which do not use the phase-locked loop. Can

be driven by a 0V to 2.4V logic signal for a frequency

shifting option.

TGL1, TGL2: High Current Gate Drives for Main Top

N-Channel MOSFET. These are the outputs of floating

drivers with a voltage swing equal to INTV_CCsuperim-

posed on the switch node voltage SW1 and SW2.

AUXFB: Feedback Input to the Auxiliary Regulator/Com-

TGS1, TGS2: Gate Drives for Small Top N-Channel parator. When used as a linear regulator, this input can

MOSFET. These are the outputs of floating drivers with a

voltage swing equal to INTV_CCsuperimposed on the

switch node voltage SW. Leaving TGS1 or TGS2 open

invokes Burst Mode operation for that controller.

either be connected to an external resistive divider or

directly to the collector of the external PNP pass device for

12V operation. When used as a comparator, this is the

8

LTC1538-AUX/ LTC1539

U

PIN FUNCTIONS

noninverting input of a comparator whose inverting input

is tied to the internal 1.19V reference. See Auxiliary Regu-

lator Application section.

AUXDR: Open Drain Output of the Auxiliary Regulator/

Comparator. The base of an external PNP device is con-

nected to this pin when used as a linear regulator. An

external pull-up resistor is required for use as a compara-

tor. A voltage > 9.5V on AUXDR causes the internal 12V

resistivedividertobeconnectedinseries withtheAUXFBpin.

AUXON: Pulling this pin high turns on the auxiliary regu-

lator/comparator. The threshold is 1.19V. This is a conve-

nient linear power supply logic-controlled on/off input.

U

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FUNCTIONAL DIAGRA

LTC1539 shown, see specific package pinout for availability of specific functions.

V

IN

INTV

CC

2.4V

PLLIN**

BOOST

TGL

DUPLICATE FOR SECOND CONTROLLER CHANNEL

PHASE

DETECTOR

f

IN

D

B

50k

PLL LPF**

R

LP

DROPOUT

DETECTOR

C

OSC

C

LP

SFB

C

B

OSCILLATOR

S

Q

C

OSC

R

POR1**

TGS**

SW

V

FB1

POWER-ON

RESET

SWITCH

LOGIC

0.6V

+

–

1.11V

SHUTDOWN

LBI**

BATTERY

SENSE

•

INTV

CC

–

LBO**

–

BG

I1

•

+

C

PGND

IN

+

–

+

C

OUT

AUXON

INTV

CC

R

V

SEC

SENSE

–

9V

–

I2

–

8k

+

30k

–

AUXDR

AUXFB

SENSE

V

LDO

+

SENSE

4k

320k

61k

180k

90.8k

10k

V

*

OSENSE

V

FB

+

V

OUT

+

–

EA

Ω

+

g

m

= 1m

C

SEC

V

PROG

*

1.19V

REF

+

SFB

SFB1*

119k

V

REF

–

+

–

1µA

V

IN

OV

†

V

IN

D

FB

4.8V

+

–

1.28V

1.19V

5V LDO

REGULATOR

C

3µA

I

TH

EXTV

CC

SHUTDOWN

R

C

RUN

SOFT START

INTV

CC

6V

RUN/SS

+

SGND

C

SS

INTERNAL

SUPPLY

^†FOLDBACK CURRENT LIMITING OPTION

BOLD LINES INDICATE HIGH CURRENT PATHS

1438 FD

*NOT AVAILABLE ON BOTH CHANNELS

**NOT AVAILABLE ON LTC1538-AUX

9

LTC1538-AUX/ LTC1539

U

(Refer to Functional Diagram)

OPERATION

Main Control Loop

Comparator OV guards against transient overshoots

> 7.5% by turning off the top MOSFET and keeping it off

until the fault is removed.

The LTC1538-AUX/LTC1539 use a constant frequency,

current mode step-down architecture. During normal op-

eration, the top MOSFET is turned on each cycle when the

oscillator sets the RS latch and turned off when the main

current comparator I1 resets the RS latch. The peak

inductor current at which I1 resets the RS latch is con-

trolled by the voltage on the I_TH1(I_TH2) pin, which is the

Low Current Operation

Adaptive Power Mode allows the LTC1539 to automati-

cally change between two output stages sized for different

load currents. The TGL1 (TGL2) and BG1 (BG2) pins drive

large synchronous N-channel MOSFETs for operation at

high currents, while the TGS1 (TGS2) pin drives a much

smaller N-channel MOSFET used in conjunction with a

Schottky diode for operation at low currents. This allows

the loop to continue to operate at normal operating fre-

quencyas theloadcurrentdecreases withoutincurringthe

large MOSFET gate charge losses. If the TGS1 (TGS2) pin

is left open, the loop defaults to Burst Mode operation in

which the large MOSFETs operate intermittently based on

load demand. Adaptive Power mode provides constant

frequency operation down to approximately 1% of rated

load current. This results in an order of magnitude reduc-

tion of load current before Burst Mode operation com-

mences. Without the small MOSFET (ie: no Adaptive

Power mode) the transition to Burst Mode operation is

approximately 10% of rated load current. The transition to

low current operation begins when comparator I2 detects

current reversal and turns off the bottom MOSFET. If the

voltage across R_SENSEdoes not exceed the hysteresis of

I2(approximately20mV)foronefullcycle, thenonfollow-

ing cycles the top drive is routed to the small MOSFET at

the TGS1 (TGS2) pin and the BG1 (BG2) pin is disabled.

This continues until an inductor current peak exceeds

20mV/R_SENSEor the I_TH1(I_TH2) voltage exceeds 0.6V,

either of which causes drive to be returned to the TGL1

(TGL2) pin on the next cycle.

output of each error amplifier (EA). The V

pin,

PROG1

described in the Pin Functions, allows the EA to receive a

selectively attenuated output feedback voltage V_FB1from

the SENSE^–1 pin while V_PROG2and V_OSENSE2allow EA to

receive an output feedback voltage V_FB2from either inter-

nal or external resistive dividers on the second controller.

When the load current increases, it causes a slight de-

crease in V relative to the 1.19V reference, which in turn

FB

causes the I_TH1(I_TH2) voltage to increase until the average

inductor current matches the new load current. After the

large top MOSFET has turned off, the bottom MOSFET is

turnedonuntileithertheinductorcurrentstarts toreverse,

as indicated by current comparator I2, or the beginning of

the next cycle.

The top MOSFET drivers are biased from floating boot

strap capacitor C_B, which normally is recharged during

each Off cycle. When V decreases to a voltage close to

V_OUT, however, the loop may enter dropout and attempt to

turn on the top MOSFET continuously. The dropout detec-

tor counts the number of oscillator cycles that the top

MOSFET remains on and periodically forces a brief off

period to allow C_Bto recharge.

IN

The main control loop is shut down by pulling the RUN/

SS1 (RUN/SS2) pin low. Releasing RUN/SS1 (RUN/SS2)

allows an internal 3µA current source to charge soft start

capacitor C_SS. When C_SSreaches 1.3V, the main control

loop is enabled with the I_TH1(I_TH2) voltage clamped at

approximately 30% of its maximum value. As C_SScontin-

ues to charge, I_TH1(I_TH2) is gradually released allowing

normal operation to resume. When both RUN/SS1 and

RUN/SS2 are low, all LTC1538-AUX/LTC1539 functions

areshutdownexceptforthe5Vstandbyregulator, internal

referenceandacomparator.RefertotheLTC1438/LTC1439

for applications which do not require a 5V standby regulator.

Twoconditions canforcecontinuous synchronous opera-

tion, even when the load current would otherwise dictate

low current operation. One is when the common mode

–

voltage of the SENSE⁺1 (SENSE⁺2) and SENSE 1

(SENSE^–2) pins are below 1.4V, and the other is when the

SFB1 pin is below 1.19V. The latter condition is used to

assistinsecondarywindingregulation,as describedinthe

Applications Information section.

10

LTC1538-AUX/ LTC1539

U

(Refer to Functional Diagram)

OPERATION

Frequency Synchronization

The AUX block can be used as a comparator having its

inverting input tied to the internal 1.19V reference. The

AUXDR pin is used as the output and requires an external

pull-up to a supply of less than 8.5V in order to inhibit the

invoking of the internal resistive divider.

A Phase-Locked Loop (PLL) is available on the LTC1539

to allow the oscillator to be synchronized to an external

source connected to the PLLIN pin. The output of the

phase detector at the PLL LPF pin is also the control input

of the oscillator, which operates over a 0V to 2.4V range

corresponding to –30% to 30% in frequency. When

locked, the PLL aligns the turn-on of the top MOSFET to

the rising edge of the synchronizing signal. When PLLIN

is left open, PLL LPF goes low, forcing the oscillator to

minimum frequency.

INTV_CC/EXTV Power

CC

Power for the top and bottom MOSFET drivers and most

of the other LTC1538-AUX/LTC1539 circuitry is derived

from the INTV pin. The bottom MOSFET driver supply is

CC

also connected to INTV_CC. When the EXTV_CCpin is left

open,aninternal5Vlowdropoutregulatorsupplies INTV

CC

Power-On Reset

power. If EXTV_CCis taken above 4.8V, the 5V regulator is

turned off and an internal switch is turned on to connect

EXTV_CCto INTV_CC. This allows the INTV_CCpower to be

derived from a high efficiency external source such as the

output of the regulator itself or a secondary winding, as

described in the Applications Information section.

The POR1 pin is an open drain output which pulls low

when the main regulator output voltage of the LTC1539

first controller is out of regulation. When the output

voltage rises to within 5% of regulation, a timer is started

which releases POR1 after 2¹⁶(65536) oscillator cycles.

The 5V/20mA INTV regulator can be used as a standby

CC

Auxiliary Linear Regulator

regulator when the two controllers are in shutdown or

when either or both controllers are on. Irrespective of the

signals on the RUN/SS pins, the INTV_CCpin will follow the

voltageappliedtotheEXTV_CCpinwhenthevoltageapplied

to the EXTV_CCpin is taken above 4.8V. The externally

applied voltage is required to be less than the voltage

The auxiliary linear regulator in the LTC1538-AUX and

LTC1539controls anexternalPNPtransistorforoperation

up to 500mA. A precise internal AUXFB resistive divider is

invoked when the AUXDR pin is above 9.5V to allow

regulated 12V VPP supplies to be easily implemented.

When AUXDR is below 8.5V an external feedback divider

may be used to set other output voltages. Taking the

AUXON pin low shuts down the auxiliary regulator provid-

ing a convenient logic-controlled power supply.

applied to the V pin at all times, even when both control-

IN

lers are shut down. This prevents a voltage backfeed

situation from the source applied to the EXTV_CCpin to the

V pin. If the EXTV pin is tied to the first controller’s 5V

IN

CC

output, the nominal INTV pin voltage will stay in the

CC

guaranteed range of 4.7V to 5.2V.

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APPLICATIONS INFORMATION

R

_SENSESelection for Output Current

The basic LTC1539 application circuit is shown in Fig-

ure 1. External component selection is driven by the load

requirementandbegins withtheselectionofR_SENSE.Once

R_SENSEis known, C_OSCand L can be chosen. Next, the

R_SENSEis chosen based on the required output current.

The LTC1538-AUX/LTC1539 current comparator has a

maximum threshold of 150mV/R_SENSEand an input com-

mon mode range of SGND to INTV_CC. The current com-

parator threshold sets the peak of the inductor current,

yielding a maximum average output current I_MAXequal to

the peak value less half the peak-to-peak ripple current, ∆I_L.

powerMOSFETs andD1areselected.Finally,C andC_OUT

IN

are selected. The circuit shown in Figure 1 can be config-

ured for operation up to an input voltage of 28V (limited by

the external MOSFETs).

11

LTC1538-AUX/ LTC1539

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APPLICATIONS INFORMATION

Allowingsomemarginforvariations intheLTC1538-AUX/

LTC1539 and external component values yield:

synchronizable applications, choose C_OSCcorresponding

to a frequency approximately 30% below your center

frequency. (See Phase-Locked Loop and Frequency

Sychronization).

100mV

R

=

SENSE

I

MAX

Inductor Value Calculation

The LTC1538-AUX/LTC1539 work well with values of

_SENSEfrom 0.005Ω to 0.2Ω.

R

The operating frequency and inductor selection are inter-

related in that higher operating frequencies allow the use

of smaller inductor and capacitor values. So why would

anyone ever choose to operate at lower frequencies with

larger components? The answer is efficiency. A higher

frequency generally results in lower efficiency because of

MOSFET gate charge losses. In addition to this basic trade

off, the effect of inductor value on ripple current and low

current operation must also be considered.

C_OSCSelection for Operating Frequency

The LTC1538-AUX/LTC1539 use a constant frequency

architecture with the frequency determined by an external

oscillatorcapacitoronC_OSC.EachtimethetopsideMOSFET

turns on,thevoltageonC_OSCis resettoground.Duringthe

on-time, C_OSCis charged by a fixed current plus an

additional current which is proportional to the output

voltage of the phase detector (V_PLLLPF)(LTC1539 only).

When the voltage on the capacitor reaches 1.19V, C_OSCis

reset to ground. The process then repeats.

Theinductorvaluehas adirecteffectonripplecurrent.The

inductor ripple current ∆I_Ldecreases with higher induc-

tanceorfrequencyandgenerallyincreases withhigherV

IN

or V_OUT

:

The value of C_OSCis calculated from the desired operating

frequency. Assuming the phase-locked loop has no exter-

nal oscillator input (V_PLLLPF= 0V):

1

V

OUT

V

IN

∆I =

V

1–

L

OUT

(f)(L)

4

1.37 10

(

)

Accepting larger values of ∆I_Lallows the use of low

inductances, but results in higher output voltage ripple

and greater core losses. A reasonable starting point for

setting ripple current is ∆I_L= 0.4(I_MAX). Remember, the

maximum ∆I_Loccurs at the maximum input voltage.

C

(pF) =

– 11

OSC

Frequency kHz

(

)

A graph for selecting C_OSCvs frequency is given in Figure

2. As the operating frequency is increased the gate charge

losses will be higher, reducing efficiency (see Efficiency

Considerations). The maximum recommended switching

frequency is 400kHz. When using Figure 2 for

The inductor value also has an effect on low current

operation. The transition to low current operation begins

60

300

V

= 5.0V

= 3.3V

= 2.5V

OUT

V

= 0V

PLLLPF

V

OUT

50

40

30

20

10

0

250

200

150

100

50

V

OUT

0

100

200

300

400

500

0

100

150

200

250

300

50

OPERATING FREQUENCY (kHz)

LTC1538 • F02

1538 F03

Figure 2. Timing Capacitor Value

Figure 3. Recommended Inductor Values

12

LTC1538-AUX/ LTC1539

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APPLICATIONS INFORMATION

when the inductor current reaches zero while the bottom

MOSFET is on. Lower inductor values (higher∆I_L) will cause

this to occur at higher load currents, which can cause a dip

in efficiency in the upper range of low current operation. In

BurstModeoperation(TGS1, 2pins open), lowerinductance

values will cause the burst frequency to decrease.

TotakeadvantageoftheAdaptivePoweroutputstage, two

topside MOSFETs must be selected. A large [low R_SD(ON)

]

MOSFET and a small [higher R_DS(ON)] MOSFET are re-

quired. The large MOSFET is used as the main switch and

works in conjunction with the synchronous switch. The

smaller MOSFET is only enabled under low load current

conditions.Thebenefitofthis is toboostlowtomidcurrent

efficiencies while continuing to operate at constant fre-

quency. Also, by using the small MOSFET the circuit will

keep switching at a constant frequency down to lower

currents and delay skipping cycles.

The Figure 3 graph gives a range of recommended induc-

tor values vs operating frequency and V_OUT

.

Inductor Core Selection

Once the value for L is known, the type of inductor must be

selected. High efficiency converters generally cannot af-

ford the core loss found in low cost powdered iron cores,

forcing the use of more expensive ferrite, molypermalloy

or Kool Mµ^®cores. Actual core loss is independent of core

size for a fixed inductor value, but it is very dependent on

inductance selected. As inductance increases, core losses

godown.Unfortunately,increasedinductancerequires more

turns of wire and therefore copper losses will increase.

The R_DS(ON)recommended for the small MOSFET is

around 0.5Ω. Be careful not to use a MOSFET with an

R_DS(ON)that is too low; remember, we want to conserve

gatecharge. (AhigherR_DS(ON)MOSFEThas asmallergate

capacitance and thus requires less current to charge its

gate). For all LTC1538-AUX and cost sensitive LTC1539

applications,thesmallMOSFETis notrequired.Thecircuit

then begins Burst Mode operation as the load current

drops.

Ferrite designs have very low core loss and are preferred

at high switching frequencies, so design goals can con-

centrate on copper loss and preventing saturation. Ferrite

core material saturates “hard,” which means that induc-

tance collapses abruptly when the peak design current is

exceeded. This results in an abrupt increase in inductor

ripple current and consequent output voltage ripple. Do

not allow the core to saturate!

The peak-to-peak drive levels are set by the INTV_CCvolt-

age. This voltage is typically 5V during start-up (see

EXTV_CCPinConnection).Consequently,logiclevelthresh-

old MOSFETs must be used in most LTC1538-AUX/

LTC1539 applications. The only exception is applications

in which EXTV_CCis powered from an external supply

greaterthan8V(mustbeless than10V), inwhichstandard

thresholdMOSFETs (V_GS(TH)<4V)maybeused.Payclose

attention to the BV_DSSspecification for the MOSFETs as well;

many of the logic level MOSFETs are limited to 30V or less.

Molypermalloy (from Magnetics, Inc.) is a very good, low

loss corematerialfortoroids,butitis moreexpensivethan

ferrite. A reasonable compromise from the same manu-

facturer is Kool Mµ. Toroids are very space efficient,

especially when you can use several layers of wire. Be-

cause they generally lack a bobbin, mounting is more

difficult. However, designs forsurfacemountareavailable

which do not increase the height significantly.

Selection criteria for the power MOSFETs include the "ON"

resistance R_SD(ON), reverse transfer capacitance C_RSS

,

input voltage and maximum output current. When the

LTC1538-AUX/LTC1539areoperatingincontinuous mode

the duty cycles for the top and bottom MOSFETs are given

by:

Power MOSFET and D1 Selection

V

OUT

Main Switch Duty Cycle =

Three external power MOSFETs must be selected for each

controllerwiththeLTC1539:apairofN-channelMOSFETs

for the top (main) switch and an N-channel MOSFET for

the bottom (synchronous) switch. Only one top MOSFET

is required for each LTC1538-AUX controller.

V

IN

V – V

(

)

IN

OUT

Synchronous Switch Duty Cycle =

V

IN

Kool Mµ is a registered trademark of Magnetics, Inc.

13

LTC1538-AUX/ LTC1539

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APPLICATIONS INFORMATION

The MOSFET power dissipations at maximum output

current are given by:

generally a good compromise for both regions of opera-

tion due to the relatively small average current.

C and C_OUTSelection

V

2

IN

OUT

P

=

I

1+δ R

+

(

) (

)

MAIN

MAX

DS(ON)

V

IN

1.85

In continuous mode, the source current of the top

N-channel MOSFET is a square wave of duty cycle V_OUT

V . To prevent large voltage transients, a low ESR input

IN

/

k V

I

(

C

f

(

)

)(

)( )

IN

MAX RSS

capacitor sized for the maximum RMS current must be

used. The maximum RMS capacitor current is given by:

V – V

2

IN

OUT

P

=

I

(

1+δ R

) (

)

SYNC

MAX

DS(ON)

V

1/ 2

]

IN

V

V – V

OUT

(

)

[

OUT IN

C Required I

≈ I

MAX

IN

RMS

V

where δ is the temperature dependency of R_DS(ON)and k

IN

is a constant inversely related to the gate drive current.

This formula has a maximum at V = 2V_OUT, where I_RMS

=

IN

I_OUT/2. This simple worst-case condition is commonly used

for design because even significant deviations do not offer

muchrelief.Notethatcapacitormanufacturer’s ripplecurrent

ratings areoftenbasedononly2000hours oflife.This makes

it advisable to further derate the capacitor or to choose a

capacitorratedatahighertemperaturethanrequired.Several

capacitors may also be paralleled to meet size or height

requirements inthedesign.Always consultthemanufacturer

if there is any question.

Both MOSFETs have I²R losses while the topside

N-channel equation includes an additional term for transi-

tion losses, which are highest at high input voltages. For

V < 20V the high current efficiency generally improves

IN

with larger MOSFETs, while for V > 20V the transition

IN

losses rapidly increase to the point that the use of a higher

R_DS(ON)device with lower C_RSSactual provides higher

efficiency. The synchronous MOSFET losses are greatest

at high input voltage or during a short circuit when the duty

cycle in this switch is nearly 100%. Refer to the Foldback

Current Limiting section for further applications information.

The selection of C_OUTis driven by the required effective

series resistance (ESR). Typically, once the ESR require-

ment is satisfied the capacitance is adequate for filtering.

The output ripple (∆V_OUT) is approximated by:

The term (1 + δ) is generally given for a MOSFET in the

form of a normalized R_DS(ON)vs Temperature curve, but

δ = 0.005/°C can be used as an approximation for low

voltageMOSFETs.C_RSSis usuallyspecifiedintheMOSFET

characteristics. The constant k = 2.5 can be used to

estimate the contributions of the two terms in the main

switch dissipation equation.

1

∆V

≈ ∆I ESR +

L

OUT

4fC

OUT

where f = operating frequency, C_OUT= output capacitance

and ∆I_L= ripple current in the inductor. The output ripple

is highest at maximum input voltage since ∆I_Lincreases

with input voltage. With ∆I_L= 0.4I_OUT(MAX)the output

The Schottky diode D1 shown in Figure 1 serves two

purposes. During continuous synchronous operation, D1

conducts during the dead-time between the conduction of

the two large power MOSFETs. This prevents the body

diode of the bottom MOSFET from turning on and storing

charge during the dead-time, which could cost as much as

1% in efficiency. During low current operation, D1 oper-

ates in conjunction with the small top MOSFET to provide

an efficient low current output stage. A 1A Schottky is

ripple will be less than 100mV at max V assuming:

IN

C_OUTRequired ESR < 2R_SENSE

Manufacturers such as Nichicon, United Chemicon and

Sanyoshouldbeconsideredforhighperformancethrough-

hole capacitors. The OS-CON semiconductor dielectric

capacitor available from Sanyo has the lowest (ESR size)

product of any aluminum electrolytic at a somewhat

14

LTC1538-AUX/ LTC1539

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APPLICATIONS INFORMATION

higher price. Once the ESR requirement for C_OUThas been

T = 70°C + (21mA)(30V)(85°C/W) = 124°C

J

met, the RMS current rating generally far exceeds the

To prevent maximum junction temperature from being

exceeded, the input supply current must be checked while

I_RIPPLE(P-P)requirement.

In surface mount applications multiple capacitors may

have to be paralleled to meet the ESR or RMS current

handling requirements of the application. Aluminum elec-

trolytic and dry tantalum capacitors are both available in

surfacemountconfigurations. Inthecaseoftantalum, itis

critical that the capacitors are surge tested for use in

switching power supplies. An excellent choice is the AVX

TPS series of surface mount tantalums, available in case

heights ranging from 2mm to 4mm. Other capacitor types

include Sanyo OS-CON, Nichicon PL series and Sprague

593D and 595D series. Consult the manufacturerforother

specific recommendations.

operating in continuous mode at maximum V .

IN

EXTV_CCConnection

The LTC1538-AUX/LTC1539 contain an internal P-chan-

nel MOSFET switch connected between the EXTV_CCand

INTV_CCpins. When the voltage applied to EXTV rises

CC

above 4.8V, the internal regulator is turned off and an

internal switch closes, connecting the EXTV_CCpin to the

INTV_CCpin thereby supplying internal power to the IC. The

switch remains closed as long as the voltage applied to

EXTV_CCremains above 4.5V. This allows the MOSFET

driver and control power to be derived from the output

during normal operation (4.8V < V_OUT< 9V) and from the

internal regulator when the output is out of regulation

(start-up, short circuit). Do not apply greater than 10V to

the EXTV_CCpin and ensure that EXTV ≤ V .

INTV /5V Standby Regulator

CC

An internal P-channel low dropout regulator produces 5V

at the INTV pin from the V supply pin. INTV powers

CC

IN

CC

IN

the drivers and internal circuitry within the LTC1538-AUX/

LTC1539,as wellas any“wake-up”circuitrytiedexternally

to the INTV_CCpin. The INTV_CCpin regulator can supply

40mA and must be bypassed to ground with a minimum

of 2.2µF tantalum or low ESR electrolytic capacitor. Good

bypassing is necessary to supply the high transient cur-

rents required by the MOSFET gate drivers.

Significant efficiency gains can be realized by powering

INTV_CCfrom the output, since the V current resulting

IN

from the driver and control currents will be scaled by a

factor of Duty Cycle/Efficiency. For 5V regulators this

supply means connecting the EXTV_CCpin directly to V_OUT

.

However, for 3.3V and other lower voltage regulators,

additional circuitry is required to derive INTV_CCpower

from the output.

To prevent any interaction due to the high transient gate

currents being drawn from the external capacitor an

additional series filter of 10Ω and 10µF to SGND can be

added.

The following list summarizes the four possible connec-

tions for EXTV_CC:

1. EXTV_CCleft open (or grounded). This will cause INTV

CC

High input voltage applications in which large MOSFETs

are being driven at high frequencies may cause the maxi-

mum junction temperature rating for the LTC1538-AUX/

LTC1539 to be exceeded. The IC supply current is domi-

nated by the gate charge supply current when not using an

to be powered from the internal 5V regulator resulting

in an efficiency penalty of up to 10% at high input

voltages.

2. EXTV_CCconnected directly to V_OUT. This is the normal

connection for a 5V regulator and provides the highest

efficiency.

output derived EXTV source. The gate charge is depen-

CC

dent on operating frequency as discussed in the Efficiency

Considerations section. The junction temperature can be

estimated by using the equations given in Note 1 of the

Electrical Characteristics. For example, the LTC1539 is

limited to less than 21mA from a 30V supply:

3. EXTV_CCconnectedtoanoutput-derivedboostnetwork.

For 3.3V and other low voltage regulators, efficiency

gains can still be realized by connecting EXTV_CCto an

output-derived voltage which has been boosted to

15

LTC1538-AUX/ LTC1539

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+

LTC1538-AUX

LTC1539*

+

V

IN

1µF

1N4148

V

C

SEC

LTC1538-AUX

LTC1539*

IN

0.22µF

BAT85

L1

+

V

IN

V

IN

+

C

L1

1:1

IN

1µF

TGL1

TGS1*

SW1

N-CH

V

IN

•

EXTV

CC

BAT85

VN2222LL

R

SENSE

N-CH

TGL1

N-CH

R6

R5

V

OUT

•

R

SENSE

TGS1*

N-CH

SFB1

V

OUT

EXTV

CC

+

BG1

SW1

BG1

C

OUT

+

SGND PGND

C

OUT

1538 F04a

PGND

OPTIONAL EXTV

*TGS1 ONLY AVAILABLE ON THE LTC1539

CC

1538 F04b

CONNECTION

5V ≤ V ≤ 9V

*TGS1 ONLY AVAILABLE ON THE LTC1539

SEC

Figure 4a. Secondary Output Loop and EXTV_CCConnection

Figure 4b. Capacitive Charge Pump for EXTV

CC

greater than 4.8V. This can be done with either the

inductive boost winding as shown in Figure 4a or the

capacitivechargepumpshowninFigure4b. Thecharge

pump has the advantage of simple magnetics.

Output Voltage Programming

The LTC1538-AUX/LTC1539 have pin selectable output

voltage programming. The output voltage is selected by

the V_PROG1(V_PROG2) pin as follows:

4. EXTV_CCconnected to an external supply. If an external

V

_PROG1,2= 0V

V_OUT1,2= 3.3V

V_OUT1,2= 5V

V_OUT2= Adjustable

supplyis availableinthe5Vto10Vrange(EXTV ≤V )

CC

IN

V_PROG1,2= INTV

CC

it may be used to power EXTV_CCproviding it is compat-

ible with the MOSFET gate drive requirements. When

driving standard threshold MOSFETs, the external sup-

ply must be always present during operation to prevent

MOSFETfailureduetoinsufficientgatedrive. Note:care

must be taken when using the connections in items 3 or

V_PROG2= Open (DC)

The top of an internal resistive divider is connected to

SENSE^–1 pin in Controller 1. For fixed output voltage

applications the SENSE^–1 pin is connected to the output

voltage as shown in Figure 5a. When using an external

resistivedividerforController2,theV_PROG2pinis leftopen

4. These connections will effect the INTV voltage

CC

when either or both controllers are on.

GND: V

= 3.3V

= 5V

OUT

V

PROG1

INTV : V

CC OUT

–

SENSE

1

V

OUT

Topside MOSFET Driver Supply (C_B,D_B)

+

LTC1538-AUX

LTC1539

C

OUT

External bootstrap capacitors C_Bconnected to the BOOST

1 and BOOST 2 pins supply the gate drive voltages for the

topside MOSFETs. Capacitor C_Bin the Functional Diagram

SGND

1538 F05a

Figure 5a. LTC1538-AUX/LTC1539 Fixed Output Applications

is charged through diode D_Bfrom INTV when the

CC

SW1(SW2) pin is low. When one of the topside MOSFETs

is to be turned on, the driver places the C_Bvoltage across

the gate source of the desired MOSFET. This enhances the

MOSFET and turns on the topside switch. The switch node

1.19V ≤ V

OUT

≤ 9V

R2

V

*

OPEN (DC)

PROG2

V

OSENSE2

voltage SW1(SW2) rises to V and the BOOST 1(BOOST

2) pin follows. With the topside MOSFET on, the boost

LTC1538-AUX

LTC1539

IN

R1

100pF

SGND

voltage is above the input supply: V_BOOST= V + V

.

1538 F05b

IN

INTVCC

R2

R1

The value of the boost capacitor C_Bneeds to be 100 times

thatofthetotalinputcapacitanceofthetopsideMOSFET(s).

The reverse breakdown on D_Bmust be greater than

V

OUT

= 1.19V 1 +

*LTC1539 ONLY

(

)

Figure 5b. LTC1538-AUX/LTC1539 Adjustable Applications

V

IN(MAX)

.

16

LTC1538-AUX/ LTC1539

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(DC) and the V_OSENSE2pin is connected to the feedback

resistors as shown in Figure 5b. Controller 2 will force the

externally attenuated output voltage to 1.19V.

required from the input power supply. If RUN/SS has been

pulled all the way to ground there is a delay before starting

of approximately 500ms/µF, followed by a similar time to

reach full current on that controller.

Power-On Reset Function (POR)

By pulling both RUN/SS controller pins below 1.3V, the

LTC1538-AUX/LTC1539 are put into shutdown

(I_Q< 200µA). These pins can be driven directly from logic

as showninFigure6.DiodeD1inFigure6reduces thestart

The power-on reset function monitors the output voltage

of the first controller and turns on an open drain device

when it is below its properly regulated voltage. An external

pull-up resistor is required on the POR1 pin.

3.3V

OR 5V

RUN/SS1

(RUN/SS2)

RUN/SS1

(RUN/SS2)

When power is first applied or when coming out of

shutdown, the POR1 output is held at ground. When the

output voltage rises above a level which is 5% below the final

regulated output value, an internal counter starts. After this

counter counts 2¹⁶(65536) clock cycles, the POR1 pull-

down device turns off. The POR1 output is active when both

D1

C

SS

1538 F06

Figure 6. RUN/SS Pin Interfacing

controllers are shut down as long as V is powered.

IN

The POR1 output will go low whenever the output voltage

of the first controller drops below 7.5% of its regulated

value for longer than approximately 30µs, signaling an

out-of-regulation condition. In shutdown, when RUN/SS1

and RUN/SS2 are both below 1.3V, the POR1 output is

pulled low even if the regulator’s output is held up by an

external source.

delay but allows C_SSto ramp up slowly providing the soft

startfunction;this diodeandC_SScanbedeletedifsoftstart

is not needed. Each RUN/SS pin has an internal 6V Zener

clamp (See Functional Diagram).

Foldback Current Limiting

As described in Power MOSFET and D1 Selection, the

worst-case dissipation for either MOSFET occurs with a

short-circuited output, when the synchronous MOSFET

conducts the current limit value almost continuously. In

most applications this will not cause excessive heating,

even for extended fault intervals. However, when heat

sinking is at a premium or higher R_DS(ON)MOSFETs are

being used, foldback current limiting should be added to

reduce the current in proportion to the severity ofthe fault.

RUN/Soft Start Function

The RUN/SS1 and RUN/SS2 pins each serve two func-

tions. Each pin provides the soft start function and a

means to shut down each controller. Soft start reduces

surgecurrents fromV byprovidingagradualramp-upof

IN

the internal current limit. Power supply sequencing can

also be accomplished using this pin.

An internal 3µA current source charges up an external

capacitor C_SS.When the voltage on RUN/SS1 (RUN/SS2)

reaches 1.3V the particular controller is permitted to start

operating. As the voltage on the pin continues to ramp

from 1.3V to 2.4V, the internal current limit is also ramped

at a proportional linear rate. The current limit begins at

approximately50mV/R_SENSE(atV_RUN/SS=1.3V)andends

at 150mV/R_SENSE(V_RUN/SS≥ 2.7V). The output current

thus ramps up slowly, reducing the starting surge current

Foldback current limiting is implemented by adding diode

D_FBbetween the output and the I_THpin as shown in the

Functional Diagram. In a hard short (V_OUT= 0V) the

current will be reduced to approximately 25% of the

maximum output current. This technique may be used for

all applications with regulated output voltages of 1.8V or

greater.

17

LTC1538-AUX/ LTC1539

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EXTERNAL

FREQUENCY

R

LP

2.4V

C

OSC

C

LP

PHASE

1.3

DETECTOR*

*PLL LPF

OSC

C

OSC

*PLLIN

f

O

DIGITAL

PHASE/

FREQUENCY

DETECTOR

0.7

SGND

50k

0

0.5

1.0

1.5

(V)

2.0

2.5

V

PLLLPF

1538 F07

1538 F08

*LTC1539 ONLY

Figure 7. Operating Frequency vs V

PLLLPF

Figure 8. Phase-Locked Loop Block Diagram

Phase-Locked Loop and Frequency Synchronization

external and internal oscillators. This type of phase detec-

tor will not lock up on input frequencies close to the

harmonics of the VCO center frequency. The PLL hold-in

range, ∆f_H, is equal to the capture range, ∆f_C:

The LTC1539 has an internal voltage-controlled oscillator

and phase detector comprising a phase-locked loop. This

allows the top MOSFET turn-on to be locked to the rising

edge of an external source. The frequency range of the

voltage-controlled oscillator is ±30% around the center

frequency f_O.

∆f_H= ∆f_C= ±0.3 f_O.

The output of the phase detector is a complementary pair

of current sources charging or discharging the external

filter network on the PLL LPF pin. A simplified block

diagram is shown in Figure 8.

The value of C_OSCis calculated from the desired operating

frequency (f_O). Assuming the phase-locked loop is locked

(V_PLLLPF= 1.19V):

If the external frequency f_PLLINis greater than the oscilla-

tor frequency f_OSC, current is sourced continuously, pull-

ingupthePLLLPFpin.Whentheexternalfrequencyis less

than f_0SC, current is sunk continuously, pulling down the

PLLLPFpin.Iftheexternalandinternalfrequencies arethe

same but exhibit a phase difference, the current sources

turn on for an amount of time corresponding to the phase

difference. Thus thevoltageonthePLLLPFpinis adjusted

until the phase and frequency of the external and internal

oscillators are identical. At this stable operating point the

phase comparator output is open and the filter capacitor

C_LPholds the voltage. The LTC1539 PLLIN pin must be

driven from a low impedance such as a logic gate located

close to the pin. Any external attenuator used needs to be

referenced to SGND.

4

2.1 10

(

)

C

(pF) =

– 11

OSC

Frequency(kHz)

Stating the frequency as a function of V_PLLLPFand C_OSC

Frequency(kHz) =

:

8

8.4 10

(

)

1

C

[

pF +11

( )

+ 2000

]

OSC

V

PLLLPF

2.4V

17µA+18µA

The phase detector used is an edge sensitive digital type

which provides zero degrees phase shift between the

The loop filter components C_LP, R_LPsmooth out the

current pulses from the phase detector and provide a

18

LTC1538-AUX/ LTC1539

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SFB1 Pin Operation

stable input to the voltage-controlled oscillator. The filter

components C_LPand R_LPdetermine how fast the loop

acquires lock.Typically,R_LP=10kandC_LPis 0.01µFto0.1µF.

The low side of the filter needs to be connected to SGND.

When the SFB1 pin drops below its ground referenced

1.19V threshold, continuous mode operation is forced. In

continuous mode, the large N-channel main and synchro-

nous switches are used regardless of the load on the main

output.

The PLL LPF pin can be driven with external logic to obtain

a 1:1.9 frequency shift. The circuit shown in Figure 9 will

provide a frequency shift from f_Oto 1.9f_Oas the voltage on

In addition to providing a logic input to force continuous

synchronous operation, the SFB1 pin provides a means to

regulate a flyback winding output. The use of a synchro-

nous switch removes the requirement that power must be

drawn from the inductor primary in order to extract power

from the auxiliary winding. With the loop in continuous

mode, the auxiliary output may be loaded without regard

to the primary output load. The SFB1 pin provides a way

to force continuous synchronous operation as needed by

the flyback winding.

V_PLLLPFincreases from 0V to 2.4V. Do not exceed 2.4V on

V

.

PLLLPF

3.3V OR 5V

2.4V

MAX

PLL LPF

18k

LTC1538 • F09

Figure 9. Directly Driving PLL LPF Pin

Low Battery Comparator

Thesecondaryoutputvoltageis setbytheturns ratioofthe

transformerinconjunctionwithapairofexternalresistors

returned to the SFB1 pin as shown in Figure 4a. The

secondary regulated voltage V_SECin Figure 4a is given by:

The LTC1539 has an on-chip low battery comparator

which can be used to sense a low battery condition when

implemented as shown in Figure 10. This comparator is

active during shutdown allowing battery charge level

interrogation prior to and after powering up part or all of

the system. The resistor divider R3/R4 sets the compara-

tor trip point as follows:

R6

V

≈ N+1 V

> 1.19V 1+

(

)

SEC

OUT

R5

where N is the turns ratio of the transformer, and V_OUTis

the main output voltage sensed by SENSE^–1.

R4

V

= 1.19V 1+

LBITRIP

R3

Auxiliary Regulator/Comparator

The auxiliary regulator/comparator can be used as a

comparator or low dropout regulator (by adding an exter-

nal PNP pass device).

The divided down voltage at the negative (–) input to the

comparator is compared to an internal 1.19V reference. A

20mV hysteresis is built in to assure rapid switching. The

output is an open drain MOSFET and requires a pull-up

resistor. This comparator is active when both the RUN/

SS1 and RUN/SS2 pins are low. The low side of the resistive

divider needs to be connected to SGND.

When the voltage present at the AUXON pin is greater than

1.19V the regulator/comparator is on. The amplifier is

stable when operating as a low dropout regulator. This

same amplifier can be used as a comparator whose

inverting input is tied to the 1.19V reference.

V

IN

The AUXDR pin is internally connected to an open drain

MOSFET which can sink up to 10mA. The voltage on

AUXDR determines whether or not an internal 12V resis-

tive divider is connected to AUXFB as described below. A

pull-up resistor is required on AUXDR and the voltage

must not exceed 28V.

R4

R3

LTC1539

LBO

LBI

–

+

1.19V REFERENCE

1538 F10

SGND

Figure 10. Low Battery Comparator

19

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With the addition of an external PNP pass device, a linear

regulator capable of supplying up to 0.5A is created. As

shown in Figure 11a, the base of the external PNP con-

nects to the AUXDR pin together with a pull-up resistor.

The output voltage V_OAUXat the collector of the external

PNP is sensed by the AUXFB pin.

in hysteresis. For other output voltages, an external resis-

tive divider is fed back to AUXFB as shown in Figure 11b.

The output voltage V_OAUXis set as follows:

R8

V

= 1.19V 1+

= 12V

< 8V AUX DR < 8.5V

AUX DR ≥ 12V

OAUX

R7

V

OAUX

The input voltage to the auxiliary regulator can be taken

from a secondary winding on the primary inductor as

shown in Figure 11a. In this application, the SFB1 pin

regulates the input voltage to the PNP regulator (see SFB1

Pin Operation) and should be set to approximately 1V to

2V above the required output voltage of the auxiliary

regulator.AZenerclampdiodemayberequiredtokeepthe

secondary winding resultant output voltage under the 28V

AUXDR pin specification when the primary is heavily

loaded and the secondary is not.

When used as a voltage comparator as shown in Figure

11c, the auxiliary block has a noninverting characteristic.

When AUXFB drops below 1.19V, the AUXDR pin will be

pulled low. A minimum current of 5µA is required to pull up

the AUXDR pin to 5V when used as a comparator output in

ordertocounteracta1.5µAinternalpull-downcurrentsource.

Efficiency Considerations

The efficiency of a switching regulator is equal to the

output power divided by the input power times 100%. It is

oftenusefultoanalyzeindividuallosses todeterminewhat

is limiting the efficiency and which change would produce

the most improvement. Efficiency can be expressed as:

The AUXFB pin is the feedback point of the regulator. An

internalresistordivideris availabletoprovidea12Voutput

bysimplyconnectingAUXFBdirectlytothecollectorofthe

external PNP. The internal resistive divider is switched in

when the voltage at AUXFB goes above 9.5V with 1V built-

Efficiency = 100% – (L1 + L2 + L3 + ...)

SECONDARY

WINDING

SECONDARY

WINDING

1:N

R6

V

SEC

= 1.19V 1 +

> 13V

V

SEC

= 1.19V 1 +

> (V

+ 1V)

OAUX

(

)

(

)

R5

V

SEC

V

SEC

V

OAUX

AUXDR

R6

R5

AUXDR

+

R6

R5

R8

R7

V

12V

+

OAUX

SFB1 AUXFB

+

LTC1538-AUX/

LTC1539

LTC1538-AUX/

LTC1539

10µF

AUXON

ON/OFF

AUXON

ON/OFF

1538 F11a

1538 F11b

Figure 11a. 12V Output Auxiliary Regulator

Using Internal Feedback Resistors

Figure 11b. 5V Output Auxiliary Regulator Using

External Feedback Resistors

V

< 7.5V

PULL-UP

LTC1538-AUX/LTC1539

ON/OFF

INPUT

AUXON

AUXDR

OUTPUT

AUXFB

–

+

1.19V REFERENCE

1538 F11c

Figure 11c. Auxiliary Comparator Configuration

20

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whereL1, L2, etc. aretheindividuallosses as apercentage

of input power.

the resistance of one MOSFET can simply be summed

with the resistances of L and R_SENSEto obtain I²R

losses. For example, if each R_DS(ON)= 0.05Ω, R_L=

0.15Ω and R_SENSE= 0.05Ω, then the total resistance is

0.25Ω. This results in losses ranging from 3% to 10%

as the output current increases from 0.5A to 2A. I²R

losses cause the efficiency to roll off at high output

currents.

Although all dissipative elements in the circuit produce

losses, four main sources usually account for most of the

losses inLTC1538-AUX/LTC1539circuits.LTC1538-AUX/

LTC1539 V current, INTV_CCcurrent, I²R losses and

IN

topside MOSFET transition losses.

1. The V current is the DC supply current given in the

IN

4. Transition losses apply only to the topside MOSFET(s)

and only when operating at high input voltages (typically

20V or greater). Transition losses can be estimated from:

ElectricalCharacteristics whichexcludes MOSFETdriver

and control currents. V current typically results in a

IN

small (<< 1%) loss which increases with V .

IN

Transition Loss ≈ 2.5(V )^1.85(I_MAX)(C_RSS)(f)

IN

2. INTV_CCcurrent is the sum of the MOSFET driver and

control currents. The MOSFET driver current results

from switching the gate capacitance of the power

MOSFETs. Each time a MOSFET gate is switched from

low to high to low again, a packet of charge dQ moves

Other losses including C and C_OUTESR dissipative

IN

losses, Schottky conduction losses during dead-time,

and inductor core losses, generally account for less

than 2% total additional loss.

from INTV to ground. The resulting dQ/dt is a current

CC

Checking Transient Response

out of INTV which is typically much larger than the

CC

control circuit current. In continuous mode, I_GATECHG

=

The regulator loop response can be checked by looking at

the load transient response. Switching regulators take

several cycles to respond to a step in DC (resistive) load

current. When a load step occurs, V_OUTshifts by an

amount equal to (∆I_LOAD)(ESR) where ESR is the effective

series resistance of C_OUT. ∆I_LOADalso begins to charge or

dischargeC_OUTgeneratingthefeedbackerrorsignalwhich

forces the regulator loop to adapt to the current change

and return V_OUTto its steady-state value. During this

recovery time V_OUTcan be monitored for overshoot or

ringing which would indicate a stability problem. The I_TH

external components shown in Figure 1 will prove ad-

equate compensation for most applications.

f(Q_T+ Q_B), where Q_Tand Q_Bare the gate charges of the

topside and bottom side MOSFETs. It is for this reason

that the large topside and synchronous MOSFETs are

turned off during low current operation in favor of the

small topside MOSFET and external Schottky diode,

allowing efficient, constant-frequency operation at low

output currents.

By powering EXTV_CCfrom an output-derived source,

the additional V current resulting from the driver and

IN

control currents will be scaled by a factor of Duty Cycle/

Efficiency. For example, in a 20V to 5V application,

10mA of INTV current results in approximately 3mA

CC

of V current. This reduces the midcurrent loss from

10% or more (if the driver was powered directly from

IN

A second, more severe transient is caused by switching in

loads with large (> 1µF) supply bypass capacitors. The

dischargedbypass capacitors areeffectivelyputinparallel

with C_OUT, causing a rapid drop in V_OUT. No regulator can

deliver enough current to prevent this problem if the load

switch resistance is low and it is driven quickly. The only

solution is to limit the rise time of the switch drive so that

the load rise time is limited to approximately (25)(C_LOAD).

Thus a 10µF capacitor would require a 250µs rise time,

limiting the charging current to about 200mA.

V ) to only a few percent.

IN

3. I²R losses are predicted from the DC resistances of the

MOSFET, inductor and current sense R. In continuous

mode the average output current flows through L and

R_SENSE, but is “chopped” between the topside main

MOSFET and the synchronous MOSFET. If the two

MOSFETs have approximately the same R_DS(ON), then

21

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Automotive Considerations: Plugging into the

Cigarette Lighter

and C_OSCcan immediately be calculated:

R

_SENSE= 100mV/3A = 0.033Ω

As battery-powered devices go mobile, there is a natural

interest in plugging into the cigarette lighter in order to

conserveorevenrechargebatterypacks duringoperation.

But before you connect, be advised: you are plugging into

the supply from hell. The main battery line in an automo-

bile is the source ofa numberofnastypotentialtransients,

including load dump, reverse battery and double battery.

C_OSC= (1.37(10⁴)/250) – 11 ≈ 43pF

Referring to Figure 3, a 10µH inductor falls within the

recommended range. To check the actual value of the

ripple current the following equation is used :

V

OUT

V

IN

OUT

∆I =

1–

L

(f)(L)

Load dump is the result of a loose battery cable. When the

cablebreaks connection,thefieldcollapseinthealternator

can cause a positive spike as high as 60V which takes

several hundred milliseconds to decay. Reverse battery is

just what it says, while double battery is a consequence of

tow-truck operators finding that a 24V jump start cranks

cold engines faster than 12V.

The highest value of the ripple current occurs at the

maximum input voltage:

3.3V

22V

∆I =

1–

= 1.12A

L

250kHz(10µH)

The power dissipation on the topside MOSFET can be

easily estimated. Using a Siliconix Si4412DY for example;

R_DS(ON)= 0.042Ω, C_RSS= 100pF. At maximum input

voltage with T(estimated) = 50°C:

The network shown in Figure 12 is the most straightfor-

ward approach to protect a DC/DC converter from the

ravages of an automotive battery line. The series diode

prevents current from flowing during reverse battery,

while the transient suppressor clamps the input voltage

during load dump. Note that the transient suppressor

should not conduct during double battery operation, but

must still clamp the input voltage below breakdown of the

converter. Although the LTC1538-AUX/LTC1539 has a

maximum input voltage of 36V, most applications will be

3.3V

22V

2

P

=

3

1+ 0.005 50°C − 25°C 0.042Ω

)( ) ( )

( )

(

[

]

MAIN

1.85

+ 2.5 22V

3A 100pF 250kHz = 122mW

)( )(

(

)

(

)

The most stringent requirement for the synchronous

N-channel MOSFET is with V_OUT= 0V (i.e. short circuit).

During a continuous short circuit, the worst-case dissipa-

tion rises to:

limited to 30V by the MOSFET BV

.

DSS

Design Example

As a design example, assume V = 12V(nominal), V =

P_SYNC= [I_SC(AVG)]²(1 + δ)R_DS(ON)

IN

22V(max), V_OUT= 3.3V, I_MAX= 3A and f = 250kHz, R_SENSE

12V

50A I RATING

PK

V

IN

LTC1538-AUX/

LTC1539

TRANSIENT VOLTAGE

SUPPRESSOR

GENERAL INSTRUMENT

1.5KA24A

1538 F12

Figure 12. Automotive Application Protection

22

LTC1538-AUX/ LTC1539

U

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APPLICATIONS INFORMATION

With the 0.033Ω sense resistor I_SC(AVG)= 4A will result,

increasing the Si4412DY dissipation to 950mW at a die

temperature of 105°C.

4. Do the (+) plates of C connect to the drains of the

IN

topside MOSFETs as closely as possible? This capaci-

tor provides the AC current to the MOSFETs.

C will require an RMS current rating of at least 1.5A at

5. Is the INTV_CCdecoupling capacitor connected closely

between INTV_CCand the power ground pin? This ca-

pacitor carries the MOSFET driver peak currents.

IN

temperatureandC_OUTwillrequireanESRof0.03Ωforlow

outputripple.Theoutputrippleincontinuous modewillbe

highest at the maximum input voltage. The output voltage

ripple due to ESR is approximately:

6. Keep the switching nodes, SW1 (SW2), away from

sensitive small-signal nodes. Ideally the switch nodes

should be placed at the furthest point from the

LTC1538-AUX/LTC1539.

V

_ORIPPLE= R_ESR(∆I_L) = 0.03Ω(1.12A) = 34mV

P-P

PC Board Layout Checklist

7. Use a low impedance source such as a logic gate to drive

the PLLIN pin and keep the lead as short as possible.

When laying out the printed circuit board, the following

checklistshouldbeusedtoensureproperoperationofthe

LTC1538-AUX/LTC1539. These items are also illustrated

graphically in the layout diagram of Figure 13. Check the

following in your layout:

PC BOARD LAYOUT SUGGESTIONS

Switching power supply printed circuit layouts are cer-

tainly among the most difficult analog circuits to design.

The following suggestions will help to get a reasonably

close solution on the first try.

1. Are the high current power ground current paths using

or running through any part of signal ground? The

LTC1438/LTC1438X/LTC1439 IC’s have their sensitive

pins on one side of the package. These pins include the

signal ground for the reference, the oscillator input, the

voltageandcurrentsensingforbothcontrollers andthe

low battery/comparator input. The signal ground area

used on this side of the IC must return to the bottom

plates of all of the output capacitors. The high current

power loops formed by the input capacitors and the

ground returns to the sources of the bottom

N-channel MOSFETs, anodes of the Schottky diodes,

The output circuits, including the external switching

MOSFETs, inductor, secondary windings, sense resistor,

input capacitors and output capacitors all have very large

voltage and/or current levels associated with them. These

components and the radiated fields (electrostatic and/or

electromagnetic) must be kept away from the very sensi-

tive control circuitry and loop compensation components

required for a current mode switching regulator.

The electrostatic or capacitive coupling problems can be

reduced by increasing the distance from the radiator,

typically a very large or very fast moving voltage signal.

The signal points that cause problems generally include:

the “switch” node, any secondary flyback winding voltage

and any nodes which also move with these nodes. The

switch,MOSFETgate,andboostnodes movebetweenVIN

andPgndeachcyclewithless thana100ns transitiontime.

The secondary flyback winding output has an AC signal

and (-) plates of C , should be as short as possible and

IN

tied through a low resistance path to the bottom plates

of the output capacitors for the ground return.

2. DotheLTC1538-AUX/LTC1539SENSE^–1andV_OSENSE2

pins connect to the (+) plates of C_OUT? In adjustable

applications, the resistive divider R1/R2 must be con-

nected between the (+) plate of C_OUTand signal ground

and the HF decoupling capacitor should be as close as

possible to the LTC1538-AUX/LTC1539.

component of –V times the turns ratio of the trans-

IN

former, and also has a similar < 100ns transition time. The

feedbackcontrolinputsignals needtohaveless thanafew

millivolts of noise in order for the regulator to perform

properly. A rough calculation shows that 80dB of isolation

at 2MHz is required from the switch node for low noise

switcheroperation.Thesituationis worsebyafactorofthe

–

3. AretheSENSE andSENSE⁺leads routedtogetherwith

minimum PC trace spacing? The filter capacitors be-

tween SENSE⁺1 (SENSE⁺2) and SENSE^–1 (SENSE^–2)

should be as close as possible to the LTC1538-AUX/

LTC1539.

23

LTC1538-AUX/ LTC1539

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APPLICATIONS INFORMATION

turns ratio for the secondary flyback winding. Keep these

switch-node-related PC traces small and away from the

“quiet” side of the IC (not just above and below each other

on the opposite side of the board).

duced is picked up by the current comparator input filter

circuit(s),as wellas bythevoltagefeedbackcircuit(s).The

current comparator’s filter capacitor placed across the

sense pins attenuates the radiated current signal. It is

important to place this capacitor immediately adjacent to

the IC sense pins. The voltage sensing input(s) minimizes

the inductive pickup component by using an input capaci-

tance filter to SGND. The capacitors in both case serve to

integrate the induced current, reducing the susceptibility

to both the “loop” radiated magnetic fields and the trans-

former or inductor leakage fields.

The electromagnetic or current-loop induced feedback

problems can be minimized by keeping the high AC-

current (transmitter) paths and the feedback circuit (re-

ceiver)pathsmalland/orshort.Maxwell’s equations areat

work here, trying to disrupt our clean flow of current and

voltage information from the output back to the controller

input. It is crucial to understand and minimize the suscep-

tibility of the control input stage as well as the more

obvious reduction of radiation from the high-current out-

put stage(s). An inductive transmitter depends upon the

frequency, current amplitude and the size of the current

loop to determine the radiation characteristic of the gen-

erated field. The current levels are set in the output stage

oncetheinputvoltage,outputvoltageandinductorvalue(s)

have been selected. The frequency is set by the output-

stage transition times. The only parameter over which we

have some control is the size of the antenna we create on

the PC board, i.e., the loop. A loop is formed with the input

capacitance, the top MOSFET, the Schottky diode, and the

pathfromtheSchottkydiode’s groundconnectionandthe

input capacitor’s ground connection. A second path is

formed when a secondary winding is used comprising the

secondary output capacitor, the secondary winding and

the rectifier diode or switching MOSFET (in the case of a

synchronous approach). These “loops” should be kept as

small and tightly packed as possible in order to minimize

their “far field” radiation effects. The radiated field pro-

The capacitor on INTV acts as a reservoir to supply the

CC

high transient currents to the bottom gates and to re-

charge the boost capacitor. This capacitor should be a

4.7µFtantalumcapacitorplacedas closeas possibletothe

INTV and PGND pins of the IC. Peak current driving the

CC

MOSFET gates exceeds 1A. The power ground pin of the

IC, connected to this capacitor, should connect directly to

the lower plates of the output capacitors to minimize the

AC ripple on the INTV IC power supply.

CC

The previous instructions will yield a PC layout which has

three separate ground regions returning separately to the

bottom plates of the output capacitors: a signal ground, a

MOSFET gate/INTV_CCground and the ground from the

input capacitors, Schottky diode and synchronous

MOSFET. In practice, this may produce a long power

ground path from the input and output capacitors. A long,

low resistance path between the input and output capaci-

tor power grounds will not upset the operation of the

switching controllers as long as the signal and power

grounds from the IC pins does not “tap in” along this path.

24

LTC1538-AUX/ LTC1539

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APPLICATIONS INFORMATION

C

LP

R

10k

0.01µF

LP

C

0.1µF

SS

EXT

CLOCK

1

2

36

35

34

33

32

31

30

29

28

27

26

25

24

23

22

21

20

19

RUN/SS1 PLL LPF

+

1000pF

C

0.1µF

B1

SENSE 1

–

PLLIN

1000pF

C

C1B

3

SENSE 1 BOOST 1

220pF

C

C1A

4

C

IN1

M1

INTV

V

PROG1

TGL1

SW1

TGS1

1000pF

CC

5

I

TH1

L1

100k

R

10k

C1

6

V

IN

M3

POR2

C

OSC

+

LTC1539

7

R

SENSE1

C

OSC

V

IN

D

B1

8

V

OUT1

+

V

M2

M5

D1

SGND

LBI

BG1

INTV

IN

+

C

OUT1

4.7µF

9

–

CC

C

–

C2B

GROUND PLANE

470pF

10

11

12

13

14

15

16

17

18

–

LBO

SFB1

PGND

BG2

C

OUT2

C2A

+

D2

INT V

R

1000pF

CC

C2

V

OUT2

10k

D

B2

I

EXTV

CC

TH2

R

SENSE2

+

100pF

L2

V

PROG2

TGS2

M6

56pF

V

SW2

OSENSE2

22pF

–

SENSE 2

TGL2

M4

C

IN2

OUTPUT DIVIDER

REQUIRED WITH

1000pF

+

SENSE 2 BOOST 2

V

OPEN

PROG

C

B2

RUN/SS2

AUXDR

AUXON

AUXFB

0.1µF

C

0.1µF

SS

10Ω

220pF

10Ω

1538 F13

NOT ALL PINS CONNECTED FOR CLARITY

BOLD LINES INDICATE HIGH CURRENT PATHS

Figure 13. LTC1539 Physical Layout Diagram

25

LTC1538-AUX/ LTC1539

U

TYPICAL APPLICATIONS

+

26

LTC1538-AUX/ LTC1539

U

TYPICAL APPLICATIONS

27

LTC1538-AUX/ LTC1539

U

TYPICAL APPLICATIONS

28

LTC1538-AUX/ LTC1539

U

TYPICAL APPLICATIONS

29

LTC1538-AUX/ LTC1539

U

W

PCB LAYOUT A D FIL

(Gerber files for this circuit board are available. Call the LTC factory.)

Silkscreen Top

Silkscreen Bottom

Copper Layer 1

Copper Layer 2 Ground Plane

Copper Layer 3

Copper Layer 4

30

LTC1538-AUX/ LTC1539

U

PACKAGE DESCRIPTION

Dimensions in inches (millimeters) unless otherwise noted.

G Package

28-Lead Plastic SSOP (0.209)

(LTC DWG # 05-08-1640)

0.397 – 0.407*

(10.07 – 10.33)

28 27 26 25 24 23 22 21 20 19 18

16 15

17

0.301 – 0.311

(7.65 – 7.90)

5

7

8

1

2

3

4

6

9 10 11 12 13 14

0.205 – 0.212**

(5.20 – 5.38)

0.068 – 0.078

(1.73 – 1.99)

0° – 8°

0.0256

(0.65)

BSC

0.005 – 0.009

(0.13 – 0.22)

0.022 – 0.037

(0.55 – 0.95)

0.002 – 0.008

(0.05 – 0.21)

0.010 – 0.015

(0.25 – 0.38)

*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD

FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

G28 SSOP 0694

GW Package

36-Lead Plastic SSOP (Wide 0.300)

(LTC DWG # 05-08-1642)

0.602 – 0.612*

(15.290 – 15.544)

36 35 34 33 32 31 30 29 28 27 26 25 24 23 22 21 20 19

0.400 – 0.410

(10.160 – 10.414)

0.292 – 0.299**

(7.417 – 7.595)

1

2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18

0.090 – 0.094

(2.286 – 2.387)

0.097 – 0.104

(2.463 – 2.641)

0.010 – 0.016

(0.254 – 0.406)

× 45°

0° – 8° TYP

0.005 – 0.012

(0.127 – 0.305)

0.024 – 0.040

(0.610 – 1.016)

0.031

(0.800)

TYP

0.009 – 0.012

(0.231 – 0.305)

0.012 – 0.017

(0.304 – 0.431)

*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD

SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

GW36 SSOP 0795

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tationthattheinterconnectionofits circuits as describedhereinwillnotinfringeonexistingpatentrights.

31

LTC1538-AUX/ LTC1539

U

TYPICAL APPLICATION

3.3V to 2.9V at 3A Low Noise Linear Regulator

5V

27Ω

6.8nF

47k

3.3V

ZETEX

Q1

MMBT2907ALTI

FZT849

(SURFACE MOUNT)

10Ω

100Ω

2.9V

3A

AUXDR

LTC1539

AUXFB

316k

1%

22pF

+

330µF

× 2

2.9V

ON/OFF

221k

1%

AUXON

1538 TA05

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DESCRIPTION

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Constant-Voltage/Constant-Current Battery Charger

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LT/GP 0896 7K • PRINTED IN USA

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

32

●

(408) 432-1900 FAX: (408) 434-0507 TELEX: 499-3977

LINEAR TECHNOLOGY CORPORATION 1996


型号：	LTC1539CGW
厂家：	Linear
描述：	Dual High Efficiency, Low Noise, Synchronous Step-Down Switching Regulators 双通道高效率，低噪声，同步降压型开关稳压器稳压器开关
文件：	总32页 (文件大小：454K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

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