LT1228CJ8#PBF_技术文档

LT1228

100MHz Current Feedback

Amplifier with DC Gain Control

U

FEATURES

DESCRIPTIO

The LT^®1228 makes it easy to electronically control the

gain of signals from DC to video frequencies. The LT1228

implements gain control with a transconductance ampli-

fier (voltage to current) whose gain is proportional to an

externally controlled current. A resistor is typically used to

convert the output current to a voltage, which is then

amplified with a current feedback amplifier. The LT1228

combines both amplifiers into an 8-pin package, and

operates on any supply voltage from 4V (± 2V) to 30V

(±15V). A complete differential input, gain controlled

amplifier can be implemented with the LT1228 and just a

few resistors.

■

Very Fast Transconductance Amplifier

Bandwidth: 75MHz

g_m= 10 × I_SET

Low THD: 0.2% at 30mV_RMSInput

Wide I_SETRange: 1µA to 1mA

■

Very Fast Current Feedback Amplifier

Bandwidth: 100MHz

Slew Rate: 1000V/µs

Output Drive Current: 30mA

Differential Gain: 0.04%

Differential Phase: 0.1°

High Input Impedance: 25MΩ, 6pF

■

Wide Supply Range: ±2V to ±15V

The LT1228 transconductance amplifier has a high imped-

ancedifferentialinputandacurrentsourceoutputwithwide

output voltage compliance. The transconductance, g_m, is

setbythecurrentthatflowsintoPin5, I_SET. Thesmallsignal

g_misequaltotentimesthevalueofI_SETandthisrelationship

holds over several decades of set current. The voltage at Pin

5 is two diode drops above the negative supply, Pin 4.

■

Inputs Common Mode to Within 1.5V of Supplies

■

Outputs Swing Within 0.8V of Supplies

Supply Current: 7mA

■

Available in 8-Lead PDIP and SOIC Packages

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APPLICATIO S

■

Video DC Restore (Clamp) Circuits

■

The LT1228 current feedback amplifier has very high input

impedance and therefore it is an excellent buffer for the

output of the transconductance amplifier. The current feed-

back amplifier maintains its wide bandwidth over a wide

range of voltage gains making it easy to interface the

transconductance amplifier output to other circuitry. The

current feedback amplifier is designed to drive low imped-

ance loads, such as cables, with excellent linearity at high

frequencies.

Video Differential Input Amplifiers

■

Video Keyer/Fader Amplifiers

■

AGC Amplifiers

■

Tunable Filters

■

Oscillators

, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.

All other trademarks are the property of their respective owners.

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Frequency Response

TYPICAL APPLICATIO

6

V

= ±15V

= 100Ω

S

L

Differential Input Variable Gain Amp

R

3

0

I

= 1mA

15V

SET

4.7µF

+

–3

R3A

10k

7

4

3

2

–6

+

–

+

1

8

–9

g

R2A

10k

V

m

+

–

IN

I

= 300µA

SET

6

–12

–15

–18

–

5

CFA

V

OUT

I

SET

R

–15V

R2

F

470Ω

= 100µA

R4

1.24k

R5

10k

SET

R3

100Ω

–21

–24

4.7µF

+

100Ω

R1

270Ω

R

G

HIGH INPUT RESISTANCE

EVEN WHEN POWER IS OFF

–18dB < GAIN < 2dB

10Ω

100k

1M

10M

100M

R6

6.19k

V

≤ 3V

RMS

FREQUENCY (Hz)

IN

LT1228 • TA02

LT1228 • TA01

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LT1228

W W U W

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U

ABSOLUTE AXI U RATI GS

PACKAGE/ORDER I FOR ATIO

(Note 1)

TOP VIEW

ORDER PART

NUMBER

LT1228CN8

LT1228CS8

LT1228IN8

LT1228IS8

Supply Voltage ...................................................... ±18V

Input Current, Pins 1, 2, 3, 5, 8 (Note 8) ............ ±15mA

Output Short Circuit Duration (Note 2) .........Continuous

Operating Temperature Range

LT1228C................................................ 0°C to 70°C

LT1228I............................................. –40°C to 85°C

LT1228M (OBSOLETE) .............. –55°C to 125°C

Storage Temperature Range ..................–65°C to 150°C

Junction Temperature

I

1

2

3

4

8

7

6

5

GAIN

OUT

+

–

+

V

–IN

g

m

+IN

V

OUT

–

V

I

SET

N8 PACKAGE

8-LEAD PLASTIC DIP

S8 PACKAGE

8-LEAD PLASTIC SOIC

T_{J MAX}= 150°C, θ_JA= 100°C/W (N)

T_{J MAX}= 150°C, θ_JA= 150°C/W (S)

S8 PART MARKING

1228

1228I

Plastic Package .............................................. 150°C

Ceramic Package (OBSOLETE) ................ 175°C

Lead Temperature (Soldering, 10 sec).................. 300°C

ORDER PART

NUMBER

J8 PACKAGE

8-LEAD CERAMIC DIP

T_{J MAX}= 175°C, θ_JA= 100°C/W (J)

LT1228MJ8

LT1228CJ8

OBSOLETE PACKAGE

Consider the N8 or S8 Packages for Alternate Source.

Order Options Tape and Reel: Add #TR

Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF

Lead Free Part Marking: http://www.linear.com/leadfree/

ELECTRICAL CHARACTERISTICS ^The

●

denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at T = 25°C. Current Feedback Amplifier, Pins 1, 6, 8. ± 5V ≤ V ≤ ± 15V, I = 0µA,

A

S

SET

V

CM

= 0V unless otherwise noted.

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP MAX

UNITS

V

Input Offset Voltage

T = 25°C

±3

±10

±15

mV

OS

+

A

●

Input Offset Voltage Drift

Noninverting Input Current

10

µV/°C

I

T = 25°C

A

±0.3

±3

±10

µA

IN

●

–

Inverting Input Current

T = 25°C

A

±10

±65

±100

µA

IN

e

Input Noise Voltage Density

Input Noise Current Density

Input Resistance

f = 1kHz, R = 1k, R = 10Ω, R = 0Ω

6

nV/√Hz

pV/√Hz

n

F

G

S

i

f = 1kHz, R = 1k, R = 10Ω, R = 10k

1.4

n

F

G

S

R

V

= ±13V, V = ±15V

●

2

25

MΩ

IN

S

= ±3V, V = ±5V

S

C

Input Capacitance (Note 3)

Input Voltage Range

V = ±5V

6

pF

IN

S

V = ±15V, T = 25°C

S

±13 ±13.5

±12

V

A

●

V = ±5V, T = 25°C

S

±3

±2

±3.5

A

CMRR

Common Mode Rejection Ratio

V = ±15V, V = ±13V, T = 25°C

55

69

dB

S

CM

A

V = ±15V, V = ±12V

●

S

CM

V = ±5V, V = ±3V, T = 25°C

S

CM

A

V = ±5V, V = ±2V

S

CM

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LT1228

ELECTRICAL CHARACTERISTICS

The

●

denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at T = 25°C. Current Feedback Amplifier, Pins 1, 6, 8. ± 5V ≤ V ≤ ±15V, I = 0µA,

A

S

SET

V

= 0V unless otherwise noted.

CM

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP MAX

UNITS

Inverting Input Current

Common Mode Rejection

V = ±15V, V = ±13V, T = 25°C

2.5

10

µA/V

S

CM

A

V = ±15V, V = ±12V

●

10

S

CM

V = ±5V, V = ±3V, T = 25°C

2.5

S

CM

A

V = ±5V, V = ±2V

S

CM

PSRR

Power Supply Rejection Ratio

V = ±2V to ±15V, T = 25°C

S

60

80

10

dB

S

A

V = ±3V to ±15V

●

Noninverting Input Current

Power Supply Rejection

V = ±2V to ±15V, T = 25°C

50

nA/V

S

A

V = ± 3V to ±15V

S

Inverting Input Current

Power Supply Rejection

V = ±2V to ±15V, T = 25°C

0.1

5

µA/V

S

A

V = ± 3V to ±15V

S

A

Large-Signal Voltage Gain

V = ±15V, V

= ±10V, R = 1k

LOAD

●

55

65

dB

V

S

OUT

V = ±5V, V

= ±2V, R

= 150Ω

S

OUT

LOAD

–

R

Transresistance, ∆V /∆I

V = ±15V, V

= ±10V, R = 1k

LOAD

●

100

200

kΩ

OL

OUT IN

S

OUT

V = ±5V, V

= ±2V, R

= 150Ω

S

OUT

LOAD

V

Maximum Output Voltage Swing

V = ±15V, R

= 400Ω, T = 25°C

±12

±10

±3

±13.5

V

OUT

S

LOAD

A

●

V = ±5V, R

= 150Ω, T = 25°C

A

±3.7

S

LOAD

±2.5

I

Maximum Output Current

R

= 0Ω, T = 25°C

30

25

65

125

mA

OUT

s

LOAD

A

●

Supply Current

V

= 0V, I = 0V

6

11

mA

V/µs

ns

OUT

SET

SR

Slew Rate (Notes 4 and 6)

Slew Rate

T = 25°C

A

300

500

3500

10

V = ±15V, R = 750Ω, R = 750Ω, R = 400Ω

S

F

G

L

t

Rise Time (Notes 5 and 6)

Small-Signal Bandwidth

Small-Signal Rise Time

Propagation Delay

T = 25°C

A

20

r

BW

V = ±15V, R = 750Ω, R = 750Ω, R = 100Ω

S

100

3.5

MHz

ns

F

G

L

t

V = ±15V, R = 750Ω, R = 750Ω, R = 100Ω

S F G L

r

V = ±15V, R = 750Ω, R = 750Ω, R = 100Ω

S

3.5

ns

F

G

L

Small-Signal Overshoot

Settling Time

V = ±15V, R = 750Ω, R = 750Ω, R = 100Ω

S

15

%

F

G

L

0.1%, V

= 10V, R =1k, R = 1k, R =1k

45

ns

s

OUT

F

G

L

Differential Gain (Note 7)

Differential Phase (Note 7)

Differential Gain (Note 7)

Differential Phase (Note 7)

V = ±15V, R = 750Ω, R = 750Ω, R = 1k

0.01

0.04

0.1

%

S

F

G

L

V = ±15V, R = 750Ω, R = 750Ω, R = 1k

DEG

%

S

F

G

L

V = ±15V, R = 750Ω, R = 750Ω, R = 150Ω

S

F

G

L

V = ±15V, R = 750Ω, R = 750Ω, R = 150Ω

DEG

S

F

G

L

ELECTRICAL CHARACTERISTICS ^The

●

denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at T = 25°C. Transconductance Amplifier, Pins 1, 2, 3, 5. ± 5V ≤ V ≤ ±15V, I

=

A

S

SET

100µA, V = 0V unless otherwise noted.

CM

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

V

Input Offset Voltage

I

= 1mA, T = 25°C

±0.5

±5

mV

OS

SET

A

●

±10

Input Offset Voltage Drift

Input Offset Current

10

40

µV/°C

I

T = 25°C

200

500

nA

OS

A

●

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LT1228

ELECTRICAL CHARACTERISTICS ^The

●

denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at T = 25°C. Transconductance Amplifier, Pins 1, 2, 3, 5. ± 5V ≤ V ≤ ± 15V, I

=

A

S

SET

100µA, V = 0V unless otherwise noted.

CM

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

I

Input Bias Current

T = 25°C

0.4

1

5

µA

B

A

●

e

Input Noise Voltage Density

f = 1kHz

20

nV/√Hz

kΩ

n

R

Input Resistance-Differential Mode

Input Resistance-Common Mode

V

≈ ±30mV

IN

30

200

IN

V = ±15V, V = ±12V

●

50

1000

MΩ

S

CM

V = ±5V, V = ± 2V

S

CM

C

Input Capacitance

3

pF

IN

Input Voltage Range

V = ±15V, T = 25°C

S

±13

±12

±3

±14

V

S

A

V = ±15V

●

V = ±5V, T = 25°C

±4

S

A

V = ±5V

S

±2

CMRR

PSRR

Common Mode Rejection Ratio

Power Supply Rejection Ratio

V = ±15V, V = ±13V, T = 25°C

60

100

dB

S

CM

A

V = ±15V, V = ±12V

●

S

CM

V = ±5V, V = ±3V, T = 25°C

S

CM

A

V = ±5V, V = ±2V

S

CM

V = ±2V to ±15V, T = 25°C

60

100

dB

S

A

V = ±3V to ±15V

S

●

g

Transconductance

I

= 100µA, I

= ±30µA, T = 25°C

0.75

1.00

–0.33

100

1.25

130

µA/mV

%/°C

µA

m

SET

OUT

A

Transconductance Drift

Maximum Output Current

Output Leakage Current

●

I

= 100µA

70

OUT

OL

SET

= 0µA (+I of CFA), T = 25°C

0.3

3

10

µA

IN

A

●

V

Maximum Output Voltage Swing

Output Resistance

V = ±15V , R1 = ∞

S

●

±13

±14

±4

V

OUT

S

V = ±5V , R1 = ∞

±3

R

V = ±15V, V = ±13V

OUT

●

2

8

MΩ

O

S

V = ±5V, V

= ±3V

S

OUT

Output Capacitance (Note 3)

Supply Current, Both Amps

Total Harmonic Distortion

Small-Signal Bandwidth

Small-Signal Rise Time

Propagation Delay

V = ±5V

6

pF

mA

%

S

I

= 1mA

SET

●

9

15

S

THD

BW

V

= 30mV

at 1kHz, R1 = 100k

0.2

80

5

IN

RMS

R1 = 50Ω, I

= 500µA

MHz

ns

SET

t

= 500µA, 10% to 90%

= 500µA, 50% to 50%

r

5

ns

Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 5: Rise time is measured from 10% to 90% on a ±500mV output

signal while operating on ±15V supplies with R = 1k, R = 110Ω and

F

G

R = 100Ω. This condition is not the fastest possible, however, it does

L

guarantee the internal capacitances are correct and it makes automatic

testing practical.

Note 2: A heat sink may be required depending on the power supply

voltage.

Note 6: AC parameters are 100% tested on the ceramic and plastic DIP

packaged parts (J and N suffix) and are sample tested on every lot of

the SO packaged parts (S suffix).

Note 3: This is the total capacitance at Pin 1. It includes the input

capacitance of the current feedback amplifier and the output capacitance

of the transconductance amplifier.

Note 7: NTSC composite video with an output level of 2V.

Note 4: Slew rate is measured at ±5V on a ±10V output signal while

operating on ±15V supplies with R = 1k, R = 110Ω and R = 400Ω. The

Note 8: Back to back 6V Zener diodes are connected between Pins 2

and 3 for ESD protection.

F

G

L

slew rate is much higher when the input is overdriven, see the applications

section.

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LT1228

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TYPICAL PERFOR A CE CHARACTERISTICS ^{Transconductance Amplifier, Pins 1, 2, 3 & 5}

Small-Signal Bandwidth vs

Set Current

Small-Signal Transconductance

and Set Current vs Bias Voltage

Small-Signal Transconductance

vs DC Input Voltage

100

10

1

100

10

10000

1000

100

10

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

R1 = 100Ω

V

= ±15V

V

= ±2V TO ±15V

= 25°C

S

A

V

SET

= ±2V TO ±15V

S

T

I

= 100µA

R1 = 1k

–55°C

1

R1 = 10k

25°C

0.1

125°C

0.01

0.001

1.0

R1 = 100k

0.2

0

0.1

1.5

10

100

SET CURRENT (µA)

1000

0.9

1.0

1.1

1.2

1.3

1.4

–200 –150 –100 –50

0

50 100 150 200

BIAS VOLTAGE, PIN 5 TO 4, (V)

INPUT VOLTAGE (mVDC)

LT1228 • TPC01

LT1228 • TPC02

LT1228 • TPC03

Total Harmonic Distortion vs

Input Voltage

Spot Output Noise Current vs

Frequency

Input Common Mode Limit vs

Temperature

+

V

10

1

1000

100

10

V

= ±2V TO ±15V

= 25°C

+

V

= ±15V

S

A

V

= 2V TO 15V

S

–0.5

–1.0

–1.5

–2.0

T

I

= 1mA

SET

I

= 100µA

SET

2.0

1.5

1.0

0.5

0.1

0.01

–

V

= –2V TO –15V

I

= 100µA

SET

I

= 1mA

10

SET

–

V

–50 –25

0

25

50

75 100 125

1

100

1000

10

100

1k

FREQUENCY (Hz)

10k

100k

TEMPERATURE (°C)

INPUT VOLTAGE (mV

)

LT1228 • TPC04

P–P

LT1228 • TPC06

LT1228 • TPC05

Small-Signal Control Path

Bandwidth vs Set Current

Small-Signal Control Path

Gain vs Input Voltage

Output Saturation Voltage vs

Temperature

+

V

100

10

1

1.0

0.9

0.8

0.7

0.6

0.5

0.4

0.3

0.2

0.1

0

V

= ±2V TO ±15V

S

= 200mV

IN

–0.5

–1.0

(PIN 2 TO 3)

±2V ≤ V ≤ ±15V

S

∆I

R1 =

∞

OUT

∆I

SET

+1.0

+0.5

–

V

–50 –25

0

25

50

75 100 125

10

100

SET CURRENT (µA)

1000

0

40

80

120

160

200

TEMPERATURE (°C)

INPUT VOLTAGE, PIN 2 TO 3, (mVDC)

LT1228 • TPC09

LT1228 • TPC07

LT1228 • TPC08

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LT1228

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TYPICAL PERFOR A CE CHARACTERISTICS_{Current Feedback Amplifier, Pins 1, 6, 8}

Voltage Gain and Phase vs

Frequency, Gain = 6dB

–3dB Bandwidth vs Supply

Voltage, Gain = 2, R = 1k

Voltage, Gain = 2, R = 100Ω

L

8

7

6

0

180

160

140

180

160

140

PHASE

GAIN

PEAKING ≤ 0.5dB

PEAKING ≤ 5dB

45

90

R

= 500Ω

F

R

= 500Ω

R

= 750Ω

F

135

180

225

F

5

4

120

100

80

60

40

20

0

120

100

80

60

40

20

0

= 750Ω

3

2

1

R

= 1k

F

PEAKING ≤ 0.5dB

PEAKING ≤ 5dB

V

R

= ±15V

= 100Ω

= 750Ω

S

L

F

0

–1

–2

R

= 2k

R

= 1k

6

F

R

= 2k

F

0.1

1

10

100

0

2

4

6

8

10 12 14 16 18

0

2

4

8

10 12 14 16 18

FREQUENCY (MHz)

SUPPLY VOLTAGE (±V)

LT1228 • TPC10

LT1228 • TPC11

LT1228 • TPC12

Voltage Gain and Phase vs

Frequency, Gain = 20dB

–3dB Bandwidth vs Supply

Voltage, Gain = 10, R = 100Ω

Voltage, Gain = 10, R = 1kΩ

L

22

21

20

0

180

160

140

180

160

140

PHASE

PEAKING ≤ 0.5dB

PEAKING ≤ 5dB

PEAKING ≤ 0.5dB

PEAKING ≤ 5dB

45

90

GAIN

135

180

225

19

18

120

100

80

60

40

20

0

120

100

80

60

40

20

0

R

= 250Ω

R = 500Ω

F

R

= 250Ω

F

17

16

15

R

= 500Ω

R

= 750Ω

F

R

= 750Ω

F

R

= 1k

F

V

R

= ±15V

= 100Ω

= 750Ω

S

L

F

R

= 1k

= 2k

F

14

13

12

R

= 2k

F

R

F

2

4

6

8

10 12 14 16 18

0.1

1

10

100

0

2

4

6

8

10 12 14 16 18

SUPPLY VOLTAGE (±V)

FREQUENCY (MHz)

SUPPLY VOLTAGE (±V)

LT1228 • TPC13

LT1228 • TPC15

LT1228 • TPC14

Voltage Gain and Phase vs

Frequency, Gain = 40dB

–3dB Bandwidth vs Supply

Voltage, Gain = 100, R = 100Ω

–3dB Bandwidth vs Supply

Voltage, Gain = 100, R = 1kΩ

L

18

16

14

42

41

40

0

18

16

14

PHASE

45

90

R

= 500Ω

GAIN

F

135

180

225

39

38

12

10

8

12

10

8

R

= 500Ω

F

R

= 1k

F

R

= 1k

= 2k

37

36

35

F

R

= 2k

F

6

R

6

F

V

R

= ±15V

= 100Ω

= 750Ω

S

L

F

4

34

33

32

2

0

2

4

6

8

10 12 14 16 18

0.1

1

10

100

2

4

6

8

10 12 14 16 18

SUPPLY VOLTAGE (±V)

FREQUENCY (MHz)

SUPPLY VOLTAGE (±V)

LT1228 • TPC18

LT1228 • TPC17

LT1228 • TPC16

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6

LT1228

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TYPICAL PERFOR A CE CHARACTERISTICS ^{Current Feedback Amplifier, Pins 1, 6, 8}

Maximum Capacitive Load vs

Feedback Resistor

Total Harmonic Distortion vs

Frequency

2nd and 3rd Harmonic

Distortion vs Frequency

10k

0.10

–20

–30

V

R

A

= ±15V

S

O

L

F

V

= ±15V

= 400Ω

S

L

= 2V

P–P

R

= 100Ω

= 750Ω

= 10dB

V

= ±5V

S

R = R = 750Ω

F

G

1k

2nd

V

–40

–50

–60

–70

V

= ±15V

S

100

10

1

0.01

V

= 7V

= 1V

3rd

O

RMS

R

= 1k

L

PEAKING ≤ 5dB

GAIN = 2

0.001

0

1

2

3

10

100

1k

FREQUENCY (Hz)

10k

100k

1

10

FREQUENCY (MHz)

100

FEEDBACK RESISTOR (kΩ)

LT1228 • TPC19

LT1228 • TPC20

LT1228 • TPC21

Input Common Mode Limit vs

Temperature

Output Saturation Voltage vs

Temperature

Output Short-Circuit Current vs

Temperature

+

V

70

60

50

40

30

–0.5

–1.0

–1.5

–2.0

–0.5

–1.0

+

V

= 2V TO 15V

R

=

∞

L

± 2V ≤ V ≤ ±15V

S

2.0

1.5

1.0

0.5

–

= –2V TO –15V

1.0

0.5

–

V

–50 –25

0

25

50

75 100 125

–50 –25

0

25

50

75 100 125

–50 –25

0

25 50 75 100 125 150 175

TEMPERATURE (°C)

LT1228 • TPC22

LT1228 • TPC23

LT1228 • TPC24

Spot Noise Voltage and Current vs

Frequency

Power Supply Rejection vs

Frequency

Output Impedance vs

Frequency

100

80

60

100

10

V

= ±15V

S

V

R

= ±15V

= 100Ω

= R = 750Ω

S

L

F

G

POSITIVE

1.0

–i

n

R = R = 2k

F

G

10

40

20

0

R = R = 750Ω

F

G

e

0.1

0.01

n

NEGATIVE

+i

n

1

0.001

10

100

1k

FREQUENCY (Hz)

10k

100k

10k

100k

1M

10M

100M

10k

100k

1M

10M

100M

FREQUENCY (Hz)

LT1228 • TPC25

LT1228 • TPC26

LT1228 • TPC27

1228fc

7

LT1228

TYPICAL PERFOR A CE CHARACTERISTICS _{Current Feedback Amplifier, Pins 1, 6 & 8}

U W

Settling Time to 10mV vs

Output Step

Settling Time to 1mV vs

Output Step

Supply Current vs Supply Voltage

10

8

10

8

10

9

NONINVERTING

INVERTING

6

8

–55°C

INVERTING

4

7

2

25°C

6

V

= ±15V

= R = 1k

G

V

= ±15V

= R = 1k

G

S

F

S

F

0

5

R

125°C

R

–2

–4

–2

–4

4

3

2

1

0

175°C

INVERTING

–6

NONINVERTING

–8

NONINVERTING

20

INVERTING

60 80

–10

0

4

8

12

16

20

0

40

100

0

2

4

6

8

10 12 14 16 18

SETTLING TIME (µs)

SETTLING TIME (ns)

SUPPLY VOLTAGE (±V)

LT1228 • TPC29

LT1228 • TPC28

LT1228 • TPC30

W

SI PLIFIED SCHE ATIC

+

V

7

6

4

BIAS

+IN

3

–IN

2

I

OUT

1

8

GAIN

OUT

I

SET

5

–

LT1228 • TA03

1228fc

8

LT1228

O U

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PPLICATI

A

S I FOR ATIO

The LT1228 contains two amplifiers, a transconductance

amplifier (voltage-to-current) and a current feedback am-

plifier (voltage-to-voltage). The gain of the transconduc-

tance amplifier is proportional to the current that is exter-

nallyprogrammedintoPin5. Bothamplifiersaredesigned

to operate on almost any available supply voltage from 4V

(±2V) to 30V (±15V). The output of the transconductance

amplifier is connected to the noninverting input of the

current feedback amplifier so that both fit into an eight pin

package.

Resistance Controlled Gain

If the set current is to be set or varied with a resistor or

potentiometer it is possible to use the negative tempera-

ture coefficient at Pin 5 (with respect to Pin 4) to compen-

sateforthenegativetemperaturecoefficientofthetranscon-

ductance. The easiest way is to use an LT1004-2.5, a 2.5V

reference diode, as shown below:

Temperature Compensation of g with a 2.5V Reference

m

R

TRANSCONDUCTANCE AMPLIFIER

I

SET

g

m

I

V

TheLT1228transconductanceamplifierhasahighimped-

ance differential input (Pins 2 and 3) and a current source

output (Pin 1) with wide output voltage compliance. The

voltage to current gain or transconductance (g_m) is set by

the current that flows into Pin 5, I_SET. The voltage at Pin 5

is two forward biased diode drops above the negative

supply, Pin 4. Therefore the voltage at Pin 5 (with

respect to V^–) is about 1.2V and changes with the log of

the set current (120mV/decade), see the characteristic

curves. The temperature coefficient of this voltage is

about –4mV/°C (–3300ppm/°C) and the temperature co-

efficient of the logging characteristic is 3300ppm/°C. It is

important that the current into Pin 5 be limited to less than

15mA. THE LT1228 WILL BE DESTROYED IF PIN 5 IS

SHORTED TO GROUND OR TO THE POSITIVE SUPPLY. A

limiting resistor (2k or so) should be used to prevent more

than 15mA from flowing into Pin 5.

be

4

2.5V

2E

g

5

R

SET

be

LT1004-2.5

–

V

LT1228 ¥ TA04

The current flowing into Pin 5 has a positive temperature

coefficient that cancels the negative coefficient of the

transconductance. The following derivation shows why a

2.5V reference results in zero gain change with tempera-

ture:

I_SET

q

Since g_m=

×

= 10 •I_SET

kT 3.87

akT

n

⎛

⎞

cT

and V_be= E_g–

where a = In

⎜

⎟

q

Ic

⎝

⎠

The small-signal transconductance (g_m) is equal to ten

times the value of I_SET(in mA/mV) and this relationship

holdsovermanydecadesofsetcurrent(seethecharacter-

istic curves). The transconductance is inversely propor-

tional to absolute temperature (–3300ppm/°C). The input

stage of the transconductance amplifier has been de-

signedtooperatewithmuchlargersignalsthanispossible

with an ordinary diff-amp. The transconductance of the

input stage varies much less than 1% for differential input

signals over a ±30 mV range (see the characteristic curve

Small-Signal Transconductance vs DC Input Voltage).

≈ 19.4 at 27°C c = 0.001, n = 3,Ic = 100µA

(

)

E_gis about 1.25V so the 2.5V reference is 2E_g. Solving

the loop for the set current gives:

⎛

⎞

akT

q

2E_g– 2 E_g–

⎜

⎟

⎝

⎠

2akT

Rq

I_SET

=

orI_SET=

R

1228fc

9

LT1228

PPLICATI

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W

U

A

S I FOR ATIO

Substitutingintothe equationfortransconductancegives:

tance amplifier requires converting the voltage into a

current that flows into Pin 5. Because the voltage at Pin 5

is two diode drops above the negative supply, a single

resistor from the control voltage source to Pin 5 will

suffice in many applications. The control voltage is refer-

enced to the negative supply and has an offset of about

900mV. The conversion will be monotonic, but the linear-

ity is determined by the change in the voltage at Pin 5

(120mV per decade of current). The characteristic is very

repeatable since the voltage at Pin 5 will vary less than

±5% from part to part. The voltage at Pin 5 also has a

negative temperature coefficient as described in the pre-

vious section. When the gain of several LT1228s are to be

varied together, the current can be split equally by using

equal value resistors to each Pin 5.

a

10

R

g_m=

=

1.94R

The temperature variation in the term “a” can be ignored

since it is much less than that of the term “T” in the

equation for V_be. Using a 2.5V source this way will main-

tain the gain constant within 1% over the full temperature

range of –55°C to 125°C. If the 2.5V source is off by 10%,

the gain will vary only about ±6% over the same tempera-

ture range.

We can also temperature compensate the transconduc-

tance without using a 2.5V reference if the negative power

supply is regulated. A Thevenin equivalent of 2.5V is

generated from two resistors to replace the reference. The

two resistors also determine the maximum set current,

approximately 1.1V/R_TH. By rearranging the Thevenin

equations to solve for R4 and R6 we get the following

equations in terms of R_THand the negative supply, V_EE.

For more accurate (and linear) control, a voltage-to-

current converter circuit using one op amp can be used.

The following circuit has several advantages. The input no

longer has to be referenced to the negative supply and the

input can be either polarity (or differential). This circuit

works on both single and split supplies since the input

voltage and the Pin 5 voltage are independent of each

other. The temperature coefficient of the output current is

set by R5.

R_TH

R_THV_EE

2.5V

R4 =

and R6 =

⎛

⎞

2.5V

1–

⎜

⎟

V_EE

⎝

⎠

R3

1M

Temperature Compensation of g with a Thevenin Voltage

m

R1

1M

1.03k

R'

R5

1k

V1

V2

+

–

I

OUT

SET

R2

1M

LT1006

50pF

TO PIN 5

OF LT1228

g

m

V

be

R4

1M

4

R6

V

= 2.5V

TH

6.19kΩ

5

I

R'

SET

be

R1 = R2

R3 = R4

R4

1.24kΩ

(V1 – V2) R3

I

=

•

= 1mA/V

–15V

OUT

R5

R1

LT1228 ¥ TA19

LT1228 ¥ TA05

Voltage Controlled Gain

Digital control of the transconductance amplifier gain is

donebyconvertingtheoutputofaDACtoacurrentflowing

into Pin 5. Unfortunately most current output DACs

sink rather than source current and do not have output

To use a voltage to control the gain of the transconduc-

1228fc

10

LT1228

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PPLICATI

A

S I FOR ATIO

Transconductance Amp Small-Signal Response

= 500µA, R1 = 50Ω

compliance compatible with Pin 5 of the LT1228. There-

fore, the easiest way to digitally control the set current is

to use a voltage output DAC and a voltage-to-current

circuit.Thepreviousvoltage-to-currentconverterwilltake

the output of any voltage output DAC and drive Pin 5 with

a proportional current. The R, 2R CMOS multiplying DACs

operatinginthevoltageswitchingmodeworkwellonboth

single and split supplies with the above circuit.

I

SET

Logarithmic control is often easier to use than linear

control. A simple circuit that doubles the set current for

each additional volt of input is shown in the voltage

controlled state variable filter application near the end of

this data sheet.

Transconductance Amplifier Frequency Response

CURRENT FEEDBACK AMPLIFIER

The bandwidth of the transconductance amplifier is a

function of the set current as shown in the characteristic

curves. At set currents below 100µA, the bandwidth is

approximately:

The LT1228 current feedback amplifier has very high

noninverting input impedance and is therefore an excel-

lent buffer for the output of the transconductance ampli-

fier. The noninverting input is at Pin 1, the inverting input

at Pin 8 and the output at Pin 6. The current feedback

amplifier maintains its wide bandwidth for almost all

voltage gains making it easy to interface the output levels

of the transconductance amplifier to other circuitry. The

current feedback amplifier is designed to drive low imped-

ance loads such as cables with excellent linearity at high

frequencies.

–3dB bandwidth = 3 • 10¹¹I_SET

The peak bandwidth is about 80MHz at 500µA. When a

resistor is used to convert the output current to a voltage,

the capacitance at the output forms a pole with the

resistor. The best case output capacitance is about 5pF

with ±15V supplies and 6pF with ±5V supplies. You must

add any PC board or socket capacitance to these values to

get the total output capacitance. When using a 1k resistor

at the output of the transconductance amp, the output

capacitance limits the bandwidth to about 25MHz.

Feedback Resistor Selection

The small-signal bandwidth of the LT1228 current feed-

backamplifierissetbytheexternalfeedbackresistorsand

the internal junction capacitors. As a result, the bandwidth

is a function of the supply voltage, the value of the

feedback resistor, the closed-loop gain and load resistor.

The characteristic curves of bandwidth versus supply

voltage are done with a heavy load (100Ω) and a light load

(1k) to show the effect of loading. These graphs also show

The output slew rate of the transconductance amplifier is

the set current divided by the output capacitance, which is

6pF plus board and socket capacitance. For example with

the set current at 1mA, the slew rate would be over

100V/µs.

1228fc

11

LT1228

U

W U U

APPLICATIO S I FOR ATIO

the family of curves that result from various values of the

feedback resistor. These curves use a solid line when the

response has less than 0.5dB of peaking and a dashed line

for the response with 0.5dB to 5dB of peaking. The curves

stop where the response has more than 5dB of peaking.

Capacitance on the Inverting Input

Current feedback amplifiers want resistive feedback from

the output to the inverting input for stable operation. Take

care to minimize the stray capacitance between the output

and the inverting input. Capacitance on the inverting input

to ground will cause peaking in the frequency response

(and overshoot in the transient response), but it does not

degrade the stability of the amplifier. The amount of

capacitance that is necessary to cause peaking is a func-

tion of the closed-loop gain taken. The higher the gain, the

more capacitance is required to cause peaking. For ex-

ample, in a gain of 100 application, the bandwidth can be

increased from 10MHz to 17MHz by adding a 2200pF

capacitor, as shown below. C_Gmust have very low series

resistance, such as silver mica.

Current Feedback Amp Small-Signal Response

V = ±15V, R = R = 750Ω, R = 100Ω

S

F

G

L

1

+

V

IN

6

CFA

V

OUT

8

–

R

F

510Ω

At a gain of two, on ±15V supplies with a 750Ω feedback

resistor, the bandwidth into a light load is over 160MHz

without peaking, but into a heavy load the bandwidth

reduces to 100MHz. The loading has so much effect

because there is a mild resonance in the output stage that

enhances the bandwidth at light loads but has its Q

reduced by the heavy load. This enhancement is only

usefulatlowgainsettings, atagainoftenitdoesnotboost

the bandwidth. At unity gain, the enhancement is so

effective the value of the feedback resistor has very little

effect on the bandwidth. At very high closed-loop gains,

the bandwidth is limited by the gain-bandwidth product of

about 1GHz. The curves show that the bandwidth at a

closed-loop gain of 100 is 10MHz, only one tenth what it

is at a gain of two.

R

G

5.1Ω

C

G

LT1228 • TA08

Boosting Bandwidth of High Gain Amplifier

with Capacitance On Inverting Input

49

46

C

= 4700pF

G

43

40

37

34

31

28

25

22

19

C

= 2200pF

G

C

= 0

G

1

10

FREQUENCY (MHz)

100

LT1228 • TA09

1228fc

12

LT1228

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PPLICATI

S I FOR ATIO

A

Capacitive Loads

The output slew rate is set by the value of the feedback

resistors and the internal capacitance. At a gain of ten with

a 1k feedback resistor and ±15V supplies, the output slew

rate is typically 500V/µs and –850V/µs. There is no input

stage enhancement because of the high gain. Larger

feedback resistors will reduce the slew rate as will lower

supply voltages, similar to the way the bandwidth is

reduced.

The LT1228 current feedback amplifier can drive capaci-

tive loads directly when the proper value of feedback

resistor is used. The graph of Maximum Capacitive Load

vs Feedback Resistor should be used to select the appro-

priate value. The value shown is for 5dB peaking when

driving a 1k load, at a gain of 2. This is a worst case

condition, the amplifier is more stable at higher gains, and

driving heavier loads. Alternatively, a small resistor (10Ω

to 20Ω) can be put in series with the output to isolate the

capacitive load from the amplifier output. This has the

advantage that the amplifier bandwidth is only reduced

when the capacitive load is present and the disadvantage

that the gain is a function of the load resistance.

Current Feedback Amp Large-Signal Response

V = ±15V, R = 1k, R = 110Ω, R = 400Ω

S

F

G

L

Slew Rate

The slew rate of the current feedback amplifier is not

independent of the amplifier gain configuration the way it

is in a traditional op amp. This is because the input stage

and the output stage both have slew rate limitations. The

inputstageoftheLT1228currentfeedbackamplifierslews

at about 100V/µs before it becomes nonlinear. Faster

input signals will turn on the normally reverse biased

emittersontheinputtransistorsandenhancetheslewrate

significantly. This enhanced slew rate can be as much as

3500V/µs!

Settling Time

The characteristic curves show that the LT1228 current

feedback amplifier settles to within 10mV of final value in

40ns to 55ns for any output step less than 10V. The curve

ofsettlingto1mVoffinalvalueshowsthatthereisaslower

thermal contribution up to 20µs. The thermal settling

component comes from the output and the input stage.

Theoutputcontributesjustunder1mV/Vofoutputchange

and the input contributes 300µV/V of input change.

Fortunately the input thermal tends to cancel the output

thermal. For this reason the noninverting gain of two

configuration settles faster than the inverting gain of one.

Current Feedback Amp Large-Signal Response

V = ±15V, R = R = 750ΩSlew Rate Enhanced

S

F

G

1228fc

13

LT1228

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APPLICATIO S I FOR ATIO

Power Supplies

For example, let’s calculate the worst case power dissipa-

tion in a variable gain video cable driver operating on ±12V

supplies that delivers a maximum of 2V into 150Ω. The

maximum set current is 1mA.

The LT1228 amplifiers will operate from single or split

supplies from ±2V (4V total) to ±18V (36V total). It is not

necessary to use equal value split supplies, however the

offset voltage and inverting input bias current of the

current feedback amplifier will degrade. The offset voltage

changesabout350µV/Vofsupplymismatch,theinverting

bias current changes about 2.5µA/V of supply mismatch.

V_OMAX

P_D= 2V_SI_SMAX+ 3.5I_SET+ V_S– V_OMAX

(

) (

)

R_L

2V

P_D= 2 •12V • 7mA + 3.5 •1mA + 12V – 2V

(

)

]

(

)

[

150Ω

Power Dissipation

= 0.252 + 0.133 = 0.385W

The total power dissipation times the thermal resistance of

the package gives the temperature rise of the die above

ambient. The above example in SO-8 surface mount pack-

age (thermal resistance is 150°C/W) gives:

The worst case amplifier power dissipation is the total of

the quiescent current times the total power supply voltage

plus the power in the IC due to the load. The quiescent

supply current of the LT1228 transconductance amplifier

isequalto3.5timesthesetcurrentatalltemperatures.The

quiescent supply current of the LT1228 current feedback

amplifier has a strong negative temperature coefficient

and at 150°C is less than 7mA, typically only 4.5mA. The

power in the IC due to the load is a function of the output

voltage, the supply voltage and load resistance. The worst

case occurs when the output voltage is at half supply, if it

can go that far, or its maximum value if it cannot reach half

supply.

Temperature Rise = P_Dθ_JA= 0.385W • 150°C/W

= 57.75°C

Therefore the maximum junction temperature is 70°C

+57.75°C or 127.75°C, well under the absolute maximum

junction temperature for plastic packages of 150°C.

U

TYPICAL APPLICATIO S

Basic Gain Control

g_m•R1= 10 •I_SET•R1

The basic gain controlled amplifier is shown on the front

page of the data sheet. The gain is directly proportional to

the set current. The signal passes through three stages

from the input to the output.

Lastly the signal is buffered and amplified by the current

feedback amplifier (CFA). The voltage gain of the current

feedback amplifier is:

R_F

1+

First the input signal is attenuated to match the dynamic

range of the transconductance amplifier. The attenuator

should reduce the signal down to less than 100mV peak.

The characteristic curves can be used to estimate how

muchdistortiontherewillbeatmaximuminputsignal. For

single ended inputs eliminate R2A or R3A.

R_G

The overall gain of the gain controlled amplifier is the

product of all three stages:

⎛

⎞

⎛

⎞

R_F

R3

R3 + R3A

A _V=

•10 •I_SET•R1• 1+

⎜

⎟

⎜

⎟

R_G

⎝

⎠

⎝

⎠

The signal is then amplified by the transconductance

amplifier (g_m) and referred to ground. The voltage gain of

the transconductance amplifier is:

More than one output can be summed into R1 because the

output of the transconductance amplifier is a current. This

is the simplest way to make a video mixer.

1228fc

14

LT1228

U

TYPICAL APPLICATIO S

Video Fader

Video DC Restore (Clamp) Circuit

NOT NECESSARY IF THE SOURCE RESISTANCE IS LESS THAN 50Ω

1k

3

2

V

+

–

IN1

200Ω

1

+

V

g

1000pF

m

+

–

LT1223

CFA

3

2

7

V

OUT

+

5

1

8

g

+

–

m

6

1k

100Ω

V

0.01µF

–

CFA

R

5

OUT

4

10k

5.1k

F

–

V

= ±5V

10k

5.1k

–5V

1k

S

R

10k

G

5V

3k

VIDEO

INPUT

1k

3

2

5

V

+

IN2

1

g

m

100Ω

LOGIC

INPUT

2N3906

–

LT1228 • TA12

RESTORE

LT1228 • TA13

The video fader uses the transconductance amplifiers

fromtwoLT1228sinthefeedbackloopofanothercurrent

feedback amplifier, the LT1223. The amount of signal

from each input at the output is set by the ratio of the

set currents of the two LT1228s, not by their absolute

value. The bandwidth of the current feedback amplifier

is inversely proportional to the set current in this

configuration. Therefore, the set currents remain high

over most of the pot’s range, keeping the bandwidth over

15MHz even when the signal is attenuated 20dB. The pot

is set up to completely turn off one LT1228 at each end of

the rotation.

The video restore (clamp) circuit restores the black level of

the composite video to zero volts at the beginning of every

line. This is necessary because AC coupled video changes

DC level as a function of the average brightness of the

picture. DC restoration also rejects low frequency noise

such as hum.

The circuit has two inputs: composite video and a logic

signal.Thelogicsignalishighexceptduringthebackporch

time right after the horizontal sync pulse. While the logic is

high, the PNP is off and I_SETis zero. With I_SETequal to zero

the feedback to Pin 2 has no affect. The video input drives

the noninverting input of the current feedback amplifier

whose gain is set by R_Fand R_G. When the logic signal is

low,thePNPturnsonandI_SETgoestoabout1mA.Thenthe

transconductance amplifier charges the capacitor to force

the output to match the voltage at Pin 3, in this case zero

volts.

This circuit can be modified so that the video is DC coupled

by operating the amplifier in an inverting configuration.

Just ground the video input shown and connect R_Gto the

video input instead of to ground.

1228fc

15

LT1228

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TYPICAL APPLICATIO S

Single Supply Wien Bridge Oscillator

resistorandthetransconductanceamplifiermustbeabout

11, resulting in a set current of about 600µA at oscillation.

At start-up there is no set current and therefore no attenu-

ation for a net gain of about 11 around the loop. As the

output oscillation builds up it turns on the PNP transistor

which generates the set current to regulate the output

voltage.

100Ω

2N3906

+

V

6V TO 30V

+

V

+

470Ω

10µF

10kΩ

7

3

2

+

–

5

1

g

12MHz Negative Resistance LC Oscillator

+

m

0.1µF

51Ω

6

CFA

R

8

4

+

V

–

9.1k

F

3

2

7

+

–

680Ω

V

O

1

8

g

+

–

m

1k

R

G

20Ω

V

51Ω

O

1.8k

6

5

CFA

+

4

160Ω

1000pF

10µF

1000pF

+

–

50Ω

V

750Ω

10µF

1k

160Ω

50Ω

4.7µH

30pF

4.3k

330Ω

2N3906

f = 1MHz

= 6dBm (450mV

V

)

RMS

O

2nd HARMONIC = –38dBc

3rd HARMONIC = –54 dBc

2N3904

LT1228 • TA14

FOR 5V OPERATION SHORT OUT 100Ω RESISTOR

0.1µF

10k

In this application the LT1228 is biased for operation from

a single supply. An artificial signal ground at half supply

voltage is generated with two 10k resistors and bypassed

withacapacitor.AcapacitorisusedinserieswithR_Gtoset

the DC gain of the current feedback amplifier to unity.

–

V

= 10dB

O

AT V = ±5V ALL HARMONICS 40dB DOWN

AT V = ±12V ALL HARMONICS 50dB DOWN

S

LT1228 • TA15

This oscillator uses the transconductance amplifier as a

negative resistor to cause oscillation. A negative resistor

results when the positive input of the transconductance

amplifier is driven and the output is returned to it. In this

example a voltage divider is used to lower the signal level

at the positive input for less distortion. The negative

resistor will not DC bias correctly unless the output of the

transconductance amplifier drives a very low resistance.

HereitseesaninductortogroundsothegainatDCiszero.

The oscillator needs negative resistance to start and that

isprovidedbythe4.3kresistortoPin5. Astheoutputlevel

rises it turns on the PNP transistor and in turn the NPN

which steals current from the transconductance amplifier

bias input.

The transconductance amplifier is used as a variable

resistor to control gain. A variable resistor is formed by

driving the inverting input and connecting the output back

to it. The equivalent resistor value is the inverse of the g_m.

This works with the 1.8k resistor to make a variable

attenuator. The 1MHz oscillation frequency is set by the

Wien bridge network made up of two 1000pF capacitors

and two 160Ω resistors.

For clean sine wave oscillation, the circuit needs a net gain

ofonearoundtheloop. Thecurrentfeedbackamplifierhas

a gain of 34 to keep the voltage at the transconductance

amplifier input low. The Wien bridge has an attenuation of

3 at resonance; therefore the attenuation of the 1.8k

1228fc

16

LT1228

U

TYPICAL APPLICATIO S

Filters

Single Pole Low/High/Allpass Filter

R3A

1k

V

IN

3

LOWPASS

INPUT

+

1

g

+

m

R3

120Ω

2

C

6

V

–

CFA

R

OUT

330pF

5

8

–

I

SET

R

G

F

V

1k

IN

HIGHPASS

INPUT

R2A

1k

10

2π

9

I

R + 1

R2

R2 + R2A

FOR THE VALUES SHOWN

R2

120Ω

SET

C

F

f

=

×

C

R

G

LT1228 • TA16

f

= 10

I

C

SET

Allpass Filter Phase Response

0

–45

1mA SET CURRENT

–90

–135

–180

100µA SET CURRENT

10k

100k

1M

10M

FREQUENCY (Hz)

LT1228 • TA17

Using the variable transconductance of the LT1228 to

make variable filters is easy and predictable. The most

straight forward way is to make an integrator by putting a

capacitor at the output of the transconductance amp and

buffering it with the current feedback amplifier. Because

the input bias current of the current feedback amplifier

must be supplied by the transconductance amplifier, the

setcurrentshouldnotbeoperatedbelow10µA.Thislimits

the filters to about a 100:1 tuning range.

values shown give a 100kHz corner frequency for 100µA

set current. The circuit has two inputs, a lowpass filter

input and a highpass filter input. To make a lowpass filter,

ground the highpass input and drive the lowpass input.

Conversely for a highpass filter, ground the lowpass input

and drive the highpass input. If both inputs are driven, the

result is an allpass filter or phase shifter. The allpass has

flat amplitude response and 0° phase shift at low frequen-

cies, going to –180° at high frequencies. The allpass filter

has –90° phase shift at the corner frequency.

The Single Pole circuit realizes a single pole filter with a

corner frequency (f_C) proportional to the set current. The

1228fc

17

LT1228

U

TYPICAL APPLICATIO S

Voltage Controlled State Variable Filter

+

–

1k

LT1006

10k

2N3906

V

C

100pF

180Ω

51k

5V

3k

–5V

3k

7

3.3k

100Ω

3

2

+

V

5

IN

1

8

g

+

m

6

BANDPASS

OUTPUT

–

CFA

4

18pF

–

–5V

1k

3.3k

5

100Ω

5V

7

3

2

+

–

1

8

g

100Ω

+

m

LOWPASS

OUTPUT

6

CFA

4

18pF

3.3k

–

–5V

1k

f

= 100kHz AT V = 0V

C

O

= 200kHz AT V = 1V

C

= 400kHz AT V = 2V

C

= 800kHz AT V = 3V

C

= 1.6MHz AT V = 4V

LT1228 • TA18

C

The state variable filter has both lowpass and bandpass

outputs. Each LT1228 is configured as a variable integra-

tor whose frequency is set by the attenuators, the capaci-

torsandthesetcurrent. Becausetheintegratorshaveboth

positive and negative inputs, the additional op amp nor-

mally required is not needed. The input attenuators set the

circuit up to handle 3V_P–Psignals.

best accuracy. If discrete transistors are used, the 51k

resistor should be trimmed to give proper frequency

response with V_Cequal zero. The circuit generates 100µA

for V_Cequal zero volts and doubles the current for every

additional volt. The two 3k resistors divide the current

between the two LT1228s. Therefore the set current of

each amplifier goes from 50µA to 800µA for a control

voltage of 0V to 4V. The resulting filter is at 100kHz for V_C

equal zero, and changes it one octave/V of control input.

Thesetcurrentisgeneratedwithasimplecircuitthatgives

logarithmic voltage to current control. The two PNP tran-

sistors should be a matched pair in the same package for

1228fc

18

LT1228

U

PACKAGE DESCRIPTIO

J8 Package

8-Lead CERDIP (Narrow .300 Inch, Hermetic)

(Reference LTC DWG # 05-08-1110)

.405

(10.287)

MAX

CORNER LEADS OPTION

(4 PLCS)

.005

(0.127)

MIN

.200

(5.080)

MAX

.300 BSC

(7.62 BSC)

6

5

4

8

7

.023 – .045

(0.584 – 1.143)

HALF LEAD

OPTION

.015 – .060

(0.381 – 1.524)

.025

.220 – .310

(5.588 – 7.874)

.045 – .068

(0.635)

RAD TYP

(1.143 – 1.650)

FULL LEAD

OPTION

.008 – .018

(0.203 – 0.457)

0° – 15°

1

2

3

.045 – .065

(1.143 – 1.651)

.125

3.175

MIN

NOTE: LEAD DIMENSIONS APPLY TO SOLDER DIP/PLATE

OR TIN PLATE LEADS

.014 – .026

(0.360 – 0.660)

.100

(2.54)

BSC

J8 0801

OBSOLETE PACKAGE

N8 Package

8-Lead PDIP (Narrow .300 Inch)

(Reference LTC DWG # 05-08-1510)

.400*

(10.160)

MAX

.130 ± .005

.300 – .325

.045 – .065

(3.302 ± 0.127)

(1.143 – 1.651)

(7.620 – 8.255)

8

1

7

6

3

5

4

.065

(1.651)

TYP

.255 ± .015*

(6.477 ± 0.381)

.008 – .015

(0.203 – 0.381)

.120

.020

(0.508)

MIN

(3.048)

MIN

+.035

–.015

2

.325

.018 ± .003

.100

(2.54)

BSC

+0.889

8.255

(0.457 ± 0.076)

(

)

N8 1002

–0.381

NOTE:

INCHES

1. DIMENSIONS ARE

MILLIMETERS

*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.

MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .010 INCH (0.254mm)

S8 Package

8-Lead Plastic Small Outline (Narrow .150 Inch)

(Reference LTC DWG # 05-08-1610)

.189 – .197

(4.801 – 5.004)

NOTE 3

.010 – .020

(0.254 – 0.508)

7

5

8

6

× 45°

.053 – .069

(1.346 – 1.752)

.045 ±.005

.160 ±.005

.050 BSC

.004 – .010

(0.101 – 0.254)

.008 – .010

0°– 8° TYP

(0.203 – 0.254)

.150 – .157

(3.810 – 3.988)

NOTE 3

.228 – .244

(5.791 – 6.197)

.016 – .050

(0.406 – 1.270)

.245

MIN

.050

(1.270)

BSC

.014 – .019

(0.355 – 0.483)

TYP

NOTE:

INCHES

1. DIMENSIONS IN

(MILLIMETERS)

2. DRAWING NOT TO SCALE

3. THESE DIMENSIONS DO NOT INCLUDE

MOLD FLASH OR PROTRUSIONS.

MOLD FLASH OR PROTRUSIONS

SHALL NOT EXCEED .006" (0.15mm)

1

2

3

4

.030 ±.005

TYP

SO8 0303

RECOMMENDED SOLDER PAD LAYOUT

1228fc

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.

19

LT1228

TYPICAL APPLICATIO S

U

RF AGC Amplifier (Leveling Loop)

15V

10k

RF INPUT

RMS

25MHz

7

3

2

+

–

0.6V

RMS

to 1.3V

1

8

g

+

–

100Ω

m

OUTPUT

2V

300Ω

CFA

5

P–P

4

470Ω

0.01µF

10k

–15V

10k

4pF

10

Ω

0.01µF

15V

10k

100k

4.7k

–

–15V

A3

LT1006

AMPLITUDE

ADJUST

1N4148’s

COUPLE THERMALLY

LT1004

1.2V

+

LT1228 • TA20

–15V

Inverting Amplifier with DC Output Less Than 5mV

Amplitude Modulator

5V

4.7µF

+

V

2

3

7

3

7

–

+

1

8

g

1

8

+

–

m

g

+

–

m

+

6

V

5

OUT

CFA

2

6

V

5

O

CFA

100µF

0dBm(230mV) AT

MODULATION = 0V

–

CARRIER

INPUT

30mV

4

R5

4

10k

1k

–

V

R

1k

F

R

750Ω

F

4.7µF

+

V

= ±5V, R5 = 3.6k

S

R

G

–5V

G

= ±15V, R5 = 13.6k

1k

750Ω

MUST BE LESS THAN

OUT

200mV

MODULATION

INPUT ≤ 8V

LT1228 • TA22

FOR LOW OUTPUT OFFSET

P–P

BW = 30Hz TO 20MHz

V

P–P

IN

INCLUDES DC

LT1228 • TA21

RELATED PARTS

PART NUMBER

DESCRIPTION

COMMENTS

LT1227

140MHz Current Feedback Amplifier

40MHz Video Fader

1100V/µs Slew Rate, 0.01% Differential Gain, 0.03% Differential Phase

Accurate Linear Gain Control: ±1% Typ, ±3% Max

800V/µs Slew Rate, 80mA Output Current

LT1251/LT1256

LT1399

400MHz Current Feedback Amplifier

1228fc

LT 0107 REV C • PRINTED IN USA

LinearTechnology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

20

●

(408) 432-1900 FAX: (408) 434-0507 www.linear.com


型号：	LT1228CJ8#PBF
厂家：	Linear
描述：	IC 1 CHANNEL, VIDEO AMPLIFIER, CDIP8, 0.300 INCH, HERMETIC SEALED, LEAD FREE, CERAMIC, DIP-8, Audio/Video Amplifier 放大器 CD 商用集成电路
文件：	总20页 (文件大小：406K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

LT1228CJ8#PBF [Linear]

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